LT1223 100MHz Current Feedback Amplifier U DESCRIPTIO FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ 100MHz Bandwidth at AV = 1 1000V/µs Slew Rate Wide Supply Range: ±5V to ±15V 1mV Input Offset Voltage 1µA Input Bias Current 5MΩ Input Resistance 75ns Settling Time to 0.1% 50mA Output Current 6mA Quiescent Current The LT1223 is a 100MHz current feedback amplifier with very good DC characteristics. The LT1223’s high slew rate, 1000V/µs, wide supply range, ±15V, and large output drive, ±50mA, make it ideal for driving analog signals over double- terminated cables. The current feedback amplifier has high gain bandwidth at high gains, unlike conventional op amps. The LT1223 comes in the industry standard pinout and can upgrade the performance of many older products. UO APPLICATI ■ ■ ■ ■ The LT1223 is manufactured on Linear Technology’s proprietary complementary bipolar process. Video Amplifiers Buffers IF and RF Amplification Cable Drivers 8-, 10-, 12-Bit Data Acquisition Systems UO ■ S TYPICAL APPLICATI Video Cable Driver Voltage Gain vs Frequency 60 LT1223 40 – RF 1k 75Ω CABLE VOUT RG 1k 100MHz GAIN BANDWIDTH 50 75Ω VOLTAGE GAIN (dB) + V IN 30 20 10 0 + RG = 10 – RG = 33 RG 1k RG = 110 RG = 470 RG = ∞ 75Ω –10 –20 100k R AV = 1 + F RG AT AMPLIFIER OUTPUT. 6dB LESS AT VOUT . 1M 10M 100M 1G FREQUENCY (Hz) LT1223 • TPC01 LT1223 • TA02 1 LT1223 W U U W W W AXI U U ABSOLUTE PACKAGE/ORDER I FOR ATIO RATI GS Supply Voltage ...................................................... ±18V Differential Input Voltage ......................................... ±5V Input Voltage ............................ Equal to Supply Voltage Output Short Circuit Duration (Note 1) ......... Continuous Operating Temperature Range LT1223M ........................................ –55°C to 125°C LT1223C ................................................ 0°C to 70°C Storage Temperature Range ................. –65°C to 150°C Junction Temperature Plastic Package ........... 150°C Junction Temperature Ceramic Package ........ 175°C Lead Temperature (Soldering, 10 sec.)................. 300°C ORDER PART NUMBER TOP VIEW NULL 1 8 SHUTDOWN –IN 2 7 V+ +IN 3 6 OUT V– 4 5 NULL LT1223MJ8 LT1223CJ8 LT1223CN8 LT1223CS8 J8 PACKAGE N8 PACKAGE 8-LEAD CERAMIC DIP 8-LEAD PLASTIC DIP S8 PART MARKING S8 PACKAGE 8-LEAD PLASTIC SOIC LT1223 • POI01 1223 TJ MAX = 175°C, θJA = 100°C/W(J8) TJ MAX = 150°C, θJA = 100°C/W(N8) TJ MAX = 150°C, θJA = 150°C/W(S8) ELECTRICAL CHARACTERISTICS VS = ± 15V, TA = 25°C, unless otherwise noted. LT1223M/C TYP MAX VCM = 0V ±1 ±3 mV Noninverting Input Current VCM = 0V ±1 ±3 µA Inverting Input Current VCM = 0V ±1 ±3 µA en Input Noise Voltage Density f = 1kHz, RF = 1k, RG = 10Ω 3.3 nV/√Hz in Input Noise Current Density f = 1kHz, RF = 1k, RG = 10Ω 2.2 pA/√Hz RIN Input Resistance VIN = ±10V 1 10 MΩ CIN Input Capacitance 1.5 pF ±10 ±12 V SYMBOL PARAMETER CONDITIONS VOS Input Offset Voltage IIN+ IIN– Input Voltage Range CMRR PSRR MIN UNITS Common-Mode Rejection Ratio VCM = ±10V Inverting Input Current Common-Mode Rejection VCM = ±10V Power Supply Rejection Ratio VS = ±4.5V to ±18V Noninverting Input Current Power Supply Rejection VS = ±4.5V to ±18V 12 100 nA/V Inverting Input Current Power Supply Rejection VS = ±4.5V to ±18V 60 500 nA/V 56 63 30 68 dB 100 80 nA/V dB AV Large Signal Voltage Gain RLOAD = 400Ω, VOUT = ±10V 70 89 dB ROL Transresistance, ∆VOUT/∆IIN– RLOAD = 400Ω, VOUT = ±10V 1.5 5 MΩ VOUT Maximum Output Voltage Swing RLOAD = 200Ω ±10 ±12 IOUT Maximum Output Current RLOAD = 200Ω 50 60 mA SR Slew Rate RF = 1.5k, RG = 1.5k, (Note 2) 800 1300 V/µs BW Bandwidth RF = 1k, RG = 1k, VOUT = 100mV 100 MHz tr Rise Time RF = 1.5k, RG = 1.5k, VOUT = 1V 6.0 ns tPD Propagation Delay RF = 1.5k, RG = 1.5k, VOUT = 1V 6.0 ns Overshoot RF = 1.5k, RG = 1.5k, VOUT = 1V 5 % Settling Time, 0.1% RF = 1k, RG = 1k, VOUT = 10V 75 ns Differential Gain RF = 1k, RG = 1k, RL = 150Ω 0.02 % Differential Phase RF = 1k, RG = 1k, RL = 150Ω 0.12 Deg ROUT Open-Loop Output Resistance VOUT = 0, IOUT = 0 IS Supply Current VIN = 0V 6 10 mA Supply Current, Shutdown Pin 8 Current = 200µA 2 4 mA ts 2 V Ω 35 LT1223 ELECTRICAL CHARACTERISTICS VS = ± 15V, VCM = 0V, 0°C ≤ TA ≤ 70°C, unless otherwise noted. MIN LT1223C TYP MAX ● ±1 ±3 mV ● ±1 ±3 µA VCM = 0V ● ±1 ±3 µA VIN = ±10V ● 1 10 MΩ ● ±10 ±12 V 56 SYMBOL PARAMETER CONDITIONS VOS Input Offset Voltage VCM = 0V IIN+ Noninverting Input Current VCM = 0V IIN– Inverting Input Current RIN Input Resistance Input Voltage Range CMRR PSRR UNITS Common-Mode Rejection Ratio VCM = ±10V ● Inverting Input Current Common-Mode Rejection VCM = ±10V ● Power Supply Rejection Ratio VS = ±4.5V to ±18V ● Noninverting Input Current Power Supply Rejection VS = ±4.5V to ±18V ● 12 100 nA/V Inverting Input Current Power Supply Rejection VS = ±4.5V to ±18V ● 60 500 nA/ V 63 30 68 dB 100 80 nA/V dB AV Large-Signal Voltage Gain RLOAD = 400Ω, VOUT = ±10V ● 70 89 dB ROL Transresistance, ∆VOUT/∆IIN– RLOAD = 400Ω, VOUT = ±10V ● 1.5 5 MΩ VOUT Maximum Output Voltage Swing RLOAD = 200Ω ● ±10 ±12 IOUT Maximum Output Current RLOAD = 200Ω ● 50 60 IS Supply Current VIN = 0V ● 6 10 mA Supply Current, Shutdown Pin 8 Current = 200µA ● 2 4 mA V mA ELECTRICAL CHARACTERISTICS VS = ± 15V, VCM = 0V, – 55°C ≤ TA ≤ 125°C, unless otherwise noted. LT1223M TYP MAX ● ±1 ±5 mV VCM = 0V ● ±1 ±5 µA VCM = 0V ● ±1 ±10 µA VIN = ±10V ● 1 10 ● ±10 ±12 V ● 56 63 dB SYMBOL PARAMETER CONDITIONS VOS Input Offset Voltage VCM = 0V IIN+ Noninverting Input Current IIN– Inverting Input Current RIN Input Resistance CMRR Common-Mode Rejection Ratio VCM = ±10V Inverting Input Current Common-Mode Rejection VCM = ±10V ● PSRR Power Supply Rejection Ratio VS = ±4.5V to ±15V ● Noninverting Input Current Power Supply Rejection VS = ±4.5V to ±15V ● Input Voltage Range MIN 30 68 UNITS MΩ 100 80 nA/V dB 12 200 nA/V 60 500 nA/V Inverting Input Current Power Supply Rejection VS = ±4.5V to ±15V ● AV Large-Signal Voltage Gain RLOAD = 400Ω, VOUT = ±10V ● 70 ROL Transresistance, ∆VOUT/∆IIN– RLOAD = 400Ω, VOUT = ±10V ● VOUT Maximum Output Voltage Swing RLOAD = 200Ω ● IOUT Maximum Output Current RLOAD = 200Ω ● IS Supply Current VIN = 0V ● 6 10 mA Supply Current, Shutdown Pin 8 Current = 200µA ● 2 4 mA 89 dB 1.5 5 MΩ ±7 ±12 V 35 60 mA The ● denotes the specifications which apply over the full operating temperature range. Note 1: A heat sink may be required. Note 2: Noninverting operation, VOUT = ±10V, measured at ±5V. 3 LT1223 U W TYPICAL PERFOR A CE CHARACTERISTICS Supply Current vs Supply Voltage, VIN = 0 (Operating) Supply Current vs Supply Voltage (Shutdown) 4 100 OUTPUT SHORT CIRCUIT CURRENT (mA) 10 PIN 8 = 0V 125°C SUPPLY CURRENT (mA) SUPPLY CURRENT (mA) 8 25°C 6 –55°C 4 3 25°C 125°C 2 –55°C 1 2 0 0 2 4 6 8 2 0 10 12 14 16 18 20 4 6 8 5 –1 4 V S = 15V +4 +3 VS = –5V 0 100 125 –1 –3 –6 –8 –10 –5 0 5 15 OUTPUT VOLTAGE SWING (V) VOS (mV) 15 25°C –55°C –10 –15 –5 0 5 10 15 COMMON MODE VOLTAGE (V) LT1223 • TPC08 –5 0 5 10 15 LT1223 • TPC07 Output Voltage Swing vs Supply Voltage 20 VS = ±15V 10 15 125°C 25°C, –55°C 5 0 –5 –10 25°C, –55°C 125°C –15 –10 –10 COMMON MODE VOLTAGE (V) 20 0 –20 –15 10 –10 –15 Output Voltage Swing vs Load Resistor V S = ±15V 125 25°C LT1223 • TPC06 20 100 –55°C –2 –4 VOS vs Common-Mode Voltage 125°C 75 0 COMMON MODE VOLTAGE (V) 5 50 V S = ±15V 2 –2 LT1223 • TPC05 10 25 4 125°C TEMPERATURE (°C) –5 0 125°C 25°C 1 –5 –15 15 –25 6 V– –50 75 10 8 –55°C –4 50 20 –IB vs Common-Mode Voltage VS = ±15V +1 25 30 LT1223 • TPC04 –l B (µA) –4 0 40 CASE TEMPERATURE (°C) 3 VS = 5V –25 50 10 2 VS = –15V 60 +IB vs Common-Mode Voltage V+ +lB (µA) COMMON MODE RANGE (V) Input Common-Mode Limit vs Temperature +2 70 LT1223 • TPC03 LT1223 • TPC02 –3 80 SUPPLY VOLTAGE (±V) SUPPLY VOLTAGE (±V) –2 90 0 –50 10 12 14 16 18 20 OUTPUT VOLTAGE SWING (V) 0 4 Output Short Circuit-Current vs Temperature –20 100 125°C 10 –55°C 25°C 5 0 25°C –5 –10 125°C –55°C –15 –20 1000 10000 0 2 4 6 8 10 12 14 16 18 20 SUPPLY VOLTAGE (±V) LOAD RESISTOR (Ω) LT1223 • TPC09 LT1223 • TPC10 LT1223 U W TYPICAL PERFOR A CE CHARACTERISTICS 100 70 60 50 40 30 80 70 40 10 1 2 RF = 1.5k 30 10 0 RF = 1k 50 20 RF = 2k 5 10 1k 100 3 80 70 125°C 60 1000 25°C 6 125°C 5 4 –55°C 3 1 0 100 10000 1000 LT1223 • TPC16 Output Impedance vs Frequency 80 MAGNITUDE OF OUTPUT IMPEDANCE (Ω) 100 VS = ±15V RF = 1k 60 POSITIVE 40 NEGATIVE 20 +i n 10k 0 10k 100k 1M 10M 100M FREQUENCY (Hz) FREQUENCY (Hz) LT1223 • TPC17 10000 LOAD RESISTOR (Ω) Power Supply Rejection vs Frequency POWER SUPPLY REJECTION (dB) SPOT NOISE (nV/√Hz OR pA/√Hz) 7 2 VS = ±15V VO = ± 10V 50 8 LT1223 • TPC15 1000 60 VS = ± 15V VO = ± 10V 9 –55°C Spot Noise Voltage and Current vs Frequency 1 50 Transimpedance vs Load Resistor LOAD RESISTOR (Ω) en 40 10 LT1223 • TPC14 –i n 30 LT1223 • TPC13 90 FEEDBACK RESISTOR (kΩ) 1k 20 VOLTAGE GAIN (V/V) 25°C 40 100 10 100 10 0 TRANSIMPEDANCE (MΩ) OPEN LOOP VOLTAGE GAIN (dB) CAPACITIVE LOAD (pF) A V = 2; RF = RG R L = 100; VS = ± 15V PEAKING < 5dB 10 300 Open-Loop Voltage Gain vs Load Resistor 100 10 2dB PEAKING 400 LT1223 • TPC12 10k 100 500 SUPPLY VOLTAGE (± V) Maximum Capacitive Load vs Feedback Resistor 2 0dB PEAKING 600 15 LT1223 • TPC11 1 700 100 0 FEEDBACK RESISTOR (k Ω) 0 800 200 0 3 VS = ±15V R L = 100 900 RF = 750 60 20 0 RF = RG AV = 2 RL = 100 Ω TA = 25°C 90 –3dB BANDWIDTH (MHz) 80 Minimum Feedback Resistor vs Voltage Gain 1000 100 A V = 2; RF = RG R L = 100 Ω ; VS = ±15V NO CAPACITIVE LOAD 90 –3dB BANDWIDTH (MHz) –3dB Bandwidth vs Supply Voltage FEEDBACK RESISTOR (Ω) –3dB Bandwidth vs Feedback Resistor VS = ±15V 10 1 RF = RG = 3k RF = RG = 1k 0.1 0.01 10k 100k 1M 10M 100M FREQUENCY (Hz) LT1223 • TPC18 LT1223 • TPC19 5 LT1223 U W TYPICAL PERFOR A CE CHARACTERISTICS Voltage Gain and Phase vs Frequency Total Harmonic Distortion vs Frequency 225 VOLTAGE GAIN (dB) 135 GAIN 5 RL = 100Ω 90 RL ≥ 1k 0 45 PHASE –5 –10 RL = 100Ω 0 RL ≥ 1k –45 PHASE SHIFT (DEGREES) 10 0.1 180 –15 –90 –20 –135 –25 –180 –225 –30 1M 10M 100M –20 VS = ±15V VO = 7VRMS RL = 400 Ω RF = RG =1k 0.01 THD 100 10 FREQUENCY (Hz) 1k 10k 1 TO 10mV 6 0 –2 –4 Inverting Amplifier Settling Time vs Output Step 4 6 TO 1mV 2 0 –2 –4 TO 1mV –6 TO 10mV 4 0 –4 –8 –10 60 80 100 0 LT1223 • TPC23 2 0 20 40 60 80 100 SETTLING TIME (ns) LT1223 • TPC24 LT1223 • TPC25 U W U UO APPLICATI 1 SETTLING TIME (µs) TO 1mV TO 10mV –6 –8 40 TO 1mV –2 –10 SETTLING TIME (ns) TO 10mV 2 –10 20 A V = –1 RF = 1k VS = ± 15V RL = 1k 8 –8 0 100 LT1223 • TPC22 OUTPUT STEP (V) 2 –6 10 FREQUENCY (MHz) 10 A V = +1 R F = 1k VS = ± 15V RL = 1k 8 OUTPUT STEP (V) OUTPUT STEP (V) –70 100k 10 4 –50 Noninverting Amplifier Settling Time to 1mV vs Output Step 10 6 3RD LT1223 • TPC21 Noninverting Amplifier Settling Time to 10mV vs Output Step A V = +1 RF = 1k VS = ± 15V RL = 1k 2ND –40 FREQUENCY (Hz) LT1223 • TPC20 8 = ± 15V = 2VP-P = 100 = 1k = 10dB –60 0.001 1G VS VO RL RF AV –30 DISTORTION (dBc) VS = ±15V RF = RG = 1k 15 TOTAL HARMONIC DISTORTION (%) 20 2nd and 3rd Harmonic Distortion vs Frequency S I FOR ATIO Current Feedback Basics The small-signal bandwidth of the LT1223, like all current feedback amplifiers, isn’t a straight inverse function of the closed-loop gain. This is because the feedback resistors determine the amount of current driving the amplifier’s internal compensation capacitor. In fact, the amplifier’s feedback resistor (RF) from output to inverting input works with internal junction capacitances of the LT1223 to set the closed-loop bandwidth. Even though the gain set resistor (RG) from inverting input to ground works with RF to set the voltage gain just like it 6 does in a voltage feedback op amp, the closed-loop bandwidth does not change. This is because the equivalent gain bandwidth product of the current feedback amplifier is set by the Thevenin equivalent resistance at the inverting input and the internal compensation capacitor. By keeping RF constant and changing the gain with RG, the Thevenin resistance changes by the same amount as the change in gain. As a result, the net closed-loop bandwidth of the LT1223 remains the same for various closed-loop gains. LT1223 U W U UO APPLICATI S I FOR ATIO The curve on the first page shows the LT1223 voltage gain versus frequency while driving 100Ω, for five gain settings from 1 to 100. The feedback resistor is a constant 1k and the gain resistor is varied from infinity to 10Ω. Shown for comparison is a plot of the fixed 100MHz gain bandwidth limitation that a voltage feedback amplifier would have. It is obvious that for gains greater than one, the LT1223 provides 3 to 20 times more bandwidth. It is also evident that second order effects reduce the bandwidth somewhat at the higher gain settings. Feedback Resistor Selection Because the feedback resistor determines the compensation of the LT1223, bandwidth and transient response can be optimized for almost every application. To increase the bandwidth when using higher gains, the feedback resistor (and gain resistor) can be reduced from the nominal 1k value. The Minimum Feedback Resistor versus Voltage Gain curve shows the values to use for ±15V supplies. Larger feedback resistors can also be used to slow down the LT1223 as shown in the –3dB Bandwidth versus Feedback Resistor curve. Capacitive Loads The LT1223 can be isolated from capacitive loads with a small resistor (10Ω to 20Ω) or it can drive the capacitive load directly if the feedback resistor is increased. Both techniques lower the amplifier’s bandwidth about the same amount. The advantage of resistive isolation is that the bandwidth is only reduced when the capacitive load is present. The disadvantage of resistor isolation is that resistive loading causes gain errors. Because the DC accuracy is not degraded with resistive loading, the desired way of driving capacitive loads, such as flash converters, is to increase the feedback resistor. The Maximum Capacitive Load versus Feedback Resistor curve shows the value of feedback resistor and capacitive load that gives 5dB of peaking. For less peaking, use a larger feedback resistor. Power Supplies The LT1223 may be operated with single or split supplies as low as ±4V (8V total) to as high as ±18V (36V total). It is not necessary to use equal value split supplies, however, the offset voltage will degrade about 350µV per volt of mismatch. The internal compensation capacitor decreases with increasing supply voltage. The –3dB Bandwidth versus Supply Voltage curve shows how this affects the bandwidth for various feedback resistors. Generally, the bandwidth at ±5V supplies is about half the value it is at ±15V supplies for a given feedback resistor. The LT1223 is very stable even with minimal supply bypassing, however, the transient response will suffer if the supply rings. It is recommended for good slew rate and settling time that 4.7µF tantalum capacitors be placed within 0.5 inches of the supply pins. Input Range The noninverting input of the LT1223 looks like a 10M resistor in parallel with a 3pF capacitor until the common mode range is exceeded. The input impedance drops somewhat and the input current rises to about 10µA when the input comes too close to the supplies. Eventually, when the input exceeds the supply by one diode drop, the base collector junction of the input transistor forward biases and the input current rises dramatically. The input current should be limited to 10mA when exceeding the supplies. The amplifier will recover quickly when the input is returned to its normal common mode range unless the input was over 500mV beyond the supplies, then it will take an extra 100ns. Offset Adjust Output offset voltage is equal to the input offset voltage times the gain plus the inverting input bias current times the feedback resistor. For low gain applications (3 or less) a 10kΩ pot connected to pins 1 and 5 with wiper to V+ will trim the inverting input current (±10µA) to null the output; it does not change the offset voltage very much. If the LT1223 is used in a high gain application, where input offset voltage is the dominate error, it can be nulled by pulling approximately 100µA from pin 1 or 5. The easy way to do this is to use a 10kΩ pot between pin 1 and 5 with a 150k resistor from the wiper to ground for 15V supply applications. Use a 47k resistor when operating on a 5V supply. 7 LT1223 U W U UO APPLICATI S I FOR ATIO Shutdown Output Slew Rate of 500V/µs Pin 8 activates a shutdown control function. Pulling more than 200µA from pin 8 drops the supply current to less than 3mA, and puts the output into a high impedance state. The easy way to force shutdown is to ground pin 8, using an open collector (drain) logic stage. An internal resistor limits current, allowing direct interfacing with no additional parts. When pin 8 is open, the LT1223 operates normally. Slew Rate The slew rate of a current feedback amplifier is not independent of the amplifier gain configuration the way it is in a traditional op amp. This is because the input stage and the output stage both have slew rate limitations. Inverting amplifiers do not slew the input and are therefore limited only by the output stage. High gain, noninverting amplifiers are similar. The input stage slew rate of the LT1223 is about 350V/µs before it becomes nonlinear and is enhanced by the normally reverse-biased emitters on the input transistors. The output slew rate depends on the size of the feedback resistors. The peak output slew rate is about 2000V/µs with a 1k feedback resistor and drops proportionally for larger values. At an output slew rate of 1000V/µs or more, the transistors in the “mirror circuits” will begin to saturate due to the large feedback currents. This causes the output to have slew induced overshoot and is somewhat unusual looking; it is in no way harmful or dangerous to the device. The photos show the LT1223 in a noninverting gain of three (RF = 1k, RG = 500Ω) with a 20V peak-to-peak output slewing at 500V/µs, 1000V/µs and 2000V/µs. Settling Time The Inverting Amplifier Settling Time versus Output Step curve shows that the LT1223 will settle to within 1mV of final value in less than 100ns for all output changes of 10V or less. When operated as an inverting amplifier there is less than 500µV of thermal settling in the amplifier. However, when operating the LT1223 as a noninverting amplifier, there is an additional thermal settling component that is about 200µV for every volt of input common mode change. So a noninverting gain of one amplifier will 8 Output Slew Rate of 1000V/µs Output Slew Rate at 2000V/µs Shows Aberrations (See Text) LT1223 W U U UO APPLICATI S I FOR ATIO have about 2.5mV thermal tail on a 10V step. Unfortunately, reducing the input signal and increasing the gain always results in a thermal tail of about the same amount for a given output step. For this reason we show separate graphs of 10mV and 1mV non-inverting amplifier settling times. Just as the bandwidth of the LT1223 is fairly constant for various closed-loop gains, the settling time remains constant as well. Adjustable Gain Amplifier To make a variable gain amplifier with the LT1223, vary the value of RG. The implementation of RG can be a pot, a light controlled resistor, a FET, or any other low capacitance variable resistor. The value of RF should not be varied to change the gain. If RF is changed, then the bandwidth will be reduced at maximum gain and the circuit will oscillate when RF is very small. Accurate Bandwidth Limiting The LT1223 It is very common to limit the bandwidth of an op amp by putting a small capacitor in parallel with RF. DO NOT PUT A SMALL CAPACITOR FROM THE INVERTING INPUT OF A CURRENT FEEDBACK AMPLIFIER TO ANYWHERE ELSE, ESPECIALLY NOT TO THE OUTPUT. The capacitor on the inverting input will cause peaking or oscillations. If you need to limit the bandwidth of a current feedback amplifier, use a resistor and capacitor at the noninverting input (R1 & C1). This technique will also cancel (to a degree) the peaking caused by stray capacitance at the inverting input. Unfortunately, this will not limit the output noise the way it does for the op amp. V IN R1 + LT1223 C1 VOUT – V IN + RF VOUT LT1223 – RG R1 = 300Ω C1 = 100pF BW = 5MHz RF LT1223 • TA05 RG LT1223 • TA03 Current Feedback Amplifier Integrator Adjustable Bandwidth Amplifier Because the resistance at the inverting input determines the bandwidth of the LT1223, an adjustable bandwidth circuit can be made easily. The gain is set as before with RF and RG; the bandwidth is maximum when the variable resistor is at a minimum. V IN + LT1223 + VOUT LT1223 – VOUT = 1 sCI RI VIN 5k RG Since we remember that the inverting input wants to see a resistor, we can add one to the standard integrator circuit. This generates a new summing node where we can apply capacitive feedback. The LT1223 integrator has excellent large signal capability and accurate phase shift at high frequencies. RF VIN LT1223 • TA04 RI VOUT – RF 1k CI LT1223 • TA06 9 LT1223 W U U UO APPLICATI S I FOR ATIO Summing Amplifier (DC Accurate) The summing amplifier is easily made by adding additional inputs to the basic inverting amplifier configuration. The LT1223 has no IOS spec because there is no correlation between the two input bias currents. Therefore, we will not improve the DC accuracy of the inverting amplifier by putting in the extra resistor in the noninverting input. + VOUT LT1223 R 1 G V I1 – R 2 G R V I2 • • • R n G F VOUT = –R F ( RV I1 + VI2 + VIn R Gn G1 R G2 VIn ) inverting input (A1) senses the shield and the non-inverting input (A2) senses the center conductor. Since this amplifier does not load the cable (take care to minimize stray capacitance) and it rejects common mode hum and noise, several amplifiers can sense the signal with only one termination at the end of the cable. The design equations are simple. Just select the gain you need (it should be two or more) and the value of the feedback resistor (typically 1k) and calculate RG1 and RG2. The gain can be tweaked with RG2 and the CMRR with RG1 if needed. The bandwidth of the noninverting input signal is not reduced by the presence of the other amplifier, however, the inverting input signal bandwidth is reduced since it passes two amplifiers. The CMRR is good at high frequencies because the bandwidth of the amplifiers are about the same even though they do not necessarily operate at the same gain. LT1223 • TA07 RG1 1k Difference Amplifier RF1 1k The LT1223 difference amplifier delivers excellent performance if the source impedance is very low. This is because the common mode input resistance is only equal to RF + RG. RG2 1k – – A1 LT1223 RG (RF –50) 100 OPTIONAL TRIM FOR CMRR A2 LT1223 + VIN – V1 RF2 1k VOUT = G (VIN+ – VIN–) R RF1 = RF2; RG1 = (G – 1) RF2; RG2 = F2 G–1 TRIM GAIN (G) WITH RG2; TRIM CMRR WITH RG1 VOUT + VIN + LT1223 • TA09 + LT1223 RG V2 VOUT = VOUT – RF (V1 – V2 ) RG RF LT1223 • TA08 Video Instrumentation Amplifier This instrumentation amplifier uses two LT1223s to increase the input resistance to well over 1M. This makes an excellent “loop through” or cable sensing amplifier if the 10 Cable Driver The cable driver circuit is shown on the front page. When driving a cable it is important to properly terminate both ends if even modest high frequency performance is required. The additional advantage of this is that it isolates the capacitive load of the cable from the amplifier so it can operate at maximum bandwidth. LT1223 UO TYPICAL APPLICATI 150mA Output Current Video Amp V+ V+ V IN + LT1223 IN LT1010 20 Ω BIAS OUT 75Ω 75Ω 75Ω 75Ω 75Ω 75Ω 75Ω 75Ω 75Ω 75Ω – V– V– 2k 2k R f = 2k TO STABILIZE CIRCUIT DIFFERENTIAL GAIN = 1% DIFFERENTIAL PHASE = 1° LT1223 • TA10 W W SI PLIFIED SCHE ATIC 7 15k 1 5 BIAS 10k 8 3 6 2 BIAS 4 LT1223 • TA01 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 11 LT1223 U PACKAGE DESCRIPTIO Dimensions in inches (millimeters) unless otherwise noted. J8 Package 8-Lead Ceramic DIP 0.005 (0.127) MIN 0.200 (5.080) MAX 0.290 – 0.320 (7.366 – 8.128) 0.015 – 0.060 (0.381 – 1.524) 0.008 – 0.018 (0.203 – 0.460) 0.405 (10.287) MAX 8 6 7 5 0.025 (0.635) RAD TYP 0.220 – 0.310 (5.588 – 7.874) 0° – 15° 1 0.038 – 0.068 (0.965 – 1.727) 0.385 ± 0.025 (9.779 ± 0.635) 0.125 3.175 0.100 ± 0.010 MIN (2.540 ± 0.254) 0.014 – 0.026 (0.360 – 0.660) 2 3 4 0.055 (1.397) MAX J8 0392 N8 Package 8-Lead Plastic DIP 0.300 – 0.320 (7.620 – 8.128) 0.045 – 0.065 (1.143 – 1.651) 0.130 ± 0.005 (3.302 ± 0.127) 8 7 6 5 0.065 (1.651) TYP 0.009 – 0.015 (0.229 – 0.381) ( 0.400 (10.160) MAX +0.025 0.325 –0.015 +0.635 8.255 –0.381 0.250 ± 0.010 (6.350 ± 0.254) 0.045 ± 0.015 (1.143 ± 0.381) ) 0.100 ± 0.010 (2.540 ± 0.254) 0.125 (3.175) MIN 0.020 (0.508) MIN 1 2 3 4 0.018 ± 0.003 (0.457 ± 0.076) N8 0392 S8 Package 8-Lead Plastic SOIC 0.189 – 0.197 (4.801 – 5.004) 0.010 – 0.020 × 45° (0.254 – 0.508) 0°– 8° TYP 0.014 – 0.019 (0.355 – 0.483) 0.050 (1.270) BSC 6 5 0.228 – 0.244 (5.791 – 6.197) 0.150 – 0.157 (3.810 – 3.988) 1 12 7 0.004 – 0.010 (0.101 – 0.254) 0.008 – 0.010 (0.203 – 0.254) 0.016 – 0.050 0.406 – 1.270 8 0.053 – 0.069 (1.346 – 1.752) Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7487 (408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977 2 3 4 SO8 0392 LT/GP 1092 5K REV A LINEAR TECHNOLOGY CORPORATION 1992