a 5 MHz–500 MHz 100 dB Demodulating Logarithmic Amplifier with Limiter Output AD8309 FEATURES Complete Multistage Log-Limiting IF Amplifier 100 dB Dynamic Range: –78 dBm to +22 dBm (Re 50 ⍀) Stable RSSI Scaling Over Temperature and Supplies: 20 mV/dB Slope, –95 dBm Intercept ⴞ0.4 dB RSSI Linearity up to 200 MHz Programmable Limiter Gain and Output Current Differential Outputs to 10 mA, 2.4 V p-p Overall Gain 100 dB, Bandwidth 500 MHz Constant Phase (Typical ⴞ80 ps Delay Skew) Single Supply of +2.7 V to +6.5 V at 16 mA Typical Fully Differential Inputs, RIN = 1 k⍀, CIN = 2.5 pF 500 ns Power-Up Time, <1 A Sleep Current FUNCTIONAL BLOCK DIAGRAM SIX STAGES TOTAL GAIN 72dB TYP GAIN 18dB LMHI INHI 12dB 12dB 12dB LIM INLO LMLO LADR ATTEN 4 3 DET DET DET DET BIAS CTRL LMDR I-V VLOG TEN DETECTORS SPACED 12dB ENBL GAIN BIAS BAND-GAP REFERENCE SLOPE BIAS FLTR INTERCEPT TEMP COMP APPLICATIONS Receivers for Frequency and Phase Modulation Very Wide Range IF and RF Power Measurement Receiver Signal Strength Indication (RSSI) Low Cost Radar and Sonar Signal Processing Instrumentation: Network and Spectrum Analyzers PRODUCT DESCRIPTION The AD8309 is a complete IF limiting amplifier, providing both an accurate logarithmic (decibel) measure of the input signal (the RSSI function) over a dynamic range of 100 dB, and a programmable limiter output, useful from 5 MHz to 500 MHz. It is easy to use, requiring few external components. A single supply voltage of +2.7 V to +6.5 V at 16 mA is needed, corresponding to a power consumption of under 50 mW at 3 V, plus the limiter bias current, determined by the application and typically 2 mA, providing a limiter gain of 100 dB when using 200 Ω loads. A CMOS-compatible control interface can enable the AD8309 within about 500 ns and disable it to a standby current of under 1 µA. The six cascaded amplifier/limiter cells in the main path have a small signal gain of 12.04 dB (×4), with a –3 dB bandwidth of 850 MHz, providing a total gain of 72 dB. The programmable output stage provides a further 18 dB of gain. The input is fully differential and presents a moderately high impedance (1 kΩ in parallel with 2.5 pF). The input-referred noise-spectral-density, when driven from a terminated 50 Ω, source is 1.28 nV/√Hz, equivalent to a noise figure of 3 dB. The sensitivity of the AD8309 can be raised by using an input matching network. Each of the main gain cells includes a full-wave detector. An additional four detectors, driven by a broadband attenuator, are used to extend the top end of the dynamic range by over 48 dB. The overall dynamic range for this combination extends from –91 dBV (–78 dBm at the 50 Ω level) to a maximum permissible value of +9 dBV, using a balanced drive of antiphase inputs each of 2 V in amplitude, which would correspond to a sine wave power of +22 dBm if the differential input were terminated in 50 Ω. The slope of the RSSI output is closely controlled to 20 mV/dB, while the intercept is set to –108 dBV (–95 dBm re 50 Ω). These scaling parameters are determined by a bandgap voltage reference and are substantially independent of temperature and supply. The logarithmic law conformance is typically within ± 0.4 dB over the central 80 dB of this range at any frequency between 10 MHz and 200 MHz, and is degraded only slightly at 500 MHz. The RSSI response time is nominally 67 ns (10%–90%). The averaging time may be increased without limit by the addition of an external capacitor. The full output of 2.34 V at the maximum input of +9 dBV can drive any resistive load down to 50 Ω and this interface remains stable with any value of capacitance on the output. The AD8309 is fabricated on an advanced complementary bipolar process using silicon-on-insulator isolation techniques and is available in the industrial temperature range of –40°C to +85°C, in a 16-lead TSSOP package. REV. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 1999 AD8309–SPECIFICATIONS (V = +5 V, T = +25ⴗC, unless otherwise noted) S A Parameter Conditions Min1 Typ INPUT STAGE Maximum Input2 (Inputs INHI, INLO) Differential Drive, p-p ± 3.5 ±4 +9 +22 1.28 –78 1000 2.5 1.725 Equivalent Power in 50 Ω Noise Floor Equivalent Power in 50 Ω Input Resistance Input Capacitance DC Bias Voltage LIMITING AMPLIFIER Usable Frequency Range At Limiter Output Phase Variation at 100 MHz Limiter Output Current Terminated in 52.3 Ω Terminated 50 Ω Source 500 MHz Bandwidth From INHI to INLO From INHI to INLO Either Input 800 LOGARITHMIC AMPLIFIER ± 3 dB Error Dynamic Range Transfer Slope Intercept (Log Offset) Linearity Error (Ripple) Output Voltage Minimum Load Resistance, RL Maximum Sink Current Output Resistance Small-Signal Bandwidth Output Settling Time to 1% Rise/Fall Time (10%–90%) POWER INTERFACES Supply Voltage, VPOS Quiescent Current Over Temperature Disable Current Additional Bias for Limiter Logic Level to Enable Power Input Current when HI Logic Level to Disable Power 1200 Units V dBV dBm nV/√Hz dBm Ω pF V (Outputs LMHI, LMLO) RLOAD = RLIM = 50 Ω to –10 dB Point Over Input Range –60 dBm to +10 dBm Nominally 400 mV/RLIM Versus Temperature –40°C ≤ TA ≤ +85°C Input Range3 Maximum Output Voltage Rise/Fall Time (10%–90%) Max1 Equivalent dBm At Either LMHI or LMLO, wrt VPS2 RLOAD ≤ 50 Ω, 40 Ω ≤ RLIM ≤ 400 Ω (Output VLOG) From Noise Floor to Maximum Input 5 MHz ≤ f ≤ 200 MHz Over Temperature –40°C < TA < +85°C 5 MHz ≤ f ≤ 200 MHz Equivalent dBm (re 50 Ω) Over Temperature –40°C ≤ TA ≤ +85°C Equivalent dBm (re 50 Ω) Temperature Sensitivity Input from –83 dBV (–70 dBm) to +7 dBV (+20 dBm) Input = –91 dBV (–78 dBm) VS = +5 V, +2.7 V Input = +9 dBV (+22 dBm) VS = +5 V Input = +9 dBV (+22 dBm) VS = +2.75 V 5 0 –78 –65 1 18 17 –116 –103 –117 –104 40 0.75 To Ground Large Scale Input, +3 dBV (+16 dBm), RL ≥␣ 50 Ω, CL ≤␣ 100 pF Large Scale Input, +3 dBV (+16 dBm), RL ≥␣ 50 Ω, CL ≤␣ 100 pF Zero-Signal, LMDR Open –40°C < TA < +85°C –40°C < TA < +85°C RLIM = 400 Ω (See Text) HI Condition, –40°C < TA < +85°C 3 V at ENBL, –40°C < TA < +85°C LO Condition, –40°C < TA < +85°C 2.7 13 11 500 875 ±3 1 –0.008 +9 +22 1.25 0.4 100 20 20 –108 –95 –108 –95 –0.009 ± 0.4 0.34 2.34 2.10 50 1.0 0.3 3.5 22 23 –100 –87 –99 –86 2.75 1.25 dB mV/dB mV/dB dBV dBm dBV dBm dB/°C dB V V V Ω mA Ω MHz 120 220 ns 67 100 ns 5 16 16 0.01 1.4 6.5 20 23 4 1.6 VPOS 60 1 V mA mA µA mA V µA V 1.8 40 –0.5 10 MHz MHz Degrees mA %/°C dBV dBm V ns NOTES 1 Minimum and maximum specified limits on parameters that are guaranteed but not tested are six sigma values. 2 The input level is specified in “dBV” since logarithmic amplifiers respond strictly to voltage, not power. 0 dBV corresponds to a sinusoidal single-frequency input of 1 V rms. A power level of 0 dBm (1 mW) in a 50 Ω termination corresponds to an input of 0.2236 V rms. Hence, the relationship between dBV and dBm is a fixed offset of +13 dBm in the special case of a 50 Ω termination. 3 Due to the extremely high Gain Bandwidth Product of the AD8309, the output of either LMHI or LMLO will be unstable for levels below –78 dBV (–65 dBm, re 50 Ω). Specifications subject to change without notice. –2– REV. B AD8309 ABSOLUTE MAXIMUM RATINGS* Operating Temperature Range . . . . . . . . . . . . –40°C to +85°C Storage Temperature Range . . . . . . . . . . . . . –65°C to +150°C Lead Temperature Range (Soldering 60 sec) . . . . . . . . +300°C Supply Voltage VS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.5 V Input Level, Differential (re 50 Ω) . . . . . . . . . . . . . . . +26 dBm Input Level, Single-Ended (re 50 Ω) . . . . . . . . . . . . . +20 dBm Internal Power Dissipation . . . . . . . . . . . . . . . . . . . . . 500 mW θJA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C/W θJC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .27.6°C/W Maximum Junction Temperature . . . . . . . . . . . . . . . . +125°C *Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may effect device reliability. ORDERING GUIDE Model Temperature Range Package Description Package Option AD8309ARU –40°C to +85°C 16-Lead TSSOP RU-16 AD8309ARU-REEL –40°C to +85°C 13" Tape and Reel RU-16 AD8309ARU-REEL7 –40°C to +85°C 7" Tape and Reel RU-16 AD8309-EVAL Evaluation Board CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD8309 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. Name Function 1 2 COM2 Special Common Pin for RSSI Output. VPS1 Supply Pin for First Five Amplifier Stages and the Main Biasing System. 3, 6, 11, 14 PADL Four Tie-Downs to the Paddle on Which the IC Is Mounted; Grounded. 4 INHI Signal Input, HI or Plus Polarity. 5 INLO Signal Input, LO or Minus Polarity. 7 COM1 Main Common Connection. 8 ENBL Chip Enable; Active When HI. 9 LMDR Limiter Drive Programming Pin. 10 FLTR RSSI Bandwidth-Reduction Pin. 12 LMLO Limiter Output, LO or Minus Polarity. 13 LMHI Limiter Output, HI or Plus Polarity. 15 VPS2 Supply Pin for Sixth Gain Stage, Limiter and RSSI Output Stage Load Current. 16 VLOG Logarithmic (RSSI) Output. REV. B ESD SENSITIVE DEVICE PIN CONFIGURATION PIN FUNCTION DESCRIPTIONS Pin WARNING! COM2 1 16 VLOG VPS1 2 15 VPS2 PADL 3 14 PADL AD8309 LMHI TOP VIEW INLO 5 (Not to Scale) 12 LMLO INHI 4 –3– 13 PADL 6 11 PADL COM1 7 10 FLTR ENBL 8 9 LMDR AD8309–Typical Performance Characteristics 100 10 SUPPLY CURRENT – mA VLOG 1 500mV PER VERTICAL DIVISION +258C 0.1 +858C GROUND REFERENCE 0.01 +10dBm INPUT LEVEL SHOWN HERE –408C 0.001 500mV PER VERTICAL DIVISION 0.0001 0.00001 0.5 100ns PER HORIZONTAL DIVISION 0.7 0.9 1.1 1.3 1.5 1.7 1.9 ENABLE VOLTAGE – V 2.1 2.3 2.5 Figure 1. Supply Current vs. Enable Voltage @ TA = –40 °C, +25 °C and +85 °C Figure 4. RSSI Pulse Response for Inputs Stepped from Zero to –63 dBV, –43 dBV, –23 dBV, –3 dBV –13dBV VLOG –33dBV –53dBV 500mV PER VERTICAL DIVISION VLOG 500mV PER VERTICAL DIVISION –73dBV –93dBV GROUND REFERENCE GROUND REFERENCE ENBL 5V PER VERTICAL DIVISION 100ns PER HORIZONTAL DIVISION 500ns PER HORIZONTAL DIVISION INPUT 1V PER VERTICAL DIVISION Figure 5. Large Signal RSSI Pulse Response with RL = 100 Ω and CL = 33 pF, 100 pF and 330 pF (Curves Overlap) Figure 2. Power On/Off Response Time with RF Input of –93 dBV to –13 dBV 27pF 270pF VLOG VLOG 3300pF 200mV PER VERTICAL DIVISION 500mV PER VERTICAL DIVISION GROUND REFERENCE GROUND REFERENCE INPUT 500mV PER VERTICAL DIVISION 100ms PER HORIZONTAL DIVISION 200ns PER HORIZONTAL DIVISION Figure 3. Large Signal RSSI Pulse Response with CL = 100 pF and RL = 50 Ω and 75 Ω (Curves Overlap) Figure 6. Small Signal AC Response of RSSI Output with External Filter Capacitance of 27 pF, 270 pF and 3300 pF –4– REV. B AD8309 2.5 5 4 3 2 ERROR – dB RSSI OUTPUT – V 2.0 1.5 1.0 TA = +858C TA = +258C TA = –408C –3 TA = –408C –4 TA = +258C –80 –60 –40 –20 0 INPUT LEVEL – dBm Re 50V 20 –5 –100 40 Figure 7. RSSI Output vs. Input Level, 100 MHz Sine Input, at TA = –40 °C, +25 °C and +85 °C, Single-Ended Input 0 –60 –40 –20 INPUT LEVEL – dBm Re 50V –80 5 100MHz 50MHz 200MHz 5MHz 2.0 20 40 Figure 10. Log Linearity of RSSI Output vs. Input Level, 100 MHz Sine Input, at TA = –40°C, +25°C, and +85°C 2.5 DYNAMIC RANGE 5MHz 50MHz 100MHz 200MHz 4 3 5MHz 61dB 85 91 97 96 63dB 93 99 103 102 2 ERROR – dB RSSI OUTPUT – V –1 –2 0.5 0 –100 TA = +858C 1 0 1.5 1.0 1 50MHz 0 –1 100MHz 200MHz –2 –3 0.5 –4 0 –100 –80 –60 –40 –20 0 INPUT LEVEL – dBm Re 50V 20 –5 –90 –80 –70 –60 –50 –40 –30 –20 –10 0 INPUT LEVEL – dBm Re 50V 40 Figure 8. RSSI Output vs. Input Level, at TA = +25 °C, for Frequencies of 5 MHz, 50 MHz, 100 MHz and 200 MHz 5 DYNAMIC RANGE 300MHz 400MHz 500MHz 4 300MHz 400MHz 500MHz 3 30 61dB 90 65 66 63dB 102 100 100 2 ERROR – dB RSSI OUTPUT – V 20 Figure 11. Log Linearity of RSSI Output vs. Input Level, at TA = +25 °C, for Frequencies of 5 MHz, 50 MHz, 100 MHz and 200 MHz 2.5 2.0 10 1.5 1.0 1 300MHz 0 –1 400MHz –2 500MHz 0.5 –3 –4 0 –100 –80 –60 –40 –20 0 INPUT LEVEL – dBm Re 50V 20 –5 –90 –80 –70 –60 –50 –40 –30 –20 –10 0 INPUT LEVEL – dBm Re 50V 40 Figure 9. RSSI Output vs. Input Level, at TA = +25°C, for Frequencies of 300 MHz, 400 MHz and 500 MHz REV. B 10 20 30 Figure 12. Log Linearity of RSSI Output vs. Input Level, at TA = +25°C, for Frequencies of 300 MHz, 400 MHz and 500 MHz –5– 25 –103 24 –104 23 –105 RSSI INTERCEPT – dBV RSSI SLOPE – mV/dB AD8309 22 21 20 19 18 –106 –107 –108 –109 –110 17 –111 16 –112 15 –113 10 100 FREQUENCY – MHz 1 1000 Figure 13. RSSI Slope vs. Frequency Using Termination of 52.3 Ω in Series with 4.7 nH 10 100 FREQUENCY – MHz 1 1000 Figure 16. RSSI Intercept vs. Frequency Using Termination of 52.3 Ω in Series with 4.7 nH 14 2ns PER HORIZONTAL DIVISION 12 10 100mV PER VERTICAL DIVISION CURRENT – mA LIMITER OUTPUTS ADDITIONAL SUPPLY CURRENT 8 6 4 2 LIMITER OUTPUT CURRENT 2mV PER VERTICAL DIVISION 0 Figure 14. Limiter Output at 300 MHz for a Sine Wave Input of –60 dBV (–47 dBm), Using an RLOAD of 50 Ω and an RLIM of 100 Ω 0 50 100 150 200 250 RLIM – V 300 350 400 450 Figure 17. Additional Supply Current and Limiter Output Current vs. RLIM NORMALIZED LIMITER PHASE RESPONSE – Degrees 10 LMLO LMHI LIMITER OUTPUTS 50mV PER VERTICAL DIVISION INPUT 1mV PER VERTICAL DIVISION 12.5ns PER HORIZONTAL DIVISION 8 6 TA = –408C 2 0 –2 –4 TA = +258C –6 –8 –10 –60 Figure 15. Limiter Response at LMHI, LMLO with Pulsed Sine Input of –70 dBV (–57 dBm) at 50 MHz; RLOAD = 50 Ω, RLIM = 200 Ω TA = +858C 4 –50 –40 –30 –20 –10 INPUT LEVEL – dBm Re 50V 0 10 Figure 18. Normalized Limiter Phase Response vs. Input Level. Frequency = 100 MHz; TA = –40°C, +25°C and +85 °C –6– REV. B AD8309 references. Note that (1) is mathematically incomplete in representing the behavior of a demodulating log amp such as the AD8309, where VIN has an alternating sign. However, the basic principles are unaffected. THEORY OF OPERATION The AD8309 is an advanced IF signal processing IC, intended for use in high performance receivers, combining two key functions. First, it provides a large voltage gain combined with progressive compression, through which an IF signal of high dynamic range is converted into a square-wave (that is, hard limited) output, from which frequency and phase information modulated on this input can be recovered by subsequent signal processing. For this purpose, the noise level referred to the input must be very low, since it determines the detection threshold for the receiver. Figure 19 shows the input/output relationship of an ideal log amp, conforming to Equation (1). The horizontal scale is logarithmic, and spans a very wide dynamic range, shown here as over 120 dB, that is, six decades of voltage or twelve decades of input-referred power. The output passes through zero (the “log-intercept”) at the unique value VIN = VX and becomes negative for inputs below the intercept. In the ideal case, the straight line describing VOUT for all values of VIN would continue indefinitely in both directions. The dotted line shows that the effect of adding an offset voltage VSHIFT to the output is to lower the effective intercept voltage VX. Further, it is often important that the group delay in this amplifier be essentially independent of the signal level, to minimize the risk of amplitude-to-phase conversion. Finally, it is also desirable that the amplitude of the limited output be well defined and temperature stable. In the AD8309, this amplitude can be controlled by the user, or even completely shut off, providing greater flexibility. VOUT 5VY The second function is to provide a demodulated (baseband) output proportional to the decibel value of the signal input, which may be used to measure the signal strength. This output, which typically runs from a value close to the ground level to a few volts above ground, is called the Received Signal Strength Indication, or RSSI. The provision of this function requires the use of a logarithmic amplifier (log amp). For this output to be suitable for measuring signal strength, it is important that its scaling attributes are well controlled. 4VY VSHIFT 3VY 2VY VY LOG VIN VOUT = 0 VIN = 10–2VX –40dBc These are the logarithmic slope, specified in mV/dB, and the intercept, often specified as an equivalent power level at the amplifier input, although a log amp is inherently a voltageresponding device. (See further discussion, below). Also important is the law conformance, that is, how well the RSSI approximates an ideal function. Many low quality log amps provide only an approximate solution, resulting in large errors in law conformance and scaling. All Analog Devices log amps are designed with close attention to matters affecting accuracy of the overall function. VIN = 102VX +40dBc VIN = 104VX +80dBc Figure 19. Ideal Log Amp Function Exactly the same modification could be achieved raising the gain (or signal level) ahead of the log amp by the factor VSHIFT/VY. For example, if VY is 400 mV/decade (that is, 20 mV/dB, as for the AD8309), an offset of 120 mV added to the output will appear to lower the intercept by two tenths of a decade, or 6 dB. Adding an offset to the output is thus indistinguishable from applying an input level that is 6 dB higher. The log amp function described by (1) differs from that of a linear amplifier in that the incremental gain DVOUT/DVIN is a very strong function of the instantaneous value of VIN, as is apparent by calculating the derivative. For the case where the logarithmic base is e, it is easy to show that Log-Amp Fundamentals The essential purpose of a logarithmic amplifier is to reduce a signal of wide dynamic range to its decibel equivalent. It is thus primarily a measurement device. The logarithmic representation leads to situations that may be confusing or even paradoxical. For example, a voltage offset added to the RSSI output of a log amp is equivalent to a gain increase ahead of its input. ∆VOUT VY = ∆VIN VIN (2) That is, the incremental gain of a log amp is inversely proportional to the instantaneous value of the input voltage. This remains true for any logarithmic base. A “perfect” log amp would be required to have infinite gain under classical “small-signal” (zero-amplitude) conditions. This demonstrates that, whatever means might be used to implement a log amp, accurate HF response under small signal conditions (that is, at the lower end of the full dynamic range) demands the provision of a very high gain-bandwidth product. A wideband log amp must therefore use many cascaded gain cells each of low gain but high bandwidth. For the AD8309, the gain-bandwidth (–10 dB) product is 52,500 GHz. When all the variables expressed as voltages, then, regardless of the particular structure, the output can be expressed as (1) where VY is the “slope voltage.” VIN is the input voltage, and VX is the “intercept voltage.” The logarithm is usually to base-10, which is appropriate to a decibel-calibrated device, in which case VY is also the “volts-per-decade.” It will be apparent from (1) that a log amp requires two references, here VX and VY, that determine the scaling of the circuit. The absolute accuracy of a log amp cannot be any better than the accuracy of its scaling REV. B VIN = VX 0dBc –2VY In the AD8309, these two basic signal-processing functions are combined to provide the necessary voltage gain with progressive compression and hard limiting, and the determination of the logarithmic magnitude of the input (RSSI). This combination is called a log limiting amplifier. A good grasp of how this product works will avoid many pitfalls in their application. VOUT = VY log (VIN /VX) LOWER INTERCEPT –7– AD8309 As a consequence of this high gain, even very small amounts of thermal noise at the input of a log amp will cause a finite output for zero input, resulting in the response line curving away from the ideal (Figure 19) at small inputs, toward a fixed baseline. This can either be above or below the intercept, depending on the design. Note that the value specified for this intercept is invariably an extrapolated one: the RSSI output voltage will never attain a value of exactly zero in a single supply implementation. Voltage (dBV) and Power (dBm) Response While Equation 1 is fundamentally correct, a simpler formula is appropriate for specifying the RSSI calibration attributes of a log amp like the AD8309, which demodulates an RF input. The usual measure is input power: VOUT = VSLOPE (PIN – P0 ) (3) VOUT is the demodulated and filtered RSSI output, VSLOPE is the logarithmic slope, expressed in volts/dB, PIN is the input power, expressed in decibels relative to some reference power level and P0 is the logarithmic intercept, expressed in decibels relative to the same reference level. The most widely used convention in RF systems is to specify power in decibels above 1 mW in 50 Ω, written dBm. (However, that the quantity [PIN – P0 ] is simply dB). The logarithmic function disappears from this formula because the conversion has already been implicitly performed in stating the input in decibels. Specification of log amp input level in terms of power is strictly a concession to popular convention: they do not respond to power (tacitly “power absorbed at the input”), but to the input voltage. In this connection, note that the input impedance of the AD8309 is much higher that 50 Ω, allowing the use of an impedance transformer at the input to raise the sensitivity, by up to 13 dB. The use of dBV, defined as decibels with respect to a 1 V rms sine amplitude, is more precise, although this is still not unambiguous complete as a general metric, because waveform is also involved in the response of a log amp, which, for a complex input (such as a CDMA signal) will not follow the rms value exactly. Since most users specify RF signals in terms of power—more specifically, in dBm/50 Ω—we use both dBV and dBm in specifying the performance of the AD8309, showing equivalent dBm levels for the special case of a 50 Ω environment. STAGE 1 STAGE 2 A VX STAGE N –1 A STAGE N A A VW Figure 20. Cascade of Nonlinear Gain Cells Theory of Logarithmic Amplifiers To develop the theory, we will first consider a somewhat different scheme to that employed in the AD8309, but which is simpler to explain, and mathematically more straightforward to analyze. This approach is based on a nonlinear amplifier unit, which we may call an A/1 cell, having the transfer characteristic shown in Figure 21. We here use lowercase variables to define the local inputs and outputs of these cells, reserving uppercase for external signals. The small signal gain ∆VOUT/∆VIN is A, and is maintained for inputs up to the knee voltage EK, above which the incremental gain drops to unity. The function is symmetrical: the same drop in gain occurs for instantaneous values of VIN less than –EK. The large signal gain has a value of A for inputs in the range –EK < VIN < +EK, but falls asymptotically toward unity for very large inputs. In logarithmic amplifiers based on this simple function, both the slope voltage and the intercept voltage must be traceable to the one reference voltage, EK. Therefore, in this fundamental analysis, the calibration accuracy of the log amp is dependent solely on this voltage. In practice, it is possible to separate the basic references used to determine VY and VX. In the AD8309, VY is traceable to an on-chip band-gap reference, while VX is derived from the thermal voltage kT/q and later temperature-corrected by a precise means. Let the input of an N-cell cascade be VIN, and the final output VOUT. For small signals, the overall gain is simply AN. A sixstage system in which A = 5 (14 dB) has an overall gain of 15,625 (84 dB). The importance of a very high small-signal ac gain in implementing the logarithmic function has already been noted. However, this is a parameter of only incidental interest in the design of log amps; greater emphasis needs to be placed on the nonlinear behavior. High speed, high dynamic range log amps use a cascade of nonlinear amplifier cells (Figure 20) to generate the logarithmic function from a series of contiguous segments, a type of piecewise-linear technique. This basic topology offers enormous gainbandwidth products. For example, the AD8309 employs in its main signal path six cells each having a small-signal gain of 12.04 dB (×4) and a –3 dB bandwidth of 850 MHz, followed by a final limiter stage whose gain is typically 18 dB. The overall gain is thus 100,000 (100 dB) and the bandwidth to –10 dB point at the limiter output is 525 MHz. This very high gainbandwidth product (52,500 GHz) is an essential prerequisite to accurate operation under small signal conditions and at high frequencies: Equation (2) reminds us that the incremental gain decreases rapidly as VIN increases. The AD8309 exhibits a logarithmic response over most of the range from the noise floor of –91 dBV, or 28 µV rms, (or –78 dBm/50 Ω) to a breakdownlimited peak input of 4 V (requiring a balanced drive at the differential inputs INHI and INLO). A/1 OUTPUT Progressive Compression AEK SLOPE = 1 SLOPE = A 0 EK INPUT Figure 21. The A/1 Amplifier Function Thus, rather than considering gain, we will analyze the overall nonlinear behavior of the cascade in response to a simple dc input, corresponding to the VIN of Equation (1). For very small inputs, the output from the first cell is V1 = AVIN; from the second, V2 = A2 VIN, and so on, up to VN = AN VIN. At a certain value of VIN, the input to the Nth cell, VN–1, is exactly equal to the knee voltage EK. Thus, VOUT = AEK and since there are N–1 cells of gain A ahead of this node, we can calculate that VIN = EK /AN–1. This unique point corresponds to the lin-log transition, –8– REV. B AD8309 labeled ① on Figure 22. Below this input, the cascade of gain cells is acting as a simple linear amplifier, while for higher values of VIN, it enters into a series of segments which lie on a logarithmic approximation. VOUT (4A-3) EK (3A-2) EK (A-1) EK (2A-1) EK A/0 OUTPUT Continuing this analysis, we find that the next transition occurs when the input to the (N–1)th stage just reaches EK, that is, when VIN = EK /AN–2. The output of this stage is then exactly AEK. It is easily demonstrated (from the function shown in Figure 21) that the output of the final stage is (2A–1)EK (labeled ≠ on Figure 22). Thus, the output has changed by an amount (A–1)EK for a change in VIN from EK /AN–1 to EK /AN–2, that is, a ratio change of A. SLOPE = 0 AEK SLOPE = A 0 EK INPUT Figure 23. A/0 Amplifier Functions (Ideal and tanh) Care is needed in the interpretation of this parameter. It was earlier defined as the input voltage at which the output passes through zero (see Figure 19). Clearly, in the absence of noise and offsets, the output of the amplifier chain shown in Figure 20 can only be zero when VIN = 0. This anomaly is due to the finite gain of the cascaded amplifier, which results in a failure to maintain the logarithmic approximation below the “lin-log transition” (Point ① in Figure 22). Closer analysis shows that the voltage given by Equation (5) represents the extrapolated, rather than actual, intercept. Demodulating Log Amps AEK RATIO OF A LOG VIN 0 EK/AN–1 EK/AN–2 EK/AN–3 EK/AN–4 Figure 22. The First Three Transitions At the next critical point, labeled ③, the input is A times larger and VOUT has increased to (3A–2)EK, that is, by another linear increment of (A–1)EK. Further analysis shows that, right up to the point where the input to the first cell reaches the knee voltage, VOUT changes by (A–1)EK for a ratio change of A in VIN. Expressed as a certain fraction of a decade, this is simply log10(A). For example, when A = 5 a transition in the piecewise linear output function occurs at regular intervals of 0.7 decade (log10(A), or 14 dB divided by 20 dB). This insight allows us to immediately state the “Volts per Decade” scaling parameter, which is also the “Scaling Voltage” VY when using base-10 logarithms: VY = Linear Change inVOUT ( A – 1)E K = Decades Change inVIN log10( A) (4) Note that only two design parameters are involved in determining VY, namely, the cell gain A and the knee voltage EK, while N, the number of stages, is unimportant in setting the slope of the overall function. For A = 5 and EK = 100 mV, the slope would be a rather awkward 572.3 mV per decade (28.6 mV/dB). A well designed practical log amp will provide more rational scaling parameters. The intercept voltage can be determined by solving Equation (4) for any two pairs of transition points on the output function (see Figure 22). The result is: VX = EK A( N +1/[ A–1]) (5) For the example under consideration, using N = 6, VX evaluates to 4.28 µV, which thus far in this analysis is still a simple dc voltage. REV. B Log amps based on a cascade of A/1 cells are useful in baseband (pulse) applications, because they do not demodulate their input signal. Demodulating (detecting) log-limiting amplifiers such as the AD8309 use a different type of amplifier stage, which we will call an A/0 cell. Its function differs from that of the A/1 cell in that the gain above the knee voltage EK falls to zero, as shown by the solid line in Figure 23. This is also known as the limiter function, and a chain of N such cells is often used alone to generate a hard limited output, in recovering the signal in FM and PM modes. The AD640, AD606, AD608, AD8307, AD8309, AD8313 and other Analog Devices communications products incorporating a logarithmic IF amplifier all use this technique. It will be apparent that the output of the last stage cannot now provide a logarithmic output, since this remains unchanged for all inputs above the limiting threshold, which occurs at VIN = EK /AN–1. Instead, the logarithmic output is generated by summing the outputs of all the stages. The full analysis for this type of log amp is only slightly more complicated than that of the previous case. It can be shown that, for practical purpose, the intercept voltage VX is identical to that given in Equation (5), while the slope voltage is: VY = A EK log10( A) (6) An A/0 cell can be very simple. In the AD8309 it is based on a bipolar-transistor differential pair, having resistive loads RL and an emitter current source IE. This amplifier limiter cell exhibits an equivalent knee-voltage of EK = 2kT/q and a small-signal gain of A = IERL /EK. The large signal transfer function is the hyperbolic tangent (see dotted line in Figure 23). This function is very precise, and the deviation from an ideal A/0 form is not detrimental. In fact, the “rounded shoulders” of the tanh function beneficially result in a lower ripple in the logarithmic conformance than that which would be obtained using an ideal A/0 function. A practical amplifier chain built of these cells is differential in structure from input to final output, and has a low –9– AD8309 sensitivity to disturbances on the supply lines. With careful design, the sensitivities to many other parametric variations, and the effects of temperature and supply voltage, can be reduced to negligible proportions. STAGE 1 VIN A/0 gm STAGE 2 A/0 gm STAGE N VLIM A/0 gm gm – +TOP-END DETECTORS CURRENT-SUMMING LINE Intercept Calibration Monolithic log amps from Analog Devices incorporate accurate means to position the intercept voltage VX (or equivalent sinewave power for a demodulating log amp, when driven at a specific impedance level). Using the scheme shown in Figure 24, the value of the intercept level departs considerably from that predicted by the simple theory. Nevertheless, the intrinsic intercept voltage is still proportional to EK, which is PTAT (proportional to absolute temperature). Recalling that the addition of an offset to the output produces an effect which is indistinguishable from a change in the position of the intercept, it will be apparent that we can cancel the “left-right” motion of VX resulting from the temperature variation of EK by simply adding an offset at its demodulated output having the required temperature behavior. VLOG RSLOPE Figure 24. Basic Log Amp Structure Using A/0 Stages and Transconductance (gm) Cells for Summing The output of each gain cell has an associated transconductance (gm) cell, which converts the differential output voltage of the cell to a pair of differential currents; these are summed by simply connecting the outputs of all the gm (detector) stages in parallel. The total current is then converted back to a voltage by a transresistance stage, which determines the slope of the logarithmic output. This general scheme is depicted, in a simplified single-sided form, in Figure 24. Additional detectors, driven by a passive attenuator, may be added to extend the top end of the dynamic range. The precise temperature-shaping of the intercept-positioning offset can result in a log amp having stable scaling parameters, making it a true measurement device, for example, as a calibrated Received Signal Strength Indicator (RSSI). In this application, one is more interested in the value of the output for an input waveform which is often sinusoidal (CW). The input level be stated as an equivalent power, in dBm, but it is essential to know the impedance level at which this “power” is presumed to be measured. In an impedance of 50 Ω, 0 dBm (1 mW) corresponds to a sinusoidal amplitude of 316.2 mV (223.6 mV rms). The slope voltage may now be decoupled from the knee-voltage EK = 2kT/q, which is inherently PTAT. The detector stages are biased with currents (not shown in the Figure) which can be derived from a band-gap reference and thus be stable with temperature. This is the architecture used in the AD8309. It affords complete control over the magnitude and temperature behavior of the logarithmic slope. For the AD8309, the intercept may be specified in dBm when the input impedance is lowered to 50 Ω, by the addition of a shunt resistor of 52.3 Ω, in which case it occurs at –95 dBm. However, the response is actually to the voltage at the input, not the power in the termination resistor, and should be specified in dBV. A –95 dBm sine input across a 50 Ω resistance corresponds to an amplitude of 5.6 µV, or –108 dBV, where 0 dBV is specified as a sine waveform of 1 V rms, that is, 2.8 V p-p. A further step is yet needed to achieve the demodulation response, required in a log-limiter amp is to convert an alternating input into a quasi- dc baseband output. This is achieved by modifying the gm cells used for summation purposes to implement the rectification function. Early log amps based on the progressive compression technique used half-wave rectifiers, which made post-detection filtering difficult. The AD640 was the first commercial monolithic log amp to use a full-wave rectifier; this proprietary practice has been used in all subsequent Analog Devices types. Note that a log amp’s intercept is a function of waveform. For example, a square-wave input will read 6 dB higher than a sinewave of the same amplitude, and a Gaussian noise input 0.5 dB higher than a sine wave of the same rms value. Further, a log amp driven by the sum of two sinusoidal voltages of equal amplitude will show an output that is only 2.1 dB higher than the response for a single sine wave drive, rather than the 3 dB that might be expected if the device truly responded to input power. These are characteristics exhibited by all demodulating log amps. We can model these detectors as being essentially linear gm cells, but producing an output current that is independent of the sign of the voltage applied to the input. That is, they implement the absolute-value function. Since the output from the later A/0 stages closely approximates an amplitude symmetric square wave for even moderate input levels, the current output from each detector is almost constant over each period of the input. Somewhat earlier detectors stages in the chain produce a waveform having only very brief “dropouts” at twice the input frequency. Only those detectors nearest the log amp’s input produce a low level waveform that is approximately sinusoidal. When all these (current mode) outputs are summed, the resulting signal has a waveform which is readily filtered, to provide a low residual ripple on the output. Dynamic Range The lower end of the dynamic range is determined largely by the thermal noise floor, measured at the input of the amplifier chain. For the AD8309, the short-circuit input-referred noise-spectral density is 1.1 nV/√Hz, and 1.275 nV/√Hz when driven from a net source impedance of 25 Ω (a terminated 50 Ω). This corresponds to a noise power of –78 dBm in a 500 MHz bandwidth. The upper end of the dynamic range is extended upward by the addition of top-end detectors driven by a tapped attenuator. These smaller signals are applied to additional full-wave detectors whose outputs are summed with those of the main detectors. With care in design, this extension in the dynamic range can be ‘seamless’ over the full frequency range. For the AD8309 it amounts to a further 48 dB. When using a supply of 4.5 V or greater, an input amplitude of 4 V can be accommodated, corresponding to a power level of +22 dBm in 50 Ω. (A larger input voltage may cause damage.) –10– REV. B AD8309 a sine wave input having an amplitude of 100 mV or greater, and RLOAD = 50 Ω. The change in time-delay (“phase skew”) over the input range –83 dBV (100 mV in amplitude, or –70 dBm in 50 Ω) to –3 dBV (1 V or +10 dBm) is ±83 ps (±3° at 100 MHz). The total dynamic range of the AD8309, defined as the ratio of the maximum permissible input to the noise floor, is thus 100 dB. Good accuracy is provided over a substantial part of this range. Input Matching SIX STAGES TOTAL GAIN 72dB Monolithic log amps present a nominal input impedance much higher than 50 Ω. For the AD8309, this can be modeled as 1 kΩ shunted by 2.5 pF, at frequencies up to 300 MHz. Thus, a simple input matching network can considerably improve the basic sensitivity , when driving from a low-impedance source, by increasing the voltage applied to the input. For a 50:1000 Ω transformation, the voltage gain is 13 dB, and the whole dynamic range moves downward by this amount; that is, the intercept is shifted to –121 dBV (–108 dBm at the primary 50 Ω input). Note that while useful voltage gain is achieved in this way, it does not follow that the noise-figure is minimal at the optimum power match. Offset Control In a monolithic log amp, direct-coupling between the stages is invariably utilized for practical reasons. Now, a dc offset voltage in the early stages of the chain is indistinguishable from a “real” signal. If as high as 400 µV, it would be 20 dB larger than the smallest resolvable ac signal (40 µV), reducing the dynamic range by this amount. This problem is solved by using a global feedback path from the last stage to the first. The high-frequency components of the signal must be removed; this achieved in the AD8309 by an on-chip low-pass filter, providing sufficient suppression of HF feedback to allow accurate operation down to at least 5 MHz. Useful operation at lower frequencies remains possible, although a particular device having a large dc offset will exhibit a reduction in the low end region of the dynamic range. PRODUCT OVERVIEW The AD8309 is built on an advanced dielectrically-isolated complementary bipolar process using thin-film resistor technology for accurate scaling. It follows well-developed foundations proven over a period of some fifteen years, with constant refinement. The backbone of the AD8309 (Figure 25) comprises a chain of six main amplifier/limiter stages, each having a gain of 12.04 dB (×4) and small-signal –3 dB bandwidth of 850 MHz. The input interface at INHI and INLO (Pins 4 and 5) is fully differential. Thus it may be driven from either single-sided or balanced inputs, the latter being required at the very top end of the dynamic range, where the total differential drive may be as large as 4 V in amplitude. The first six stages, also used in developing the logarithmic RSSI output, are followed by a versatile programmable output, and thus programmable gain, final limiter section. Its opencollector outputs are also fully differential, at LMHI and LMLO (Pins 12 and 13). This output stage provides a gain of 18 dB when using equal valued load and bias setting resistors and the pin-to-pin output is used. The overall voltage gain is thus 100 dB. When using RLIM = RLOAD = 200 Ω, the additional current consumption in the limiter is approximately 2.8 mA, of which 2 mA goes to the load. The ratio depends on RLIM (for example, when 20 Ω, the efficiency is 90%), and the voltage at the pin LMDR is rather more than 400 mV, but the total load current is accurately (400 mV)/RLIM. The rise and fall times of the hard-limited (essentially squarewave) voltage at the outputs are typically 0.4 ns, when driven by REV. B TYP GAIN 18dB LMHI INHI 12dB 12dB 12dB LIM INLO LMLO LADR ATTEN 4 3 DET DET DET DET BIAS CTRL LMDR I-V VLOG TEN DETECTORS SPACED 12dB ENBL GAIN BIAS BAND-GAP REFERENCE SLOPE BIAS FLTR INTERCEPT TEMP COMP Figure 25. Main Features of the AD8309 The six main cells and their associated full-wave detectors, having a transconductance (gm) form, handle the lower part of the dynamic range. Biasing for these cells is provided by two references, one of which determines their gain, the other being a band-gap cell which determines the logarithmic slope, and stabilizes it against supply and temperature variations. A special dcoffset-sensing cell (not shown in Figure 25) is placed at the end of this main section, and used to null any residual offset at the input, ensuring accurate response down to the noise floor. The first amplifier stage provides a short-circuited voltage-noise spectral-density of 1.07 nV/√Hz. The last detector stage includes a modification to temperaturestabilize the log-intercept, which is accurately positioned so as to make optimal use of the full output voltage range. Four further “top end” detectors are placed at 12.04 dB taps along a passive attenuator, to handle the upper part of the range. The differential current-mode outputs of all ten detectors stages are summed with equal weightings and converted to a single-sided voltage by the output stage, generating the logarithmic (or RSSI) output at VLOG (Pin 16), nominally scaled 20 mV/dB (that is, 400 mV per decade). The junction between the lower and upper regions is seamless, and the logarithmic law-conformance is typically well within ± 0.4 dB from –83 dBV to +7 dBV (–70 dBm to +10 dBm). The full-scale rise time of the RSSI output stage, which operates as a two-pole low-pass filter with a corner frequency of 3.5 MHz, is about 200 ns. A capacitor connected between FLTR (Pin 10) and VLOG can be used to lower the corner frequency (see below). The output has a minimum level of about 0.34 V (corresponding to a noise power of –78 dBm, or 17 dB above the nominal intercept of –95 dBm). This rather high baseline level ensures that the pulse response remains unimpaired at very low inputs. The maximum RSSI output depends on the supply voltage and the load. An output of 2.34 V, that is, 20 mV/dB × (12 + 105) dB, is guaranteed when using a supply voltage of 4.5 V or greater and a load resistance of 50 Ω or higher, for a differential input of 9 dBV (a 4 V sine amplitude, using balanced drives). When using a 3 V supply, the maximum differential input may still be as high as –3 dBV (1 V sine amplitude), and the corresponding RSSI output of 2.1 V, that is, 20 mV/dB × (0 + 105) dB is also guaranteed. –11– AD8309 A fully-programmable output interface is provided for the hardlimited signal, permitting the user to establish the optimal output current from its differential current-mode output. Its magnitude is determined by the resistor RLIM placed between LMDR (Pin 9) and ground, across which a nominal bias voltage of ~400 mV appears. Using RLIM = 200 Ω, this dc bias current, which is commutated alternately to the output pins, LMHI and LMLO, by the signal, is 2 mA. (The total supply current is somewhat higher). Input Interface Figure 27 shows the essentials of the signal input interface. The parasitic capacitances to ground are labeled CP; the differential input capacitance, CD, mainly due to the diffusion capacitance of Q1 and Q2. In most applications both input pins are accoupled. The switch S closes when Enable is asserted. When disabled, the inputs float, bias current IE is shut off, and the coupling capacitors remain charged. If the log amp is disabled for long periods, small leakage currents will discharge these capacitors. If they are poorly matched, charging currents at power-up can generate a transient input voltage which may block the lower reaches of the dynamic range until it has become much less than the signal. These currents may readily be converted to voltage form by the inclusion of load resistors, which will typically range from a few tens of ohms at 500 MHz to as high as 2 kΩ in lower frequency applications. Alternatively, a resonant load may be used to extract the fundamental signal and modulation sidebands, minimizing the out-of-band noise. A transformer or impedance matching network may also be used at this output. The peak voltage swing down from the supply voltage may be 1.2 V, before the output transistors go into saturation. (The Applications section provides further information on the use of this interface). In most applications, the input signal will be single-sided, and may be applied to either Pin 4 or 5, with the remaining pin accoupled to ground. Under these conditions, the largest input signal that can be handled is –3 dBV (sine amplitude of 1 V) when operating from a 3 V supply ; a +3 dBV input may be handled using a supply of 4.5 V or greater. When using a fullybalanced drive, the +3 dBV level may be achieved for the supplies down to 2.7 V and +9 dBV using >4.5 V. For frequencies in the range 10 MHz to 200 MHz these high drive levels are easily achieved using a matching network (see later). Using such a network, having an inductor at the input, the input transient is eliminated. The supply current for all sections except the limiter output stage, and with no load attached to the RSSI output, is nominally 16 mA at TA = 27°C, substantially independent of supply voltage. It varies in direct proportion to the absolute temperature (PTAT). The RSSI load current is simply the voltage at VLOG divided by the load resistance (e.g., 2.4 mA max in a 1 kΩ load). The limiter supply current is 1.1 times that flowing in RLIM. The AD8309 may be enabled/disabled by a CMOScompatible level at ENBL (Pin 8). VPS1 TO STAGES 1 THRU 5 67V S 1.78V 3.65kV In the following simplified interface diagrams, the components denoted with an uppercase “R” are thin-film resistors having a very low temperature-coefficient of resistance and high linearity under large-signal conditions. Their absolute value is typically within ± 20%. Capacitors denoted using an uppercase “C” have a typical tolerance of ± 15% and essentially zero temperature or voltage sensitivity. Most interfaces have additional small junction capacitances associated with them, due to active devices or ESD protection; these may be neither accurate nor stable. Component numbering in each of these interface diagrams is local. CC INHI SIGNAL INPUT CC INLO 1.725V RIN = 1kV 67V 3.65kV IB = 15mA CD 2.5pF Q1 TO 2ND STAGE 20e RIN = 3kV 1.725V Q2 20e 2.6kV GAIN BIAS 1.26V (TOP-END DETECTORS) CP CP 130V 3.4mA PTAT COMM Figure 27. Signal Input Interface Enable Interface The chip-enable interface is shown in Figure 26. The current in R1 controls the turn-on and turn-off states of the band-gap reference and the bias generator, and is a maximum of 100 µA when Pin 8 is taken to 5 V. Left unconnected, or at any voltage below 1 V, the AD8309 will be disabled, when it consumes a sleep current of much less than 1 µA (leakage currents only); when tied to the supply, or any voltage above 2 V, it will be fully enabled. The internal bias circuitry requires approximately 300 ns for either OFF or ON, while a delay of some 6 µs is required for the supply current to fall below 10 µA. ENBL R1 60kV TO BIAS ENABLE 1.3kV 50kV 4kV COMM Limiter Output Interface The simplified limiter output stage is shown in Figure 28. The bias for this stage is provided by a temperature-stable reference voltage of nominally 400 mV which is forced across the external resistor RLIM connected from Pin 9 (LMDR, or limiter drive) by a special op amp buffer stage. The biasing scheme also introduces a slight “lift” to this voltage to compensate for the finite current gain of the current source Q3 and the output transistors Q1 and Q2. A maximum current of 10 mA is permissible (RLIM = 40 Ω). In special applications, it may be desirable to modulate the bias current; an example of this is provided in the Applications section. Note that while the bias currents are temperature stable, the ac gain of this stage will vary with temperature, by –6 dB over a 120°C range. A pair of supply and temperature stable complementary currents is generated at the differential output LMHI and LMLO (Pins 12 and 13), having a square wave form with rise and fall times of typically 0.4 ns, when load resistors of 50 Ω are used. The voltage at these output pins may swing to 1.2 V below the supply voltage applied to VPS2 (Pin 15). Figure 26. Enable Interface –12– REV. B AD8309 Because of the very high gain bandwidth product of this amplifier considerable care must be exercised in using the limiter outputs. The minimum necessary bias current and voltage swings should be used. These outputs are best utilized in a fullydifferential mode. A flux-coupled transformer, a balun, or an output matching network can be selected to transform these voltages to a single-sided form. Equal load resistors are recommended, even when only one output pin is used, and these should always be returned to the same well decoupled node on the PC board. When the AD8309 is used only to generate an RSSI output, the limiter should be completely disabled by omitting RLIM and strapping LMHI and LMLO to VPS2. VPS2 1.3kV LMHI LMLO 1.3kV Q1 Q2 4e 400mV OA ZERO-TC Q3 2.6kV 1.3kV 1.3kV COM1 RLIM Figure 28. Limiter Output Interface RSSI Output Interface The outputs from the ten detectors are differential currents, having an average value that is dependent on the signal input level, plus a fluctuation at twice the input frequency. The currents are summed at the internal nodes LGP and LGN shown in Figure 29. A further current ITC is added to LGP, to position the intercept to –108 dBV, by raising the RSSI output voltage for zero input, and to provide temperature compensation , resulting in a stable intercept. For zero signal conditions, all the detector output currents are equal. For a finite input, of either polarity, their difference is converted by the output interface to a single-sided voltage nominally scaled 20 mV/dB (400 mV per decade), at the output VLOG (Pin 16). This scaling is controlled by a separate feedback stage, having a tightly controlled transconductance. A small uncertainty in the log slope and intercept remains (see Specifications); the intercept may be adjusted (see Applications). VPS2 SUMMED 1.3kV DETECTOR OUTPUTS 1.3kV ISOURCE >50mA ON DEMAND FLTR C1 3.5pF LGP LGN IT 3.3kV VLOG 250ms 3.3kV ISINK FIXED 1mA CF VLOG 20mV/dB 125mA COMM TRANSCONDUCTANCE DETERMINES SLOPE Figure 29. Simplified RSSI Output Interface REV. B 12.7 × 10–10 – 3.5 pF ; f LP f LP = 12.7 × 10−6 CF + 3.5 pF (7) Using a load resistance of 50 Ω or greater, and at any temperature, the peak output voltage may be at least 2.4 V when using a supply of 4.5 V, and at least 2.1 V for a 3 V supply, which are consistent with the maximum permissible input levels. The incremental output resistance is approximately 0.3 Ω at low frequencies, rising to 1 Ω at 150 kHz and 18 Ω at very high frequencies. USING THE AD8309 LMDR CURRENT MIRROR CF = The output is unconditionally stable with load capacitance, but it should be noted while the peak sourcing current is over 100 mA, and able to rapidly charge even large capacitances, the internally provided sinking current is only 1 mA. Thus, the fall time from the 2 V level will be as long as 2 µs for a 1 nF load. This may be reduced by adding a grounded load resistance. 4e FROM FINAL LIMITER STAGE The RSSI output bandwidth, fLP, is nominally 3.5 MHz. This is controlled by the compensation capacitor C1, which may be increased by adding an external capacitor, CF, between FLTR (Pin 10) and VLOG (Pin 16). An external 33 pF will reduce fLP to 350 kHz, while 360 pF will set it to 35 kHz, in each case with an essentially one-pole response. In general, the relationships are: The AD8309 exhibits very high gain from 1 MHz to over 1 GHz, at which frequency the gain of the main path is still over 65 dB. Consequently, it is susceptible to all signals within this very broad frequency range which find their way to the input terminals. It is important to remember that these are quite indistinguishable from the “wanted” signal, and will have the effect of raising the apparent noise floor (that is, lowering the useful dynamic range). Therefore, while the signal of interest may be an IF of, say, 200 MHz, any of the following could easily be larger than this signal at the lower extremities of its dynamic range: a 60 Hz hum, picked up due to poor grounding techniques; spurious coupling from digital logic on the same PC board; a strong EMI source; etc. Very careful shielding is essential to guard against such unwanted signals, and also to minimize the likelihood of instability due to HF feedback from the limiter outputs to the input. With this in mind, the minimum possible limiter gain should be used. Where only the logarithmic amplifier (RSSI) function is required, the limiter should be disabled by omitting RLIM and tying the outputs LMHI and LMLO directly to VPS2. A good ground plane should be used to provide a low impedance connection to the common pins, for the decoupling capacitor(s) used at VPS1 and VPS2, and at the output ground. It is inadvisable to assume that any ground plane is an equipotential, however, and neither of the signal inputs should be accoupled directly to it, but kept separate, being returned instead to the “low” associated with the source. This requires isolating the “low”’ side of an input connector with a small resistance to the ground plane. Note that COM2 is a special ground pin serving just the RSSI output. The voltages at the two supply pins should not be allowed to differ greatly; up to 500 mV is permissible It is desirable to allow VPS1 to be slightly more negative than VPS2. When the primary supply is greater than 2.7 V, the decoupling resistors R1 and R2 may be increased to improve the isolation and lower dissipation in the IC. However, since VPS2 supports the RSSI –13– AD8309 2.5 The four pins labeled PADL tie down directly to the metallic lead frame, and are thus connected to the back of the chip. The process on which the AD8309 is fabricated uses a bonded-wafer technique to provide a silicon-on-insulator isolation, and there is no junction or other dc path from the back side to the circuitry on the surface. These paddle pins must be connected directly to the ground plane using the shortest possible lead lengths to minimize inductance. 2.0 RSSI OUTPUT – V load current, which may be large, the value of R2 should take this into account. Figure 30 shows the connections required for most applications. The inputs are ac-coupled by C1 and C2, which normally should have the same value, say, CO. The coupling time constant is ROCO /2, where RO = RS + RIN, thus forming a high pass corner with a 3 dB attenuation at fHP = 1/(π RT CC ). In highfrequency applications, fHP should be chosen as large as possible, to minimize the coupling of unwanted signals. On the other hand, in low frequency applications, a simple RC network forming a low-pass filter should be added at the input for the same reason. 1 COM2 SEE TEXT FOR MORE ABOUT DECOUPLING 0.1mF VLOG 16 2 VPS1 VPS2 15 3 PADL PADL 14 LMHI 4 INHI LMHI 13 RLOAD 5 INLO LMLO 12 RLOAD 6 PADL PADL 11 7 COM1 FLTR 10 NC RLIM LMDR 9 40 5 4 3 5MHz DYNAMIC RANGE 5MHz 50MHz 100MHz 200MHz 1dB 85 91 97 96 3dB 93 99 103 102 1 50MHz 0 –1 100MHz 200MHz –3 LMLO 8 ENBL –4 –5 –90 –80 –70 –60 –50 –40 –30 –20 –10 0 INPUT LEVEL – dBm Re 50V 52.3V ENABLE 20 –2 RT 4.7nH –60 –40 –20 0 INPUT LEVEL – dBm Re 50V 2 RSSI AD8309 –80 Figure 31. RSSI Output vs. Input Level at TA = +25 °C, for Frequencies of 5 MHz, 50 MHz, 100 MHz and 200 MHz VS R2 10V 0.1mF C1 FOR BROADBAND 50V TERMINATION TO 1GHz 1.0 0 –100 ERROR – dB R1 10V C2 1.5 0.5 Basic Connections SIGNAL INPUTS 100MHz 50MHz 200MHz 5MHz 10 20 30 Figure 32. Log Linearity vs. Input Level at TA = +25 °C, for Frequencies of 5 MHz, 50 MHz, 100 MHz and 200 MHz NC = NO CONNECT Input Matching Figure 30. Basic Connections Where either a higher sensitivity or a better high frequency match is required, an input matching network is valuable. Using a flux-coupled transformer to achieve the impedance transformation also eliminates the need for coupling capacitors, lowers any dc offset voltages generated directly at the input, and usefully balances the drives to INHI and INLO, permitting full utilization of the unusually large input voltage capacity of the AD8309. Where it is necessary to terminate the source at a low impedance, the resistor RT should be added, with allowance for the shunting effect of the 1 kΩ input resistance (RIN) of the AD8309. For example, to terminate a 50 Ω source, a 52.3 Ω␣ resistor should be used for signal frequencies up to about 50 MHz. The termination means may be placed either at the input or at the log amp side of the coupling capacitors. In the former case smaller capacitors can be used for a given frequency range; in the latter case, the dc resistance is lowered directly at the log amp inputs, which helps to keep offsets to a minimum. At higher frequencies, the reactance of the 2.5 pF input capacitance must be accounted for. A 4.7 nH inductor in series with the 52.3 Ω termination resistor provides an essentially flat 50 Ω input impedance to 1 GHz. An impedance-transforming network is preferably used to provide a 50 Ω interface, since this also introduces a balanced voltage gain of typically 13 dB and the AD8309 has a very high capacity for large input voltages. Figure 31 shows the output versus the input level, with the axis marked in dBm (correct only when terminated in 50 Ω), for sine inputs at 5 MHz, 50 MHz, 100 MHz and 200 MHz. Figure 32 shows the typical logarithmic linearity (law conformance) under the same conditions. The choice of turns ratio will depend somewhat on the frequency. At frequencies below 30 MHz, the reactance of the input capacitance is much higher than the real part of the input impedance. In this frequency range, a turns ratio of 2:9 will lower the effective input impedance to 50 Ω while raising the input voltage by 13 dB. However, this does not lower the effect of the short circuit noise voltage by the same factor, since there will be a contribution from the input noise current. Thus, the total noise will be reduced by a smaller factor. The intercept at the primary input will be lowered to –120 dBV (–107 dBm). Impedance matching and drive balancing using a flux-coupled transformer is useful whenever broadband coupling is required. However, this may not always be convenient. At high frequencies, it will often be preferable to use a narrow-band matching network, as shown in Figure 33, which has several advantages. First, the same voltage gain can be achieved, providing increased –14– REV. B AD8309 sensitivity, but now a measure of selectively is simultaneously introduced. Second, the component count is low: two capacitors and an inexpensive chip inductor are needed. Third, the network also serves as a balun. Analysis of this network shows that the amplitude of the voltages at INHI and INLO are quite similar when the impedance ratio is fairly high (say, 50 Ω to 1000 Ω). 14 13 12 11 DECIBELS 10 VS 10V 10V 1 COM2 VPS2 15 3 PADL PADL 14 7 6 5 INPUT AT TERMINATION 3 RSSI 0.1mF 2 VPS1 9 8 4 VLOG 16 0.1mF GAIN 2 1 0 C1 = CM ZIN –1 60 AD8309 4 INHI LMHI 13 5 INLO LMLO 12 LIMITER OUTPUT LM C2 = CM PADL 11 7 COM1 FLTR 10 NC RLIM LMDR 9 130 140 150 Figure 34. Response of 100 MHz Matching Network Step 1: Tune Out CIN Figure 33. High Frequency Input Matching Network Figure 34 shows the response for a center frequency of 100 MHz. The response is down by 50 dB at one-tenth the center frequency, falling by 40 dB per decade below this. The very high frequency attenuation is relatively small, however, since in the limiting case it is determined simply by the ratio of the AD8309’s input capacitance to the coupling capacitors. Table I provides solutions for a variety of center frequencies fC and matching from impedances ZIN of nominally 50 Ω and 100 Ω. Exact values are shown, and some judgment is needed in utilizing the nearest standard values. At a center frequency fC, the shunt impedance of the input capacitance CIN can be made to disappear by resonating with a temporary inductor LIN, whose value is given by LIN = 1/{(2 π fC)2CIN} = 1010/fC2 fC MHz Match to 50 ⍀ (Gain = 13 dB) CM LM pF nH Match to 100 ⍀ (Gain = 10 dB) CM LM pF nH 10 10.7 15 20 21.4 25 30 35 40 45 50 60 80 100 120 150 200 250 300 350 400 450 500 140 133 95.0 71.0 66.5 57.0 47.5 40.7 35.6 31.6 28.5 23.7 17.8 14.2 11.9 9.5 7.1 5.7 4.75 4.07 3.57 3.16 2.85 100.7 94.1 67.1 50.3 47.0 40.3 33.5 28.8 25.2 22.4 20.1 16.8 12.6 10.1 8.4 6.7 5.03 4.03 3.36 2.87 2.52 2.24 2.01 4790 4460 3120 2290 2120 1790 1460 1220 1047 912 804 644 448 335 261 191 125 89.1 66.8 52.1 41.8 34.3 28.6 (8) when CIN = 2.5 pF. For example, at fC = 100 MHz, LIN = 1 µH. Step 2: Calculate CO and LO Now having a purely resistive input impedance, we can calculate the nominal coupling elements CO and LO, using CO = Table I. REV. B 90 100 110 120 FREQUENCY – MHz For other center frequencies and source impedances, the following method can be used to calculate the basic matching parameters. NC = NO CONNECT 3500 3200 2250 1660 1550 1310 1070 904 779 682 604 489 346 262 208 155 104 75.3 57.4 45.3 36.7 30.4 25.6 80 General Matching Procedure 6 PADL 8 ENBL 70 1 2 πfC (R IN RM ) ; LO = (R IN RM ) 2 πfC (9) For the AD8309, RIN is 1 kΩ. Thus, if a match to 50 Ω is needed, at fC = 100 MHz, CO must be 7.12 pF and LO must be 356 nH. Step 3: Split CO Into Two Parts Since we wish to provide the fully-balanced form of network shown in Figure 33, two capacitors C1 = C2 each of nominally twice CO, shown as CM in the figure, can be used. This requires a value of 14.24 pF in this example. Under these conditions, the voltage amplitudes at INHI and INLO will be similar. A somewhat better balance in the two drives may be achieved when C1 is made slightly larger than C2, which also allows a wider range of choices in selecting from standard values. For example, capacitors of C1 = 15 pF and C2 = 13 pF may be used (making CO = 6.96 pF). Step 4: Calculate LM The matching inductor required to provide both LIN and LO is just the parallel combination of these: LM = LINLO/(LIN + LO) (10) With LIN = 1 µH and LO = 356 nH, the value of LM to complete this example of a match of 50 Ω at 100 MHz is 262.5 nH. The nearest standard value of 270 nH may be used with only a slight loss of matching accuracy. The voltage gain at resonance depends only on the ratio of impedances, as is given by R IN GAIN = 20 log R S –15– RIN = 10 log RS (11) AD8309 Slope and Intercept Adjustment VS 10V The AD8309 provides limited opportunities for adjustment of its basic scaling parameters, which are controlled to within tight limits through robust design. In applications involving the observation of measured signal levels on a DVM a slope of 10 mV per decade is convenient: the reading is then directly in decibels, needing only the positioning of the decimal point. This may be simply achieved and at the same time trimmed to this exact value using the scheme shown in Figure 35. A large filter capacitor CFILT may be added as shown when the voltage is to be measured on a DVM; this lowers the fluctuation in the lowerorder display digits. VS 10V 10V 1 COM2 VLOG 16 2 VPS1 VPS2 15 3 PADL PADL 14 8.87kV AD8309 INPUT COUPLING 4 INHI LMHI 13 5 INLO LMLO 12 6 PADL PADL 11 7 COM1 FLTR 10 8 ENBL 0.1mF LMDR 9 0.1mF CFILT 10mF VR1 2kV SLOPE RSSI OUTPUT 10mV/dB 6 10% 8.87kV DVM LIMITER MAY BE DISABLED FOR RSSI ONLY MODE VPS2 15 3 PADL PADL 14 RSSI 0.1mF IN914 OR SIMILAR 4 INHI LMHI 13 5 INLO LMLO 12 6 PADL PADL 11 7 COM1 FLTR 10 VR2 2kV INTERCEPT 47kV 8 ENBL LMDR 9 9.6kV RLIM Figure 36. Trimming Intercept to –113 dBV ± 4 dB APPLICATIONS The AD8309 is a versatile and easily applied log-limiting amp. Being complete, it can be used with very few external components, and most applications can be accommodated using the simple connections shown in the preceding section. A few examples of more specialized applications are provided here. Log Amp with High Slope Voltage Where a higher RSSI slope voltage is required, and/or complete calibration with good temperature stability and minimal interaction between trims, the interface shown in Figure 37 may be used. Note that at 50 mV/dB, the full 100 dB dynamic range of the AD8309 requires a 5 V swing. This can be provided by a single supply operational amplifier having a rail-to-rail output stage and operating from a 6 V supply. Where a lower range is sufficient, or when using the 40 mV/dB option, a 5 V supply will be adequate. In this application, the supply current into the VPS2 pin is only slightly dependent on the current delivered to the load resistance, RL, so a voltage dropping resistor, RD, may be added to lower the supply to the AD8309, which can meet all of its specifications with a 2.7 V supply. The lower chip dissipation and the resulting reduction in operating temperature will minimize degradation of noise figure at high ambient temperatures. RD is calculated as follows: Figure 35. Trimming Slope to 10 mV/dB ± 10% The intercept can be adjusted by the use of the auxiliary circuit shown in Figure 36, without changing the slope, which remains 20 mV/dB. This circuit provides a range of about ± 4 dB on a nominal intercept of –113 dBV (–100 dBm), with a fairly low residual temperature sensitivity (+0.008 dB/°C). This is sufficient to absorb the worst-case intercept error in the AD8309 plus system-level gain errors. VR2 is adjusted while applying an accurately known CW signal near the lower end of the dynamic range, in order to minimize the effect of any residual uncertainty in the slope. For example, to position the intercept to exactly –100 dBm, a test level of –60 dBm may be applied and VR2 adjusted to produce a dc output of 40 dB above the intercept, which is +0.8 V. This trim can optionally be combined with the slope trim described above. VLOG 16 2 VPS1 AD8309 (12) For example, using an rms signal level of –40 dBm with a 70% modulation depth (M = 0.7), the decibel range is 15 dB, as the signal varies from –47.5 dBm to –32.5 dBm. The output would thus be adjusted to have a peak-to-peak amplitude of 150 mV. 1 COM2 0.1mF A precision attenuator or signal generator is required to provide several test levels at 10 dB intervals. The adjustment may also made using an AM modulated signal, at about the center of the dynamic range. For a modulation depth M, expressed as a fraction, the decibel range between the peaks and troughs over one cycle of the modulation period is given by ∆dB = 20 log10 (1+M)/(1–M) 10V RD = VS − 3 25 mA (13) RLIM ≥ 100 Ω which allows for operation at ambient temperatures up to +85°C. Table II may be used to select the component values for various different operating conditions. The slope adjustment range is ± 10% and the intercept adjustment range is ± 3 dB. Since the intercept offset bias is derived from the supply, there is a sensitivity to this voltage. Where supply stability is poor, a regulator may be needed to bias VR2 and R4. –16– REV. B AD8309 AD8309 SUPPLY DROPPED TO 3V RD = (VS –3V)/25mA VS 10V 10V 1 COM2 VLOG 16 2 VPS1 VPS2 15 3 PADL PADL 14 R1 R4 0.1mF AD8309 4 INHI LMHI 13 5 INLO LMLO 12 6 PADL PADL 11 VR1 2kV SLOPE INPUT 7 COM1 FLTR 10 8 ENBL LMDR 9 0.1mF AD8031 VR2 10kV INT R3 33.2kV RSSI R6 1.96kV R5 R2 0.1mF GND Figure 37. Buffered RSSI Output with Slope and Intercept Adjustments Table II. High Output Limiter Loading Slope Intercept R1 mV/dB dBV k⍀ R2 R4 R5 k⍀ k⍀ k⍀ VOUT (V) at –88 dBV +12 dBV 40 50 40 50 8.87 9.53 8.87 9.53 0.56 0.75 0.08 0.1 –102 –103 –90 –90 3.92 1.05 3.92 1.05 O/C O/C 20.5 15.4 1 1 1.05 1.07 4.56 5.75 4.08 5.10 Setting the Limiter Output Level The limiter output is a pair of differential currents of magnitude, IOUT, from high␣ impedance (open-collector) sources. These are converted to equal-amplitude voltages by supplyreferenced load resistors, RLOAD. The limiter output current is set by RLIM, the resistor connected between Pin 9 (LMDR) and ground depending on the application, the resulting voltage may be used in a fully balanced or unbalanced manner. It is good practice to retain the both resistors, whichever output mode is used. The unbalanced, or single sided mode, is more inclined to result in instabilities caused by the very high gain of the signal path. If the limiter output is not needed, LMDR should be left open with LMHI and LMLO being tied to VPS2. The AD8309 can generate a fairly large output power at its differential limiter output interface. This may be coupled into a 50 Ω grounded load using the narrow-band coupling network following similar lines to those provided for input matching. Alternatively, a flux-linked transformer, having a center-tapped primary, may be used. Even higher output powers can be obtained using emitter-followers. In Figure 38, the supply voltage to the AD8309 is dropped from 5 V to about 4.2 V, by the diode. This increases the available swing at each output to about 2 V. Taking both outputs differentially, a square wave output of 4 V p-p can be generated. IN914 10V RLOAD 10V 1 COM2 VLOG 16 2 VPS1 VPS2 15 3 PADL PADL 14 0.1mF RLOAD RSSI 0.1mF SET RL = 5*RLIM AD8309 The limiter output current is set by the equation: IOUT = –400 mV/RLIM and has an absolute accuracy of ± 5%. 4 INHI LMHI 13 5 INLO LMLO 12 5V TO 3V 6 PADL PADL 11 DIFFERENTIAL OUTPUT = 4V pk-pk 7 COM1 FLTR 10 8 ENBL LMDR 9 3V TO 5V RLIM Figure 38. Increasing Limiter Output Voltage The voltage on each of the limiter pins will be given by: VLIM = VS – 400 mV × RLOAD /RLIM The limiter current may be set as high as 10 mA, which requires RLIM to be 40 ohms, and can be optionally increased somewhat beyond this level. It is inadvisable, however, to use high bias currents, since the gain of this wide bandwidth signal path is proportional to it, and the risk of instability is elevated as RLIM is reduced (recommended value is 400 Ω). The limiter output is specified for input levels between –78 dBV and +9 dBV. The output of the limiter will be unstable for levels below –78 dBV (–65 dBm). REV. B +5V APPROX. 4.2V When operating at high output power levels and high frequencies, very careful attention must be paid to the issue of stability. Oscillation is likely to be observed when the input signal level is low, due to the extremely high gain-bandwidth product of the AD8309 under such conditions. These oscillations will be less evident when signal-balancing networks are used, operating at frequencies below 200 MHz, and they will generally be fully quenched by the signal at input levels of a few dB above the noise floor. –17– AD8309 the case of the AD8309 being alternately fed by an unmodulated sine wave and by a single CDMA channel of the same rms power. The AD8309’s output voltage will differ by the equivalent of 3.55 dB (71 mV) over the complete dynamic range of the device (the output for a CDMA input being lower). Modulated Limiter Output The limiter output stage of the AD8309 also provides an analog multiplication capability: the amplitude of the output square wave can be controlled by the current withdrawn from LMDR (Pin 9). An analog control input of 0 V to +1 V is used to generate an exactly-proportional current of 0 mA to 10 mA in the npn transistor, whose collector is held at a fixed voltage of ∼400 mV by the internal bias in the AD8309. When the input signal is above the limiting threshold, the output will then be a squarewave whose amplitude is proportional to the control bias. Table III shows the correction factors that should be applied to measure the rms signal strength of a various signal types. A sine wave input is used as a reference. To measure the rms power of a square wave, for example, the mV equivalent of the dB value given in the table (20 mV/dB times 3.01 dB) should be subtracted from the output voltage of the AD8309. VS 10V 10V 1 COM2 VLOG 16 RSSI Table III. Shift in AD8309 Output for Signals with Differing Crest Factors 0.1mF 2 VPS1 VPS2 15 3 PADL PADL 14 0.1mF 0V TO +1V AD8309 0.1mF 4 INHI LMHI 13 5 INLO LMLO 12 6 PADL PADL 11 7 COM1 FLTR 10 8 ENBL LMDR 9 Correction Factor (Add to Output Reading) Sine Wave Square Wave or DC Triangular Wave GSM Channel (All Time Slots On) CDMA Channel PDC Channel (All Time Slots On) Gaussian Noise 0 dB –3.01 dB +0.9 dB +0.55 dB +3.55 dB +0.58 dB +2.51 dB 8.2kV VARIABLE OUTPUT 2N3904 AD8031 1.8kV 18V 0mA TO 10mA Signal Type Figure 39. Variable Limiter Output Programming Evaluation Board Effect of Waveform Type on Intercept An evaluation board, carefully laid out and tested to demonstrate the specified high speed performance of the AD8309 is available. Figure 40 shows the schematic of the evaluation board which fairly closely follows the basic connections schematic shown in Figure 30. For ordering information, please refer to the Ordering Guide. Links, switches and component settings for different setups are described in Table IV. The AD8309 fundamentally responds to voltage and not to power. A direct consequence of this characteristic is that input signals of equal rms power, but differing crest factors, will produce different results at the log amp’s output. The effect of differing signal waveforms is to shift the effective value of the log amp’s intercept. Graphically, this looks like a vertical shift in the log amp’s transfer function. The device’s logarithmic slope however is not affected. For example, consider R3 0V R2 4.7V +VS C1 0.01mF SIG INHI C3 0.1mF R1 0V 2 VPS1 VPS2 15 3 PADL PADL 14 4 INHI R5 4.7V R6 402V C4 0.1mF R7 402V +VS C5 0.01mF LMHI 13 LMHI C7 (OPEN) 5 INLO LMLO 12 6 PADL PADL 11 VRSSI L1 (OPEN) LMLO C2 0.01mF A EXT ENABLE VLOG 16 AD8309 R/L 52.3V SIG INLO 1 COM2 R4 (OPEN) 7 COM1 FLTR 10 8 ENBL LMDR 9 C6 0.01mF LK1 R8 402V B Figure 40. Evaluation Board Schematic –18– REV. B AD8309 Table IV. Evaluation Board Setup Options Component Function Default Condition SW1 Device Enable. When in position A, the ENBL pin is connected to +VS and the AD8309 is in normal operating mode. In position B, the ENBL pin is connected to an SMA connector labeled Ext Enable. An applied signal can be applied to this connector to enable/disable the AD8309. If left open, the ENBL pin will float to ground putting the device in power-down mode. SW1 = A R1 This pad is used to ac-couple to ground for single-ended input drive. To drive the AD8309 differentially, R1 should be removed. R1 = 0 Ω R/L, C1, C2 Input Interface. The 52.3 Ω resistor in position R/L along with C1 and C2 create a high pass input filter whose corner frequency (640 kHz) is equal to 1/(πRC), where C = C1 = C2 and R is the parallel combination of 52.3 Ω and the AD8309’s input impedance of 1000 Ω. Alternatively, the 52.3 Ω resistor can be replaced by an inductor to form an input matching network. See Input Matching Network section for more details. R/L= 52.3 Ω C1 = C2 = 0.01 µF R3/R4 Slope Adjust. A simple slope adjustment can be implemented by adding a resistive divider at the VLOG output. R3 and R4, whose sum should be about 1 kΩ, and never less than 40 Ω (see specs), set the slope according to the equation: Slope = 20 mV/dB × R4/(R3+R4). R3 = 0 Ω R4 = ⬁ L1, C5, C6 Limiter Output Coupling. C5 and C6 ac-couple the limiter’s differential outputs. By adjusting these values and installing an inductor in L1, an output matching network can be implemented. L1 = Open C5 = 0.01 µF C6 = 0.01 µF R8, LK1 Limiter Output Current. With LK2 installed, R8 enables and sets the limiter output current. The limiter’s output current is set according to the equation (IOUT = 400 mV/R8). The limiter current can be as high as 10 mA (R8 = 40 Ω). To disable the limiter (recommended if the limiter is not being used), LK3 should be removed. LK1 Installed. R8 = 402 Ω C7 RSSI Bandwidth Adjust. The addition of C7 will lower the RSSI bandwidth of the VLOG output according to the equation: fCORNER = 12.7 × 10–6/(CFILT + 3.5 pF). C7 = Open Figure 42. Layout of Power Layer Figure 41. Layout of Signal Layer REV. B –19– C3440b–0–8/99 AD8309 Figure 43. Signal Layer Silkscreen Figure 44. Power Layer Silkscreen OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 16-Lead TSSOP (RU-16) 0.201 (5.10) 0.193 (4.90) 9 0.256 (6.50) 0.246 (6.25) 0.177 (4.50) 0.169 (4.30) 16 1 8 0.0256 SEATING (0.65) PLANE BSC 0.0433 (1.10) MAX 0.0118 (0.30) 0.0075 (0.19) 8° 0° 0.0079 (0.20) 0.0035 (0.090) –20– PRINTED IN U.S.A. PIN 1 0.006 (0.15) 0.002 (0.05) 0.028 (0.70) 0.020 (0.50) REV. B

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