LINER LT1676 Wide input range, high efficiency, step-down switching regulator Datasheet

LT1676
Wide Input Range,
High Efficiency, Step-Down
Switching Regulator
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FEATURES
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DESCRIPTIO
Wide Input Range: 7.4V to 60V
700mA Peak Switch Current Rating
Adaptive Switch Drive Maintains Efficiency at High
Load Without Pulse Skipping at Light Load
True Current Mode Control
100kHz Fixed Operating Frequency
Synchronizable to 250kHz
Low Supply Current in Shutdown: 30µA
Available in 8-Pin SO and PDIP Packages
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APPLICATIO S
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Automotive DC/DC Converters
Telecom 48V Step-Down Converters
Cellular Phone Battery Charger Accessories
IEEE 1394 Step-Down Converters
The LT®1676 is a wide input range, high efficiency Buck
(step-down) switching regulator. The monolithic die includes all oscillator, control and protection circuitry. The
part can accept input voltages as high as 60V and contains
an output switch rated at 700mA peak current. Current
mode control offers excellent dynamic input supply rejection and short-circuit protection.
The LT1676 contains several features to enhance efficiency. The internal control circuitry is normally powered
via the VCC pin, thereby minimizing power drawn directly
from the VIN supply (see Applications Information). The
action of the LT1676 switch circuitry is also load dependent. At medium to high loads, the output switch circuitry
maintains high rise time for good efficiency. At light loads,
rise time is deliberately reduced to avoid pulse skipping
behavior.
The available SO-8 package and 100kHz switching frequency allow for minimal PC board area requirements.
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATIO
Efficiency vs VIN and ILOAD
VIN
8V TO 50V
5
+
39µF
63V
VIN
2
SHDN
VCC
3
VSW
80
220µH*
+
MBR160
LT1676
100µF
10V
5V
400mA
36.5k
1%
7
6
FB
8
VC
SYNC
2200pF
GND
100pF
22k
12.1k
1%
4
70
EFFICIENCY (%)
1
90
60
50
40 VIN = 12V
VIN = 24V
30
VIN = 36V
VIN = 48V
20
*65T #30 ON MAGNETICS
MPP #55030
1676 F01
1
10
100
1000
ILOAD (mA)
1676 TA01
Figure 1
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LT1676
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
Supply Voltage ........................................................ 60V
Switch Voltage ......................................................... 60V
SHDN, SYNC Pin Voltage ........................................... 7V
VCC Pin Voltage ....................................................... 30V
FB Pin Voltage ........................................................... 3V
Operating Junction Temperature Range
LT1676C ................................................ 0°C to 125°C
LT1676I ............................................ – 40°C to 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
SHDN 1
8
VC
VCC 2
7
FB
VSW 3
6
SYNC
GND 4
5
VIN
N8 PACKAGE
8-LEAD PDIP
S8 PACKAGE
8-LEAD PLASTIC SO
LT1676CN8
LT1676CS8
LT1676IN8
LT1676IS8
S8 PART MARKING
TJMAX = 125°C, θJA = 130°C/ W (N8)
TJMAX = 125°C, θJA = 110°C/ W (S8)
1676
1676I
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VIN = 48V, VSW open, VCC = 5V, VC = 1.4V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
6.7
7.0
7.4
V
V
620
800
900
µA
µA
3.2
4.0
5.0
mA
mA
2.8
3.1
V
30
50
75
µA
µA
1.240
1.255
1.265
V
V
Power Supplies
VIN(MIN)
Minimum Input Voltage
●
IVIN
VIN Supply Current
VC = 0V
●
IVCC
VCC Supply Current
VC = 0V
●
VVCC
VCC Dropout Voltage
(Note 2)
Shutdown Mode IVIN
VSHDN = 0V
●
●
Feedback Amplifier
VREF
Reference Voltage
IIN
FB Pin Input Bias Current
gm
Feedback Amplifier Transconductance
ISRC, ISNK
VCL
●
1.225
1.215
600
1500
nA
650
●
400
200
1000
1500
µmho
µmho
60
45
100
●
170
220
µA
µA
∆lc = ±10µA
Feedback Amplifier Source or Sink Current
Feedback Amplifier Clamp Voltage
Reference Voltage Line Regulation
2.0
12V ≤ VIN ≤ 60V
0.01
●
Voltage Gain
V
%/V
200
600
V/V
1.0
1.5
V
0.55
0.70
1.0
A
0.9
1.1
1.25
Output Switch
VON
Output Switch On Voltage
ISW = 0.5A
ILIM
Switch Current Limit
(Note 3)
●
Current Amplifier
Control Pin Threshold
Control Voltage to Switch Transconductance
2
Duty Cycle = 0%
2
V
A/V
LT1676
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VIN = 48V, VSW open, VCC = 5V, VC = 1.4V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
90
85
100
●
110
115
kHz
kHz
●
85
90
%
300
ns
VC Pin Boost Threshold
1.35
V
dV/dt Below Threshold
0.2
V/ns
dV/dt Above Threshold
1.6
V/ns
Timing
f
Switching Frequency
Maximum Switch Duty Cycle
tON(MIN)
Minimum Switch On Time
High dV/dt Mode, RL = 50Ω (Note 4)
Boost Operation
Sync Function
Minimum Sync Amplitude
●
Synchronization Range
●
1.5
130
SYNC Pin Input R
2.2
V
250
kHz
40
kΩ
SHDN Pin Function
VSHDN
Shutdown Mode Threshold
0.5
0.2
●
ISHDN
0.8
Upper Lockout Threshold
Switching Action On
1.260
Lower Lockout Threshold
Switching Action Off
1.245
Shutdown Pin Current
VSHDN = 0V
VSHDN = 1.25V
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: Control circuitry powered from VCC.
12
2.5
V
V
V
V
20
10
µA
µA
Note 3: Switch current limit is DC trimmed and tested in production.
Inductor dl/dt rate will cause a somewhat higher current limit in actual
application.
Note 4: Minimum switch on time is production tested with a 50Ω resistive
load to ground.
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TYPICAL PERFORMANCE CHARACTERISTICS
Minimum Input Voltage vs
Temperature
Switch-On Voltage vs
Switch Current
7.4
1.50
7.2
1.25
Switch Current Limit vs
Duty Cycle
1000
7.0
6.8
6.6
6.4
25°C
– 55°C
1.00
0.75
125°C
0.50
0.25
6.2
6.0
–50 –25
SWITCH CURRENT LIMIT (mA)
SWITCH VOLTAGE (V)
INPUT VOLTAGE (V)
TA = 25°C
50
25
75
0
TEMPERATURE (°C)
100
125
LT1676 G01
0
0
100
200 300 400 500 600
SWITCH CURRENT (mA)
700
1676 G02
800
600
400
200
0
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
1676 G03
3
LT1676
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TYPICAL PERFORMANCE CHARACTERISTICS
SHDN Pin Shutdown Threshold
vs Temperature
SHDN Pin Input Current
vs Voltage
SHDN PIN INPUT CURRENT (µA)
700
600
500
400
300
200
–50 –25
1.30
0
1.28
–5
–10
25°C
–55°C
125°C
–15
100
125
0
1
3
4
2
SHDN PIN VOLTAGE (V)
LT1676 G04
102
100
98
96
50
100
75
1.75
1.50
1.25
1.00
0
TEMPERATURE (°C)
25
50
75
100
1676 G07
1.2
CLAMP
VOLTAGE
BOOST
THRESHOLD
SWITCHING
THRESHOLD
1.0
0.8
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
LT1676 G10
4
400
300
200
100
0
–50 –25
125
50
25
75
0
TEMPERATURE (°C)
100
125
1676 G09
Feedback Amplifier Output
Current vs FB Pin Voltage
1.6
1.4
VIN = 48V
RL = 50Ω
FB =
1676 G08
FEEDBACK AMPLIFIER OUTPUT CURRENT (µA)
VC PIN VOLTAGE (V)
1.8
500
TEMPERATURE (°C)
2.0
125
600
2.00
VC Pin Switching Threshold,
Boost Threshold, Clamp Voltage
vs Temperature
100
Switch Minimum On-Time
vs Temperature
2.25
0.75
–50 –25
125
2.2
50
25
75
0
TEMPERATURE (°C)
LT1676 G06
SWITCH MINIMUM ON-TIME (ns)
MINIMUM SYNCHRONIZATION VOLTAGE (V)
SWITCHING FREQUENCY (kHz)
104
25
5
Minimum Synchronization Voltage
vs Temperature
106
0
1.24
1676 G05
Switching Frequency
vs Temperature
94
–50 –25
LOWER THRESHOLD
1.20
–50 –25
–20
50
25
75
0
TEMPERATURE (°C)
UPPER THRESHOLD
1.26
1.22
Error Amplifier Transconductance
vs Temperature
750
100
25°C
–55°C
125°C
50
TRANSCONDUCTANCE (µmho)
SHDN PIN VOLTAGE (mV)
800
5
SHDN PIN VOLTAGE (V)
900
SHDN Pin Lockout Thresholds
vs Temperature
0
–50
–100
700
650
600
550
500
450
–150
1.0
1.1
1.3
1.4
1.2
FB PIN VOLTAGE (V)
1.5
1676 G11
400
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
LT1676 G12
LT1676
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PIN FUNCTIONS
SHDN (Pin 1): When pulled below the shutdown mode
threshold, nominally 0.30V, this pin turns off the regulator and reduces VIN input current to a few tens of microamperes (shutdown mode).
When this pin is held above the shutdown mode threshold, but below the lockout threshold, the part will be
operational with the exception that output switching
action will be inhibited (lockout mode). A user-adjustable
undervoltage lockout can be implemented by driving this
pin from an external resistor divider to VIN. This action is
logically “ANDed” with the internal UVLO, set at nominally
6.7V, such that minimum VIN can be increased above
6.7V, but not decreased (see Applications Information).
short as possible to minimize electromagnetic radiation
and voltage spikes.
GND (Pin 4): This is the device ground pin. The internal
reference and feedback amplifier are referred to it. Keep
the ground path connection to the FB divider and the VC
compensation capacitor free of large ground currents.
VIN (Pin 5): This is the high voltage supply pin for the
output switch. It also supplies power to the internal control
circuitry during start-up conditions or if the VCC pin is left
open. A high quality bypass capacitor that meets the input
ripple current requirements is needed here. (See Applications Information.)
If unused, this pin should be left open. However, the high
impedance nature of this pin renders it susceptible to
coupling from the high speed VSW node, so a small
capacitor to ground, typically 100pF or so is recommended when the pin is left “open.”
SYNC (Pin 6): Pin used to synchronize internal oscillator
to the external frequency reference. It is directly logic
compatible and can be driven with any signal between
10% and 90% duty cycle. The sync function is internally
disabled if the FB pin voltage is low enough to cause
oscillator slowdown. If unused, this pin should be grounded.
VCC (Pin 2): This pin is used to power the internal control
circuitry off of the switching supply output. Proper use of
this pin enhances overall power supply efficiency. During
start-up conditions, internal control circuitry is powered
directly from VIN. If the output capacitor is located more
than an inch from the VCC pin, a separate 0.1µF bypass
capacitor to ground may be required right at the pin.
FB (Pin 7): This is the inverting input to the feedback
amplifier. The noninverting input of this amplifier is internally tied to the 1.24V reference. This pin also slows down
the frequency of the internal oscillator when its voltage is
abnormally low, e.g., 2/3 of normal or less. This feature
helps maintain proper short-circuit protection.
VSW (Pin 3): This is the emitter node of the output switch
and has large currents flowing through it. This node
moves at a high dV/dt rate, especially when in “boost”
mode. Keep the traces to the switching components as
VC (Pin 8): This is the control voltage pin which is the
output of the feedback amplifier and the input of the
current comparator. Frequency compensation of the overall loop is effected by placing a capacitor, (or in most cases
a series RC combination) between this node and ground.
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TIMING DIAGRAMS
High dV/dt Mode
Low dV/dt Mode
VIN
VIN
VSW
VSW
0
0
SWDR
SWDR
SWON
SWON
BOOST
BOOST
SWOFF
SWOFF
1676 TD01
1676 TD02
5
LT1676
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BLOCK DIAGRA
VCC 2
R1
5
VIN
3
VSW
RSENSE
VBG
SHDN 1
BIAS
VB
OSC
SYNC 6
LOGIC
Q3
I
COMP
SWDR
Q4
SWDR
SWON
BOOST
SWOFF
Q2
Q1
D1
GND 4
SWON
I
BOOST
COMP
I
I
VC 8
FB
AMP
FB 7
gm
VTH
BOOST
I
SWOFF
Q5
VBG
1676 BD
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OPERATIO
The LT1676 is a current mode switching regulator IC that
has been optimized for high efficiency operation in high
input voltage, low output voltage Buck topologies. The
Block Diagram shows an overall view of the system.
Several of the blocks are straightforward and similar to
those found in traditional designs, including: Internal Bias
Regulator, Oscillator and Feedback Amplifier. The novel
portion includes an elaborate Output Switch section and
Logic Section to provide the control signals required by
the switch section.
The LT1676 operates much the same as traditional
current mode switchers, the major difference being its
specialized output switch section. Due to space constraints, this discussion will not reiterate the basics of
current mode switcher/controllers and the “Buck” topology. A good source of information on these topics is
Application Note 19.
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Output Switch Theory
One of the classic problems in delivering low output
voltage from high input voltage at good efficiency is that
minimizing AC switching losses requires very fast voltage (dV/dt) and current (dI/dt) transition at the output
device. This is in spite of the fact that in a bipolar
implementation, slow lateral PNPs must be included in
the switching signal path.
Fast positive-going slew rate action is provided by lateral
PNP Q3 driving the Darlington arrangement of Q1 and Q2.
The extra β available from Q2 greatly reduces the drive
requirements of Q3.
Although desirable for dynamic reasons, this topology
alone will yield a large DC forward voltage drop. A second
lateral PNP, Q4, acts directly on the base of Q1 to reduce
the voltage drop after the slewing phase has taken place.
To achieve the desired high slew rate, PNPs Q3 and Q4 are
“force-fed” packets of charge via the current sources
controlled by the boost signal.
LT1676
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OPERATIO
Please refer to the High dV/dt Mode Timing Diagram. A
typical oscillator cycle is as follows: The logic section first
generates an SWDR signal that powers up the current
comparator and allows it time to settle. About 1µs later, the
SWON signal is asserted and the BOOST signal is pulsed
for a few hundred nanoseconds. After a short delay, the
VSW pin slews rapidly to VIN. Later, after the peak switch
current indicated by the control voltage VC has been
reached (current mode control), the SWON and SWDR
signals are turned off, and SWOFF is pulsed for several
hundred nanoseconds. The use of an explicit turn-off
device, i.e., Q5, improves turn-off response time and thus
aids both controllability and efficiency.
The system as previously described handles heavy loads
(continuous mode) at good efficiency, but it is actually
counterproductive for light loads. The method of jamming charge into the PNP bases makes it difficult to turn
them off rapidly and achieve the very short switch ON
times required by light loads in discontinuous mode.
Furthermore, the high leading edge dV/dt rate similarly
adversely affects light load controllability.
The solution is to employ a “boost comparator” whose
inputs are the VC control voltage and a fixed internal
threshold reference, VTH. (Remember that in a current
mode switching topology, the VC voltage determines the
peak switch current.) When the VC signal is above VTH, the
previously described “high dV/dt” action is performed.
When the VC signal is below VTH, the boost pulses are
absent, as can be seen in the Low dV/dt Mode Timing
Diagram. Now the DC current, activated by the SWON
signal alone, drives Q4 and this transistor drives Q1 by
itself. The absence of a boost pulse, plus the lack of a
second NPN driver, result in a much lower slew rate which
aids light load controllability.
A further aid to overall efficiency is provided by the
specialized bias regulator circuit, which has a pair of
inputs, VIN and VCC. The VCC pin is normally connected to
the switching supply output. During start-up conditions,
the LT1676 powers itself directly from VIN. However, after
the switching supply output voltage reaches about 2.9V,
the bias regulator uses this supply as its input. Previous
generation Buck controller ICs without this provision
typically required hundreds of milliwatts of quiescent
power when operating at high input voltage. This both
degraded efficiency and limited available output current
due to internal heating.
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APPLICATIONS INFORMATION
Selecting a Power Inductor
There are several parameters to consider when selecting
a power inductor. These include inductance value, peak
current rating (to avoid core saturation), DC resistance,
construction type, physical size, and of course, cost.
In a typical application, proper inductance value is dictated
by matching the discontinuous/continuous crossover point
with the LT1676 internal low-to-high dV/dt threshold. This
is the best compromise between maintaining control with
light loads while maintaining good efficiency with heavy
loads. The fixed internal dV/dt threshold has a nominal
value of 1.4V, which referred to the VC pin threshold and
control voltage to switch transconductance, corresponds
to a peak current of about 200mA. Standard Buck converter theory yields the following expression for inductance at the discontinuous/continuous crossover:
V
 V – V 
L =  OUT   IN OUT 
VIN
 f • IPK  

For example, substituting 48V, 5V, 200mA and 100kHz
respectively for VIN, VOUT, IPK and f yields a value of about
220µH. Note that the left half of this expression is independent of input voltage while the right half is only a weak
function of VIN when VIN is much greater than VOUT. This
means that a single inductor value will work well over a
range of “high” input voltage. And although a progressively smaller inductor is suggested as VIN begins to
approach VOUT, note that the much higher ON duty cycles
under these conditions are much more forgiving with
respect to controllability and efficiency issues. Therefore
when a wide input voltage range must be accommodated,
say 10V to 50V for 5VOUT, the user should choose an
inductance value based on the maximum input voltage.
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LT1676
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APPLICATIONS INFORMATION
Once the inductance value is decided, inductor peak
current rating and resistance need to be considered. Here,
the inductor peak current rating refers to the onset of
saturation in the core material, although manufacturers
sometimes specify a “peak current rating” which is
derived from a worst-case combination of core saturation
and self-heating effects. Inductor winding resistance alone
limits the inductor’s current carrying capability as the I2R
power threatens to overheat the inductor. If applicable,
remember to include the condition of output short circuit.
Although the peak current rating of the inductor can be
exceeded in short-circuit operation, as core saturation per
se is not destructive to the core, excess resistive selfheating is still a potential problem.
The final inductor selection is generally based on cost,
which usually translates into choosing the smallest physical size part that meets the desired inductance value,
resistance and current carrying capability. An additional
factor to consider is that of physical construction. Briefly
stated, “open” inductors built on a rod- or barrel-shaped
core generally offer the smallest physical size and lowest
cost. However their open construction does not contain
the resulting magnetic field, and they may not be acceptable in RFI-sensitive applications. Toroidal style inductors, many available in surface mount configuration, offer
improved RFI performance, generally at an increase in
cost and physical size. And although custom design is
always a possibility, most potential LT1676 applications
can be handled by the array of standard, off-the-shelf
inductor products offered by the major suppliers.
Selecting Freewheeling Diode
Highest efficiency operation requires the use of a Schottky
type diode. DC switching losses are minimized due to its
low forward voltage drop, and AC behavior is benign due
to its lack of a significant reverse recovery time. Schottky
diodes are generally available with reverse voltage ratings
of 60V and even 100V, and are price competitive with other
types.
The use of so-called “ultrafast” recovery diodes is generally not recommended. When operating in continuous
mode, the reverse recovery time exhibited by “ultrafast”
diodes will result in a slingshot type effect. The power
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internal switch will ramp up VIN current into the diode in an
attempt to get it to recover. Then, when the diode has
finally turned off, some tens of nanoseconds later, the VSW
node voltage ramps up at an extremely high dV/dt, perhaps 5 to even 10V/ns ! With real world lead inductances,
the VSW node can easily overshoot the VIN rail. This can
result in poor RFI behavior and if the overshoot is severe
enough, damage the IC itself.
Selecting Bypass Capacitors
The basic topology as shown in Figure 1 uses two bypass
capacitors, one for the VIN input supply and one for the
VOUT output supply.
User selection of an appropriate output capacitor is relatively easy, as this capacitor sees only the AC ripple current
in the inductor. As the LT1676 is designed for Buck or
step-down applications, output voltage will nearly always
be compatible with tantalum type capacitors, which are
generally available in ratings up to 35V or so. These
tantalum types offer good volumetric efficiency and many
are available with specified ESR performance. The product
of inductor AC ripple current and output capacitor ESR will
manifest itself as peak-to-peak voltage ripple on the output
node. (Note: If this ripple becomes too large, heavier
control loop compensation, at least at the switching frequency, may be required on the VC pin.) The most
demanding applications, requiring very low output ripple,
may be best served not with a single extremely large
output capacitor, but instead by the common technique of
a separate L/C lowpass post filter in series with the output.
(In this case, “Two caps are better than one.”)
The input bypass capacitor is normally a more difficult
choice. In a typical application e.g., 48VIN to 5VOUT,
relatively heavy VIN current is drawn by the power switch
for only a small portion of the oscillator period (low ON
duty cycle). The resulting RMS ripple current, for which
the capacitor must be rated, is often several times the DC
average VIN current. Similarly, the “glitch” seen on the VIN
supply as the power switch turns on and off will be related
to the product of capacitor ESR, and the relatively high
instantaneous current drawn by the switch. To compound
these problems is the fact that most of these applications
will be designed for a relatively high input voltage, for
LT1676
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APPLICATIONS INFORMATION
Minimum Load Considerations
As discussed previously, a lightly loaded LT1676 with VC
pin control voltage below the boost threshold will operate
in low dV/dt mode. This affords greater controllability at
light loads, as minimum tON requirements are relaxed. In
many applications, it is possible to operate the LT1676
down to zero external load without “pulse skipping”!
In these cases, the LT1676’s modest VCC current
requirement of several milliamperes provides enough of a
load to avoid pulse skipping.
However, some users may be indifferent to pulse skipping
behavior, but instead may be concerned with maintaining
maximum possible efficiency at light loads. This requirement can be satisfied by forcing the part into Burst ModeTM
operation. The use of an external comparator whose
output controls the shutdown pin allows high efficiency at
light loads through Burst Mode operation behavior (see
Typical Applications and Figure 8).
Maximum Load/Short-Circuit Considerations
The LT1676 is a current mode controller. It uses the VC
node voltage as an input to a current comparator which
turns off the output switch on a cycle-by-cycle basis as
this peak current is reached. The internal clamp on the VC
node, nominally 2V, then acts as an output switch peak
current limit. This action becomes the switch current limit
specification. The maximum available output power is
then determined by the switch current limit.
A potential controllability problem could occur under
short-circuit conditions. If the power supply output is
short circuited, the feedback amplifier responds to the low
output voltage by raising the control voltage, VC, to its
peak current limit value. Ideally, the output switch would
be turned on, and then turned off as its current exceeded
the value indicated by VC. However, there is finite response
time involved in both the current comparator and turnoff
of the output switch. These result in a minimum on time
Burst Mode is a trademark of Linear Technology Corporation.
tON(MIN). When combined with the large ratio of VIN to
(VF + I • R), the diode forward voltage plus inductor I • R
voltage drop, the potential exists for a loss of control.
Expressed mathematically the requirement to maintain
control is:
V +I•R
f • tON ≤ F
VIN
where:
f = switching frequency
tON = switch ON time
VF = diode forward voltage
VIN = Input voltage
I • R = inductor I • R voltage drop
If this condition is not observed, the current will not be
limited at IPK, but will cycle-by-cycle ratchet up to some
higher value. Using the nominal LT1676 clock frequency
of 100KHz, a VIN of 48V and a (VF + I • R) of say 0.7V, the
maximum tON to maintain control would be approximately
140ns, an unacceptably short time.
The solution to this dilemma is to slow down the oscillator
when the FB pin voltage is abnormally low thereby indicating some sort of short-circuit condition. Figure 2 shows
the typical response of Oscillator Frequency vs FB divider
Thevenin voltage and impedance. Oscillator frequency is
unaffected until FB voltage drops to about 2/3 of its normal
value. Below this point the oscillator frequency decreases
roughly linearly down to a limit of about 25kHz. This lower
120
100
RTH = 22k
80
fOSC (kHz)
which tantalum capacitors are generally unavailable. Relatively bulky “high frequency” aluminum electrolytic types,
specifically constructed and rated for switching supply
applications, may be the only choice.
RTH = 10k
RTH = 4.7k
60
40
RTH
LT1676
FB
20
0
0
0.25
0.50
0.75
1.00
FB DIVIDER THEVENIN VOLTAGE (V)
1.25
1676 F02
Figure 2. Oscillator Frequency vs FB Divider
Thevenin Voltage and Impedance
9
LT1676
U
W
U
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APPLICATIONS INFORMATION
oscillator frequency during short-circuit conditions can
then maintain control with the effective minimum ON time.
A further potential problem with short-circuit operation
might occur if the user were operating the part with its
oscillator slaved to an external frequency source via the
SYNC pin. However, the LT1676 has circuitry that automatically disables the sync function when the oscillator is
slowed down due to abnormally low FB voltage.
Feedback Divider Considerations
An LT1676 application typically includes a resistive divider
between VOUT and ground, the center node of which drives
the FB pin to the reference voltage VREF. This establishes
a fixed ratio between the two resistors, but a second
degree of freedom is offered by the overall impedance
level of the resistor pair. The most obvious effect this has
is one of efficiency—a higher resistance feedback divider
will waste less power and offer somewhat higher efficiency, especially at light load.
However, remember that oscillator slowdown to achieve
short-circuit protection (discussed above) is dependent
on FB pin behavior, and this in turn, is sensitive to FB node
external impedance. Figure 2 shows the typical relationship between FB divider Thevenin voltage and impedance,
and oscillator frequency. This shows that as feedback
network impedance increases beyond 10k, complete oscillator slowdown is not achieved, and short-circuit protection may be compromised. And as a practical matter,
the product of FB pin bias current and larger FB network
impedances will cause increasing output voltage error.
(Nominal cancellation for 10k of FB Thevenin impedance
is included internally.)
Thermal Considerations
Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause excessive die temperatures. The packages are rated at 110°C/W
for the 8-pin SO (S8) and 130°C/W for 8-pin PDIP (N8).
Quiescent power is given by:
PQ = IVIN • VIN + IVCC • VOUT
(This assumes that the VCC pin is connected to VOUT.)
10
Power loss internal to the LT1676 related to actual output
current is composed of both DC and AC switching losses.
These can be roughly estimated as follows:
DC switching losses are dominated by output switch “ON
voltage”, i.e.,
PDC = VON • IOUT • DC
VON = Output switch ON voltage, typically 1V at 500mA
IOUT = Output current
DC = ON duty cycle
AC switching losses are typically dominated by power lost
due to the finite rise time and fall time at the VSW node.
Assuming, for simplicity, a linear ramp up of both voltage
and current and a current rise/fall time equal to 15ns,
PAC = 1/2 • VIN • IOUT • (tr + tf + 30ns) • f
tr = (VIN/1.6)ns in high dV/dt mode
(VIN/0.16)ns in low dV/dt mode
tf = (VIN/1.6)ns (irrespective of dV/dt mode)
f = switching frequency
Total power dissipation of the die is simply the sum of
quiescent, DC and AC losses previously calculated.
PD(TOTAL) = PQ + PDC + PAC
Frequency Compensation
Loop frequency compensation is performed by connecting a capacitor, or in most cases a series RC, from the
output of the error amplifier (VC pin) to ground. Proper
loop compensation may be obtained by empirical methods as described in detail in Application Note 19. Briefly,
this involves applying a load transient and observing the
dynamic response over the expected range of VIN and
ILOAD values.
As a practical matter, a second small capacitor, directly
from the VC pin to ground is generally recommended to
attenuate capacitive coupling from the VSW pin. A typical
value for this capacitor is 100pF. (See Switch Node Considerations).
Switch Node Considerations
For maximum efficiency, switch rise and fall times are
made as short as practical. To prevent radiation and high
LT1676
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APPLICATIONS INFORMATION
frequency resonance problems, proper layout of the components connected to the IC is essential, especially the
power path. B field (magnetic) radiation is minimized by
keeping output diode, switch pin and intput bypass
capacitor leads as short as possible. E field radiation is
kept low by minimizing the length and area of all traces
connected to the switch pin (VSW). A ground plane should
always be used under the switcher circuitry to prevent
interplane coupling.
The high speed switching current path is shown schematically in Figure 3. Minimum lead length in these paths is
essential to ensure clean switching and minimal EMI. The
paths containing the input capacitor, output switch and
output diode are the only ones containing nanosecond rise
and fall times. Keep these paths as short as possible.
Additionally, it is possible for the LT1676 to cause EMI
problems by “coupling to itself”. Specifically, this can
occur if the VSW pin is allowed to capacitively couple in an
uncontrolled manner to the part’s high impedance nodes,
VIN
C1
LT1676
L1
VSW
VOUT
+
D1
As an example, assume that the capacitance between the
VSW node and a high impedance pin node is 0.1pF, and
further assume that the high impedance node in question
exhibits a capacitance of 1pF to ground. Due to the high
dV/dt, large excursion behavior of the VSW node, this will
couple a nearly 5V transient to the high impedance pin,
causing abnormal operation. (This assumes the “typical”
48VIN to 5VOUT application.) An explicit 100pF capacitor
added to the node will reduce the amplitude of the disturbance to more like 50mV (although settling time will
increase).
Specific pin recommendations are as follows:
SHDN: If unused, add a 100pF capacitor to ground.
SYNC: Ground if unused.
VC: Add a capacitor directly to ground in addition to the
explicit compensation network. A value of one-tenth of
the main compensation capacitor is recommended, up
to a maximum of 100pF.
+
VIN
i.e., SHDN, SYNC, VC and FB. This can cause erratic
operation such as odd/even cycle behavior, pulse width
“nervousness”, improper output voltage and/or premature current limit action.
C2
1676 F03
Figure 3. High Speed Current Switching Paths
FB: Assuming the VC pin is handled properly, this pin
usually requires no explicit capacitor of its own, but
keep this node physically small to minimize stray capacitance.
U
TYPICAL APPLICATIONS
Minimum Component Count Application
User Programmable Undervoltage Lockout
Figure 4a shows a basic “minimum component count”
application. The circuit produces 5V at up to 500mA IOUT
with input voltages in the range of 12V to 48V. The typical
POUT/PIN efficiency is shown in Figure 4b. No pulse
skipping is observed down to zero external load. As
shown, the SHDN and SYNC pins are unused, however
either (or both) can be optionally driven by external signals
as desired.
Figure 5 adds a resistor divider to the basic application.
This is a simple, cost-effective way to add a user-programmable undervoltage lockout (UVLO) function. Resistor R5
is chosen to have approximately 200µA through it at the
nominal SHDN pin lockout threshold of roughly 1.25V.
The somewhat arbitrary value of 200µA was chosen to be
significantly above the SHDN pin input current to minimize
its error contribution, but significantly below the typical
3.2mA the LT1676 draws in lockout mode. Resistor R4 is
then chosen to yield this same 200µA, less 2.5µA, with the
11
LT1676
U
TYPICAL APPLICATIONS
VIN
12V TO
48V
5
+
C1
39µF
63V
SHDN
VCC
VSW
C5
100pF
6
2
3
L1
D1
MBRS1100 220µH
LT1676
7
FB
8
VC
SYNC
4
C2
100µF
10V
R1
36.5k
1%
80
R2
12.1k
1%
C4
100pF
60
50
40
1676 F04a
C1: PANASONIC HFQ
C2: AVX D CASE TPSD107M010R0080
C4, C5: X7R OR COG/NPO
D1: MOTOROLA 100V, 1A, SMD SCHOTTKY
L1: COILCRAFT DO3316P-224
90
70
C3
2200pF
X7R
R3
22k
5%
GND
+
VOUT
5V
0mA to 500mA
EFFICIENCY (%)
1
VIN
FOR 3.3V VOUT VERSION:
R1: 24.3K, R2: 14.7k
L1: 150µH, DO3316P-154
IOUT: 0mA TO 500mA
VIN = 12V
VIN = 24V
30 VIN = 36V
VIN = 48V
20
1
10
100
1000
ILOAD (mA)
1676 F04b
Figure 4a. Minimum Component Count Application
VIN
R4
210k
1%
+
5
C1
1
VIN
2
SHDN
VCC
VSW
L1
3
VOUT
+
D1
LT1676
C5
R5
6.19k
1%
Figure 4b. POUT/PIN Efficiency
C2
R1
7
6
FB
8
VC
SYNC
R2
C3
GND
4
C4
R3
1676 F05
Figure 5. User Programmable Undervoltage Lockout
desired VIN UVLO voltage minus 1.25V applied across it.
(The 2.5µA factor is an allowance to minimize error due to
SHDN pin input current.)
Behavior is as follows: Normal operation is observed at the
nominal input voltage of 48V. As the input voltage is
decreased to roughly 43V, switching action will stop, VOUT
will drop to zero, and the LT1676 will draw its VIN and VCC
quiescent currents from the VIN supply. At a much lower
input voltage, typically 18V or so at 25°C, the voltage on
12
the SHDN pin will drop to the shutdown threshold, and the
part will draw its shutdown current only from the VIN rail.
The resistive divider of R4 and R5 will continue to draw
power from VIN. (The user should be aware that while the
SHDN pin lockout threshold is relatively accurate including temperature effects, the SHDN pin shutdown threshold is more coarse, and exhibits considerably more
temperature drift. Nevertheless the shutdown threshold
will always be well below the lockout threshold.)
LT1676
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TYPICAL APPLICATIONS
Micropower Undervoltage Lockout
Certain applications may require very low current drain
when in undervoltage lockout mode. This can be accomplished with the addition of a few more external components. Figure 6 shows an LTC®1440 micropower
comparator/reference added to control the LT1676 via its
SHDN pin. The extremely low input bias current of the
CMOS comparator allows the impedance of the resistor
divider R4/R5 to be increased, thereby minimizing power
drain. Hysteresis is externally programmable via resistor
divider R6/R7. The LTC1440 output directly controls the
LT1676 via its shutdown pin, driving it to either 5V (ON) or
0V (Full Shutdown). A simple linear voltage regulator to
power the LTC1440 is provided by Q1, Q2 and R7. Just
below the UVLO threshold, nominally 43V, total current
drain is typically 50µA.
Burst Mode Operation Configuration
Figure 4b demonstrates that power supply efficiency degrades with lower output load current. This is not surprising, as the LT1676 itself represents a fixed power overhead.
A possible way to improve light load efficiency is in Burst
Mode operation.
Figure 7 shows the LT1676 configured for Burst Mode
operation. Output voltage regulation is now provided in a
“bang-bang” digital manner, via comparator U2, an
LTC1440. Resistor divider R3/R4 provides a scaled version of the output voltage, which is compared against U2’s
internal reference. Intentional hysteresis is set by the R5/
R6 divider. As the output voltage falls below the regulation
range, the LT1676 is turned on. The output voltage rises,
and as it climbs above the regulation range, the LT1676 is
turned off. Efficiency is maximized, as the LT1676 is only
powered up while it is providing heavy output current.
Figure 7b shows that efficiency is typically maintained at
75% or better down to a load current of 10mA. Even at a
load of 1mA, efficiency is still a respectable 59% to 68%,
depending on VIN.
Resistor divider R1/R2 is still present, but does not
directly influence output voltage. It is chosen to ensure
that the LT1676 delivers high output current throughout
the voltage regulation range. Its presence is also required
VIN
5
6
R8
10M
+
Q1
PN2484
VIN
2
SYNC
VCC
VSW
C1
39µF
63V
3
L1
D1
MBRS1100 220µH
U1
LT1676
1
7
FB
8
VC
SHDN
GND
Q2
2N2369
4
NC
C3
2200pF
R3
22k
+
C2
100µF
10V
R1
36.5
1%
VOUT
5V
R2
12.1k
1%
C4
100pF
7
8
V+
OUT
IN +
–
3
IN
U2
LTC1440
6
REF
5
HYST
–
V
GND
2
1
VIN
4
R6
22k
R4
6.8M
R7
2.4M
R5
240k
C1: PANASONIC HFQ
C2: AVX D CASE TPSD107M010R0080
C4, C5: X7R OR COG/NPO
D1: MOTOROLA 100V, 1A, SMD SCHOTTKY
L1: COILCRAFT DO3316P-224
1676 F06
Figure 6. Micropower Undervoltage Lockout
13
LT1676
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TYPICAL APPLICATIONS
to maintain proper short-circuit protection. Transistors
Q1, Q2 and resistor R7 form a high VIN, low quiescent
current voltage regulator to power U2.
Minimum Size Inductor Application
Figure 4a employs power path parts that are capable of
delivering the full rated output capability of the LT1676.
Potential users with low output current requirements may
be interested in substituting a physically smaller and less
costly power inductor. The circuit shown on the last page
of this data sheet is topologically identical to the basic
application, but specifies a much smaller inductor, and, a
somewhat smaller input electrolytic capacitor. This circuit
is capable of delivering up to 150mA at 5V, or, up to 200mA
at 3.3V. The only disadvantage is that due to the increased
resistance in the inductor, the circuit is no longer capable
of withstanding indefinite short circuits to ground. The
LT1676 will still current limit at its nominal ILIM value, but
this will overheat the inductor. Momentary short circuits
of a few seconds or less can still be tolerated.
Burst Mode Operation Configuration with UVLO
Figure 7a uses an external comparator to control the
LT1676 via its SHDN pin. As such, the user’s ability to set
an undervoltage lockout (UVLO) threshold with a resistor
divider from VIN to SHDN pin to ground is lost. This ability
is regained in the slightly more complicated circuit shown
in Figure 8.
A dual comparator, the LTC1442, replaces the previous
single comparator. The second comparator monitors a
resistive divider between VIN and ground to provide the
(user-adjustable) UVLO function. The two comparator
outputs are logically combined in a CMOS NOR gate (U3)
to drive the LT1676 SHDN pin.
VIN
+
5
C1
6
R7
10M
VIN
2
SYNC
VCC
VSW
L1
3
+
U1
LT1676
Q1
PN2484
D1
7
1
FB
8
VC
SHDN
GND
Q2
2N2369
C3
100pF
4
C2
VOUT
5V
R1
39k
5%
R2
10k
5%
90
R3
323k
1%
80
VIN = 12V
NC
EFFICIENCY (%)
70
7
8
C1: PANASONIC HFQ 39µF AT 63V
C2: AVX D CASE 100µF 10V
TPSD107M010R0080
D1: MOTOROLA 100V, 1A,
SMD SCHOTTKY
MBRS1100 (T3)
L1: COILCRAFT DO3316-224
V+
OUT
+
3
IN
4
IN –
U2
LTC1440
6
REF
5
HYST
–
V
GND
2
1
R5
22k
VIN = 48V
VIN = 36V
VIN = 24V
60
50
40
R4
100k
1%
30
20
R6
2.4M
1
10
100
1000
ILOAD (mA)
1676 F07a
(a)
Figure 7. Burst Mode Operation Configuration for High Efficiency at Light Load
14
1676 F07b
(b)
LT1676
U
TYPICAL APPLICATIONS
VIN
5
6
+
R7
10M
VIN
SYNC
VCC
C1
VSW
2
L1
3
Q1
PN2484
1
Q2
2N2369
D1
FB
SHDN
VC
VOUT
5V
+
U1
LT1676
R1
39k
C2
7
8
R2
10k
GND
C3
4
VIN
NC
V
C1: PANASONIC HFQ 39µF AT 63V
C2: AVX D CASE 100µF 10V
TPSD107M010R0080
D1: MOTOROLA 100V, 1A,
SMD SCHOTTKY
MBRS1100 (T3)
L1: COILCRAFT DO3316-224
5
4
U3 3
7S02
OUTA
1
2
R3
323k
1%
R8
6.8M
+
INA+
INB–
U2
LTC1442
REF
OUTB HYST
V–
R5
22k
R4
100k
1%
R9
240k
R6
2.4M
1676 F08
Figure 8. Burst Mode Operation Configuration with Micropower UVLO
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.300 – 0.325
(7.620 – 8.255)
0.045 – 0.065
(1.143 – 1.651)
(
0.130 ± 0.005
(3.302 ± 0.127)
0.065
(1.651)
TYP
0.009 – 0.015
(0.229 – 0.381)
+0.035
0.325 –0.015
+0.889
8.255
–0.381
0.400*
(10.160)
MAX
)
8
7
6
5
1
2
3
4
0.255 ± 0.015*
(6.477 ± 0.381)
0.125
(3.175) 0.020
MIN
(0.508)
MIN
0.018 ± 0.003
(0.457 ± 0.076)
N8 1197
0.100 ± 0.010
(2.540 ± 0.254)
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.053 – 0.069
(1.346 – 1.752)
0.008 – 0.010
(0.203 – 0.254)
0°– 8° TYP
0.016 – 0.050
0.406 – 1.270
0.014 – 0.019
(0.355 – 0.483)
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
8
7
6
5
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
TYP
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
2
3
4
SO8 0996
15
LT1676
U
TYPICAL APPLICATION
Minimum Inductor Size Application
VIN
12V TO
48V
5
1
+
VIN
SHDN
VCC
C5
100pF
VSW
C1
12µF
63V
2
3
LT1676
6
L1
D1 220µH
+
7
FB
8
VC
SYNC
C3
2200pF
X7R
R3
22k
5%
GND
4
C2
100µF
10V
C4
100pF
R1
36.5k
1%
VOUT
5V
0mA to 150mA
R2
12.1k
1%
1676 TA02
C1: PANASONIC HFQ
C2: AVX D CASE TPSD107M010R0080
C4, C5: X7R OR COG/NPO
D1: MOTOROLA 100V, 1A, SMD SCHOTTKY
MBRS1100 (T3)
L1: COILCRAFT DO1608C-224
FOR 3.3V VOUT VERSION:
IOUT: 0mA TO 200mA
L1: 150µH, DO1608C-154
R1: 24.3K, R2: 14.7k
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1076
2A, 100kHz Step-Down Switching Regulator
Operation Up to 45V Input (64V for HV Version)
LT1149
High Efficiency Synchronous Step-Down Switching Regulator
Operation Up to 48V Input, 95% Efficiency, 100% Duty Cycle
LT1176
1.2A, 100kHz Step-Down Switching Regulator
Operation Up to 38V Input, Adjustable and Fixed 5V Versions
LT1339
High Power Synchronous DC/DC Controller
Operation Up to 60V, High Power Anti-Shoot-Through Drivers
LT1375/LT1376
1.5A, 500kHz Step-Down Switching Regulators
Operation Up to 25V Input, Synchronizable (LT1375)
LT1620
Rail-to-Rail Current Sense Amplifier
Transforms Switching Regulators Into High Efficiency
Battery Chargers
LT1776
Wide Input Range, High Efficiency, Step-Down
Switching Regulator
LT1676 with 200kHz Switching Frequency (High Current
Applications Generally Restricted to ≤ 40V)
LT1777
Low Noise Buck Regulator
Operation up to 48V, Controlled Voltage and Current Slew Rates
16
Linear Technology Corporation
1676f LT/TP 0499 4K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
 LINEAR TECHNOLOGY CORPORATION 1998
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