LINER LTC1475CS8-5 Low quiescent current high efficiency step-down converter Datasheet

LTC1474/LTC1475
Low Quiescent Current
High Efficiency Step-Down
Converters
DESCRIPTION
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FEATURES
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The LT C®1474/LTC1475 series are high efficiency stepdown converters with internal P-channel MOSFET power
switches that draw only 10µA typical DC supply current at
no load while maintaining output voltage. The LTC1474
uses logic-controlled shutdown while the LTC1475 features pushbutton on/off.
High Efficiency: Over 92% Possible
Very Low Standby Current: 10µA Typ
Available in Space Saving 8-Lead MSOP Package
Internal 1.4Ω Power Switch (VIN = 10V)
Wide VIN Range: 3V to 18V Operation
Very Low Dropout Operation: 100% Duty Cycle
Low-Battery Detector Functional During Shutdown
Programmable Current Limit with Optional
Current Sense Resistor (10mA to 400mA Typ)
Short-Circuit Protection
Few External Components Required
Active Low Micropower Shutdown: IQ = 6µA Typ
Pushbutton On/Off (LTC1475 Only)
3.3V, 5V and Adjustable Output Versions
The low supply current coupled with Burst ModeTM operation enables the LTC1474/LTC1475 to maintain high efficiency over a wide range of loads. These features, along
with their capability of 100% duty cycle for low dropout
and wide input supply range, make the LTC1474/LTC1475
ideal for moderate current (up to 300mA) battery-powered
applications.
The peak switch current is user-programmable with an
optional sense resistor (defaults to 325mA minimum if not
used) providing a simple means for optimizing the design
for lower current applications. The peak current control
also provides short-circuit protection and excellent startup behavior. A low-battery detector that remains functional
in shutdown is provided .
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APPLICATIONS
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Cellular Telephones and Wireless Modems
4mA to 20mA Current Loop Step-Down Converter
Portable Instruments
Battery-Operated Digital Devices
Battery Chargers
Inverting Converters
Intrinsic Safety Applications
The LTC1474/LTC1475 series availability in 8-lead MSOP
and SO packages and need for few additional components
provide for a minimum area solution.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
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TYPICAL APPLICATION
LTC1474 Efficiency
100
VIN
4V TO 18V
LOW BATTERY OUT
VIN = 5V
90
10µF
25V
0.1µF
VIN = 10V
7
VIN
1
SENSE
VFB
LTC1474-3.3
3
2
LBI
LBO
6
LOW BATTERY IN
RUN SHDN
100k
8
RUN
GND
4
SW
VOUT
3.3V AT 250mA
L1
100µH
+
100µF
6.3V
5
D1
MBR0530
EFFICIENCY (%)
+
80
VIN = 15V
70
60
L = 100µH
VOUT = 3.3V
RSENSE = 0Ω
1474/75 F01
L1 = SUMIDA CDRH74-101
50
Figure 1. High Efficiency Step-Down Converter
0.03
0.3
30
3
LOAD CURRENT (mA)
300
1474/75 TA01
1
LTC1474/LTC1475
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ABSOLUTE MAXIMUM RATINGS
Input Supply Voltage (VIN).........................– 0.3V to 20V
Switch Current (SW, SENSE) .............................. 750mA
Switch Voltage (SW).............. (VIN – 20V) to (VIN + 0.3V)
VFB (Adjustable Versions) ..........................– 0.3V to 12V
VOUT (Fixed Versions) ................................ –0.3V to 20V
LBI, LBO ....................................................– 0.3V to 20V
RUN, SENSE .................................. – 0.3V to (VIN + 0.3V)
Operating Ambient Temperature Range
Commercial ............................................ 0°C to 70°C
Industrial ............................................ – 40°C to 85°C
Junction Temperature (Note 1) ............................ 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
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PACKAGE/ORDER INFORMATION
TOP VIEW
TOP VIEW
VOUT/VFB
LBO
LBI
GND
TOP VIEW
8
7
6
5
1
2
3
4
RUN
VIN
SENSE
SW
MS8 PACKAGE
8-LEAD PLASTIC MSOP
VOUT/VFB
LBO
LBI/OFF
GND
8 RUN
VOUT/VFB 1
8
7
6
5
1
2
3
4
ON
VIN
SENSE
SW
MS8 PACKAGE
8-LEAD PLASTIC MSOP
TOP VIEW
LBO 2
7 VIN
LBI 3
6 SENSE
GND 4
5 SW
S8 PACKAGE
8-LEAD PLASTIC SO
VOUT/VFB 1
8 ON
LBO 2
7 VIN
LBI/OFF 3
6 SENSE
GND 4
5 SW
S8 PACKAGE
8-LEAD PLASTIC SO
TJMAX = 125°C, θJA = 150°C/ W
TJMAX = 125°C, θJA = 150°C/ W
TJMAX = 125°C, θJA = 110°C/ W
TJMAX = 125°C, θJA = 110°C/ W
ORDER PART NUMBER
ORDER PART NUMBER
ORDER PART NUMBER
ORDER PART NUMBER
LTC1474CMS8
LTC1474CMS8-3.3
LTC1474CMS8-5
LTC1475CMS8
LTC1475CMS8-3.3
LTC1475CMS8-5
LTC1474CS8
LTC1474IS8
LTC1474CS8-3.3
LTC1474CS8-5
LTC1474IS8-3.3
LTC1474IS8-5
LTC1475CS8
LTC1475IS8
LTC1475CS8-3.3
LTC1475CS8-5
MS8 PART MARKING
MS8 PART MARKING
S8 PART MARKING
S8 PART MARKING
1474
1474I
14743
14745
14743I
14745I
1475
1475I
14753
14755
LTBW
LTCR
LTCS
Consult factory for Military grade parts.
2
LTBK
LTCP
LTCQ
LTC1474/LTC1475
ELECTRICAL CHARACTERISTICS
TA = 25°C, VIN = 10V, VRUN = open, RSENSE = 0, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VFB
Feedback Voltage
LTC1474/LTC1475
ILOAD = 50mA
VOUT
Regulated Output Voltage
LTC1474-3.3/LTC1475-3.3
LTC1474-5/LTC1475-5
ILOAD = 50mA
MIN
TYP
MAX
UNITS
●
1.205
1.230
1.255
V
●
●
3.234
4.900
3.300
5.000
3.366
5.100
V
V
0
30
nA
IFB
Feedback Current
LTC1474/LTC1475 Only
ISUPPLY
No Load Supply Current (Note 3)
ILOAD = 0 (Figure 1 Circuit)
10
∆VOUT
Output Voltage Line Regulation
VIN = 7V to 12V, ILOAD = 50mA
5
20
mV
Output Voltage Load Regulation
ILOAD = 0mA to 50mA
2
15
mV
Output Ripple
ILOAD = 10mA
50
Input DC Supply Current (Note 2)
Active Mode (Switch On)
Sleep Mode (Note 3)
Shutdown
(Exclusive of Driver Gate Charge Current)
VIN = 3V to 18V
VIN = 3V to 18V
VIN = 3V to 18V, VRUN = 0V
100
9
6
175
15
12
µA
µA
µA
RON
Switch Resistance
ISW = 100mA
1.4
1.6
Ω
IPEAK
Current Comp Max Current Trip Threshold
RSENSE = 0Ω
RSENSE = 1.1Ω
325
70
400
76
85
mA
mA
90
100
110
mV
IQ
●
µA
mVP-P
VSENSE
Current Comp Sense Voltage Trip Threshold
VHYST
Voltage Comparator Hysteresis
tOFF
Switch Off-Time
VLBI, TRIP
Low Battery Comparator Threshold
VRUN
Run/ON Pin Threshold
0.4
0.7
1.0
V
VLBI, OFF
OFF Pin Threshold (LTC1475 Only)
0.4
0.7
1.0
V
ILBO, SINK
Sink Current into Pin 2
VLBI = 0V, VLBO = 0.4V
0.45
0.70
IRUN, SOURCE
Source Current from Pin 8
VRUN = 0V
0.4
ISW, LEAK
Switch Leakage Current
VIN = 18V, VSW = 0V, VRUN = 0V
ILBI, LEAK
Leakage Current into Pin 3
ILBO, LEAK
Leakage Current into Pin 2
The ● denotes specifications which apply over the full operating
temperature range.
Note 1: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC1474CS8/LTC1475CS8: TJ = TA + (PD • 110°C/W)
LTC1474CMS8/LTC1475CMS8: TJ = TA + (PD • 150°C/W)
●
5
VOUT at Regulated Value
VOUT = 0V
●
mV
3.5
4.75
65
6.0
µs
µs
1.16
1.23
1.27
V
mA
0.8
1.2
µA
0.015
1
µA
VLBI = 18V, VIN = 18V
0
0.1
µA
VLBI = 2V, VLBO = 5V
0
0.5
µA
Note 2: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 3: No load supply current consists of sleep mode DC current (9µA
typical) plus a small switching component (about 1µA for Figure 1 circuit)
necessary to overcome Schottky diode and feedback resistor leakage.
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LTC1474/LTC1475
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TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage
Line Regulation
40
FIGURE 1 CIRCUIT
L: CDRH73-101
90
30
ILOAD = 200mA
85
FIGURE 1 CIRCUIT
30
20
RSENSE = 0Ω
20
ILOAD = 25mA
∆VOUT (mV)
EFFICIENCY (%)
95
40
FIGURE 1 CIRCUIT
ILOAD = 100mA
∆VOUT (mV)
100
Load Regulation
10
RSENSE = 0.33Ω
80
0
70
–10
0
4
–20
16
12
8
INPUT VOLTAGE (V)
10
VIN = 10V
0
VIN = 5V
–10
ILOAD = 1mA
75
VIN = 15V
–20
–30
0
4
0
16
12
8
INPUT VOLTAGE (V)
1474/75 G01
50
250
150
200
100
LOAD CURRENT (mA)
300
1474/75 G02
1474/75 G03
Switch Resistance vs
Input Voltage
Current Trip Threshold vs
Temperature
500
Supply Current in Shutdown
10.0
5
RSENSE = 0Ω
4
300
200
RSENSE = 1.1Ω
100
0
SUPPLY CURRENT (µA)
400
RDS(ON) (Ω)
CURRENT TRIP THRESHOLD (mA)
VIN = 10V
3
2
T = 70°C
1
T = 25°C
0
0
20
40
60
80
5
10
2.5
0
20
15
5
10
1474/75 G05
Switch Leakage Current vs
Temperature
1474/75 G06
Off-Time vs Output Voltage
VIN DC Supply Current
80
120
1.0
20
15
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
1474/75 G04
VIN = 18V
VIN = 10V
ACTIVE MODE
100
0.6
0.4
0.2
60
80
OFF-TIME (µs)
0.8
SUPPLY CURRENT (µA)
LEAKAGE CURRENT (µA)
5.0
0
0
TEMPERATURE (°C)
60
0
20
40
60
TEMPERATURE (°C)
80
100
1474/75 G07
40
40
20
20
0
4
7.5
0
SLEEP MODE
0
0
4
12
8
INPUT VOLTAGE (V)
16
20
1474/75 G08
0
40
100
20
60
80
% OF REGULATED OUTPUT VOLTAGE (%)
1474/75 G09
LTC1474/LTC1475
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PIN FUNCTIONS
VOUT/VFB (Pin 1): Feedback of Output Voltage. In the fixed
versions, an internal resistive divider divides the output
voltage down for comparison to the internal 1.23V reference. In the adjustable versions, this divider must be
implemented externally.
SW (Pin 5): Drain of Internal PMOS Power Switch. Cathode of Schottky diode must be closely connected to this
pin.
SENSE (Pin 6): Current Sense Input for Monitoring Switch
Current and Source of Internal PMOS Power Switch.
Maximum switch current is programmed with a resistor
between SENSE and VIN pins.
LBO (Pin 2): Open Drain Output of the Low Battery
Comparator. This pin will sink current when Pin 3 is below
1.23V.
VIN (Pin 7): Main Supply Pin.
LBI/OFF (Pin 3): Input to Low Battery Comparator. This
input is compared to the internal 1.23V reference. For the
LTC1475, a momentary ground on this pin puts regulator
in shutdown mode.
RUN/ON (Pin 8): On LTC1474, voltage level on this pin
controls shutdown/run mode (ground = shutdown, open/
high = run). On LTC1475, a momentary ground on this pin
puts regulator in run mode. A 100k series resistor must be
used between Pin 8 and the switch or control voltage.
GND (Pin 4): Ground Pin.
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FUNCTIONAL DIAGRA
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LBI/OFF
100mV
1µA
VIN
–
×
ON
VIN
7
C
RSENSE
(OPTIONAL)
+
+
VCC
–
6
V
ON
SENSE
5Ω
+
LTC1474: RUN 8
LTC1475: ON
4.75µs
1×
1-SHOT
TRIGGER
20×
OUT
STRETCH WAKEUP
LBO
SW
2
5
VOUT
+
+
3M
(5V VERSION)
1.68M
(3.3V VERSION)
LB
1.23V
–
READY
3
LTC1474: LBI
LTC1475: LBI/OFF
× CONNECTION NOT PRESENT IN LTC1474 SERIES
CONNECTION PRESENT IN LTC1474 SERIES ONLY
VOUT/VFB
1
1.23V
REFERENCE
4
GND
1M
OUTPUT DIVIDER IS
IMPLEMENTED EXTERNALLY IN
ADJUSTABLE VERSIONS
1474/75 FD
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LTC1474/LTC1475
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OPERATIO (Refer to Functional Diagram)
The LTC1474/LTC1475 are step-down converters with
internal power switches that use Burst Mode operation to
keep the output capacitor charged to the proper output
voltage while minimizing the quiescent current. Burst
Mode operation functions by using short “burst” cycles to
ramp the inductor current through the internal power
switch and external Schottky diode, followed by a sleep
cycle where the power switch is off and the load current is
supplied by the output capacitor. During sleep mode, the
LTC1474/LTC1475 draw only 9µA typical supply current.
At light loads, the burst cycles are a small percentage of
the total cycle time; thus the average supply current is very
low, greatly enhancing efficiency.
Peak Inductor Current Programming
Burst Mode Operation
Off-Time
At the beginning of the burst cycle, the switch is turned on
and the inductor current ramps up. At this time, the internal
current comparator is also turned on to monitor the switch
current by measuring the voltage across the internal and
optional external current sense resistors. When this voltage reaches 100mV, the current comparator trips and
pulses the 1-shot timer to start a 4.75µs off-time during
which the switch is turned off and the inductor current
ramps down. At the end of the off-time, if the output
voltage is less than the voltage comparator threshold, the
switch is turned back on and another cycle commences. To
minimize supply current, the current comparator is turned
on only during the switch-on period when it is needed to
monitor switch current. Likewise, the 1-shot timer will only
be on during the 4.75µs off-time period.
The off-time duration is 4.75µs when the feedback voltage
is close to the reference; however, as the feedback voltage
drops, the off-time lengthens and reaches a maximum
value of about 65µs when this voltage is zero. This ensures
that the inductor current has enough time to decay when
the reverse voltage across the inductor is low such as
during short circuit.
The average inductor current during a burst cycle will
normally be greater than the load current, and thus the
output voltage will slowly increase until the internal voltage comparator trips. At this time, the LTC1474/LTC1475
go into sleep mode, during which the power switch is off
and only the minimum required circuitry is left on: the
voltage comparator, reference and low battery comparator. During sleep mode, with the output capacitor supplying the load current, the VFB voltage will slowly decrease
until it reaches the lower threshold of the voltage comparator (about 5mV below the upper threshold). The
voltage comparator then trips again, signaling the LTC1474/
LTC1475 to turn on the circuitry necessary to begin a new
burst cycle.
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The current comparator provides a means for programming the maximum inductor/switch current for each switch
cycle. The 1X sense MOSFET, a portion of the main power
MOSFET, is used to divert a sample (about 5%) of the
switch current through the internal 5Ω sense resistor. The
current comparator monitors the voltage drop across the
series combination of the internal and external sense
resistors and trips when the voltage exceeds 100mV. If the
external sense resistor is not used (Pins 6 and 7 shorted),
the current threshold defaults to the 400mA maximum due
to the internal sense resistor.
Shutdown Mode
Both LTC1474 and LTC1475 provide a shutdown mode
that turns off the power switch and all circuitry except for
the low battery comparator and 1.23V reference, further
reducing DC supply current to 6µA typical. The LTC1474’s
run/shutdown mode is controlled by a voltage level at the
RUN pin (ground = shutdown, open/high = run). The
LTC1475’s run/shutdown mode, on the other hand, is
controlled by an internal S/R flip-flop to provide pushbutton
on/off control. The flip-flop is set (run mode) by a momentary ground at the ON pin and reset (shutdown mode) by
a momentary ground at the LBI/OFF pin.
Low Battery Comparator
The low battery comparator compares the voltage on the
LBI pin to the internal reference and has an open drain
N-channel MOSFET at its output. If LBI is above the
reference, the output FET is off and the LBO output is high
impedance. If LBI is below the reference, the output FET is
on and sinks current. The comparator is still active in
shutdown.
LTC1474/LTC1475
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APPLICATIONS INFORMATION
The basic LTC1474/LTC1475 application circuit is shown
in Figure 1, a high efficiency step-down converter. External
component selection is driven by the load requirement and
begins with the selection of RSENSE. Once RSENSE is
known, L can be chosen. Finally D1, CIN and COUT are
selected.
RSENSE Selection
The current sense resistor (RSENSE) allows the user to
program the maximum inductor/switch current to optimize the inductor size for the maximum load. The LTC1474/
LTC1475 current comparator has a maximum threshold of
100mV/(RSENSE + 0.25). The maximum average output
current IMAX is equal to this peak value less half the peakto-peak ripple current ∆IL.
Allowing a margin for variations in the LTC1474/LTC1475
and external components, the required RSENSE can be
calculated from Figure 2 and the following equation:
RSENSE = (0.067/IMAX) – 0.25
(1)
for 10mA < IMAX < 200mA.
ments. Lower peak currents have the advantage of lower
output ripple (∆VOUT = IPEAK • ESR), lower noise, and less
stress on alkaline batteries and other circuit components.
Also, lower peak currents allow the use of inductors with
smaller physical size.
Peak currents as low as 10mA can be programmed with
the appropriate sense resistor. Increasing RSENSE above
10Ω, however, gives no further reduction of IPEAK.
For RSENSE values less than 1Ω, it is recommended that
the user parallel standard resistors (available in values ≥
1Ω) instead of using a special low valued shunt resistor.
Although a single resisor could be used with the desired
value, these low valued shunt resistor types are much
more expensive and are currently not available in case
sizes smaller than 1206. Three or four 0603 size standard
resistors require about the same area as one 1206 size
current shunt resistor at a fraction of the cost.
At higher supply voltages and lower inductances, the peak
currents may be slightly higher due to current comparator
overshoot and can be predicted from the second term in
the following equation:
5
0.1
IPEAK =
+
0.25 + RSENSE
4
RSENSE (Ω)
FOR LOWEST NOISE
3
FOR BEST EFFICIENCY
2
1
0
0
250
100
150
200
50
MAXIMUM OUTPUT CURRENT (mA)
300
1474/75 F02
Figure 2. RSENSE Selection
For IMAX above 200mA, RSENSE is set to zero by shorting
Pins 6 and 7 to provide the maximum peak current of
400mA (limited by the fixed internal sense resistor). This
400mA default peak current can be used for lower IMAX if
desired to eliminate the need for the sense resistor and
associated decoupling capacitor. However, for optimal
performance, the peak inductor current should be set to no
more than what is needed to meet the load current require-
(2.5)10−7 (VIN − VOUT)
L
(2)
Note that RSENSE only sets the maximum inductor current
peak. At lower dI/dt (lower input voltages and higher
inductances), the observed peak current at loads less than
IMAX may be less than this calculated peak value due to the
voltage comparator tripping before the current ramps up
high enough to trip the current comparator. This effect
improves efficiency at lower loads by keeping the I2R
losses down (see Efficiency Considerations section).
Inductor Value Selection
Once RSENSE and IPEAK are known, the inductor value can
be determined. The minimum inductance recommended
as a function of IPEAK and IMAX can be calculated from:
(
)
0.75 VOUT + VD tOFF 

L MIN ≥ 
 IPEAK − IMAX



where tOFF = 4.75µs.
(3)
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LTC1474/LTC1475
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APPLICATIONS INFORMATION
If the LMIN calculated is not practical, a larger IPEAK should
be used. Although the above equation provides the minimum, better performance (efficiency, line/load regulation,
noise) is usually gained with higher values. At higher
inductances, peak current and frequency decrease (improving efficiency) and inductor ripple current decreases
(improving noise and line/load regulation). For a given
inductor type, however, as inductance is increased, DC
resistance (DCR) increases, increasing copper losses,
and current rating decreases, both effects placing an
upper limit on the inductance. The recommended range of
inductances for small surface mount inductors as a function of peak current is shown in Figure 3. The values in this
range are a good compromise between the trade-offs
discussed above. If space is not a premium, inductors with
larger cores can be used, which extends the recommended range of Figure 3 to larger values.
INDUCTOR VALUE (µH)
1000
500
100
50
10
100
PEAK INDUCTOR CURRENT (mA)
1000
1474/75 F03
Figure 3. Recommended Inductor Values
Inductor Core Selection
Once the value of L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, as discussed in the previous
section, increased inductance requires more turns of wire
and therefore copper losses will increase.
Ferrite and Kool Mµ designs have very low core loss and
are preferred at high switching frequencies, so design
goals can concentrate on copper loss and preventing
saturation. Ferrite core material saturates “hard,” which
means that inductance collapses abruptly when the peak
design current is exceeded. This results in an abrupt
increase in inductor current above IPEAK and consequent
increase in voltage ripple. Do not allow the core to saturate! Coiltronics, Coilcraft, Dale and Sumida make high
performance inductors in small surface mount packages
with low loss ferrite and Kool Mµ cores and work well in
LTC1474/LTC1475 regulators.
Catch Diode Selection
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As VIN approaches VOUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition, the diode must
safely handle IPEAK at close to 100% duty cycle.
To maximize both low and high current efficiency, a fast
switching diode with low forward drop and low reverse
leakage should be used. Low reverse leakage current is
critical to maximize low current efficiency since the leakage can potentially approach the magnitude of the LTC1474/
LTC1475 supply current. Low forward drop is critical for
high current efficiency since loss is proportional to forward drop. These are conflicting parameters (see Table 1),
but a good compromise is the MBR0530 0.5A Schottky
diode specified in the application circuits.
Table 1. Effect of Catch Diode on Performance
DIODE (D1)
BAS85
LEAKAGE
FORWARD
NO LOAD
DROP
SUPPLY CURRENT EFFICIENCY*
200nA
0.6V
9.7µA
77.9%
MBR0530
1µA
0.4V
10µA
83.3%
MBRS130
20µA
0.3V
16µA
84.6%
*Figure 1 circuit with VIN = 15V, IOUT = 0.1A
Kool Mµ is a registered trademark of Magnetics, Inc.
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LTC1474/LTC1475
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APPLICATIONS INFORMATION
CIN and COUT Selection
At higher load currents, when the inductor current is
continuous, the source current of the P-channel MOSFET
is a square wave of duty cycle VOUT/VIN. To prevent large
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum
capacitor current is given by:
CIN Required IRMS =
[
(
IMAX VOUT VIN − VOUT
)]
1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Do not underspecify this component. An additional 0.1µF ceramic capacitor is also required on VIN for
high frequency decoupling.
The selection of COUT is driven by the required effective
series resistance (ESR) to meet the output voltage ripple
and line regulation requirements. The output voltage ripple
during a burst cycle is dominated by the output capacitor
ESR and can be estimated from the following relation:
25mV < ∆VOUT, RIPPLE = ∆IL • ESR
where ∆IL ≤ IPEAK and the lower limit of 25mV is due to the
voltage comparator hysteresis. Line regulation can also
vary with COUT ESR in applications with a large input
voltage range and high peak currents.
ESR is a direct function of the volume of the capacitor.
Manufacturers such as Nichicon, AVX and Sprague should
be considered for high performance capacitors. The
OS-CON semiconductor dielectric capacitor available from
SANYO has the lowest ESR for its size at a somewhat
higher price. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. For lower
current applications with peak currents less than 50mA,
10µF ceramic capacitors provide adequate filtering and
are a good choice due to their small size and almost
negligible ESR. AVX and Marcon are good sources for
these capacitors.
In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalums, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include SANYO OS-CON, Nichicon PL series and Sprague
595D series. Consult the manufacturer for other specific
recommendations.
To avoid overheating, the output capacitor must be sized
to handle the ripple current generated by the inductor. The
worst-case ripple current in the output capacitor is given
by:
IRMS = IPEAK / 2
Once the ESR requirement for COUT has been met, the
RMS current rating generally far exceeds the IRIPPLE(P-P)
requirement.
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting efficiency and which change would produce the
most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, three main sources usually account for most of the
losses in LTC1474/LTC1475 circuits: VIN current, I2R
losses and catch diode losses.
1. The VIN current is due to two components: the DC bias
current and the internal P-channel switch gate charge
current. The DC bias current is 9µA at no load and
increases proportionally with load up to a constant
100µA during continuous mode. This bias current is so
9
LTC1474/LTC1475
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small that this loss is negligible at loads above a
milliamp but at no load accounts for nearly all of the
loss. The second component, the gate charge current,
results from switching the gate capacitance of the
internal P-channel switch. Each time the gate is switched
from high to low to high again, a packet of charge dQ
moves from VIN to ground. The resulting dQ/dt is the
current out of VIN which is typically much larger than the
DC bias current. In continuous mode, IGATECHG = fQP
where QP is the gate charge of the internal switch. Both
the DC bias and gate charge losses are proportional to
VIN and thus their effects will be more pronounced at
higher supply voltages.
2. I2R losses are predicted from the internal switch,
inductor and current sense resistor. At low supply
voltages where the switch on-resistance is higher and
the switch is on for longer periods due to higher duty
cycle, the switch losses will dominate. Keeping the peak
currents low with the appropriate RSENSE and with
larger inductance helps minimize these switch losses.
At higher supply voltages, these losses are proportional
to load and result in the flat efficiency curves seen in
Figure 1.
3. The catch diode loss is due to the VDID loss as the diode
conducts current during the off-time and is more pronounced at high supply voltage where the on-time is
short. This loss is proportional to the forward drop.
However, as discussed in the Catch Diode section,
diodes with lower forward drops often have higher
leakage current, so although efficiency is improved, the
no load supply current will increase.
Adjustable Applications
For adjustable versions, the output voltage is programmed
with an external divider from VOUT to VFB (Pin 1) as shown
in Figure 4. The regulated voltage is determined by:
 R2
VOUT = 1.23 1+ 
 R1
To minimize no-load supply current, resistor values in the
megohm range should be used. The increase in supply
current due to the feedback resistors can be calculated
from:

 V
V
∆I VIN =  OUT   OUT 
 R1 + R2  VIN 
A 10pF feedforward capacitor across R2 is necessary due
to the high impedances to prevent stray pickup and
improve stability.
VOUT
10pF
R2
1
LTC1474 V
FB
LTC1475
R1
GND
4
1474/75 F04
Figure 4. LTC1474/LTC1475 Adjustable Configuration
Low Battery Comparator
The LTC1474/LTC1475 have an on-chip low battery comparator that can be used to sense a low battery condition
when implemented as shown in Figure 5. The resistive
divider R3/R4 sets the comparator trip point as follows:
 R4 
VTRIP = 1.23 1 + 
 R3 
The divided down voltage at the LBI pin is compared to the
internal 1.23V reference. When VLBI < 1.23V, the LBO
output sinks current. The low battery comparator is active
all the time, even during shutdown mode.
VIN
R4
LBI
(4)
R3
LTC1474/LTC1475
LBO
–
+
1.23V
REFERENCE
1474/75 F05
Figure 5. Low Battery Comparator
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LTC1475 Pushbutton On/Off and
Microprocessor Interface
The LTC1475 provides pushbutton control of power on/off
for use with handheld products. A momentary ground on
the ON pin sets an internal S/R latch to run mode while a
momentary ground on the LBI/OFF pin resets the latch to
shutdown mode. See Figure 6 for a comparsion of on/off
operation of the LTC1474 and LTC1475 and Figure 7 for a
comparison of the circuit implementations.
In the LTC1475, the LBI/OFF pin has a dual function as
both the shutdown control pin and the low battery comparator input. Since the “OFF” pushbutton is normally
open, it does not affect the normal operation of the low
battery comparator. In the unpressed state, the LBI/OFF
input is the divided down input voltage from the resistive
divider to the internal low battery comparator and will
normally be above or just below the trip threshold of
1.23V. When shutdown mode is desired, the LBI/OFF pin
is pulled below the 0.7V threshold to invoke shutdown.
the depressed switch state is detected by the microcontroller through its input. The microcontroller then pulls the
LBI/OFF pin low with the connection to one of its ouputs.
With the LBI/OFF pin low, the LTC1475 powers down
turning the microcontroller off. Note that since the I/O pins
of most microcontrollers have a reversed bias diode
between input and supply, a blocking diode with less than
1µA leakage is necessary to prevent the powered down
microcontroller from pulling down on the ON pin.
Figure 19 in the Typical Applications section shows how to
use the low battery comparator to provide a low battery
lockout on the “ON” switch. The LBO output disconnects
the pushbutton from the ON pin when the comparator has
tripped, preventing the LTC1475 from attempting to start
up again until VIN is increased.
100k
100k
ON
RUN
VIN
RUN
LTC1474
ON
LTC1475
LBI/OFF
RUN
OFF
LTC1474
MODE
RUN
SHUTDOWN
RUN
1474/75 F07
ON OVERRIDES LBI/OFF
WHILE ON IS LOW
Figure 7. Simplified Implementation of
LTC1474 and LTC1475 On/Off
ON
Absolute Maximum Ratings and Latchup Prevention
LBI/OFF
LTC1475
MODE
RUN
SHUTDOWN
RUN
1474/75 F06
Figure 6. Comparison of LTC1474 and LTC1475
Run/Shutdown Operation
The ON pin has precedence over the LBI/OFF pin. As seen
in Figure 6, if both pins are grounded simultaneously, run
mode wins.
Figure 18 in the Typical Applications section shows an
example for the use of the LTC1475 to control on/off of a
microcontroller with a single pushbutton. With both the
microcontroller and LTC1475 off, depressing the
pushbutton grounds the LTC1475 ON pin and starts up the
LTC1475 regulator which then powers up the microcontroller. When the pushbutton is depressed a second time,
The absolute maximum ratings specify that SW (Pin 5) can
never exceed VIN (Pin 7) by more than 0.3V. Normally this
situation should never occur. It could, however, if the
output is held up while the supply is pulled down. A
condition where this could potentially occur is when a
battery is supplying power to an LTC1474 or LTC1475
regulator and also to one or more loads in parallel with the
the regulator’s VIN. If the battery is disconnected while the
LTC1474 or LTC1475 regulator is supplying a light load
and one of the parallel circuits is a heavy load, the input
capacitor of the LTC1474 or LTC1475 regulator could be
pulled down faster than the output capacitor, causing the
absolute maximum ratings to be exceeded. The result is
often a latchup which can be destructive if VIN is reapplied.
Battery disconnect is possible as a result of mechanical
stress, bad battery contacts or use of a lithium-ion battery
11
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with a built-in internal disconnect. The user needs to
assess his/her application to determine whether this situation could occur. If so, additional protection is necessary.
where P is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to the
ambient temperature.
Prevention against latchup can be accomplished by simply connecting a Schottky diode across the SW and VIN
pins as shown in Figure 8. The diode will normally be
reverse biased unless VIN is pulled below VOUT at which
time the diode will clamp the (VOUT – VIN) potential to less
than the 0.6V required for latchup. Note that a low leakage
Schottky should be used to minimize the effect on no-load
supply current. Schottky diodes such as MBR0530, BAS85
and BAT84 work well. Another more serious effect of the
protection diode leakage is that at no load with nothing to
provide a sink for this leakage current, the output voltage
can potentially float above the maximum allowable tolerance. To prevent this from occuring, a resistor must be
connected between VOUT and ground with a value low
enough to sink the maximum possible leakage current.
The junction temperature is given by:
LATCHUP
PROTECTION
SCHOTTKY
TJ = TA + TR
As an example consider the LTC1474/LTC1475 in dropout
at an input voltage of 3.5V, a load current of 300mA, and
an ambient temperature of 70°C. From the typical performance graph of switch resistance, the on-resistance of the
P-channel switch at 70°C is 3.5Ω. Therefore, power dissipated by the part is:
P = I2 • RDS(ON) = 0.315W
For the MSOP package, the θJA is 150°C/W. Thus the
junction temperature of the regulator is:
TJ = 70°C + (0.315)(150) = 117°C
which is near the maximum junction temperature of 125oC.
Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance.
PC Board Layout Checklist
VIN
VOUT
SW
LTC1474
LTC1475
+
1474/75 F08
Figure 8. Preventing Absolute Maximum
Ratings from Being Exceeded
Thermal Considerations
In the majority of the applications, the LTC1474/LTC1475
do not dissipate much heat due to their high efficiency.
However, in applications where the switching regulator is
running at high ambient temperature with low supply
voltage and high duty cycles, such as dropout with the
switch on continuously, the user will need to do some
thermal analysis. The goal of the thermal analysis is to
determine whether the power dissipated by the regulator
exceeds the maximum junction temperature of the part.
The temperature rise is given by:
TR = P • θJA
12
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC1474/LTC1475. These items are also illustrated
graphically in the layout diagram of Figure 9. Check the
following in your layout:
1. Is the Schottky diode cathode closely connected to SW
(Pin 5)?
2. Is the 0.1µF input decoupling capacitor closely connected between VIN (Pin 7) and ground (Pin 4)? This
capacitor carries the high frequency peak currents.
3. When using adjustable version, is the resistive divider
closely connected to the (+) and (–) plates of COUT with
a 10pF capacitor connected across R2?
4. Is the 1000pF decoupling capacitor for the current
sense resistor connected as close as possible to Pins 6
and 7? If no current sense resistor is used, Pins 6 and
7 should be shorted.
LTC1474/LTC1475
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OUTPUT DIVIDER REQUIRED WITH
ADJUSTABLE VERSION ONLY
10pF
LTC1474
1
VOUT
R2
R1
+
2
3
COUT
4
VFB
8
RUN
VIN 7
100k
L
LBO
LBI
GND
SENSE
SW
1000pF
6
RSENSE
5
1474/75 F09
0.1µF
CIN
D1
+
VIN
BOLD LINES INDICATE HIGH PATH CURRENTS
Figure 9. LTC1474/LTC1475 Layout Diagram (See Board Layout Checklist)
5. Are the signal and power grounds segregated? The
signal ground consists of the (–) plate of COUT, Pin 4 of
the LTC1474/LTC1475 and the resistive divider. The
power ground consists of the Schottky diode anode,
the (–) plate of CIN and the 0.1µF decoupling capacitor.
6. Is a 100k resistor connected in series between RUN
(Pin 8) and the RUN control voltage? The resistor
should be as close as possible to Pin 8.
Design Example (Refer to RSENSE and Inductor
Selection)
As a design example, assume VIN = 10V, VOUT = 3V, and
a maximum average output current IMAX = 100mA. With
this information, we can easily calculate all the important
components:
From the equation (1),
RSENSE = (0.067/0.1) – 0.25 = 0.42Ω
Using the standard resistors (1Ω, 1Ω and 2Ω) in parallel
provides 0.4Ω without having to use a more expensive
low value current shunt type resistor (see RSENSE Selection section).
With RSENSE = 0.4Ω, the peak inductor current IPEAK is
calculated from (2), neglecting the second term, to be
150mA. The minimum inductance is, therefore, from the
equation (3) and assuming VD = 0.4V,
L MIN =
(
)(
) = 264µH
0.75 3.3 + 0.4 4.75µs
0.15 − 0.1
From Figure 3, an inductance of 270µH is chosen from the
recommended region. The CDRH73-271 or CD54-271 is a
good choice for space limited applications.
For the feedback resistors, choose R1 = 1M to minimize
supply current. R2 can then be calculated from the equation (4) to be:
V

R2 =  OUT − 1 • R1 = 1.43 M
 1.23 
For the catch diode, the MBR0530 will work well in this
application.
For the input and output capacitors, AVX 4.7µF and 100µF,
respectively, low ESR TPS series work well and meet the
RMS current requirement of 100mA/2 = 50mA. They are
available in small “C” case sizes with 0.15Ω ESR. The
0.15Ω output capacitor ESR will result in 25mV of output
voltage ripple.
Figure 10 shows the complete circuit for this example.
13
LTC1474/LTC1475
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TYPICAL APPLICATIONS
VIN
3.5V TO 18V
10pF
+
4.7µF†
35V
1Ω**
1Ω**
2Ω**
1000pF
6
3
* SUMIDA CDRH73-271
** 3 PARALLEL STANDARD RESISTORS
PROVIDE LEAST EXPENSIVE SOLUTION
(SEE R SENSE SELECTION SECTION)
† AVX TPSC475M035
†† AVX TPSC107M006
RUN
100k 8
0.1µF
7
VIN
SENSE
LBI
RUN
VFB
LTC1474 LBO
GND
SW
1.43M
1%
1
1M
1%
2
L*
270µH
+
5
VOUT
3V
100mA
100µF††
6.3V
D1
MBR0530
4
1474/75 F10
Figure 10. High Efficiency 3V/100mA Regulator (Design Example)
IN +
4mA TO 20mA
D2††
12V
1µF
×3
2Ω
7.5M
SENSE
VOUT
LTC1474-3.3
3
LBI
LBO
TO A/D
MBR0530
IN
4mA TO 20mA
7
VIN
6
†
–
1000pF
1M
100k
RUN
8
RUN
GND
SW
VOUT
3.3V
10mA
1
2
L*
330µH
4
D1
MBR0530
* COILCRAFT DO1608-334
** MARCON THCS50E1E106Z,
AVX 1206ZG106Z
† OPTIONAL RESISTOR FOR SENSING LOOP CURRENT BY A/D CONVERTER
† † MOTOROLA MMBZ5242BL
Figure 11. High Efficiency 3.3V/10mA Output from 4mA to 20mA Loop
14
10µF**
5
1474/75 F11
LTC1474/LTC1475
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TYPICAL APPLICATIONS
MBR0530
VIN
3.5V TO 6V
10pF
+
+
0.1µF
22µF**
16V
4.7M
1%
7
6
VIN
SENSE
3
RUN
VOUT
–12V
70mA
22µF††
25V
LBI
100k 8
VFB
LTC1474 LBO
RUN
SW
GND
1
536k
1%
2
+
22µF††
25V
+
5
VIN (V)
10µF†
L*
200µH 25V
4
* COILTRONICS CTX200-4
** AVX TPSC226M016
† AVX TPSC106M025
†† AVX TPSD226M025
L*
200µH
VOUT
12V
70mA
D1
MBR0530
I LOAD(MAX)
3.5
30mA
4
50mA
5
70mA
6
90mA
1474/75 F12
Figure 12. 5V to ±12V Regulator
+
0.1µF
10µF**
25V
7
6
3
RUN
100k 8
SENSE
VIN
LTC1474-5
LBI
RUN
LBO
GND
4
* COILTRONICS CTX100-4
** AVX TPSC106MO25
† AVX TPSC336M010
VOUT
SW
1
2
10µF**
25V
L*
100µH
+
33µF†
10V
VOUT
5V
200mA AT VIN = 10V
+
VIN
3.5V TO 12V
5
VIN (V)
L*
100µH
D1
MBR0530
3.5
I LOAD(MAX)
70mA
4
95mA
5
125mA
8
180mA
10
200mA
12
225mA
1474/75 F13
Figure 13. 5V Buck-Boost Converter
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LTC1474/LTC1475
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TYPICAL APPLICATIONS
VIN
3.5V TO 12V
ON/OFF††
+
0.1µF
10µF**
25V
TP0610
7
6
3
8
10M
SENSE
VIN
VOUT
LTC1474-5
LBI
LBO
RUN
SW
GND
1
2
VIN (V)
L*
100µH
+
33µF†
10V
5
D1
MBR0530
4
I LOAD(MAX)
3.5
100mA
5
140mA
8
190mA
12
240mA
VOUT
–5V
140mA AT VIN = 5V
* SUMIDA CDRH74-101
** AVX TPSC106M025
† AVX TPSC336M010
†† RUN: ON/OFF = 0, SHUTDOWN: 0N/OFF = V
IN
Figure 14. Positive-to-Negative (– 5V) Converter
VIN
8V TO 18V
10pF
+
4.7µF**
35V
0.1µF
7
6
3
CHARGER
ON/OFF
100k 8
* SUMIDA CDRH73-101
** AVX TPSC475M035
† AVX TPSD476M016
SENSE
LBI
RUN
VIN
VFB
LTC1474 LBO
GND
SW
1
2
1M
L*
100µH
+
47µF†
16V
VOUT
4-NiCd
200mA
5
4
D1
MBR0530
1474/75 F15
Figure 15. 4-NiCd Battery Charger
16
MBR0530
4.69M
1474/75 F14
LTC1474/LTC1475
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TYPICAL APPLICATIONS
VIN
4V TO 18V
+
4.7µF†
35V
0.1µF
2.2M
7
6
3
1M
100k
8
RUN
VIN
SENSE
VOUT
LTC1474-3.3
LBI
LBO
RUN
SW
GND
1
2
L*
100µH
+
100µF††
6.3V
VOUT
3.3V
250mA
5
D1
MBR0530
4
* SUMIDA CDRH73-101
† AVX TPSC475M035
†† AVX TPSC107M006
1474/75 F16
Figure 16. High Efficiency 3.3V Regulator with Low Battery Lockout
VIN
5.7V TO 18V
+
4.7µF**
35V
0.1µF
7
3.65M
6
3
100k 8
OFF
1M
VIN
SENSE
VOUT
LTC1475-5
LBO
LBI/OFF
ON
SW
GND
ON
1
+
2
L*
100µH
33µF†
10V
VOUT
5V
250mA
5
D1
MBR0530
4
* SUMIDA CDRH73-101
** AVX TPSC475M035
† AVX TPSC336M010
1474/75 F17
Figure 17. Pushbutton On/Off 5V/250mA Regulator
VCC
VIN
4V TO 18V
+
MMBD914LT1
0.1µF
0.1µF
4.7µF**
35V
100k
ON/OFF
2.2M
8
2
3
1M
µC
ON
7
VIN
6
SENSE
VOUT
1
+
LTC1475-3.3
L*
100µH
LBO
LBI/OFF
GND
SW
VOUT
3.3V
250mA
100µF†
6.3V
5
4
* SUMIDA CDRH73-101
** AVX TPSC475M035
† AVX TPSC107M006
D1
MBR0530
1474/75 F18
Figure 18. LTC1475 Regulator with 1-Button Toggle On/Off
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LTC1474/LTC1475
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PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
MS8 Package
8-Lead Plastic MSOP
(LTC DWG # 05-08-1660)
0.118 ± 0.004*
(3.00 ± 0.102)
8
7 6
5
0.118 ± 0.004**
(3.00 ± 0.102)
0.192 ± 0.004
(4.88 ± 0.10)
1
0.040 ± 0.006
(1.02 ± 0.15)
0.007
(0.18)
2 3
4
0.034 ± 0.004
(0.86 ± 0.102)
0° – 6° TYP
0.021 ± 0.006
(0.53 ± 0.015)
SEATING
PLANE 0.012
(0.30)
0.0256
REF
(0.65)
TYP
0.006 ± 0.004
(0.15 ± 0.102)
* DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH,
PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
18
MSOP (MS8) 1197
LTC1474/LTC1475
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PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
8
7
6
5
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0°– 8° TYP
0.016 – 0.050
0.406 – 1.270
0.014 – 0.019
(0.355 – 0.483)
2
3
4
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
SO8 0996
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LTC1474/LTC1475
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TYPICAL APPLICATION
10pF
VIN
3.5V to 18V
+
0.1µF
4.7µF**
35V
7
1.8M
6
1M
100k
MMBT2N2222LT1
8
3
VIN
SENSE
ON
LTC1475
LBI/OFF
1M
ON
OFF
VFB
LBO
GND
SW
1
2
1.02M
1%
1M
1%
+
L*
100µH
VOUT
2.5V
250mA
100µF†
6.3V
5
4
D1
MBR0530
* SUMIDA CDRH73-101
** AVX TPSC475M035
† AVX TPSC107M006
1474/75 F19
Figure 19. Pushbutton On/Off with Low Battery Lockout
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1096/LTC1098
Micropower Sampling 8-Bit Serial I/O A/D Converter
IQ = 80µA Max
LT1121/LT1121-3.3/LT1121-5
150mA Low Dropout Regulator
Linear Regulator, IQ = 30µA
LTC1174/LTC1174-3.3/LTC1174-5 High Efficiency Step-Down and Inverting DC/DC Converters
Selectable IPEAK = 300mA or 600mA
LTC1265
1.2A High Efficiency Step-Down DC/DC Converter
Burst Mode Operation, Internal MOSFET
LT1375/LT1376
1.5A 500kHz Step-Down Switching Regulators
500kHz, Small Inductor, High
Efficiency Switchers, 1.5A Switch
LTC1440/LTC1441/LTC1442
Ultralow Power Comparator with Reference
IQ = 2.8µA Max
LT1495/LT1496
1.5µA Precision Rail-to-Rail Op Amps
IQ = 1.5µA Max
LT1521/LT1521-3/LT1521-3.3/
LT1521-5
300mA Low Dropout Regulator
Linear Regulator, IQ = 12µA
LTC1574/LTC1574-3.3/LTC1574-5 High Efficiency Step-Down DC/DC Converters with Internal Schottky Diode LTC1174 with Internal Schottky Diode
LT1634-1.25
20
Micropower Precision Shunt Reference
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● (408) 432-1900
FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com
IQ(MIN) = 10µA
14745fa LT/TP 0398 4K REV A • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 1997
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