LINER LTC1701B 1mhz step-down dc/dc converters in sot-23 Datasheet

LTC1701/LTC1701B
1MHz Step-Down
DC/DC Converters in SOT-23
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FEATURES
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DESCRIPTIO
Tiny 5-Lead SOT-23 Package
Uses Tiny Capacitors and Inductor
High Frequency Operation: 1MHz
High Output Current: 500mA
Low RDS(ON) Internal Switch: 0.28Ω
High Efficiency: Up to 94%
Current Mode Operation for Excellent Line
and Load Transient Response
Short-Circuit Protected
Low Quiescent Current: 135µA (LTC1701)
Low Dropout Operation: 100% Duty Cycle
Ultralow Shutdown Current: IQ < 1µA
Peak Inductor Current Independent of Inductor Value
Output Voltages from 5V Down to 1.25V
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APPLICATIO S
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PDAs/Palmtop PCs
Digital Cameras
Cellular Phones
Portable Media Players
PC Cards
Handheld Equipment
The LTC®1701/LTC1701B are the industry’s first SOT-23
step-down, current mode, DC/DC converters. Intended for
low to medium power applications, they operate from 2.5V
to 5.5V input voltage range and switch at 1MHz, allowing
the use of tiny, low cost capacitors and inductors 2mm or
less in height. The output voltage is adjustable from 1.25V
to 5V. A built-in 0.28Ω switch allows up to 0.5A of output
current at high efficiency. OPTI-LOOPTM compensation
allows the transient response to be optimized over a wide
range of loads and output capacitors.
The LTC1701 incorporates automatic power saving Burst
ModeTM operation to reduce gate charge losses when the
load current drops below the level required for continuous
operation. The LTC1701B operates continuously to very
low load currents to provide low ripple at the expense of
light load efficiency. With no load, the LTC1701 draws only
135µA. In shutdown, both devices draw less than 1µA,
making them ideal for current sensitive applications.
Their small size and switching frequency enables a
complete DC/DC converter function to consume less than
0.3 square inches of PC board area.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode and OPTI-LOOP are trademarks of Linear Technology Corporation.
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TYPICAL APPLICATION
VIN
+
SW
D1
R4
1M
VOUT
(2.5V/
500mA)
LTC1701
R3
5.1k
C3
330pF
VFB
VIN = 3.3V
VOUT = 2.5V
95
90
R2
121k
+
ITH/RUN
GND
100
LTC1701
85
C2
47µF
R1
121k
EFFICIENCY (%)
C1
10µF
Efficiency Curve
L1
4.7µH
VIN
2.5V TO
5.5V
80
75
LTC1701B
70
65
60
C1: TAIYO YUDEN JMK316BJ106ML
C2: SANYO POSCAP 6TPA47M
D1: MBRM120L
L1: SUMIDA CD43-4R7
Figure 1. 2.5V/500mA Step-Down Regulator
1701 F01a
55
50
1
10
100
LOAD CURRENT (mA)
1000
1701 F01b
1
LTC1701/LTC1701B
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ABSOLUTE
AXI U RATI GS
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PACKAGE/ORDER I FOR ATIO
(Note 1)
(Voltages Referred to GND Pin)
VIN Voltage (Pin 5).......................................– 0.3V to 6V
ITH/RUN Voltage (Pin 4) ..............................– 0.3V to 3V
VFB Voltage (Pin 3) ......................................– 0.3V to 3V
VIN – SW (Max Switch Voltage) ................8.5V to – 0.3V
Operating Temperature Range (Note 2) .. – 40°C to 85°C
Junction Temperature (Note 5) ............................. 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
SW 1
5 VIN
LTC1701ES5
LTC1701BES5
GND 2
VFB 3
4 ITH/RUN
S5 PACKAGE
5-LEAD PLASTIC SOT-23
S5 PART
MARKING
TJMAX = 125°C, θJA = 250°C/W
LTKG
LTUD
SEE THE APPLICATION
INFORMATION SECTION
Consult factory for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 3.3V, RITH/RUN = 1Meg (from VIN to ITH/RUN) unless otherwise
specified. (Note 2)
SYMBOL
PARAMETER
CONDITIONS
MIN
VIN
Operating Voltage Range
IFB
Feedback Pin Input Current
(Note 3)
VFB
Feedback Voltage
(Note 3)
∆VLINE REG
Reference Voltage Line Regulation
VIN = 2.5V to 5V (Note 3)
∆VLOAD REG
Output Voltage Load Regulation
Measured in Servo Loop, VITH = 1.5V, (Note 3)
Measured in Servo Loop, VITH = 1.9V, (Note 3)
TYP
2.5
●
1.22
MAX
UNITS
5.5
V
±0.1
µA
1.25
1.28
V
0.04
0.1
%/V
0.01
– 0.80
0.70
–1.50
%
%
185
135
0.25
300
200
1
µA
µA
µA
Input DC Supply Current (Note 4)
Active Mode
Sleep Mode
Shutdown
VFB = 0V
VFB = 1.4V (LTC1701 only)
VITH/RUN = 0V
Run Threshold High
Run Threshold Low
ITH/RUN Ramping Down
ITH/RUN Ramping Up
1.4
0.6
1.6
0.3
V
V
IITH/RUN
Run Pullup Current
VITH/RUN = 1V
50
100
300
µA
ISW(PEAK)
Peak Switch Current Threshold
VFB = 0V
0.9
1.1
A
RDS(ON)
Switch ON Resistance
VIN = 5V, VFB = 0V
VIN = 3.3V, VFB = 0V
VIN = 2.5V, VFB = 0V
0.28
0.30
0.35
Ω
Ω
Ω
ISW(LKG)
Switch Leakage Current
VIN = 5V, VITH/RUN = 0V, VFB = 0V
0.01
1
µA
tOFF
Switch Off-Time
500
600
ns
VITH/RUN
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC1701E/LTC1701BE guaranteed to meet performance
specifications from 0°C to 70°C. Specifications over the – 40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls.
2
400
Note 3: The LTC1701/LTC1701B are tested in a feedback loop which
servos VFB to the midpoint for the error amplifier without RITH/RUN = 1MHz
(VITH = 1.7V unless otherwise specified).
Note 4: Dynamic supply current is higher due to the internal gate charge
being delivered at the switching frequency.
Note 5: TJ is calculated from the ambient TA and power dissipation PD
according to the following formula:
LTC1701ES5/LTC1701BES5: TJ = TA + (PD • 250°C/W)
LTC1701/LTC1701B
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TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage
Efficiency vs Load Current
100
100
VOUT = 2.5V
V IN = 3.3V
95
DC Supply Current
300
VOUT = 2.5V
95
ILOAD =100mA
250
V IN = 5.0V
85
80
85
SUPPLY CURRENT (µA)
90
EFFICIENCY (%)
EFFICIENCY (%)
90
ILOAD =10mA
80
75
70
75
70
65
LTC1701
LTC1701B
1
10
100
LOAD CURRENT (mA)
2
4
3
150
SLEEP
100
50
LTC1701
LTC1701B
60
1000
ACTIVE
200
0
6
5
2
3
INPUT VOLTAGE (V)
1701 • G01
5
4
INPUT VOLTAGE (V)
1701 • G03
1701 • G02
Switch Resistance vs
Supply Voltage
Load Regulation
Line Regulation
0.60
370
6
0.30
0.40
310
290
ILOAD = 200mA
0.00
VOUT ERROR (%)
330
0.25
VOUT = 5.0V
0.20
VOUT ERROR (%)
ON RESISTANCE (mΩ)
350
–0.20
–0.40
–0.60
VOUT = 3.3V
–0.80
0.15
0.10
ILOAD = 400mA
–1.00
270
0.20
0.05
–1.20
250
2
3
5
4
SUPPLY VOLTAGE (V)
6
–1.40
0
400
200
LOAD CURRENT (mA)
1701 • G04
600
0
2
3
4
VIN (V)
6
1701 • G06
1701 • G05
Dropout Characteristics
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Transient Response
Start-Up
3.4
3.3
ILOAD = 100mA
VOUT
1V/DIV
VOUT (V)
3.2
VOUT
50mV/DIV
AC COUPLED
3.1
ILOAD = 200mA
3.0
ITH
2V/DIV
IL
200mA/DIV
2.9
ILOAD = 500mA
IL
500mA/DIV
2.8
VOUT = 3.3V
FIGURE 1
2.7
2.6
3.0
3.2
3.4
3.6
3.8
VIN (V)
701 • G07
VIN = 3.3V, VOUT = 2.5V
CIRCUIT OF FIGURE 1
RLOAD = 6Ω
1701 G08
VIN = 3.3V, VOUT = 2.5V
CIRCUIT OF FIGURE 1
ILOAD = 100mA TO 500mA STEP
1701 G09
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LTC1701/LTC1701B
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PI FU CTIO S
SW (Pin 1): The Switch Node Connection to the Inductor.
This pin swings from VIN to a Schottky diode (external)
voltage drop below ground. The cathode of the Schottky
diode must be closely connected to this pin.
ITH/RUN (Pin 4): Combination of Error Amplifier Compensation Point and Run Control Input. The current comparator threshold increases with this control voltage. Nominal
voltage range for this pin is 1.25V to 2.25V. Forcing this
pin below 0.8V causes the device to be shut down. In
shutdown all functions are disabled.
GND (Pin 2): Ground Pin. Connect to the (–) terminal of
COUT, the Schottky diode and (–) terminal of CIN.
VIN (Pin 5): Main Supply Pin and the (+) Input to the
Current Comparator. Must be closely decoupled to ground.
VFB (Pin 3): Receives the feedback voltage from the
external resistive divider across the output. Nominal voltage for this pin is 1.25V.
Pin Limit Table
PIN
NAME
DESCRIPTION
MIN
1
SW
Switch Node
– 0.3
NOMINAL (V)
TYP
MAX
ABSOLUTE MAX (V)
MIN
MAX
VIN
VIN – 8.5
VIN + 0.3
1.35
– 0.3
3
2
GND
Ground Pin
3
VFB
Output Feedback Pin
0
0
4
ITH/RUN
Error Amplifier Compensation and RUN Pin
0
2.25
– 0.3
3
5
VIN
Main Power Supply
2.5
5.5
– 0.3
6
1.25
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BLOCK DIAGRA
VIN
VIN
VIN
1.25V
BANDGAP
REFERENCE
50µA
VREF
+
+
CURRENT
SENSE
AMP
+
ITH/REF
CLAMP
–
1.5V
CURRENT
COMP
+
ITH
COMP
SHDN
ITH/RUN
–
–
VREF
–
VREF
+
(1.25V TO 2.25V)
ERROR
AMP
VFB
VREF
(1.25V)
(LTC1701 only)
–
SW
+
1.4V
OFF-TIMER
AND GATE
CONTROL LOGIC
OVER
VOLTAGE
COMP
–
PULSE
STRETCHER
VFB < 0.6V
4
GATE
DRIVER
GND
1701 BD
LTC1701/LTC1701B
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OPERATIO
The LTC1701 uses a contant off-time, current mode architecture. The operating frequency is then determined by the
off-time and the difference between VIN and VOUT.
The output voltage is set by an external divider returned to
the VFB pin. An error amplfier compares the divided output
voltage with a reference voltage of 1.25V and adjusts the
peak inductor current accordingly.
Main Control Loop
During normal operation, the internal PMOS switch is
turned on when the VFB voltage is below the reference
voltage. The current into the inductor and the load increases until the current limit is reached. The switch turns
off and energy stored in the inductor flows through the
external Schottky diode into the load. After the constant
off-time interval, the switch turns on and the cycle repeats.
The peak inductor current is controlled by the voltage on
the ITH/RUN pin, which is the output of the error
amplifier.This amplifier compares the VFB pin to the 1.25V
reference. When the load current increases, the FB voltage
decreases slightly below the reference. This decrease
causes the error amplifier to increase the ITH/RUN voltage
until the average inductor current matches the new load
current.
The main control loop is shut down by pulling the ITH/RUN
pin to ground. When the pin is released an external resistor
is used to charge the compensation capacitor. When the
voltage at the ITH/RUN pin reaches 0.8V, the main control
loop is enabled and the error amplifier drives the ITH/RUN
pin. Soft-start can be implemented by ramping the voltage
on the ITH/RUN pin (see Applications Information section).
Low Current Operation
To optimize efficiency when the load is relatively light, the
LTC1701 automatically switches to Burst Mode operation
in which the internal PMOS switch operates intermittently
based on load demand. The main control loop is interrupted when the output voltage reaches the desired regulated value. The hysteretic voltage comparator trips when
ITH/RUN is below 1.5V, shutting off the switch and reducing the power consumed. The output capacitor and the
inductor supply the power to the load until the output
voltage drops slightly and the ITH/RUN pin exceeds 1.5V,
turning on the switch and the main control loop which
starts another cycle.
For reduced output ripple, the LTC1701B doesn't use
Burst Mode operation and operates continuously down to
very low currents where the part starts skipping cycles.
Dropout Operation
In dropout, the internal PMOS switch is turned on continuously (100% duty cycle) providing low dropout operation
with VOUT at VIN. Since the LTC1701 does not incorporate
an under voltage lockout, care should be taken to shut
down the LTC1701 for VIN < 2.5V.
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APPLICATIO S I FOR ATIO
The basic LTC1701 application circuit is shown in
Figure␣ 1. External component selection is driven by the
load requirement and begins with the selection of L1. Once
L1 is chosen, the Schottky diode D1 can be selected
followed by CIN and COUT.
L Selection and Operating Frequency
The operating frequency is fixed by VIN, VOUT and the
constant off-time of about 500ns. The complete expression for operating frequency is given by:
V −V   1 
fO =  IN OUT  

 VIN + VD   TOFF 
Although the inductor does not influence the operating
frequency, the inductor value has a direct effect on ripple
current. The inductor ripple current ∆IL decreases with
higher inductance and increases with higher VIN or VOUT:
 V − V  V
+V 
∆IL =  IN OUT   OUT D 
fL

  VIN + VD 
where VD is the output Schottky diode forward drop.
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LTC1701/LTC1701B
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APPLICATIO S I FOR ATIO
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.4A.
The inductor value also has an effect on low current
operation. Lower inductor values (higher ∆IL) will cause
Burst Mode operation to begin at higher load currents,
which can cause a dip in efficiency in the upper range of
low current operation. In Burst Mode operation, lower
inductance values will cause the burst frequency to decrease.
Inductor Core Selection
Once the value for L is selected, the type of inductor must
be chosen. Basically, there are two kinds of losses in an
inductor —core and copper losses.
Catch Diode Selection
The diode D1 shown in Figure 1 conducts during the offtime. It is important to adequately specify the diode peak
current and average power dissipation so as not to exceed
the diode ratings.
Losses in the catch diode depend on forward drop and
switching times. Therefore, Schottky diodes are a good
choice for low drop and fast switching times.
Since the catch diode carries the load current during the
off-time, the average diode current is dependent on the
switch duty cycle. At high input voltages, the diode conducts most of the time. As VIN approaches VOUT, the diode
conducts only a small fraction of the time. The most
stressful condition for the diode is when the regulator
output is shorted to ground.
Core losses are dependent on the peak-to-peak ripple
current and core material. However, it is independent of
the physical size of the core. By increasing inductance, the
peak-to-peak inductor ripple current will decrease, therefore reducing core loss. Unfortunately, increased inductance requires more turns of wire and, therefore, copper
losses will increase.
Under short-circuit conditions (VOUT = 0V), the diode
must safely handle ISC(PK) at close to 100% duty cycle.
Under normal load conditions, the average current conducted by the diode is simply:
High efficiency converters generally cannot afford the core
loss found in low cost powdered iron cores, forcing the
use of more expensive ferrite, molypermalloy or Kool Mµ®
cores. Ferrite designs have very low core loss and are
preferred at high switching frequencies. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded.
This results in an abrupt increase in inductor ripple current
and consequent output voltage ripple. Do not allow the
core to saturate!
Remember to keep lead lengths short and observe proper
grounding (see Board Layout Considerations) to avoid
ringing and increased dissipation.
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ core material. Toroids are very space
efficient, expecially when you can use several layers of
wire. Because they generally lack a bobbin, mounting is
more difficult. However, surface mount designs that do
not increase the height significantly are available
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V −V 
IDIODE(avg) = ILOAD(avg)  IN OUT 
 VIN + VD 
The forward voltage drop allowed in the diode is calculated
from the maximum short-circuit current as:
 P  V + V 
VD ≈  D   IN D 
 ISC(avg)   VIN 
where PD is the allowable diode power dissipation and will
be determined by efficiency and/or thermal requirements
(see Efficiency Considerations).
Most LTC1701 circuits will be well served by either an
MBR0520L or an MBRM120L. An MBR0520L is a good
choice for IOUT(MAX) ≤ 500mA, as long as the output
doesn’t need to sustain a continuous short.
Kool Mµ is a registered trademark of Magnetics, Inc.
LTC1701/LTC1701B
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APPLICATIO S I FOR ATIO
Input Capacitor (CIN) Selection
In continuous mode, the input current of the converter is
a square wave with a duty cycle of approximately VOUT/
VIN. To prevent large voltage transients, a low equivalent
series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS
capacitor current is given by:
IRMS ≈ IMAX
(
VOUT VIN − VOUT
)
VIN
where the maximum average output current IMAX equals
the peak current (1 Amp) minus half the peak-to-peak
ripple current, IMAX = 1 – ∆IL/2.
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT/2. This simple worst-case is commonly used to
design because even significant deviations do not offer
much relief. Note that capacitor manufacturer’s ripple
current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature
than required. Several capacitors may also be paralleled to
meet the size or height requirements of the design. An
additional 0.1µF to 1µF ceramic capacitor is also recommended on VIN for high frequency decoupling.
Output Capacitor (COUT) Selection
The selection of COUT is driven by the required ESR.
Typically, once the ESR requirement is satisfied, the
capacitance is adequate for filtering. The output ripple
(∆VOUT) is determined by:

1 
∆VOUT ≈ ∆IL  ESR +


8 fCOUT 
where f = operating frequency, COUT = output capacitance
and ∆IL = ripple current in the inductor. With ∆IL = 0.4
IOUT(MAX) the output ripple will be less than 100mV with:
ESRCOUT < 100mΩ
Once the ESR requirements for COUT have been met, the
RMS current rating generally far exceeds the IRIPPLE(P-P)
requirement.
When the capacitance of COUT is made too small, the
output ripple at low frequencies will be large enough to trip
the ITH comparator. This causes Burst Mode operation to
be activated when the LTC1701 would normally be in
continuous mode operation. The effect can be improved at
higher frequencies with lower inductor values.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the capacitance, ESR or RMS
current handling requirement of the application. Aluminum electrolyte and dry tantulum capacitors are both
available in surface mount configurations. In the case of
tantalum, it is critical that the capacitors are surge tested
for use in switching power supplies. An excellent choice is
the AVX TPS, AVX TPSV and KEMET T510 series of
surface mount tantalums, avalable in case heights ranging
from 2mm to 4mm. Other capacitor types include Nichicon
PL series, Sanyo POSCAP and Panasonic SP.
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generates a loop “zero” at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually
resonate with their ESL before ESR becomes effective.
Also, ceramic caps are prone to temperature effects which
requires the designer to check loop stability over the
operating temperature range.
For these reasons, most of the input and output capacitance should be composed of tantalum capacitors for
stability combined with about 0.1µF to 1µF of ceramic
capacitors for high frequency decoupling. Great care must
be taken when using only ceramic input and output capacitors. The OPTI-LOOP compensation allows transient response to be optimized for all types of output capacitors,
including low ESR ceramics.
Setting the Output Voltage
The LTC1701 develops a 1.25V reference voltage between
the feedback pin, VFB, and the signal ground as shown in
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LTC1701/LTC1701B
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APPLICATIO S I FOR ATIO
Figure 2. The output voltage is set by a resistive divider
according to the following formula:
VOUT = 1.25V(1 + R2/R1)
To prevent stray pickup, a capacitor of about 5pF can be
added across R1, located close to the LTC1701. Unfortunately, the load step response is degraded by this capacitor. Using a good printed circuit board layout eliminates
the need for this capacitor. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
VOUT
CF
LTC1701
R2
1%
VFB
SGND
5pF
R1
1%
1701 F02
Figure 2. Setting the Output Voltage
Transient Response
The OPTI-LOOP compensation allows the transient response to be optimized for a wide range of loads and
output capacitors. The availability of the ITH pin not only
allows optimization of the control loop behavior but also
provides a DC coupled and AC filtered closed-loop response test point. The DC step, rise time and settling at this
test point truly reflects the closed-loop response. Assuming a predominately second order system, phase margin
and/or damping factor can be estimated using the percentage of overshoot seen at this pin. The bandwidth can also
be estimated by examining the rise time at the pin.
The ITH external components shown in the Figure 1 circuit
will provide an adequate starting point for most applications. The series R3-C3 filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and the
particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
feedback factor gain and phrase. An output current pulse
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of 20% to 100% of full-load current having a rise time of
1µs to 10µs will produce output voltage and ITH pin
waveforms that will give a sense of the overall loop
stability without breaking the feedback loop.
The initial output voltage step may not be within the
bandwidth of the feedback loop, so the standard secondorder overshoot/DC ratio cannot be used to determine
phase margin. The gain of the loop increases with R3 and
the bandwidth of the loop increases with decreasing C3. If
R3 is increased by the same factor that C3 is decreased,
the zero frequency will be kept the same, thereby keeping
the phase the same in the most critical frequency range of
the feedback loop. In addition, a feed-forward capacitor,
CF, can be added to improve the high frequency response,
as shown in Figure 2. Capacitor CF provides phase lead by
creating a high frequency zero with R2 which improves the
phase margin.
The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance. For a detailed
explanation of optimizing the compensation components,
including a review of control loop theory, refer to Application Note 76.
RUN Function
The ITH/RUN pin is a dual purpose pin that provides the
loop compensation and a means to shut down the LTC1701.
Soft-start can also be implemented with this pin. Soft-start
reduces surge currents from VIN by gradually increasing
the internal peak inductor current. Power supply sequencing can also be accomplished using this pin.
An external pull-up is required to charge the external
capacitor C3 in Figure 1. Typically, a 1M resistor between
VIN and ITH/RUN is used. When the voltage on ITH/RUN
reaches about 0.8V the LTC1701 begins operating. At this
point the error amplifier pulls up the ITH/RUN pin to the
normal operating range of 1.25V to 2.25V.
Soft-start can be implemented by ramping the voltage on
ITH/RUN during start-up as shown in Figure 3(b). As the
voltage on ITH/RUN ramps through its operating range the
internal peak current limit is also ramped at a proportional
linear rate.
LTC1701/LTC1701B
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APPLICATIO S I FOR ATIO
During normal operation the voltage on the ITH/RUN pin
will vary from 1.25V to 2.25V depending on the load
current. Pulling the ITH/RUN pin below 0.8V puts the
LTC1701 into a low quiescent current shutdown mode
(IQ < 1µA). This pin can be driven directly from logic as
shown in Figures 3(a).
ITH/RUN
ITH/RUN
R1
D1
CC
RC
(a)
CC
C1
(b)
RC
1701 F03
Figure 3. ITH/RUN Pin Interfacing
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and what change would
produce the most improvement. Percent efficiency can be
expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, 4 main sources usually account for most of the
losses in LTC1701 circuits: 1) LTC1701 VIN current,
2)␣ switching losses, 3) I2R losses, 4) Schottky diode
losses.
1) The VIN current is the DC supply current given in the
electrical characteristics which excludes MOSFET driver
and control currents. VIN current results in a small (< 0.1%)
loss that increases with VIN, even at no load.
2) The switching current is the sum of the internal MOSFET
driver and control currents. The MOSFET driver current
results from switching the gate capacitance of the power
MOSFET. Each time a MOSFET gate is switched from low
to high to low again, a packet of charge dQ moves from VIN
to ground. The resulting dQ/dt is a current out of VIN that
is typically much larger than the control circuit current. In
continuous mode, IGATECHG = f • QP, where QP is the gate
charge of the internal MOSFET switch.
3) I2R Losses are predicted from the DC resistances of the
MOSFET and inductor. In continuous mode the average
output current flows through L, but is “chopped” between
the topside internal MOSFET and the Schottky diode. At
low supply voltages where the switch on-resistance is
higher and the switch is on for longer periods due to the
higher duty cycle, the switch losses will dominate. Using
a larger inductance helps minimize these switch losses. At
high supply voltages, these losses are proportional to the
load. I2R losses cause the efficiency to drop at high output
currents.
4) The Schottky diode is a major source of power loss at
high currents and gets worse at low output voltages. The
diode loss is calculated by multiplying the forward voltage
drop times the diode duty cycle multiplied by the load
current.
Other “hidden” losses such as copper trace and internal
battery resistances can account for additional efficiency
degradations in portable systems. It is very important to
include these “system” level losses in the design of a
system. The internal battery and fuse resistance losses
can be minimized by making sure that CIN has adequate
charge storage and very low ESR at the switching frequency. Other losses including Schottky conduction losses
during dead-time and inductor core losses generally account for less than 2% total additional loss.
THERMAL CONSIDERATIONS
The power handling capability of the device at high ambient temperatures will be limited by the maximum rated
junction temperature (125°C). It is important to give
careful consideration to all sources of thermal resistance
from junction to ambient. Additional heat sources mounted
nearby must also be considered.
For surface mount devices, heat sinking is accomplished
by using the heat spreading capabilities of the PC board
and its copper traces. Copper board stiffeners and plated
through-holes can also be used to spread the heat generated by power devices.
9
LTC1701/LTC1701B
U
W
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APPLICATIO S I FOR ATIO
The following table lists thermal resistance for several
different board sizes and copper areas. All measurements
were taken in still air on 3/32" FR-4 board with one ounce
copper.
Table 1. Measured Thermal Resistance
COPPER AREA
THERMAL RESISTANCE
TOPSIDE*
BACKSIDE
BOARD AREA
θJA
2500mm2
2500mm2
2500mm2
125°C/W
1000mm2
2500mm2
2500mm2
125°C/W
225mm2
2500mm2
2500mm2
130°C/W
100mm2
2500mm2
2500mm2
135°C/W
50mm2
2500mm2
2500mm2
150°C/W
*Device is mounted on topside.
Calculating Junction Temperature
In a majority of applications, the LTC1701 does not dissipate much heat due to its high efficiency. However, in
applications where the switching regulator is running at
high duty cycles or the part is in dropout with the switch
turned on continuously (DC), some thermal analysis is
required. The goal of the thermal analysis is to determine
whether the power dissipated by the regulator exceeds the
maximum junction temperature. The temperature rise is
given by:
TRISE = PD • θJA
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to the
ambient temperature.
The junction temperature is given by:
TJ = TRISE + TAMBIENT
As an example, consider the case when the LTC1701 is in
dropout at an input voltage of 3.3V with a load current of
0.5A. The ON resistance of the P-channel switch is approximately 0.30Ω. Therefore, power dissipated by the part is:
PD = I2 • RDS(ON) = 75mW
The SOT package junction-to-ambient thermal resistance,
θJA, will be in the range of 125°C/W to 150°C/W. Therefore,
the junction temperature of the regulator operating in a
25°C ambient temperature is approximately:
TJ = 0.075 • 150 + 25 = 36°C
10
Remembering that the above junction temperature is obtained from a RDS(ON) at 25°C, we might recalculate the
junction temperature based on a higher RDS(ON) since it
increases with temperature. However, we can safely assume that the actual junction temperature will not exceed
the absolute maximum junction temperature of 125°C.
Board Layout Considerations
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1701. These items are also illustrated graphically in the
layout diagram of Figure 4. Check the following in your
layout:
1. Does the capacitor CIN connect to the power VIN (Pin 5)
and GND (Pin 2) as close as possible? This capacitor
provides the AC current to the internal P-channel MOSFET
and its driver.
2. Is the Schottky diode closely connected between the
ground (Pin 2) and switch output (Pin 1)?
3. Are the COUT, L1 and D1 closely connected? The Schottky
anode should connect directly to the input capacitor ground.
4. The resistor divider, R1 and R2, must be connected
between the (+) plate of COUT and a ground line terminated
near GND (Pin 2). The feedback signal FB should be routed
away from noisy components and traces, such as the SW
line (Pin 1).
5. Keep sensitive components away from the SW pin. The
input capacitor CIN, the compensation capacitor CC and all
the resistors R1, R2, RC and RS should be routed away from
the SW trace and the components L1 and D1.
L1
VOUT
1
SW
+
5
VIN
+
LTC1701
D1
COUT
VIN
CIN
2 GND
R2
3
VFB
R1
ITH/RUN
4
RS
RC
CC
BOLD LINES INDICATE HIGH CURRENT PATHS
1701 F04
Figure 4. LTC1701 Layout Diagram (See Board Layout Checklist)
LTC1701/LTC1701B
U
TYPICAL APPLICATIO S
2mm Nominal Height 1.5V Converter
90
L1
4.7µH
VIN
2.5V TO
5.5V
VIN
SW
R2
20k
ITH/RUN
GND
R3
5.1k
C1: AVX TAJA156M010R
C2: AVX TAJA226M006R
C2
22µF
VFB
+
C5
4.7µF
R1
100k
C3
330pF
V IN = 3.3V
80
LTC1701
C4
1µF
V IN = 2.5V
85
D1
R4
1M
+
VOUT
(1.5V/0.5A)
EFFICIENCY (%)
C1
15µF
Efficiency Curve
75
70
65
60
C4: TAIYO YUDEN LMK212BJ105MG
C5: TAIYO YUDEN JMK212BJ475MG
D1: MBRM120L
L1: MURATA LQH3C4R7M24
55
1701 TA01a
LTC1701
LTC1701B
VOUT = 1.5V
50
10
100
LOAD CURRENT (mA)
1
1000
1701TA01b
Efficiency Curve
All Ceramic Capacitor 2.5V Converter
L1
4.7µH
VIN
2.5V TO
5.5V
VIN
SW
VOUT = 2.5V
95
V IN = 3.3V
LTC1701
C4
1µF
ITH/RUN
R3
5.1k
R2
121k
C5
1µF
C2
10µF
VFB
EFFICIENCY (%)
D1
R4
1M
C1
10µF
100
VOUT
(2.5V/0.5A)
GND
C6
33pF
R1
121k
C3
180pF
C1, C2: TAIYO YUDEN JMK316BJ106ML
C4, C5: TAIYO YUDEN LMK212BJ105MG
90
V IN = 5.0V
85
80
75
L1: MURATA LQH3C4R7M24
D1: MBRM120L
LTC1701
LTC1701B
1701 TA02
70
1
10
100
LOAD CURRENT (mA)
1000
1701 TA02b
LTC1701B Low Current Pulse Skip
5V to 3.3V Converter with Push-Button On/Off
L1
4.7µH
VIN
3.3V TO
5.5V
VIN
VOUT
(3.3V/
0.5A)
SW
D1
ON
+
C1
15µF
LTC1701
R4
1M
C4
1µF
OFF
R2
34k
ITH/RUN
GND
R5
5.1M
R3
5.1k
VFB
C2
22µF
+
VOUT
20mV/DIV
C5
1µF
R1
20.5k
C3
330pF
C1: AVX TAJA156M010R
C2: AVX TAJA226M006R
C4, C5: TAIYO YUDEN LMK212BJ105MG
IL
50mA/DIV
D1: MBRM120L
L1: MURATA LQH3C4R7M24
1701 TA03a
VIN = 5V
VOUT = 2.5V
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
5µs/DIV
1701 TA03b
11
LTC1701/LTC1701B
U
TYPICAL APPLICATIO
Single Cell Li-Ion to 3.3V Zeta Converter
C6
4.7µF
+
VIN
2.5V TO 4.2V
VIN
+
C4
1µF
×5R
VOUT
(3.3V)
SW
R4
1M
L2
D1
+
LTC1701
C1
22µF
R2
34k
ITH/RUN
GND
R3
5.1k
U
PACKAGE DESCRIPTIO
IOUT(MAX)
2.5V
200mA
3.0V
225mA
3.5V
250mA
4.0V
280mA
4.2V
290mA
1701 TA04
Dimensions in inches (millimeters) unless otherwise noted.
S5 Package
5-Lead Plastic SOT-23
2.60 – 3.00
(0.102 – 0.118)
1.50 – 1.75
(0.059 – 0.069)
VIN
R1
20.5k
D1: MBR0520L
L1, L2: SUMIDA CLQ72-4R7
DRG NO 6333-JPS-010
C1, C2: AVX TAJA226M006R
C6: TAIYO YUDEN JMK212BJ475MG
C2
22µF
VFB
C3
330pF
0.35 – 0.55
(0.014 – 0.022)
L1
4.7µH
(LTC DWG # 05-08-1633)
0.00 – 0.15
(0.00 – 0.006)
0.09 – 0.20
(0.004 – 0.008)
(NOTE 2)
2.80 – 3.00
(0.110 – 0.118)
(NOTE 3)
0.90 – 1.45
(0.035 – 0.057)
0.35 – 0.50
0.90 – 1.30
(0.014 – 0.020)
(0.035 – 0.051)
FIVE PLACES (NOTE 2)
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DIMENSIONS ARE INCLUSIVE OF PLATING
3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
4. MOLD FLASH SHALL NOT EXCEED 0.254mm
5. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ)
1.90
(0.074)
REF
0.95
(0.037)
REF
S5 SOT-23 0599
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12
Linear Technology Corporation
1701Bfa LT/TP 1100 REV A 2K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
 LINEAR TECHNOLOGY CORPORATION 1999
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