Microsemi LX1681CDM Voltage-mode pwm controller Datasheet

LX1681/1682
®
TM
Voltage-Mode PWM Controllers
P RODUCTION D ATA S HEET
KEY FEATURES
DESCRIPTION
ƒ Fixed 200kHz Switching Frequency
ƒ Constant Frequency Voltage-Mode
Control Requires NO External
Compensation
ƒ Hiccup-Mode Over-Current
Protection
ƒ High Efficiency
ƒ Output Voltage Set By Resistor
Divider
ƒ Under-Voltage Lockout
ƒ Soft-Start And Enable
ƒ Synchronous Rectification
(LX1682)
ƒ Non-Synchronous Rectification
(LX1681)
ƒ Small, 8-pin Surface Mount
Package
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Hiccup-mode fault protection reduces
average power to the power elements
during short-circuit conditions.
Switching frequency is fixed at
200kHz for optimal cost and space.
Under-voltage lockout and soft-start
for optimal start-up performance. Pulling
the soft-start pin to ground can disable
the LX1681/82.
Small 8-pin SOIC packaging reduces
board space. Optimized for 5V-to-3.3V
or
5V-to-2.5V
conversion,
the
LX1681/82 can also be used for
converting 12V to 5V, 3.3V or other
voltages
with
high
efficiency,
eliminating the need for bulky heat sinks.
The LX1681/1682 are monolithic,
pulse-width modulator controller ICs.
They are designed to implement a
flexible, low cost buck (step-down)
regulator supply with minimal external
components.
The LX1681 is a non-synchronous
controller;
the
LX1682
has
a
synchronous driver for higher efficiency.
The output voltage is adjustable by
means of a resistor divider to set the
voltage between 1.25V and 4.5V.
Short-circuit current limiting can be
implemented without expensive current
sense resistors. Current is sensed using
the voltage drop across the RDS(ON) of the
MOSFET — sensing is delayed for 1µs
to eliminate MOSFET ringing errors.
APPLICATIONS
ƒ 5V to 3.3V Or Less Buck
Regulators
ƒ FPGA Supplies
ƒ Microprocessor Chipset Supplies
(e.g. Camino, Whitney, etc.)
©
ƒ Rambus RIMM™ Supplies
ƒ Hard Disk Drives
ƒ Computer Add-on Cards
IMPORTANT: For the most current data, consult MICROSEMI’s website: http://www.microsemi.com
PRODUCT HIGHLIGHT
V BO OST
12V
V IN
5V
C3
C3
V BOO ST
12V
1µF
1µF
C1
C SS
V FB
V C2
SS
CS
C1
1500µF
x3
V FB
C SS
N.C.
GND
C2
L1
5µH
R1
1500µF
x3
V OU T
V C1
G ND
Q1
IRL3103S
TDRV
CS
LX1682
V OU T
V C1
1500µF
x3
V C2
SS
0.1µF
LX1681
0.1µF
V IN
5V
BDRV
Q1
IRL3103S
C2
L1
5µH
TDRV
R SET
R SET
D2
R2
R1
1500µF
x3
R2
Q2
IRL3103S
MBR2545
LX1681/1682
LX1682 Synchronous Controller
LX1681 Non-Synchronous Controller
PACKAGE ORDER INFO
TA (°C)
OUTPUT
0 to 70
Non-Synchronous
Synchronous
DM
Plastic SOIC
8-PIN
RoHS Compliant / Pb-free
Transition D/C: 0440
LX1681CDM
LX1682CDM
Note: Available in Tape & Reel. Append the letters “TR” to the part number. (i.e. LX1681CDM-TR)
Copyright © 2000
Rev. 1.1b,2005-03-09
Microsemi
Linfinity Microelectronics Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 1
LX1681/1682
®
TM
Voltage-Mode PWM Controllers
P RODUCTION D ATA S HEET
ABSOLUTE MAXIMUM RATINGS (NOTE 1)
V FB
1
8
V CC
SS
2
7
CS
N.C.
3
6
V C1
GND
4
5
TDRV
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Supply Voltage (VC1).................................................................................................... 18V
Supply Voltage (VCC) ..................................................................................................... 7V
Output Drive Peak Current Source (500ns) ................................................................. 1.0A
Output Drive Peak Current Sink (500ns) ..................................................................... 1.0A
Input Voltage (SS/ENABLE Pin) ........................................................................ -0.3 to 6V
Operating Junction Temperature................................................................................ 150°C
Storage Temperature................................................................................. -65°C to +150°C
PACKAGE PIN OUT
LX1681
DM P ACKAGE
RoHS / Pb-free Peak Package Solder Reflow Temp (40 second max. exposure) ........... 260°C (+0, -5)
(Top View)
Note:
Exceeding these ratings could cause damage to the device. All voltages are with respect to
Ground. Currents are positive into, negative out of specified terminal.
V FB
1
8
V CC
SS
2
7
CS
GND
3
6
V C1
BDRV
4
5
TDRV
THERMAL DATA
LX1682
DM P ACKAGE
(Top View)
DM
RoHS / Pb-free 100% Matte Tin Lead Finish
PACKAGE
THERMAL RESISTANCE-JUNCTION TO AMBIENT, θJA
165°C/W
Junction Temperature Calculation: TJ = TA + (PD x θJA).
The θJA numbers are guidelines for the thermal performance of the device/pc-board
system. All of the above assume no ambient airflow.
FUNCTIONAL PIN DESCRIPTION
PIN NAME
DESCRIPTION
VFB
Voltage Feedback. A 1.25V reference is connected to a resistor divider to set desired output voltage.
SS
Soft-Start And Hiccup Capacitor Pin. During start up the voltage of this pin controls the output voltage. An
internal 20kΩ resistor and the external capacitor set the time constant for soft-startup. Soft-start does not begin
until the supply voltage exceeds the UVLO threshold. When over-current occurs, this capacitor is used for timing
hiccup. The PWM can be disabled by pulling the SS pin below 0.3V
Ground for IC.
TDRV
Gate Drive For Upper MOSFET.
BDRV
Gate Drive For Lower MOSFET.
VC1
Separate Supply For MOSFET Gate Drive. Connect to 12V.
CS
Over-Current Set. Connect resistor between CS pin and the source of the upper MOSFET to set current-limit
point.
VCC
IC Supply Voltage (nominal 5V) And High Side Drain Sense Voltage.
Copyright © 2000
Rev. 1.1b,2005-03-09
Microsemi
Linfinity Microelectronics Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
PACKAGE DATA
GND
Page 2
LX1681/1682
®
TM
Voltage-Mode PWM Controllers
P RODUCTION D ATA S HEET
ELECTRICAL CHARACTERISTICS
Parameter
`
`
VFB
Frequency
FOSC
Ramp Amplitude
VRAMP
VOUT=VFB, TA=25°C
1.237
VOUT=VFB, 0°C < TA < 70°C
1.231
1.262
V
1.269
V
170
190
230
kHz
1.25
VPP
20
kΩ
VOUT=VFB
ISET
VCS = VCC –0.4V
40
45
µA
Reference to VCC
40
45
µA
1.1
µsec
50
Ns
11
V
CURRENT SENSE
Current Sense Delayed
TCSD
OUTPUT DRIVERS
Drive Rise Time, Fall Time
TRF
CL=3000pF
Drive High
VDH
ISOURCE=20mA, VC1=12V
Drive Low
VDL
ISINK=20mA, VC1=12V
VST
VC1 > 4.0V
10
0.1
0.2
V
4.25
4.5
V
UVLO AND SOFT-START (SS)
VCC5 Start-Up Threshold
4.0
Hysteresis
SS Resistor
RSS
SS Output Enable
VEN
Hiccup Duty Cycle
DCHIC
0.25
0.10
V
20
kΩ
0.3
CSS = 0.1µF, FREQ=100Hz
10
0.35
V
%
SUPPLY CURRENT
VCC12 Dynamic Supply Current
ICD
Out Freq = 200kHz, CL=3000pF, Synch., VSS
> 0.3V
24
28
mA
Static Supply Current 12CV
IVC1
VSS < 0.3V
5
7
mA
5V
IVCC
VSS > 0.3V
10
12
mA
Copyright © 2000
Rev. 1.1b,2005-03-09
Microsemi
Linfinity Microelectronics Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 3
ELECTRICALS
`
1.25
RIN
VTRIP
`
Units
ERROR AMPLIFIER
Current Set
`
LX1681/1682
Min
Typ
Max
OSCILLATOR
Input Resistance
`
Test Conditions
REFERENCE
Reference Voltage
`
Symbol
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Unless otherwise specified, the following specifications apply over the operating ambient temperature 0°C ≤ TA ≤ 70°C except where
otherwise noted. Test conditions: VCC=5V, VC1=12V, T=25°C
LX1681/1682
®
TM
Voltage-Mode PWM Controllers
P RODUCTION D ATA S HEET
THEORY OF OPERATION
OVER-CURRENT PROTECTION (OCP) and HICCUP
The LX1681/82 are voltage-mode pulse-width modulation
controller integrated circuits. The internal oscillator and ramp
generator frequency is fixed at 200kHz. The devices have internal
compensation, so that no external compensation is required.
The LX1681/1682 family uses the RDS(ON) of the upper
MOSFET, together with a resistor (RSET) to set the actual current
limit point. The comparator senses the current 1µs after the top
MOSFET is switched on. Experiments have shown that the
MOSFET drain voltage will ring for 200-500ns after the gate is
turned on. In order to reduce inaccuracies due to ringing, a 1µs
delay after gate turn-on is built into the current sense comparator.
The comparator draws a current (ISET), whose magnitude is 45µA.
The set resistor is selected to set the current limit for the
application. When the sensed voltage across the RDS(ON) plus the
set resistor exceeds the 400mV V TRIP threshold, the OCP
comparator outputs a signal to reset the PWM latch and to start
hiccup mode. The soft-start capacitor (CSS) is discharged slowly
(10 times slower than when being charged up by RSS). When the
voltage on the SS/ENABLE pin reaches a 0.3V threshold, hiccup
finishes and the circuit soft-starts again. During hiccup, the top
MOSFET is OFF and the bottom MOSFET remains ON. Hiccup is
disabled during the soft-start interval, allowing the circuit to start
up with the maximum current. If the rise speed of the output
voltage is too fast, the required charging current to the output
capacitor may be higher than the limit-current. In this case, the
peak MOSFET current is regulated to the limit-current by the
current-sense comparator. If the MOSFET current still reaches its
limit after the soft-start finishes, the hiccup is triggered again. The
hiccup ensures the average heat generation on both MOSFET’s and
the average current to be much less than that in normal operation,
if the output has a short circuit. Over-current protection can also be
implemented using a sense resistor, instead of using the RDS(ON) of
the upper MOSFET, for greater set-point accuracy. See
Application Information section.
POWER UP and INITIALIZATION
At power up, the LX1681/82 monitors the supply voltage to
both the +5V and the +12V pins (there is no special requirement
for the sequence of the two supplies). Before both supplies reach
their under-voltage lock-out (UVLO) thresholds, the soft-start (SS)
pin is held low to prevent soft-start from beginning; the oscillator
control is disabled and the top MOSFET is kept OFF.
SOFT-START
Once the supplies are above the UVLO threshold, the soft-start
capacitor begins to be charged up by the reference through a 20k
internal resistor. The capacitor voltage at the SS pin rises as a
simple RC circuit. The SS pin is connected to the amplifier's noninverting input that controls the output voltage. The output voltage
will follow the SS pin voltage if sufficient charging current is
provided to the output capacitor. The simple RC soft-start allows
the output to rise faster at the beginning and slower at the end of
the soft-start interval. Thus, the required charging current into the
output capacitor is less at the end of the soft-start interval so
decreasing the possibility of an over-current. A comparator
monitors the SS pin voltage and indicates the end of soft-start
when SS pin voltage reaches 95% of VREF.
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GENERAL DESCRIPTION
OSCILLATOR FREQUENCY
An internal oscillator sets the switching frequency at 200 kHz.
DESCRIPTION
Copyright © 2000
Rev. 1.1b,2005-03-09
Microsemi
Linfinity Microelectronics Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 4
LX1681/1682
®
TM
Voltage-Mode PWM Controllers
P RODUCTION D ATA S HEET
OUTPUT INDUCTOR
OUTPUT CAPACITOR (continued)
The output inductor should be selected to meet the requirements
of the output voltage ripple in steady-state operation and the
inductor current slew-rate during transient. The peak-to-peak
output voltage ripple is:
Electrolytic capacitors can be used for the output capacitor, but
are less stable with age than tantalum capacitors. As they age, their
ESR degrades, reducing the system performance and increasing the
risk of failure. It is recommended that multiple parallel capacitors
be used, so that, as ESR increases with age, overall performance
will still meet the processor’s requirements.
There is frequently strong pressure to use the least expensive
components possible, however, this could lead to degraded longterm reliability, especially in the case of filter capacitors.
Linfinity’s demonstration boards use Sanyo MV-GX filter
capacitors, which are aluminum electrolytic, and have
demonstrated reliability. The Oscon series from Sanyo generally
provides the very best performance in terms of long term ESR
stability and general reliability, but at a substantial cost penalty.
The MV-GX series provides excellent ESR performance at a
reasonable cost. Beware of off-brand, very low-cost filter
capacitors, which have been shown to degrade in both ESR and
general electrolytic characteristics over time.
VRIPPLE = ESR × IRIPPLE
where
IRIPPLE =
(VIN - VOUT ) × VOUT
fSW × L
VIN
IRIPPLE is the inductor ripple current, L is the output inductor
value and ESR is the Effective Series Resistance of the output
capacitor.
IRIPPLE should typically be in the range of 20% to 40% of the
maximum output current. Higher inductance results in lower
output voltage ripple, allowing slightly higher ESR to satisfy the
transient specification. Higher inductance also slows the inductor
current slew rate in response to the load-current step change, ∆I,
resulting in more output-capacitor voltage droop. The inductorcurrent rise and fall times are:
TRISE =
L × ∆I
(VIN − VOUT )
and
TFALL =
L × ∆I
VOUT
INPUT CAPACITOR
The input capacitor and the input inductor are to filter the
pulsating current generated by the buck converter to reduce
interference to other circuits connected to the same 5V rail. In
addition, the input capacitor provides local de-coupling the buck
converter. The capacitor should be rated to handle the RMS current
requirement. The RMS current is:
IRMS = IL d (1 − d )
When using electrolytic capacitors, the capacitor voltage droop
is usually negligible, due to the large capacitance.
OUTPUT CAPACITOR
ESR × (IRIPPLE + ∆I ) < VEX
where IRIPPLE is the inductor ripple current, ∆I is the maximum
load current step change, and VEX is the allowed output voltage
excursion in the transient.
where IL is the inductor current and the d is the duty cycle. The
maximum value, when d = 50%, IRMS = 0.5IL . For 5V input and
output in the range of 2 to 3V, the required RMS current is very
close to 0.5IL.
SOFT-START CAPACITOR
The value of the soft-start capacitor determines how fast the
output voltage rises and how large the inductor current is required
to charge the output capacitor. The output voltage will follow the
voltage at SS pin if the required inductor current does not exceed
the maximum current in the inductor. The SS pin voltage can be
expressed as:
VSS = VSET (1 − e −t / RSSCSS )
where VSET is the reference voltage. RSS and CSS are soft start
resistor and capacitor. The required inductor current for the output
capacitor to follow the SS-pin voltage equals the required capacitor
current plus the load current. The soft-start capacitor should be
selected so that the overall inductor current does not exceed it
maximum.
Microsemi
Linfinity Microelectronics Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 5
APPLICATION
The output capacitor is sized to meet ripple and transient
performance specifications. Effective Series Resistance (ESR) is a
critical parameter. When a step load current occurs, the output
voltage will have a step that equals the product of the ESR and the
current step, ∆I. In an advanced microprocessor power supply, the
output capacitor is usually selected for ESR instead of capacitance
or RMS current capability. A capacitor that satisfies the ESR
requirement usually has a larger capacitance and current capability
than strictly needed. The allowed ESR can be found by:
Copyright © 2000
Rev. 1.1b,2005-03-09
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APPLICATION INFORMATION
LX1681/1682
®
TM
Voltage-Mode PWM Controllers
P RODUCTION D ATA S HEET
SOFT-START CAPACITOR (continued)
OUTPUT ENABLE
The capacitor current to follow the SS-pin voltage is:
The LX1681/82 FET driver outputs are driven to ground by
pulling the soft-start pin below 0.3V.
ICOUT = COUT
dV COUT −(t / RSSCSS )
=
×e
dt
CSS
PROGRAMMING THE OUTPUT VOLTAGE
where COUT is the output capacitance. The typical value of CSS
should be in the range of 0.1 to 0.2µF.
During the soft-start interval the load current from a microprocessor is negligible; therefore, the capacitor current is
approximately the required inductor current.
The output voltage is sensed by the feedback pin (VFB ) which
has a 1.25V reference. The output voltage can be set to any voltage
above 1.25V (and lower than the input voltage) by means of a
resistor divider (see Product Highlight).
VOUT = VREF (1 +
OVER-CURRENT PROTECTION
Current limiting occurs at current level ICL , when the voltage
detected by the current sense comparator is greater than the current
sense comparator threshold, VTRIP (400mV).
ICL × RDS (ON ) + ISET × RSET = VTRIP
So,
RSET =
VTRIP − ICL × RDS (ON )
ISET
RSET =
400mV − ICL × RDS (ON )
45µA
Example:
For 10A current limit, using IRL3303 MOSFET (26mΩ RDS(ON) ):
RSET =
0.4 − 10 × 0.026
= 3.1kΩ
45 ×10 −6
Current Sensing Using Sense Resistor
Example:
For 10A current limit, using a 5µ sense resistor:
RSET =
VTRIP − ( ICL × RSENSE )
ISET
RSET =
0.4 − 10 × 0.005
= 7.8kΩ
45 × 10 −6
Copyright © 2000
Rev. 1.1b,2005-03-09
Note: Keep R1 and R2 close to 100(order of magnitude).
FET SELECTION
To insure reliable operation, the operating junction temperature
of the FET switches must be kept below certain limits. The Intel
specification states that 115°C maximum junction temperature
should be maintained with an ambient of 50°C. This is achieved by
properly derating the part, and by adequate heat sinking. One of the
most critical parameters for FET selection is the RDS(ON) resistance.
This parameter directly contributes to the power dissipation of the
FET devices, and thus impacts heat sink design, mechanical layout,
and reliability. In general, the larger the current handling capability
of the FET, the lower the RDS(ON) will be, since more die area is
available.
This table gives selection of suitable FETs from International Rectifier.
RDS(ON)
ID @
Max. BreakDevice
@10V(mΩ)
TC=100ºC
down Voltage
IRL3803
6
83
30
IRL22203N
7
71
30
IRL3103
14
40
30
IRL3102
13
56
20
IRL3303
26
24
30
IRL2703
40
17
30
All devices in TO-220 package. For surface mount devices (TO-263 /
D 2 -Pak), add 'S' to part number, e.g. IRL3103S.
TABLE 1 - FET Selection Guide
Heat Dissipated In Upper MOSFET
The heat dissipated in the top MOSFET will be:
PD = ( I 2 × RDS (ON ) × Duty Cycle ) + (0.5 × I × VIN × tSW × fS )
Where t SW is switching transition line for body diode (~100ns) and
fS is the switching frequency.
For the IRL3102 (13µ RDS(ON) ), converting 5V to 2.0V at 15A
will result in typical heat dissipation of 1.92W.
Microsemi
Linfinity Microelectronics Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 6
APPLICATION
The method of current sensing using the RDS(ON) of the upper
MOSFET is economical, but can have a large tolerance, since the
RDS(ON) can vary with temperature, etc. A more accurate alternative
is to use an external sense resistor (RSENSE ). Since one input to the
current sense comparator is the supply voltage to the IC (VCC - pin
8), the sense resistor could be a PCB trace (for construction details,
see Application Note AN-10 or LX1668 data sheet). The overcurrent trip point is calculated as in the equations above, replacing
RDS(ON) with RSENSE .
R1
)
R2
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APPLICATION INFORMATION
LX1681/1682
®
TM
Voltage-Mode PWM Controllers
P RODUCTION D ATA S HEET
FET SELECTION (continued)
Synchronous Rectification – Lower MOSFET
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APPLICATION INFORMATION
5V Input
The lower pass element can be either a MOSFET or a Schottky
diode. The use of a MOSFET (synchronous rectification) will
result in higher efficiency, but at higher cost than using a Schottky
diode (non-synchronous). Power dissipated in the bottom
MOSFET will be:
PD = I 2 × RDS (ON ) × [1 − Duty Cycle ] = 3.51W
Output
LX168x
[IRL3303 or 1.76W for the IRL3102]
Non-Synchronous Operation - Schottky Diode
A typical Schottky diode, with a forward drop of 0.6V will
dissipate 0.6 * 15 * [1 – 2/5] = 5.4W (compared to the 1.8 to 3.5W
dissipated by a MOSFET under the same conditions). This power
loss becomes much more significant at lower duty cycles. The use
of a dual Schottky diode in a single TO-220 package (e.g. the
MBR2535) helps improve thermal dissipation.
GND
Operation From A Single Power Supply
The LX1681/1682 needs a secondary supply voltage (VC1) to
provide sufficient drive to the upper MOSFET. In many
applications with a 5V (VCC) and a 12V (VC1) supply are present.
In situations where only 5V is present, VC1 can be generated using
a bootstrap (charge pump) circuit, as shown in Figure 4 (Typical
Applications section). The capacitor (C4) is alternatively charged
up from VCC via the Schottky diode (D2), and then boosted up
when the FET is turned on. This scheme provides a VC1 voltage
equal to 2 * VCC - VDS (D2), or approximately 9.5V with VCC = 5V.
This voltage will provide sufficient gate drive to the external
MOSFET in order to get a low RDS(ON) . Note that using the
bootstrap circuit in synchronous rectification mode is likely to
result in faster turn-on than in non-synchronous mode.
LAYOUT GUIDELINES - THERMAL DESIGN
Copyright © 2000
Rev. 1.1b,2005-03-09
General Notes
As always, be sure to provide local capacitive decoupling close
to the chip. Be sure use ground plane construction for all highfrequency work. Use low ESR capacitors where justified, but be
alert for damping and ringing problems. High-frequency designs
demand careful routing and layout, and may require several
iterations to achieve desired performance levels.
Power Traces
To reduce power losses due to ohmic resistance, careful consideration should be given to the layout of traces that carry high
currents. The main paths to consider are:
ƒ
Input power from 5V supply to drain of top MOSFET.
ƒ
Trace between top MOSFET and lower MOSFET or
Schottky diode.
ƒ
Trace between lower MOSFET or Schottky diode and
ground.
ƒ
Trace between source of top MOSFET and inductor and
load.
All of these traces should be made as wide and thick as possible,
in order to minimize resistance and hence power losses. It is also
recommended that, whenever possible, the ground, input and
output power signals should be on separate planes (PCB layers).
See Figure 2 – bold traces are power traces.
Layout Assistance
Please contact Linfinity’s Applications Engineers for assistance
with any layout or component selection issues. A Gerber file with
layout for the most popular devices is available upon request.
Evaluation boards are also available upon request. Please check
Linfinity's web site for further application notes.
Microsemi
Linfinity Microelectronics Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 7
APPLICATION
A great deal of time and effort were spent optimizing the
thermal design of the demonstration boards. Any user who intends
to implement an embedded motherboard would be well advised to
carefully read and follow these guidelines. If the FET switches
have been carefully selected, external heatsinking is generally not
required. However, this means that copper trace on the PC board
must now be used. This is a potential trouble spot; as much copper
area as possible must be dedicated to heatsinking the FET
switches, and the diode as well if a non-synchronous solution is
used. In our VRM module, heatsink area was taken from internal
ground and VCC planes which were actually split and connected
with VIAS to the power device tabs. The TO-220 and TO-263
cases are well suited for this application, and are the preferred
packages. Remember to remove any conformal coating from all
exposed PC traces which are involved in heatsinking.
FIGURE 2 — Enabling Linear Regulator
LX1681/1682
®
TM
Voltage-Mode PWM Controllers
P RODUCTION D ATA S HEET
WWW . Microsemi .C OM
TYPICAL APPLICATION
V BOO ST
12V
V IN
5V
C3
C1
V FB
C SS
V C2
CS
SS
R SET
LX1681
0.1µF
R SENSE
V OUT
V C1
N.C.
GND
Q1
C2
L1
TDRV
R1
R2
D2
V IN
5V
C1
C SS
V FB
V C2
SS
CS
LX1682
0.1µF
GND
BDRV
V C1
D2
Q1
C2
L1
TDRV
Q2
R1
APPLICATION
R SET
Copyright © 2000
Rev. 1.1b,2005-03-09
V O UT
C4
R2
Microsemi
Linfinity Microelectronics Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 8
LX1681/1682
®
TM
Voltage-Mode PWM Controllers
P RODUCTION D ATA S HEET
R SET
I SE T
CS
7
+12V
CS Com p
I RES ET
V T R IP
6
PW M
+
V CC
I S ET
R
Q
S
Q
V C1
-
+
+
Am plifier/
Com pensation
V RE SE T
-
R1
TDRV
Error Com p
Set
V FB
20k
R2
C IN
5
320k
1
V IN (5V)
4
L
V CO RE
ESR
WWW . Microsemi .C OM
BLOCK DIAGRAM
C O UT
BDRV
V REF
3
GND
R SS
+5V
Hiccup Hiccup
Ramp
Oscillator
UVLO
8
UVLO
V CC
2
SS/ENABLE
C SS
PHYSICAL DIMENSIONS
DM
8-Pin Plastic SOIC
Dim
A
P
B
G
L
M
C
INCHES
MIN
MAX
0.190
0.197
0.150
0.155
0.053
0.069
0.013
0.020
0.030
0.050 BSC
0.007
0.010
0.005
0.010
0.189
0.205
8°
0.228
0.244
0.004
BLOCK DIAGRAM
A
B
C
D
F
G
J
K
L
M
P
*LC
MILLIMETERS
MIN
MAX
4.83
5.00
3.81
3.94
1.35
1.75
0.33
0.51
0.77
1.27 BSC
0.19
0.25
0.13
0.25
4.80
5.21
8°
5.79
6.20
0.10
*Lead Coplanarity
D
K
J
F
Note:
1. Dimensions do not include mold flash or protrusions;
these shall not exceed 0.155mm(.006”) on any side. Lead
dimension shall not include solder coverage.
Copyright © 2000
Rev. 1.1b,2005-03-09
Microsemi
Linfinity Microelectronics Division
11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570
Page 9
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