Fairchild ML4803IX-X 8-pin pfc and pwm controller combo Datasheet

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ML4803
8-Pin PFC and PWM Controller Combo
Features
General Description
• Internally synchronized PFC and PWM in one 8-pin IC
• Patented one-pin voltage error amplifier with advanced
input current shaping technique
• Peak or average current, continuous boost, leading edge
PFC (Input Current Shaping Technology)
• High efficiency trailing-edge current mode PWM
• Low supply currents; start-up: 150µA typ., operating:
2mA typ.
• Synchronized leading and trailing edge modulation
• Reduces ripple current in the storage capacitor between
the PFC and PWM sections
• Overvoltage, UVLO, and brownout protection
• PFC VCCOVP with PFC Soft Start
The ML4803 is a space-saving controller for power factor
corrected, switched mode power supplies that offers very
low start-up and operating currents.
Power Factor Correction (PFC) offers the use of smaller, lower
cost bulk capacitors, reduces power line loading and stress on
the switching FETs, and results in a power supply fully compliant to IEC1000-3-2 specifications. The ML4803 includes
circuits for the implementation of a leading edge, average
current “boost” type PFC and a trailing edge, PWM.
The ML4803-1’s PFC and PWM operate at the same
frequency, 67kHz. The PFC frequency of the ML4803-2 is
automatically set at half that of the 134kHz PWM. This
higher frequency allows the user to design with smaller
PWM components while maintaining the optimum operating
frequency for the PFC. An overvoltage comparator shuts
down the PFC section in the event of a sudden decrease in
load. The PFC section also includes peak current limiting for
enhanced system reliability.
Block Diagram
7
VEAO
7V
4
+
VCC
PFC OFF
COMP
–
17.5V
16.2V
35µA
REF
+
VREF
VCC OVP
GND
COMP
–
2
M3
–1
M4
M1
PFC
CONTROL
LOGIC
M2
R1
C1
30pF
PFC OUT
1
M7
LEADING
EDGE PFC
ONE PIN ERROR AMPLIFIER
3
ISENSE
+
–
VCC
OSCILLATOR
PFC – 67kHz
PWM – 134kHz
26k
–
DUTY CYCLE
LIMIT
PWM COMPARATOR
COMP
40k
+
1.2V
6
TRAILING
EDGE PWM
PFC/PWM UVLO
VREF
VDC
SOFT START
–
PFC ILIMIT
+
–1V
5
COMP
–4
ILIMIT
PWM
CONTROL
LOGIC
–
COMP
+
PWM OUT
8
M6
1.5V
–
DC ILIMIT
+
REV. 1.1.3 8/3/01
ML4803
PRODUCT SPECIFICATION
Pin Configuration
ML4803
8-Pin PDIP (P08)
8-Pin SOIC (S08)
PFC OUT
1
8
PWM OUT
GND
2
7
VCC
ISENSE
3
6
ILIMIT
VEAO
4
5
VDC
TOP VIEW
Pin Description
PIN
NAME
1
PFC OUT
FUNCTION
PFC driver output
2
GND
Ground
3
ISENSE
Current sense input to the PFC current limit comparator
4
VEAO
PFC one-pin error amplifier input
5
VDC
6
ILIMIT
7
VCC
8
PWM OUT
PWM voltage feedback input
PWM current limit comparator input
Positive supply (may require an external shunt regulator)
PWM driver output
Absolute Maximum Ratings
Absolute maximum ratings are those values beyond which the device could be permanently damaged. Absolute maximum
ratings are stress ratings only and functional device operation is not implied.
Parameter
Min
ICC Current (average)
VCC MAX
Max
Unit
40
mA
18.3
V
-5
1
V
GND – 0.3
VCC + 0.3
V
Peak PFC OUT Current, Source or Sink
1
A
Peak PWM OUT Current, Source or Sink
1
A
ISENSE Voltage
Voltage on Any Other Pin
PFC OUT, PWM OUT Energy Per Cycle
1.5
µJ
Junction Temperature
150
°C
150
°C
260
°C
110
160
°C/W
°C/W
Storage Temperature Range
-65°
Lead Temperature (Soldering, 10 sec)
Thermal Resistance (θJA)
Plastic DIP
Plastic SOIC
Operating Conditions
Temperature Range
ML4803CX-X
0°C to 70°C
ML4803IX-X
-40°C to 85°C
(Industrial temp ranges available 2002)
2
REV. 1.1.3 8/3/01
PRODUCT SPECIFICATION
ML4803
Electrical Characteristics
Unless otherwise specified, VCC = 15V, TA = Operating Temperature Range (Note 1)
Symbol
Parameter
Conditions
Min
Typ
Max
Units
VEAO Output Current
TA = 25°C, VEAO = 6V
33.5
35.0
36.5
µA
Line Regulation
10V < VCC < 15V, VEAO = 6V
0.1
0.3
µA
16.3
16.8
V
One-pin Error Amplifier
VCC OVP Comparator
Threshold Voltage
TA = 0°C to 70°C
15.5
PFC ILIMIT Comparator
Threshold Voltage
-0.9
Delay to Output
-1
-1.15
V
150
300
ns
1.5
1.6
V
150
300
ns
67
74
kHz
DC ILIMIT Comparator
Threshold Voltage
1.4
Delay to Output
Oscillator
Initial Accuracy
TA = 25°C
Voltage Stability
10V < VCC < 15V
62
Temperature Stability
1
%
2
%
Total Variation
Over Line and Temp
60
67
74.5
kHz
Dead Time
PFC Only
0.3
0.45
0.65
µs
90
95
PFC
Minimum Duty Cycle
VEAO > 7.0V,ISENSE = -0.2V
Maximum Duty Cycle
VEAO < 4.0V,ISENSE = 0V
0
8
15
Ω
IOUT = –100mA
0.8
1.5
V
IOUT = –10mA, VCC = 8V
0.7
1.5
V
8
15
Ω
Output Low Impedance
Output Low Voltage
%
%
Output High Impedance
Output High Voltage
IOUT = 100mA, VCC = 15V
Rise/Fall Time
CL = 1000pF
Duty Cycle Range
TA = 0°C to 70°C, ML4803-2
0-43
TA = 0°C to 70°C, ML4803-1
0-49.5
13.5
14.2
V
50
ns
PWM
Output Low Impedance
Output Low Voltage
0-47
8
0-50
%
0-50
%
15
Ω
IOUT = –100mA
0.8
1.5
V
IOUT = –10mA, VCC = 8V
0.7
1.5
V
8
15
Ω
Output High Impedance
Output High Voltage
IOUT = 100mA, VCC = 15V
Rise/Fall Time
CL = 1000pF
VCC Clamp Voltage (VCCZ)
ICC = 10mA
Start-up Current
Operating Current
13.5
14.2
V
50
ns
Supply
16.7
17.5
18.3
V
VCC = 11V, CL = 0
0.2
0.4
mA
VCC = 15V, CL = 0
2.5
4
mA
Undervoltage Lockout Threshold
11.5
12
12.5
V
Undervoltage Lockout Hysteresis
2.4
2.9
3.4
V
Note:
1. Limits are guaranteed by 100% testing, sampling, or correlation with worst case test conditions.
REV. 1.1.3 8/3/01
3
ML4803
Functional Description
The ML4803 consists of an average current mode boost
Power Factor Corrector (PFC) front end followed by a synchronized Pulse Width Modulation (PWM) controller. It is
distinguished from earlier combo controllers by its low pin
count, innovative input current shaping technique, and very
low start-up and operating currents. The PWM section is
dedicated to peak current mode operation. It uses conventional trailing-edge modulation, while the PFC uses leadingedge modulation. This patented Leading Edge/Trailing Edge
(LETE) modulation technique helps to minimize ripple current in the PFC DC buss capacitor.
The ML4803 is offered in two versions. The ML4803-1
operates both PFC and PWM sections at 67kHz, while the
ML4803-2 operates the PWM section at twice the frequency
(134kHz) of the PFC. This allows the use of smaller PWM
magnetics and output filter components, while minimizing
switching losses in the PFC stage.
In addition to power factor correction, several protection features have been built into the ML4803. These include soft
start, redundant PFC over-voltage protection, peak current
limiting, duty cycle limit, and under voltage lockout
(UVLO). See Figure 12 for a typical application.
Detailed Pin Descriptions
VEAO
This pin provides the feedback path which forces the PFC
output to regulate at the programmed value. It connects to
programming resistors tied to the PFC output voltage and is
shunted by the feedback compensation network.
ISENSE
This pin ties to a resistor or current sense transformer which
senses the PFC input current. This signal should be negative
with respect to the IC ground. It internally feeds the pulseby-pulse current limit comparator and the current sense feedback signal. The ILIMIT trip level is –1V. The ISENSE feedback is internally multiplied by a gain of four and compared
against the internal programmed ramp to set the PFC duty
cycle. The intersection of the boost inductor current
downslope with the internal programming ramp determines
the boost off-time.
VDC
This pin is typically tied to the feedback opto-collector. It is
tied to the internal 5V reference through a 26kΩ resistor and
to GND through a 40kΩ resistor.
ILIMIT
This pin is tied to the primary side PWM current sense resistor or transformer. It provides the internal pulse-by-pulse
current limit for the PWM stage (which occurs at 1.5V) and
the peak current mode feedback path for the current mode
4
PRODUCT SPECIFICATION
control of the PWM stage. The current ramp is offset internally by 1.2V and then compared against the opto feedback
voltage to set the PWM duty cycle.
PFC OUT and PWM OUT
PFC OUT and PWM OUT are the high-current power drivers capable of directly driving the gate of a power MOSFET
with peak currents up to ±1A. Both outputs are actively held
low when VCC is below the UVLO threshold level.
VCC
VCC is the power input connection to the IC. The VCC startup current is 150µA . The no-load ICC current is 2mA. VCC
quiescent current will include both the IC biasing currents
and the PFC and PWM output currents. Given the operating
frequency and the MOSFET gate charge (Qg), average
PFC and PWM output currents can be calculated as IOUT =
Qg x F. The average magnetizing current required for any
gate drive transformers must also be included. The VCC pin
is also assumed to be proportional to the PFC output voltage.
Internally it is tied to the VCCOVP comparator (16.2V)
providing redundant high-speed over-voltage protection
(OVP) of the PFC stage. VCC also ties internally to the
UVLO circuitry, enabling the IC at 12V and disabling it at
9.1V. VCC must be bypassed with a high quality ceramic
bypass capacitor placed as close as possible to the IC.
Good bypassing is critical to the proper operation of the
ML4803.
VCC is typically produced by an additional winding off the
boost inductor or PFC Choke, providing a voltage that is proportional to the PFC output voltage. Since the VCCOVP max
voltage is 16.2V, an internal shunt limits VCC overvoltage to
an acceptable value. An external clamp, such as shown in
Figure 1, is desirable but not necessary.
VCC
1N4148
1N4148
1N5246B
GND
Figure 1. Optional VCC Clamp
VCC is internally clamped to 16.7V minimum, 18.3V maximum. This limits the maximum VCC that can be applied to
the IC while allowing a VCC which is high enough to trip the
VCCOVP. The max current through this zener is 10mA.
External series resistance is required in order to limit the
current through this Zener in the case where the VCC voltage
exceeds the zener clamp level.
REV. 1.1.3 8/3/01
PRODUCT SPECIFICATION
ML4803
GND
verter is the full wave rectified AC line voltage. No filtering
is applied following the bridge rectifier, so the input
voltage to the boost converter ranges, at twice line frequency,
from zero volts to the peak value of the AC input and back to
zero. By forcing the boost converter to meet two simultaneous conditions, it is possible to ensure that the current that
the converter draws from the power line matches the instantaneous line voltage. One of these conditions is that the
output voltage of the boost converter must be set higher than
the peak value of the line voltage. A commonly used value is
385VDC, to allow for a high line of 270VACRMS. The other
condition is that the current that the converter is allowed to
draw from the line at any given instant must be proportional
to the line voltage.
GND is the return point for all circuits associated with
this part. Note: a high-quality, low impedance ground is
critical to the proper operation of the IC. High frequency
grounding techniques should be used.
Power Factor Correction
Power factor correction makes a nonlinear load look like a
resistive load to the AC line. For a resistor, the current drawn
from the line is in phase with, and proportional to, the line
voltage. This is defined as a unity power factor is (one). A
common class of nonlinear load is the input of a most power
supplies, which use a bridge rectifier and capacitive input filter fed from the line. Peak-charging effect, which occurs on
the input filter capacitor in such a supply, causes brief highamplitude pulses of current to flow from the power line,
rather than a sinusoidal current in phase with the line voltage. Such a supply presents a power factor to the line of less
than one (another way to state this is that it causes significant
current harmonics to appear at its input). If the input current
drawn by such a supply (or any other nonlinear load) can be
made to follow the input voltage in instantaneous amplitude,
it will appear resistive to the AC line and a unity power factor
will be achieved.
Since the boost converter topology in the ML4803 PFC is of
the current-averaging type, no slope compensation is
required.
Leading/Trailing Modulation
Conventional Pulse Width Modulation (PWM) techniques
employ trailing edge modulation in which the switch will
turn ON right after the trailing edge of the system clock.
The error amplifier output voltage is then compared with the
modulating ramp. When the modulating ramp reaches the
level of the error amplifier output voltage, the switch will be
turned OFF. When the switch is ON, the inductor current will
ramp up. The effective duty cycle of the trailing edge modulation is determined during the ON time of the switch. Figure
2 shows a typical trailing edge control scheme.
To hold the input current draw of a device drawing power
from the AC line in phase with, and proportional to, the input
voltage, a way must be found to prevent that device from
loading the line except in proportion to the instantaneous line
voltage. The PFC section of the ML4803 uses a boost-mode
DC-DC converter to accomplish this. The input to the con-
SW2
L1
I2
I1
+
I3
I4
VIN
RL
SW1
DC
C1
RAMP
VEAO
REF
U3
+
–EA
TIME
DFF
RAMP
OSC
U4
CLK
+
–
U1
R
Q
D U2
Q
CLK
VSW1
TIME
Figure 2. Typical Trailing Edge Control Scheme.
REV. 1.1.3 8/3/01
5
ML4803
PRODUCT SPECIFICATION
In the case of leading edge modulation, the switch is turned
OFF right at the leading edge of the system clock. When the
modulating ramp reaches the level of the error amplifier
output voltage, the switch will be turned ON. The effective
duty-cycle of the leading edge modulation is determined
during the OFF time of the switch. Figure 3 shows a leading
edge control scheme.
Programming Resistor Value
One of the advantages of this control technique is that it
requires only one system clock. Switch 1 (SW1) turns OFF
and Switch 2 (SW2) turns ON at the same instant to minimize the momentary “no-load” period, thus lowering ripple
voltage generated by the switching action. With such
synchronized switching, the ripple voltage of the first stage
is reduced. Calculation and evaluation have shown that the
120Hz component of the PFC’s output ripple voltage can be
reduced by as much as 30% using this method, substantially
reducing dissipation in the high-voltage PFC capacitor.
PFC Voltage Loop Compensation
Equation 1 calculates the required programming resistor
value.
Rp =
VBOOST − VEAO 400V − 50V
.
=
= 113
. MΩ
IPGM
35µA
The voltage-loop bandwidth must be set to less than 120Hz
to limit the amount of line current harmonic distortion.
A typical crossover frequency is 30Hz. Equation 1, for
simplicity, assumes that the pole capacitor dominates
the error amplifier gain at the loop unity-gain frequency.
Equation 2 places a pole at the crossover frequency,
providing 45 degrees of phase margin. Equation 3 places a
zero one decade prior to the pole. Bode plots showing the
overall gain and phase are shown in Figures 5 and 6. Figure 4
displays a simplified model of the voltage loop.
Typical Applications
CCOMP =
One Pin Error Amp
The ML4803 utilizes a one pin voltage error amplifier in the
PFC section (VEAO). The error amplifier is in reality a current sink which forces 35µA through the output programming resistor. The nominal voltage at the VEAO pin is 5V.
The VEAO voltage range is 4 to 6V. For a 11.3MΩ resistor
chain to the boost output voltage and 5V steady state at the
VEAO, the boost output voltage would be 400V.
SW2
L1
I2
I1
+
(1)
CCOMP =
Pin
R p × V BOOST × ∆VEAO × COUT × (2 × π × f)
2
(2)
300W
11.3MΩ × 400V × 0.5V × 220µF × (2 × π × 30Hz)2
CCOMP = 16nF
I3
I4
VIN
RL
SW1
DC
C1
RAMP
VEAO
U3
+
–EA
REF
RAMP
OSC
U4
CLK
VEAO
+
–
U1
TIME
DFF
CMP
R
Q
D U2
Q
CLK
VSW1
TIME
Figure 3. Typical Leading Edge Control Scheme.
6
REV. 1.1.3 8/3/01
PRODUCT SPECIFICATION
ML4803
Internal Voltage Ramp
2 × π × f × CCOMP
1
= 330kΩ
6.28 × 30Hz × 16nF
RCOMP =
CZERO =
The internal ramp current source is programmed by way of
the VEAO pin voltage. Figure 7 displays the internal ramp
current vs. the VEAO voltage. This current source is used to
develop the internal ramp by charging the internal 30pF +12/
–10% capacitor. See Figures 10 and 11. The frequency of the
internal programming ramp is set internally to 67kHz.
1
RCOMP =
1
f
× RCOMP
2×π×
10
PFC Current Sense Filtering
In DCM, the input current wave shaping technique used by
the ML4803 could cause the input current to run away.
In order for this technique to be able to operate properly
under DCM, the programming ramp must meet the boost
inductor current down-slope at zero amps. Assuming the
programming ramp is zero under light load, the OFF-time
will be terminated once the inductor current reaches zero.
1
= 016
. µF
CZERO =
6.28 × 3Hz × 330kΩ
60
VO
40
11.3MΩ
20
RLOAD
667Ω
330kΩ
IVEAO
35µA
15nF
–
GAIN (dB)
220µF
∆VEAO +
ML4803
VEAO
IOUT
ML4803
0
–20
0.15µF
POWER
STAGE
Power Stage
Overall Gain
Compensation
Network Gain
COMPENSATION
–40
Figure 4. Voltage Control Loop
–60
0.1
10
1
1000
100
FREQUENCY (Hz)
Figure 5. Voltage Loop Gain
50
0
Power Stage
Overall
Compensation
Network
FF @ –55ºC
TYP @ –55ºC
40
TYP @ ROOM TEMP
IRAMP (µA)
PHASE (°)
50
100
150
TYP @ 155ºC
30
SS @ 155ºC
20
10
0
200
0.1
1
10
100
FREQUENCY (Hz)
Figure 6. Voltage Loop Phase
REV. 1.1.3 8/3/01
1000
0
1
2
3
4
5
6
7
VEAO (V)
Figure 7. Internal Ramp Current vs. VEAO
7
ML4803
PRODUCT SPECIFICATION
Subsequently the PFC gate drive is initiated, eliminating the
necessary dead time needed for the DCM mode. This forces
the output to run away until the VCC OVP shuts down the
PFC. This situation is corrected by adding an offset voltage
to the current sense signal, which forces the duty cycle to
zero at light loads. This offset prevents the PFC from operating in the DCM and forces pulse-skipping from CCM to noduty, avoiding DMC operation. External filtering to the current sense signal helps to smooth out the sense signal,
expanding the operating range slightly into the DCM range,
but this should be done carefully, as this filtering also
reduces the bandwidth of the signal feeding the pulse-bypulse current limit signal. Figure 9 displays a typical circuit
for adding offset to ISENSE at light loads.
sequence. Once disabled, the VEAO pin charges HIGH by
way of the external components until the PFC duty cycle
goes to zero, disabling the PFC. The VCC OVP resets once
the VCC discharges below 16.2V, enabling the VEAO current
sink and discharging the VEAO compensation components
until the steady state operating point is reached. It should be
noted that, as shown in Figure 8, once the VEAO pin exceeds
6.5V, the internal ramp is defeated. Because of this, an external Zener can be installed to reduce the maximum voltage to
which the VEAO pin may rise in a shutdown condition.
Clamping the VEAO pin externally to 7.4V will reduce the
time required for the VEAO pin to recover to its steady state
value.
UVLO
PFC Start-Up and Soft Start
During steady state operation VEAO draws 35µA. At start-up
the internal current mirror which sinks this current is defeated
until VCC reaches 12V. This forces the PFC error voltage to
VCC at the time that the IC is enabled. With leading edge
modulation VCC on the VEAO pin forces zero duty on the
PFC output. When selecting external compensation components and VCC supply circuits VEAO must not be prevented
from reaching 6V prior to VCC reaching 12V in the turn-on
sequence. This will guarantee that the PFC stage will enter
soft-start. Once VCC reaches 12V the 35µA VEAO current
sink is enabled. VEAO compensation components are then
discharged by way of the 35µA current sink until the steady
state operating point is reached. See Figure 8.
PFC Soft Recovery Following VCC OVP
The ML4803 assumes that VCC is generated from a source
that is proportional to the PFC output voltage. Once that
source reaches 16.2V the internal current sink tied to the
VEAO pin is disabled just as in the soft start turn-on
Once VCC reaches 12V both the PFC and PWM are enabled.
The UVLO threshold is 9.1V providing 2.9V of hysteresis.
Generating VCC
An internal clamp limits overvoltage to VCC. This clamp
circuit ensures that the VCC OVP circuitry of the ML4803
will function properly over tolerance and temperature while
protecting the part from voltage transients. This circuit
allows the ML4803 to deliver 15V nominal gate drive at
PWM OUT and PFC OUT, sufficient to drive low-cost
IGBTs.
It is important to limit the current through the Zener to avoid
overheating or destroying it. This can be done with a single
resistor in series with the VCC pin, returned to a bias supply
of typically 14V to 18V. The resistor value must be chosen
to meet the operating current requirement of the ML4803
itself (4.0mA max) plus the current required by the two gate
driver outputs.
10V/div.
VCC
C23
0.01µF
0
10V/div.
VEAO
0
PFC
GATE
0
200V/div.
VBOOST
CR16
1N4148
C16
1µF
R4
1kΩ
R19
10kΩ
ISENSE
10V/div.
VOUT
R28
20kΩ
R29
20kΩ
R3
0.15Ω
3W
C5
0.0082µF
VCC
RTN
0
200ms/Div.
Figure 8. PFC Soft Start
8
Figure 9. ISENSE Offset for Light Load Conditions
REV. 1.1.3 8/3/01
PRODUCT SPECIFICATION
ML4803
VCC OVP
Component Reduction
VCC is assumed to be a voltage proportional to the PFC
output voltage, typically a bootstrap winding off the boost
inductor. The VCC OVP comparator senses when this voltage exceeds 16V, and terminates the PFC output drive while
disabling the VEAO current sink. Once the VEAO current
sink is disabled, the VEAO voltage will charge unabated,
except for a diode clamp to VCC, reducing the PFC pulse
width. Once the VCC rail has decreased to below 16.2V the
VEAO sink will be enabled, discharging external VEAO
compensation components until the steady state voltage is
reached. Given that 15V on VCC corresponds to 400V on
the PFC output, 16V on VCC corresponds to an OVP level of
426V.
Components associated with the VRMS and IRMS pins of a
typical PFC controller such as the ML4824 have been eliminated. The PFC power limit and bandwidth does vary with
line voltage. Double the power can be delivered from a 220
V AC line versus a 110 V AC line. Since this is a combination PFC/PWM, the power to the load is limited by the PWM
stage.
VISENSE
VC1 RAMP
GATE
DRIVE
OUTPUT
Figure 10. Typical Peak Current Mode Waveforms
VOUT = 400V
RP
VC1
VEAO
4
+
C1
30pF
RCOMP
CCOMP
35µA
GATE
OUTPUT
COMP
–
5V
R1
CZERO
3
ISENSE
–4
VI SENSE
Figure 11. ML4803 PFC Control
REV. 1.1.3 8/3/01
9
ML4803
PRODUCT SPECIFICATION
LINE F1 5A 250V
J1-1
R24
470kΩ
0.5W
C19
4.7nF
250VAC
102T
L2
TH1
10Ω
5A
C4
0.47µF
250VAC
Q5
R1
BR1
600V
4A
36Ω
Q2
NEUTRAL
J1-2
CR1 8A, 600V
1000µH
R2
L3
C20
4.7nF
250VAC
C1
220µF
450V
36Ω
R22
10kΩ
C16
0.01µF
CR5
16V
0.5W
R3 0.15Ω 3W
Q4
CR18 51V
R8 36Ω
CR7
CR3
R30 200Ω
T2
R13
5.8MΩ
3
10
4
C3
1µF
R14
150Ω
2W
J2-2
L1 25µH
R26
20kΩ
3W
C26
0.01µF
500V
CR8
R29 20kΩ
R28
20kΩ
1
2
3
CR12
4
C8
0.15µF
CR2
30A, 60V
12VRET
R4 1kΩ
L2
C9
1µF
C22
1µF
R19
10kΩ
ML4803
PFC
PWM
GND
VCC
ISENSE ILIMIT
VEAO
VDC
8
CR10
C28
1µF
C21
1µF
CR15
R6 1.2kΩ
5
6
CR11
C27
0.01µF
U3
1
CR9
R21
10kΩ
R37
330Ω
C17
0.1µF
2
R32 100Ω
5
C14
4.7µF
R9
1.5kΩ
Q3
R5 36Ω
R11 150Ω
C10
2.2nF
C6
1µF
4T
7
C5
8.2nF
C15
0.015µF
J2-1
C2
2200µF
R31
10Ω
7.0V
12V
CR2
30A
60V
R27
20kΩ
3W
C7
0.1µF
CR16
IN4148
R25
390kΩ
C25
0.01µF
500V
R36 220Ω
T1
C11
1000µF
R12
5.8MΩ
C18 4.7nF
R23
10kΩ
C29 0.01µF
R7
10Ω
C23
0.01µF
Q1
R38 22Ω
1
4
R10
0.75Ω
3W
R15
9.09kΩ
C13 1nF
R17 3.3kΩ
CR4
R20
510Ω
3
1
U2
C12 0.1µF
2
R18 1kΩ
R16
2.37kΩ
Figure 12. Typical Application Circuit. Universal Input 240W 12V DC Output
10
REV. 1.1.3 8/3/01
PRODUCT SPECIFICATION
ML4803
Mechanical Dimensions inches (millimeters)
Package: P08
8-Pin PDIP
0.365 - 0.385
(9.27 - 9.77)
0.055 - 0.065
(1.39 - 1.65)
8
0.240 - 0.260
(6.09 - 6.60)
PIN1 ID
0.020 MIN
(0.51 MIN)
(4 PLACES)
0.299 - 0.335
(7.59 - 8.50)
1
0.100 BSC
(2.54 BSC)
0.015 MIN
(0.38 MIN)
0.170 MAX
(4.32 MAX)
0.125 MIN
(3.18 MIN)
0.016 - 0.020
(0.40 - 0.51)
0° - 15°
0.008 - 0.012
(0.20 - 0.31)
SEATING PLANE
Package: S08
8 Pin SOIC
0.189 - 0.199
(4.80 - 5.06)
8
PIN 1 ID
0.148 - 0.158 0.228 - 0.244
(3.76 - 4.01) (5.79 - 6.20)
1
0.017 - 0.027
(0.43 - 0.69)
(4 PLACES)
0.050 BSC
(1.27 BSC)
0.059 - 0.069
(1.49 - 1.75)
0°–8°
0.055 - 0.061
(1.40 - 1.55)
0.012 - 0.020
(0.30 - 0.51)
0.004 - 0.010
(0.10 - 0.26)
0.015 - 0.035
(0.38 - 0.89)
0.006 - 0.010
(0.15 - 0.26)
SEATING PLANE
REV. 1.1.3 8/3/01
11
ML4803
PRODUCT SPECIFICATION
Ordering Information
Part Number
PFC/PWM Frequency
Temperature Range
Package
ML4803CP-1
67kHz / 67kHz
0°C to 70°C
8-Pin PDIP (P08)
ML4803CS-1
67kHz / 67kHz
0°C to 70°C
8-Pin SOIC (S08)
ML4803IP-1*
67kHz / 67kHz
-40°C to 85°C
8-Pin PDIP (P08)
ML4803IS-1*
67kHz / 67kHz
-40°C to 85°C
8-Pin SOIC (S08)
ML4803CP-2
67kHz / 134kHz
0°C to 70°C
8-Pin PDIP (P08)
ML4803CS-2
67kHz / 134kHz
0°C to 70°C
8-Pin SOIC (S08)
ML4803IP-2*
67kHz / 134kHz
-40°C to 85°C
8-Pin PDIP (P08)
ML4803IS-2*
67kHz / 134kHz
-40°C to 85°C
8-Pin SOIC (S08)
*Available 2002
DISCLAIMER
FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO
ANY PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION OR DESIGN. FAIRCHILD DOES NOT ASSUME
ANY LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN;
NEITHER DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS.
LIFE SUPPORT POLICY
FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES
OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR
CORPORATION. As used herein:
1. Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body,
or (b) support or sustain life, and (c) whose failure to
perform when properly used in accordance with
instructions for use provided in the labeling, can be
reasonably expected to result in a significant injury of the
user.
2. A critical component in any component of a life support
device or system whose failure to perform can be
reasonably expected to cause the failure of the life support
device or system, or to affect its safety or effectiveness.
www.fairchildsemi.com
8/3/01 0.0m 003
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2001 Fairchild Semiconductor Corporation
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