TI1 OPA689M Gain 4 stable wideband voltage-limiting amplifier Datasheet

OPA689M
GAIN +4 STABLE WIDEBAND
VOLTAGE-LIMITING AMPLIFIER
www.ti.com
SGLS146B – MARCH 2003 – REVISED DECEMBER 2006
FEATURES
APPLICATIONS
•
•
•
•
•
•
•
•
•
High Linearity Near Limiting
Fast Recovery From Overdrive: 2.4 ns
Limiting Voltage Accuracy: ±15 mV
–3-dB Bandwidth (G = 6): 260 MHz
Stable for G ≥ 4
Slew Rate: 1400 V/µs
±5-V and 5-V Supply Operation
High-Gain Version of OPA688
•
•
•
•
•
•
Transimpedance With Fast Overdrive
Recovery
Fast Limiting ADC Input Buffers
Low Propagation Delay Comparator
Non-Linear Analog Signal Processing
Difference Amplifier
IF Limiting Amplifier
AM Signal Generation
JD PACKAGE
(TOP VIEW)
NC
INVERTING INPUT
NON-INVERTING INPUT
−VCC
1
8
2
7
3
6
4
5
VH
+VCC
OUTPUT
VL
NC - No internal connection
DESCRIPTION/ORDERING INFORMATION
The OPA689 is a wideband, voltage-feedback operational amplifier that offers bipolar output voltage limiting, and
is stable for gains ≥4. Two buffered limiting voltages take control of the output when it attempts to drive beyond
these limits. This new output limiting architecture holds the limiter offset error to ±15 mV. The operational
amplifier operates linearly to within 30 mV of the limits.
The combination of narrow nonlinear range and low limiting offset allows the limiting voltages to be set within
100 mV of the desired linear output range. A fast 2.4-ns recovery from limiting ensures that overdrive signals are
transparent to the signal channel. Implementing the limiting function at the output, as opposed to the input, gives
the specified limiting accuracy for any gain, and allows the OPA689 to be used in all standard operational
amplifier applications.
Nonlinear analog signal processing circuits benefit from the OPA689's sharp transition from linear operation to
output limiting. The quick recovery time supports high-speed applications.
The OPA689M is available in an industry-standard pinout in a CDIP-8 package. For lower gain applications
requiring output limiting with fast recovery, consider the OPA688M.
ORDERING INFORMATION
PACKAGE (1)
TA
–55°C to 125°C
(1)
CDIP – JD
Tube
ORDERABLE PART NUMBER
OPA689MJD
TOP-SIDE MARKING
OPA689MJD
Package drawings, standard packing quantities, thermal data, symbolization, and PCB design guidelines are available at
www.ti.com/sc/package.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2003–2006, Texas Instruments Incorporated
OPA689M
GAIN +4 STABLE WIDEBAND
VOLTAGE-LIMITING AMPLIFIER
www.ti.com
SGLS146B – MARCH 2003 – REVISED DECEMBER 2006
D E T A IL O F L IM IT E D O U T P U T V O LT AG E
L IM IT E D O U T P U T R E S P O N S E
2 .1 0
2 .5
V H = − V L = 2 .0 V
G = +2
2 .0 5
2 .0 0
1 .5
V IN
1 .0
O u tp u t V o lta g e ( V)
In pu t a nd O u tp ut V oltag e (V )
2 .0
V O
0 .5
0
− 0 .5
− 1 .0
1 .9 5
VO
1 .9 0
1 .8 5
1 .8 0
1 .7 5
− 1 .5
1 .7 0
− 2 .0
1 .6 5
1 .6 0
− 2 .5
T im e ( 5 0 n s / d iv )
T im e ( 2 0 0 n s / d iv )
Absolute Maximum Ratings (1)
over operating free-air temperature (unless otherwise noted)
Power supply
OPA698M
UNIT
–6.5 to 6.5
V
VIC
Common-mode input voltage
–VCC to VCC
V
VID
Differential input voltage
–VCC to VCC
V
–(VS – 0.7 V) to (VS – 0.7 V)
V
TA
Limiter voltage range
Operating free-air temperature range
–55 to 125
°C
Tstg
Storage temperature range
–65 to 150
°C
Lead temperature 1,6 mm (1/16 in) from case for 10 s
300
°C
Case temperature for 10 s
260
°C
Junction temperature
150
°C
TJ
(1)
Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
Recommended Operating Conditions
MIN
Operating voltage
Split-rail operation
Single-supply operation
Operating free-air temperature
2
–55
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NOM
MAX
±5
±6
5
12
125
UNIT
V
°C
OPA689M
GAIN +4 STABLE WIDEBAND
VOLTAGE-LIMITING AMPLIFIER
www.ti.com
SGLS146B – MARCH 2003 – REVISED DECEMBER 2006
Electrical Characteristics
(1)
VCC = ±5 V, VICM = 0 V, RL = 500 Ω, limiter pins open (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
AC Performance (See Figure 1)
G=6
Small signal bandwidth
VO < 0.5 Vp-p,
Gain bandwidth product (G ≥ 20)
VO < 0.5 Vp-p
Gain peaking
VO = 0.5 V, G = 4
Bandwidth for 0.1-dB gain flatness
VO = 0.5 V
Large signal bandwidth
VO = 2 Vp-p
Slew rate
VO = 2 V step
Rise and fall time
VO = 0.5 V step
Settling time to 0.05%
VO = 2 V step
Spurious free dynamic range
VO = 2 Vp-p, f = 5 MHz
Differential gain
Differential phase
Input noise, voltage noise density
Input noise, current noise density
260
G = 12
86
G = –6
220
MHz
720
MHz
8
dB
30
MHz
290
MHz
1400
V/µs
1.6
ns
8
ns
61
dB
RL = 500 Ω, NTSC, PAL
0.02
%
RL = 500 Ω, NTSC, PAL
0.01
°
f ≥ 1 MHz
4.6
nV/√Hz
f ≥ 1 MHz
2
pA/√Hz
DC Performance
AVOL
Open-loop voltage gain
VIO
Input offset voltage
IIB
Input bias current (2)
IIO
Input offset current
VO = ±0.5 V
TA = 25°C
50
TA = Full range
47
56
±1
TA = 25°C
dB
±7
±10
TA = Full range
±8
TA = 25°C
±12
±20
TA = Full range
±0.3
TA = 25°C
±2
±4
TA = Full range
mV
µA
µA
Input
CMRR Common-mode rejection ratio
VICR
Common-mode input voltage
range (3)
VICM = ±0.5 V,
Input referred
TA = 25°C
53
TA = Full range
50
TA = 25°C
±3.2
TA = Full range
±3.1
Input impedance, differential mode
Input impedance, common mode
60
±3.3
dB
V
0.4
1
MΩ
pF
1
1
MΩ
pF
Output
VOH,
VOL
Output voltage range
VH = 4.3 V, VL = –4.3 V,
RL ≥ 500 Ω
TA = 25°C
±3.9
TA = Full range
±3.7
IOH
Current output, sourcing
VH = 4.3 V, VL = –4.3 V,
RL ≥ 20 Ω
TA = 25°C
90
TA = Full range
80
IOL
Current output, sinking
VH = 4.3 V, VL = –4.3 V,
RL ≥ 20 Ω
TA = 25°C
–70
TA = Full range
–60
Closed-loop output impedance
G = 4, f < 100 kHz
(1)
(2)
(3)
±4.1
105
–85
0.2
V
mA
mA
Ω
All typical limits are at TA = 25°C (unless otherwise specified).
Current is considered positive out of node.
CMIR tested as <3-dB degradation from minimum CMRR at specified limits.
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VOLTAGE-LIMITING AMPLIFIER
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SGLS146B – MARCH 2003 – REVISED DECEMBER 2006
Electrical Characteristics (continued)
VCC = ±5 V, VICM = 0 V, RL = 500 Ω, limiter pins open (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
±5
±6
TA = 25°C
14
15.8
17
TA = Full range
11
UNIT
Power Supply
VCC
Operating voltage
ICC
Quiescent current
PSRR
Power-supply rejection ratio
Input referred,
VS = ±4.5 V to ±5.5 V
TA = 25°C
58
TA = Full range
55
20
70
V
mA
dB
Output Voltage Limiters (Pins 5 and 8)
Default output limited voltage
Limiter pins open
Limiter output offset voltage
(VO – VH) or (VO – VL)
Limiter input bias current
magnitude (4)
VO = 0 V
TA = 25°C
TA = Full range
±3
TA = Full range
TA = 25°C
35
TA = Full range
31
±15
±50
54
65
70
f = 5 MHz
400
Operational amplifier bias current
shift (6)
Limter slew
4
µA
450
MHz
100
V/µs
Limiter step response, overshoot
VI = ±2 V
250
mV
Limiter step response, recovery time
VI = ±2 V
2.4
ns
VO = 2 Vp-p, f = 5 MHz
30
mV
Linearity
(4)
(5)
(6)
(7)
(8)
VI = ±2 V, VO < 0.02 Vp-p
V
mV
3
rate (7)
µA
dB
±4.3
Minimum limiter voltage separation
mV
MΩ
pF
–60
Maximum limiter voltage
Limiter small signal bandwidth
V
2
1
Limiter input impedance
Limiter feedthrough (5)
±3.3
±2.8
guardband (8)
IVH (VH bias current) is positive and IVL (VL bias current) is negative under these conditions (see Note 3, Figure 30, and Figure 37).
Limiter feedthrough is the ratio of the output magnitude to the sinewave added to VH (or VL) when VIN = 0.
Current is considered positive out of node.
VH slew rate conditions are VIN = 0.7 V, G = 6, VL = –2 V, VH = step between 2 V and 0 V. VL slew rate conditions are similar.
Linearity guardband is defined for an output sinusoid (f = 1 MHz, VO = 2 Vp-p) centered between the limiter levels (VH and VL). It is the
difference between the limiter level and the peak output voltage where SFDR decreases by 3 dB (see Figure 38).
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OPA689M
GAIN +4 STABLE WIDEBAND
VOLTAGE-LIMITING AMPLIFIER
www.ti.com
SGLS146B – MARCH 2003 – REVISED DECEMBER 2006
Electrical Characteristics
(1)
VCC = 5 V, VICM = 2.5 V, RL = 500 Ω, limiter pins open (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
AC Performance (See Figure 2)
G=6
Small signal bandwidth
VO < 0.5 Vp-p
Gain bandwidth product
VO < 0.5 Vp-p, G ≥ 20
Gain peaking
VO = 0.5 Vp-p, G = 4
Bandwidth for 0.1-dB gain flatness
VO = 0.5 Vp-p
Large signal bandwidth
VO = 2 Vp-p
Slew rate
VO = 2-V step
Rise and fall time
VO = 0.5-V step
Settling time to 0.05%
2-V step
Spurious free dynamic range
210
G = 12
70
G = –6
180
MHz
440
MHz
8
dB
30
MHz
175
MHz
1050
V/µs
1.9
ns
7
ns
VO = 2 Vp-p, f = 5 MHz
59
dB
Input noise, voltage noise density
f ≥ 1 MHz
4.6
nV/√Hz
Input noise, current noise density
f ≥ 1 MHz
2
pA/√Hz
DC Performance
AVOL
Open-loop voltage gain
VIO
Input offset voltage
IIB
Input bias current
IIO
Input offset current
VO = ±0.4 V
TA = 25°C
50
TA = Full range
47
56
±1
TA = 25°C
dB
±7
±10
TA = Full range
±8
TA = 25°C
±12
±20
TA = Full range
±0.3
TA = 25°C
±2
±4
TA = Full range
mV
µA
µA
Input
CMRR Common-mode rejection ratio
VICR
Common-mode input voltage
range (2)
VICM = ±0.5 V,
Input referred
TA = 25°C
51
TA = Full range
48
TA = 25°C
VICM
±0.7
TA = Full range
VICM
±0.6
Input impedance, differential mode
Input impedance, common mode
58
VICM
± 0.8
dB
V
0.4
1
MΩ
pF
1
1
MΩ
pF
Output
VH = VICM + 1.8 V,
VL = VICM – 1.8 V,
RL ≥ 500 Ω
VOH,
VOL
Output voltage range
IOH
Current output, sourcing
VCC = ±2.5 V, RL ≥ 20 Ω
IOL
Current output, sinking
VCC = ±2.5 V, RL ≥ 20 Ω
Closed-loop output impedance
G = 4, f < 100 kHz
(1)
(2)
TA = 25°C
VICM
±1.4
TA = Full range
VICM
±1.3
TA = 25°C
60
TA = Full range
50
TA = 25°C
–50
TA = Full range
–40
VICM
±1.6
70
–60
0.8
V
mA
mA
Ω
All typical limits are at TA = 25°C (unless otherwise specified).
CMIR tested as <3-dB degradation from minimum CMRR at specified limits.
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GAIN +4 STABLE WIDEBAND
VOLTAGE-LIMITING AMPLIFIER
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SGLS146B – MARCH 2003 – REVISED DECEMBER 2006
Electrical Characteristics (continued)
VCC = 5 V, VICM = 2.5 V, RL = 500 Ω, limiter pins open (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
5
12
11
13
15
UNIT
Power Supply
VCC
Operating voltage
ICC
Quiescent current
PSRR
Power-supply rejection ratio
TA = 25°C
TA = Full range
Input referred,
VCC = ±2 V to ±3 V
9
TA = Full range
16.5
55
70
TA = 25°C
VICM±
0.6
VICM±
0.9
TA = Full range
VICM±
0.4
V
mA
dB
Output Voltage Limiters (Pins 5 and 8)
Default output limited voltage
Limiter output offset voltage
Limiter input bias current
magnitude (3)
Limiter pins open
(VO– VH) or (VO– VL)
VO = 2.5 V
TA = Full range
TA = 25°C
0
TA = Full range
0
Limiter input bias current drift
Limiter input impedance
±15
±50
35
65
85
nA/°C
MΩ
pF
f = 5 MHz
–60
VI = ±2 V, limit mode
±15
400
VI = VICM± 0.4 V, VO < 0.02 Vp-p
V
µA
300
MHz
20
V/µs
Limiter step response, overshoot
VI = VICM± 0.4 V
55
mV
Limiter step response, recovery time
VI = VICM± 0.4 V
15
ns
VO = 2 Vp-p, f = 5 MHz
30
mV
Linearity
6
mV
mV
5
Limter slew rate (6)
(3)
(4)
(5)
(6)
(7)
dB
±40
VICM±
1.8 V
Maximum limiter voltage
Output bias current shift (5)
µA
2
1
Limiter offset
Minimum limiter voltage separation
mV
30
Limiter feedthrough (4)
Limiter small signal bandwidth
V
guardband (7)
IVH (VH bias current) is positive and IVL (VL bias current) is negative under these conditions (see Note 3, Figure 31, and Figure 37).
Limiter feedthrough is the ratio of the output magnitude to the sinewave added to VH (or VL) when VIN = 0.
Current is considered positive out of node.
VH slew rate conditions are VIN = 0.7 V, G = 6, VL = –2 V, VH = step between 2 V and 0 V. VL slew rate conditions are similar.
Linearity guardband is defined for an output sinusoid (f = 1 MHz, VO = 2 Vp-p) centered between the limiter levels (VH and VL). It is the
difference between the limiter level and the peak output voltage where SFDR decreases by 3 dB (see Figure 38).
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OPA689M
GAIN +4 STABLE WIDEBAND
VOLTAGE-LIMITING AMPLIFIER
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SGLS146B – MARCH 2003 – REVISED DECEMBER 2006
TYPICAL CHARACTERISTICS
NON-INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
INVERTING SMALL-SIGNAL
FREQUENCY RESPONSE
9
3
0
0
−3
−3
−6
−9
G = +12
−12
−15
G = +20
G = −6
−6
−9
G = −12
−12
−15
−18
−18
−21
−21
−24
1M
10M
100M
1G
1M
Figure 1.
Figure 2.
LARGE−SIGNAL PULSE RESPONSE
V O = 0.5Vp−p
0.3
1.5
0.2
1.0
0.1
0
−0.1
−0.2
V O = 2Vp−p
2.0
Output Voltage (V)
Output Voltage (V)
1G
2.5
0.4
0.5
0
−0.5
−1.0
−0.3
−1.5
−0.4
−2.0
−2.5
−0.5
Time (5ns/div)
Time (5ns/div)
Figure 3.
Figure 4.
V L − LIMITED PULSE RESPONSE
V H − LIMITED PULSE RESPONSE
2.5
2.5
V O
2.0
2.0
1.5
1.5
V IN
0.5
0
−0.5
−1.0
V H +2 V
G = +6
Input and Output Voltages (V)
Input and Output Voltages (V)
100M
Frequency (Hz)
SMALL−SIGNAL PULSE RESPONSE
−1.5
10M
Frequency (Hz)
0.5
1.0
G = −4
V O = 0.5Vp−p
3
G = +6
Normalized Gain (dB)
Normalized Gain (dB)
G = +4
V O = 0.5Vp−p
6
6
1.0
V L −2 V
G = +6
0.5
0
−0.5
−1.0
V IN
−1.5
−2.0
−2.0
−2.5
−2.5
V O
Time (20ns/div)
Time (20ns/div)
Figure 5.
Figure 6.
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TYPICAL CHARACTERISTICS (continued)
HARMONIC DISTORTION NEAR LIMIT VOLTAGES
HARMONIC DISTORTION vs FREQUENCY
−40
−40
V O = 2Vp-p
R L = 500 Ω
−50
V O = 0 V DC ± 1 Vp
f 1 = 5 MHz
R L = 500 Ω
−45
2nd and 3rd Harmonic Distortion (dBc)
2nd and 3rd Harmonic Distortion (dBc)
−45
HD2
−55
−60
−65
−70
HD3
−75
−80
−80
−50
−55
HD2
−60
−65
−70
−75
−80
HD3
−80
−90
−90
1M
10M
0.9
20M
1.1
1.2
1.3
Figure 8.
1.6
1.7
1.8
1.9
2.0
3RD HARMONIC DISTORTION vs OUTPUT SWING
−40
R L = 500 Ω
−50
f 1 = 10MHz
−55
−60
−65
f 1 = 5MHz
−70
f 1 = 2MHz
−75
−80
R L = 500 Ω
−45
f 1 = 20MHz
−50
3rd Harmonic Distortion (dBc)
−45
f 1 = 1MHz
f 1 = 20MHz
−55
f 1 = 10MHz
−60
−65
f 1 = 5MHz
−70
f 1 = 2MHz
−75
−80
f 1 = 1MHz
−80
−80
−90
−90
0.1
1.0
0.1
5.0
1.0
5.0
Output Swing (Vp−p)
Output Swing (Vp−p)
Figure 9.
Figure 10.
HARMONIC DISTORTION vs LOAD RESISTANCE
LARGE SIGNAL FREQUENCY RESPONSE
−40
21.6
V O = 2 Vp-p
f 1 = 5 MHz
−45
−50
2Vp−p
15.6
HD2
−55
12.6
−60
9.6
−65
G = +6
18.6
Gain (dB)
2nd and 3rd Harmonic Distortion (dBc)
1.5
Figure 7.
2ND HARMONIC DISTORTION vs OUTPUT SWING
HD3
−70
≤ 0.5Vp−p
6.6
3.6
−75
0.6
−80
−2.4
−85
−5.4
−90
−8.4
50
100
1000
0.1
Load Resistance ( Ω )
10M
100M
Frequency (Hz)
Figure 11.
8
1.4
± Limit Voltage (V)
−40
2nd Harmonic Distortion (dBc)
1.0
Frequency (Hz)
Figure 12.
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TYPICAL CHARACTERISTICS (continued)
R S vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
21.6
45
18.6
40
15.6
Gain to Capacitive Load (dB)
50
R S (Ω )
35
30
25
20
15
10
V O = 0.5Vp−p
9.6
6.6
R
IN
1k Ω
C
O
L
−2.4
150 Ω
−8.4
1000
1k Ω is optional
0.1
10M
100M
1G
Frequency (Hz)
Figure 13.
Figure 14.
INPUT NOISE DENSITY
OPEN−LOOP FREQUENCY RESPONSE
60
0
50
100
−60
Phase
30
−90
20
−120
V O = 0.5 Vp-p
10
−150
0
−180
−10
−210
−20
−240
10k
100k
1M
10M
100M
Open−Loop Phase (deg)
40
Input Voltage Noise Density (nV/√Hz)
Input Current Noise Density (pA/ √Hz)
−30
Gain
Voltage Noise
10
4.6 nV/√ Hz
Current Noise
2.0 pA/ √ Hz
1
1G
100
Frequency (Hz)
Figure 15.
10k
100k
1M
10M
Figure 16.
LIMITER FEEDTHROUGH
LIMITER SMALL−SIGNAL FREQUENCY RESPONSE
−30
6
V O = 0.02Vp−p
3
−35
0
−40
−3
−45
−6
−9
1k
Frequency (Hz)
V
H
= 0.02Vp−p + 2.0V
Feedthrough (dB)
Open−Loop Gain (dB)
V
750 Ω
Capacitive Load (pF)
Limiter Gain (dB)
S
OPA689
0.6
0
100
125 Ω
V
3.6
−5.4
10
C L = 10 pF
C L = 100 pF
C L = 1000 pF
12.6
5
1
C L =0
DC
125 Ω
0.7V
8
DC
V
−12
−15
O
750 Ω
−50
V
H
= 0.02Vp−p + 2V
DC
125 Ω
−55
8
−60
V
−65
O
750 Ω
−70
−18
150 Ω
150 Ω
−75
−21
−80
−24
1M
10M
100M
1G
1M
10M
50M
Frequency (Hz)
Frequency (Hz)
Figure 17.
Figure 18.
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TYPICAL CHARACTERISTICS (continued)
L im te r In p u t B ia s C u r r e n t ( ∝
CLOSED−LOOP OUTPUT IMPEDANCE
100
Output Impedance ( Ω )
G = +4
V O = 0.5Vp−p
10
1
0.1
100k
1M
10M
100M
L IM IT E R IN P U T B IA S C U R R E N T v s B IA S V O LT A G E
100
M a x im u m O v e r T e m p e ra ture
75
50
25
M in im u m O ve r T e m p e ra tu r e
0
− 25
− 50
L im ite r H e a d r o o m = + V S − V H
= V L − ( − V S)
− 75
C u r r e n t = I VH o r − I VL
− 100
0 .0
1G
0 .5
1 .0
1 .5
2 .0
2 .5
3 .0
3 .5
4 .0
4 .5
5 .0
L im ite r H e a d r o o m ( V )
Frequency (Hz)
Figure 19.
Figure 20.
PSR AND CMR vs TEMPERATURE
SUPPLY AND OUTPUT CURRENTS vs TEMPERATURE
20
200
18
180
100
16
160
Supply Current
14
140
| Output Current, Sinking |
12
120
10
PSR and CMR, Input Referred (dB)
Output Current, Sourcing
Output Current (mA)
Supply Current (mA)
95
−25
0
25
50
75
PSR−
85
PSRR
80
75
PSR+
70
65
60
CMRR
55
50
100
−50
90
−50
100
−25
0
Figure 21.
50
75
100
Figure 22.
NON−INVERTING SMALL−SIGNAL
FREQUENCY RESPONSE
VOLTAGE RANGES vs TEMPERATURE
5.0
9
V H = −V L = 4.3 V
V O = 0.5Vp−p
6
4.5
3
Output Voltage Range
Normalized Gain (dB)
Voltage Range (V)
25
Ambient Temperature ( ° C)
Ambient Temperature (¡C)
4.0
3.5
0
G = +6
G = +4
−3
−6
G = +20
−9
G = +12
−12
−15
Common−Mode Input Range
−18
3.0
−50
−25
0
25
50
Ambient Temperature (°C)
75
100
−21
1M
10M
100M
Frequency (Hz)
Figure 23.
10
Figure 24.
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TYPICAL CHARACTERISTICS (continued)
LARGE−SIGNAL FREQUENCY RESPONSE
INVERTING SMALL−SIGNAL
21.6
FREQUENCY RESPONSE
6
18.6
V O = 0.5Vp−p
3
15.6
G = −4
≤ 0.5Vp−p
12.6
−3
G = −12
−6
G = −6
−9
6.6
3.6
−12
0.6
−15
−2.4
−18
−5.4
−21
−8.4
0.1
−24
1M
10M
100M
2Vp−p
9.6
Gain (dB)
Normalized Gain (dB)
0
10M
100M
1G
1G
Frequency (Hz)
Figure 25.
Figure 26.
V H A N D V H − L IM IT E D P U LS E R E S P O N S E
2nd and 3rd Harmonic Distortion (dBc)
V O
3 .5
3 .0
2 .5
V IN
V IN
2 .0
1 .5
V C M = 2 .5 V
VO
1 .0
V O = 2 Vp-p
R L = 500 Ω
−45
V L = V CM − 1 .2 V
0 .5
0
−50
HD2
−55
−60
HD3
−65
−70
−75
−80
−85
−90
1M
T im e ( 2 0 n s / d iv )
10M
20M
Frequency (Hz)
Figure 27.
Figure 28.
HARMONIC DISTORTION NEAR LIMIT VOLTAGES
−40
V O = 2.5V DC ± 1 Vp
f 1 = 5 MHz
R L = 500 Ω
−45
2nd and 3rd Harmonic Distortion (dBc)
In p u t a n d O u tp u t V o lta g e s ( V)
V H = V CM + 1 .2 V
4 .5
4 .0
HARMONIC DISTORTION vs FREQUENCY
−40
5 .0
−50
−55
HD2
−60
−65
−70
HD3
−75
−80
−85
−90
0.9
1.0
1.1
1.2
1.3
1.4
1.5
1.6
1.7
1.8
| Limit Voltages − 2.5 V DC |
Figure 29.
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APPLICATION INFORMATION
Dual-Supply, Non-Inverting Amplifier
Figure 30 shows a non-inverting gain amplifier for dual-supply operation. This circuit was used for AC
characterization of the OPA689, with a 50-Ω source, which it matches, and a 500-Ω load. The power-supply
bypass capacitors are shown explicitly in Figure 30 and Figure 31, but will be assumed in the other figures. The
limiter voltages (VH and VL) and their bias currents (IVH and IVL) have the polarities shown.
3.01kΩ
1.91kΩ
+V S = +5V
+
µ F
2.2 µ F 0.1
0.1 µ F
V H = +2V
Ω
174
3
7
V IN
8
Ω
49.9
R G
402 Ω
6
OPA689
2
R F
402 Ω
0.1 µ F
I VH
5
I VL
V O
500
Ω
4
0.1 µ F
V L = − 2V
2.2 µ F
3.01kΩ
1.91kΩ
+
− V S = − 5V
Figure 30. DC-Coupled, Dual Supply Amplifier
Single-Supply, Non-Inverting Amplifier
Figure 31 shows an AC-coupled, non-inverting gain amplifier for single 5-V supply operation. This circuit was
used for AC characterization of the OPA689, with a 50-Ω source, which it matches, and a 500-Ω load. The
power-supply bypass capacitors are shown explicitly in Figure 30 and Figure 31, but will be assumed in the
other figures. The limiter voltages (VH and VL) and their bias currents (IVH and IVL) have the polarities shown.
Notice that the single-supply circuit can use three resistors to set VH and VL, where the dual-supply circuit
usually uses four to reference the limit voltages to ground.
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APPLICATION INFORMATION (continued)
V S = +5V
0.1 µ F
2.2 µ F
523 Ω
+
0.1 µ F
V H = 3.7V
Ω
806
0.1 µ F
3
I VH
7
V IN
976 Ω
8
Ω
57.6
Ω
806
0.1 µ F
6
OPA689
2
5
R F
500 Ω
I VL
4
402
V O
Ω
0.1 µ F
R G
402 Ω
V L = 1.3V
523 Ω
0.1 µ F
Figure 31. AC-Coupled, Single Supply Amplifier
Limited Output, ADC Input Driver
The circuit in Figure 32 shows an inverting, low distortion ADC driver that operates on a single supply. The
converter's internal references bias the op amp input. The 4.0-pF and 18-pF capacitors form a compensation
network that allows the OPA689 to have a flat frequency response at a gain of –2. This increases the loop gain
of the op amp feedback network, which reduces the distortion products below their specified values.
V S = +5V
787Ω
4 .0 p F
0 .1 µ F 3 7 4 Ω
750Ω
100Ω
V IN
RE FT
2
V S = +5V
+ 3 .5 V
V S = +5V
18pF
0 .1 µ F
V H = + 3 .6 V
R S EL
+V S
7
8
O P A 689
5
3
2 4 .9 Ω
A DS 822
6
IN
1 0 −B it
1 0 −B i t
D a ta
40M S P S
100pF
4
0 .1 µ F
R EF B
1 .4 0 k Ω
GND
100Ω
0 .1 µ F
1 .4 0 k Ω
+ 2 .5 V
I NT /E X T
+ 1 .5 V
V L = + 1 .4 V
787Ω
Figure 32. Low Distortion, Limiting ADC Input Driver
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APPLICATION INFORMATION (continued)
Precision Half Wave Rectifier
Figure 33 shows a half wave rectifier with outstanding precision and speed. VH (pin 8) will default to a voltage
between 3.1 and 3.8 V if left open, while the negative limit is set to ground.
+V S = +5V
Ω
124
7
2
NC
V IN
8
OPA689
6
V O
5
3
4
150
Ω
Ω
402
− V S = − 5V
Figure 33. Precision Half Wave Rectifier
Very High Speed Comparator
Figure 34 shows a very high speed comparator with hysterisis. The output level are precisely defined, and the
recovery time is exceptional. The output voltage swings between 0.5 V and 3.5 V to provide a logic level output
that switches as VIN crosses VREF.
+V S = +5V
100
Ω
2.00k
604 Ω
Ω
V O
V REF
3
0.1µ F
7
8
95.3
Ω
6
OPA689
2
1.21k
Ω
200k
Ω
0.1µ F
5
V IN
4
−V S = −5 V
Figure 34. Very High Speed Comparator
Transimpedance Amplifier
Figure 35 shows a transimpedance amplifier that has exceptional overdrive characteristics. The feedback
capacitor (CF) stabilizes the circuit for the assumed diode capacitance (CD).
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APPLICATION INFORMATION (continued)
C F
1 .0 p F
4 .3 2 kΩ
λ
V O
C D
5 .0 p F
ID
+V S = + 5V
3
7
N C
− V B
8
O P A 689
4
0 .1 µ F
6
5
2
N C
4 .3 2 k Ω
− VS = − 5V
Figure 35. Transimpedance Amplifier
Design-In Tools
Applications Support
The Texas Instruments web site (http://www.ti.com) has the latest data sheets and other design aids.
Demonstration Boards
Two PC boards are available to assist in the initial evaluation of circuit performance of the OPA689 in both
package styles. These are available as an unpopulated PCB with descriptive documentation. See the
demonstration board literature for more information. The summary information for these boards are shown in
Table 1.
Table 1. Evaluation Module Ordering Information
PRODUCT
PACKAGE
BOARD PART NUMBER
LITERATURE NUMBER
OPA689U
SO-8
DEM-OPA-SO-1A
SBOU009
SPICE Models
Computer simulation of circuit performance using SPICE is often useful when analyzing analog circuit or system
performance. This is particularly true for high speed amplifier circuits where parasitic capacitance and
inductance can have a major effect on frequency response.
SPICE models are available through the Texas Instruments web site (www.ti.com). These models do a good job
of predicting small-signal AC and transient performance under a wide variety of operating conditions. They do
not do as well in predicting the harmonic distortion, temperature effects, or different gain and phase
characteristics. These models do not distinquish between the AC performance of different package types.
Theory of Operation
The OPA689 is a voltage feedback op amp that is stable for gains ≥4. The output voltage is limited to a range
set by the limiter pins (5 and 8). When the input tries to overdrive the output, the limiters take control of the
output buffer. This avoids saturating any parts in the signal path, gives quick overdrive recovery, and gives
consistent limiter accuracy for any gain.
This part is de-compensated (stable for gains ≥4). This gives greater bandwidth, higher slew rate, and lower
noise than the unity gain stable companion part OPA689.
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The limiters have a very sharp transition from the linear region of operation to output limiting. This allows the
limiter voltages to be set very near (<100 mV) the desired signal range. The distortion performance is also very
good near the limiter voltages.
Circuit Layout
Achieving optimum performance with the high-frequency OPA689 requires careful attention to layout design and
component selection. Recommended PCB layout techniques and component selection criteria are:
• Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Open a window in the
ground and power planes around the signal I/O pins, and leave the ground and power planes unbroken
elsewhere.
• Provide a high quality power supply. Use linear regulators, ground plane and power planes to provide
power. Place high-frequency 0.1 µF decoupling capacitors <0.2" away from each power-supply pin. Use
wide, short traces to connect to these capacitors to the ground and power planes. Also use larger (2.2 µF to
6.8 µF) high-frequency decoupling capacitors to bypass lower frequencies. They may be somewhat further
from the device, and be shared among several adjacent devices.
• Place external components close to the OPA689. This minimizes inductance, ground loops, transmission
line effects and propagation delay problems. Be extra careful with the feedback (RF), input and output
resistors.
• Use high-frequency components to minimize parasitic elements. Resistors should be a very low reactance
type. Surface-mount resistors work best and allow a tighter layout. Metal film or carbon composition
axially-leaded resistors can also provide good performance when their leads are as short as possible. Never
use wirewound resistors for high-frequency applications. Remember that most potentiometers have large
parasitic
capacitances
and
inductances.
Multilayer ceramic chip capacitors work best and take up little space. Monolithic ceramic capacitors also work
very well. Use RF type capacitors with low ESR and ESL. The large power pin bypass capacitors (2.2 µF to
6.8 µF) should be tantalum for better high-frequency and pulse performance.
• Choose low resistor values to minimize the time constant set by the resistor and its parasitic parallel
capacitance. Good metal film or surface mount resistors have approximately 0.2 pF parasitic parallel
capacitance. For resistors >1.5 kΩ, this adds a pole and/or zero below 500 MHz.
Make sure that the output loading is not too heavy. The recommended 402-Ω feedback resistor is a good
starting point in your design.
• Use short direct traces to other wideband devices on the board. Short traces act as a lumped capacitive
load. Wide traces (50 to 100 mils) should be used. Estimate the total capacitive load at the output, and use
the series isolation resistor recommended in the typical performance curve "RS vs Capacitive Load". Parasitic
loads <2 pF may not need the isolation resistor.
• When long traces are necessary, use transmission line design techniques (consult an ECL design
handbook for microstrip and stripline layout techniques). A 50-Ω transmission line is not required on
board—a higher characteristic impedance will help reduce output loading. Use a matching series resistor at
the output of the op amp to drive a transmission line, and a matched load resistor at the other end to make
the line appear as a resistor. If the 6 dB of attenuation that the matched load produces is not acceptable, and
the line is not too long, use the series resistor at the source only. This will isolate the source from the
reactive load presented by the line, but the frequency response will be degraded.
Multiple destination devices are best handled as separate transmission lines, each with its own series source
and shunt load terminations. Any parasitic impedances acting on the terminating resistors will alter the
transmission line match, and can cause unwanted signal reflections and reactive loading.
• Do not use sockets for high-speed parts like the OPA689. The additional lead length and pin-to-pin
capacitance introduced by the socket creates an extremely troublesome parasitic network. Best results are
obtained by soldering the part onto the board.
Power Supplies
The OPA689 is nominally specified for operation using either ±5-V supplies or a single 5-V supply. The
maximum specified total supply voltage of 13 V allows reasonable tolerances on the supplies. Higher supply
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voltages can break down internal junctions, possibly leading to catastrophic failure. Single-supply operation is
possible as long as common mode voltage constraints are observed. The common mode input and output
voltage specifications can be interpreted as a required headroom to the supply voltage. Observing this input and
output headroom requirement will allow design of non-standard or single-supply operation circuits.Figure 31
shows one approach to single-supply operation.
ESD Protection
ESD damage has been known to damage MOSFET devices, but any semiconductor device is vulnerable to ESD
damage. This is particularly true for very high-speed, fine geometry processes.
ESD damage can cause subtle changes in amplifier input characteristics without necessarily destroying the
device. In precision operational amplifiers, this may cause a noticeable degradation of offset voltage and drift.
Therefore, ESD handling precautions are required when handling the OPA689.
Output Limiters
The output voltage is linearly dependent on the input(s) when it is between the limiter voltages VH (pin 8) and VL
(pin 5). When the output tries to exceed VH or VL, the corresponding limiter buffer takes control of the output
voltage and holds it at VH or VL.
Because the limiters act on the output, their accuracy does not change with gain. The transition from the linear
region of operation to output limiting is sharp—the desired output signal can safely come to within 30 mV of VH
or VL. Distortion performance is also good over the same range.
The limiter voltages can be set to within 0.7 V of the supplies (VL≥– VS + 0.7 V, VH≤ +VS– 0.7 V). They must
also be at least 400 mV apart (VH– VL≥ 0.4 V).
L IM IT E R IN P U T B IA S C U R R EN T vs B IA S V O L T A G E
L im te r In p u t B ia s C u r re n t ( ∝ A)
100
M a x im u m O v e r T e m p e ra tur e
75
50
25
M in im u m O v e r T e m p e r a tu re
0
− 25
L im ite r H e a d r o o m = + V S − V H
− 50
= V L − ( − V S)
C u r re n t = I VH o r − I VL
− 75
− 100
0 .0
0 .5
1 .0
1 .5
2 .0
2 .5
3 .0
3 .5
4 .0
4 .5
5 .0
L im ite r H e a d r o o m ( V )
Figure 36. Limiter Bias Current vs Bias Voltage
When pins 5 and 8 are left open, VH and VL go to the Default Voltage Limit; the minimum values are in the
Specifications. Looking at Figure 37 for the zero bias current case will show the expected range of (VS– default
limit voltages) = headroom.
When the limiter voltages are more than 2.1 V from the supplies (VL≥– VS + 2.1 V or VH≤ VS– 2.1 V), you can
use simple resistor dividers to set VH and VL (see Figure 30). Make sure you include the Limiter Input Bias
Currents (Figure 37) in the calculations (i.e., IVL≥– 50 µA out of pin 5, and IVH≤ 50 µA out of pin 8). For good
limiter voltage accuracy, run at least 1-mA quiescent bias current through these resistors.
When the limiter voltages need to be within 2.1 V of the supplies (VL≤– VS + 2.1 V or VH≥ VS– 2.1 V), consider
using low impedance buffers to set VH and VL to minimize errors due to bias current uncertainty. This will
typically be the case for single supply operation (VS = 5 V). Figure 31 runs 2.5 mA through the resistive divider
that sets VH and VL. This keeps errors due to IVH and IVL <±1% of the target limit voltages.
The limiters' DC accuracy depends on attention to detail. The two dominant error sources can be improved as
follows:
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•
•
Power supplies, when used to drive resistive dividers that set VH and VL, can contribute large errors (e.g.,
5%). Using a more accurate source, and bypassing pins 5 and 8 with good capacitors, will improve limiter
PSRR.
The resistor tolerances in the resistive divider can also dominate. Use 1% resistors.
Other error sources also contribute, but should have little impact on the limiters' DC accuracy:
• Reduce offsets caused by the Limiter Input Bias Currents. Select the resistors in the resistive divider(s) as
described above.
• Consider the signal path DC errors as contributing to uncertainty in the useable output swing.
• The Limiter Offset Voltage only slightly degrades limiter accuracy.
Figure 37 shows how the limiters affect distortion performance. Virtually no degradation in linearity is observed
for output voltage swinging right up to the limiter voltages.
HARMONIC DISTORTION NEAR LIMIT VOLTAGES
−40
V O = 0V DC ± 1 Vp
f 1 = 5 MHz
R L = 500 Ω
2nd and 3rd Harmonic Distortion (dBc)
−45
−50
−55
HD2
−60
−65
−70
−75
−80
HD3
−85
−90
0.9
1.0
1.1
1.2
1.3
1.4
1.5
1.6
1.7
1.8
1.9
2.0
± Limit Voltage (V)
Figure 37. Harmonic Distortion Near Limit Voltages
Offset Voltage Adjustment
The circuit in Figure 38 allows offset adjustment without degrading offset drift with temperature. Use this circuit
with caution since power supply noise can inadvertently couple into the op amp.
+V
R 2
S
R
T R IM
47 kΩ
V
OPA689
− V
S
0 .1 µ F
O
R 1 (1 )
R 3 (2 ) = R
V
IN
1
|| R
2
o r G ro u n d
NOT ES: (1) Set R 1 << R TRIM. (2) R 3 is optional and
minimizes output offset due to input bias currents.
Figure 38. Offset Voltage Trim
Remember that additional offset errors can be created by the amplifier's input bias currents. Whenever possible,
match the impedance seen by both DC input bias currents using R3. This minimizes the output offset voltage
caused by the input bias currents.
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Output Drive
The OPA689 has been optimized to drive 500-Ω loads, such as ADCs. It still performs very well driving 100-Ω
loads; the specifications are shown for the 500-Ω load. This makes the OPA689 an ideal choice for a wide range
of high-frequency applications.
Many high-speed applications, such as driving ADCs, require op amps with low output impedance. As shown in
the typical performance curve "Output Impedance vs Frequency", the OPA689 maintains very low closed-loop
output impedance over frequency. Closed-loop output impedance increases with frequency, since loop gain
decreases with frequency.
Thermal Considerations
The OPA689 will not require heat-sinking under most operating conditions. Maximum desired junction
temperature will set a maximum allowed internal power dissipation as described below. In no case should the
maximum junction temperature be allowed to exceed 150°C.
The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and the additional power dissipated
in the output stage (PDL) while delivering load power. PDQ is simply the specified no-load supply current times
the total supply voltage across the part. PDL depends on the required output signals and loads. For a grounded
resistive load, and equal bipolar supplies, it is at a maximum when the output is at 1/2 either supply voltage. In
this condition, PDL = VS2/(4RL) where RL includes the feedback network loading. Note that it is the power in the
output stage, and not in the load, that comprises PDL.
The operating junction temperature is: TJ = TA + PDθJA, where TA is the ambient temperature.
For example, the maximum TJ for a OPA689M with G = 6, RFB = 750 Ω, RL = 100 Ω, and ±VS = ±5 V at the
maximum TA = 85°C is calculated as:
P DQ + (10 V
20 mA) + 200 mW
2
P DL +
4
(5 V)
ǒ100 W ø 850 WǓ
P D + 200 mW ) 70 mW + 270 mW
TJ = 85°C + 270 mW × (119°C/W) = 117°C
Capacitive Loads
Capacitive loads, such as the input to ADCs, will decrease the amplifier's phase margin, which may cause
high-frequency peaking or oscillations. Capacitive loads × 2 pF should be isolated by connecting a small resistor
in series with the output as shown in Figure 39. Increasing the gain from +2 will improve the capacitive drive
capabilities due to increased phase margin.
R
S
V
OPA688
R
L
C
O
L
R L is optional
Figure 39. Driving Capacitive Loads
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In general, capacitive loads should be minimized for optimum high-frequency performance. The capacitance of
coax cable (29 pF/foot for RG-58) will not load the amplifier when the coaxial cable, or transmission line, is
terminated in its characteristic impedance.
Frequency Response Compensation
The OPA689 is internally compensated to be unity-gain stable at a gain of +4, and has a nominal phase margin
of 60 degrees at a gain of +6. Phase margin and peaking improve at higher gains. Recall that an inverting gain
of –5 is equivalent to a gain of +6 for bandwidth purposes (i.e., noise gain = 6).
Standard external compensation techniques work with this device. For example, in the inverting configuration,
the bandwidth may be limited without modifying the inverting gain by placing a series RC network to ground on
the inverting node. This has the effect of increasing the noise gain at high frequencies, which limits the
bandwidth.
To maintain a wide bandwidth at high gains, cascade several op amps.
In applications where a large feedback resistor is required, such as photodiode transimpedance amplifier, the
parasitic capacitance from the inverting input to ground causes peaking or oscillations. To compensate for this
effect, connect a small capacitor in parallel with the feedback resistor. The bandwidth will be limited by the pole
that the feedback resistor and this capacitor create. In other high gain applications, use a three resistor "Tee"
network to reduce the RC time constants set by the parasitic capacitances. Be careful to not increase the noise
generated by this feedback network too much.
Pulse Settling Time
The OPA689 is capable of an extremely fast settling time in response to a pulse input. Frequency response
flatness and phase linearity are needed to obtain the best settling times. For capacitive loads, such as an ADC,
use the recommended RS in the typical performance curve "RS vs Capacitive Load". Extremely fine-scale settling
(0.01%) requires close attention to ground return current in the supply decoupling capacitors.
The pulse settling characteristics when recovering from overdrive are very good.
Distortion
The OPA689's distortion performance is specified for a 500-Ω load, such as an ADC. Driving loads with smaller
resistance will increase the distortion as shown in Figure 40. Remember to include the feedback network in the
load resistance calculations.
HARMONIC DISTORTION vs LOAD RESISTANCE
−40
V O = 2 Vp-p
f 1 = 5MHz
2nd and 3rd Harmonic Distortion (dBc)
−45
−50
HD2
−55
−60
−65
HD3
−70
−75
−80
−85
−90
50
100
1000
Load Resistance ( Ω )
Figure 40. 5-MHz Harmonic Distortion vs Load Resistance
20
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PACKAGE OPTION ADDENDUM
www.ti.com
1-Jul-2009
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
OPA689MJD
NRND
CDIP SB
JD
Pins Package Eco Plan (2)
Qty
8
1
TBD
Lead/Ball Finish
MSL Peak Temp (3)
POST-PLATE N / A for Pkg Type
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take
reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on
incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF OPA689M :
• Catalog: OPA689
NOTE: Qualified Version Definitions:
• Catalog - TI's standard catalog product
Addendum-Page 1
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