LT6109-1/LT6109-2 High Side Current Sense Amplifier with Reference and Comparators DESCRIPTION FEATURES n n n n n n n n Current Sense Amplifier – Fast Step Response: 500ns – Low Offset Voltage: 125µV Maximum – Low Gain Error: 0.2% Maximum Internal 400mV Precision Reference Internal Latching Comparators with Reset – Fast Response Time: 500ns – Total Threshold Error: ±1.25% Maximum – Two Comparator Polarity Options Wide Supply Range: 2.7V to 60V Supply Current: 550µA Low Shutdown Current: 5µA Maximum Specified for –40°C to 125°C Temperature Range Available in 10-Lead MSOP Package The LT®6109 is a complete high side current sense device that incorporates a precision current sense amplifier, an integrated voltage reference and two comparators. Two versions of the LT6109 are available. The LT6109-1 has the comparators connected in opposing polarity and the LT6109-2 has the comparators connected in the same polarity. In addition, the current sense amplifier and comparator inputs and outputs are directly accessible. The amplifier gain and comparator trip points are configured by external resistors. The open-drain comparator outputs allows for easy interface to other system components. The overall propagation delay of the LT6109 is typically only 1.4µs, allowing for quick reaction to overcurrent and undercurrent conditions. The 1MHz bandwidth allows the LT6109 to be used for error detection in critical applications such as motor control. The high threshold accuracy of the comparators, combined with the ability to latch both comparators, ensures the LT6109 can capture high speed events. APPLICATIONS n n n n n n n Overcurrent, Undercurrent and Fault Detection Current Shunt Measurement Battery Monitoring Motor Control Automotive Monitoring and Control Remote Sensing Industrial Control The LT6109 is fully specified for operation from –40°C to 125°C, making it suitable for industrial and automotive applications. The LT6109 is available in a small 10-lead MSOP. L, LT, LTC, LTM, TimerBlox, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. TYPICAL APPLICATION Response to Overcurrent Event Circuit Fault Protection with Latching Load Disconnect and Early Warning Indication 0.1Ω 12V 6.2V* 1k 3.3V IRF9640 TO LOAD 0.1µF 100Ω SENSEHI SENSELO 10k 1k 2N2700 1.62k 100k V+ VOUT OUTA LT6109-2 EN/RST RESET 100mA WARNING OUTC2 250mA DISCONNECT OUTC1 ILOAD 200mA/DIV 0mA VOUTC1 5V/DIV 0V VOUTC2 5V/DIV 0V 6.04k INC2 2.37k V– INC1 1.6k 610912 TA01a *CMH25234B VLOAD 10V/DIV 0V 250mA DISCONNECT 100mA WARNING 5µs/DIV 610912 TA01b 610912fa 1 LT6109-1/LT6109-2 ABSOLUTE MAXIMUM RATINGS (Note 1) PIN CONFIGURATION Total Supply Voltage (V+ to V–)..................................60V Maximum Voltage (SENSELO, SENSEHI, OUTA)................................ V+ + 1V Maximum V+ – (SENSELO or SENSEHI).....................33V Maximum EN/RST Voltage.........................................60V Maximum Comparator Input Voltage.........................60V Maximum Comparator Output Voltage......................60V Input Current (Note 2)...........................................–10mA SENSEHI, SENSELO Input Current........................ ±10mA Differential SENSEHI or SENSELO Input Current....±2.5mA Amplifier Output Short-Circuit Duration (to V–)... Indefinite Operating Temperature Range (Note 3) LT6109I.................................................–40°C to 85°C LT6109H............................................. –40°C to 125°C Specified Temperature Range (Note 3) LT6109I.................................................–40°C to 85°C LT6109H............................................. –40°C to 125°C Maximum Junction Temperature........................... 150°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec).................... 300°C TOP VIEW SENSELO EN/RST OUTC2 OUTC1 V– 1 2 3 4 5 10 9 8 7 6 SENSEHI V+ OUTA INC2 INC1 MS PACKAGE 10-LEAD PLASTIC MSOP θJA = 160°C/W, θJC = 45°C/W ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION SPECIFIED TEMPERATURE RANGE LT6109AIMS-1#PBF LT6109AIMS-1#TRPBF LTFNJ 10-Lead Plastic MSOP –40°C to 85°C LT6109IMS-1#PBF LT6109IMS-1#TRPBF LTFNJ 10-Lead Plastic MSOP –40°C to 85°C LT6109AHMS-1#PBF LT6109AHMS-1#TRPBF LTFNJ 10-Lead Plastic MSOP –40°C to 125°C LT6109HMS-1#PBF LT6109HMS-1#TRPBF LTFNJ 10-Lead Plastic MSOP –40°C to 125°C LT6109AIMS-2#PBF LT6109AIMS-2#TRPBF LTFWY 10-Lead Plastic MSOP –40°C to 85°C LT6109IMS-2#PBF LT6109IMS-2#TRPBF LTFWY 10-Lead Plastic MSOP –40°C to 85°C LT6109AHMS-2#PBF LT6109AHMS-2#TRPBF LTFWY 10-Lead Plastic MSOP –40°C to 125°C LT6109HMS-2#PBF LT6109HMS-2#TRPBF LTFWY 10-Lead Plastic MSOP –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 610912fa 2 LT6109-1/LT6109-2 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. V+ = 12V, VPULLUP = V+, VEN/RST = 2.7V, RIN = 100Ω, ROUT = R1 + R2 + R3 = 10k, gain = 100, RC = 25.5k, CL = CLC = 2pF, unless otherwise noted. (See Figure 3) SYMBOL PARAMETER V+ Supply Voltage Range IS Supply Current (Note 4) CONDITIONS MIN l TYP 2.7 V+ = 2.7V, RIN = 1k, VSENSE = 5mV V+ = 60V, RIN = 1k, VSENSE = 5mV 60 475 V+ = 2.7V, VEN/RST = 0V, RIN = 1k, VSENSE = 0.5V VIH VIL VEN/RST = 0V, V+ = 60V EN/RST Pin Input High V+ = 2.7V to 60V l EN/RST Pin Input Low V+ = 2.7V to 60V l µA µA µA 3 5 7 µA µA 7 11 13 µA µA l EN/RST Pin Current V 700 1000 l V+ = 60V, VEN/RST = 0V, RIN = 1k, VSENSE = 0.5V UNITS 600 l Supply Current in Shutdown MAX –200 nA 1.9 V 0.8 V 125 350 250 450 µV µV µV µV Current Sense Amplifier VSENSE = 5mV, LT6109A VSENSE = 5mV, LT6109 VSENSE = 5mV, LT6109A VSENSE = 5mV, LT6109 l l Input Offset Voltage Drift VSENSE = 5mV l Input Bias Current (SENSELO, SENSEHI) V+ = 2.7V to 60V IOS Input Offset Current V+ = 2.7V to 60V IOUTA Output Current (Note 5) PSRR Power Supply Rejection Ratio (Note 6) V+ = 2.7V to 60V Common Mode Rejection Ratio V+ = 36V, VSENSE = 5mV, VICM = 2.7V to 36V VOS ∆VOS/∆T IB CMRR Input Offset Voltage –125 –350 –250 –450 ±0.8 60 l V+ = 60V, V µV/°C 300 350 ±5 l 1 l 120 114 nA mA 127 dB dB 125 dB 125 dB dB l 110 103 RIN = 500Ω l 500 V+ = 2.7V to 12V V+ = 12V to 60V, VSENSE = 5mV to 100mV l –0.2 SENSELO Voltage (Note 8) V+ = 2.7V, VSENSE = 100mV, ROUT = 2k V+ = 60V, VSENSE = 100mV l l 2.5 27 Output Swing High (V+ to VOUTA) V+ = 2.7V, VSENSE = 27mV l 0.2 V V+ = 12V, V l 0.5 V VSENSE(MAX) Full-Scale Input Sense Voltage (Note 5) Gain Error (Note 7) SENSE = 5mV, VICM = 27V to 60V nA nA SENSE = 120mV mV –0.08 0 % % V V BW Signal Bandwidth IOUT = 1mA IOUT = 100µA 1 140 MHz kHz tr Input Step Response (to 50% of Final Output Voltage) V+ = 2.7V, VSENSE = 24mV Step, Output Rising Edge V+ = 12V to 60V, VSENSE = 100mV Step, Output Rising Edge 500 500 ns ns tSETTLE Settling Time to 1% VSENSE = 10mV to 100mV, ROUT = 2k 2 µs 610912fa 3 LT6109-1/LT6109-2 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. V+ = 12V, VPULLUP = V+, VEN/RST = 2.7V, RIN = 100Ω, ROUT = R1 + R2 + R3 = 10k, gain = 100, RC = 25.5k, CL = CLC = 2pF, unless otherwise noted. (See Figure 3) SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Reference and Comparator VTH(R) (Note 9) Rising Input Threshold Voltage (LT6109-1 Comparator 1 LT6109-2 Both Comparators) V+ = 2.7V to 60V, LT6109A V+ = 2.7V to 60V, LT6109 l l 395 392 400 400 405 408 mV mV VTH(F) (Note 9) Falling Input Threshold Voltage (LT6109-1 Comparator 2) V+ = 2.7V to 60V, LT6109A V+ = 2.7V to 60V, LT6109 l l 395 392 400 400 405 408 mV mV VHYS VHYS = VTH(R) – VTH(F) V+ = 2.7V to 60V 3 10 15 mV Comparator Input Bias Current VOL Output Low Voltage VINC1,2 = 0V, V+ = 60V IOUTC1,C2 l –50 = 500µA, V+ = 2.7V nA 60 l High to Low Propagation Delay 5mV Overdrive 100mV Overdrive 150 220 mV mV 3 0.5 µs µs Output Fall Time 0.08 µs tRESET Reset Time 0.5 µs tRPW Valid RST Pulse Width Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: Input and output pins have ESD diodes connected to ground. The SENSEHI and SENSELO pins have additional current handling capability specified as SENSEHI, SENSELO input current. Note 3: The LT6109I is guaranteed to meet specified performance from –40°C to 85°C. LT6109H is guaranteed to meet specified performance from –40°C to 125°C. Note 4: Supply current is specified with the comparator outputs high. When the comparator outputs go low the supply current will increase by 75µA typically per comparator. l 2 15 µs Note 5: The full-scale input sense voltage and the maximum output current must be considered to achieve the specified performance. Note 6: Supply voltage and input common mode voltage are varied while amplifier input offset voltage is monitored. Note 7: Specified gain error does not include the effects of external resistors RIN and ROUT. Although gain error is only guaranteed between 12V and 60V, similar performance is expected for V+ < 12V, as well. Note 8: Refer to SENSELO, SENSEHI Range in the Applications Information section for more information. Note 9: The input threshold voltage which causes the output voltage of the comparator to transition from high to low is specified. The input voltage which causes the comparator output to transition from low to high is the magnitude of the difference between the specified threshold and the hysteresis. 610912fa 4 LT6109-1/LT6109-2 TYPICAL PERFORMANCE CHARACTERISTICS Performance characteristics taken at TA = 25°C, + + V = 12V, VPULLUP = V , VEN/RST = 2.7V, RIN = 100Ω, ROUT = R1 + R2 + R3 = 10k, gain = 100, RC = 25.5k, CL = CLC = 2pF, unless otherwise noted. (See Figure 3) Supply Current vs Supply Voltage Start-Up Supply Current Enable/Disable Response 700 SUPPLY CURRENT (µA) 600 V+ 5V/DIV 500 VEN/RST 2V/DIV 0V 400 0V 300 200 IS 500µA/DIV 100 0µA 0 10 40 30 20 SUPPLY VOLTAGE (V) 50 60 10µs/DIV 610912 G03 Input Offset Voltage vs Temperature Amplifier Offset Voltage vs Supply Voltage 100 5 TYPICAL UNITS 200 Offset Voltage Drift Distribution 12 5 TYPICAL UNITS 80 60 OFFSET VOLTAGE (µV) INPUT OFFSET VOLTAGE (µV) 300 100µs/DIV 610912 G02 610912 G01 100 0 –100 40 20 0 –20 –40 –60 –200 10 PERCENTAGE OF UNITS (%) 0 IS 500µA/DIV 0µA 8 6 4 2 –80 –300 –40 –25 –10 5 20 35 50 65 80 95 110 125 TEMPERATURE (°C) –100 0 10 30 40 20 SUPPLY VOLTAGE (V) Amplifier Gain Error vs Temperature 25 Amplifier Output Swing vs Temperature RIN = 1k –0.05 RIN = 100Ω –0.10 –0.15 0.50 VSENSE = 5mV TO 100mV 0.45 0.40 20 V+ = 12V VSENSE = 120mV 0.35 V+ – VOUTA (V) PERCENTAGE OF UNITS (%) 0 –2 –1.5 –1 –0.5 0 0.5 1 1.5 2 OFFSET VOLTAGE DRIFT (µV/°C) 610912 G06 Amplifier Gain Error Distribution VSENSE = 5mV TO 100mV GAIN ERROR (%) 0 60 610912 G05 610912 G04 0.05 50 0.30 15 0.25 0.20 10 V+ = 2.7V VSENSE = 27mV 0.15 0.10 5 0.05 –0.20 –50 –25 50 0 75 25 TEMPERATURE (°C) 100 125 610912 G07 0 –0.048 –0.052 –0.056 –0.060 –0.064 –0.68 GAIN ERROR (%) 610912 G08 0 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 610912 G18 610912fa 5 LT6109-1/LT6109-2 TYPICAL PERFORMANCE CHARACTERISTICS Performance characteristics taken at TA = 25°C, + + V = 12V, VPULLUP = V , VEN/RST = 2.7V, RIN = 100Ω, ROUT = R1 + R2 + R3 = 10k, gain = 100, RC = 25.5k, CL = CLC = 2pF, unless otherwise noted. (See Figure 3) Common Mode Rejection Ratio vs Frequency Power Supply Rejection Ratio vs Frequency 120 100 80 60 40 20 1 10 100 1k 10k 100k FREQUENCY (Hz) 1M 120 100 80 60 28 G = 20, ROUT = 2k 22 20 1 10 100 1k 10k 100k FREQUENCY (Hz) 1M 10M 16 IOUTA = 1mA IOUTA = 100µA 1k 10k 100k 1M FREQUENCY (Hz) Amplifier Step Response (VSENSE = 0mV to 100mV) Amplifier Input Bias Current vs Temperature System Step Response 100 VSENSE 100mV/DIV 0V RIN = 100Ω G = 100V/V INPUT BIAS CURRENT (nA) 90 VOUTA 1V/DIV 0V VOUTC1 2V/DIV 0V ROUT = 2k,100mV INC1 OVERDRIVE 2µs/DIV 610912 G12 10M 610912 G11 610912 G10 610912 G09 VEN/RST 5V/DIV 0V G = 50, ROUT = 5k 34 40 0 10M G = 100 40 GAIN (dB) COMMON MODE REJECTION RATIO (dB) POWER SUPPLY REJECTION RATIO (dB) 140 0 Amplifier Gain vs Frequency 46 140 160 80 70 SENSEHI 60 50 VOUTA 2V/DIV SENSELO 0V 40 30 20 VSENSE 50mV/DIV 10 0V 0 –40 –25 –10 5 20 35 50 65 80 95 110 125 TEMPERATURE (°C) 2µs/DIV 610912 G14 610912 G13 Amplifier Step Response (VSENSE = 10mV to 100mV) Amplifier Step Response (VSENSE = 10mV to 100mV) Amplifier Step Response (VSENSE = 0mV to 100mV) RIN = 1k ROUT = 20k G = 20V/V RIN = 100Ω G = 100V/V VOUTA 2V/DIV RIN = 1k ROUT = 20k G = 20V/V VOUTA 1V/DIV VOUTA 1V/DIV 0V 0V VSENSE 100mV/DIV 0V VSENSE 100mV/DIV 0V 0V VSENSE 50mV/DIV 0V 2µs/DIV 2µs/DIV 610912 G15 2µs/DIV 610912 G16 610912 G17 610912fa 6 LT6109-1/LT6109-2 TYPICAL PERFORMANCE CHARACTERISTICS Performance characteristics taken at TA = 25°C, + + V = 12V, VPULLUP = V , VEN/RST = 2.7V, RIN = 100Ω, ROUT = R1 + R2 + R3 = 10k, gain = 100, RC = 25.5k, CL = CLC = 2pF, unless otherwise noted. (See Figure 3) Comparator Threshold vs Temperature Comparator Threshold Distribution 408 20 15 10 5 0 396 397.6 399.2 400.8 402.8 COMPARATOR THRESHOLD (mV) Hysteresis Distribution 30 5 TYPICAL PARTS 406 25 PERCENTAGE OF UNITS (%) COMPARATOR THRESHOLD (mV) PERCENTAGE OF UNITS (%) 25 404 402 400 398 396 394 14 12 10 8 6 4 2 0 –40 –25 –10 5 20 35 50 65 80 95 110 125 TEMPERATURE (°C) 0 10 8 6 4 –50 –100 –150 –200 2 0 0 10 20 30 V+ (V) 40 50 60 –250 –10 125°C 25°C –40°C 20 40 COMPARATOR INPUT VOLTAGE (V) 60 610912 G25 1.00 0 –5 –10 125°C 25°C –40°C –15 –20 0 0.2 0.4 0.6 0.8 COMPARATOR INPUT VOLTAGE (V) 50 60 125°C 25°C –40°C 5 VOLOUTC1, OUTC2 (V) COMPARATOR INPUT BIAS CURRENT (nA) –5 20 30 40 EN/RST VOLTAGE (V) Comparator Output Low Voltage vs Output Sink Current 10 0 10 610912 G24 Comparator Input Bias Current vs Input Voltage 5 0 610912 G23 10 COMPARATOR INPUT BIAS CURRENT (nA) EN/RST Current vs Voltage 12 Comparator Input Bias Current vs Input Voltage 0 5 50 5 TYPICAL PARTS 610912 G22 –20 10 610912 G21 EN/RST CURRENT (nA) COMPARATOR HYSTERESIS (mV) COMPARATOR HYSTERESIS (mV) 18 –15 15 Hysteresis vs Supply Voltage Hysteresis vs Temperature 20 14 125°C 610912 G20 610912 G19 16 25°C 0 3.0 4.6 6.2 7.7 9.3 10.9 12.5 14.1 15.7 17.3 COMPARATOR HYSTERESIS (mV) 392 –40 –25 –10 5 20 35 50 65 80 95 110 125 TEMPERATURE (°C) 404 –40°C 20 1.0 610912 G26 0.75 0.50 0.25 0 0 2 1 3 IOUTC (mA) 610912 G27 610912fa 7 LT6109-1/LT6109-2 TYPICAL PERFORMANCE CHARACTERISTICS Performance characteristics taken at TA = 25°C, + + V = 12V, VPULLUP = V , VEN/RST = 2.7V, RIN = 100Ω, ROUT = R1 + R2 + R3 = 10k, gain = 100, RC = 25.5k, CL = CLC = 2pF, unless otherwise noted. (See Figure 3) Comparator Propagation Delay vs Input Overdrive Comparator Output Leakage Current vs Pull-Up Voltage 5.0 18 125°C 13 8 3 –2 –40°C AND 25°C 0 50 60 20 30 40 10 COMPARATOR OUTPUT PULL-UP VOLTAGE (V) 10000 4.0 3.5 3.0 2.5 2.0 H TO L 1.5 1000 RISE TIME FALL TIME 100 1.0 L TO H 0.5 0 0 40 120 160 200 80 COMPARATOR INPUT OVERDRIVE (mV) 610912 G28 10 1 10 100 RC PULL-UP RESISTOR (kΩ) 1000 610912 G30 610912 G29 Comparator Step Response (5mV INC1 Overdrive) Comparator Step Response (100mV INC1 Overdrive) Comparator Reset Response VINC 0.5V/DIV 0V VINC 0.5V/DIV 0V VOH = 0.9 • VPULLUP VOL = 0.1 • VPULLUP 100mV INC1 OVERDRIVE CL = 2pF 4.5 RISE/FALL TIME (ns) COMPARATOR PROPAGATION DELAY (µs) OUTC1, OUTC2 LEAKAGE CURRENT (nA) 23 Comparator Rise/Fall Time vs Pull-Up Resistor VOUTC 5V/DIV 0V VOUTC 2V/DIV VOUTC 2V/DIV 0V 0V VEN/RST 5V/DIV 0V VEN/RST 2V/DIV VEN/RST 5V/DIV 0V 0V 5µs/DIV 5µs/DIV 610912 G31 5µs/DIV 610912 G32 610912 G33 PIN FUNCTIONS SENSELO (Pin 1): Sense Amplifier Input. This pin must be tied to the load end of the sense resistor. EN/RST (Pin 2): Enable and Latch Reset Input. When the EN/RST pin is pulled high the LT6109 is enabled. When the EN/RST pin is pulled low for longer than typically 40µs, the LT6109 will enter the shutdown mode. Pulsing this pin low for between 2µs and 15µs will reset the comparators of the LT6109. OUTC2 (Pin 3): Open-Drain Comparator 2 Output. Offstate voltage may be as high as 60V above V–, regardless of V+ used. OUTC1 (Pin 4): Open-Drain Comparator 1 Output. Offstate voltage may be as high as 60V above V–, regardless of V+ used. V– (Pin 5): Negative Supply Pin. This pin is normally connected to ground. 610912fa 8 LT6109-1/LT6109-2 PIN FUNCTIONS V+ (Pin 9): Positive Supply Pin. The V+ pin can be connected directly to either side of the sense resistor, RSENSE. When V+ is tied to the load end of the sense resistor, the SENSEHI pin can go up to 0.2V above V+. Supply current is drawn through this pin. INC1 (Pin 6): This is the inverting input of comparator 1. The second input of this comparator is internally connected to the 400mV reference. INC2 (Pin 7): This is the input of comparator 2. For the LT6109-1 this is the noninverting input of comparator 2. For the LT6109-2 this is the inverting input of comparator 2. The second input of each of these comparators is internally connected to the 400mV reference. SENSEHI (Pin 10): Sense Amplifier Input. The internal sense amplifier will drive SENSEHI to the same potential as SENSELO. A resistor (typically RIN) tied from supply to SENSEHI sets the output current, IOUT = VSENSE/RIN, where VSENSE is the voltage developed across RSENSE. OUTA (Pin 8): Current Output of the Sense Amplifier. This pin will source a current that is equal to the sense voltage divided by the external gain setting resistor, RIN. BLOCK DIAGRAMS 9 V+ LT6109-1 100Ω 10 1 SENSEHI 3k SENSELO 3k 34V – + OUTA V– V– 6V 8 V– V+ 200nA 2 EN/RST ENABLE AND RESET TIMING RESET V+ OUTC2 INC2 + 3 UNDERCURRENT FLAG 7 – V– 400mV REFERENCE V+ OUTC1 + OVERCURRENT FLAG – 4 INC1 6 V– 5 610912 F01 Figure 1. LT6109-1 Block Diagram (Comparators with Opposing Polarity) 610912fa 9 LT6109-1/LT6109-2 BLOCK DIAGRAMS 9 V+ LT6109-2 100Ω 10 1 SENSEHI 3k SENSELO 3k 34V – + OUTA V– V– 6V 8 V– V+ 200nA 2 EN/RST ENABLE AND RESET TIMING RESET V+ INC2 – 3 OUTC2 OVERCURRENT FLAG 7 + V– 400mV REFERENCE V+ OUTC1 + 4 OVERCURRENT FLAG INC1 6 – V– 5 610912 F02 Figure 2. LT6109-2 Block Diagram (Comparators with the Same Polarity) APPLICATIONS INFORMATION The LT6109 high side current sense amplifier provides accurate monitoring of currents through an external sense resistor. The input sense voltage is level-shifted from the sensed power supply to a ground referenced output and is amplified by a user-selected gain to the output. The output voltage is directly proportional to the current flowing through the sense resistor. The LT6109 comparators have a threshold set with a built-in 400mV precision reference and have 10mV of hysteresis. The open-drain outputs can be easily used to level shift to digital supplies. Amplifier Theory of Operation An internal sense amplifier loop forces SENSEHI to have the same potential as SENSELO as shown in Figure 3. Connecting an external resistor, RIN, between SENSEHI and VSUPPLY forces a potential, VSENSE, across RIN. A corresponding current, IOUTA, equal to VSENSE/RIN, will flow through RIN. The high impedance inputs of the sense amplifier do not load this current, so it will flow through an internal MOSFET to the output pin, OUTA. 610912fa 10 LT6109-1/LT6109-2 APPLICATIONS INFORMATION The output current can be transformed back into a voltage by adding a resistor from OUTA to V–(typically ground). The output voltage is then: VOUT = V– + I OUTA • ROUT where ROUT = R1 + R2 + R3 as shown in Figure 3. Table 1. Example Gain Configurations GAIN RIN ROUT VSENSE FOR VOUT = 5V IOUTA AT VOUT = 5V 20 499Ω 10k 250mV 500µA 50 200Ω 10k 100mV 500µA 100 100Ω 10k 50mV 500µA Useful Equations Input Voltage: VSENSE = ISENSE •RSENSE Voltage Gain: VOUT R = OUT VSENSE RIN I R Current Gain: OUTA = SENSE ISENSE RIN Note that VSENSE(MAX) can be exceeded without damaging the amplifier, however, output accuracy will degrade as VSENSE exceeds VSENSE(MAX), resulting in increased output current, IOUTA. Selection of External Current Sense Resistor The external sense resistor, RSENSE, has a significant effect on the function of a current sensing system and must be chosen with care. First, the power dissipation in the resistor should be considered. The measured load current will cause power dissipation as well as a voltage drop in RSENSE. As a result, the sense resistor should be as small as possible while still providing the input dynamic range required by the measurement. Note that the input dynamic range is the difference between the maximum input signal and the minimum accurately reproduced signal, and is limited primarily by input DC offset of the internal sense amplifier of the LT6109. To ensure the specified performance, RSENSE should be small enough that VSENSE does not exceed VSENSE(MAX) under peak load conditions. As an example, an application may require the maximum sense voltage be 100mV. If this application is expected to draw 2A at peak load, RSENSE should be set to 50mΩ. Once the maximum RSENSE value is determined, the minimum sense resistor value will be set by the resolution or dynamic range required. The minimum signal that can be accurately represented by this sense amplifier is limited by the input offset. As an example, the LT6109 has a maximum input offset of 125µV. If the minimum current is 20mA, a sense resistor of 6.25mΩ will set VSENSE to 125µV. This is the same value as the input offset. A larger sense resistor will reduce the error due to offset by increasing the sense voltage for a given load current. Choosing a 50mΩ RSENSE will maximize the dynamic range and provide a system that has 100mV across the sense resistor at peak load (2A), while input offset causes an error equivalent to only 2.5mA of load current. In the previous example, the peak dissipation in RSENSE is 200mW. If a 5mΩ sense resistor is employed, then the effective current error is 25mA, while the peak sense voltage is reduced to 10mV at 2A, dissipating only 20mW. The low offset and corresponding large dynamic range of the LT6109 make it more flexible than other solutions in this respect. The 125µV maximum offset gives 72dB of dynamic range for a sense voltage that is limited to 500mV max. Sense Resistor Connection Kelvin connection of the SENSEHI and SENSELO inputs to the sense resistor should be used in all but the lowest power applications. Solder connections and PC board interconnections that carry high currents can cause significant error in measurement due to their relatively large resistances. One 10mm × 10mm square trace of 1oz copper is approximately 0.5mΩ. A 1mV error can be caused by as little as 2A flowing through this small interconnect. This will cause a 1% error for a full-scale VSENSE of 100mV. A 10A load current in the same interconnect will cause a 5% error for the same 100mV signal. By isolating the sense traces from the high current paths, this error can be reduced by orders of magnitude. A sense resistor with integrated Kelvin sense terminals will give the best results. Figure 3 illustrates the recommended method for connecting the SENSEHI and SENSELO pins to the sense resistor. 610912fa 11 LT6109-1/LT6109-2 APPLICATIONS INFORMATION VSUPPLY + RIN VSENSE – + LOAD V ISENSE = SENSE RSENSE VPULLUP VRESET RC – V– V+ V+ 2 EN/RST 9 C1 OUTA 8 V+ 3 OUTC2 CLC IOUTA V– RC INC2 7 VOUT R3* CL 4 OUTC1 5 V– – CLC 400mV REFERENCE V+ R2* + OVERCURRENT FLAG SENSEHI 10 – UNDERCURRENT FLAG LT6109-1 1 SENSELO + RSENSE INC1 6 R1* V– 610912 F03 *ROUT = R1 + R2 + R3 Figure 3. LT6109-1 Typical Connection Selection of External Input Gain Resistor, RIN RIN should be chosen to allow the required speed and resolution while limiting the output current to 1mA. The maximum value for RIN is 1k to maintain good loop stability. For a given VSENSE, larger values of RIN will lower power dissipation in the LT6109 due to the reduction in IOUT while smaller values of RIN will result in faster response time due to the increase in IOUT . If low sense currents must be resolved accurately in a system that has a very wide dynamic range, a smaller RIN may be used if the maximum IOUTA current is limited in another way, such as with a Schottky diode across RSENSE (Figure 4). This will reduce the high current measurement accuracy by limiting the result, while increasing the low current measurement resolution. This approach can be helpful in cases where occasional bursts of high currents can be ignored. V+ RSENSE DSENSE 610912 F04 LOAD Figure 4. Shunt Diode Limits Maximum Input Voltage to Allow Better Low Input Resolution Without Overranging Care should be taken when designing the board layout for RIN, especially for small RIN values. All trace and interconnect resistances will increase the effective RIN value, causing a gain error. The power dissipated in the sense resistor can create a thermal gradient across a printed circuit board and consequently a gain error if RIN and ROUT are placed such that they operate at different temperatures. If significant power is being dissipated in the sense resistor then care 610912fa 12 LT6109-1/LT6109-2 APPLICATIONS INFORMATION should be taken to place RIN and ROUT such that the gain error due to the thermal gradient is minimized. Selection of External Output Gain Resistor, ROUT The output resistor, ROUT , determines how the output current is converted to voltage. VOUT is simply IOUTA • ROUT . Typically, ROUT is a combination of resistors configured as a resistor divider which has voltage taps going to the comparator inputs to set the comparator thresholds. In choosing an output resistor, the maximum output voltage must first be considered. If the subsequent circuit is a buffer or ADC with limited input range, then ROUT must be chosen so that IOUTA(MAX) • ROUT is less than the allowed maximum input range of this circuit. In addition, the output impedance is determined by ROUT . If another circuit is being driven, then the input impedance of that circuit must be considered. If the subsequent circuit has high enough input impedance, then almost any useful output impedance will be acceptable. However, if the subsequent circuit has relatively low input impedance, or draws spikes of current such as an ADC load, then a lower output impedance may be required to preserve the accuracy of the output. More information can be found in the Output Filtering section. As an example, if the input impedance of the driven circuit, RIN(DRIVEN), is 100 times ROUT, then the accuracy of VOUT will be reduced by 1% since: VOUT = IOUTA • ROUT •RIN(DRIVEN) ROUT +RIN(DRIVEN) = IOUTA •ROUT • 100 = 0.99 •IOUTA •ROUT 101 Amplifier Error Sources The current sense system uses an amplifier and resistors to apply gain and level-shift the result. Consequently, the output is dependent on the characteristics of the amplifier, such as gain error and input offset, as well as the matching of the external resistors. In this case, the only error is due to external resistor mismatch, which provides an error in gain only. However, offset voltage, input bias current and finite gain in the amplifier can cause additional errors: Output Voltage Error, ∆VOUT(VOS), Due to the Amplifier DC Offset Voltage, VOS ∆VOUT(VOS) = VOS • ROUT RIN The DC offset voltage of the amplifier adds directly to the value of the sense voltage, VSENSE. As VSENSE is increased, accuracy improves. This is the dominant error of the system and it limits the available dynamic range. Output Voltage Error, ∆VOUT(IBIAS), Due to the Bias Currents IB+ and IB– The amplifier bias current IB+ flows into the SENSELO pin while IB– flows into the SENSEHI pin. The error due to IB is the following: R ∆VOUT(IBIAS) = ROUT IB+ • SENSE –IB– RIN Since IB+ ≈ IB– = IBIAS, if RSENSE << RIN then, ∆VOUT(IBIAS) = –ROUT (IBIAS) It is useful to refer the error to the input: ∆VVIN(IBIAS) = –RIN (IBIAS) For instance, if IBIAS is 100nA and RIN is 1k, the input referred error is 100µV. This error becomes less significant as the value of RIN decreases. The bias current error can be reduced if an external resistor, RIN+, is connected as shown in Figure 5, the error is then reduced to: VOUT(IBIAS) = ±ROUT • IOS; IOS = IB+ – IB– Minimizing low current errors will maximize the dynamic range of the circuit. Ideally, the circuit output is: VOUT = VSENSE • ROUT ; VSENSE = RSENSE •ISENSE RIN 610912fa 13 LT6109-1/LT6109-2 APPLICATIONS INFORMATION There is also power dissipated due to the quiescent power supply current: V+ 9 V+ LT6109 VBATT RSENSE PS = IS • V+ – RIN 10 SENSEHI + 1 SENSELO RIN+ OUTA 8 ISENSE 5 VOUT ROUT V– 610912 F05 Figure 5. RIN+ Reduces Error Due to IB Output Voltage Error, ∆VOUT(GAIN ERROR), Due to External Resistors The LT6109 exhibits a very low gain error. As a result, the gain error is only significant when low tolerance resistors are used to set the gain. Note the gain error is systematically negative. For instance, if 0.1% resistors are used for RIN and ROUT then the resulting worst-case gain error is –0.4% with RIN = 100Ω. Figure 6 is a graph of the maximum gain error which can be expected versus the external resistor tolerance. RESULTING GAIN ERROR (%) 10 The comparator output current flows into the comparator output pin and out of the V– pin. The power dissipated in the LT6109 due to each comparator is often insignificant and can be calculated as follows: POUTC1,C2 = (VOUTC1,C2 – V–) • IOUTC1,C2 The total power dissipated is the sum of these dissipations: PTOTAL = POUTA + POUTC1 + POUTC2 + PS At maximum supply and maximum output currents, the total power dissipation can exceed 100mW. This will cause significant heating of the LT6109 die. In order to prevent damage to the LT6109, the maximum expected dissipation in each application should be calculated. This number can be multiplied by the θJA value, 160°C/W, to find the maximum expected die temperature. Proper heat sinking and thermal relief should be used to ensure that the die temperature does not exceed the maximum rating. Output Filtering 1 RIN = 100Ω 0.1 0.01 0.01 RIN = 1k 0.1 1 RESISTOR TOLERANCE (%) 10 610912 F06 Figure 6. Gain Error vs Resistor Tolerance Output Current Limitations Due to Power Dissipation The LT6109 can deliver a continuous current of 1mA to the OUTA pin. This current flows through RIN and enters the current sense amplifier via the SENSEHI pin. The power dissipated in the LT6109 due to the output signal is: POUT = (VSENSEHI – VOUTA) • IOUTA Since VSENSEHI ≈ V+, POUTA ≈ (V+ – VOUTA) • IOUTA The AC output voltage, VOUT, is simply IOUTA • ZOUT. This makes filtering straightforward. Any circuit may be used which generates the required ZOUT to get the desired filter response. For example, a capacitor in parallel with ROUT will give a lowpass response. This will reduce noise at the output, and may also be useful as a charge reservoir to keep the output steady while driving a switching circuit such as a MUX or ADC. This output capacitor in parallel with ROUT will create an output pole at: f –3dB = 1 2 • π •ROUT • CL SENSELO, SENSEHI Range The difference between VBATT (see Figure 7) and V+, as well as the maximum value of VSENSE, must be considered to ensure that the SENSELO pin doesn’t exceed the range listed in the Electrical Characteristics table. The SENSELO and SENSEHI pins of the LT6109 can function from 0.2V 610912fa 14 LT6109-1/LT6109-2 APPLICATIONS INFORMATION 60 ALLOWABLE OPERATING VOLTAGES ON SENSELO AND SENSEHI INPUTS (V) above the positive supply to 33V below it. These operating voltages are limited by internal diode clamps shown in Figures 1 and 2. On supplies less than 35.5V, the lower range is limited by V– + 2.5V. This allows the monitored supply, VBATT , to be separate from the LT6109 positive supply as shown in Figure 7. Figure 8 shows the range of operating voltages for the SENSELO and SENSEHI inputs, for different supply voltage inputs (V+). The SENSELO and SENSEHI range has been designed to allow the LT6109 to monitor its own supply current (in addition to the load), as long as VSENSE is less than 200mV. This is shown in Figure 9. 50 40.2V 40 VALID SENSELO/ SENSEHI RANGE 30 27 20.2V 20 10 2.8V 2.5V Minimum Output Voltage The output of the LT6109 current sense amplifier can produce a non-zero output voltage when the sense voltage is zero. This is a result of the sense amplifier VOS being forced across RIN as discussed in the Output Voltage Error, ∆VOUT(VOS) section. Figure 10 shows the effect of the input offset voltage on the transfer function for parts at the VOS limits. With a negative offset voltage, zero input sense voltage produces an output voltage. With a positive offset voltage, the output voltage is zero until the input sense voltage exceeds the input offset voltage. Neglecting VOS, the output circuit is not limited by saturation of pull-down circuitry and can reach 0V. 2.7 10 20 30 35.5 40 V + (V) 60 610912 F08 Figure 8. Allowable SENSELO, SENSEHI Voltage Range 9 V+ LT6109 VBATT RIN 10 SENSEHI RSENSE 1 SENSELO – + OUTA 8 5 VOUT ROUT V– ISENSE Response Time The LT6109 amplifier is designed to exhibit fast response to inputs for the purpose of circuit protection or current monitoring. This response time will be affected by the external components in two ways, delay and speed. 50 610912 F09 Figure 9. LT6109 Supply Current Monitored with Load 120 G = 100 100 OUTPUT VOLTAGE (mV) V+ 9 V+ LT6109 VBATT RIN 10 SENSEHI RSENSE ISENSE 1 SENSELO – + OUTA 8 V– 5 VOUT ROUT 610912 F07 80 VOS = –125µV 60 40 VOS = 125µV 20 0 0 100 200 300 400 500 600 700 800 900 1000 INPUT SENSE VOLTAGE (µV) 610912 F10 Figure 7. V+ Powered Separately from Load Supply (VBATT) Figure 10. Amplifier Output Voltage vs Input Sense Voltage 610912fa 15 LT6109-1/LT6109-2 APPLICATIONS INFORMATION If the output current is very low and an input transient occurs, there may be an increased delay before the output voltage begins to change. The Typical Performance Characteristics show that this delay is short and it can be improved by increasing the minimum output current, either by increasing RSENSE or decreasing RIN. Note that the Typical Performance Characteristics are labeled with respect to the initial sense voltage. The speed is also affected by the external components. Using a larger ROUT will decrease the response time, since VOUT = IOUTA • ZOUT where ZOUT is the parallel combination of ROUT and any parasitic and/or load capacitance. Note that reducing RIN or increasing ROUT will both have the effect of increasing the voltage gain of the circuit. If the output capacitance is limiting the speed of the system, RIN and ROUT can be decreased together in order to maintain the desired gain and provide more current to charge the output capacitance. The response time of the comparators is the sum of the propagation delay and the fall time. The propagation delay is a function of the overdrive voltage on the input of the comparators. A larger overdrive will result in a lower propagation delay. This helps achieve a fast system response time to fault events. The fall time is affected by the load on the output of the comparator as well as the pull-up voltage. The LT6109 amplifier has a typical response time of 500ns and the comparators have a typical response time of 500ns. When configured as a system, the amplifier output drives the comparator input causing a total system response time which is typically greater than that implied by the individually specified response times. This is due to the overdrive on the comparator input being determined by the speed of the amplifier output. Internal Reference and Comparators The integrated precision reference and comparators combined with the high precision current sense allow for rapid and easy detection of abnormal load currents. This is often critical in systems that require high levels of safety and reliability. The LT6109 comparators are optimized for fault detection and are designed with latching outputs. Latching outputs prevent faults from clearing themselves and require a separate system or user to reset the outputs. In applications where the comparator output can intervene and disconnect loads from the supply, latched outputs are required to avoid oscillation. Latching outputs are also useful for detecting problems that are intermittent. The comparator outputs on the LT6109 are always latching and there is no way to disable this feature. Each of the comparators has one input available externally, with the two versions of the part differing by the polarity of those available inputs. The other comparator inputs are connected internally to the 400mV precision reference. The input threshold (the voltage which causes the output to transition from high to low) is designed to be equal to that of the reference. The reference voltage is established with respect to the device V– connection. Comparator Inputs The comparator inputs can swing from V– to 60V regardless of the supply voltage used. The input current for inputs well above the threshold is just a few pAs. With decreasing input voltage, a small bias current begins to be drawn out of the input near the threshold, reaching 50nA max when at ground potential. Note that this change in input bias current can cause a small nonlinearity in the OUTA transfer function if the comparator inputs are coupled to the amplifier output with a voltage divider. For example, if the maximum comparator input current is 50nA, and the resistance seen looking out of the comparator input is 1k, then a change in output voltage of 50µV will be seen on the analog output when the comparator input voltage passes through its threshold. If both comparator inputs are connected to the output then they must both be considered. Setting Comparator Thresholds The comparators have an internal precision 400mV reference. In order to set the trip points of the LT6109-1 comparators, the output currents, IOVER and IUNDER, as well as the maximum output current, IMAX, must be calculated: IOVER = IMAX = VSENSE(OVER) RIN , IUNDER = VSENSE(UNDER) RIN , VSENSE(MAX) RIN 610912fa 16 LT6109-1/LT6109-2 APPLICATIONS INFORMATION where IOVER and IUNDER are the over and under currents through the sense resistor which cause the comparators to trip. IMAX is the maximum current through the sense resistor. R1= R2 = Depending on the desired maximum amplifier output voltage (VMAX) the three output resistors, R1, R2 and R3, can be configured in two ways. If: 400mV 400mV –IUNDER (R1) VMAX > + IMAX IUNDER IOVER If: 400mV IOVER 400mV – IUNDER (R1) IUNDER V –I (R1+ R2 ) R3 = MAX MAX IMAX 400mV 400mV –IUNDER (R1) VMAX < + IMAX IUNDER IOVER then use the configuration shown in Figure 3. The desired trip points and full-scale analog output voltage for the circuit in Figure 3 can then be achieved using the following equations: then use the configuration shown in Figure 11. VSUPPLY + RIN VSENSE – LT6109-1 1 SENSELO + LOAD V ISENSE = SENSE RSENSE VPULLUP RC – V– V+ 2 EN/RST C1 OUTA 8 V+ 3 OUTC2 CLC IOUTA CL INC2 7 400mV REFERENCE V+ V– R2 – 5 VOUT + 4 OUTC1 CLC R3 V– RC OVERCURRENT FLAG 9 V+ – UNDERCURRENT FLAG VRESET SENSEHI 10 + RSENSE V– INC1 6 R1 610912 F11 Figure 11. Typical Configuration with Alternative ROUT Configuration 610912fa 17 LT6109-1/LT6109-2 APPLICATIONS INFORMATION The desired trip points and full-scale analog output voltage for the circuit in Figure 13 can be achieved as follows: R1= R2 = R3 = OUTC1 (LT6109-1/LT6109-2) OUTC2 (LT6109-2) 400mV OUTC2 (LT6109-1) IOVER INCREASING VINC1,2 VHYS VHYS VMAX – IMAX (R1) 610912 F12 VTH IMAX 400mV – IUNDER (R1+ R2 ) Figure 12. Comparator Output Transfer Characteristics IUNDER ing input thresholds, VTH (the actual internal threshold remains unaffected). Trip points for the LT6109-2 can be set by replacing IUNDER with a second overcurrent, IOVER2. Figure 13 shows how to add additional hysteresis to a noninverting comparator. Hysteresis R6 can be calculated from the extra hysteresis being added, VHYS(EXTRA) and the amplifier output current which you want to cause the comparator output to trip, IUNDER. Note that the hysteresis being added, VHYS(EXTRA), is in addition to the typical 10mV of built-in hysteresis. Each comparator has a typical built-in hysteresis of 10mV to simplify design, ensure stable operation in the presence of noise at the inputs, and to reject supply noise that might be induced by state change load transients. The hysteresis is designed such that the threshold voltage is altered when the output is transitioning from low to high as is shown in Figure 12. R6 = External positive feedback circuitry can be employed to increase the effective hysteresis if desired, but such circuitry will have an effect on both the rising and fall- V+ 400mV – VHYS(EXTRA) IUNDER R1 should be chosen such that R1 >> R6 so that VOUTA does not change significantly when the comparator trips. 9 V+ LT6109-1 V+ 10 SENSEHI – 1 SENSELO + RSENSE ILOAD V– V+ R3 OUTA 8 V+ R5 INC2 7 + RIN 3 OUTC2 R1 VTH R6 400mV REFERENCE – V– R2 5 610912 F13 Figure 13. Noninverting Comparator with Added Hysteresis 18 610912fa LT6109-1/LT6109-2 APPLICATIONS INFORMATION R3 should be chosen to allow sufficient VOL and comparator output rise time due to capacitive loading. R2 can be calculated: R2 = ( )( R1• V + – 400mV – VHYS(EXTRA) •R3 In the previous example, this is an error of 4.3mV at the output of the amplifier or 43µV at the input of the amplifier assuming a gain of 100. When using the comparators with their inputs decoupled from the output of the amplifier, they may be driven directly by a voltage source. It is useful to know the threshold voltage equations with the additional hysteresis. The input falling edge threshold which causes the output to transition from high to low is: ) VHYS(EXTRA) For very large values of R2 PCB related leakage may become an issue. A tee network can be implemented to reduce the required resistor values. 1 V + •R1 1 VTH(F) = 400mV •R1• + – R1 R2 +R3 R2 +R3 The approximate total hysteresis will be: V + – 400mV VHYS = 10mV +R1• R2 +R3 The input rising edge threshold which causes the output to transition from low to high is: For example, to achieve IUNDER = 100µA with 50mV of total hysteresis, R6 = 3.57k. Choosing R1 = 35.7k, R3 = 10k and V+ = 5V results in R2 = 4.12M. 1 1 VTH(R) = 410mV •R1• + R1 R2 Figure 14 shows how to add additional hysteresis to an inverting comparator. The analog output voltage will also be affected when the comparator trips due to the current injected into R6 by the positive feedback. Because of this, it is desirable to have (R1 + R2 + R3) >> R6. The maximum VOUTA error caused by this can be calculated as: R7 can be calculated from the amplifier output current which is required to cause the comparator output to trip, IOVER. R7 = R6 ∆VOUTA = V • R1+R2 +R3 +R6 + V+ 400mV , Assuming (R1+R2) >> R7 IOVER 9 V+ LT6109-1 V+ 10 SENSEHI – 1 SENSELO + RSENSE ILOAD V– V+ V+ R6 INC1 6 4 OUTC1 R1 VTH R7 400mV REFERENCE + R3 OUTA 8 – RIN V– VDD 5 R2 610912 F14 Figure 14. Inverting Comparator with Added Hysteresis 610912fa 19 LT6109-1/LT6109-2 APPLICATIONS INFORMATION To ensure (R1 + R2) >> R7, R1 should be chosen such that R1 >> R7 so that VOUTA does not change significantly when the comparator trips. R3 should be chosen to allow sufficient VOL and comparator output rise time due to capacitive loading. R2 can be calculated: V – 390mV R2 = R1• DD VHYS(EXTRA) Note that the hysteresis being added, VHYS(EXTRA), is in addition to the typical 10mV of built-in hysteresis. For very large values of R2 PCB related leakage may become an issue. A tee network can be implemented to reduce the required resistor values. The approximate total hysteresis is: V – 390mV VHYS = 10mV +R1• DD R2 For example, to achieve IOVER = 900µA with 50mV of total hysteresis, R7 = 442Ω. Choosing R1 = 4.42k, R3 = 10k and VDD = 5V results in R2 = 513k. The analog output voltage will also be affected when the comparator trips due to the current injected into R7 by the positive feedback. Because of this, it is desirable to have (R1 + R2) >> R7. The maximum VOUTA error caused by this can be calculated as: R7 ∆VOUTA = VDD • R1+R2+R7 In the previous example, this is an error of 4.3mV at the output of the amplifier or 43µV at the input of the amplifier assuming a gain of 100. When using the comparators with their inputs decoupled from the output of the amplifier they may be driven directly by a voltage source. It is useful to know the threshold voltage equations with additional hysteresis. The input rising edge threshold which causes the output to transition from high to low is: R1 VTH(R) = 400mV • 1+ R2 The input falling edge threshold which causes the output to transition from low to high is: R1 R1 VTH(F) = 390mV • 1+ – VDD R2 R2 Comparator Outputs The comparator outputs can maintain a logic low level of 150mV while sinking 500µA. The outputs can sink higher currents at elevated VOL levels as shown in the Typical Performance Characteristics. Load currents are conducted to the V– pin. The output off-state voltage may range between 0V and 60V with respect to V–, regardless of the supply voltage used. As with any open-drain device, the outputs may be tied together to implement wire-OR logic functions. The LT6109-1 can be used as a single-output window comparator in this way. EN/RST Pin The EN/RST pin performs the two functions of resetting the latch on the comparators as well as shutting down the LT6109. After powering on the LT6109, the comparators must be reset in order to guarantee a valid state at their outputs. Applying a pulse to the EN/RST pin will reset the comparators from their tripped state as long as the input on the comparator is below the threshold and hysteresis for an inverting comparator or above the threshold and hysteresis for a noninverting comparator. For example, if VINC1 is pulled higher than 400mV and latches the comparator, a reset pulse will not reset that comparator unless its input is held below the threshold by a voltage greater than the 10mV typical hysteresis. The comparator outputs typically unlatch in 0.5µs with 2pF of capacitive load. Increased capacitive loading will cause increased unlatch time. Figure 15 shows the reset functionality of the EN/RST pin. The width of the pulse applied to reset the comparators must be greater than tRPW(MIN) (2µs) but less than tRPW(MAX) (15µs). Applying a pulse that is longer than 40µs typically (or tying the pin low) will cause the part to enter shutdown. Once the part has entered shutdown, the supply current will be reduced to 3µA typically and the amplifier, comparators and reference will cease to function 610912fa 20 LT6109-1/LT6109-2 APPLICATIONS INFORMATION until the EN/RST pin is transitioned high. When the part is disabled, both the amplifier and comparator outputs are high impedance. on VOUTA. Circuitry connected to OUTA can be protected from these transients by using an external diode to clamp VOUTA or a capacitor to filter VOUTA. When the EN/RST pin is transitioned from low to high to enable the part, the amplifier output PMOS can turn on momentarily causing typically 1mA of current to flow into the SENSEHI pin and out of the OUTA pin. Once the amplifier is fully on, the output will go to the correct current. Figure 16 shows this behavior and the impact it has Power Up RESET PULSE WIDTH LIMITS EN/RST tRPW(MIN) 2µs COMPARATOR RESET Reverse-Supply Protection tRPW(MAX) 15µs OUTC1 OUTC2 610912 F15 tRESET 0.5µs (TYPICAL) Figure 15. Comparator Reset Functionality V+ = 60V RIN = 100Ω ROUT = 10k VEN/RST 2V/DIV 0V VOUTA 2V/DIV 0V 50µs/DIV 610912 F14 Figure 16. Amplifier Enable Response After powering on the LT6109, the comparators must be reset in order to guarantee a valid state at their outputs. Fast supply ramps may cause a supply current transient during start-up as shown in the Typical Performance Characteristics. This current can be lowered by reducing the edge speed of the supply. The LT6109 is not protected internally from external reversal of supply polarity. To prevent damage that may occur during this condition, a Schottky diode should be added in series with V— (Figure 17). This will limit the reverse current through the LT6109. Note that this diode will limit the low voltage operation of the LT6109 by effectively reducing the supply voltage to the part by VD. Also note that the comparator reference, comparator output and EN/RST input are referenced to the V– pin. In order to preserve the precision of the reference and to avoid driving the comparator inputs below V–, R2 must connect to the V– pin. This will shift the amplifier output voltage up by VD. VOUTA can be accurately measured differentially across R1 and R2. The comparator output low voltage will also be shifted up by VD. The EN/RST pin threshold is referenced to the V– pin. In order to provide valid input levels to the LT6109 and avoid driving EN/RST below V– the negative supply of the driving circuit should be tied to V–. 610912fa 21 LT6109-1/LT6109-2 APPLICATIONS INFORMATION V+ 9 V+ LT6109-1 V+ RIN 10 SENSEHI – 1 SENSELO + RSENSE VDD OUTA 8 R3 + V+ V– R1 4 OUTC INC 6 – VDD ILOAD VOUTA R2 400mV REFERENCE + VDD 2 EN/RST – V– 5 + 610912 F17 VD – Figure 17. Schottky Prevents Damage During Supply Reversal TYPICAL APPLICATIONS Overcurrent and Undervoltage Battery Fault Protection 12 LITHIUM 40V CELL STACK IRF9640 0.1Ω + 10µF + + 1M 0.1µF 100k R10 100Ω INC2 10 13.3k 9 5V + 10k SENSEHI SENSELO V+ OUTA LT6109-1 RESET 2 4 3 EN/RST INC1 OUTC1 OUTC2 V– 5 INC2 TO LOAD 6.2V* 1 8 0.8A OVERCURRENT 6 DETECTION 7 VOUT 9.53k 100k 475Ω 2N7000 30V UNDERVOLTAGE DETECTION 6109 TA02 *CMH25234B The comparators monitor for overcurrent and undervoltage conditions. If either fault condition is detected the battery will immediately be disconnected from the load. The latching comparator outputs ensure the battery stays disconnected from the load until an outside source resets the LT6109 comparator outputs. 610912fa 22 LT6109-1/LT6109-2 TYPICAL APPLICATIONS MCU Interfacing with Hardware Interupts 0.1Ω V+ Example: TO LOAD 5V 100Ω 10 9 AtMega1280 5 PB0 6 PB1 7 PCINT2 2 PCINT3 3 ADC2 1 PB5 SENSEHI SENSELO V+ 5V 8 LT6109-1 10k RESET 2 3 4 5V VOUT/ADC IN OUTA EN/RST INC2 OUTC2 OUTC1 10k V– INC1 7 6 5 OUTC2 GOES LOW 0V 1 VOUT ADC IN MCU INTERUPT 2k 6.65k UNDERCURRENT ROUTINE 1.33k RESET COMPARATORS 6109 TA03 The comparators are set to have a 50mA undercurrent threshold and a 300mA overcurrent threshold. The MCU 610912 TA03b will receive the comparator outputs as hardware interrupts and immediately run an appropriate fault routine. Simplified DC Motor Torque Control VMOTOR 100µF 1k SENSEHI SENSELO V+ OUTA LT6109 RESET EN/RST CURRENT SET POINT (0V TO 5V) VOUT 0.47µF 100k 5.62k 1µF 5V INC2 3.4k OUTC2 OUTC1 0.1Ω V– INC1 2 3 1k – + 4 7 6 1 LTC6246 78.7k BRUSHED DC MOTOR (0A TO 5A) MABUCHI RS-540SH 5 V+ 6 MOD OUT LTC6992-1 3 1N5818 SET DIV GND 2 4 IRF640 100k 1M 280k 5V 610912 TA04 The figure shows a simplified DC motor control circuit. The circuit controls motor current, which is proportional to motor torque; the LT6109 is used to provide current feedback to a difference amplifier that controls the current in the motor. The LTC®6992 is used to convert the output of the difference amp to the motors PWM control signal. 610912fa 23 LT6109-1/LT6109-2 TYPICAL APPLICATIONS Power-On Reset or Disconnect Using a TimerBlox® Circuit 5V 9 V+ LT6109-1 RIN 100Ω 10 RSENSE 1 SENSEHI – SENSELO + ILOAD R5 10k V+ OUTA 8 – V V+ 3 R1 8.06k INC2 7 + OUTC2 – 5V R8 30k Q1 2N2222 OPTIONAL: DISCHARGES C1 WHEN SUPPLY IS DISCONNECTED V– R4 10k CREATES A DELAYED C1 10µs RESET PULSE 0.1µF ON START-UP R7 1M TRIG OUT LTC6993-3 GND V+ SET 400mV REFERENCE V+ 4 OUTC1 2 EN/RST R2 1.5k + – INC1 6 R3 499Ω V– 5 610912 TA06 DIV R6 487k The LTC6993-1 provides a 10µS reset pulse to the LT6109‑1. The reset pulse is delayed by R7 and C1 whose time constant must be greater than 10ms and longer than the supply turn-on time. Optional components R8 and Q1 discharge capacitor C1 when the supply and/or ground are disconnected. This ensures that when the power supply and/or ground are restored, capacitor C1 can fully recharge and trigger the LTC6993-3 to produce another comparator reset pulse. These optional components are particularly useful if the power and/or ground connections are intermittent, as can occur when PCB are plugged into a connector. 610912fa 24 LT6109-1/LT6109-2 TYPICAL APPLICATIONS Precision Power-On Reset Using a TimerBlox® Circuit 5V 9 V+ LT6109-1 RIN 100Ω 10 RSENSE 1 SENSEHI – SENSELO + ILOAD R5 10k V+ OUTA 8 V– V+ 3 R1 8.06k INC2 + OUTC2 7 – V– R4 10k R8 100k C1 0.1µF 1 SECOND DELAY ON START-UP 10µs RESET PULSE GENERATOR TRIG OUT LTC6994-1 GND V+ TRIG OUT LTC6993-1 GND V+ SET R7 191k DIV R6 1M R5 681k SET DIV C2 0.1µF 400mV REFERENCE V+ 4 OUTC1 2 EN/RST R2 1.5k + – INC1 6 R3 499Ω V– 5 610912 TA07 R4 487k 610912fa 25 LT6109-1/LT6109-2 PACKAGE DESCRIPTION MS Package 10-Lead Plastic MSOP (Reference LTC DWG # 05-08-1661 Rev E) 0.889 ± 0.127 (.035 ± .005) 5.23 (.206) MIN 3.20 – 3.45 (.126 – .136) 3.00 ± 0.102 (.118 ± .004) (NOTE 3) 0.50 0.305 ± 0.038 (.0197) (.0120 ± .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 10 9 8 7 6 3.00 ± 0.102 (.118 ± .004) (NOTE 4) 4.90 ± 0.152 (.193 ± .006) DETAIL “A” 0.497 ± 0.076 (.0196 ± .003) REF 0° – 6° TYP GAUGE PLANE 1 2 3 4 5 0.53 ± 0.152 (.021 ± .006) DETAIL “A” 0.18 (.007) SEATING PLANE 0.86 (.034) REF 1.10 (.043) MAX 0.17 – 0.27 (.007 – .011) TYP 0.50 (.0197) BSC NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 0.1016 ± 0.0508 (.004 ± .002) MSOP (MS) 0307 REV E 610912fa 26 LT6109-1/LT6109-2 REVISION HISTORY REV DATE DESCRIPTION PAGE NUMBER A 12/12 Addition of A-grade Performance and Electrical Characteristics 1, 3, 4, 11, 13, 15 (Fig10), 28 Correction to Typical Application diagram 1 Addition of A-grade Order Information 2 Clarification to Absolute Maximum Short Circuit Duration 2 Edits to Electrical Characteristics conditions and notes 3, 4 Clarification to nomenclature used in Typical Performance Characteristics 5-8 Clarification to Description of Pin Functions Internal Reference Block redrawn for consistency Edits to Applications Information Addition of LT6108 to Related Parts 8, 9 9, 10, 12, 17, 18, 19, 25, 26 10-16, 18, 20-25 28 610912fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 27 LT6109-1/LT6109-2 TYPICAL APPLICATION ADC Driving Application IN SENSE HIGH SENSE LOW 0.1Ω 0.1µF OUT VCC VREF 100Ω 10 9 SENSEHI SENSELO V+ VCC VCC 10k OUTA 8 LT6109-1 RESET 2 3 10k 4 EN/RST INC2 OUTC2 OUTC1 V– INC1 COMP 1 7 6 5 IN+ 2k LTC2470 TO MCU 0.1µF 6.65k 1.33k OVERCURRENT 6109 TA05 UNDERCURRENT The low sampling current of the LTC2470 16-bit delta sigma ADC is ideal for the LT6109. RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1787 Bidirectional High Side Current Sense Amplifier 2.7V to 60V, 75µV Offset, 60µA Quiescent, 8V/V Gain LTC4150 Coulomb Counter/Battery Gas Gauge Indicates Charge Quantity and Polarity LT6100 Gain-Selectable High Side Current Sense Amplifier 4.1V to 48V, Gain Settings: 10, 12.5, 20, 25, 40, 50V/V LTC6101 High Voltage High Side Current Sense Amplifier Up to 100V, Resistor Set Gain, 300µV Offset, SOT-23 LTC6102 Zero Drift High Side Current Sense Amplifier Up to 100V, Resistor Set Gain, 10µV Offset, MSOP8/DFN LTC6103 Dual High Side Current Sense Amplifier 4V to 60V, Resistor Set Gain, 2 Independent Amps, MSOP8 LTC6104 Bidirectional High Side Current Sense Amplifier 4V to 60V, Separate Gain Control for Each Direction, MSOP8 LT6105 Precision Rail-to-Rail Input Current Sense Amplifer –0.3V to 44V Input Range, 300µV Offset, 1% Gain Error LT6106 Low Cost High Side Current Sense Amplifier 2.7V to 36V, 250µV Offset, Resistor Set Gain, SOT-23 LT6107 High Temperature High Side Current Sense Amplifier 2.7V to 36V, –55°C to 150°C, Fully Tested: –55°C, 25°C, 150°C LT6108 High Side Current Sense Amplifier with Reference and Comparator 2.7V to 60V, 125µV Offset, Resistor Set Gain, ±1.25% Threshold Error LT6700 Dual Comparator with 400mV Reference 1.4V to 18V, 6.5µA Supply Current 610912fa 28 Linear Technology Corporation LT 1212 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2011