TI1 OPA691IDBV Ultra-wideband, current-feedback operational amplifier with disable Datasheet

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OPA695
SBOS293H – DECEMBER 2003 – REVISED DECEMBER 2015
OPA695 Ultra-Wideband, Current-Feedback Operational Amplifier With Disable
1 Features
3 Description
•
•
•
•
•
•
•
•
The OPA695 is a high bandwidth, current-feedback
operational amplifier that combines an exceptional
4300-V/μs slew rate and a low input voltage noise to
deliver a precision, low-cost, high dynamic range
intermediate frequency (IF) amplifier. Optimized for
high gain operation, the OPA695 is ideally suited to
buffering surface acoustic wave (SAW) filters in an IF
strip, or delivering high output power at low distortion
for cable-modem upstream line drivers. At lower
gains, a higher bandwidth of 1400 MHz is achievable,
making the OPA695 an excellent video line driver for
supporting high-resolution RGB applications.
1
Gain = +2 Bandwidth (1400 MHz)
Gain = +8 Bandwidth (450 MHz)
Output Voltage Swing: ±4.2 V
Ultra-High Slew Rate: 4300 V/μs
3RD-Order Intercept: > 40 dBm (f < 50 MHz)
Low Power: 129 mW
Low Disabled Power: 0.5 mW
Packages: SOIC-8, VSSOP-8, SOT23-6
2 Applications
•
•
•
•
•
•
•
The OPA695 low 12.9-mA supply current is precisely
trimmed at +25°C. This trim, along with a low
temperature drift, gives low system power over
temperature. System power may be further reduced
using the optional disable control pin. Leaving this pin
open, or holding it HIGH, gives normal operation. If
pulled LOW, the OPA695 supply current drops to less
than 170 μA. This power-saving feature, along with
exceptional single +5-V operation and ultra-small
SOT23-6 packaging, make the OPA695 ideal for
portable applications.
Very Wideband ADC Drivers
Low-Cost Precision IF Amplifiers
Broadband Video Line Drivers
Portable Instruments
Active Filters
ARB Waveform Output Drivers
OPA685 Performance Upgrades
Device Information(1)
PART NUMBER
OPA695
PACKAGE
BODY SIZE (NOM)
SOT-23 (6)
1.60 mm × 2.90 mm
VSSOP (8)
3.00 mm × 3.00 mm
SOIC (8)
3.91 mm × 4.90 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Gain 2V/V Video Line Driver
Gain of +2V/V Video Line Driver Pulse Response
+5V
1.2
125MHz Input
Voltage at
Matched Load
VIN
75Ω
VLOAD
RG-59
OPA695
75Ω
75Ω
511Ω
511Ω
−5V
Input/Load Voltage (V)
1
0.8
0.6
0.4
0.2
0
−0.2
Time (1ns/div)
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
OPA695
SBOS293H – DECEMBER 2003 – REVISED DECEMBER 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
6.1
6.2
6.3
6.4
6.5
6.6
1
1
1
2
3
3
Absolute Maximum Ratings ...................................... 3
ESD Ratings ............................................................ 4
Recommended Operating Conditions....................... 4
Thermal Information .................................................. 4
Electrical Characteristics........................................... 5
Typical Characteristics ............................................ 12
7
Parameter Measurement Information ................ 20
8
Detailed Description ............................................ 21
7.1 Differential Small Signal Measurement................... 20
8.1 Overview ................................................................. 21
8.2 Functional Block Diagram ....................................... 21
8.3 Feature Description................................................. 21
8.4 Device Functional Modes........................................ 28
9
Application and Implementation ........................ 29
9.1 Application Information............................................ 29
9.2 Typical Application ................................................. 42
10 Power Supply Recommendations ..................... 43
11 Layout................................................................... 44
11.1 Layout Guidelines ................................................. 44
11.2 Layout Example .................................................... 45
12 Device and Documentation Support ................. 46
12.1
12.2
12.3
12.4
12.5
12.6
Device Support......................................................
Documentation Support .......................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
46
46
46
46
46
46
13 Mechanical, Packaging, and Orderable
Information ........................................................... 47
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision G (April 2009) to Revision H
Page
•
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section. ................................................................................................ 1
•
Removed lead temperature ................................................................................................................................................... 3
Changes from Revision F (July 2006) to Revision G
•
Added DGK (MSOP-8) package to Package Ordering Information table and to Thermal Resistance specification in
the Electrical Characteristics tables........................................................................................................................................ 1
Changes from Revision E (March 2006) to Revision F
•
2
Page
Page
Changed Storage Temperature Range from −40°C to +125C to −65°C to +125C. .............................................................. 3
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SBOS293H – DECEMBER 2003 – REVISED DECEMBER 2015
5 Pin Configuration and Functions
DGK or D Package
8-Pin VSSOP or SOIC
Top View
DBV Package
6-Pin SOT-23
Top View
NC
1
8
DIS
Inverting Input
2
7
+VS
Noninverting Input
3
6
Output
–VS
4
5
NC
Output
1
6
+VS
–VS
2
5
DIS
Noninverting Input
3
4
Inverting Input
NC = No Connection
6
5
4
A71L
1
2
3
Pin Orientation/Package Marking
Pin Functions
PIN
NAME
I/O
DESCRIPTION
VSSOP, SOIC NO.
SOT-23 NO.
1, 5
—
NC
—
Not connected
2
4
Inverting input
I
Inverting input
3
3
Non-inverting input
I
Non-inverting input
4
2
–VS
P
Negative supply
6
1
Output
O
Output
7
6
+VS
P
Positive supply
8
5
DIS
I
Not disable (Enable)
6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
(1)
MIN
Power supply
Internal power dissipation
MAX
UNIT
±6.5
V
See Thermal Analysis
Differential input voltage
±1.2
Input common-mode voltage
±VS
V
TJ
Junction temperature
150
°C
Tstg
Storage temperature; D, DBV
125
°C
(1)
–65
V
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
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OPA695
SBOS293H – DECEMBER 2003 – REVISED DECEMBER 2015
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6.2 ESD Ratings
VALUE
UNIT
OPA695 in DGK or D package
Human-body model (HBM), per
ANSI/ESDA/JEDEC JS-001 (1)
V(ESD)
Electrostatic discharge
All pins except pin 2
±1500
Pin 2
±500
Charged-device model (CDM), per JEDEC
All pins
specification JESD22-C101 (2)
±1000
Machine Model (MM)
All pins
±100
Human-body model (HBM), per
ANSI/ESDA/JEDEC JS-001 (1)
All pins except pin 4
±1500
Pin 4
±500
V
OPA695 in DBV package
V(ESD)
(1)
(2)
Electrostatic discharge
Charged-device model (CDM), per JEDEC
All pins
specification JESD22-C101 (2)
±1000
Machine Model (MM)
±100
All pins
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. Manufacturing with
less than 500-V HBM is possible if necessary precautions are taken.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. Manufacturing with
less than 250-V CDM is possible if necessary precautions are taken.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
NOM
MAX
±2.5
±5
±6
V
Single supply voltage
5
10
12
V
Ambient temperature
–40
25
85
°C
VS
Split supply voltage
VS
TA
UNIT
6.4 Thermal Information
OPA695
THERMAL METRIC
(1)
D (SOIC)
DGK (VSSOP) DBV (SOT-23)
UNIT
8 PINS
8 PINS
6 PINS
150
°C/W
RθJA
Junction-to-ambient thermal resistance
125
135
RθJC(top)
Junction-to-case (top) thermal resistance
63
81
108
°C/W
RθJB
Junction-to-board thermal resistance
58
56
26.4
°C/W
ψJT
Junction-to-top characterization parameter
12
8.5
15
°C/W
ψJB
Junction-to-board characterization parameter
57
48
26
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
—
—
—
—
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
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SBOS293H – DECEMBER 2003 – REVISED DECEMBER 2015
6.5 Electrical Characteristics
RF = 348 Ω, RL = 100 Ω to VS/2, and G = +8, (see Figure 50 for AC performance only), unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEST
LEVEL
(1)
AC PERFORMANCE (see Gain 2V/V Video Line Driver)
Small-signal bandwidth
(VO = 0.5 VPP)
G = +1, RF = 523 Ω
25°C
G = +2, RF = 511 Ω
25°C
G = +8, RF = 402 Ω
25°C (2)
400
0°C to 70°C (3)
380
–40°C to +85°C
G = +16, RF = 249 Ω
Bandwidth for 0.2-dB gain
flatness
(3)
1400
C
450
G = +8, VO = 4 VPP
4.6
0°C to 70°C (3)
Slew Rate
G = +8, VO = 4-V Step
Rise-and-fall time
Settling time
to 0.02%
to 0.1%
450
3700
0°C to 70°C (3)
3600
–40°C to +85°C (3)
3500
25°C
2600
0°C to 70°C (3)
2500
–40°C to +85°C (3)
2400
G = +8, VO = 0.5-V Step
25°C
0.8
G = +8, VO = 4-V Step
25°C
1
G = +8, VO = 2-V Step
RL = 100 Ω
2ndharmonic
25°C
10
Harmonic
distortion (G = +8,
f = 10 MHz, VO = 2
VPP)
–60
–59
3rdharmonic
–74
–73
–86
–75
–40°C to +85°C (3)
–72
–86
0°C to 70°C (3)
25°C (2)
f > 1 MHz
0°C to 70°C
(3)
Noninverting input current noise
f > 1 MHz
0°C to 70°C
(1)
(2)
(3)
ns
C
ns
C
dBc
B
–82
2
B
2.9
18
–40°C to +85°C (3)
B
2.7 nV/√Hz
–40°C to +85°C (3)
(3)
V/μs
–80
1.8
25°C (2)
c
–81
–40°C to +85°C (3)
Input voltage noise
–84
0°C to 70°C (3)
(2)
MHz
–76
–40°C to +85°C (3)
25°C
RL ≥ 500 Ω
–78
0°C to 70°C (3)
(2)
B
–62
–40°C to +85°C (3)
25°C
RL = 100 Ω
–65
0°C to 70°C (3)
25°C
RL ≥ 500 Ω
2900
16
(2)
dB
4300
25°C
25°C (2)
B
6
25°C (2)
(2)
MHZ
5.4
5.8
25°C
G = –8, VO = 4-V Step
B
C
320
–40°C to +85°C (3)
Large-signal bandwidth
MHz
350
G = +2, VO = 0.5 VPP, RF =523 Ω 25°C
RF = 523 Ω, VO = 0.5 VPP
C
350
25°C
25°C (2)
Peaking at a gain of +1
1700
19
21 nV/√Hz
B
22
Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
Junction temperature = ambient for +25°C specifications.
Junction temperature = ambient at low temperature limit; junction temperature = ambient +15°C at high temperature limit for over
temperature specifications.
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SBOS293H – DECEMBER 2003 – REVISED DECEMBER 2015
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Electrical Characteristics (continued)
RF = 348 Ω, RL = 100 Ω to VS/2, and G = +8, (see Figure 50 for AC performance only), unless otherwise noted.
PARAMETER
TEST CONDITIONS
25°C
Inverting input current noise
MIN
(2)
TYP
MAX
22
24
0°C to 70°C (3)
f > 1 MHz
–40°C to +85°C
26 pA/√Hz
(3)
TEST
LEVEL
(1)
B
27
Differential gain
G = +2, NTSC, VO = 1.4 Vp,
RL = 150 Ω
25°C
0.04%
Differntial phase
G = +2, NTSC, VO = 1.4 Vp,
RL = 150 Ω
25°C
0.007
DC PERFORMANCE
UNIT
C
deg
deg
kΩ
A
mV
A
μV/°C
B
μA
A
µA
A
µA
A
nA/°C
B
V
A
dB
A
kΩ || pF
C
Ω
C
V
A
(4)
25°C (2)
Open-loop transimpedance gain
(ZOL)
VO = 0 V, RL = 100
0°C to 70°C
45 (5)
(3)
43
–40°C to +85°C (3)
41
25°C (2)
Input offset voltage
VCM = 0 V
0°C to 70°C
85
±0.3 ±3.0 (5)
(3)
±3.5
–40°C to +85°C (3)
Average offset voltage drift
±4
0°C to 70°C (3)
VCM = 0 V
–40°C to +85°C
±10
(3)
±15
25°C (2)
Noninverting input bias current
Average noninventing input bias
current drift
VCM = 0 V
0°C to 70°C
VCM = 0 V
±13
(3)
±37
–40°C to +85°C (3)
±41
0°C to 70°C (3)
150
–40°C to +85°C (3)
150
25°C (2)
Inverting input bias current
VCM = 0 V
0°C to 70°C
±20
(3)
±70
0°C to 70°C (3)
VCM = 0 V
–40°C to +85°C
±60 (5)
±66
–40°C to +85°C (3)
Average inventing bias current
drift
±30 (5)
±120
(3)
±160
INPUT
Common-mode input range (6)
(CMIR)
25°C (2)
±3.1 (5)
0°C to 70°C (3)
–40°C to +85°C
(3)
±3
25°C (2)
Common-mode rejection ratio
(CMRR)
VCM = 0 V
Noninverting input impedance
25°C (2)
Inverting input resistance (Rl)
Open-loop
±3.3
±3
51 (5)
0°C to 70°C (3)
–40°C to +85°C
56
50
(3)
50
280 || 1.2
25°C (2)
29
OUTPUT
25°C (2)
No load
–40°C to +85°C (3)
Voltage output swing
25°C
100-Ω load
(4)
(5)
(6)
6
0°C to 70°C
±4 (5)
(3)
(2)
±4.2
±3.9
±3.9
±3.7 (5)
0°C to 70°C (3)
±3.7
–40°C to +85°C (3)
±3.6
±3.9
Current is considered positive out-of-node. VCM is the input common-mode voltage.
Limits are tested at +25°C.
Tested < 3 dB below minimum specified CMRR at ± CMIR limits.
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Electrical Characteristics (continued)
RF = 348 Ω, RL = 100 Ω to VS/2, and G = +8, (see Figure 50 for AC performance only), unless otherwise noted.
PARAMETER
TEST CONDITIONS
25°C
Current output, sourcing
(2)
–40°C to +85°C
Closed-loop output impedance
VO = 0
G = +8, f = 100 kHz
(5)
120
MAX
80
(3)
25°C (2)
Current output, sinking
TYP
90
0°C to 70°C (3)
VO = 0
MIN
UNIT
TEST
LEVEL
mA
A
mA
A
Ω
C
µA
A
(1)
70
90 (5)
0°C to 70°C (3)
–80
–40°C to
+85°C (3)
–70
25°C
0.04
DISABLE (Disabled LOW)
25°C
Power-down supply current
(+VS)
–100
25°C (2)
VDIS = 0
–170 (5)
0°C to 70°C (3)
–186
–40°C to +85°C (3)
–192
Disable time
VIN = ±0.25 VDC
25°C
1
µs
C
Enable time
VIN = ±0.25 VDC
25°C
1
ns
C
Off isolation
G = +8, 10 MHz
25°C
70
dB
C
Output capacitance in disable
25°C
4
pF
C
Turn on glitch
G = +2, RL = 150 Ω, VIN = 0
25°C
±100
mV
C
Turn off glitch
G = +2, RL = 150 Ω, VIN = 0
25°C
±20
mV
C
V
A
V
A
µA
A
V
C
V
A
mA
A
mA
A
dB
A
25°C
Enable voltage
(2)
3.5
(5)
0°C to 70°C (3)
3.6
–40°C to +85°C (3)
3.7
25°C (2)
Disable voltage
0°C to 70°C
3.3
1.8
(3)
1.6
–40°C to +85°C (3)
1.5
25°C (2)
Control pin input bias current
(DIS)
1.7 (5)
VDIS = 0
0°C to 70°C
75
(3)
130 (5)
143
–40°C to +85°C (3)
145
POWER SUPPLY
Specified operating voltage
25°C
±5
25°C (2)
Maximum operating voltage
range
±6 (5)
0°C to 70°C (3)
–40°C to +85°C
±6
(3)
±6
25°C
Maximum quiescent
VS = ±5 V
12.9
25°C (2)
13.3 (5)
0°C to 70°C (3)
13.7
–40°C to +85°C (3)
14.1
25°C
Minimum quiescent current
VS = ±5 V
12.9
25°C (2)
0°C to 70°C
12.6 (5)
(3)
–40°C to +85°C (3)
11.8
11
25°C
Power-supply rejection radio
(–PSRR)
Input referred
25°C
55
(2)
51
(5)
0°C to 70°C (3)
48
–40°C to +85°C (3)
48
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Electrical Characteristics (continued)
RF = 348 Ω, RL = 100 Ω to VS/2, and G = +8, (see Figure 50 for AC performance only), unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEST
LEVEL
°C
C
(1)
TEMPERATURE RANGE
Specification: ID, IDBV
–40 to
85
25°C
AC PERFORMANCE (see Figure 50)
Small-signal bandwidth
(VO = 0.5 VPP)
G = +1, RF = 511 Ω
25°C
G = +2, RF = 487 Ω
25°C
G = +8, RF = 348 Ω
G = +16, RF = 162 Ω
Bandwith for 0.2-dB gain flatness G = +2, VO <0.5 VPP, RF = 487Ω
1400
960
VO <0.5 VPP, RF = 511 Ω
C
25°C
395
25°C (2)
980
0°C to 70°C (3)
330
–40°C to +85°C (3)
300
25°C
235
25°C
230
25°C (2)
180
0°C to 70°C (3)
135
–40°C to +85°C (3)
110
25°C
Peaking at a gain of +1
Slew rate
G = +8, VO = 2 VPP
2.5
Settling time
310
25°C
1700
0°C to 70°C
1300
(3)
1200
25°C
1
G = +8, VO = 2-V Step
25°C
1
to 0.02%
G = +8, VO = 2-V Step
25°C
16
to 0.1%
G = +8, VO = 2-V Step
25°C
10
25°C
2ndHarmonic
25°C
Harmonic
distortion
(G = +8, f = 10
MHz, VO = 2 VPP)
–58
–40°C to +85°C (3)
–57
v/μs
B
ns
C
ns
C
dBc
B
–66
(3)
–65
–66
25°C (2)
–64
0°C to 70°C (3)
–64
(3)
25°C
25°C (2)
0°C to 70°C
–63
–65
–63
(3)
–40°C to +85°C (3)
8
C
–66
0°C to 70°C (3)
–40°C to +85°C
RL ≥ 500 Ω to VS
MHz
–70
25°C (2)
25°C
3rdHarmonic
B
–58
0°C to 70°C (3)
–40°C to +85°C
RL = 100 Ω to VS
dB
–62
(2)
25°C
RL ≥ 500 Ω to VS/2
B
1100
G = +8, VO = 0.5-V Step
RL = 100 Ω to VS/2
MHz
3
25°C
–40°C to +85°C (3)
Rise-and-fall-time
B
C
2
0°C to 70°C (3)
25°C (2)
G = +8, 2-V Step
MHz
1
25°C (2)
–40°C to +85°C (3)
Large-signal bandwidth
C
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Electrical Characteristics (continued)
RF = 348 Ω, RL = 100 Ω to VS/2, and G = +8, (see Figure 50 for AC performance only), unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
25°C
Input voltage noise
TYP
0°C to 70°C
2
(3)
2.7
–40°C to +85°C (3)
f > 1 MHz
DC PERFORMANCE
f > 1 MHz
nV/√Hz
B
nV/√Hz
B
pA/√Hz
B
kΩ
A
mV
A
μV/°C
B
µA
A
nA/°C
B
nA/°C
B
V
A
V
A
(1)
18
25°C (2)
19
0°C to 70°C (3)
21
–40°C to +85°C (3)
22
25°C
Inverting input current noise
TEST
LEVEL
2.9
25°C
Noninverting input current noise
UNIT
1.8
25°C (2)
f > 1 MHz
MAX
22
25°C (2)
24
0°C to 70°C (3)
26
–40°C to +85°C (3)
27
(4)
25°C
Open-loop transimpedance gain
(ZOL)
VO = VS/2, RL = 100 Ω to VS/2
70
25°C (2)
40
0°C to 70°C (3)
38
–40°C to +85°C (3)
38
25°C
Input offset voltage
25°C
VCM = VS/2
±0.3
(2)
0°C to 70°C
±3 (5)
(3)
±3.5
–40°C to +85°C (3)
Average offset voltage drift
VCM = VS
±4
0°C to 70°C (3)
±10
–40°C to +85°C (3)
±15
25°C
Noninventing input bias current
±5
25°C (2)
VCM = VS
0°C to 70°C
±40 (5)
(3)
±45
–40°C to +85°C (3)
Average noninventing input bias
current drift
VCM = VS
±50
0°C to 70°C (3)
±110
–40°C to +85°C (3)
±170
25°C
Inverting input bias current
VCM = VS
±60 (5)
0°C to 70°C (3)
–40°C to +85°C
Average inverting input bias
current drift
±5
25°C (2)
VCM = VS
±66
(3)
±70
0°C to 70°C (3)
±120
–40°C to +85°C (3)
±160
INPUT
25°C
Least positive input voltage (6)
1.7
25°C (2)
1.8 (5)
0°C to 70°C (3)
1.9
–40°C to +85°C (3)
1.9
25°C
Most positive input voltage (6)
3.3
25°C (2)
0°C to 70°C
3.2 (5)
(3)
3.1
–40°C to +85°C (3)
3.1
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Electrical Characteristics (continued)
RF = 348 Ω, RL = 100 Ω to VS/2, and G = +8, (see Figure 50 for AC performance only), unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
25°C
Common-mode rejection ratio
(CMRR)
25°C
Inverting input resistance (Rl)
Open-loop
dB
A
kΩ || pF
C
Ω
C
V
A
V
A
mA
A
mA
A
Ω
C
µA
C
µs
C
25
ns
C
70
dB
C
4
pF
C
51 (5)
0°C to 70°C (3)
50
–40°C to +85°C (3)
Noinverting input impedance
TEST
LEVEL
MAX
(1)
54
25°C (2)
VCM = VS/2
UNIT
TYP
50
280 || 1.2
25°C
32
OUTPUT
25°C
25°C
No load
Most positive output voltage
4.2
(2)
4.0
(5)
0°C to 70°C (3)
3.9
–40°C to +85°C (3)
3.8
25°C
4
25°C (2)
RL = 100 Ω to VS/2
0°C to 70°C
3.9 (5)
(3)
3.8
–40°C to +85°C (3)
3.7
25°C
0.8
25°C (2)
No load
0°C to 70°C
1 (5)
(3)
–40°C to +85°C
Least positive output voltage
1.1
(3)
1.2
25°C
1
25°C (2)
RL = 100 Ω to VS/2
1.1 (5)
0°C to 70°C (3)
–40°C to +85°C
1.2
(3)
1.3
25°C
Current output, sourcing
90
25°C (2)
VO = VS/2
0°C to 70°C
70 (5)
(3)
–40°C to +85°C (3)
67
66
25°C
Current output, sinking
Closed-loop output impedance
25°C
VO = VS/2
–90
(2)
–70
(5)
0°C to 70°C (3)
–67
–40°C to +85°C (3)
–66
G = +2, f = 100 kHz
0.05
DISABLE (Disabled LOW)
25°C
–95
25°C (2)
Power down supply current (+VS) VDIS = 0
0°C to 70°C
–160
(3)
–40°C to +85°C (3)
–175
–180
Disable time
25°C
Enable time
25°C
Off isolation
G = +8, 10 MHz
Output capacitance in disable
25°C
Tun on glitch
G = +2, RL = 150 Ω, VIN = VS /2
25°C
±100
mV
C
Turn off glitch
G = +2, RL = 150 Ω, VIN = VS /2
25°C
±20
mV
C
10
1
25°C
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SBOS293H – DECEMBER 2003 – REVISED DECEMBER 2015
Electrical Characteristics (continued)
RF = 348 Ω, RL = 100 Ω to VS/2, and G = +8, (see Figure 50 for AC performance only), unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
25°C
Enable voltage
V
A
V
A
µA
C
V
C
V
A
mA
A
mA
A
56
dB
A
–40 to
+85
°C
°C
3.5 (5)
0°C to 70°C (3)
3.6
–40°C to +85°C (3)
3.7
25°C
TEST
LEVEL
(1)
1.8
(2)
0°C to 70°C
1.7 (5)
(3)
1.6
–40°C to +85°C (3)
1.5
25°C
Control pin input bias current
(DIS)
MAX
3.3
25°C (2)
25°C
Disable voltage
UNIT
TYP
VDIS = 0
75
25°C (2)
130
0°C to 70°C (3)
143
–40°C to +85°C (3)
149
POWER SUPPLY
Specified single-supply operating
25°C
voltage
5
25°C (2)
Max single-supply operating
voltage
0°C to 70°C
12 (5)
(3)
12
–40°C to +85°C (3)
12
25°C
Max quiescent current
VS = +5 V
25°C
11.4
(2)
0°C to 70°C
12 (5)
(3)
12.5
–40°C to +85°C (3)
12.9
25°C
Min quiescent current
Power-supply rejection ratio
(–PSRR)
VS = +5 V
Input referred
25°C (2)
11.4
10.9 (5)
0°C to 70°C (3)
9.4
–40°C to +85°C (3)
9.1
25°C
TEMPERATURE RANGE
Specification: ID, IDBV
25°C
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6.6 Typical Characteristics
G = +8, RF = 402 Ω, RL = 100 Ω, unless otherwise noted.
6
–3
–6
G = +4, RF = 480Ω
–12
G = +8, RF = 402Ω
–15
–18
G = –2, RF = 499Ω
0
–6
G = –8, RF = 442Ω
–9
–12
–15
G = –16, RF = 806Ω
–18
–21
–24
–24
0
200
400
600
800
1000
1200
1400
0
Frequency (MHz)
24
800
1000
1200
1400
G = –8, RF = 442Ω
21
VO = 2VPP
18
12
Gain (3dB/div)
VO = 1VPP
and 2VPP
15
Gain (3dB/div)
600
Figure 2. Inverting Small-Signal Frequency Response
18
9
6
VO = 4VPP
3
0
15
12
VO = 7VPP
9
VO = 4VPP
6
VO = 1VPP
3
VO = 7VPP
0
–3
–3
–6
–6
0
500MHz
1GHz
0
Frequency (100MHz/div)
3
125MHz Square Wave Input
3
G = +8, RF = 402Ω
125MHz Square Wave Input
1
1
Output Voltage
2
–1
1GHz
Figure 4. Inverting Large-Signal Frequency Response
2
0
500MHz
Frequency (100MHz/div)
Figure 3. Noninverting Large-Signal Frequency Response
Output Voltage
400
24
G = +8, RF = 402Ω
21
200
Frequency (MHz)
Figure 1. Noninverting Small-Signal Frequency Response
Small-Signal ±500mV
Large-Signal ±2V
0
G = +8, RF = 402Ω
Small-Signal ±500mV
–1
Large-Signal ±2V
–2
–2
–3
12
G = –4,
RF = 475Ω
–3
G = +16, RF = 249Ω
–21
VO = 500mVPP
3
Normalized Gain (3dB/div)
Normalized Gain (3dB/div)
0
–9
6
VO = 500mVPP
G = +2, RF = 523Ω
3
–3
Time (1ns/div)
Time (1ns/div)
Figure 5. Noninverting Large and Small-Signal Frequency
Response
Figure 6. Inverting Large and Small-Signal Frequency
Response
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Typical Characteristics (continued)
G = +8, RF = 402 Ω, RL = 100 Ω, unless otherwise noted.
–50
–55
–60
–60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
VO = 2VPP
G = 8V/V
2nd-Harmonic
–70
–80
3rd-Harmonic
–90
VO = 2VPP, G = 8V/V
RL = 100Ω
2nd-Harmonic
–65
–70
–75
–80
3rd-Harmonic
–85
–90
–100
–95
50
100
500
2.5
3.0
3.5
Load Resistance (Ω)
Figure 7. 10-MHz Harmonic Distortion vs Load Resistance
5.0
5.5
6.0
–50
G = +8V/V
RL = 100Ω
Harmonic Distortion (dBc)
VO = 2VPP, G = +8V/V
RL = 100Ω
Harmonic Distortion (dBc)
4.5
Figure 8. 10-MHz Harmonic Distortion vs Supply Voltage
–50
–60
2nd-Harmonic
–70
3rd-Harmonic
–80
–90
–60
–70
2nd-Harmonic
–80
3rd-Harmonic
–90
–100
–100
0.5
1
10
0.1
100
1
Figure 9. Harmonic Distortion vs Frequency
–60
Figure 10. 10-MHz Harmonic Distortion vs Output Voltage
–55
VO = 2VPP
RL = 100Ω
VO = 2VPP, RL = 100Ω
–60
Harmonic Distortion (dBc)
–65
2nd-Harmonic
–70
–75
–80
5
Output Voltage (V PP)
Frequency (MHz)
Harmonic Distortion (dBc)
4.0
Supply Voltage (±V)
3rd-Harmonic
–85
2nd-Harmonic
–65
–70
–75
–80
–85
3rd-Harmonic
–90
–90
2
10
20
2
10
20
Inverting Gain (|V/V|)
Noninverting Gain (V/V)
Figure 11. 10-MHz Harmonic Distortion vs Noninverting
Gain
Figure 12. 10-MHz Harmonic Distortion vs Inverting Gain
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Typical Characteristics (continued)
G = +8, RF = 402 Ω, RL = 100 Ω, unless otherwise noted.
45
100
Inverting
50 Ω
G = –8
22pA/√Hz
Inverting Input Current Noise
19pA/√Hz
Noninverting Input Current Noise
10
Input Voltage Noise
Output Intercept (+dBm)
Current Noise (pA/√Hz)
Voltage Noise (nV/√Hz)
40
1.7nV/√Hz
PO
OPA695
402Ω
50 Ω
Noninverting
35
50 Ω
PI
G = 12dB to matched load.
G = +8
30
PI
50 Ω
PO
OPA695
50 Ω
25
402Ω
50 Ω
56.2Ω
20
G = 12dB to matched load.
15
1
103
104
105
106
107
108
20
40
60
80
100 120 140 160 180 200 220 240
Frequency (MHz)
Frequency (Hz)
Figure 13. Input Voltage and Current Noise Density
Figure 14. Two-Tone 3rd-Order Intermodulation Intercept ±5
V
0
0
–10
–10
G = –8
Return Loss (5dB/div)
Return Loss (5dB/div)
G = ±8V/V
–20
VSWR < 1.2:1
–30
–40
G = +8
Without
Trim Cap
–20
VSWR < 1.2:1
With
Trim Cap
–30
–40
50Ω
OPA695
–50
–50
S22
Trim Cap
–60
10M
100M
–60
10M
1G
100M
Figure 15. Input Return Loss vs Frequency (S11)
Figure 16. Output Return Loss vs Frequency (S22)
35
21
0.5dB Peaking
Allowed
CL = 10pF
Normalized Gain (dB)
30
25
RS (Ω)
1G
Frequency (Hz)
Frequency (Hz)
20
15
10
18
CL = 20pF
CL = 100pF
15
+5V
CL = 50pF
RS
VI
VO
50 Ω OPA695
12
CL
1k Ω
– 5V 402 Ω
5
57.4 Ω
0
5
14
2.5pF
10
100
9
10M
1kΩ load is optional
100M
1G
Capacitive Load (pF)
Frequency (Hz)
Figure 17. RS vs Capacitive Load
Figure 18. Small-Signal Frequency Response vs Capacitive
Load
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SBOS293H – DECEMBER 2003 – REVISED DECEMBER 2015
Typical Characteristics (continued)
G = +8, RF = 402 Ω, RL = 100 Ω, unless otherwise noted.
–PSRR
CMRR
45
40
35
30
25
–40
70
–60
60
–80
–100
50
∠ Z OL
40
–120
30
–140
20
–160
10
–180
20
–200
0
103
104
105
106
Frequency (Hz)
107
105
108
Figure 19. CMRR and PSRR vs Frequency
106
107
Frequency (Hz)
108
109
Figure 20. Open-Loop Transimpedance Gain and Phase
5
130
4
14
Sourcing Output Current
1 Watt
Internal Power
Left Scale
Output Current (mA)
3
2
1
50Ω Load Line
0
25Ω Load Line
–1
–2
100Ω Load Line
–3
13
Supply Current
Sinking Output
Current
120
Right Scale
12
Left Scale
11
1 Watt
Internal Power
–4
–5
–250 –200 –150 –100 –50
110
0
50
100
150
200
250
10
–25
0
IO (mA)
6
50
75
100
125
Figure 22. Supply and Output Current vs Temperature
6
G = +8V/V
G = –8V/V
Output
Input
Output
4
2
Input
Linear Input Range
–2
–4
Input/Output Voltage
4
0
25
Ambient Temperature (°C)
Figure 21. Output Voltage and Current Limitations
Input/Output Voltage
–20
80
Supply Current (mA)
50
Open-Loop Transimpedance
Gain (dBΩ)
Rejection Ratio (dB)
20 log| ZOL|
90
55
VO (V)
0
100
+PSRR
Open-Loop Phase ( °)
60
2
0
Linear Input Range
–2
–4
–6
–6
Time (50ns/div)
Time (50ns/div)
Figure 23. Noninverting Overdrive Recovery
Figure 24. Inverting Overdrive Recovery
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Typical Characteristics (continued)
G = +8, RF = 402 Ω, RL = 100 Ω, unless otherwise noted.
20
15
Input/Output Voltage (5mV/div)
–40
G = +8V/V
VO = 2V Step
G = +8V/V
–50
Forward
10
Input
–60
Gain (dB)
5
0
–5
–70
Reverse
–80
Output
–10
–90
–15
–20
–100
1
Time (1ns/div)
10
100
Frequency (MHz)
Figure 25. Settling Time
Figure 26. Disabled Feedthrough vs Frequency
20
0.5
10
Noninverting Input Bias Current
Right Scale
0
0
Input Offset Voltage
Left Scale
–0.5
–10
–1.0
Input/Output Swing (±) Volts
Inverting Input Bias Current
Right Scale
6
Input Bias Currents (µA)
Input Offset Voltage (mV)
1.0
–20
–50
–25
0
25
50
75
100
5
Output Voltage Range
4
3
Input Voltage Range
2
1
0
125
2.0
Ambient Temperature ( °C)
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
Power Supplies (±) Volts
Figure 27. Typical DC Drift Over Temperature
Figure 28. Common-Mode Input and Output Swing vs
Supply Voltage
0.08
5
VI
1kΩ
511 Ω
0.06
511 Ω
VDIS
VO
75 Ω OPA695
0.07
Video
Loads
4
– 5V
1kΩ, optional pulldown
3
0.05
dG
Volts
dG/dφ (%/°)
2.5
0.04
0.03
1
dG, 1kΩ Pulldown
0.02
VO
2
VIN = 0.25VDC
dφ
0
0.01
dφ, 1kΩ Pulldown
0
–1
1
2
3
4
Time (500ns/div)
Number of 150Ω Loads
Figure 29. Composite Video dG/dφ
16
Figure 30. Large-Signal Disable/Enable Response
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Typical Characteristics (continued)
G = +8, RF = 402 Ω, RL = 100 Ω, unless otherwise noted.
2
GD = 5
1
−2
19.5
GD = 20
−3
VO = 2VPP
and 4VPP
20.0
GD = 10
−1
GD = 10
20.5
Gain (dB)
Normalized Gain (dB)
0
21.0
VO = 2VPP
−4
18.5
18.0
−5
17.5
−6
17.0
−7
16.5
−8
VO = 8VPP
19.0
VO = 12VPP
VO = 16VPP
16.0
1
10
100
1000
1
10
100
1000
Frequency (MHz)
Frequency (MHz)
See Figure 47
Figure 31. Differential Small-Signal Frequency Response
Figure 32. Large-Signal Bandwidth
−65
−65
GD = 10V/V
VO = 2VPP
GD = 10V/V
F = 20MHz
RL = 800Ω
−70
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−70
−75
−80
3rd-Harmonic
−85
−90
−95
2nd-Harmonic
−75
3rd-Harmonic
−80
2nd-Harmonic
−85
−90
−95
−100
−100
−105
−105
10
0
100
2
4
Figure 33. Distortion vs Frequency
8
10
Figure 34. Distortion vs VOUT
55
6
45
40
35
30
G = +2, RF = 487Ω
3
Normalized Gain (3dB/div)
RL = 800Ω
GD = 10
50
Intercept (dBm)
6
VO (VPP)
Frequency (MHz)
G = +4, RF = 450Ω
0
–3
–6
–9
G = +8, RF = 348Ω
–12
–15
–18
G = +16, RF = 162Ω
–21
25
–24
0
20
40
60
80
100 120 140 160 180 200
0
200
400
600
800
1GHz
Center Frequency (MHz)
Frequency (200MHz/div)
Figure 35. 2-Tone, 3rd-Order Intermodulation Intercept
Figure 36. Noninverting Small-Signal Frequency Response
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Typical Characteristics (continued)
G = +8, RF = 402 Ω, RL = 100 Ω, unless otherwise noted.
4.0
6
G = –2V/V, R F = 453Ω
Normalized Gain (3dB/div)
3
G = +8V/V
100MHz, Square Wave Input
3.5
0
G = –4, RF = 442Ω
Output Voltage
–3
–6
–9
–12
G = –16, RF = 806Ω
RG = 50Ω
–15
3.0
2.5
2.0
G = –8, RF = 422Ω
–18
1.5
–21
1.0
–24
0
200
400
600
800
1GHz
Time (1ns/div)
Frequency (200MHz/div)
Figure 37. Inverting Small-Signal Frequency Response
4.0
100MHz, Square Wave Input
Figure 38. Noninverting Pulse Response
25
G = –8V/V
0.5dB Peaking
Allowed
20
3.0
RS ( Ω)
Output Voltage
3.5
2.5
15
10
2.0
5
1.5
1.0
0
Time (1ns/div)
5
10
100
Capacitive Load (pF)
Figure 40. RS vs Capacitive Load
Figure 39. Inverting Pulse Response
–50
21
18
CL = 20pF
CL = 100pF
15
CL = 50pF
+5V
1000pF
2k Ω
DIS
RS
VI
50 Ω
12
VO
2k Ω OPA695
RF
348 Ω
50 Ω
1000pF
CL
VO = 2VPP
RL = 100Ω
G = +8V/V
–55
Harmonic Distortion (dBc)
Normalized Gain (dB)
CL = 10pF
1k Ω
–60
3rd-Harmonic
–65
–70
–75
–80
–85
1kΩ load is optional
9
–90
10
100
1k
0.5
Frequency (MHz)
1
10
100
Frequency (MHz)
Figure 41. Small-Signal Frequency Response vs Capacitive
Load
18
2nd-Harmonic
Figure 42. Harmonic Distortion vs Frequency
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Typical Characteristics (continued)
G = +8, RF = 402 Ω, RL = 100 Ω, unless otherwise noted.
–50
–50
G = +8V/V
RL = 100Ω
VO = 2VPP
G = +8V/V
–55
2nd-Harmonic
–60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
–55
–65
–70
3rd-Harmonic
–75
–80
2nd-Harmonic
–60
–65
–70
3rd-Harmonic
–75
–80
–85
–85
–90
–90
0
0.5
1.0
1.5
Output Voltage (VPP)
2.0
50
2.5
Figure 43. 10-MHz Harmonic Distortion vs Output Voltage
500
Figure 44. 10-MHz Harmonic Distortion vs Load Resistance
500
40
RF = 348Ω
VO = 500mVPP
G = +8V/V
480
35
460
440
30
BW (MHz)
Intercept Point (+dBm)
100
Load Resistance (Ω)
See Figure 53
25
See Figure 52
420
400
380
360
20
340
320
15
300
20
40
60
80
100 120 140 160 180 200 220 240
4
5
6
7
8
9
10
11
12
Single Power Supply Voltage
Frequency (MHz)
Figure 45. Two-Tone, 3rd-Order Intermodulation Intercept
Figure 46. Small-Signal BW vs Single-Supply Voltage
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7 Parameter Measurement Information
7.1 Differential Small Signal Measurement
+5V
OPA695
–5V
ZI = RT || 2RG
1:1
VI
RG
RF
500Ω
RG
RF
500Ω
RT
VO
RL
800Ω
+5V
VO 500Ω
=
= GD
RG
VI
OPA695
–5V
Figure 47. Schematic for Differential Small-Signal Frequency Response
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8 Detailed Description
8.1 Overview
The OPA695, seen below in the Functional Block Diagram, is an operational amplifier with time-proven current
feedback architecture. Advantages of current feedback include no gain bandwidth product limitations, fast slew
rate, high large signal bandwidth and excellent distortion performance at high frequencies and large amplitudes.
Common applications for current feedback operational amplifiers include coaxial cable drivers, ADC drivers,
video amplifiers and high frequency gain blocks.
8.2 Functional Block Diagram
8.3 Feature Description
8.3.1 Wideband Current Feedback Operation
The OPA695 provides a new level of performance in wideband current feedback operational amplifiers. Nearly
constant AC performance over a wide gain range, along with 4300-V/μs slew rate, gives a lower power and cost
solution for high-intercept IF amplifier requirements. While optimized at a gain of +8 V/V (12 dB to a matched 50Ω load) to give 450-MHz bandwidth, applications from gains of 1 to 40 can be supported. As a gain of +2 video
line driver, the bandwidth extends to 1.4 GHz with a slew rate to support the highest pixel rates. At gains above
20, the signal bandwidth starts to decrease, but still exceeds 180 MHz up to a gain of 40 V/V (26 dB to a
matched 50-Ω load). Single +5-V supply operation is also supported with similar bandwidths but reduced output
power capability. For lower speed (< 250-MHz) requirements with higher output powers, consider the OPA691.
Figure 48 shows the DC-coupled, gain of +8 V/V, dual-power supply circuit used as the basis of the ±5-V
Specifications and Typical Characteristic curves. For test purposes, the input impedance is set to 50 Ω with a
resistor to ground, and the output impedance is set to 50 Ω with a series output resistor. Voltage swings reported
in the specifications are taken directly at the input and output pins, while load powers (dBm) are defined at a
matched 50-Ω load. For the circuit of Figure 48, the total effective load is 100 Ω || 458 Ω = 82 Ω. The disable
control line (DIS) is typically left open for normal amplifier operation. The disable line must be asserted low to
shut off the OPA695. One optional component is included in Figure 48. In addition to the usual power supply
decoupling capacitors to ground, a 0.01-μF capacitor is included between the two power supply pins. In practical
PCB layouts, this optional added capacitor typically improves the 2nd-harmonic distortion performance by 3 dB to
6 dB for bipolar supply operation.
Figure 49 shows the DC-coupled, gain of –8 V/V, dual-power supply circuit used as the basis of the Inverting
Typical Characteristic curves. Inverting operation offers several performance benefits. Because there is no
common-mode signal across the input stage, the slew rate for inverting operation is higher and the distortion
performance is slightly improved. An additional input resistor, RT, is included in Figure 49 to set the input
impedance equal to 50 Ω. The parallel combination of RT and RG set the input impedance. Both the non-inverting
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Feature Description (continued)
and inverting applications of Figure 48 and Figure 49 benefit from optimizing the feedback resistor (RF) value for
bandwidth (see the discussion in Setting Resistor Values to Optimize Bandwidth). The typical design sequence is
to select the RF value for best bandwidth, set RG for the gain, then set RT for the desired input impedance. As the
gain increases for the inverting configuration, a point is reached where RG equals 50 Ω, where RT is removed,
and the input match is set by RG only. With RG fixed to achieve an input match to 50 Ω, RF is increased to
increase gain. This quickly reduces the achievable bandwidth, as shown by the inverting gain of –16 frequency
response in the Typical Characteristic curves. For gains > 10 V/V (14 dB at the matched load), noninverting
operation is recommended to maintain broader bandwidth.
+5V
+
0.1µF
6.8µF
50Ω Source
DIS
VI
50Ω
VO
50Ω
50Ω Load
OPA695
Optional
0.01µF
RF
402Ω
RG
56.2Ω
0.1µF
+
6.8µF
–5V
Figure 48. DC-Coupled, G = +8 V/V, Bipolar Supply Specifications and Test Circuit
+5V
+VS
+
0.1µF
20Ω
6.8µF
DIS
50Ω Load
VO
50Ω
OPA695
Optional
0.01µF
50Ω Source
RF
442Ω
RG
54.9Ω
VI
RT
562Ω
0.1µF
+
6.8µF
–VS
–5V
Figure 49. DC-Coupled, G = –8 V/V, Bipolar Supply Specifications and Test Circuit
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Feature Description (continued)
Figure 50 shows the AC-coupled, single +5-V supply, gain of +8 V/V circuit configuration used as a basis for the
+5V-only Specifications and Typical Characteristic curves. The key requirement for broadband single-supply
operation is to maintain input and output signal swings within the useable voltage ranges at both the input and
the output. The circuit of Figure 50 establishes an input midpoint bias using a simple resistive divider from the
+5-V supply (two 806-Ω resistors) to the noninverting input. The input signal is then AC-coupled into this
midpoint-voltage bias. The input voltage can swing to within 1.6 V of either supply pin, giving a 1.8-VPP input
signal range centered between the supply pins. The input impedance matching resistor (57.6 Ω) used in
Figure 50 is adjusted to give a 50-Ω input match when the parallel combination of the biasing divider network is
included. The gain resistor (RG) is AC-coupled, giving the circuit a DC gain of +1. This puts the input DC bias
voltage (2.5 V) on the output as well. The feedback resistor value has been adjusted from the bipolar supply
condition to re-optimize for a flat frequency response in +5 V only, gain of +8 operation (see Setting Resistor
Values to Optimize Bandwidth). On a single +5-V supply, the output voltage can swing to within 1.0 V of either
supply pin while delivering more than 90-mA output current, giving 3-V output swing into 100 Ω (7-dBm maximum
at the matched load). The circuit in Figure 50 shows a blocking capacitor driving into a 50-Ω output resistor, then
into a 50-Ω load. Alternatively, the blocking capacitor could be removed with the load tied to a supply midpoint, or
to ground if the DC current required by this grounded load is acceptable.
Figure 51 shows the AC-coupled, single +5-V supply, gain of –8 V/V circuit configuration used as a basis for the
+5V-only Typical Characteristic curves. In this case, the midpoint DC bias on the noninverting input is also decoupled with an additional 0.1-μF decoupling capacitor. This reduces the source impedance at higher
frequencies for the noninverting input bias current noise. This 2.5-V bias on the noninverting input pin appears on
the inverting input pin and, because RG is DC-blocked by the input capacitor, also appears at the output pin. One
advantage to inverting operation is that as there is no signal swing across the input stage, higher slew rates and
operation to lower supply voltages are possible. To retain a 1-VPP output capability, operation down to a 3-V
supply is allowed. At a +3-V supply, the input common mode range is 0 V. However, for the inverting
configuration of a current feedback amplifier, wideband operation is retained even with the input stage saturated.
+5V
+VS
+
0.1µF
6.8µF
806Ω
50Ω Source
0.1µF
DIS
50Ω Load
VI
57.6Ω
1000pF
OPA695
806Ω
RF
348Ω
VO
0.1µF
50Ω
1000pF
RG
50Ω
1000pF
0.1µF
Figure 50. AC-Coupled, G = +8 V/V, Single-Supply Specifications and Test Circuit
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Feature Description (continued)
+5V
+VS
+
0.1µF
6.8µF
806Ω
20Ω
DIS
50Ω Load
1000pF
0.1µF
806Ω
OPA695
VO
0.1µF
50Ω
1000pF
0.1µF
RF
400Ω
RG
50Ω
VI
1000pF
Figure 51. AC-Coupled, G = –8 V/V, Single-Supply Specifications and Test Circuit
The single-supply test circuits of Figure 50 and Figure 51 show +5-V operation. These same circuits can be used
over a single-supply range of +5 V to +12 V. Operating on a single +12-V supply, with the Absolute Maximum
Supply voltage specification of +13 V, gives adequate design margin for the typical ±5% supply tolerance.
8.3.2 RF Specifications and Applications
The ultra-high, full-power bandwidth and 3rd-order intercept of the OPA695 are ideal for IF amplifier applications.
The advantage of a wideband operational amplifier such as the OPA695 include good (and independent) I/O
impedance matching, as well as high reverse isolation. A designer accustomed to fixed-gain RF amplifiers will
get almost perfect gain accuracy, higher I/O return loss, and 3rd-order intercept points exceeding 30 dBm (up to
110 MHz) using only a 13-mA supply current for the OPA695. Using the considerable design freedom achieved
by adjusting the external resistors, the OPA695 can replace a wide range of fixed-gain RF amplifiers with a
single part. To understand (in RF amplifier terms) how to take advantage of this, consider first the 4-S
parameters (see the example circuits of Figure 48 and Figure 49 on ±5-V supplies, but similar results can be
obtained on a single +5-V to +12-V supply).
8.3.3 Input Return Loss (S11)
Input return loss is a measure of how closely (over frequency) the input impedance matches the source
impedance. This is relatively independent of gain setting for both the noninverting and inverting configurations.
The Typical Characteristics show the magnitude of S11 for the circuits of Figure 48 and Figure 49 through 1 GHz
(noninverting gain of +8 and inverting gain of –8 operation, respectively). Noninverting operation offers much
better matching to higher frequencies, with the only deviation due to the parasitic input capacitance of the input
pin. The noninverting input match is simply set by the resistor to ground on the noninverting input, as the
amplifier itself shows a very high input impedance. Inverting operation is also good, but rises more quickly due to
loop gain roll-off effects appearing at the inverting node. The inverting mode input match is set by the parallel
combination of RG and RT in Figure 49, as the inverting amplifier node may be considered a virtual ground. A
good, fixed-gain, RF amplifier would have an input, Voltage Standing Wave Ratio (VSWR) < 1.2:1. This
corresponds to an S11 of –21 dB. The OPA695 exceeds this performance through 100 MHz for the inverting
mode of operation, and through 400 MHz for the noninverting mode.
8.3.4 Output Return Loss (S22)
Output return loss is a measure of how closely (over frequency) the output impedance matches the load
impedance. This is relatively independent of gain setting for both the noninverting and inverting configurations.
The output matching impedance, to a first order, is set by adding a series resistor to the low impedance output of
the operational amplifier.
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Feature Description (continued)
Because the operational amplifier itself shows a low output impedance that increases with frequency, an
improvement in the output match can therefore be obtained by adding a small equalizing capacitor across this
output resistor. The Typical Characteristics show the measured S22 with and without this 2.5-pF capacitor (across
the 50-Ω output resistor). Again, a good match for a fixed-gain RF amplifier would give a VSWR of 1.2:1 (S22 <
–21 dB). The Typical Characteristic curves show that a simple 50-Ω output resistor holds better than –21 dB to
140 MHz, but up to 380 MHz with the tuning capacitor.
8.3.5 Forward Gain (S21)
In all high-speed amplifier data sheets, forward gain is the small signal gain plotted over frequency. The
difference between noninverting and inverting operation is that the phase of S21 starts out at 0° for the
noninverting and –180° for the inverting. This initial phase shift for inverting mode is inconsequential to most IF
strip applications. The phase of S21 was not shown in Typical Characteristics, but is linear with frequency and
may be accurately modeled as a constant time delay through the amplifier.
The Typical Characteristics show S21 over a range of signal gains, where the external resistors have been
adjusted to re-optimize flatness at each gain setting. Because this is a current feedback operational amplifier, the
signal bandwidth can be held relatively constant as the desired gain setting is changed. The plot of the
noninverting bandwidth versus gain shows some change in bandwidth versus gain (due to parasitic capacitive
effects on the inverting node) with very little change showing up for the inverting mode of operation.
Signal gains are most often referred to as V/V in operational amplifier data sheets. This is the voltage gain from
input to output and is set by external resistor ratios. Because the output impedance is set by a physical series
resistor, the voltage gain to the matched load is cut in half by this resistor divider. The log gain to the matched
load for the noninverting circuit of Figure 48 is:
G + = 20 log
R
1
1 + F dB
2
RG
(
)
(1)
The log gain to the matched load for the inverting circuit of Figure 49 is:
G – = 20 log
1 RF
dB
2 RG
( )
(2)
The specific resistor values used in Figure 48 and Figure 49 give both a maximally-flat bandwidth and a 12-dB
gain to the matched load. The design tables located in the Noise Figure section summarize the required resistor
values over a range of desired gains for the circuits of Figure 48 and Figure 49.
As the desired signal gain increases, the achievable bandwidths decrease. In the noninverting case, it decreases
relatively quickly as shown in Typical Characteristics. The inverting configuration holds almost constant
bandwidth (with correctly selected external resistor values) until RG reduces to equal 50 Ω, and remains at that
value to satisfy the input impedance matching requirement, with further increases in gain achieved by increasing
RF in Figure 49. The bandwidth then decreases rapidly as shown by the gain of –16 V/V plot in Typical
Characteristics.
8.3.6 Reverse Isolation (S12)
Reverse isolation is a measure of how much power injected into the output pin returns to the source. This is
rarely specified for an operational amplifier because operational amplifiers are nearly uni-directional signal
devices. Below 300 MHz, the noninverting configuration of Figure 48 gives much better isolation than the
inverting of Figure 49. Both are well below 40-dB isolation through 350 MHz.
8.3.7 Limits to Dynamic Range
The next set of considerations for RF amplifier applications are the defined limits to dynamic range. Typical fixedgain RF amplifiers include:
• –1-dB compression (a measure of maximum output power)
• Two-tone, 3rd-order, output intermodulation intercept (a measure of achievable spurious-free dynamic range)
• Noise figure (a measure of degradation in signal to noise ratio in passing through the amplifier)
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Feature Description (continued)
8.3.7.1 –1-dB Compression
The definition for –1-dB compression power is output power where the actual power is 1 dB less than the input
power, plus the log gain. In classic RF amplifiers, this is typically 10 dB less than the 3rd-order intercept. That
relationship does not hold for operational amplifiers, as their intercept is improved by loop gain to be far more
than 10 dB higher than the –1-dB compression. A simple estimate for –1-dB compression for the OPA695 is the
maximum non-slew limited output voltage swing available at the matched load, converted into a power with 1 dB
added to satisfy the definition. For the OPA695 on ±5-V supplies, its output will deliver approximately ±4.0 V at
the output pin or ±2.0 V at the matched load. The conversion from VPP to power (for a sine wave) is:
( 2V√PP2 )
2
[ ]
PO (dBm) = 10 log
0.001(50Ω)
(3)
Converting this 4.0-VPP swing at the load to dBm gives 16 dBm; adding 1 dB to this (to satisfy the definition)
gives a –1-dB compression of 17 dBm for the OPA695 operating on ±5-V supplies. This is a good estimate for
frequencies that require less than the full slew rate of the OPA695.
The maximum frequency of operation given an available slew rate and desired peak output swing (at the output
pin for a sine wave) is:
FMAX =
Slew Rate
2 π Vp (0.707)
(4)
Putting in the 4600-V/μs slew rate available in the inverting mode of operation and the 4.0-V peak output swing
at the output pin gives a maximum frequency of 259 MHz. This is the maximum frequency where the –1-dB
compression would be 17 dBm at the matched load. Higher useable bandwidths are possible at lower output
powers, as shown in the Large Signal Bandwidth curves. As those graphs show, 7-VPP outputs are possible with
almost perfect frequency response flatness through 100 MHz for both non-inverting or inverting operation.
8.3.7.2 Two-Tone 3rd-Order Output Intermodulation Intercept (OP3)
In narrowband IF strips, each amplifier typically feeds into a bandpass filter that attenuates most harmonic
distortion terms. The most troublesome remaining distortion is the 3rdorder, two-tone intermodulations that can
fall very close (in frequency) to the desired signals and cannot be filtered out. If two test frequencies are defined
at FO + ΔF and FO – ΔF, the 3rd-order intermodulation distortion products will fall at FO + 3ΔF and FO – 3ΔF. If
the two test power levels (PT) are equal, the OPA695 produces 3rd-order spurious terms (PS) at these
frequencies, and at a power level below the test power levels given by:
PT – PS = 2 (OP3 – PT)
(5)
The 3rd-order intercept plot shown in Typical Characteristics shows a very high intercept at low frequencies that
decreases with increasing frequency. This intercept is defined at the matched load to allow direct comparison
with fixed-gain RF amplifiers. To produce a 2-VPP total two-tone envelope at the matched load, each power level
must be 4 dBm at the matched load (1 VPP). Using Equation 5, and the performance curve for inverting
operation, at 50 MHz (41.5-dBm intercept) the 3rd-order spurious will be 2 × (41.5 – 4) = 75 dB below these 4dBm test tones. This is an exceptionally low distortion for an amplifier that only uses 13-mA supply current.
Considerable improvement from this level of performance is also possible if the output drives directly into the
lighter load of an ADC input.
This very high intercept versus quiescent power is achieved by the high loop gain of the OPA695. This loop gain
does, however, decrease with frequency, giving the decreasing OP3 performance shown in Typical
Characteristics. Application as an IF amplifier through 200 MHz is possible with output intercepts exceeding 21
dBm at 200 MHz. Intercept performance varies slightly with gain setting, decreasing at higher gains (that is, gains
greater than the 8 V/V or 12 dB gain used in the Typical Characteristic curves) and increasing at lower gains.
8.3.7.3 Noise Figure
All fixed-gain RF amplifiers show a very good noise figure (typically < 5 dB). For broadband amplifiers, this is
achieved by a low-noise input transistor and an input match set by feedback. This feedback greatly reduces the
noise figure for fixed-gain RF amplifiers, but also makes the input match dependent on the load and the output
match dependent on the source impedance at the input.
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Feature Description (continued)
The noise figure for an operational amplifier is always higher than for fixed-gain RF amplifiers, due to the more
complex internal circuits of an operational amplifier (giving higher input noise voltage and current terms). Also, for
simple circuits, the input match is set resistively. What is gained is an almost perfect I/O impedance match, much
better load isolation, and very high 3rd-order intercepts versus quiescent power. These higher noise figures can
be acceptable if the OPA695 has enough gain preceding it in the IF chain.
Operational amplifier noise figure equations include at least six terms (see Noise Performance), due to the
external resistors. As a point of reference, the circuit of Figure 48 has an input noise figure of 14 dB, while the
inverting configuration of Figure 49 has an input noise figure of 11 dB. At higher gains, it is typical for the
inverting noise figure to be slightly better than for an equivalent gain, noninverting configuration. Improve the
noise figure for the noninverting configuration of the OPA695 by including a step-up, 1:2 turns ratio transformer at
the input. This configuration is shown in Figure 52.
Supply decoupling
not shown.
+5V
50Ω Source
DIS
1:2
VI
50Ω Load
50Ω
OPA695
200Ω
VO
RF
–5V
RG
Figure 52. IF Amplifier With Improved Noise Figure
The transformer provides a noiseless voltage gain at the expense of higher source impedance for the OPA695
noninverting input current noise. The input impedance is still set to 50 Ω by the 200-Ω resistor on the transformer
secondary. A 1:2 turns ratio transformer will reflect the 200 Ω to the input side as a 50-Ω impedance over the
bandwidth of the transformer. Using a 1:2 step-up transformer also reduces the required amplifier gain by 1/2 for
any particular desired overall gain.
Table 1, Table 2, and Table 3 summarize the recommended resistor values and resulting noise figures over the
desired gain setting for three circuit options for the OPA695 operated as a precision IF amplifier. In each case,
RF and RG are adjusted for both best bandwidth and required gain.
In all cases, exact computed values for resistors are shown; in an application, pick standard resistor values that
are closest to those in the tables.
Table 1. Noninverting Wideband Operational Amplifier
GAIN TO LOAD (dB)
RF (Ω)
RG (Ω)
NOISE FIGURE
6
478
159
17.20
7
468
134
16.55
8
458
113
15.95
9
446
96
15.40
10
433
81
14.91
11
419
68
14.47
12
402
57
14.09
13
384
48
13.76
14
363
40
13.23
15
340
33
13.23
16
314
27
13.03
17
284
21
12.86
18
252
16
12.72
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Feature Description (continued)
Table 1. Noninverting Wideband Operational Amplifier (continued)
GAIN TO LOAD (dB)
RF (Ω)
RG (Ω)
NOISE FIGURE
19
215
12
12.60
20
174
9
12.51
Table 2. Noninverting With a 1:2 Input Step-Up Transformer
GAIN TO LOAD (dB)
RF (Ω)
RG (Ω)
NOISE FIGURE
6
516
518
16.34
7
511
412
15.54
8
506
334
14.78
9
500
275
14.07
10
493
228
13.40
11
486
190
12.78
12
478
160
12.21
13
469
135
11.70
14
458
114
11.25
15
447
96
10.85
16
434
81
10.15
17
419
69
10.21
18
403
58
9.96
19
384
48
9.74
20
364
40
9.57
Table 3. Inverting Wideband RF Amplifier
GAIN TO LOAD (dB)
Optimum RF (Ω)
RG (Ω)
Input Match RT
NOISE FIGURE
6
463.27
116
87
16.94
7
454.61
101
98
16.06
8
444.91
88
114
15.16
9
434.07
77
142
14.23
10
421.95
66
199
13.24
11
408.42
57
380
12.16
12
398.11
50
Infinite
11.03
13
446.68
50
Infinite
10.92
14
501.19
50
Infinite
10.83
15
562.34
50
Infinite
10.75
16
630.96
50
Infinite
10.67
17
707.95
50
Infinite
10.61
18
794.33
50
Infinite
10.55
19
891.25
50
Infinite
10.49
20
1000.00
50
Infinite
10.45
8.4 Device Functional Modes
The OPA695 has two functional modes. The first functional mode is accessed by applying a logic 1 (>3.3 V) to
the not Disable (Disable bar) pin. In this mode the amplifier is fully enabled and will draw a supply current of 13
mA.
The second functional mode is the disabled state. The disabled state is accessed by applying a logic 0 (<1.8 V)
to the not Disable pin. In this mode, the amplifier is fully disabled and draws a current of only 100 µA.
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
9.1.1 SAW Filter Buffer
One common requirement in an IF strip is to buffer the output of a mixer with enough gain to recover the
insertion loss of a narrowband SAW filter. Figure 65 shows one possible configuration driving a SAW filter.
Figure 53 shows the intercept at the 50-Ω load. Operating in the inverting mode at a voltage gain of –8 V/V, this
circuit provides a 50-Ω input match using the gain set resistor, has the feedback optimized for maximum
bandwidth (700 MHz in this case), and drives through a 50-Ω output resistor into the matching network at the
input of the SAW filter. If the SAW filter gives a 12-dB insertion loss, a net gain of 0 dB to the 50-Ω load at the
output of the SAW (which could be the input impedance of the next IF amplifier or mixer) is delivered in the
passband of the SAW filter. Using the OPA695 in this application isolates the first mixer from the impedance of
the SAW filter and provides very low two-tone, 3rd-order spurious levels in the SAW filter bandwidth. Inverting
operation gives the broadest bandwidth up to a gain of –12 V/V (15.6 dB). Noninverting operation gives higher
bandwidth at gain settings higher than this, but will also give a slight reduction in intercept and noise figure
performance.
Output Intercept (dBm)
50
40
30
20
10
0
50
100
150
200
250
Center Frequency (MHz)
Figure 53. 2-Tone, 3rd-Order Intermodulation Intercept
9.1.2 LO Buffer Amplifier
The OPA695 can also be used to buffer the Local Oscillator (LO) from the mixer. Operating at a voltage gain of
+2, the OPA695 provides almost perfect load isolation for the LO, with a net gain of 0 dB to the mixer.
Applications through 1.4-GHz LOs may be considered, but best operation would be for LOs < 1.0 GHz at a gain
of +2. Gain can also be provided by the OPA695 to drive higher power levels into the mixer. One option for the
OPA695 as an LO buffer is shown in Figure 54. Because the OPA695 can drive multiple output loads, two
identical LO signals may be delivered to the mixers in a diversity receiver by tapping the output off through two
series 50-Ω output resistors. This circuit is set up for a voltage gain of +2 V/V to the output pin for a gain of +1
V/V (0 dB) to the mixers, but could easily be adjusted to deliver higher gains as well.
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Antenna
IF1
LNA
Diversity Receiver
Bandpass
Filter
Antenna
LNA
IF2
+5V
DIS
LO
Bandpass
Filter
50Ω
OPA695
50Ω
50Ω
–5V
RG
511Ω
RF
511Ω
Power supply decoupling not shown.
Figure 54. Dual Output LO Buffer
9.1.3 Wideband Cable Driving Applications
The high slew rate and bandwidth of the OPA695 can be used to meet the most demanding cable driving
applications.
9.1.3.1 Cable Modem Return Path Driver
The standard cable modem upstream driver is typically required to drive high power over a 5-MHz to 65-MHz
bandwidth while delivering < –50-dBc distortion. Highly-integrated solutions (including programmable gain
stages) often fall short of this target due to high losses from the amplifier output to the line. The higher gainoperating capability of the OPA695 and its very high slew rate provide a low-cost solution for delivering this
signal with the required spurious-free dynamic range. Figure 55 shows one example of using the OPA695 as an
upstream driver for a cable modem return path. In this case, the input impedance of the driver is set to 75 Ω by
the gain resistor (RG). The required input level from the adjustable gain stage is significantly reduced by the 15.5dB gain provided by the OPA695. In this example, the physical 75-Ω output matching resistor, along with the 3dB loss in the diplexer, attenuate the output swing by 9 dB on the line. In this example, a single +12-V supply
was used to achieve the lowest harmonic distortion for the 6-VPP output pin voltage through 65 MHz. Measured
performance for this example gave 600-MHz small-signal bandwidth and < –54-dBc distortion through 65 MHz
for a 6-VPP output pin voltage swing.
An alternative to this circuit that gives even lower distortion is a differential driver using two OPA695s driving into
an output transformer. This can be used either to double the available line power, or to improve distortion by
cutting the required output swing in half for each stage. The channel disable required by the MCNS specification
must be implemented by using the PGA disable feature. The MCNS disable specification requires that an output
impedance match be maintained with the signal channel shut off. The disable feature of the OPA695 is intended
principally for power savings and puts the output and inverting input pins into a high impedance mode. This does
not maintain the required output-impedance matching. Turning off the signal at the input of Figure 55, while
keeping the OPA695 active, maintains the impedance matching while putting very little noise on the line. The line
noise in disable for the circuit of Figure 55 (with the PGA source turned off, but still presenting a 75-Ω source
impedance) will be a very low 4 nV/√Hz (–157 dBm/Hz) due to the low input noise of the OPA695.
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Application Information (continued)
Receive Channel
+12V
67dBmV
6kΩ
0.1µF
6kΩ
Diplexer
–3dB
DIS
75Ω
0.01µF
1000pF
58dBmV
Supply decoupling
not shown
20Ω
75Ω
OPA695
1000pF
PGA Output
0.1µF
RG
75Ω
RF
450Ω
51.5dBmV
1000pF
Figure 55. Cable Modem Upstream Driver
9.1.3.2 RGB Video Line Driver
The extremely high bandwidth of the OPA695 operating at a gain of +2 supports the fastest RAMDAC outputs for
applications such as auxiliary monitor driving. Gain 2V/V Video Line Driver shows measured performance for a
0 → +1-V input square wave at 125 MHz. As a general rule, the required full-power bandwidth for the amplifier
must be at least one-half the pixel rate. With its noninverting gain of +2, slew rate of 2900 V/μs, and a 1.4-VPP
output pin voltage swing for standard RGB video levels, the OPA695 gives a bandwidth of 600 MHz, which then
supports up to 1.26-GHz pixel rates. Figure 56 shows an example where three OPA695s provide an auxiliary
monitor output for a highresolution RGB RAMDAC.
An alternative circuit that takes advantage of the higher inverting slew rate of the OPA695 (4300 V/μs) takes the
complementary current output from the RAMDAC and converts it to positive video to give a very high, full-power
bandwidth RGB line driver. This will give sharper pixel edges than the circuit of Figure 56. Most high-speed
DACs are current-steering designs with both an output current signal used for the video, and a complementary
output that is typically discarded into a matching resistor. The complementary current output can be used as an
auxiliary output if it is inverted, as shown in Figure 57. In the circuit of Figure 57, the complementary current
output is terminated by an equivalent 75-Ω impedance (the parallel combination of RT and RG) that also provides
a current division to reduce the signal current through the feedback resistor, RF. This allows RF to be increased
to a value which holds a flat frequency response. Since the complementary current output is essentially an
inverted video signal, this circuit sets up a white video level at the output of the OPA695 for zero DAC output
current (using the 0.77-V DC bias on the noninverting input), then inverts the complementary output current to
produce a signal that ranges from this 1.4 V at zero output current down to 0 V at maximum output current level
(assuming a 20-mA maximum output current). This gives a very wideband (> 800-MHz) video signal capability.
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Red
75Ω
RAMDAC
Green
Power supply decoupling not shown.
75Ω
+5V
Blue
75Ω
DIS
20Ω
75Ω
OPA695
RF
511Ω
–5V
Addtional
OPA695
Stages
511Ω
Figure 56. Gain of +2, High-Resolution RGB Monitor Output
+5V
Power supply decoupling not shown.
4.22kΩ
DIS
20Ω
0.77V
75Ω
OPA695
0.1µF
768Ω
–5V
RAMDAC
RG
536Ω
RF
500Ω
IO
RT
86.6Ω
Figure 57. High-Resolution RGB Driver Using DAC Complementary Output Current
9.1.3.3 Arbitrary Waveform Driver
The OPA695 can be used as the output stage for moderate output power arbitrary waveform driver applications.
Driving out through a series 50-Ω matching resistor into a 50-Ω matched load allows up to a 4.0-VPP swing at the
matched load (15 dBm) when operating the OPA695 on a ±5-V power supply. This level of power is available for
gains of either ±8 with a flat response through 100 MHz. When interfacing directly from a complementary current
output DAC, consider the circuit of Figure 57, modified for the peak output currents of the particular DAC being
considered. Where purely AC-coupled output signals are required from a complementary current output DAC,
consider a push-pull output stage using the circuit of Figure 58. The resistor values here have been calculated
for a 20-mA peak output current DAC, which produces up to a 5-VPP swing at the matched load (18 dBm). This
approach gives higher power at the load, with lower 2nd-harmonic distortion.
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For a 20-mA peak output current DAC, the mid-scale current of 10 mA gives a 2-V DC output common-mode
operating voltage, due to the 200-Ω resistor to ground at the outputs. The total AC impedance at each output is
50 Ω, giving a ±0.5-V swing around this 2-V common-mode voltage for the DAC. These resistors also act as a
current divider, sending 75% of the DAC output current through the feedback resistor (464 Ω). The blocking
capacitor references the OPA695 output voltage to ground, and turns the unipolar DAC output current into a
bipolar swing of 0.75 × 20 mA × 464 Ω = 7 VPP at each amplifier output. Each output is exactly 180° out-of-phase
from the other, producing double 7 VPP into the matching resistors. To limit the peak output current and improve
distortion, the circuit of Figure 58 is set up with a 1.4:1 stepdown transformer. This reflects the 50-Ω load to be
100 Ω at the primary side of the transformer. For the maximum 14-VPP swing across the outputs of the two
amplifiers, the matching resistors will drop this to 7 VPP at the input of the transformer, then down to 5-VPP
maximum at the 50-Ω load at the output of the transformer. This step-down approach reduces the peak output
current to 14 VP/(200 Ω) = 70 mA.
+5V
Power supply decoupling not shown.
20Ω
DIS
OPA695
±3.5V
50Ω Source
0.01µF
66.5Ω
464Ω
50Ω
1.4:1
IO
200Ω
–5V
DAC
0.01µF
+5V
464Ω
66.5Ω
50Ω
Differential
Filter
IO
200Ω
OPA695
20mA Peak Output
±3.5V
20Ω
DIS
–5V
Figure 58. High Power, Wideband AC-Coupled Arbitrary Waveform Driver
9.1.4 Differential I/O Applications
The OPA695 offers very low 3rd-order distortion terms with a dominant 2nd-order distortion for the single
amplifier operation. For the lowest distortion, particularly where differential outputs are needed, operating two
OPA695s in a differential I/O design suppresses these even-order terms, delivering extremely low harmonic
distortion through high frequencies and powers. Differential outputs are often preferred for high performance
ADCs, twisted-pair driving, and mixer interfaces. Two basic approaches to differential I/Os are the noninverting or
inverting configurations. Because the output is differential, the signal polarity is somewhat meaningless; the
noninverting and inverting terminology applies here to where the input is brought into the two OPA695s. Each
approach has its advantages and disadvantages. Figure 59 shows a basic starting point for non-inverting
differential I/O applications.
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Application Information (continued)
+VCC
OPA695
–VCC
VI
RG
+VCC
RF
500Ω
RF
500Ω
VO
OPA695
–VCC
Figure 59. Noninverting Input Differential I/O Amplifier
This approach allows for a source termination impedance independent of the signal gain. For instance, simple
differential filters may be included in the signal path right up to the non-inverting inputs without interacting with
the gain setting. The differential signal gain for the circuit of Figure 59 is:
AD = 1 + 2 × RF/RG
(6)
Because the OPA695 is a current feedback amplifier, its bandwidth is principally controlled with the feedback
resistor value: Figure 59 shows a typical value of 500 Ω. However, the differential gain may be adjusted with
considerable freedom using just the RG resistor. RG can be a reactive network providing an isolated shaping to
the differential frequency response. AC-coupled applications often include a blocking capacitor in series with RG.
This reduces the gain to 1 at low frequency, rising to the AD expression shown above at higher frequencies. The
noninverting input approach of Figure 59 can be used for higher gains than the inverting input approach, but may
have a reduced full-power bandwidth due to the lower slew rate of the OPA695 running a noninverting versus
inverting input mode of operation.
Various combinations of single-supply or AC-coupled gain can also be delivered using the basic circuit of
Figure 59. Common-mode bias voltages on the two noninverting inputs pass on to the output with a gain of 1, as
an equal DC voltage at each inverting node creates no current through RG. This circuit shows a common-mode
gain of 1 from input to output. The source connection must either remove this common-mode signal if it is
unnecessary (using an input transformer), or the common-mode voltage at the inputs can set the output
common-mode bias. If the low common-mode rejection of this circuit is a problem, the output interface may also
be used to reject that common-mode. For instance, most modern differential input ADCs reject common-mode
signals well, while a line driver application through a transformer also removes the common-mode signal at the
secondary of the transformer.
Figure 60 shows a differential I/O stage configured as an inverting amplifier. In this case, the gain resistors (RG)
become part of the input resistance for the source. This provides a better noise performance than the noninverting configuration, but limits the flexibility in setting the input impedance separately from the gain.
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Application Information (continued)
+VCC
VCM
OPA695
RG
VI
–VCC
RF
500Ω
RF
500Ω
RG
VO
OPA695
VCM
–VCC
Figure 60. Inverting Input Differential I/O Amplifier
The two noninverting inputs provide an easy common-mode control input, particularly if the source is AC-coupled
through either blocking caps or a transformer. In either case, the common-mode input voltages on the two
noninverting inputs again have a gain of 1 to the output pins, giving easy common-mode control for single-supply
operation. The OPA695 in this configuration constrains the feedback to the 500-Ω region for best frequency
response. With RF fixed, the input resistors may be adjusted to the desired gain, but will also be changing the
input impedance. The high-frequency common-mode gain for this circuit from input to output is the same as for
the signal gain. Again, if the source might include an undesired common-mode signal, that could be rejected at
the input using blocking caps (for low-frequency and DC common-mode) or a transformer coupling. The
differential performance plots shown in the Typical Characteristics used the configuration of Figure 60 and an
input 1:1 transformer. The differential signal gain in the circuit of Figure 60 is:
AD = RF/RG
(7)
Using this configuration suppresses the 2nd-harmonics, leaving only 3rd-harmonic terms as the limit to output
SFDR. The higher slew rate of the inverting configuration also extends the full-power bandwidth and the range of
low intermodulation distortion over the performance bandwidth available from the circuit of Figure 59. The Typical
Characteristics show that the circuit of Figure 60 operating at an AD = 10 can deliver a 16 VPP signal with over
500-MHz –3-dB bandwidth. Using Equation 4, this implies a differential output slew of 18000 V/μsec, or 9000
V/μsec at each output. This output slew rate is far higher than specified, and probably due to the lighter load
used in the differential tests.
This inverting input differential configuration is suited to high SFDR converter interfaces, specifically narrowband
IF channels. The Typical Characteristics show the 2-tone, 3rd-order intermodulation intercept exceeding 45 dBm
through 90 MHz. Although this data was taken with an 800-Ω load, the intercept model appears to work for this
circuit, treating the power level as if it were into 50 Ω. For example, at 70 MHz, the differential Typical
Characteristic plots show a 48 dBm intercept. To predict the 2-tone intermodulation SFDR, assuming a –1-dB
below full-scale envelope to a 2-VPP maximum differential input converter, the test power level would be 9 dBm –
6 dBm = 3 dBm for each tone. Putting this into the intercept equation, gives:
ΔdBc = 2 × (48 – 3) = 90 dBc
(8)
The single-tone distortion data shows approximately 72-dB SFDR at 70 MHz for a 2-VPP output into this light
800-Ω load. A modest post filter after the amplifier can reduce these harmonics (2nd at 140 MHz, 3rd at 210
MHz) to the point where the full SFDR to a converter can be in the 85-dB range for a 70-MHz IF operation.
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Application Information (continued)
9.1.5 Operating Suggestions
9.1.5.1 Setting Resistor Values to Optimize Bandwidth
A current-feedback operational amplifier such as the OPA695 can hold an almost constant bandwidth over signal
gain settings with the proper adjustment of the external resistor values. This is shown in Typical Characteristics.
The small-signal bandwidth decreases only slightly with increasing gain. These curves also show that the
feedback resistor has been changed for each gain setting. The resistor values on the inverting side of the circuit
for a current-feedback operational amplifier can be treated as frequency response compensation elements, while
their ratios set the signal gain. Figure 15 shows the analysis circuit for the OPA695 small-signal frequency
response.
The key elements of this current feedback operational amplifier model are:
• α ⇒ Buffer gain from the noninverting input to the inverting input.
• RI ⇒ Buffer output impedance
• iERR ⇒ Feedback error current signal
• Z(s) ⇒ Frequency-dependent, open-loop transimpedance gain from iERR to VO
VI
α
VO
RI
iERR
Z(S) iERR
RF
RG
Figure 61. Current-Feedback Transfer Function Analysis Circuit
The buffer gain is typically very close to 1.00 and is normally neglected from signal gain considerations. It will,
however, set the CMRR for a single operational amplifier differential amplifier configuration. For the buffer gain α
< 1.0, the CMRR = –20 × log (1 – α).
RI, the buffer output impedance, is a critical portion of the bandwidth control equation. For the OPA695, it is
typically about 28 Ω for ±5-V operation, and 31 Ω for single +5-V operation.
A current-feedback operational amplifier senses an error current in the inverting node (as opposed to a
differential input error voltage for a voltage-feedback operational amplifier) and passes this on to the output
through an internal frequency-dependent transimpedance gain. Typical Characteristics show this open-loop
transimpedance response. This is analogous to the open-loop voltage gain curve for a voltage-feedback
operational amplifier. Developing the transfer function for the circuit of Figure 64 gives Equation 9:
(
α 1+
VO
=
VI
RF
RG
)
( )
RF + RI 1 +
1+
RF
RG
=
α • NG
RF + RI • NG
1+
Z (S)
Z (S)
where
•
NC = 1 + RF/RG = Noise Gain
(9)
This is written in a loop gain analysis format, where the errors arising from a non-infinite open-loop gain are
shown in the denominator. If Z(s) were infinite over all frequencies, the denominator of Equation 9 would reduce
to 1, and the ideal desired signal gain shown in the numerator would be achieved. The fraction in the
denominator of Equation 9 determines the frequency response. Equation 10 shows this as the loop gain
equation:
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Z (S )
RF + RI • NG
= Loop Gain
(10)
If 20 × log (RF + NG × RI) were superimposed on the open-loop transimpedance plot, the difference between the
two would be the loop gain at a given frequency. Eventually, Z(s) rolls off to equal the denominator of
Equation 10, at which point the loop gain has reduced to 1 (and the curves have intersected). This point of
equality is where the amplifier closed-loop frequency response given by Equation 9 starts to roll off, and is
exactly analogous to the frequency at which the noise gain equals the open-loop voltage gain for a voltagefeedback operational amplifier. The difference is that the total impedance in the denominator of Equation 10 may
be controlled separately from the desired signal gain (or NG).
The OPA695 is internally compensated to give a maximally flat frequency response for RF = 402 Ω at NG = 8 on
±5-V supplies. Evaluating the denominator of Equation 7 (the feedback transimpedance) gives an optimal target
of 663 Ω. As the signal gain changes, the contribution of the NG ×RI term in the feedback transimpedance
changes, but the total can be held constant by adjusting RF. Equation 11 gives an approximate equation for
optimum RF over signal gain:
RF = 663 Ω – NG × RI
(11)
As the desired signal gain increases, this equation will eventually predict a negative RF. A subjective limit to this
adjustment can be set by holding RG to a minimum value of 10 Ω. Lower values will load both the buffer stage at
the input and the output stage if RF gets too low, decreasing the bandwidth. Figure 62 shows the recommended
RF versus NG for both ±5 V and a single +5-V operation. The optimum target feedback impedance for +5-V
operation used in Equation 8 is 663 Ω, while the typical buffer output impedance is 32 Ω. The values for RF
versus gain shown are approximately equal to the values used to generate the typical characteristic curves. In
some cases, the values used differ slightly from that shown here, in that the values used in the typical
characteristics are also correcting for board parasitics not considered in the simplified analysis leading to
Equation 11. The values shown in Figure 62 give a good starting point for designs where bandwidth optimization
is desired and a flat frequency response is needed.
600
Feedback Resistor (Ω)
500
VS = ±5V
400
VS = +5V
300
200
100
0
0
2
4
6
8
10
12
14
Noise Gain (V/V)
16
18
20
Figure 62. Recommended Feedback Resistor vs Noise Gain
The total impedance presented to the inverting input can adjust the closed-loop signal bandwidth. Inserting a
series resistor between the inverting input and the summing junction increases the feedback impedance
(denominator of Equation 10), decreasing the bandwidth. The internal buffer output impedance for the OPA695 is
slightly influenced by the source impedance looking out of the noninverting input terminal. High source resistors
increase RI, decreasing the bandwidth. For those single-supply applications which develop a midpoint bias at the
non-inverting input through high-valued resistors, the decoupling capacitor is essential for power-supply ripple
rejection, non-inverting input noise current shunting, and minimizing the high-frequency value for RI in Figure 61.
Inverting feedback optimization is complicated by the impedance matching requirement at the input, as shown in
Figure 49. The resistor values shown in Table 3 must be used in this case.
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Application Information (continued)
9.1.5.2 Output Current and Voltage
The OPA695 provides output voltage and current capabilities consistent with driving doubly-terminated 50-Ω
lines. For a 100-Ω load at a gain of +8 (see Figure 48), the total load is the parallel combination of the 100-Ω
load and the 456-Ω total feedback network impedance. This 82-Ω load requires no more than 45-mA output
current to support the ±3.7-V minimum output voltage swing specified for 100-Ω loads. This is well below the
minimum ±90-mA specifications.
The specifications described above, though familiar in the industry, consider voltage and current limits
separately. In many applications, it is the voltage × current, or V-I, product which is more relevant to circuit
operation. Refer to Figure 21. The X and Y axes of this graph show the zero-voltage output current limit and the
zero-current output voltage limit, respectively. The four quadrants provide a more detailed view of the OPA695
output drive capabilities. Superimposing resistor load lines onto the plot shows the available output voltage and
current for specific loads.
The minimum specified output voltage and current overtemperature are set by worst-case simulations at the cold
temperature extreme. Only at cold startup does the output current and voltage decrease to the numbers shown in
the specification tables. As the output transistors deliver power, the junction temperatures increase, decreasing
the VBEs (increasing the available output voltage swing) and increasing the current gains (increasing the
available output current). In steady-state operation, the available output voltage and current are always be
greater than that shown in the over-temperature specifications, because the output stage junction temperatures
are higher than the minimum specified operating ambient.
To maintain maximum output-stage linearity, no output short-circuit protection is provided. This is not normally a
problem, as most applications include a series-matching resistor at the output that limits the internal power
dissipation if the output side of this resistor is shorted to ground. However, shorting the output pin directly to the
adjacent positive power supply pin will, in most cases, destroy the amplifier. If additional short-circuit protection is
required, consider a small series resistor in the power-supply leads. Under heavy output loads, this reduces the
available output voltage swing. A 5-Ω series resistor in each power-supply lead limits the internal power
dissipation to less than 1W for an output short circuit, while decreasing the available output voltage swing only
0.25 V for up to 50-mA desired load currents. Always place the 0.1-μF power supply decoupling capacitors
directly on the supply pins after these supply current-limiting resistors.
9.1.5.3 Driving Capacitive Loads
One of the most demanding, and yet very common, load conditions for an operational amplifier is capacitive
loading. Often, the capacitive load is the input of an A/D converter,including additional external capacitance
which may be recommended to improve A/D linearity. A high-speed, high open-loop gain amplifier like the
OPA695 can be susceptible to decreased stability and closed-loop response peaking when a capacitive load is
placed directly on the output pin. When the open-loop output resistance of the amplifier is considered, this
capacitive load introduces an additional pole in the signal path that can decrease the phase margin. Several
external solutions to this problem have been suggested. When the primary considerations are frequency
response flatness, pulse response fidelity, and distortion, the simplest and most effective solution is to isolate the
capacitive load from the feedback loop by inserting a series isolation resistor between the amplifier output and
the capacitive load. This does not eliminate the pole from the loop response, but shifts it and adds a zero at a
higher frequency. The additional zero acts to cancel the phase lag from the capacitive load pole, thus increasing
the phase margin and improving stability.
The typical characteristics show the recommended RS versus capacitive load and the resulting frequency
response at the load. Parasitic capacitive loads greater than 2 pF can begin to degrade the performance of the
OPA695. Long PCB traces, unmatched cables, and connections to multiple devices can exceed this value.
Always consider this effect carefully and add the recommended series resistor as close as possible to the
OPA695 output pin (see Layout Guidelines).
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Application Information (continued)
9.1.5.4 Distortion Performance
The OPA695 provides good distortion performance into a 100-Ω load on ±5-V supplies. Compared to other
solutions, the OPA695 holds lower distortion at higher frequencies (> 20 MHz). Generally, until the fundamental
signal reaches very high frequency or power levels, the 2nd-harmonic dominates the distortion with a negligible
3rd-harmonic component. Focusing on the 2nd-harmonic, increasing the load impedance directly improves
distortion: the total load includes the feedback network. In the non-inverting configuration (see Figure 48), this is
the sum of RF + RG, while in the inverting configuration, it is only RF. Also, providing an additional supply
decoupling capacitor (0.01 μF) between the supply pins (for bipolar operation) improves the 2nd-order distortion
slightly (3 dB to 6 dB).
In most operational amplifiers, increasing the output voltage swing directly increases harmonic distortion. The
typical performance curves show the 2nd-harmonic increasing at a little less than the expected 2x rate, while the
3rd-harmonic increases at a little less than the expected 3x rate. Where the test power doubles, the difference
between it and the 2nd harmonic decreases less than the expected 6 dB, while the difference between it and the
3rd decreases by less than the expected 12 dB.
The OPA695 has extremely low 3rd-order harmonic distortion. This also gives a high 2-tone, 3rd-order
intermodulation intercept, as shown in the typical characteristic curves. This intercept curve is defined at the 50-Ω
load when driven through a 50-Ω matching resistor to allow direct comparisons to RF MMIC devices, and is
shown for both gains of ±8. There is a slight improvement in intercept by operating the OPA695 in the inverting
mode. The output matching resistor attenuates the voltage swing from the output pin to the load by 6 dB. If the
OPA695 drives directly into the input of a high impedance device, such as an ADC, this 6-dB attenuation is not
taken. Under these conditions, the intercept increases by a minimum 6 dBm.
The intercept predicts the intermodulation products for two closely-spaced frequencies. If the two test
frequencies, F1 and F2, are specified in terms of average and delta frequency, FO = (F1 + F2)/2 and ΔF = |F2 – F1|
/2, the two 3rd-order, close-in spurious tones will appear at FO ±3 × ΔF. The difference between two equal testtone power levels and these intermodulation spurious power levels is given by ΔdBc = 2 × (OP3 – PO), where
OP3 is the intercept taken from the typical characteristic curve and PO is the power level in dBm at the 50-Ω load
for one of the two closely-spaced test frequencies. For example, at 50 MHz, gain of –8, the OPA695 has an
intercept of 42 dBm at a matched 50-Ω load. If the full envelope of the two frequencies must be 2 VPP, this
requires each tone to be 4 dBm. The 3rd-order intermodulation spurious tones are then 2 × (42 – 4) = 76 dBc
below the test-tone power level (–72 dBm). If this same 2-VPP 2-tone envelope were delivered directly into the
input of an ADC without the matching loss or the loading of the 50-Ω network, the intercept would increase to at
least 48 dBm. With the same signal and gain conditions, but now driving directly into a light load, the 3rd-order
spurious tones are then at least 2 × (48 – 4) = 88 dBc below the 4-dBm test-tone power levels centered on 50
MHz. Tests have shown that, in reality, the 3rd-order spurious levels are much lower due to the lighter loading
presented by most ADCs.
9.1.5.5 Noise Performance
The OPA695 offers an excellent balance between voltage and current noise terms to achieve low output noise.
The inverting current noise (22 pA/√Hz) is lower than most other current-feedback operational amplifiers, while
the input voltage noise (1.8 nV/√Hz) is lower than any unity-gain stable, wideband, voltage-feedback operational
amplifier. This low-input voltage noise was achieved at the price of a higher noninverting input current noise (18
pA/√Hz). As long as the AC source impedance looking out of the noninverting node is less than 50 Ω, this current
noise does not contribute significantly to the total output noise. The operational amplifier input voltage noise and
the two input current noise terms combine to give low output noise under a wide variety of operating conditions.
Figure 63 shows the operational amplifier noise analysis model with all the noise terms included. In this model,
all noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz.
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Application Information (continued)
ENI
EO
OPA695
RS
IBN
ERS
RF
√4kTRS
RG
4kT
RG
IBI
√4kTRF
4kT = 1.6E –20J
at 290°K
Figure 63. Operational Amplifier Noise Figure Analysis Model
The total output spot-noise voltage can be computed as the square root of the sum of all squared output noise
voltage contributors. Equation 12 shows the general form for the output noise voltage using the terms shown in
Figure 59.
(E
EO =
NI
2
2
)
2
+ (IBNR S ) + 4kTRS GN2 + (IBIRF ) + 4kTRF GN
(12)
Dividing this expression by the noise gain (NG = (1 + RF/RG)) gives the equivalent input referred spot-noise
voltage at the noninverting input, as shown in Equation 13:
√
2
2
( I NGR )
EN = ENI2 + (IBNR S ) + 4kTR S +
BI F
+
4kTRF
NG
(13)
Evaluating these two equations for the OPA695 circuit and component values shown in Figure 48 gives a total
output spot-noise voltage of 18.7 nV/√Hz and a total equivalent input spot-noise voltage of 2.3 nV/√Hz. This total
input referred spot-noise voltage is higher than the 1.8-nV/√Hz specification for the operational amplifier voltage
noise alone. This reflects the noise added to the output by the inverting current noise times the feedback resistor.
If the feedback resistor is reduced in high-gain configurations (as suggested previously), the total input referred
voltage noise given by Equation 13 just approaches the 1.8 nV/√Hz of the operational amplifier itself. For
example, going to a gain of +20 (using RF = 200 Ω) gives a total input referred noise of 2.0 nV/√Hz.
For a more complete discussion of operational amplifier noise calculation, see TI Application Note, SBOA066,
Noise Analysis for High Speed Op Amps, available through www.ti.com.
9.1.5.6 DC Accuracy and Offset Control
A current-feedback operational amplifier such as the OPA695 provides exceptional bandwidth in high gains,
giving fast pulse settling but only moderate DC accuracy. The typical specifications show an input offset voltage
comparable to high-speed voltage-feedback amplifiers; however, the two input bias currents are somewhat
higher and are unmatched. Although bias current cancellation techniques are effective with most voltagefeedback operational amplifiers, they do not generally reduce the output DC offset for wideband current-feedback
operational amplifiers. Because the two input bias currents are unrelated in both magnitude and polarity,
matching the source impedance looking out of each input to reduce their error contribution to the output is
ineffective. Evaluating the configuration of Figure 48, using a worst-case +25°C input offset voltage and the two
input bias currents, gives a worst-case output offset range equal to:
±(NG × VOS) + (IBN × RS/2 × NG) ±(IBI × RF)
where
•
NG = noninverting signal gain
(14)
= ±(8 × 3.0 mV) ± (30 µA × 25 Ω × 8) ±(402 Ω × 60 µA)
= ±24 mV ± 1.6 mV ± 24 mV
= ±54 mV
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Application Information (continued)
A fine-scale output offset null, or DC operating point adjustment, is often required. Numerous techniques are
available for introducing DC offset control into an operational amplifier circuit. Most simple adjustment techniques
do not correct for temperature drift.
9.1.5.7 Power Shutdown Operation
The OPA695 provides an optional power shutdown feature that can be used to reduce system power. If the VDIS
control pin is left unconnected, the OPA695 operates normally. This shutdown is intended only as a powersaving feature. Forward path isolation is effective for small signals. Large signal isolation is not ensured. Using
this feature to multiplex two or more outputs together is not recommended. Large signals applied to the shutdown
output stages can turn on parasitic devices, degrading signal linearity for the desired channel.
Turn-on time is quick from the shutdown condition, typically < 60 ns. Turn-off time is strongly dependent on the
external circuit configuration, but is typically 200 ns for the circuit of Figure 48.
To shut down, the control pin must be asserted low. This logic control is referenced to the positive supply, as
shown in the simplified circuit of Figure 64.
+VS
8kΩ
Q1
120kΩ
17kΩ
VDIS
IS
Control
–VS
Figure 64. Operational Amplifier Noise Figure Analysis Model
In normal operation, base current to Q1 is provided through the 120-kΩ resistor, while the emitter current through
the 8-kΩ resistor sets up a voltage drop that is inadequate to turn on the two diodes in the Q1 emitter. As VDIS is
pulled low, additional current is pulled through the 8-kΩ resistor, eventually turning on these two diodes (≈ 180
μA). At this point, any further current pulled out of VDIS goes through those diodes holding the emitter-base
voltage of Q1 at approximately 0 V. This shuts off the collector current out of Q1, turning the amplifier off. The
supply current in the shutdown mode is only that required to operate the circuit of Figure 64.
When disabled, the output and input nodes go to a high impedance state. If the OPA695 is operating in a gain of
+1, this will show a very high impedance (3 pF || 1 MΩ) at the output and exceptional signal isolation. If operating
at a gain greater than +1, the total feedback network resistance (RF + RG) appears as the impedance looking
back into the output, but the circuit will still show very high forward and reverse isolation. If configured as an
inverting amplifier, the input and output are connected through the feedback network resistance (RF + RG), giving
relatively poor input to output isolation.
9.1.5.8 Thermal Analysis
The OPA695 does not require external heatsinking for most applications. Maximum desired junction temperature
sets the maximum allowed internal power dissipation as described below. In no case should the maximum
junction temperature be allowed to exceed 150°C.
Operating junction temperature (TJ) is given by TA + PD × θJA. The total internal power dissipation (PD) is the sum
of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver load power.
Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL
depends on the required output signal and load. However, for a grounded resistive load, PDL would be at a
maximum when the output is fixed at a voltage equal to one-half of either supply voltage (for equal bipolar
supplies). Under this condition, PDL = VS 2/(4 × RL), where RL includes feedback network loading.
Note that it is the power in the output stage and not into the load that determines internal power dissipation.
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Application Information (continued)
As an absolute worst-case example, compute the maximum TJ using an OPA695IDBV (SOT23-6 package) in the
circuit of Figure 48 operating at the maximum specified ambient temperature of +85°C and driving a grounded
100-Ω load.
PD = 10 V × 14.1 mA + 52/(4 × (100 Ω || 458 Ω)) = 217 mW
(15)
Maximum TJ = +85°C + (0.22 W × 150°C/W) = 118°C
(16)
This maximum operating junction temperature is well below most system level targets. Most applications are
lower as an absolute worst-case output stage power was assumed in this calculation.
9.2 Typical Application
+12V
5kΩ
50Ω
1000pF
5kΩ OPA695
0.1µF
PO
Matching
Network
50Ω
50Ω
Source 1000pF
SAW
Filter
50Ω
400Ω
PO
PI
PI
= 12dB – (SAW Loss)
Figure 65. IF Amplifier Driving SAW Filter
9.2.1 Design Requirements
9.2.1.1 Saw Filter Buffer
One common requirement in an IF strip is to buffer the output of a mixer with enough gain to recover the
insertion loss of a narrowband SAW filter. Figure 65 shows one possible configuration driving a SAW filter.
Figure 53 shows the intercept at the 50-Ω load. Operating in the inverting mode at a voltage gain of –8 V/V, this
circuit provides a 50-Ω input match using the gain set resistor, has the feedback optimized for maximum
bandwidth (700 MHz in this case), and drives through a 50-Ω output resistor into the matching network at the
input of the SAW filter. If the SAW filter gives a 12-dB insertion loss, a net gain of 0 dB to the 50-Ω load at the
output of the SAW (which could be the input impedance of the next IF amplifier or mixer) is delivered in the
passband of the SAW filter. Using the OPA695 in this application isolates the first mixer from the impedance of
the SAW filter and provides very low two-tone, 3rd-order spurious levels in the SAW filter bandwidth. Inverting
operation gives the broadest bandwidth up to a gain of –12 V/V (15.6 dB). Noninverting operation gives higher
bandwidth at gain settings higher than this, but will also give a slight reduction in intercept and noise figure
performance.
9.2.2 Detailed Design Procedure
The design procedure begins with calculating the required signal gain and signal swing. Once the gain and swing
requirements are determined the appropriate amplifier is selected along with the required supply voltage. Due to
the input impedance of 50 Ω the gain and the input impedance require a feedback resistor value of 400 Ω.
In this application the supply voltage is 12 V single ended. In order to provide the proper DC operating point it is
necessary to apply a mid supply voltage to the non inverting input. This is accomplished by using a resistive
voltage divider composed of two 1% precision 5-kΩ resistors along with two ceramic bypass capacitors. These
components provide an accurate and low AC impedance reference voltage for the non inverting input. The
inverting input requires only an AC coupling capacitor to isolate the 6 V operating voltage from the signal source.
In this example a ceramic 1000-pF capacitor is used.
The circuit shown in Figure 65 shows an output resistor value of 50 Ω. This resistor will need to be adjusted to
accommodate the SAW input impedance. Additional L/C components may be required as well, consult the SAW
manufacturer's design guidelines for more details.
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Typical Application (continued)
9.2.3 Application Curve
Output Intercept (dBm)
50
40
30
20
10
0
50
100
150
200
250
Center Frequency (MHz)
Figure 66. 2-Tone, 3rd-Order Intermodulation Intercept
10 Power Supply Recommendations
High-speed amplifiers require low inductance power supply traces and low ESR bypass capacitors. When
possible both power and ground planes must be used in the printed circuit board design and the power plane
must be adjacent to the ground plane in the board stack-up. The power supply voltage must be centered on the
desired amplifier output voltage, so for ground referenced output signals, split supplies are required. The power
supply voltage must be from 5 V to 12 V.
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11 Layout
11.1 Layout Guidelines
Achieving optimum performance with a high-frequency amplifier like the OPA695 requires careful attention to
board layout parasitics and external component types. Recommendations that will optimize performance include:
• Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on
the output and inverting input pins can cause instability; on the non-inverting input, it can react with the
source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around
the signal I/O pins must be opened in all of the ground and power planes around those pins. Otherwise,
ground and power planes must be unbroken elsewhere on the board.
• Minimize the distance (< 0.25") from the power supply pins to high frequency 0.1-μF decoupling
capacitors. At the device pins, the ground and power plane layout must not be in close proximity to the
signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the
decoupling capacitors. The power-supply connections must always be decoupled with these capacitors. An
optional supply-decoupling capacitor across the two power supplies (for bipolar operation) improves 2ndharmonic distortion performance. Larger (2.2 μF to 6.8 μF) decoupling capacitors, effective at a lower
frequency, must also be used on the main supply pins. These may be placed somewhat farther from the
device, and may be shared among several devices in the same area of the PCB.
• Careful selection and placement of external components will preserve the high frequency
performance of the OPA695. Resistors must be a low reactance type. Surface-mount resistors work best
and allow a tighter overall layout. Metal-film and carbon composition, axially-leaded resistors can also provide
good high frequency performance. Keep their leads and PCB trace length as short as possible. Never use
wirewound-type resistors in a high frequency application. Because the output pin and inverting input pin are
the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as
close as possible to the output pin. Other network components, such as noninverting input termination
resistors, must also be placed close to the package. Where double-side component mounting is allowed,
place the feedback resistor directly under the package on the other side of the board between the output and
inverting input pins. The frequency response is primarily determined by the feedback resistor value.
Increasing its value reduces the bandwidth, while decreasing it gives a more peaked frequency response.
The 402-Ω feedback resistor (used in the typical performance specifications at a gain of +8 on ±5-V supplies)
is a good starting point for design. Note that a 523-Ω feedback resistor, rather than a direct short, is required
for the unity gain follower application. A current-feedback operational amplifier requires a feedback resistor,
even in the unity gain follower configuration, to control stability.
• Connections to other wideband devices on the board may be made with short direct traces or through
onboard transmission lines. For short connections, consider the trace and the input to the next device as a
lumped capacitive load. Relatively wide traces (50 mils to 100 mils) must be used, preferably with ground and
power planes opened up around them. Estimate the total capacitive load and set RS from the plot of
Figure 40. Low parasitic capacitive loads (< 5 pF) may not need an RS as the OPA695 is nominally
compensated to operate with a 2-pF parasitic load. If a long trace is required, and the 6-dB signal loss
intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission
line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline
layout techniques). A 50-Ω environment is usually not necessary on board. In fact, a higher impedance
environment improves distortion, as shown in the distortion versus load plots. With a characteristic board
trace impedance defined (based on board material and trace dimensions), use a matching series resistor into
the trace from the output of the OPA695. Also use terminating shunt resistor at the input of the destination
device. Remember that the terminating impedance will be the parallel combination of the shunt resistor and
the input impedance of the destination device; this total effective impedance must be set to match the trace
impedance. The high output voltage and current capability of the OPA695 allows multiple destination devices
to be handled as separate transmission lines, each with their own series and shunt terminations. If the 6-dB
attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at
the source end only. Treat the trace as a capacitive load in this case, and set the series resistor value as
shown in the plot of Figure 40. This will not preserve signal integrity as well as a doubly-terminated line. If the
input impedance of the destination device is low, there will be some signal attenuation due to the voltage
divider formed by the series output into the terminating impedance.
• Socketing a high-speed part like the OPA695 is not recommended. The additional lead length and pin-topin capacitance introduced by the socket can create a troublesome parasitic network, which can make it
almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the
OPA695 directly onto the board.
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Layout Guidelines (continued)
11.1.1 Input and ESD Protection
The OPA695 is built using a very high-speed, complementary bipolar process. The internal junction breakdown
voltages are relatively low for these small geometry devices. These breakdowns are reflected in the Absolute
Maximum Ratings where an absolute maximum ±6.5-V supply is reported. All device pins have limited ESD
protection using internal diodes to the power supplies, as shown in Figure 67.
These diodes also provide moderate protection to input overdrive voltages above the supplies. The protection
diodes can typically support 30-mA continuous current. Where higher currents are possible (for example, in
systems with ±15-V supply parts driving into the OPA695), current-limiting series resistors must be added into
the two inputs. Keep these resistor values as low as possible as high values degrade both noise performance
and frequency response.
+V CC
External
Pin
Internal
Circuitry
–V CC
Figure 67. Internal ESD Protection
11.2 Layout Example
As detailed in Layout Guidelines and illustrated in Figure 68, the input termination resistor, output resistor and
bypass capacitors must be placed close to the amplifier. Power and ground planes are placed under the
amplifier, but must be removed under the input and output pins as shown in Figure 68.
Figure 68. SBOS293 Layout
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12 Device and Documentation Support
12.1 Device Support
12.1.1 Design-In Tools
12.1.1.1 Demonstration Fixtures
Two printed circuit boards (PCBs) are available to assist in the initial evaluation of circuit performance using the
OPA695 in its two package options. Both of these are offered free of charge as unpopulated PCBs, delivered
with a user's guide. The summary information for these fixtures is shown in Table 4.
Table 4. Demonstration Boards
ORDERING NUMBER
USER'S GUIDE
LITERATURE NUMBER
VSSOP-8
DEM-OPA-SO-1B
SBOU026
SOT23-6
DEM-OPA-SOT-1B
SBOU027
PRODUCT
PACKAGE
OPA695ID
OPA691IDBV
The demonstration fixtures can be requested at the Texas Instruments web site (www.ti.com) through the
OPA695 product folder.
12.2 Documentation Support
12.2.1 Related Documentation
For related documentation, see the following:
• Absolute Maximum Ratings for Soldering, SNOA549
• Current Feedback Op Amp Applications Circuit Guide, Application Note OA--07, SNOA365
• Frequent Faux Pas in Applying Wideband Current Feedback Amplifiers, Application Note OA-15, SNOA367
• Noise Analysis for Comlinear Amplifiers, Application Note OA-12, SNOA375
• Semiconductor and IC Package Thermal Metrics, SPRA953
12.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.4 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
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13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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