AD EVAL-ADF7021DBZ6 High performance narrow-band transceiver ic Datasheet

High Performance
Narrow-Band Transceiver IC
ADF7021
Data Sheet
FEATURES
On-chip VCO and fractional-N PLL
On-chip, 7-bit ADC and temperature sensor
Fully automatic frequency control loop (AFC)
Digital received signal strength indication (RSSI)
Integrated Tx/Rx switch
0.1 µA leakage current in power-down mode
Low power, narrow-band transceiver
Frequency bands using dual VCO
80 MHz to 650 MHz
862 MHz to 950 MHz
Modulation schemes
2FSK, 3FSK, 4FSK, MSK
Spectral shaping
Gaussian and raised cosine filtering
Data rates supported
0.05 kbps to 32.8 kbps
2.3 V to 3.6 V power supply
Programmable output power
−16 dBm to +13 dBm in 63 steps
Automatic PA ramp control
Receiver sensitivity
−130 dBm at 100 bps, 2FSK
−122 dBm at 1 kbps, 2FSK
−113 dBm at 25 kbps, raised cosine 2FSK
Patent pending, on-chip image rejection calibration
APPLICATIONS
Narrow-band standards
ETSI EN 300 220, FCC Part 15, FCC Part 90, FCC Part 95,
ARIB STD-T67
Low cost, wireless data transfer
Remote control/security systems
Wireless metering
Private mobile radio
Wireless medical telemetry service (WMTS)
Keyless entry
Home automation
Process and building control
Pagers
FUNCTIONAL BLOCK DIAGRAM
CE
RSET
TEMP
SENSOR
RLNA
MUX
2FSK
3FSK
4FSK
RSSI/
LOG AMP
IF FILTER
RFINB
MUXOUT
LDO(1:4)
TEST MUX
7-BIT ADC
LNA
RFIN
CREG(1:4)
CLOCK
AND DATA
RECOVERY
DEMODULATOR
TxRxCLK
Tx/Rx
CONTROL
TxRxDATA
SWD
GAIN
AGC
CONTROL
SLE
SERIAL
PORT
AFC
CONTROL
PA RAMP
DIV P
VCO1
VCO2
VCOIN
SCLK
GAUSSIAN/
RAISED COSINE
FILTER
3FSK
ENCODING
CP
PFD
DIV R
L2
2FSK
3FSK
4FSK
MOD CONTROL
Σ-Δ
MODULATOR
÷2
MUX
L1
N/N + 1
SREAD
CPOUT
CLK
DIV
OSC
OSC1
OSC2
CLKOUT
05876-001
÷1/÷2
RFOUT
SDATA
Figure 1.
Rev. B
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ADF7021
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Receiver Setup............................................................................. 34
Applications ....................................................................................... 1
Demodulator Considerations ................................................... 36
Functional Block Diagram .............................................................. 1
AFC Operation ........................................................................... 36
Revision History ............................................................................... 3
Automatic Sync Word Detection (SWD) ................................ 37
General Description ......................................................................... 4
Applications Information .............................................................. 38
Specifications..................................................................................... 5
IF Filter Bandwidth Calibration ............................................... 38
RF and PLL Specifications........................................................... 5
LNA/PA Matching ...................................................................... 38
Transmission Specifications ........................................................ 6
Image Rejection Calibration ..................................................... 39
Receiver Specifications ................................................................ 8
Packet Structure and Coding .................................................... 41
Digital Specifications ................................................................. 10
Programming After Initial Power-Up ..................................... 41
General Specifications ............................................................... 11
Applications Circuit ................................................................... 44
Timing Characteristics .............................................................. 11
Serial Interface ................................................................................ 45
Absolute Maximum Ratings.......................................................... 15
Readback Format........................................................................ 45
ESD Caution ................................................................................ 15
Interfacing to Microcontroller/DSP ........................................ 46
Pin Configuration and Function Descriptions ........................... 16
Register 0—N Register............................................................... 47
Typical Performance Characteristics ........................................... 18
Register 1—VCO/Oscillator Register ...................................... 48
Frequency Synthesizer ................................................................... 22
Register 2—Transmit Modulation Register ............................ 49
Reference Input ........................................................................... 22
Register 3—Transmit/Receive Clock Register ........................ 50
MUXOUT.................................................................................... 23
Register 4—Demodulator Setup Register ............................... 51
Voltage Controlled Oscillator (VCO) ...................................... 24
Register 5—IF Filter Setup Register ......................................... 52
Choosing Channels for Best System Performance ................. 25
Register 6—IF Fine Cal Setup Register ................................... 53
Transmitter ...................................................................................... 26
Register 7—Readback Setup Register ...................................... 54
RF Output Stage .......................................................................... 26
Register 8—Power-Down Test Register .................................. 55
Modulation Schemes .................................................................. 26
Register 9—AGC Register ......................................................... 56
Spectral Shaping ......................................................................... 28
Register 10—AFC Register ....................................................... 57
Modulation and Filtering Options ........................................... 29
Register 11—Sync Word Detect Register ................................ 58
Transmit Latency ........................................................................ 29
Register 12—SWD/Threshold Setup Register ........................ 58
Test Pattern Generator ............................................................... 29
Register 13—3FSK/4FSK Demod Register ............................. 59
Receiver Section .............................................................................. 30
Register 14—Test DAC Register ............................................... 60
RF Front End ............................................................................... 30
Register 15—Test Mode Register ............................................. 61
IF Filter......................................................................................... 30
Outline Dimensions ....................................................................... 62
RSSI/AGC .................................................................................... 31
Ordering Guide .......................................................................... 62
Demodulation, Detection, and CDR ....................................... 32
Rev. B | Page 2 of 64
Data Sheet
ADF7021
REVISION HISTORY
4/13—Rev. A to Rev. B
Changes to Figure 10 ......................................................................16
Updated Outline Dimensions ........................................................62
Changes to Ordering Guide ...........................................................62
9/07—Rev. 0 to Rev. A
Change to UART/SPI Mode Section.............................................14
Changes to Figure 10 ......................................................................16
Change to Table 8 ............................................................................16
Changes to Figure 12 ......................................................................18
Change to Internal Inductor VCO Section ..................................24
Changes to Figure 40 ......................................................................26
Changes to Figure 47 ......................................................................32
Change to Table 19 ..........................................................................34
Changes to Figure 56 ......................................................................44
Change to SPI Mode Section .........................................................46
Changes to Figure 59 ......................................................................46
Changes to Figure 60 ......................................................................46
Change to Register 3—Transmit/Receive Clock
Register Section ............................................................................... 50
Change to Register 4—Demodulator Setup
Register Section ............................................................................... 51
Change to Register 7—Readback Setup Register ........................ 54
Change to Register 13—3FSK/4FSK Demod
Register Heading ............................................................................. 59
3/07—Revision 0: Initial Version
Rev. B | Page 3 of 64
ADF7021
Data Sheet
GENERAL DESCRIPTION
The ADF7021 is a high performance, low power, highly integrated
2FSK/3FSK/4FSK transceiver. It is designed to operate in the
narrow-band, license-free ISM bands, and in the licensed bands
with frequency ranges of 80 MHz to 650 MHz and 862 MHz to
950 MHz. The part has both Gaussian and raised cosine transmit
data filtering options to improve spectral efficiency for narrowband applications. It is suitable for circuit applications targeted
at European ETSI EN 300 220, the Japanese ARIB STD-T67,
the Chinese short range device regulations, and the North
American FCC Part 15, Part 90, and Part 95 regulatory standards.
A complete transceiver can be built using a small number of
external discrete components, making the ADF7021 very
suitable for price sensitive and area sensitive applications.
The range of on-chip FSK modulation and data filtering options
allows users greater flexibility in their choice of modulation
schemes while meeting tight spectral efficiency requirements.
The ADF7021 also supports protocols that dynamically switch
between 2FSK/3FSK/4FSK to maximize communication range
and data throughput.
The transmit section contains dual voltage controlled oscillators
(VCOs) and a low noise fractional-N PLL with an output resolution
of <1 ppm. The ADF7021 has a VCO using an internal LC tank
(431 MHz to 475 MHz, 862 MHz to 950 MHz) and a VCO using
an external inductor as part of its tank circuit (80 MHz to
650 MHz). The dual VCO design allows dual-band operation
where the user can transmit and/or receive at any frequency
supported by the internal inductor VCO and can also transmit
and/or receive at a particular frequency band supported by the
external inductor VCO.
The frequency agile PLL allows the ADF7021 to be used in
frequency hopping spread spectrum (FHSS) systems.
Both VCOs operate at twice the fundamental frequency to
reduce spurious emissions and frequency pulling problems.
The transmitter output power is programmable in 63 steps
from −16 dBm to +13 dBm, and has an automatic power ramp
control to prevent spectral splatter and help meet regulatory
standards. The transceiver RF frequency, channel spacing, and
modulation are programmable using a simple 3-wire interface.
The device operates with a power supply range of 2.3 V to 3.6 V
and can be powered down when not in use.
A low IF architecture is used in the receiver (100 kHz), which
minimizes power consumption and the external component
count, yet avoids dc offset and flicker noise at low frequencies.
The IF filter has programmable bandwidths of 12.5 kHz, 18.75 kHz,
and 25 kHz. The ADF7021 supports a wide variety of programmable features including Rx linearity, sensitivity, and IF bandwidth,
allowing the user to trade off receiver sensitivity and selectivity
against current consumption, depending on the application.
The receiver also features a patent-pending automatic frequency
control (AFC) loop with programmable pull-in range that allows
the PLL to track out the frequency error in the incoming signal.
The receiver achieves an image rejection performance of 56 dB
using a patent-pending IR calibration scheme that does not
require the use of an external RF source.
An on-chip ADC provides readback of the integrated temperature sensor, external analog input, battery voltage, and RSSI
signal, which provides savings on an ADC in some applications.
The temperature sensor is accurate to ±10°C over the full
operating temperature range of −40°C to +85°C. This accuracy
can be improved by performing a 1-point calibration at room
temperature and storing the result in memory.
Rev. B | Page 4 of 64
Data Sheet
ADF7021
SPECIFICATIONS
VDD = 2.3 V to 3.6 V, GND = 0 V, TA = TMIN to TMAX, unless otherwise noted. Typical specifications are at VDD = 3 V, TA = 25°C.
All measurements are performed with the EVAL-ADF7021DBx using the PN9 data sequence, unless otherwise noted.
RF AND PLL SPECIFICATIONS
Table 1.
Parameter
RF CHARACTERISTICS
Max
Unit
650
950
325
475
26/30
MHz
MHz
MHz
MHz
MHz
Test Conditions/Comments
See Table 9 for required VCO_BIAS and
VCO_ADJUST settings
External inductor VCO
Internal inductor VCO
External inductor VCO, RF divide-by-2 enabled
Internal inductor VCO, RF divide-by-2 enabled
Crystal reference/external reference
58
29
27
6
MHz/V
MHz/V
MHz/V
MHz/V
VCO_ADJUST = 0, VCO_BIAS = 8
VCO_ADJUST = 0, VCO_BIAS = 8
VCO_ADJUST = 0, VCO_BIAS = 3
VCO_ADJUST = 0, VCO_BIAS = 2
−97
dBc/Hz
433 MHz, Internal Inductor VCO
−103
dBc/Hz
426 MHz, External Inductor VCO
−95
dBc/Hz
Phase Noise (Out-of-Band)
−124
dBc/Hz
10 kHz offset, PA = 10 dBm, VDD = 3.0 V,
PFD = 19.68 MHz, VCO_BIAS = 8
10 kHz offset, PA = 10 dBm, VDD = 3.0 V,
PFD = 19.68 MHz, VCO_BIAS = 8
10 kHz offset, PA = 10 dBm, VDD = 3.0 V,
PFD = 9.84 MHz, VCO_BIAS = 3
1 MHz offset, fRF = 433 MHz, PA = 10 dBm,
VDD = 3.0 V, PFD = 19.68 MHz, VCO_BIAS = 8
Normalized In-Band Phase Noise Floor 3
PLL Settling
−203
40
dBc/Hz
µs
Frequency Ranges (Direct Output)
Frequency Ranges (RF Divide-by-2 Mode)
Phase Frequency Detector (PFD) Frequency 1
PHASE-LOCKED LOOP (PLL)
VCO Gain 2
868 MHz, Internal Inductor VCO
434 MHz, Internal Inductor VCO
426 MHz, External Inductor VCO
160 MHz, External Inductor VCO
Phase Noise (In-Band)
868 MHz, Internal Inductor VCO
REFERENCE INPUT
Crystal Reference 4
External Oscillator4, 5
Crystal Start-Up Time 6
XTAL Bias = 20 µA
XTAL Bias = 35 µA
Input Level for External Oscillator 7
OSC1
OSC2
ADC PARAMETERS
INL
DNL
Min
Typ
160
862
80
431
RF/256
3.625
3.625
26
30
Measured for a 10 MHz frequency step to within
5 ppm accuracy, PFD = 19.68 MHz, loop
bandwidth (LBW) = 100 kHz
MHz
MHz
0.930
0.438
ms
ms
10 MHz XTAL, 33 pF load capacitors, VDD = 3.0 V
10 MHz XTAL, 33 pF load capacitors, VDD = 3.0 V
0.8
CMOS levels
V p-p
V
Clipped sine wave
±0.4
±0.4
LSB
LSB
VDD = 2.3 V to 3.6 V, TA = 25°C
VDD = 2.3 V to 3.6 V, TA = 25°C
The maximum usable PFD at a particular RF frequency is limited by the minimum N divide value.
VCO gain measured at a VCO tuning voltage of 1 V. The VCO gain varies across the tuning range of the VCO. The software package ADIsimPLL™ can be used to model this
variation.
3
This value can be used to calculate the in-band phase noise for any operating frequency. Use the following equation to calculate the in-band phase noise performance
as seen at the PA output: −203 + 10 log(fPFD) + 20 logN.
4
Guaranteed by design. Sample tested to ensure compliance.
5
A TCXO, VCXO, or OCXO can be used as an external oscillator.
6
Crystal start-up time is the time from chip enable (CE) being asserted to correct clock frequency on the CLKOUT pin.
7
Refer to the Reference Input section for details on using an external oscillator.
1
2
Rev. B | Page 5 of 64
ADF7021
Data Sheet
TRANSMISSION SPECIFICATIONS
Table 2.
Parameter
DATA RATE
2FSK, 3FSK
4FSK
MODULATION
Frequency Deviation (fDEV) 3
Deviation Frequency Resolution
Gaussian Filter BT
Raised Cosine Filter Alpha
TRANSMIT POWER
Maximum Transmit Power 4
Transmit Power Variation vs.
Temperature
Transmit Power Variation vs. VDD
Transmit Power Flatness
Programmable Step Size
ADJACENT CHANNEL POWER (ACP)
426 MHz, External Inductor VCO
12.5 kHz Channel Spacing
25 kHz Channel Spacing
868 MHz, Internal Inductor VCO
12.5 kHz Channel Spacing
25 kHz Channel Spacing
433 MHz, Internal Inductor VCO
12.5 kHz Channel Spacing
25 kHz Channel Spacing
Min
Typ
Max
Unit
Test Conditions/Comments
0.05
0.05
25 1
32.8 2
kbps
kbps
IF_BW = 25 kHz
IF_BW = 25 kHz
0.056
0.306
56
28.26
156
kHz
kHz
Hz
PFD = 3.625 MHz
PFD = 20 MHz
PFD = 3.625 MHz
0.5
0.5/0.7
Programmable
+13
±1
dBm
dB
VDD = 3.0 V, TA = 25°C
−40°C to +85°C
±1
±1
0.3125
dB
dB
dB
2.3 V to 3.6 V at 915 MHz, TA = 25°C
902 MHz to 928 MHz, 3 V, TA = 25°C
−20 dBm to +13 dBm
−50
dBc
−50
dBc
−46
dBm
−43
dBm
−50
dBm
−47
dBm
3.9
9.9
kHz
kHz
4.4
10.2
kHz
kHz
3.9
9.5
kHz
kHz
13.2
kHz
OCCUPIED BANDWIDTH
2FSK Gaussian Data Filtering
12.5 kHz Channel Spacing
25 kHz Channel Spacing
2FSK Raised Cosine Data Filtering
12.5 kHz Channel Spacing
25 kHz Channel Spacing
3FSK Raised Cosine Filtering
12.5 kHz Channel Spacing
25 kHz Channel Spacing
4FSK Raised Cosine Filtering
25 kHz Channel Spacing
PFD = 9.84 MHz
Gaussian 2FSK modulation, measured in a ±4.25 kHz bandwidth
at ±12.5 kHz offset, 2.4 kbps PN9 data, 1.2 kHz frequency deviation,
compliant with ARIB STD-T67
Gaussian 2FSK modulation, measured in a ±8 kHz bandwidth at
±25 kHz offset, 9.6 kbps PN9 data, 2.4 kHz frequency deviation,
compliant with ARIB STD-T67
PFD = 19.68 MHz
Gaussian 2FSK modulation, 10 dBm output power, measured in
a ±6.25 kHz bandwidth at ±12.5 kHz offset, 2.4 kbps PN9 data,
1.2 kHz frequency deviation, compliant with ETSI EN 300-220
Gaussian 2FSK modulation, 10 dBm output power, measured in
a ±12.5 kHz bandwidth at ±25 kHz offset, 9.6 kbps PN9 data,
2.4 kHz frequency deviation, compliant with ETSI EN 300-220
PFD = 19.68 MHz
Gaussian 2FSK modulation, 10 dBm output power, measured in
a ±6.25 kHz bandwidth at ±12.5 kHz offset, 2.4 kbps PN9 data,
1.2 kHz frequency deviation, compliant with ETSI EN 300-220
Gaussian 2FSK modulation, 10 dBm output power, measured in
a ±12.5 kHz bandwidth at ±25 kHz offset, 9.6 kbps PN9 data,
2.4 kHz frequency deviation, compliant with ETSI EN 300-220
99.0% of total mean power; 12.5 kHz channel spacing (2.4 kbps
PN9 data, 1.2 kHz frequency deviation); 25 kHz channel spacing
(9.6 kbps PN9 data, 2.4 kHz frequency deviation)
19.2 kbps PN9 data, 1.2 kHz frequency deviation
Rev. B | Page 6 of 64
Data Sheet
Parameter
SPURIOUS EMISSIONS
Reference Spurs
HARMONICS 5
Second Harmonic
Third Harmonic
All Other Harmonics
OPTIMUM PA LOAD IMPEDANCE 6
fRF = 915 MHz
fRF = 868 MHz
fRF = 450 MHz
fRF = 426 MHz
fRF = 315 MHz
fRF = 175 MHz
ADF7021
Min
Typ
Max
Unit
Test Conditions/Comments
−65
dBc
100 kHz loop bandwidth
13 dBm output power, unfiltered conductive/filtered conductive
−35/−52
−43/−60
−36/−65
dBc
dBc
dBc
39 + j61
48 + j54
98 + j65
100 + j65
129 + j63
173 + j49
Ω
Ω
Ω
Ω
Ω
Ω
Using Gaussian or raised cosine filtering. The frequency deviation should be chosen to ensure that the transmit occupied signal bandwidth is within the receiver
IF filter bandwidth.
2
Using raised cosine filtering with an alpha = 0.7. The inner frequency deviation = 1.78 kHz, and the POST_DEMOD_BW = 24.6 kHz.
3
For the definition of frequency deviation, refer to the Register 2—Transmit Modulation Register section.
4
Measured as maximum unmodulated power.
5
Conductive filtered harmonic emissions measured on the EVAL-ADF7021DBx, which includes a T-stage harmonic filter (two inductors and one capacitor).
6
For matching details, refer to the LNA/PA Matching section.
1
Rev. B | Page 7 of 64
ADF7021
Data Sheet
RECEIVER SPECIFICATIONS
Table 3.
Parameter
SENSITIVITY
Unit
Test Conditions/Comments
Bit error rate (BER) = 10−3, low noise amplifier (LNA) and
power amplifier (PA) matched separately
−130
−127
−122
−115
−110
dBm
dBm
dBm
dBm
dBm
fDEV = 1 kHz, high sensitivity mode, IF_BW = 12.5 kHz
fDEV = 1 kHz, high sensitivity mode, IF_BW = 12.5 kHz
fDEV = 1 kHz, high sensitivity mode, IF_BW = 12.5 kHz
fDEV = 4 kHz, high sensitivity mode, IF_BW = 18.75 kHz
fDEV = 10 kHz, high sensitivity mode, IF_BW = 25 kHz
−129
−127
−121
−114
−111
dBm
dBm
dBm
dBm
dBm
fDEV = 1 kHz, high sensitivity mode, IF_BW = 12.5 kHz
fDEV = 1 kHz, high sensitivity mode, IF_BW = 12.5 kHz
fDEV = 1 kHz, high sensitivity mode, IF_BW = 12.5 kHz
fDEV = 4 kHz, high sensitivity mode, IF_BW = 18.75 kHz
fDEV = 10 kHz, high sensitivity mode, IF_BW = 25 kHz
−113
dBm
fDEV = 2.4 kHz, high sensitivity mode, IF_BW = 18.75 kHz
−127
−121
−114
−113
dBm
dBm
dBm
dBm
fDEV = 1 kHz, high sensitivity mode, IF_BW = 12.5 kHz
fDEV = 1 kHz, high sensitivity mode, IF_BW = 12.5 kHz
fDEV = 4 kHz, high sensitivity mode, IF_BW = 18.75 kHz
fDEV = 10 kHz, high sensitivity mode, IF_BW = 25 kHz
−110
dBm
fDEV = 2.4 kHz, high sensitivity mode, IF_BW = 18.75 kHz,
Viterbi detection on
−110
dBm
−106
dBm
fDEV = 2.4 kHz, high sensitivity mode, IF_BW = 12.5 kHz,
alpha = 0.5, Viterbi detection on
fDEV = 4.8 kHz, high sensitivity mode, IF_BW = 18.75 kHz,
alpha = 0.5, Viterbi detection on
−112
dBm
Sensitivity at 19.6 kbps
−107
dBm
Raised Cosine 4FSK
Sensitivity at 9.6 kbps
−109
dBm
Sensitivity at 19.2 kbps
−103
dBm
Sensitivity at 32.8 kbps
−100
dBm
−3
dBm
fDEV (inner) = 1.2 kHz, high sensitivity mode,
IF_BW = 12.5 kHz, alpha = 0.5
fDEV (inner) = 1.2 kHz, high sensitivity mode,
IF_BW = 18.75 kHz, alpha = 0.5
fDEV (inner) = 1.8 kHz, high sensitivity mode, IF_BW = 25 kHz,
alpha = 0.7
Two-tone test, fLO = 860 MHz, F1 = fLO + 100 kHz,
F2 = fLO − 800 kHz
LNA_GAIN = 3, MIXER_LINEARITY = 1
−13.5
−24
dBm
dBm
LNA_GAIN = 10, MIXER_LINEARITY = 0
LNA_GAIN = 30, MIXER_LINEARITY = 0
2FSK
Sensitivity at 0.1 kbps
Sensitivity at 0.25 kbps
Sensitivity at 1 kbps
Sensitivity at 9.6 kbps
Sensitivity at 25 kbps
Gaussian 2FSK
Sensitivity at 0.1 kbps
Sensitivity at 0.25 kbps
Sensitivity at 1 kbps
Sensitivity at 9.6 kbps
Sensitivity at 25 kbps
GMSK
Sensitivity at 9.6 kbps
Raised Cosine 2FSK
Sensitivity at 0.25 kbps
Sensitivity at 1 kbps
Sensitivity at 9.6 kbps
Sensitivity at 25 kbps
3FSK
Sensitivity at 9.6 kbps
Raised Cosine 3FSK
Sensitivity at 9.6 kbps
Sensitivity at 19.6 kbps
4FSK
Sensitivity at 9.6 kbps
Min
Typ
Max
INPUT IP3
Low Gain Enhanced Linearity
Mode
Medium Gain Mode
High Sensitivity Mode
fDEV (inner) = 1.2 kHz, high sensitivity mode,
IF_BW = 12.5 kHz
fDEV (inner) = 2.4 kHz, high sensitivity mode,
IF_BW = 25 kHz
Rev. B | Page 8 of 64
Data Sheet
Parameter
ADJACENT CHANNEL REJECTION
868 MHz
ADF7021
Min
12.5 kHz Channel Spacing
25 kHz Channel Spacing
25 kHz Channel Spacing
426 MHz, External Inductor VCO
Typ
Max
Unit
25
27
39
dB
dB
dB
12.5 kHz Channel Spacing
25 kHz Channel Spacing
25 kHz Channel Spacing
CO-CHANNEL REJECTION
25
30
41
dB
dB
dB
868 MHz
IMAGE CHANNEL REJECTION
−3
dB
900 MHz
450 MHz
450 MHz, External Inductor VCO
BLOCKING
23/39
29/50
38/53
dB
dB
dB
±1 MHz
±2 MHz
±5 MHz
±10 MHz
SATURATION
(MAXIMUM INPUT LEVEL)
RSSI
Range at Input 2
69
75
78
78.5
12
dB
dB
dB
dB
dBm
−120 to
−47
±2
±3
300
dBm
Linearity
Absolute Accuracy
Response Time
AFC
Pull-In Range
Response Time
Accuracy
Rx SPURIOUS EMISSIONS 3
Internal Inductor VCO
External Inductor VCO
Test Conditions/Comments
Wanted signal is 3 dB above the sensitivity point (BER = 10−3);
unmodulated interferer is at the center of the adjacent
channel; rejection measured as the difference between
interferer level and wanted signal level in dB
12.5 kHz IF_BW
25 kHz IF_BW
18 kHz IF_BW
Wanted signal 3 dB above reference sensitivity point
(BER = 10−2); modulated interferer (1 kHz sine, ±2 kHz deviation)
at the center of the adjacent channel; rejection measured as
the difference between interferer level and reference
sensitivity level in dB
12.5 kHz IF_BW
25 kHz IF_BW
18 kHz IF_BW, compliant with ARIB STD-T67
Wanted signal (2FSK, 9.6 kbps, ±4 kHz deviation) is 10 dB
above the sensitivity point (BER = 10−3), modulated interferer
Wanted signal (2FSK, 9.6 kbps, ±4 kHz deviation) is 10 dB
above the sensitivity point (BER = 10−3); modulated interferer
(2FSK, 9.6 kbps, ±4 kHz deviation) is placed at the image
frequency of fRF − 200 kHz; interferer level is increased until
BER = 10−3
Uncalibrated/calibrated 1, VDD = 3.0 V, TA = 25°C
Uncalibrated/calibrated1,VDD = 3.0 V, TA = 25°C
Uncalibrated/calibrated1, VDD = 3.0 V, TA = 25°C
Wanted signal is 10 dB above the input sensitivity level;
CW interferer level is increased until BER = 10−3
2FSK mode, BER = 10−3
dB
dB
µs
Input power range = −100 dBm to −47 dBm
Input power range = −100 dBm to −47 dBm
See the RSSI/AGC section
kHz
Bits
kHz
The range is programmable, R10_DB[24:31]
48
0.5
−91/−91
dBm
−52/−70
dBm
−62/−72
dBm
−64/−85
dBm
<1 GHz at antenna input, unfiltered conductive/filtered
conductive
>1 GHz at antenna input, unfiltered conductive/filtered
conductive
<1 GHz at antenna input, unfiltered conductive/filtered
conductive
>1 GHz at antenna input, unfiltered conductive/filtered
conductive
0.5
1.5 × IF_BW
Input power range = −100 dBm to +12 dBm
Rev. B | Page 9 of 64
ADF7021
Parameter
LNA INPUT IMPEDANCE
fRF = 915 MHz
fRF = 868 MHz
fRF = 450 MHz
fRF = 426 MHz
fRF = 315 MHz
fRF = 175 MHz
1
2
3
Data Sheet
Min
Typ
Max
Unit
24 − j60
26 − j63
63 − j129
68 − j134
96 − j160
178 − j190
Test Conditions/Comments
RFIN to RFGND
Ω
Ω
Ω
Ω
Ω
Ω
Calibration of the image rejection used an external RF source.
For received signal levels < −100 dBm, it is recommended to average the RSSI readback value over a number of samples to improve the RSSI accuracy at low input powers.
Filtered conductive receive spurious emissions measured on the EVAL-ADF7021DBx, which includes a T-stage harmonic filter (two inductors and one capacitor).
DIGITAL SPECIFICATIONS
Table 4.
Parameter
TIMING INFORMATION
Chip Enabled to Regulator Ready
Chip Enabled to Tx Mode
TCXO Reference
XTAL
Chip Enabled to Rx Mode
Min
Max
Unit
Test Conditions/Comments
10
µs
CREG = 100 nF
32-bit register write time = 50 µs
1
2
ms
ms
32-bit register write time = 50 µs, IF filter coarse
calibration only
TCXO Reference
XTAL
Tx to Rx Turnaround Time
LOGIC INPUTS
Input High Voltage, VINH
Input Low Voltage, VINL
Input Current, IINH/IINL
Input Capacitance, CIN
Control Clock Input
LOGIC OUTPUTS
Output High Voltage, VOH
Output Low Voltage, VOL
CLKOUT Rise/Fall
CLKOUT Load
Typ
1.2
2.2
300 µs + (5 × tBIT)
ms
ms
Time to synchronized data out, includes AGC
settling and CDR synchronization; see AGC
Information and Timing section for more details;
tBIT = data bit period
0.7 × VDD
0.2 × VDD
±1
10
50
V
V
µA
pF
MHz
0.4
5
10
V
V
ns
pF
DVDD − 0.4
Rev. B | Page 10 of 64
IOH = 500 µA
IOL = 500 µA
Data Sheet
ADF7021
GENERAL SPECIFICATIONS
Table 5.
Parameter
TEMPERATURE RANGE (TA)
POWER SUPPLIES
Voltage Supply, VDD
TRANSMIT CURRENT CONSUMPTION 1
868 MHz
0 dBm
5 dBm
10 dBm
450 MHz, Internal Inductor VCO
0 dBm
5 dBm
10 dBm
426 MHz, External Inductor VCO
0 dBm
5 dBm
10 dBm
RECEIVE CURRENT CONSUMPTION
868 MHz
Low Current Mode
High Sensitivity Mode
433MHz, Internal Inductor VCO
Low Current Mode
High Sensitivity Mode
426 MHz, External Inductor VCO
Low Current Mode
High Sensitivity Mode
POWER-DOWN CURRENT CONSUMPTION
Low Power Sleep Mode
1
Min
−40
Typ
2.3
Max
+85
Unit
°C
Test Conditions/Comments
3.6
V
All VDD pins must be tied together
VDD = 3.0 V, PA is matched into 50 Ω
VCO_BIAS = 8
20.2
24.7
32.3
mA
mA
mA
19.9
23.2
29.2
mA
mA
mA
13.5
17
23.3
mA
mA
mA
VCO_BIAS = 8
VCO_BIAS = 2
VDD = 3.0 V
VCO_BIAS = 8
22.7
24.6
mA
mA
24.5
26.4
mA
mA
17.5
19.5
mA
mA
VCO_BIAS = 8
VCO_BIAS = 2
0.1
1
µA
CE low
The transmit current consumption tests used the same combined PA and LNA matching network as that used on the EVAL-ADF7021DBx evaluation boards. Improved
PA efficiency is achieved by using a separate PA matching network.
TIMING CHARACTERISTICS
VDD = 3 V ± 10%, DGND = AGND = 0 V, TA = 25°C, unless otherwise noted. Guaranteed by design but not production tested.
Table 6.
Parameter
t1
t2
t3
t4
t5
t6
t8
t9
t10
t11
t12
t13
t14
t15
Limit at TMIN to TMAX
>10
>10
>25
>25
>10
>20
<25
<25
>10
5 < t11 < (¼ × tBIT)
>5
>5
>¼ × tBIT
>¼ × tBIT
Unit
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
µs
µs
Test Conditions/Comments
SDATA to SCLK setup time
SDATA to SCLK hold time
SCLK high duration
SCLK low duration
SCLK to SLE setup time
SLE pulse width
SCLK to SREAD data valid, readback
SREAD hold time after SCLK, readback
SCLK to SLE disable time, readback
TxRxCLK negative edge to SLE
TxRxDATA to TxRxCLK setup time (Tx mode)
TxRxCLK to TxRxDATA hold time (Tx mode)
TxRxCLK negative edge to SLE
SLE positive edge to positive edge of TxRxCLK
Rev. B | Page 11 of 64
ADF7021
Data Sheet
Timing Diagrams
Serial Interface
t3
t4
SCLK
t1
SDATA
DB31 (MSB)
t2
DB30
DB1
(CONTROL BIT C2)
DB2
DB0 (LSB)
(CONTROL BIT C1)
t6
05876-002
SLE
t5
Figure 2. Serial Interface Timing Diagram
t1
t2
SCLK
SDATA
REG7 DB0
(CONTROL BIT C1)
SLE
t3
t10
RV16
RV2
RV15
RV1
X
05876-003
X
SREAD
t9
t8
Figure 3. Serial Interface Readback Timing Diagram
2FSK/3FSK Timing
±1 × DATA RATE/32
1/DATA RATE
TxRxCLK
TxRxDATA
05876-004
DATA
Figure 4. TxRxDATA/TxRxCLK Timing Diagram in Receive Mode
1/DATA RATE
TxRxCLK
TxRxDATA
FETCH
05876-005
DATA
SAMPLE
Figure 5. TxRxDATA/TxRxCLK Timing Diagram in Transmit Mode
Rev. B | Page 12 of 64
Data Sheet
ADF7021
4FSK Timing
In 4FSK receive mode, MSB/LSB synchronization should be guaranteed by SWD in the receive bit stream.
REGISTER 0 WRITE
SWITCH FROM Rx TO Tx
tSYMBOL
t13
t12
t11
tBIT
SLE
TxRxCLK
Rx SYMBOL
MSB
Rx SYMBOL
LSB
Rx SYMBOL
MSB
Rx SYMBOL
LSB
Tx SYMBOL
MSB
Tx SYMBOL
LSB
Rx MODE
Tx/Rx MODE
Tx SYMBOL
MSB
Tx MODE
05876-074
TxRxDATA
Figure 6. Receive-to-Transmit Timing Diagram in 4FSK Mode
REGISTER 0 WRITE
SWITCH FROM Tx TO Rx
t15
tSYMBOL
t14
tBIT
SLE
TxRxCLK
Tx/Rx MODE
Tx SYMBOL
MSB
Tx SYMBOL
LSB
Tx SYMBOL
MSB
Tx SYMBOL
LSB
Tx MODE
Figure 7. Transmit-to-Receive Timing Diagram in 4FSK Mode
Rev. B | Page 13 of 64
Rx SYMBOL
MSB
Rx SYMBOL
LSB
Rx MODE
05876-075
TxRxDATA
ADF7021
Data Sheet
UART/SPI Mode
UART mode is enabled by setting R0_DB28 to 1. SPI mode is enabled by setting R0_DB28 to 1 and setting R15_DB[17:19] to 0x7.
The transmit/receive data clock is available on the CLKOUT pin.
tBIT
CLKOUT
(TRANSMIT/RECEIVE DATA
CLOCK IN SPI MODE.
NOT USED IN UART MODE.)
Tx BIT
SAMPLE
Tx BIT
TxRxDATA
(RECEIVE DATA OUTPUT
IN UART/SPI MODE.)
Tx BIT
Tx BIT
Tx BIT
HIGH-Z
Tx/Rx MODE
05876-082
TxRxCLK
(TRANSMIT DATA INPUT
IN UART/SPI MODE.)
FETCH
Tx MODE
Figure 8. Transmit Timing Diagram in UART/SPI Mode
tBIT
CLKOUT
(TRANSMIT/RECEIVE DATA
CLOCK IN SPI MODE.
NOT USED IN UART MODE.)
FETCH SAMPLE
TxRxCLK
(TRANSMIT DATA INPUT
IN UART/SPI MODE.)
Tx/Rx MODE
Rx BIT
Rx BIT
Rx BIT
Rx BIT
Rx MODE
Figure 9. Receive Timing Diagram in UART/SPI Mode
Rev. B | Page 14 of 64
Rx BIT
05876-078
TxRxDATA
(RECEIVE DATA OUTPUT
IN UART/SPI MODE.)
HIGH-Z
Data Sheet
ADF7021
ABSOLUTE MAXIMUM RATINGS
TA = 25°C, unless otherwise noted.
Table 7.
Parameter
VDD to GND1
Analog I/O Voltage to GND
Digital I/O Voltage to GND
Operating Temperature Range
Industrial (B Version)
Storage Temperature Range
Maximum Junction Temperature
MLF θJA Thermal Impedance
Reflow Soldering
Peak Temperature
Time at Peak Temperature
1
Rating
−0.3 V to +5 V
−0.3 V to AVDD + 0.3 V
−0.3 V to DVDD + 0.3 V
−40°C to +85°C
−65°C to +125°C
150°C
26°C/W
260°C
40 sec
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
This device is a high performance RF integrated circuit with an
ESD rating of <2 kV and it is ESD sensitive. Proper precautions
should be taken for handling and assembly.
ESD CAUTION
GND = CPGND = RFGND = DGND = AGND = 0.
Rev. B | Page 15 of 64
ADF7021
Data Sheet
48
47
46
45
44
43
42
41
40
39
38
37
CVCO
GND1
L1
GND
L2
VDD
CPOUT
CREG3
VDD3
OSC1
OSC2
MUXOUT
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
ADF7021
TOP VIEW
(Not to Scale)
36
35
34
33
32
31
30
29
28
27
26
25
CLKOUT
TxRxCLK
TxRxDATA
SWD
VDD2
CREG2
ADCIN
GND2
SCLK
SREAD
SDATA
SLE
NOTES
1. THE EXPOSED PAD MUST BE CONNECTED TO GND.
05876-006
MIX_I
MIX_I
MIX_Q
MIX_Q
FILT_I
FILT_I
GND4
FILT_Q
FILT_Q
GND4
TEST_A
CE
13
14
15
16
17
18
19
20
21
22
23
24
VCOIN 1
CREG1 2
VDD1 3
RFOUT 4
RFGND 5
RFIN 6
RFINB 7
RLNA 8
VDD4 9
RSET 10
CREG4 11
GND4 12
Figure 10. Pin Configuration
Table 8. Pin Function Descriptions
Pin No.
1
Mnemonic
VCOIN
2
CREG1
3
VDD1
4
RFOUT
5
6
RFGND
RFIN
7
8
9
10
11
RFINB
RLNA
VDD4
RSET
CREG4
12, 19, 22
13 to 18
24
GND4
MIX_I, MIX_I,
MIX_Q, MIX_Q,
FILT_I, FILT_I
FILT_Q, FILT_Q,
TEST_A
CE
25
SLE
26
SDATA
27
SREAD
28
SCLK
20, 21, 23
Description
The tuning voltage on this pin determines the output frequency of the voltage controlled oscillator (VCO).
The higher the tuning voltage, the higher the output frequency.
Regulator Voltage for PA Block. Place a series 3.9 Ω resistor and a 100 nF capacitor between this pin and
ground for regulator stability and noise rejection.
Voltage Supply for PA Block. Place decoupling capacitors of 0.1 μF and 100 pF as close as possible to this pin.
Tie all VDD pins together.
The modulated signal is available at this pin. Output power levels are from −16 dBm to +13 dBm. The output
should be impedance matched to the desired load using suitable components (see the Transmitter section).
Ground for Output Stage of Transmitter. All GND pins should be tied together.
LNA Input for Receiver Section. Input matching is required between the antenna and the differential LNA
input to ensure maximum power transfer (see the LNA/PA Matching section).
Complementary LNA Input (see the LNA/PA Matching section).
External Bias Resistor for LNA. Optimum resistor is 1.1 kΩ with 5% tolerance.
Voltage Supply for LNA/MIXER Block. This pin should be decoupled to ground with a 10 nF capacitor.
External Resistor. Sets charge pump current and some internal bias currents. Use a 3.6 kΩ resistor with 5% tolerance.
Regulator Voltage for LNA/MIXER Block. Place a 100 nF capacitor between this pin and GND for regulator
stability and noise rejection.
Ground for LNA/MIXER Block.
Signal Chain Test Pins. These pins are high impedance under normal conditions and should be left unconnected.
Signal Chain Test Pins. These pins are high impedance under normal conditions and should be left unconnected.
Chip Enable. Bringing CE low puts the ADF7021 into complete power-down. Register values are lost when CE
is low, and the part must be reprogrammed once CE is brought high.
Load Enable, CMOS Input. When SLE goes high, the data stored in the shift registers is loaded into one of the
four latches. A latch is selected using the control bits.
Serial Data Input. The serial data is loaded MSB first with the 4 LSBs as the control bits. This pin is a high
impedance CMOS input.
Serial Data Output. This pin is used to feed readback data from the ADF7021 to the microcontroller. The SCLK
input is used to clock each readback bit (for example, AFC or ADC) from the SREAD pin.
Serial Clock Input. This serial clock is used to clock in the serial data to the registers. The data is latched into
the 32-bit shift register on the CLK rising edge. This pin is a digital CMOS input.
Rev. B | Page 16 of 64
Data Sheet
Pin No.
29
30
Mnemonic
GND2
ADCIN
31
CREG2
32
33
VDD2
SWD
34
TxRxDATA
35
TxRxCLK
36
CLKOUT
37
MUXOUT
38
OSC2
39
OSC1
40
41
VDD3
CREG3
42
CPOUT
43
44, 46
VDD
L2, L1
45, 47
48
49
GND, GND1
CVCO
EPAD
ADF7021
Description
Ground for Digital Section.
Analog-to-Digital Converter Input. The internal 7-bit ADC can be accessed through this pin. Full scale is 0 V to
1.9 V. Readback is made using the SREAD pin.
Regulator Voltage for Digital Block. Place a 100 nF capacitor between this pin and ground for regulator
stability and noise rejection.
Voltage Supply for Digital Block. Place a decoupling capacitor of 10 nF as close as possible to this pin.
Sync Word Detect. The ADF7021 asserts this pin when it has found a match for the sync word sequence
(see the Register 11—Sync Word Detect Register section). This provides an interrupt for an external
microcontroller indicating valid data is being received.
Transmit Data Input/Received Data Output. This is a digital pin and normal CMOS levels apply. In UART/SPI
mode, this pin provides an output for the received data in receive mode. In transmit UART/SPI mode, this pin
is high impedance (see the Interfacing to Microcontroller/DSP section).
Outputs the data clock in both receive and transmit modes. This is a digital pin and normal CMOS levels
apply. The positive clock edge is matched to the center of the received data. In transmit mode, this pin
outputs an accurate clock to latch the data from the microcontroller into the transmit section at the exact
required data rate. In UART/SPI mode, this pin is used to input the transmit data in transmit mode. In receive
UART/SPI mode, this pin is high impedance (see the Interfacing to Microcontroller/DSP section).
A divided-down version of the crystal reference with output driver. The digital clock output can be used to drive
several other CMOS inputs such as a microcontroller clock. The output has a 50:50 mark-space ratio and is
inverted with respect to the reference. Place a series 1 kΩ resistor as close as possible to the pin in
applications where the CLKOUT feature is being used.
Provides the DIGITAL_LOCK_DETECT Signal. This signal is used to determine if the PLL is locked to the correct
frequency. It also provides other signals such as REGULATOR_READY, which is an indicator of the status of the
serial interface regulator (see the MUXOUT section for more information).
Connect the reference crystal between this pin and OSC1. A TCXO reference can be used by driving this pin
with CMOS levels and disabling the internal crystal oscillator.
Connect the reference crystal between this pin and OSC2. A TCXO reference can be used by driving this pin
with ac-coupled 0.8 V p-p levels and by enabling the internal crystal oscillator.
Voltage Supply for the Charge Pump and PLL Dividers. Decouple this pin to ground with a 10 nF capacitor.
Regulator Voltage for Charge Pump and PLL Dividers. Place a 100 nF capacitor between this pin and ground
for regulator stability and noise rejection.
Charge Pump Output. This output generates current pulses that are integrated in the loop filter. The
integrated current changes the control voltage on the input to the VCO.
Voltage Supply for VCO Tank Circuit. Decouple this pin to ground with a 10 nF capacitor.
External VCO Inductor Pins. If using an external VCO inductor, connect a chip inductor across these pins to set
the VCO operating frequency. If using the internal VCO inductor, these pins can be left floating. See the
Voltage Controlled Oscillator (VCO) section for more information.
Grounds for VCO Block.
Place a 22 nF capacitor between this pin and CREG1 to reduce VCO noise.
Exposed Pad. The exposed pad must be connected to GND.
Rev. B | Page 17 of 64
ADF7021
Data Sheet
TYPICAL PERFORMANCE CHARACTERISTICS
–70
PHASE NOISE (dBc/Hz)
ICP = 0.8mA
–90
DR = 9.6kbps
DATA = PRBS9
RF FREQ = 900MHz
VDD = 2.3V
TEMPERATURE = 25°C
VCO BIAS = 8
VCO ADJUST = 3
–80
fDEV = 2.4kHz
RF FREQ = 869.5MHz
ICP = 1.4mA
FSK
–100
GFSK
–110
ICP = 2.2mA
–120
–130
1
10
100
1000
10000
FREQUENCY OFFSET (MHz)
CENTER 869.5 25MHz
RES BW 300Hz
VBW 300Hz
Figure 14. Output Spectrum in 2FSK and GFSK Modes
Figure 11. Phase Noise Response at 900 MHz, VDD = 2.3 V
16
8
PA BIAS = 9µA
4
0
PA BIAS = 5µA
–4
2FSK
PA BIAS = 7µA
–8
–12
–16
–20
–24
RC2FSK
–28
–32
0
4
8
05876-051
–40
12 16 20 24 28 32 36 40 44 48 52 56 60
PA SETTING
Figure 12. RF Output Power vs. PA Setting
1R
CENTER 869.5 25MHz
RES BW 300Hz
VBW 300Hz
SPAN 50kHz
SWEEP 2.118s (601pts)
05876-048
–36
Figure 15. Output Spectrum in 2FSK and Raised Cosine 2FSK Modes
RF FREQ = 440MHz
OUTPUT POWER = 10dBm
FILTER = T-STAGE LC FILTER
MARKER Δ = 52.2dB
SR = 4.8ksym/s
DATA = PRBS9
fDEV = 2.4kHz
RF FREQ = 869.5MHz
4FSK
1
START 300MHz
RES BW 100Hz
STOP 3.5GHz
VBW 100Hz SWEEP 385.8ms (601pts)
CENTER 869.493 8MHz
RES BW 300Hz
Figure 13. PA Output Harmonic Response with T-Stage LC Filter
VBW 300Hz
SPAN 100kHz
SWEEP 4.237s (601pts)
05876-049
RC4FSK
05876-050
RF OUTPUT POWER (dBm)
DR = 9.6kbps
DATA = PRBS9
fDEV = 2.4kHz
RF FREQ = 869.5MHz
PA BIAS = 11µA
12
SPAN 50kHz
SWEEP 2.118s (601pts)
05876-047
–150
05876-060
–140
Figure 16. Output Spectrum in 4FSK and Raised Cosine 4FSK Modes
Rev. B | Page 18 of 64
Data Sheet
ADF7021
REF 15dBm
SAMP LOG 10dB/
0
ATTEN 25dB
DR = 9.6kbps
DATA = PRS9
fDEV = 2.4kHz
RF FREQ = 869.5MHz
DATA RATE = 1kbps
fDEV = 1kHz
RF FREQ = 135MHz
IF BW = 12.5kHz
–1
–2
VAVG 100
V1 V2
S3 FC
LOG BER
–3
3FSK
3.0V, +25°C
2.3V, +85°C
–4
–5
3.6V, –40°C
RC3FSK
–6
VBW 300Hz
SPAN 50Hz
SWEEP2.226s (401pts)
05876-070
CENTER 869.5MHz
RES BW 300Hz
–8
–130 –128 –126 –124 –122 –120 –118 –116 –114 –112 –110 –108
RF INPUT POWER (dBm)
Figure 20. 2FSK Sensitivity vs. VDD and Temperature, fRF = 135 MHz
Figure 17. Output Spectrum in 3FSK and Raised Cosine 3FSK Modes
10
0
TRACE = MAX HOLD
PA ON/OFF RATE = 3Hz
PA ON/OFF CYCLES = 10000
VDD = 3.0V
3FSK MODULATION
DATA RATE = 9.6kbps
MOD INDEX = 0.5
RF FREQ = 440 MHz
–2
–10
–30
–4
–5
–40
–6
–50
–7
–50
0
50
100
FREQUENCY OFFSET (kHz)
–8
–120
0
–2
3.0V, +25°C
3.6V, –40°C
fDEV (inner) = 2.4kHz
–2
MOD INDEX = 0.5
RF FREQ = 420MHz
IF BW = 12.5kHz
–4
–6
–6
–7
–7
–118
–116
–114
–112
–110
–108
–106
–104
RF INPUT POWER (dBm)
05876-052
–5
–120
–95
DATA RATE = 19.6kbps
SYMBOL RATE = 9.8ksym/s
–1
–5
–8
–122
–100
–8
–120
2.3V +25°C
3.0V +25°C
3.6V +25°C
2.3V –40°C
3.0V –40°C
3.6V –40°C
2.3V +85°C
3.0V +85°C
3.6V +85°C
–115
–110
–105
–100
–95
RF INPUT POWER (dBm)
Figure 22. 4FSK Sensitivity vs. VDD and Temperature, fRF = 420 MHz
Figure 19. 2FSK Sensitivity vs. VDD and Temperature, fRF = 868 MHz
Rev. B | Page 19 of 64
05876-066
2.3V, +85°C
–4
–105
–3
LOG BER
–3
–110
Figure 21. 3FSK Sensitivity vs. VDD and Temperature, fRF = 440 MHz
DATA RATE = 9.6kbps
fDEV = 4kHz
RF FREQ = 868MHz
IF BW = 25kHz
–1
–115
RF INPUT POWER (dBm)
Figure 18. Output Spectrum in Maximum Hold
for Various PA Ramp Rate Options
0
2.3V +25°C
3.0V +25°C
3.6V +25°C
2.3V –40°C
3.0V –40°C
3.6V –40°C
2.3V +85°C
3.0V +85°C
3.6V +85°C
05876-065
LOG BER
–3
–20
–60
–100
LOG BER
fDEV = 2.4kHz
–1
05876-068
OUTPUT POWER (dBm)
0
RAMP RATE:
CW ONLY
256 CODES/BIT
128 CODES/BIT
64 CODES/BIT
32 CODES/BIT
05876-053
–7
ADF7021
Data Sheet
90
–100
80
–102
50
RF FREQ = 868MHz
WANTED SIGNAL
(10dB ABOVE SENSITIVITY
POINT) = 2FSK,
fDEV = 4kHz,
DATA RATE = 9.8kbps
BLOCKER = 2FSK,
fDEV = 4kHz,
DATA RATE = 9.8kbps
VDD = 3.0V
TEMPERATURE = 25°C
40
30
20
10
0
–10
–22
–18
–14
–6
–10
–2 0 2
10
6
14
18
22
FREQUENCY OFFSET (MHz)
–104
–106
–108
DISCRIMINATOR BANDWIDTH =
2× FSK FREQUENCY DEVIATION
–110
–112
–114
–116
DISCRIMINATOR BANDWIDTH =
1× FSK FREQUENCY DEVIATION
–118
05876-059
BLOCKING (dB)
60
0
0.2
0.4
0.6
0.8
1.0
1.2
MODULATION INDEX
Figure 23. Wideband Interference Rejection
05876-058
SENSITIVITY POINT (dBm)
70
RF FREQ = 860MHz
2FSK MODULATION
DATA RATE = 9.6kbps
IF BW = 25kHz
VDD = 3.0V
TEMPERATURE = 25°C
Figure 26. 2FSK Sensitivity vs. Modulation Index vs. Correlator
Discriminator Bandwidth
–20
0
RSSI
READBACK LEVEL
–40
–1
THRESHOLD DETECTION
–60
VITERBI DETECTION
LOG BER
–100
–4
ACTUAL RF INPUT LEVEL
–5
–120
–6
–62.5
–52.5
–42.5
INPUT POWER (dBm)
Figure 27. 3FSK Receiver Sensitivity Using Viterbi Detection and
Threshold Detection
Figure 24. Digital RSSI Readback Linearity
CALIBRATED
60
RF FREQ = 430MHz
EXTERNAL VCO INDUCTOR
DATA RATE = 9.6kbps
TEMPERATURE = 25°C, VDD = 3.0V
+3
RECEIVER SYMBOL LEVEL
70
40
30
UNCALIBRATED
20
10
+1
0
–1
–3
RF I/P LEVEL = –70dBm
DATA RATE = 9.7kbps
0
fDEV (inner) = 1.2kHz
RF FREQUENCY (MHz)
Figure 25. Image Rejection, Uncalibrated vs. Calibrated
22452 ACQS
05876-054
BLOCKING (dB)
50
–10
429.80 429.85 429.90 429.95 430.00 430.05 430.10 430.15 430.20
TYPICALLY
3dB
–7
–120 –118 –116 –114 –112 –110 –108 –106 –104 –102 –100
05876-055
–140
–122.5 –112.5 –102.5 –92.5 –82.5 –72.5
RF INPUT (dBm)
3FSK MODULATION
VDD = 3.0V, TEMP = 25°C
DATA RATE = 9.6kbps
fDEV = 2.4kHz
RF FREQ = 868MHz
IF BW = 18.75kHz
05876-062
–80
–3
IF BW = 25kHz
POST DEMOD BW = 12.4kHz
05876-064
RSSI LEVEL (dBm)
–2
M 50µs
Figure 28. 4FSK Receiver Eye Diagram Measure Using the Test DAC Output
Rev. B | Page 20 of 64
Data Sheet
ADF7021
SENSITIVITY (dBm)
0
–90
20834 ACQS
IF BW = 12.5kHz
POST DEMOD BW = 12.4kHz
M 20µs
C13
MODULATION = 2FSK
DATA RATE = 9.6kbps
fDEV = 4kHz
IF BW = 12.5kHz
DEMOD = CORRELATOR
SENSITIVITY @ 1E-3 BER
IP3 = –9dBm
IP3 = –3dBm
IP3 = –20dBm
–110
–120
DEFAULT
MIXER
LINEARITY
IP3 = –13.5dBm
IP3 = –24dBm
–130
1.7V
3, 72
10, 72
30, 72
(LOW GAIN MODE)
(MEDIUM GAIN MODE)
(HIGH GAIN MODE)
LNA GAIN, FILTER GAIN
Figure 29. 3FSK Receiver Eye Diagram Measured Using the Test DAC Output
05876-069
RF I/P LEVEL = –70dBm
DATA RATE = 10kbps
fDEV = 2.5kHz
IP3= –5dBm
–100
–1
4
HIGH MIXER
LINEARITY
–80
+1
05876-063
RECEIVER SYMBOL LEVEL
–70
Figure 30. Receive Sensitivity vs. LNA/IF Filter Gain and Mixer Linearity Settings
(The Input IP3 at Each Setting is Also Shown)
Rev. B | Page 21 of 64
ADF7021
Data Sheet
FREQUENCY SYNTHESIZER
CLKOUT Divider and Buffer
The on-board crystal oscillator circuitry (see Figure 31) can
use a quartz crystal as the PLL reference. Using a quartz crystal
with a frequency tolerance of ≤10 ppm for narrow-band applications is recommended. It is possible to use a quartz crystal
with >10 ppm tolerance, but to comply with the absolute
frequency error specifications of narrow-band regulations
(for example, ARIB STD-T67 and ETSI EN 300-220), compensation for the frequency error of the crystal is necessary.
The CLKOUT circuit takes the reference clock signal from the
oscillator section, shown in Figure 31, and supplies a divideddown, 50:50 mark-space signal to the CLKOUT pin. The CLKOUT
signal is inverted with respect to the reference clock. An even
divide from 2 to 30 is available. This divide number is set in
R1_DB[7:10]. On power-up, the CLKOUT defaults to divide-by-8.
DVDD
CLKOUT
ENABLE BIT
The oscillator circuit is enabled by setting R1_DB12 high. It is
enabled by default on power-up and is disabled by bringing CE
low. Errors in the crystal can be corrected by using the automatic
frequency control feature or by adjusting the fractional-N value
(see the N Counter section).
05876-083
CP1
Two parallel resonant capacitors are required for oscillation at
the correct frequency. Their values are dependent upon the
crystal specification. They should be chosen to make sure that
the series value of capacitance added to the PCB track capacitance
adds up to the specified load capacitance of the crystal, usually
12 pF to 20 pF. Track capacitance values vary from 2 pF to 5 pF,
depending on board layout. When possible, choose capacitors
that have a very low temperature coefficient to ensure stable
frequency operation over all conditions.
Using a TCXO Reference
A single-ended reference (TCXO, VCXO, or OCXO) can also be
used with the ADF7021. This is recommended for applications
having absolute frequency accuracy requirements of <10 ppm, such
as ARIB STD-T67 or ETSI EN 300-220. There are two options
for interfacing the ADF7021 to an external reference oscillator.
•
CLKOUT
Figure 32. CLKOUT Stage
R Counter
Figure 31. Oscillator Circuit on the ADF7021
•
÷2
To disable CLKOUT, set the divide number to 0. The output
buffer can drive up to a 20 pF load with a 10% rise time at
4.8 MHz. Faster edges can result in some spurious feedthrough
to the output. A series resistor (1 kΩ) can be used to slow the
clock edges to reduce these spurs at the CLKOUT frequency.
OSC2
CP2
DIVIDER
1 TO 15
An oscillator with CMOS output levels can be applied to
OSC2. The internal oscillator circuit should be disabled by
setting R1_DB12 low.
An oscillator with 0.8 V p-p levels can be ac-coupled through
a 22 pF capacitor into OSC1. The internal oscillator circuit
should be enabled by setting R1_DB12 high.
Programmable Crystal Bias Current
Bias current in the oscillator circuit can be configured
between 20 µA and 35 µA by writing to the XTAL_BIAS bits
(R1_DB[13:14]). Increasing the bias current allows the crystal
oscillator to power up faster.
The 3-bit R counter divides the reference input frequency by an
integer of 1 to 7. The divided-down signal is presented as the
reference clock to the phase frequency detector (PFD). The divide
ratio is set in R1_DB[4:6]. Maximizing the PFD frequency reduces
the N value. This reduces the noise multiplied at a rate of 20 log(N)
to the output and reduces occurrences of spurious components.
Register 1 defaults to R = 1 on power-up.
PFD [Hz] = XTAL/R
Loop Filter
The loop filter integrates the current pulses from the charge
pump to form a voltage that tunes the output of the VCO to the
desired frequency. It also attenuates spurious levels generated by
the PLL. A typical loop filter design is shown in Figure 33.
CHARGE
PUMP OUT
VCO
05876-010
OSC1
OSC1
05876-008
REFERENCE INPUT
Figure 33. Typical Loop Filter Configuration
The loop should be designed so that the loop bandwidth (LBW)
is approximately 100 kHz. This provides a good compromise
between in-band phase noise and out-of-band spurious rejection.
Widening the LBW excessively reduces the time spent jumping
between frequencies, but it can cause insufficient spurious attenuation. Narrow-loop bandwidths can result in the loop taking long
periods to attain lock and can also result in a higher level of power
falling into the adjacent channel. The loop filter design on the
EVAL-ADF7021DBX should be used for optimum performance.
Rev. B | Page 22 of 64
Data Sheet
ADF7021
The free design tool ADIsimPLL can also be used to design loop
filters for the ADF7021 (go to www.analog.com/ADIsimPLL for
details).
must be stabilized. Regulator status (CREG4) can be monitored
using the REGULATOR_READY signal from MUXOUT.
N Counter
The MUXOUT pin allows access to various digital points in the
ADF7021. The state of MUXOUT is controlled by R0_DB[29:31].
The feedback divider in the ADF7021 PLL consists of an 8-bit
integer counter (R0_DB[19:26]) and a 15-bit Σ-Δ fractional_N
divider (R0_DB[4:18]). The integer counter is the standard
pulse-swallow type that is common in PLLs. This sets the
minimum integer divide value to 23. The fractional divide value
provides very fine resolution at the output, where the output
frequency of the PLL is calculated as
Fractional _ N 
XTAL 

  Integer _ N 
R
215


When RF_DIVIDE_BY_2 (see the Voltage Controlled
Oscillator (VCO) section) is selected, this formula becomes
f OUT 
REGULATOR_READY
REGULATOR_READY is the default setting on MUXOUT
after the transceiver is powered up. The power-up time of
the regulator is typically 50 μs. Because the serial interface
is powered from the regulator, the regulator must be at its
nominal voltage before the ADF7021 can be programmed.
The status of the regulator can be monitored at MUXOUT.
When the regulator ready signal on MUXOUT is high,
programming of the ADF7021 can begin.
DVDD
Fractional _ N 
XTAL

 0.5   Integer_N 

R
2 15


REGULATOR_READY (DEFAULT)
FILTER_CAL_COMPLETE
DIGITAL_LOCK_DETECT
The combination of the integer_N (maximum = 255) and
the fractional_N (maximum = 32,768/32,768) give a maximum
N divider of 255 + 1. Therefore, the minimum usable PFD is
PFD MIN Hz  
RSSI_READY
Tx_Rx
MUX
LOGIC_ZERO
Maximum Required Output Frequency
TRISTATE
255  1
LOGIC_ONE
For example, when operating in the European 868 MHz to
870 MHz band, PFDMIN equals 3.4 MHz.
DGND
Figure 35. MUXOUT Circuit
REFERENCE IN
4\R
MUXOUT
CONTROL
05876-009
f OUT 
MUXOUT
PFD/
CHARGE
PUMP
FILTER_CAL_COMPLETE
VCO
MUXOUT can be set to FILTER_CAL_COMPLETE. This signal
goes low for the duration of both a coarse IF filter calibration
and a fine IF filter calibration. It can be used as an interrupt to a
microcontroller to signal the end of the IF filter calibration.
4\N
THIRD-ORDER
Σ-∆ MODULATOR
INTEGER-N
05876-011
FRACTIONAL-N
DIGITAL_LOCK_DETECT
Figure 34. Fractional-N PLL
Voltage Regulators
The ADF7021 contains four regulators to supply stable voltages
to the part. The nominal regulator voltage is 2.3 V. Regulator 1
requires a 3.9 Ω resistor and a 100 nF capacitor in series between
CREG1 and GND, whereas the other regulators require a 100 nF
capacitor connected between CREGx and GND. When CE is
high, the regulators and other associated circuitry are powered
on, drawing a total supply current of 2 mA. Bringing the CE pin
low disables the regulators, reduces the supply current to less
than 1 μA, and erases all values held in the registers. The serial
interface operates from a regulator supply. Therefore, to write to
the part, the user must have CE high and the regulator voltage
DIGITAL_LOCK_DETECT indicates when the PLL has locked.
The lock detect circuit is located at the PFD. When the phase
error on five consecutive cycles is less than 15 ns, lock detect is
set high. Lock detect remains high until a 25 ns phase error is
detected at the PFD.
RSSI_READY
MUXOUT can be set to RSSI_READY. This indicates that the
internal analog RSSI has settled and a digital RSSI readback
can be performed.
Tx_Rx
Tx_Rx signifies whether the ADF7021 is in transmit or receive
mode. When in transmit mode, this signal is low. When in
receive mode, this signal is high. It can be used to control an
external Tx/Rx switch.
Rev. B | Page 23 of 64
ADF7021
Data Sheet
To minimize spurious emissions, both VCOs operate at twice
the RF frequency. The VCO signal is then divided by 2 inside
the synthesizer loop, giving the required frequency for the
transmitter and the required local oscillator (LO) frequency for
the receiver. A further divide-by-2 (RF_DIVIDE_BY_2) is
performed outside the synthesizer loop to allow operation in
the 431 MHz to 475 MHz band (internal inductor VCO) and
80 MHz to 325 MHz band (external inductor VCO).
The VCO needs an external 22 nF capacitor between the CVCO
pin and the regulator (CREG1) to reduce internal noise.
VCO BIAS
R1_DB(19:22)
LOOP FILTER
VCO
MUX
÷2
TO PA
÷2
CVCO PIN
TO
N DIVIDER
DIVIDE-BY-2
R1_DB18
05876-012
220µF
Figure 36. Voltage Controlled Oscillator (VCO)
Internal Inductor VCO
To select the internal inductor VCO, set R1_DB25 to Logic 0,
which is the default setting.
VCO bias current can be adjusted using R1_DB[19:22]. To
ensure VCO oscillation, the minimum bias current setting under
all conditions when using the internal inductor VCO is 0x8.
The VCO should be re-centered, depending on the required
frequency of operation, by programming the VCO_ADJUST
bits (R1_DB[23:24]). This is detailed in Table 9.
External Inductor VCO
When using the external inductor VCO, the center frequency of the
VCO is set by the internal varactor capacitance and the combined
inductance of the external chip inductor, bond wire, and PCB track.
The external inductor is connected between the L2 and L1 pins.
750
700
650
fMAX (MHz)
600
550
500
450
400
350
fMIN (MHz)
300
05876-061
The ADF7021 contains two VCO cores. The first VCO, the
internal inductor VCO, uses an internal LC tank and supports
862 MHz to 950 MHz and 431 MHz to 475 MHz operating
bands. The second VCO, the external inductor VCO, uses an
external inductor as part of its LC tank and supports the RF
operating band of 80 MHz to 650 MHz.
A plot of the VCO operating frequency vs. total external
inductance (chip inductor + PCB track) is shown in Figure 37.
FREQUENCY (MHz)
VOLTAGE CONTROLLED OSCILLATOR (VCO)
250
200
0
5
10
15
20
25
30
TOTAL EXTERNAL INDUCTANCE (nH)
Figure 37. Direct RF Output vs. Total External Inductance
The inductance for a PCB track using FR4 material is approximately 0.57 nH/mm. This should be subtracted from the total
value to determine the correct chip inductor value.
Typically, a particular inductor value allows the ADF7021 to
function over a range of ±6% of the RF operating frequency.
When the RF_DIVIDE_BY_2 bit (R1_DB18) is selected, this
range becomes ±3%. At 400 MHz, for example, an operating
range of ±24 MHz (that is, 376 MHz to 424 MHz) with a single
inductor (VCO range centered at 400 MHz) can be expected.
The VCO tuning voltage can be checked for a particular RF
output frequency by measuring the voltage on the VCOIN pin
when the part is fully powered up in transmit or receive mode.
The VCO tuning range is 0.2 V to 2 V. The external inductor
value should be chosen to ensure that the VCO is operating
as close as possible to the center of this tuning range. This is
particularly important for RF frequencies <200 MHz, where
the VCO gain is reduced and a tuning range of <±6 MHz exists.
The VCO operating frequency range can be adjusted by
programming the VCO_ADJUST bits (R1_DB[23:24]). This
typically allows the VCO operating range to be shifted up or
down by a maximum of 1% of the RF frequency.
To select the external inductor VCO, set R1_DB25 to Logic 1.
The VCO_BIAS_CURRENT should be set depending on the
frequency of operation (as indicated in Table 9).
Table 9. RF Output Frequency Ranges for Internal and External Inductor VCOs and Required Register Settings
RF Frequency
Output (MHz)
900 to 950
862 to 900
450 to 470
431 to 450
450 to 650
200 to 450
80 to 200
VCO to
Be Used
Internal L
Internal L
Internal L
Internal L
External L
External L
External L
RF Divide
by 2
No
No
Yes
Yes
No
No
Yes
(VCO_INDUCTOR)
R1_DB25
0
0
0
0
1
1
1
Register Settings
(RF_DIVIDE_BY_2)
(VCO_ADJUST)
R1_DB18
R1_DB[23:24]
0
11
0
00
1
11
1
00
0
XX
0
XX
1
XX
Rev. B | Page 24 of 64
(VCO_BIAS)
R1_DB[19:22]
8
8
8
8
4
3
2
Data Sheet
ADF7021
CHOOSING CHANNELS FOR BEST SYSTEM
PERFORMANCE
An interaction between the RF VCO frequency and the
reference frequency can lead to fractional spur creation. When
the synthesizer is in fractional mode (that is, the RF VCO and
reference frequencies are not integer related), spurs can appear
on the VCO output spectrum at an offset frequency that
corresponds to the difference frequency between an integer
multiple of the reference and the VCO frequency.
These spurs are attenuated by the loop filter. They are more
noticeable on channels close to integer multiples of the reference
where the difference frequency may be inside the loop bandwidth;
thus, the name integer boundary spurs. The occurrence of these
spurs is rare because the integer frequencies are around multiples
of the reference, which is typically >10 MHz. To avoid having
very small or very large values in the fractional register, choose
a suitable reference frequency.
Rev. B | Page 25 of 64
ADF7021
Data Sheet
TRANSMITTER
1
RF OUTPUT STAGE
2
3
4
...
8
...
16
DATA BITS
The power amplifier (PA) of the ADF7021 is based on a singleended, controlled current, open-drain amplifier that has been
designed to deliver up to 13 dBm into a 50 Ω load at a maximum
frequency of 950 MHz.
The PA output current and consequently, the output power, are
programmable over a wide range. The PA configuration is shown
in Figure 38. The output power is set using R2_DB[13:18].
PA RAMP 0
(NO RAMP)
PA RAMP 1
(256 CODES PER BIT)
PA RAMP 2
(128 CODES PER BIT)
PA RAMP 3
(64 CODES PER BIT)
PA RAMP 4
(32 CODES PER BIT)
R2_DB(11:12)
PA RAMP 5
(16 CODES PER BIT)
2
05876-014
PA RAMP 6
(8 CODES PER BIT)
R2_DB(13:18)
PA RAMP 7
(4 CODES PER BIT)
Figure 39. PA Ramping Settings
RFOUT
R2_DB7
+
PA Bias Currents
R0_DB27
05876-013
RFGND
FROM VCO
Figure 38. PA Configuration
The PA is equipped with overvoltage protection, which makes it
robust in severe mismatch conditions. Depending on the application, users can design a matching network for the PA to exhibit
optimum efficiency at the desired radiated output power level
for a wide range of antennas, such as loop or monopole antennas.
See the LNA/PA Matching section for more information.
PA Ramping
The PA_BIAS bits (R2_DB[11:12]) facilitate an adjustment of
the PA bias current to further extend the output power control
range, if necessary. If this feature is not required, the default
value of 9 µA is recommended. If output power of greater than
10 dBm is required, a PA bias setting of 11 µA is recommended.
The output stage is powered down by resetting R2_DB7.
MODULATION SCHEMES
The ADF7021 supports 2FSK, 3FSK, and 4FSK modulation.
The implementation of these modulation schemes is shown in
Figure 40.
When the PA is switched on or off quickly, its changing input
impedance momentarily disturbs the VCO output frequency.
This process is called VCO pulling, and it manifests as spectral
splatter or spurs in the output spectrum around the desired carrier
frequency. Some radio emissions regulations place limits on
these PA transient-induced spurs (for example, ETSI EN 300 220).
By gradually ramping the PA on and off, PA transient spurs are
minimized.
The ADF7021 has built-in PA ramping configurability. As
Figure 39 illustrates, there are eight ramp rate settings, defined
as a certain number of PA setting codes per one data bit period.
The PA steps through each of its 64 code levels but at different
speeds for each setting. The ramp rate is set by configuring
R2_DB[8:10].
PFD/
CHARGE
PUMP
REF
TO
PA STAGE
LOOP FILTER
÷2
VCO
÷N
FRAC_N
THIRD-ORDER
Σ-Δ MODULATOR
F_DEVIATION
INTEGER-N
2FSK
GAUSSIAN
OR
RAISED COSINE
FILTERING
If the PA is enabled/disabled by PA_ENABLE (R2_DB7), it
ramps up at the programmed rate but turns off hard. If the PA is
enabled/disabled by Tx/Rx (R0_DB27), it ramps up and down
at the programmed rate.
Rev. B | Page 26 of 64
TxDATA
MUX
3FSK
4FSK
1 – D2 PR
SHAPING
PRECODER
4FSK
BIT SYMBOL
MAPPER
Figure 40. Transmit Modulation Implementation
05876-015
6
IDAC
Data Sheet
ADF7021
Setting the Transmit Data Rate
3-Level Frequency Shift Keying (3FSK)
In all modulation modes except oversampled 2FSK mode, an
accurate clock is provided on the TxRxCLK pin to latch the data
from the microcontroller into the transmit section at the required
data rate. The exact frequency of this clock is defined by
In 3-level FSK modulation (also known as modified Duobinary
FSK), the binary data (Logic 0 and Logic 1) is mapped onto
three distinct frequencies, the carrier frequency (fC), the carrier
frequency minus a deviation frequency (fC − fDEV), and the
carrier frequency plus the deviation frequency (fC + fDEV).
DATA CLK =
A Logic 0 is mapped to the carrier frequency while a Logic 1 is
either mapped onto frequency fC − fDEV or fC + fDEV.
0
where:
XTAL is the crystal or TCXO frequency.
DEMOD_CLK_DIVIDE is the divider that sets the demodulator
clock rate (R3_DB[6:9]).
CDR_CLK_DIVIDE is the divider that sets the CDR clock rate
(R3_DB[10:17]).
Setting the FSK Transmit Deviation Frequency
In all modulation modes, the deviation from the center
frequency is set using the Tx_FREQUENCY_DEVIATION bits
(R2_DB[19:27]).
The deviation from the center frequency in Hz is as follows:
For direct RF output,
fDEV [Hz] =
PFD × Tx _ FREQUENCY _ DEVIATION
2 16
For RF_DIVIDE_BY_2 enabled,
PFD × Tx _ FREQUENCY _ DEVIATION
fDEV [Hz] = 0.5 ×
216
where Tx_FREQUENCY_DEVIATION is a number from 1 to
511 (R2_DB[19:27]).
In 4FSK modulation, the four symbols (00, 01, 11, 10) are
transmitted as ±3 × fDEV and ±1 × fDEV.
Binary Frequency Shift Keying (2FSK)
Two-level frequency shift keying is implemented by setting the
N value for the center frequency and then toggling it with the
TxDATA line. The deviation from the center frequency is set
using the Tx_FREQUENCY_DEVIATION bits, R2_DB[19:27].
fC – fDEV
fC + fDEV
Figure 41. 3FSK Symbol-to-Frequency Mapping
Compared to 2FSK, this bits-to-frequency mapping results in a
reduced transmission bandwidth because some energy is removed
from the RF sidebands and transferred to the carrier frequency.
At low modulation index, 3FSK improves the transmit spectral
efficiency by up to 25% when compared to 2FSK.
Bit-to-symbol mapping for 3FSK is implemented using a linear
convolutional encoder that also permits Viterbi detection to be
used in the receiver. A block diagram of the transmit hardware
used to realize this system is shown in Figure 42. The convolutional encoder polynomial used to implement the transmit
spectral shaping is
P(D) = 1 − D2
where:
P is the convolutional encoder polynomial.
D is the unit delay operator.
A digital precoder with transfer function 1/P(D) implements an
inverse modulo-2 operation of the 1 − D2 shaping filter in the
transmitter.
Tx DATA
0, 1
PRECODER
1/P(D)
0, 1
CONVOLUTIONAL
ENCODER
P(D)
0, +1, –1
fC
2FSK is selected by setting the MODULATION_SCHEME bits
(R2_DB[4:6]) to 000.
Minimum shift keying (MSK) or Gaussian minimum shift
keying (GMSK) is supported by selecting 2FSK modulation
and using a modulation index of 0.5. A modulation index of
0.5 is set up by configuring R2_DB[19:27] for a FREQDEVIATION =
0.25 × transmit data rate.
fC
RF FREQUENCY
fC + fDEV
FSK MOD
fC – fDEV
CONTROL
AND
DATA FILTERING
TO
N DIVIDER
05876-046
Refer to the Register 3—Transmit/Receive Clock Register
section for more programming information.
+1
–1
05876-057
XTAL
DEMOD _ CLK _ DIVIDE × CDR _ CLK _ DIVIDE × 32
Figure 42. 3FSK Encoding
The signal mapping of the input binary transmit data to the
3-level convolutional output is shown in Table 10. The
convolutional encoder restricts the maximum number of
sequential +1s or −1s to two and delivers an equal number of
+1s and −1s to the FSK modulator, thus ensuring equal spectral
energy in both RF sidebands.
Rev. B | Page 27 of 64
ADF7021
Data Sheet
Table 10. 3-Level Signal Mapping of the Convolutional Encoder
TxDATA
Precoder Output
Encoder Output
1
0
1
0
+1 0
1
0
−1
1
1
+1
0
0
0
0
1
0
1
1
+1
0
1
0
0
1
0
1
0
−1
Another property of this encoding scheme is that the transmitted
symbol sequence is dc-free, which facilitates symbol detection
and frequency measurement in the receiver. In addition, there
is no code rate loss associated with this 3-level convolutional
encoder; that is, the transmitted symbol rate is equal to the data
rate presented at the transmit data input.
3FSK is selected by setting the MODULATION_SCHEME bits
(R2_DB[4:6]) to 010. It can also be used with raised cosine
filtering to further increase the spectral efficiency of the
transmit signal.
By minimizing the separation between symbol frequencies,
4FSK can have high spectral efficiency. The bit-to-symbol
mapping for 4FSK is gray coded and is shown in Figure 43.
0
0
1
1
0
1
This is the only modulation mode that can be used with the UART
mode interface for data transmission (refer to the Interfacing to
Microcontroller/DSPsection for more information).
Gaussian Frequency Shift Keying (GFSK)
1
Gaussian frequency shift keying reduces the bandwidth occupied
by the transmitted spectrum by digitally prefiltering the transmit
data. The BT product of the Gaussian filter used is 0.5.
f
+3fDEV
+fDEV
Gaussian filtering can only be used with 2FSK modulation. This
is selected by setting R2_DB[4:6] to 001.
–fDEV
Raised Cosine Filtering
–3fDEV
t
05876-016
SYMBOL
FREQUENCIES
In oversampled 2FSK, there is no data clock from the TxRxCLK
pin. Instead, the transmit data at the TxRxDATA pin is sampled
at 32 times the programmed rate.
Gaussian or raised cosine filtering can be used to improve
transmit spectral efficiency. The ADF7021 supports Gaussian
filtering (bandwidth time [BT] = 0.5) on 2FSK modulation.
Raised cosine filtering can be used with 2FSK, 3FSK, or 4FSK
modulation. The roll off factor (alpha) of the raised cosine filter
has programmable options of 0.5 and 0.7. Both the Gaussian
and raised cosine filters are implemented using linear phase
digital filter architectures that deliver precise control over the
BT and alpha filter parameters, and guarantee a transmit spectrum
that is very stable over temperature and supply variation.
In 4FSK modulation, two bits per symbol spectral efficiency is
realized by mapping consecutive input bit-pairs in the Tx data
bit stream to one of four possible symbols (−3, −1, +1, +3). Thus,
the transmitted symbol rate is half of the input bit rate.
0
Oversampled 2FSK
SPECTRAL SHAPING
4-Level Frequency Shift Keying (4FSK)
Tx DATA
The transmit clock from Pin TxRxCLK is available after writing
to Register 3 in the power-up sequence for receive mode. The
MSB of the first symbol should be clocked into the ADF7021 on
the first transmit clock pulse from the ADF7021 after writing to
Register 3. Refer to Figure 6 for more timing information.
Figure 43. 4FSK Bit-to-Symbol Mapping
The inner deviation frequencies (+fDEV and − fDEV) are set using
the Tx_FREQUENCY_DEVIATION bits, R2_DB[19:27]. The
outer deviation frequencies are automatically set to three times
the inner deviation frequency.
Raised cosine filtering provides digital prefiltering of the transmit
data by using a raised cosine filter with a roll-off factor (alpha)
of either 0.5 or 0.7. The alpha is set to 0.5 by default, but the
raised cosine filter bandwidth can be increased to provide less
aggressive data filtering by using an alpha of 0.7 (set R2_DB30
to Logic 1). Raised cosine filtering can be used with 2FSK,
3FSK, and 4FSK.
Raised cosine filtering is enabled by setting R2_DB[4:6] as
outlined in Table 11.
Rev. B | Page 28 of 64
Data Sheet
ADF7021
MODULATION AND FILTERING OPTIONS
The various modulation and data filtering options are described
in Table 11.
Table 11. Modulation and Filtering Options
Modulation
BINARY FSK
2FSK
MSK1
OQPSK with half sine
baseband shaping2
GFSK
GMSK3
RC2FSK
Oversampled 2FSK
3-LEVEL FSK
3FSK
RC3FSK
4-LEVEL FSK
4FSK
RC4FSK
Data Filtering
R2_DB[4:6]
None
None
None
000
000
000
Gaussian
Gaussian
Raised cosine
None
001
001
101
100
None
Raised cosine
010
110
None
Raised cosine
011
111
is inverted by setting R2_DB[28:29], an additional 0.5 bit
latency can be added to all values in Table 12.
Table 12. Bit/Symbol Latency in Transmit Mode for Various
Modulation Schemes
Modulation
2FSK
GFSK
RC2FSK, Alpha = 0.5
RC2FSK, Alpha = 0.7
3FSK
RC3FSK, Alpha = 0.5
RC3FSK, Alpha = 0.7
4FSK
RC4FSK, Alpha = 0.5
RC4FSK, Alpha = 0.7
Latency
1 bit
4 bits
5 bits
4 bits
1 bit
5 bits
4 bits
1 symbol
5 symbols
4 symbols
TEST PATTERN GENERATOR
The ADF7021 has a number of built-in test pattern generators
that can be used to facilitate radio link setup or RF measurement.
A full list of the supported patterns is shown in Table 13. The
data rate for these test patterns is the programmed data rate set
in Register 3.
MSK is 2FSK modulation with a modulation index = 0.5.
2
Offset quadrature phase shift keying (OQPSK) with half sine baseband
shaping is spectrally equivalent to MSK.
3
GMSK is GFSK with a modulation index = 0.5.
1
TRANSMIT LATENCY
Transmit latency is the delay time from the sampling of a
bit/symbol by the TxRxCLK signal to when that bit/symbol
appears at the RF output. The latency without any data filtering
is 1 bit. The addition of data filtering adds a further latency as
outlined in Table 12.
It is important that the ADF7021 be left in transmit mode after
the last data bit is sampled by the data clock to account for this
latency. The ADF7021 should stay in transmit mode for a time
equal to the number of latency bit periods for the applied
modulation scheme. This ensures that all of the data sampled by
the TxRxCLK signal appears at RF.
The PN9 sequence is suitable for test modulation when carrying
out adjacent channel power (ACP) or occupied bandwidth
measurements.
Table 13. Transmit Test Pattern Generator Options
Test Pattern
Normal
Transmit Carrier
Transmit + fDEV tone
Transmit − fDEV tone
Transmit 1010 pattern
Transmit PN9 sequence
Transmit SWD pattern repeatedly
The figures for latency in Table 12 assume that the positive
TxRxCLK edge is used to sample data (default). If the TxRxCLK
Rev. B | Page 29 of 64
R15_DB[8:10]
000
001
010
011
100
101
110
ADF7021
Data Sheet
RECEIVER SECTION
RF FRONT END
The ADF7021 is based on a fully integrated, low IF receiver
architecture. The low IF architecture facilitates a very low
external component count and does not suffer from powerlineinduced interference problems.
Figure 44 shows the structure of the receiver front end. The
many programming options allow users to trade off sensitivity,
linearity, and current consumption to best suit their application.
To achieve a high level of resilience against spurious reception,
the low noise amplifier (LNA) features a differential input.
Switch SW2 shorts the LNA input when transmit mode is
selected (R0_DB27 = 0). This feature facilitates the design of a
combined LNA/PA matching network, avoiding the need for an
external Rx/Tx switch. See the LNA/PA Matching section for
details on the design of the matching network.
I (TO FILTER)
RFIN
Tx/Rx SELECT
(R0_DB27)
RFINB
SW2
LNA
LO
Q (TO FILTER)
LNA MODE
(R9_DB25)
MIXER LINEARITY
(R9_DB28)
LNA CURRENT
(R9_DB[26:27])
05876-017
LNA GAIN
(R9_DB[20:21])
LNA/MIXER ENABLE
(R8_DB6)
If the AGC loop is disabled, the gain of the IF filter can be set to one
of three levels by using the FILTER_GAIN bits (R9_DB[22:23]).
The filter gain is adjusted automatically if the AGC loop is
enabled.
IF Filter Bandwidth and Center Frequency Calibration
To compensate for manufacturing tolerances, the IF filter should be
calibrated after power-up to ensure that the bandwidth and
center frequency are correct. Coarse and fine calibration
schemes are provided to offer a choice between fast calibration
(coarse calibration) and high filter centering accuracy (fine
calibration). Coarse calibration is enabled by setting R5_DB4
high. Fine calibration is enabled by setting R6_DB4 high.
For details on when it is necessary to perform a filter
calibration, and in what applications to use either a coarse
calibration or fine calibration, refer to the IF Filter Bandwidth
Calibration section.
It is necessary to do a coarse calibration before doing a fine
calibration. If the IF_FINE_CAL bit (R6_DB4) has already been
configured high, it is possible to do a fine calibration by writing
only to Register 5. Once initiated by writing to the part, the calibration is performed automatically without any user intervention.
Calibration time is 200 μs for coarse calibration and a few
milliseconds for fine calibration, during which time the ADF7021
should not be accessed. The IF filter calibration logic requires
that the IF_FILTER_DIVIDER bits (R5_DB[5:13]) be set such that
Figure 44. RF Front End
XTAL [Hz]
The LNA is followed by a quadrature downconversion mixer,
which converts the RF signal to the IF frequency of 100 kHz.
An important consideration is that the output frequency of the
synthesizer must be programmed to a value 100 kHz below the
center frequency of the received channel. The LNA has two
basic operating modes: high gain/low noise mode and low
gain/low power mode. To switch between these two modes, use
the LNA_MODE bit (R9_DB25). The mixer is also configurable
between a low current and an enhanced linearity mode using
the MIXER_LINEARITY bit (R9_DB28).
IF _ FILTER _ DIVIDER
Based on the specific sensitivity and linearity requirements of
the application, it is recommended to adjust the LNA_MODE
bit and MIXER_LINEARITY bit as outlined in Table 14.
The gain of the LNA is configured by the LNA_GAIN bits
(R9_DB[20:21]) and can be set by either the user or the
automatic gain control (AGC) logic.
 50 kHz
IF Filter Fine Calibration Overview
The fine calibration uses two internally generated tones at
certain offsets around the IF filter. The two tones are attenuated
by the IF filter, and the level of this attenuation is measured
using the RSSI. The filter center frequency is adjusted to allow
equal attenuation of both tones. The attenuation of the two test
tones is then remeasured. This continues for a maximum of
10 RSSI measurements, at which stage the calibration algorithm
sets the IF filter center frequency to within 0.5 kHz of 100 kHz.
The frequency of these tones is set by the following bits:

IF_CAL_LOWER_TONE_DIVIDER (R6_DB[5:12])

IF_CAL_UPPER_TONE_DIVIDER (R6_DB[13:20])
It is recommended to place the lower and upper tones as close
as possible to 65.8 kHz and 131.5 kHz, respectively, as outlined
in the following equations:
IF FILTER
IF Filter Settings
Out-of-band interference is rejected by means of a fifth-order
Butterworth polyphase IF filter centered on a frequency of
100 kHz. The bandwidth of the IF filter can be programmed to
12.5 kHz, 18.75 kHz, or 25 kHz by R4_DB[30:31] and should be
chosen as a compromise between interference rejection and
attenuation of the desired signal.
Rev. B | Page 30 of 64
XTAL
 65.8 kHz
IF _ CAL _ LOWER _ TONE _ DIVIDE  2
XTAL
 131.5 kHz
IF _ CAL _ UPPER _ TONE _ DIVIDE  2
Data Sheet
ADF7021
The user has the option of changing the two threshold values
from the defaults of 30 and 70 (Register 9). The default AGC
setup values should be adequate for most applications. The
threshold values must be chosen to be more than 30 apart for
the AGC to operate correctly.
The calibration algorithm adjusts the filter center frequency
and measures the RSSI 10 times during the calibration. The
time for an adjustment plus RSSI measurement is given by
IF Tone Calibration Time =
IF_CAL_DWELL_TIME
SEQ CLK
It is recommended that the IF tone calibration time be at least
500 µs. The total time for the IF filter fine calibration is given by
IF Filter Fine Calibration Time = IF Tone Calibration Time × 10
RSSI/AGC
The RSSI is implemented as a successive compression log amp
following the baseband (BB) channel filtering. The log amp
achieves ±3 dB log linearity. It also doubles as a limiter to
convert the signal-to-digital levels for the FSK demodulator.
The offset correction circuit uses the BBOS_CLK_DIVIDE bits
(R3_DB[4:5]), which should be set between 1 MHz and 2 MHz.
The RSSI level is converted for user readback and for digitally
controlled AGC by an 80-level (7-bit) flash ADC. This level can
be converted to input power in dBm. By default, the AGC is on
when powered up in receive mode.
OFFSET
CORRECTION
A
1
IFWR
A
A
IFWR
IFWR
LATCH
In Register 3, the user should set the BBOS_CLK_DIVIDE bits
(R3_DB[4:5]) to give a baseband offset clock (BBOS CLK)
frequency between 1 MHz and 2 MHz.
BBOS CLK [Hz] = XTAL/(BBOS_CLK_DIVIDE)
where BBOS_CLK_DIVIDE can be set to 4, 8, 16, or 32.
AGC Information and Timing
AGC is selected by default and operates by setting the appropriate
LNA and filter gain settings for the measured RSSI level. It is
possible to disable AGC by writing to Register 9 if the user wants to
enter one of the modes listed in Table 14. The time for the AGC
circuit to settle and, therefore, the time it takes to measure the RSSI
accurately, is typically 300 µs. However, this depends on how many
gain settings the AGC circuit has to cycle through. After each gain
change, the AGC loop waits for a programmed time to allow
transients to settle. This AGC update rate is set according to
FSK
DEMOD
SEQ _ CLK _ DIVIDE [Hz]
AGC Update Rate [Hz] =
CLK
IFWR
Offset Correction Clock
RSSI
05876-018
ADC
R
Figure 45. RSSI Block Diagram
AGC _ CLK _ DIVIDE
where:
AGC_CLK_DIVIDE is set by R3_DB[26:31]. A value of 10 is
recommended.
SEQ_CLK_DIVIDE = 100 kHz (R3_DB[18:25]).
By using the recommended setting for AGC_CLK_DIVIDE, the
total AGC settling time is
RSSI Thresholds
When the RSSI is above AGC_HIGH_THRESHOLD
(R9_DB[11:17]), the gain is reduced. When the RSSI is
below AGC_LOW_THRESHOLD (R9_DB[4:10]), the gain
is increased. The thresholds default to 30 and 70 on power-up
in receive mode. A delay (set by AGC_CLOCK_DIVIDE,
R3_DB[26:31]) is programmed to allow for settling of the loop.
A value of 10 is recommended.
AGC Settling Time [sec] =
Number of AGC Gain Changes
AGC Update Rate [Hz]
The worst case for AGC settling is when the AGC control loop
has to cycle through all five gain settings, which gives a maximum
AGC settling time of 500 µs.
Table 14. LNA/Mixer Modes
Receiver Mode
High Sensitivity
Mode (Default)
Enhanced Linearity
High Gain
Medium Gain
Enhanced Linearity
Medium Gain
Low Gain
Enhanced Linearity
Low Gain
LNA_MODE
(R9_DB25)
0
LNA_GAIN
(R9_DB[20:21])
30
MIXER_LINEARITY
(R9_DB28)
0
Sensitivity
(2FSK, DR = 4.8 kbps,
fDEV = 4 kHz)
−118
Rx Current
Consumption (mA)
24.6
Input IP3
(dBm)
−24
0
30
1
−114.5
24.6
−20
1
1
10
10
0
1
−112
−105.5
22.1
22.1
−13.5
−9
1
1
3
3
0
1
−100
−92.3
22.1
22.1
−5
−3
Rev. B | Page 31 of 64
ADF7021
Data Sheet
FREQUENCY
CORRELATOR
Q
Input Power [dBm] =
−130 dBm + (Readback Code + Gain Mode Correction) × 0.5
MUX
LINEAR
DEMODULATOR
where:
Readback Code is given by Bit RV7 to Bit RV1 in the readback
register (see the Readback Format section).
Gain Mode Correction is given by the values in Table 15.
The LNA gain (LG2, LG1) and filter gain (FG2, FG1) values
are also obtained from the readback register, as part of an RSSI
readback.
THRESHOLD
DETECTION
2/3/4FSK
TxRxDATA
TxRx CLK
CLOCK
AND
DATA
RECOVERY
MUX
VITERBI
DETECTION
3FSK
Table 15. Gain Mode Correction
Filter Gain
(FG2, FG1)
H (1, 0)
H (1, 0)
M (0, 1)
L (0, 0)
L (0, 0)
Gain Mode
Correction
0
24
38
58
86
Figure 46. Overview of Demodulation, Detection, and CDR Process
Correlator Demodulator
The correlator demodulator can be used for 2FSK, 3FSK, and
4FSK demodulation. Figure 47 shows the operation of the
correlator demodulator for 2FSK.
FREQUENCY CORRELATOR
DISCRIM BW
An additional factor should be introduced to account for losses
in the front-end-matching network/antenna.
I
OUTPUT LEVELS:
2FSK = +1, –1
3FSK = +1, 0, –1
4FSK = +3, +1, –1, –3
LIMITERS
DEMODULATION, DETECTION, AND CDR
Q
System Overview
IF – fDEV
An overview of the demodulation, detection, and clock and
data recovery (CDR) of the received signal on the ADF7021 is
shown in Figure 46.
The quadrature outputs of the IF filter are first limited and
then fed to either the correlator FSK demodulator or the
linear FSK demodulator. The correlator demodulator is used
to demodulate 2FSK, 3FSK, and 4FSK. The linear demodulator
is used for frequency measurement and is enabled when the
AFC loop is active. The linear demodulator can also be used
to demodulate 2FSK.
Following the demodulator, a digital post demodulator filter
removes excess noise from the demodulator signal output.
Threshold/slicer detection is used for data recovery of 2FSK and
4FSK. Data recovery of 3FSK can be implemented using either
threshold detection or Viterbi detection.
R4_DB(10:19)
DISCRIMINATOR BW
IF
IF + fDEV
R4_DB9
Rx DATA INVERT
R4_DB7
DOT/CROSS PRODUCT
05876-079
LNA Gain
(LG2, LG1)
H (1, 0)
M (0, 1)
M (0, 1)
M (0, 1)
L (0, 0)
05876-080
The RSSI formula is
POST
DEMOD FILTER
LIMITERS
I
RSSI Formula (Converting to dBm)
Figure 47. 2FSK Correlator Demodulator Operation
The quadrature outputs of the IF filter are first limited and then
fed to a digital frequency correlator that performs filtering and
frequency discrimination of the 2FSK/3FSK/4FSK spectrum.
For 2FSK modulation, data is recovered by comparing the
output levels from two correlators. The performance of this
frequency discriminator approximates that of a matched filter
detector, which is known to provide optimum detection in the
presence of additive white Gaussian noise (AWGN). This
method of FSK demodulation provides approximately 3 dB to
4 dB better sensitivity than a linear demodulator.
An on-chip CDR PLL is used to resynchronize the received bit
stream to a local clock. It outputs the retimed data and clock on
the TxRxDATA and TxRxCLK pins, respectively.
Rev. B | Page 32 of 64
Data Sheet
ADF7021
Figure 48 shows a block diagram of the linear demodulator.
4FSK demodulation is implemented using the correlator
demodulator followed by the post demodulator filter and
threshold detection. The output of the post demodulation
filter is a 4-level signal that represents the transmitted symbols
(−3, −1, +1, +3). Threshold detection of 4FSK requires three
threshold settings, one that is always fixed at 0 and two that
are programmable and are symmetrically placed above and
below 0 using the 3FSK/4FSK_SLICER_THRESHOLD bits
(R13_DB[4:10]).
I
LEVEL
IF
LIMITER
Q
FREQUENCY
LINEAR
DISCRIMINATOR
ENVELOPE
DETECTOR
3FSK and 4FSK Threshold Detection
POST DEMOD
FILTER
Linear Demodulator
+
SLICER
2FSK
2FSK Rx DATA
RxCLK
R4_DB(20:29)
05876-073
FREQUENCY
READBACK
AND AFC LOOP
Figure 48. Block Diagram of Linear FSK Demodulator
A digital frequency discriminator provides an output signal that
is linearly proportional to the frequency of the limiter outputs.
The discriminator output is filtered and averaged using a combined
averaging filter and envelope detector. The demodulated 2FSK
data from the post demodulator filter is recovered by threshold
detecting the envelope detector output, as shown in Figure 48.
This method of demodulation corrects for frequency errors
between transmitter and receiver when the received spectrum is
close to or within the IF bandwidth. This envelope detector
output is also used for AFC readback and provides the
frequency estimate for the AFC control loop.
Post Demodulator Filter
A second-order, digital low-pass filter removes excess noise from
the demodulated bit stream at the output of the discriminator.
The bandwidth of this post demodulator filter is programmable
and must be optimized for the user’s data rate and received
modulation type. If the bandwidth is set too narrow, performance
degrades due to intersymbol interference (ISI). If the bandwidth
is set too wide, excess noise degrades the performance of the
receiver. The POST_DEMODULATOR_BW bits (R4_DB[20:29])
set the bandwidth of this filter.
2FSK Bit Slicer/Threshold Detection
2FSK demodulation can be implemented using the correlator
FSK demodulator or the linear FSK demodulator. In both cases,
threshold detection is used for data recovery at the output of the
post demodulation filter.
The output signal levels of the correlator demodulator are
always centered about zero. Therefore, the slicer threshold level
can be fixed at zero, and the demodulator performance is
independent of the run-length constraints of the transmit data
bit stream. This results in robust data recovery that does not
suffer from the classic baseline wander problems that exist in
the more traditional FSK demodulators.
When the linear demodulator is used for 2FSK demodulation,
the output of the envelope detector is used as the slicer threshold,
and this output tracks frequency errors that are within the IF
filter bandwidth.
3FSK demodulation is implemented using the correlator demodulator, followed by a post demodulator filter. The output of the
post demodulator filter is a 3-level signal that represents the
transmitted symbols (−1, 0, +1). Data recovery of 3FSK can be
implemented using threshold detection or Viterbi detection.
Threshold detection is implemented using two thresholds that
are programmable and are symmetrically placed above and
below zero using the 3FSK/4FSK_SLICER_THRESHOLD bits
(R13_DB[4:10]).
3FSK Viterbi Detection
Viterbi detection of 3FSK operates on a four-state trellis and is
implemented using two interleaved Viterbi detectors operating
at half the symbol rate. The Viterbi detector is enabled by
R13_DB11.
To facilitate different run length constraints in the transmitted
bit stream, the Viterbi path memory length is programmable
in steps of 4 bits, 6 bits, 8 bits, or 32 bits by setting the
VITERBI_PATH_MEMORY bits (R13_DB[13:14]). This
should be set equal to or longer than the maximum number
of consecutive 0s in the interleaved transmit bit stream.
When used with Viterbi detection, the receiver sensitivity
for 3FSK is typically +3 dB better than that obtained using
threshold detection. When the Viterbi detector is enabled,
however, the receiver bit latency is increased by twice the
Viterbi path memory length.
Clock Recovery
An oversampled digital clock and data recovery (CDR) PLL is
used to resynchronize the received bit stream to a local clock
in all modulation modes. The oversampled clock rate of the PLL
(CDR CLK) must be set at 32 times the symbol rate (see the
Register 3—Transmit/Receive Clock Register section). The
maximum data/symbol rate tolerance of the CDR PLL is
determined by the number of zero-crossing symbol transitions
in the transmitted packet. For example, if using 2FSK with a
101010 preamble, a maximum tolerance of ±3.0% of the data
rate is achieved. However, this tolerance is reduced during
recovery of the remainder of the packet where symbol transitions may not be guaranteed to occur at regular intervals.
To maximize the data rate tolerance of the CDR, some form
of encoding and/or data scrambling is recommended that
guarantees a number of transitions at regular intervals. For
Rev. B | Page 33 of 64
ADF7021
Data Sheet
example, using 2FSK with Manchester-encoded data achieves a
data rate tolerance of ±2.0%.
The CDR PLL is designed for fast acquisition of the recovered
symbols during preamble and typically achieves bit synchronization within 5-symbol transitions of preamble.
In 4FSK modulation, the tolerance using the +3, −3, +3, −3
preamble is ±3% of the symbol rate (or ±1.5% of the data rate).
However, this tolerance is reduced during recovery of the
remainder of the packet where symbol transitions may not be
guaranteed to occur at regular intervals. To maximize the
symbol/data rate tolerance, the remainder of the 4FSK packet
should be constructed so that the transmitted symbols retain close
to dc-free properties by using data scrambling and/or by inserting
specific dc balancing symbols that are inserted in the transmitted
bit stream at regular intervals such as after every 8 or 16 symbols.
In 3FSK modulation, the linear convolutional encoder scheme
guarantees that the transmitted symbol sequence is dc-free,
facilitating symbol detection. However, Tx data scrambling is
recommended to limit the run length of zero symbols in the
transmit bit stream. Using 3FSK, the CDR data rate tolerance is
typically ±0.5%.
RECEIVER SETUP
Correlator Demodulator Setup
To enable the correlator for various modulation modes, refer to
Table 16.
(DEMOD CLK × K )
400 × 10 3
where:
DEMOD CLK is as defined in the Register 3—Transmit/Receive
Clock Register section.
K is set for each modulation mode according to the following:
 100 × 10 3
K = Round 
 f DEV




 100 × 10 3 

K = Round4 FSK 

 4 × f DEV 
where:
Round is rounded to the nearest integer.
Round4FSK is rounded to the nearest of the following integers: 32,
31, 28, 27, 24, 23, 20, 19, 16, 15, 12, 11, 8, 7, 4, 3.
fDEV is the transmit frequency deviation in Hz. For 4FSK, fDEV is
the frequency deviation used for the ±1 symbols (that is, the
inner frequency deviations).
To optimize the coefficients of the correlator, R4_DB7 and
R4_DB[8:9] must also be assigned. The value of these bits
depends on whether K is odd or even. These bits are assigned
according to Table 17 and Table 18.
Table 17. Assignment of Correlator K Value for 2FSK and 3FSK
K
Even
Even
Odd
Odd
K/2
Even
Odd
N/A
N/A
(K + 1)/2
N/A
N/A
Even
Odd
R4_DB7
0
0
1
1
R4_DB[8:9]
00
10
00
10
R4_DB7
0
1
R4_DB[8:9]
00
00
Linear Demodulator Setup
To optimize receiver sensitivity, the correlator bandwidth must be
optimized for the specific deviation frequency and modulation
used by the transmitter. The discriminator bandwidth is
controlled by R4_DB[10:19] and is defined as
For 2FSK,




For 4FSK,
K
Even
Odd
DEMOD_SCHEME (R4_DB[4:6])
001
010
011
DISCRIMINATOR _ BW =
 100 × 10 3
K = Round 
 2 × f DEV
Table 18. Assignment of Correlator K Value for 4FSK
Table 16. Enabling the Correlator Demodulator
Received Modulation
2FSK
3FSK
4FSK
For 3FSK,
The linear demodulator can be used for 2FSK demodulation. To
enable the linear demodulator, set the DEMOD_SCHEME bits
(R4_DB[4:6]) to 000.
Post Demodulator Filter Setup
The 3 dB bandwidth of the post demodulator filter should be
set according to the received modulation type and data rate.
The bandwidth is controlled by R4_DB[20:29] and is given by
POST _ DEMOD _ BW =
2 11 × π × f CUTOFF
DEMOD CLK
where fCUTOFF is the target 3 dB bandwidth in Hz of the post
demodulator filter.
Table 19. Post Demodulator Filter Bandwidth Settings for
2FSK/3FSK/4FSK Modulation Schemes
Received
Modulation
2FSK
3FSK
4FSK
Rev. B | Page 34 of 64
Post Demodulator Filter Bandwidth,
fCUTOFF (Hz)
0.75 × data rate
1 × data rate
1.6 × symbol rate (= 0.8 × data rate)
Data Sheet
ADF7021
3FSK Viterbi Detector Setup
3FSK Threshold Detector Setup
The Viterbi detector can be used for 3FSK data detection. This
is activated by setting R13_DB11 to Logic 1.
To activate threshold detection of 3FSK, R13_DB11 should be
set to Logic 0. The 3FSK/4FSK_SLICER_THRESHOLD bits
(R13_DB[4:10]) should be set as outlined in the 3FSK Viterbi
Detector Setup section.
The Viterbi path memory length is programmable in steps of 4,
6, 8, or 32 bits (VITERBI_PATH_MEMORY, R13_DB[13:14]).
The path memory length should be set equal to or greater than
the maximum number of consecutive 0s in the interleaved
transmit bit stream.
The Viterbi detector also uses threshold levels to implement the
maximum likelihood detection algorithm. These thresholds are
programmable via the 3FSK/4FSK_SLICER_THRESHOLD bits
(R13_DB[4:10]).
These bits are assigned as follows:
3FSK CDR Setup
In 3FSK, a transmit preamble of at least 40 bits of continuous 1s
is recommended to ensure a maximum number of symbol
transitions for the CDR to acquire lock.
The clock and data recovery for 3FSK requires a number of
parameters in Register 13 to be set (see Table 20).
4FSK Threshold Detector Setup
The threshold for the 4FSK detector is set using the
3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]).
The threshold should be set according to
3FSK/4FSK_SLICER_THRESHOLD =
 Transmit Frequency Deviation × K 

57 × 
100 × 10 3


3FSK/4FSK_SLICER_THRESHOLD =
 4FSK Outer Tx Deviation × K 

78 × 
100 × 10 3


where K is the value calculated for correlator discriminator
bandwidth.
where K is the value calculated for correlator discriminator
bandwidth.
Table 20. 3FSK CDR Settings
Parameter
PHASE_CORRECTION (R13_DB12)
3FSK_CDR_THRESHOLD (R13_DB[15:21])
3FSK_PREAMBLE_TIME_VALIDATE (R13_DB [22:25])
Recommended Setting
1
 Transmit Frequency Deviation × K 

62 × 
100 × 10 3


where K is the value calculated for
correlator discriminator bandwidth.
1111
Rev. B | Page 35 of 64
Purpose
Phase correction on
Sets CDR decision threshold levels
Preamble detector time qualifier
ADF7021
Data Sheet
DEMODULATOR CONSIDERATIONS
2FSK Preamble
The recommended preamble bit pattern for 2FSK is a dc-free
pattern (such as a 10101010… pattern). Preamble patterns with
longer run-length constraints (such as 11001100…) can also be
used but result in a longer synchronization time of the received
bit stream in the receiver. The preamble needs to allow enough
bits for AGC settling of the receiver and CDR acquisition. A
minimum of 16 preamble bits is recommended. When the receiver
is using the internal AFC, the minimum recommended number
of preamble bits is 48.
The remaining fields that follow the preamble header do not
have to use dc-free coding. For these fields, the ADF7021 can
accommodate coding schemes with a run length of up to
eight bits without any performance degradation. If longer run
lengths are required, an encoding scheme such as 8B/10B or
Manchester encoding is recommended.
4FSK Preamble and Data Coding
The recommended preamble bit pattern for 4FSK is a repeating
00100010… bit sequence. This 2-level sequence of repeating
−3, +3, −3, +3 symbols is dc-free and maximizes the symbol
timing performance and data recovery of the 4FSK preamble in
the receiver. The minimum recommended length of the
preamble is 32 bits (16 symbols).
The remainder of the 4FSK packet should be constructed so
that the transmitted symbols retain close to dc-free property by
using data scrambling and/or by inserting specific dc balancing
symbols in the transmitted bit stream at regular intervals, such
as after every 8 or 16 symbols.
2FSK Correlator Demodulator and Frequency Errors
The ADF7021 has a number of options to combat frequency
errors that exist due to mismatches between the transmit and
receive crystals/TCXOs.
With AFC disabled, the correlator demodulator is tolerant to
frequency errors over the ±0.4 × fDEV range, where fDEV is the
FSK frequency deviation. For larger frequency errors, the
frequency tolerance can be widened to ±0.8 × fDEV by adjusting
the value of K and thus doubling the correlator bandwidth.
K should then be calculated as
 100 × 10
K = Round 
 2 × f DEV
3




The DISCRIMINATOR_BW setting should also be recalculated
using the new K value. Doubling the correlator bandwidth to
improve frequency error tolerance in this manner typically
results in a 1 dB to 2 dB loss in receiver sensitivity.
Correlator Demodulator and Low Modulation Indices
The modulation index in 2FSK is defined as
Modulation Index =
The receiver sensitivity performance can be maximized at low
modulation index by increasing the discriminator bandwidth of
the correlator demodulator. For modulation indices of less than
0.4, it is recommended to double the correlator bandwidth by
calculating K as follows:
 100 e 3
K = Round 
 2 × f DEV




The DISCRIMINATOR_BW should be recalculated using the
new K value. Figure 26 highlights the improved sensitivity that
can be achieved for 2FSK modulation, at low modulation
indices, by doubling the correlator bandwidth.
AFC OPERATION
The ADF7021 also supports a real-time AFC loop that is used
to remove frequency errors due to mismatches between the
transmit and receive crystals/TCXOs. The AFC loop uses the
linear frequency discriminator block to estimate frequency
errors. The linear FSK discriminator output is filtered and
averaged to remove the FSK frequency modulation using a
combined averaging filter and envelope detector. In receive
mode, the output of the envelope detector provides an estimate
of the average IF frequency.
Two methods of AFC supported on the ADF7021 are external
and internal.
External AFC
Here, the user reads back the frequency information through
the ADF7021 serial port and applies a frequency correction
value to the fractional-N synthesizer-N divider.
The frequency information is obtained by reading the 16-bit
signed AFC readback, as described in the Readback Format
section, and by applying the following formula:
Frequency Readback [Hz] =
(AFC_READBACK × DEMOD CLK)/218
Although the AFC_READBACK value is a signed number, under
normal operating conditions, it is positive. In the absence of
frequency errors, the frequency readback value is equal to the
IF frequency of 100 kHz.
Internal AFC
The ADF7021 supports a real-time, internal, automatic
frequency control loop. In this mode, an internal control loop
automatically monitors the frequency error and adjusts the
synthesizer-N divider using an internal proportional integral
(PI) control loop.
The internal AFC control loop parameters are controlled in
Register 10. The internal AFC loop is activated by setting
R10_DB4 to 1. A scaling coefficient must also be entered, based
on the crystal frequency in use. This is set up in R10_DB[5:16]
and should be calculated using
2 × f DEV
Data Rate
 2 24 × 500 

AFC _ SCALING _ FACTOR = Round 
 XTAL 


Rev. B | Page 36 of 64
Data Sheet
ADF7021
Maximum AFC Range
AUTOMATIC SYNC WORD DETECTION (SWD)
The maximum frequency correction range of the AFC loop is
programmable on the ADF7021. This is set by R10_DB[24:31].
The maximum AFC correction range is the difference in
frequency between the upper and lower limits of the AFC
tuning range. For example, if the maximum AFC correction
range is set to 10 kHz, the AFC can adjust the receiver LO
within the
fLO ± 5 kHz range.
The ADF7021 also supports automatic detection of the sync or
ID fields. To activate this mode, the sync (or ID) word must be
preprogrammed into the ADF7021. In receive mode, this
preprogrammed word is compared to the received bit stream.
When a valid match is identified, the external SWD pin is
asserted by the ADF7021 on the next Rx clock pulse.
However, when RF_DIVIDE_BY_2 (R1_DB18) is enabled, the
programmed range is halved. The user should account for this
halving by doubling the programmed maximum AFC range.
The recommended maximum AFC correction range should be
≤ 1.5 × IF filter bandwidth. If the maximum frequency correction
range is set to be > 1.5 × IF bandwidth, the attenuation of the IF
filter can degrade the AFC loop sensitivity.
The adjacent channel rejection (ACR) performance of the
receivers can be degraded when AFC is enabled and the AFC
correction range is close to the IF filter bandwidth. However,
because the AFC correction range is programmable, the user
can trade off correction range and ACR performance.
When AFC errors are removed using either the internal or
external AFC, further improvement in receiver sensitivity can
be obtained by reducing the IF filter bandwidth using the
IF_BW bits (R4_DB[30:31]).
This feature can be used to alert the microprocessor that a
valid channel has been detected. It relaxes the computational
requirements of the microprocessor and reduces the overall
power consumption.
The SWD signal can also be used to frame the received packet
by staying high for a preprogrammed number of bytes. The data
packet length can be set in R12_DB[8:15].
The SWD pin status can be configured by setting R12_DB[6:7].
R11_DB[4:5] are used to set the length of the sync/ID word, which
can be 12, 16, 20, or 24 bits long. A value of 24 bits is recommended
to minimize false sync word detection in the receiver that can
occur during recovery of the remainder of the packet or when
noise/no signal is present at the receiver input. The transmitter
must transmit the sync byte MSB first and the LSB last to ensure
proper alignment in the receiver sync-byte-detection hardware.
An error tolerance parameter can also be programmed that
accepts a valid match when up to 3 bits of the word are
incorrect. The error tolerance value is assigned in R11_DB[6:7].
Rev. B | Page 37 of 64
ADF7021
Data Sheet
APPLICATIONS INFORMATION
IF FILTER BANDWIDTH CALIBRATION
The IF filter should be calibrated on every power-up in receive
mode to correct for errors in the bandwidth and filter center
frequency due to process variations. The automatic calibration
requires no external intervention once it is initiated by a write
to Register 5. Depending on numerous factors, such as IF filter
bandwidth, received signal bandwidth, and temperature
variation, the user must determine whether to carry out a
coarse calibration or a fine calibration. For information on
calibration setup, refer to the IF Filter section.
This method should only be used if the successive power-ups in
receive mode are over a short duration, during which time there
is little variation in temperature (>15°C).
IF Filter Variation with Temperature
The performance of both calibration methods is outlined in
Table 21.
Table 21. IF Filter Calibration Specifications
Filter Calibration
Method
Coarse Cal
Center Frequency
Accuracy1
100 kHz ± 2.5 kHz
Calibration
Time (Typ)
200 µs
Fine Cal
100 kHz ± 0.5 kHz
5.2 ms
1
After the initial coarse calibration and fine calibration, the result of
the fine calibration can be read back through the serial interface
using the FILTER_CAL_READBACK result (refer to the Filter
Bandwidth Calibration Readback section). On subsequent
power-ups in receive mode, the filter is manually adjusted using
the previous fine filter calibration result. This manual adjust is
performed using the IF_FILTER_ADJUST bits (R5_DB[14:19]).
After calibration.
When to Use Coarse Calibration
It is recommended to perform a coarse calibration on every
receive mode power-up. This calibration typically takes 200 µs.
The FILTER_CAL_COMPLETE signal from MUXOUT can be
used to monitor the filter calibration duration or to signal the
end of calibration. The ADF7021 should not be accessed during
calibration.
When to Use a Fine Calibration
In cases where the receive signal bandwidth is very close to the
bandwidth of the IF filter, it is recommended to perform a fine
filter calibration every time the unit powers up in receive mode.
A fine calibration should be performed if
OBW + Coarse Calibration Variation > IF_FILTER_BW
where:
OBW is the 99% occupied bandwidth of the transmit signal.
Coarse Calibration Variation is 2.5 kHz.
IF_FILTER_BW is set by R4_DB[30:31].
The FILTER_CAL_COMPLETE signal from MUXOUT (set by
R0_DB[29:31]) can be used to monitor the filter calibration
duration or to signal the end of calibration. A coarse filter
calibration is automatically performed prior to a fine filter
calibration.
When to Use Single Fine Calibration
In applications where the receiver powers up numerous times in
a short period, it is only necessary to perform a one-time fine
calibration on the initial receiver power-up.
When calibrated, the filter center frequency can vary with
changes in temperature. If the ADF7021 is used in an application
where it remains in receive mode for a considerable length of
time, the user must consider this variation of filter center
frequency with temperature. This variation is typically 0.7 kHz
per 10°C, which means that if a coarse filter calibration and fine
filter calibration are performed at 25°C, the initial maximum
error is ±0.5 kHz, and the maximum possible change in the
filter center frequency over temperature (−40°C and +85°C) is
±4.5 kHz. This gives a total error of ±5 kHz.
If the receive signal occupied bandwidth is considerably less
than the IF filter bandwidth, the variation of filter center
frequency over the operating temperature range may not be
an issue. Alternatively, if the IF filter bandwidth is not wide
enough to tolerate the variation with temperature, a periodic
filter calibration can be performed, or alternatively, the on-chip
temperature sensor can be used to determine when a filter calibration is necessary by monitoring for changes in temperature.
LNA/PA MATCHING
The ADF7021 exhibits optimum performance in terms of
sensitivity, transmit power, and current consumption, only if its
RF input and output ports are properly matched to the antenna
impedance. For cost sensitive applications, the ADF7021 is
equipped with an internal Rx/Tx switch that facilitates the use
of a simple, combined passive PA/LNA matching network.
Alternatively, an external Rx/Tx switch such as the ADG919 can
be used, which yields a slightly improved receiver sensitivity
and lower transmitter power consumption.
Internal Rx/Tx Switch
Figure 49 shows the ADF7021 in a configuration where
the internal Rx/Tx switch is used with a combined LNA/PA
matching network. This is the configuration used on the EVALADF7021DBX evaluation board. For most applications, the slight
performance degradation of 1 dB to 2 dB caused by the internal
Rx/Tx switch is acceptable, allowing the user to take advantage
of the cost saving potential of this solution. The design of the
combined matching network must compensate for the reactance
presented by the networks in the Tx and the Rx paths, taking
the state of the Rx/Tx switch into consideration.
Rev. B | Page 38 of 64
Data Sheet
ADF7021
VBAT
C1
L1
PA_OUT
PA
ANTENNA
ZOPT_PA
OPTIONAL
BPF OR LPF
ZIN_RFIN
CA
RFIN
LA
LNA
RFINB
05876-022
ZIN_RFIN
CB
ADF7021
Figure 49. ADF7021 with Internal Rx/Tx Switch
The procedure typically requires several iterations until an
acceptable compromise has been reached. The successful implementation of a combined LNA/PA matching network for the
ADF7021 is critically dependent on the availability of an accurate
electrical model for the PCB. In this context, the use of a suitable
CAD package is strongly recommended. To avoid this effort, a
small form-factor reference design for the ADF7021 is provided,
including matching and harmonic filter components. The
design is on a 2-layer PCB to minimize cost. Gerber files are
available at www.analog.com.
External Rx/Tx Switch
Figure 50 shows a configuration using an external Rx/Tx switch.
This configuration allows an independent optimization of the
matching and filter network in the transmit and receive path.
Therefore, it is more flexible and less difficult to design than the
configuration using the internal Rx/Tx switch. The PA is biased
through Inductor L1, while C1 blocks dc current. Together, L1
and C1 form the matching network that transforms the source
impedance into the optimum PA load impedance, ZOPT_PA.
VBAT
OPTIONAL
LPF
C1
L1
PA_OUT
PA
ANTENNA
ZOPT_PA
ZIN_RFIN
OPTIONAL CA
BPF
(SAW)
RFIN
LA
Depending on the antenna configuration, the user may need a
harmonic filter at the PA output to satisfy the spurious emission
requirement of the applicable government regulations. The
harmonic filter can be implemented in various ways, such as
a discrete LC pi or T-stage filter. Dielectric low-pass filter
components, such as the LFL18924MTC1A052 (for operation
in the 915 MHz and 868 MHz band) by Murata Manufacturing
Co. Ltd., represent an attractive alternative to discrete designs.
The immunity of the ADF7021 to strong out-of-band interference
can be improved by adding a band-pass filter in the Rx path.
Apart from discrete designs, SAW or dielectric filter components
such as the SAFCH869MAM0T00, SAFCH915MAL0N00,
DCFB2869MLEJAA-TT1, or DCFB3915MLDJAA-TT1, all by
Murata Manufacturing Co. Ltd., are well-suited for this purpose.
Alternatively, the ADF7021 blocking performance can be improved
by selecting one of the enhanced linearity modes, as described
in Table 14.
IMAGE REJECTION CALIBRATION
The image channel in the ADF7021 is 200 kHz below the desired
signal. The polyphase filter rejects this image with an asymmetric frequency response. The image rejection performance of
the receiver is dependent on how well matched the I and Q
signals are in amplitude, and how well matched the quadrature
is between them (that is, how close to 90° apart they are). The
uncalibrated image rejection performance is approximately
29 dB (at 450 MHz). However, it is possible to improve on this
performance by as much as 20 dB by finding the optimum I/Q
gain and phase adjust settings.
ZIN_RFIN
CB
ADF7021
05876-021
Calibration Using Internal RF Source
ADG919
Rx/Tx – SELECT
LNA
RFINB
Due to the differential LNA input, the LNA matching network
must be designed to provide both a single-ended-to-differential
conversion and a complex, conjugate impedance match. The
network with the lowest component count that can satisfy these
requirements is the configuration shown in Figure 50, consisting
of two capacitors and one inductor. A first-order implementation
of the matching network can be obtained by understanding the
arrangement as two L-type matching networks in a back-toback configuration. Due to the asymmetry of the network with
respect to ground, a compromise between the input reflection
coefficient and the maximum differential signal swing at the
LNA input must be established. The use of appropriate CAD
software is strongly recommended for this optimization.
Figure 50. ADF7021 with External Rx/Tx Switch
ZOPT_PA depends on various factors, such as the required
output power, the frequency range, the supply voltage range,
and the temperature range. Selecting an appropriate ZOPT_PA
helps to minimize the Tx current consumption in the application.
Application Note AN-764 contains a number of ZOPT_PA values
for representative conditions. Under certain conditions, however,
it is recommended to obtain a suitable ZOPT_PA value by means
of a load-pull measurement.
With the LNA powered off, an on-chip generated, low level RF
tone is applied to the mixer inputs. The LO is adjusted to make
the tone fall at the image frequency where it is attenuated by the
image rejection of the IF filter. The power level of this tone is then
measured using the RSSI readback. The I/Q gain and phase adjust
DACs (R5_DB[20:31]) are adjusted and the RSSI is remeasured.
This process is repeated until the optimum values for the gain
and phase adjust are found that provide the lowest RSSI readback
level, thereby maximizing the image rejection performance of
the receiver.
Rev. B | Page 39 of 64
ADF7021
Data Sheet
ADF7021
RFIN
LNA
RFINB
GAIN ADJUST
MUX
INTERNAL
SIGNAL
SOURCE
POLYPHASE
IF FILTER
RSSI/
LOG AMP
7-BIT ADC
PHASE ADJUST
Q
I
FROM LO
SERIAL
INTERFACE
4
PHASE ADJUST
REGISTER 5
RSSI READBACK
4
GAIN ADJUST
REGISTER 5
MICROCONTROLLER
05876-072
I/Q GAIN/PHASE ADJUST AND
RSSI MEASUREMENT
ALGORITHM
Figure 51. Image Rejection Calibration Using the Internal Calibration Source and a Microcontroller
Calibration Using External RF Source
IR calibration can also be implemented using an external RF
source. The IR calibration procedure is the same as that used for
the internal RF source, except that an RF tone is applied to the
LNA input.
Calibration Procedure and Setup
applied to either the I or Q channel, depending on the value of
IR_GAIN_ADJUST_I/Q bit (R5_DB30), whereas the
IR_GAIN_ADJUST_UP/DN bit (R5_DB31) sets whether
the gain adjustment defines a gain or an attenuation adjust.
The calibration results are valid over changes in the ADF7021
supply voltage. However, there is some variation with temperature.
A typical plot of variation in image rejection over temperature
after initial calibrations at −40°C, +25°C, and +85°C is shown in
Figure 52. The internal temperature sensor on the ADF7021 can
be used to determine if a new IR calibration is required.
The IR calibration algorithm available from Analog Devices, Inc. is
based on a low complexity, 2D optimization algorithm that can
be implemented in an external microprocessor or microcontroller.
The magnitude of the phase adjust is set by using the IR_PHASE_
ADJUST_MAG bits (R5_DB[20:23]). This correction can be
applied to either the I channel or Q channel, depending on the
value of the IR_PHASE_ADJUST_DIRECTION bit (R5_DB24).
CAL AT +25°C
50
IMAGE REJECTION (dB)
To enable the internal RF source, the IR_CAL_SOURCE_
DRIVE_LEVEL bits (R6_DB[28:29]) should be set to the
maximum level. The LNA should be set to its minimum gain
setting, and the AGC should be disabled if the internal source is
being used. Alternatively, an external RF source can be used.
60
The magnitude of the I/Q gain is adjusted by the IR_GAIN_
ADJUST_MAG bits (R5_DB[25:29]). This correction can be
40
CAL AT +85°C
CAL AT –40°C
30
VDD = 3.0V
IF BW = 25kHz
20
10
WANTED SIGNAL:
RF FREQ = 430MHz
MODULATION = 2FSK
DATA RATE = 9.6kbps,
PRBS9
fDEV = 4kHz
LEVEL= –100dBm
0
–60
–40
–20
0
INTERFERER SIGNAL:
RF FREQ = 429.8MHz
MODULATION = 2FSK
DATA RATE = 9.6kbps,
PRBS11
fDEV = 4kHz
20
40
TEMPERATURE (°C)
60
80
100
05876-067
Using the internal RF source, the RF frequencies that can be
utilized for image calibration are programmable and are odd
multiples of the reference frequency.
Figure 52. Image Rejection Variation with Temperature after Initial
Calibrations at −40°C, +25°C, and +85°C
Rev. B | Page 40 of 64
Data Sheet
ADF7021
PACKET STRUCTURE AND CODING
particular application, such as setting up sync byte detection or
enabling AFC. When going from Tx to Rx or vice versa, the
user needs to toggle the Tx/Rx bit and write only to Register 0
to alter the LO by 100 kHz.
PREAMBLE
SYNC
WORD
ID
FIELD
DATA FIELD
CRC
05876-023
The suggested packet structure to use with the ADF7021 is
shown in Figure 53.
Figure 53. Typical Format of a Transmit Protocol
Refer to the Receiver Setup section for information on the
required preamble structure and length for the various modulation
schemes.
PROGRAMMING AFTER INITIAL POWER-UP
Table 22 lists the minimum number of writes needed to set up
the ADF7021 in either Tx or Rx mode after CE is brought high.
Additional registers can also be written to tailor the part to a
Table 22. Minimum Register Writes Required for Tx/Rx Setup
Mode
Tx
Rx
Tx to Rx and Rx to Tx
Reg 1
Reg 1
Reg 0
Reg 3
Reg 3
Registers
Reg 0 Reg 2
Reg 0 Reg 5
Reg 4
The recommended programming sequences for transmit and
receive are shown in Figure 54 and Figure 55, respectively. The
difference in the power-up routine for a TCXO and XTAL
reference is shown in these figures.
Rev. B | Page 41 of 64
ADF7021
Data Sheet
TCXO
REFERENCE
XTAL
REFERENCE
POWER-DOWN
CE LOW
CE HIGH
WAIT 10µs + 1ms
(REGULATOR POWER-UP + TYPICAL XTAL SETTLING)
CE HIGH
WAIT 10µs (REGULATOR POWER-UP)
WRITE TO REGISTER 1 (TURNS ON VCO)
WAIT 0.7ms (TYPICAL VCO SETTLING)
WRITE TO REGISTER 3 (TURNS ON Tx/Rx CLOCKS)
WRITE TO REGISTER 0 (TURNS ON PLL)
WAIT 40µs (TYPICAL PLL SETTLING)
WRITE TO REGISTER 2 (TURNS ON PA)
WAIT FOR PA TO RAMP UP (ONLY IF PA RAMP ENABLED)
Tx MODE
WAIT FOR Tx LATENCY NUMBER OF BITS
(REFER TO TABLE 12)
WRITE TO REGISTER 2 (TURNS OFF PA)
WAIT FOR PA TO RAMP DOWN
05876-086
CE LOW
POWER-DOWN
OPTIONAL. ONLY NECESSARY IF PA
RAMP DOWN IS REQUIRED.
Figure 54. Power-Up Sequence for Transmit Mode
Rev. B | Page 42 of 64
Data Sheet
ADF7021
TCXO
REFERENCE
XTAL
REFERENCE
POWER-DOWN
CE LOW
CE HIGH
WAIT 10µs + 1ms
(REGULATOR POWER-UP + TYPICAL XTAL SETTLING)
CE HIGH
WAIT 10µs (REGULATOR POWER-UP)
WRITE TO REGISTER 1 (TURNS ON VCO)
WAIT 0.7ms (TYPICAL VCO SETTLING)
WRITE TO REGISTER 3 (TURNS ON Tx/Rx CLOCKS)
WRITE TO REGISTER 6 (SETS UP IF FILTER CALIBRATION)
OPTIONAL:
ONLY NECESSARY IF
IF FILTER FINE CAL IS REQUIRED.
WRITE TO REGISTER 5 (STARTS IF FILTER CALIBRATION)
WAIT 0.2ms (COARSE CAL) OR WAIT 5.2ms
(COARSE CALIBRATION + FINE CALIBRATION)
WRITE TO REGISTER 11 (SET UP SWD)
WRITE TO REGISTER 12 (ENABLE SWD)
OPTIONAL:
ONLY NECESSARY IF
SWD IS REQUIRED.
WRITE TO REGISTER 0 (TURNS ON PLL)
WAIT 40µs (TYPICAL PLL SETTLING)
WRITE TO REGISTER 4 (TURNS ON DEMOD)
WRITE TO REGISTER 10 (TURNS ON AFC)
OPTIONAL:
ONLY NECESSARY IF
AFC IS REQUIRED.
Rx MODE
05876-087
CE LOW
POWER-DOWN
OPTIONAL.
Figure 55. Power-Up Sequence for Receive Mode
Rev. B | Page 43 of 64
ADF7021
Data Sheet
APPLICATIONS CIRCUIT
For recommended component values, refer to the ADF7021
evaluation board data sheet and AN-915 application note
accessible from the ADF7021 product page. Follow the
reference design schematic closely to ensure optimum
performance in narrow-band applications.
The ADF7021 requires very few external components for
operation. Figure 56 shows the recommended application
circuit. Note that the power supply decoupling and regulator
capacitors are omitted for clarity.
LOOP FILTER
VDD
TCXO
EXT VCO L*
CVCO
CAP
REFERENCE
4
RFOUT
5
RFGND
6
RFIN
38
OSC2
39
OSC1
41
40
VDD3
CREG3
42
CPOUT
44
45
L2
43
VDD
GND
46
47
37
CLKOUT 36
TxRxCLK 35
SWD 33
VDD2
32
ADF7021
ADCIN 30
RFINB
8
RLNA
GND2
29
9
VDD4
SCLK
28
10
RSET
SREAD
SDATA
CREG4
SLE
25
CE
TEST_A
TO
MICROCONTROLLER
CONFIGURATION
INTERFACE
27
26
24
23
FILT_Q
GND4
FILT_Q
22
GND4
21
20
FILT_I
RSET
RESISTOR
19
MIX_Q
FILT_I
18
17
13
RLNA
RESISTOR
MIX_Q
GND4
MIX_I
12
VDD
CREG2 31
7
11
TO
MICROCONTROLLER
Tx/Rx SIGNAL
INTERFACE
TxRxDATA 34
16
VDD
VDD1
MIX_I
T-STAGE LC
FILTER
3
15
VDD
VCOIN
CREG1
14
ANTENNA
CONNECTION
2
MUXOUT
1
MATCHING
L1
VDD
GND1
CVCO
48
VDD
CHIP ENABLE
TO MICROCONTROLLER
NOTES
1. PINS [13:18], PINS [20:21], AND PIN 23 ARE TEST PINS AND ARE NOT USED IN NORMAL OPERATION.
Figure 56. Typical Application Circuit (Regulator Capacitors and Power Supply Decoupling Not Shown)
Rev. B | Page 44 of 64
05876-084
*PIN 44 AND PIN 46 CAN BE LEFT FLOATING IF EXTERNAL INDUCTOR VCO IS NOT USED.
Data Sheet
ADF7021
SERIAL INTERFACE
RSSI Readback
The serial interface allows the user to program the 16-/32-bit
registers using a 3-wire interface (SCLK, SDATA, and SLE).
It consists of a level shifter, 32-bit shift register, and 16 latches.
Signals should be CMOS compatible. The serial interface is
powered by the regulator, and, therefore, is inactive when CE is low.
The format of the readback word is shown in Figure 57. It
comprises the RSSI-level information (Bit RV1 to Bit RV7), the
current filter gain (FG1, FG2), and the current LNA gain (LG1,
LG2) setting. The filter and LNA gain are coded in accordance
with the definitions in the Register 9—AGC Register section. For
signal levels below −100 dBm, averaging the measured RSSI values
improves accuracy. The input power can be calculated from the
RSSI readback value as outlined in the RSSI/AGC section.
Data is clocked into the register, MSB first, on the rising edge of
each clock (SCLK). Data is transferred to one of 16 latches on the
rising edge of SLE. The destination latch is determined by the
value of the four control bits (C4 to C1); these are the bottom
4 LSBs, DB3 to DB0, as shown in Figure 2. Data can also be read
back on the SREAD pin.
Battery Voltage/ADCIN/Temperature Sensor Readback
The battery voltage is measured at Pin VDD4. The readback
information is contained in Bit RV1 to Bit RV7. This also
applies for the readback of the voltage at the ADCIN pin and
the temperature sensor. From the readback information, the
battery or ADCIN voltage can be determined using
READBACK FORMAT
The readback operation is initiated by writing a valid control
word to the readback register and enabling the READBACK bit
(R7_DB8 = 1). The readback can begin after the control word
has been latched with the SLE signal. SLE must be kept high
while the data is being read out. Each active edge at the SCLK
pin successively clocks the readback word out at the SREAD
pin, as shown in Figure 57, starting with the MSB first. The data
appearing at the first clock cycle following the latch operation
must be ignored. An extra clock cycle is needed after the 16th
readback bit to return the SREAD pin to tristate. Therefore, 18
total clock cycles are needed for each read back. After the 18th
clock cycle, SLE should be brought low.
VBATTERY = (BATTERY_VOLTAGE_READBACK)/21.1
VADCIN = (ADCIN_VOLTAGE_READBACK)/42.1
The temperature can be calculated using
Temp [°C] = −40 + (68.4 − TEMP_READBACK) × 9.32
Silicon Revision Readback
The silicon revision readback word is valid without setting any
other registers. The silicon revision word is coded with four
quartets in BCD format. The product code (PC) is coded with
three quartets extending from Bit RV5 to Bit RV16. The revision
code (RC) is coded with one quartet extending from Bit RV1 to
Bit RV4. The product code for the ADF7021 should read back
as PC = 0x210. The current revision code should read as RC = 0x4.
AFC Readback
The AFC readback is valid only during the reception of FSK
signals with either the linear or correlator demodulator active.
The AFC readback value is formatted as a signed 16-bit integer
comprising Bit RV1 to Bit RV16 and is scaled according to the
following formula:
Filter Bandwidth Calibration Readback
The filter calibration readback word is contained in Bit RV1 to
Bit RV8. This readback can be used for manual filter adjust, thereby
avoiding the need to do an IF filter calibration in some instances.
The manual adjust value is programmed by R5_DB[14:19]. To
calculate the manual adjust based on a filter calibration readback,
use the following formula:
FREQ RB [Hz] = (AFC_READBACK × DEMOD CLK)/218
In the absence of frequency errors, FREQ RB is equal to the IF
frequency of 100 kHz. Note that, for the AFC readback to yield
a valid result, the down converted input signal must not fall outside
the bandwidth of the analog IF filter. At low input signal levels,
the variation in the readback value can be improved by averaging.
IF_FILTER_ADJUST = FILTER_CAL_READBACK − 128
The result should be programmed into R5_DB[14:19] as outlined
in the Register 5—IF Filter Setup Register section.
DB15
DB14
DB13
DB12
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
AFC READBACK
RV16
RV15
RV14
RV13
RV12
RV11
RV10
RV9
RV8
RV7
RV6
RV5
RV4
RV3
RV2
RV1
RSSI READBACK
X
X
X
X
X
LG2
LG1
FG2
FG1
RV7
RV6
RV5
RV4
RV3
RV2
RV1
BATTERY VOLTAGE/ADCIN/
TEMP. SENSOR READBACK
X
X
X
X
X
X
X
X
X
RV7
RV6
RV5
RV4
RV3
RV2
RV1
SILICON REVISION
RV16
RV15
RV14
RV13
RV12
RV11
RV10
RV9
RV8
RV7
RV6
RV5
RV4
RV3
RV2
RV1
FILTER CAL READBACK
0
0
0
0
0
0
0
0
RV8
RV7
RV6
RV5
RV4
RV3
RV2
RV1
Figure 57. Readback Value Table
Rev. B | Page 45 of 64
05876-029
READBACK VALUE
READBACK MODE
ADF7021
Data Sheet
INTERFACING TO MICROCONTROLLER/DSP
SPI Mode
Standard Transmit/Receive Data Interface
In SPI mode, the TxRxCLK pin is configured to input transmit
data in transmit mode. In receive mode, the receive data is available
on the TxRxDATA pin. The data clock in both transmit and receive
modes is available on the CLKOUT pin. In transmit mode, data
is clocked into the ADF7021 on the positive edge of CLKOUT.
In receive mode, the TxRxDATA data pin should be sampled by
the microcontroller on the positive edge of the CLKOUT.
The standard transmit/receive signal and configuration interface
to a microcontroller is shown in Figure 58. In transmit mode,
the ADF7021 provides the data clock on the TxRxCLK pin, and
the TxRxDATA pin is used as the data input. The transmit data
is clocked into the ADF7021 on the rising edge of TxRxCLK.
ADF7021
To enable SPI interface mode, set R0_DB28 high and set
R15_DB[17:19] to 0x7. Figure 8 and Figure 9 show the relevant
timing diagrams for SPI mode, while Figure 60 shows the
recommended interface to a microcontroller using the SPI
mode of the ADF7021.
MOSI
TxRxCLK
SCLOCK
SS
CE
P3.7
SWD
GPIO
P2.4
SREAD
P2.5
SLE
P2.6
SDATA
P2.7
SCLK
MICROCONTROLLER
05876-026
P3.2/INT0
SPI
Figure 58. ADuC84x to ADF7021 Connection Diagram
TxDATA
TxRxDATA
SCLK
Figure 60. ADF7021 (SPI Mode) to Microcontroller Interface
ADSP-BF533 interface
The suggested method of interfacing to the Blackfin® ADSPBF533 is given in Figure 61.
CE
SWD
SREAD
SLE
SDATA
SCLK
05876-085
GPIO
SLE
SDATA
ADF7021
RxDATA
CLKOUT
SREAD
In UART mode, the TxRxCLK pin is configured to input transmit
data in transmit mode. In receive mode, the receive data is available
on the TxRxDATA pin, thus providing an asynchronous data
interface. The UART mode can only be used with oversampled
2FSK. Figure 59 shows a possible interface to a microcontroller
using the UART mode of the ADF7021. To enable this UART
interface mode, set R0_DB28 high. Figure 8 and Figure 9 show
the relevant timing diagrams for UART mode.
UART
TxRxDATA
GPIO
Figure 59. ADF7021(UART Mode) to Asynchronous Microcontroller Interface
Rev. B | Page 46 of 64
ADF7021
ADSP-BF533
SCK
SCLK
MOSI
SDATA
MISO
SREAD
PF5
TxRxCLK
MOSI
SCLK
SWD
UART Mode
MICROCONTROLLER
TxRxCLK
CE
In receive mode, the ADF7021 provides the synchronized data
clock on the TxRxCLK pin. The receive data is available on the
TxRxDATA pin. The rising edge of TxRxCLK should be used to
clock the receive data into the microcontroller. Refer to Figure 4
and Figure 5 for the relevant timing diagrams.
In 4FSK transmit mode, the MSB of the transmit symbol is
clocked into the ADF7021 on the first rising edge of the data
clock from the TxRxCLK pin. In 4FSK receive mode, the MSB
of the first payload symbol is clocked out on the first negative
edge of the data clock after the SWD, and should be clocked
into the microcontroller on the following rising edge. Refer to
Figure 6 and Figure 7 for the relevant timing diagrams.
ADF7021
MISO
RSCLK1
DT1PRI
SLE
TxRxCLK
TxRxDATA
DR1PRI
RFS1
PF6
SWD
CE
Figure 61. ADSP-BF533 to ADF7021 Connection Diagram
05876-027
TxRxDATA
MISO
05876-076
ADuC84x
Data Sheet
ADF7021
M3
M2
M1
MUXOUT
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
REGULATOR_READY (DEFAULT)
FILTER_CAL_COMPLETE
DIGITAL_LOCK_DETECT
RSSI_READY
Tx_Rx
LOGIC_ZERO
TRISTATE
LOGIC_ONE
DB1
DB0
C2 (0)
C1 (0)
DB3
DB2
DB4
M1
C4 (0)
C3 (0)
DB5
M2
DB6
DB7
M4
M3
DB8
DB12
M9
M5
DB13
M10
DB9
DB14
M11
DB10
DB15
M6
DB16
M13
M12
M7
DB17
DB11
M15
M14
M13
...
M3
M2
M1
FRACTIONAL
DIVIDE RATIO
0
0
0
.
.
.
1
1
1
1
0
0
0
.
.
.
1
1
1
1
0
0
0
.
.
.
1
1
1
1
...
...
...
...
...
...
...
...
...
...
0
0
0
.
.
.
1
1
1
1
0
0
1
.
.
.
0
0
1
1
0
1
0
.
.
.
0
1
0
1
0
1
2
.
.
.
32764
32765
32766
32767
N8
N7
N6
N5
N4
N3
N2
N1
N COUNTER
DIVIDE RATIO
0
0
.
.
.
1
0
0
.
.
.
1
0
0
.
.
.
1
1
1
.
.
.
1
0
1
.
.
.
1
1
0
.
.
.
1
1
0
.
.
.
0
1
0
.
.
.
1
23
24
.
.
.
253
1
1
1
1
1
1
1
0
254
1
1
1
1
1
1
1
1
255
05876-030
DISABLED
ENABLED
M14
N2
UART MODE
0
1
DB18
DB21
DB20
N3
U1
DB19
DB22
N4
TRANSMIT
RECEIVE
N1
DB23
0
1
M15
DB24
N6
N5
TRANSMIT/
RECEIVE
M8
Tx/Rx
DB25
DB28
U1
N7
DB29
DB26
TR1
ADDRESS
BITS
15-BIT FRACTIONAL_N
DB27
DB30
M2
M1
8-BIT INTEGER_N
N8
DB31
M3
MUXOUT
TR1
UART
MODE
REGISTER 0—N REGISTER
Figure 62. Register 0—N Register Map
The RF output frequency is calculated by the following:
For the direct output
Fractional _ N 

RFOUT = PFD ×  Integer _ N +

2 15


For the RF_DIVIDE_BY_2 (DB18) selected
Fractional _ N 

RFOUT = PFD × 0.5 ×  Integer _ N +

2 15


In UART/SPI mode, the TxRxCLK pin is used to input the Tx
data. The Rx Data is available on the TxRxDATA pin.
In the MUXOUT map in Figure 62, the FILTER_CAL_COMPLETE
indicates when a coarse or coarse plus fine IF filter calibration
has finished. The DIGITAL_LOCK_DETECT indicates when
the PLL has locked. The RSSI_READY indicates that the RSSI
signal has settled and an RSSI readback can be performed.
Tx_Rx gives the status of DB27 in this register, which can be
used to control an external Tx/Rx switch.
Rev. B | Page 47 of 64
ADF7021
Data Sheet
VB4
0
0
.
1
VB3
0
0
.
1
VCO
VB1
1
0
.
1
DB8
DB7
DB6
DB5
DB4
DB3
CL2
CL1
R3
R2
R1
C4 (0)
LOOP
CONDITION
0
1
VCO OFF
VCO ON
INTERNAL L VCO
EXTERNAL L VCO
D1
0
1
CP2
0
0
1
1
CP1
RSET
0
1
0
1
ICP (mA)
3.6kΩ
0.3
0.9
1.5
2.1
CL1
0
1
0
.
.
.
1
DB0
DB9
VE1
CL2
0
0
1
.
.
.
1
C1 (1)
DB10
CL4
CL3
3.75mA
CL3
0
0
0
.
.
.
1
DB2
DB11
D1
CL4
0
0
0
.
.
.
1
DB1
DB12
X1
VCO BIAS
CURRENT
0.25mA
0.5mA
R1
1
0
.
.
.
1
C3 (0)
DB13
XB1
OFF
ON
R2
0
1
.
.
.
1
C2 (0)
DB14
XB2
XOSC_
ENABLE
XTAL_
DOUBLER
DB15
CP1
CP_
CURRENT
DB17
DB16
R3
0
0
.
.
.
1
RFD1 RF DIVIDE BY 2
0
1
ADDRESS
BITS
R_COUNTER
RF R COUNTER
DIVIDE RATIO
1
2
.
.
.
7
CLKOUT
DIVIDE RATIO
OFF
2
4
.
.
.
30
XTAL
DOUBLER
DISABLE
ENABLED
X1 XTAL OSC
OFF
0
ON
1
XB2 XB1
0
0
1
1
0
1
0
1
XTAL
BIAS
20µA
25µA
30µA
35µA
05876-031
NOMINAL
VCO ADJUST UP 1
VCO ADJUST UP 2
VCO ADJUST UP 3
CLOCKOUT_
DIVIDE
VE1
VB1
0
1
0
1
XTAL_
BIAS
CP2
DB19
VB2
VA1
0
0
1
1
VB2
0
1
.
1
RF_DIVIDE_
BY_2
VCO_
ENABLE
DB20
VB3
VCO CENTER
FREQ ADJUST
RFD1 DB18
DB22
DB21
VB4
DB23
VA1
VCO_
ADJUST
DB24
VCO_BIAS
VA2
VCL1
0
1
VA2
VCL1 DB25
VCO_
INDUCTOR
REGISTER 1—VCO/OSCILLATOR REGISTER
Figure 63. Register 1—VCO/Oscillator Register Map
The R_COUNTER and XTAL_DOUBLER relationship is
expressed as follows:
If XTAL_DOUBLER = 0,
PFD =
XTAL
R _ COUNTER
If XTAL_DOUBLER =1,
PFD =
XTAL × 2
R _ COUNTER
The CLOCKOUT_DIVIDE is a divided-down and inverted
version of the XTAL and is available on Pin 36 (CLKOUT).
Set XOSC_ENABLE high when using an external crystal. If
using an external oscillator (such as TCXO) with CMOS-level
outputs into Pin OSC2, set XOSC_ENABLE low. If using an
external oscillator with a 0.8 V p-p clipped sine wave output
into Pin OSC1, set XOSC_ENABLE high.
The VCO_BIAS bits should be set according to Table 9.
The VCO_ADJUST bits adjust the center of the VCO operating
band. Each bit typically adjusts the VCO band up by 1% of the
RF operating frequency (0.5% if RF_DIVIDE_BY_2 is enabled).
Setting VCO_INDUCTOR to external allows the use of the
external inductor VCO, which gives RF operating frequencies of
80 Hz to 650 MHz. If the internal inductor VCO is being used
for operation, set this bit low.
Rev. B | Page 48 of 64
Data Sheet
ADF7021
DI1
TxDATA INVERT
0
0
1
1
0
1
0
1
NORMAL
INVERT CLK
INVERT DATA
INV CLK AND DATA
TFD9 ...
0
0
0
0
.
1
...
...
...
...
...
...
NRC1 RAISED COSINE ALPHA
0
1
0.5 (Default)
0.7
0
0
0
0
.
1
0
0
1
1
.
1
0
1
0
1
.
1
MODULATION_
SCHEME
DB6
DB5
DB4
DB3
DB2
DB1
DB0
S2
S1
C4 (0)
C3 (0)
C2 (1)
C1 (0)
PA2
PA1
PA BIAS
PE1
POWER AMPLIFIER
0
0
1
1
0
1
0
1
5µA
7µA
9µA
11µA
0
1
OFF
ON
PR3
PR2
PR1
PA RAMP RATE
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
NO RAMP
256 CODES/BIT
128 CODES/BIT
64 CODES/BIT
32 CODES/BIT
16 CODES/BIT
8 CODES/BIT
4 CODES/BIT
P6
.
.
P2
P1
PA LEVEL
0
0
0
0
.
.
1
.
.
.
.
.
.
1
.
.
.
.
.
.
.
0
0
1
1
.
.
1
0
1
0
1
.
.
1
0 (PA OFF)
1 (–16.0 dBm)
2
3
.
.
63 (13 dBm)
ADDRESS
BITS
S3
PA_
ENABLE
DB7
DB8
PE1
PR1
DB9
PR2
DB11
PA1
DB10
DB12
PA2
PR3
DB13
DB14
P1
PA_RAMP
P2
DB16
P4
0
1
2
3
.
511
DB15
DB17
P5
TFD3 TFD2 TFD1 fDEV
PA_BIAS
P3
DB18
P6
TFD1 DB19
TFD2 DB20
TFD4 DB22
TFD3 DB21
POWER_AMPLIFIER
S3
S2
S1
MODULATION SCHEME
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
2FSK
GAUSSIAN 2FSK
3FSK
4FSK
OVERSAMPLED 2FSK
RAISED COSINE 2FSK
RAISED COSINE 3FSK
RAISED COSINE 4FSK
05876-032
DI2
TFD5 DB23
TFD6 DB24
TFD8 DB26
DB28
DI1
Tx_FREQUENCY_DEVIATION
TFD9 DB27
DB29
DI2
NRC1 DB30
TxDATA_
INVERT
TFD7 DB25
R-COSINE_
ALPHA
REGISTER 2—TRANSMIT MODULATION REGISTER
Figure 64. Register 2—Transmit Modulation Register Map
The 2FSK/3FSK/4FSK frequency deviation is expressed by the
following:
Direct output
Frequency Deviation [Hz] =
Tx_FREQUENCY_DEVIATION × PFD
2 16
With RF_DIVIDE_BY_2 (R1_DB18) enabled
In the case of 4FSK, there are tones at ±3 × the frequency
deviation and at ±1 × the deviation. The power amplifier (PA)
ramps at the programmed rate (R2_DB[8:10]) until it reaches
its programmed level DB[13:18]. If the PA is enabled/disabled
by the PA_ENABLE bit (DB7), it ramps up and down. If it is
enabled/disabled by the Tx/Rx bit (R0_DB27), it ramps up and
turns hard off.
R-COSINE_ALPHA sets the roll-off factor (alpha) of the raised
cosine data filter to either 0.5 or 0.7. The alpha is set to 0.5 by
default, but the raised cosine filter bandwidth can be increased
to provide less aggressive data filtering by using an alpha of 0.7.
Frequency Deviation [Hz] =
Tx_FREQUENCY_DEVIATION × PFD
0. 5 ×
2 16
where Tx_FREQUENCY_DEVIATION is set by DB[19:27] and
PFD is the PFD frequency.
Rev. B | Page 49 of 64
ADF7021
Data Sheet
GD4
GD3
GD2
GD1
AGC CLK DIVIDE
0
0
...
1
0
0
...
1
0
0
...
1
0
0
...
1
0
0
...
1
0
1
...
1
INVALID
1
...
127
DB5
BK2
DB0
DB6
OK1
C1 (1)
DB7
OK2
DB1
DB8
OK3
C2 (1)
DB9
OK4
DB2
DB10
FS1
DB3
DB11
FS2
C3 (0)
DB12
FS3
C4 (0)
DB13
FS4
DB4
DB14
FS5
BK1
DB15
FS6
ADDRESS
BITS
SK3
SK2
SK1
SEQ CLK DIVIDE
BK2
BK1
BBOS CLK DIVIDE
0
0
.
1
1
0
1
.
1
1
1
0
.
0
1
1
2
.
254
255
0
0
1
1
0
1
0
1
4
8
16
32
OK4 OK3 OK2 OK1
DEMOD CLK DIVIDE
0
0
...
1
INVALID
1
...
15
0
0
...
1
0
0
...
1
0
1
...
1
FS8
FS7
...
FS3
FS2
FS1
CDR CLK DIVIDE
0
0
.
1
1
0
0
.
1
1
...
...
...
...
...
0
0
.
1
1
0
1
.
1
1
1
0
.
0
1
1
2
.
254
255
05876-033
GD5
DB16
...
...
...
...
...
...
DB17
0
0
.
1
1
FS7
DB22
SK5
DB18
DB23
SK6
SK7
0
0
.
1
1
FS8
DB24
SK7
SK1
DB25
SK8
DB19
DB26
GD1
DB20
DB27
GD2
SK2
DB28
GD3
SK3
DB29
GD4
DB21
DB30
GD5
SK8
GD6
DEM_CLK_
DIVIDE
CDR_CLK_DIVIDE
SK4
DB31
SEQUENCER_CLK_DIVIDE
GD6
AGC_CLK_DIVIDE
BBOS_CLK_
DIVIDE
REGISTER 3—TRANSMIT/RECEIVE CLOCK REGISTER
Figure 65. Register 3—Transmit/Receive Clock Register Map
Baseband offset clock frequency (BBOS CLK) must be greater
than 1 MHz and less than 2 MHz, where
XTAL
BBOS CLK =
BBOS _ CLK _ DIVIDE
CDR CLK =
CDR _ CLK _ DIVIDE
The sequencer clock (SEQ CLK) supplies the clock to the digital
receive block. It should be as close to 100 kHz as possible.
Set the demodulator clock (DEMOD CLK) such that 2 MHz ≤
DEMOD CLK ≤ 15 MHz, where
DEMOD CLK =
DEMOD CLK
XTAL
DEMOD _ CLK _ DIVIDE
SEQ CLK =
XTAL
SEQ _ CLK _ DIVIDE
The time allowed for each AGC step to settle is determined by
the AGC update rate. It should be set close to 10 kHz.
For 2FSK/3FSK, the data/clock recovery frequency (CDR CLK)
needs to be within 2% of (32 × data rate). For 4FSK, the CDR
CLK needs to be within 2% of (32 × symbol rate).
Rev. B | Page 50 of 64
AGC Update Rate [Hz] =
SEQ CLK
AGC _ CLK _ DIVIDE
Data Sheet
ADF7021
DW10 .
POST DEMOD
DW6 DW5 DW4 DW3 DW2 DW1 BW
0
0
.
.
.
.
1
0
0
.
.
.
.
1
0
0
.
.
.
.
1
0
0
.
.
.
.
1
CROSS PRODUCT
DOT PRODUCT
DB4
DB3
DB2
DB1
DB0
DS1
C4(0)
C3(1)
C2(0)
C1(0)
DB10
TD1
DB5
DB11
TD2
DS2
DB12
TD3
DB6
DB13
TD4
DB7
DB14
TD5
DS3
DB15
TD6
DP1
DB16
TD7
DB8
PRODUCT
0
1
NORMAL
INVERT CLK
INVERT DATA
INVERT CLK/DATA
DS3 DS2 DS1
0
0
0
0
1
1
1
1
1
2
.
.
.
.
1023
TD10 .
TD6
TD5
TD4
TD3
TD2
TD1
CORRELATOR
DISCRIM BW
0
0
.
.
.
.
1
0
0
.
.
.
.
0
0
0
.
.
.
.
1
0
0
.
.
.
.
0
0
0
.
.
.
.
1
0
1
.
.
.
.
0
1
0
.
.
.
.
0
1
2
.
.
.
.
660
.
.
.
.
.
.
.
DP1
ADDRESS
BITS
INVERT
1
0
.
.
.
.
1
0
1
.
.
.
.
1
RI1
DB17
0
1
0
1
DEMOD_
SCHEME
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
DEMODULATOR SCHEME
2FSK LINEAR DEMODULATOR
2FSK CORRELATOR DEMODULATOR
3FSK DEMOD
4FSK DEMOD
RESERVED
RESERVED
RESERVED
RESERVED
05876-034
0
0
.
.
.
.
1
0
0
1
1
DB9
DB18
TD9
TD8
RI2 RI1
RI2
DB19
TD10
DW1 DB20
DW2 DB21
DW3 DB22
DW4 DB23
DW5 DB24
DW6 DB25
DW7 DB26
DW8 DB27
DW9 DB28
DB30
IFB1
DW10 DB29
DB31
IFB2
IF FILTER
IFB2 IFB1 BW
0
0
12.5kHz
1
0
18.75kHz
0
1
25kHz
1
1
INVALID
.
.
.
.
.
.
.
Rx_
INVERT
DISCRIMINATOR_BW
POST_DEMOD_BW
IF_BW
DOT_PRODUCT
REGISTER 4—DEMODULATOR SETUP REGISTER
Figure 66. Register 4—Demodulator Setup Register Map
To solve for DISCRIMINATOR_BW, use the following
equation:
DISCRIMINATOR_BW =
DEMOD CLK × K
where the maximum value = 660.
For 2FSK,
 100 × 10 3
K = Round 
 f DEV
400 × 10 3
where:
Round is rounded to the nearest integer.
Round4FSK is rounded to the nearest of the following integers: 32,
31, 28, 27, 24, 23, 20, 19, 16, 15, 12, 11, 8, 7, 4, 3.
fDEV is the transmit frequency deviation in Hz. For 4FSK, fDEV is
the frequency deviation used for the ±1 symbols (that is, the
inner frequency deviations).
Rx_INVERT (DB[8:9]) and DOT_PRODUCT (DB7) need to be
set as outlined in Table 17 and Table 18.




POST_DEMOD_BW =
For 3FSK,
 100 × 10 3
K = Round 
 2 × f DEV




For 4FSK,
 100 × 10 3
K = Round 4 FSK 
 4 × f DEV




211 × π × f CUTOFF
DEMOD CLK
where the cutoff frequency (fCUTOFF) of the post demodulator
filter should typically be 0.75 × the data rate in 2FSK. In 3FSK,
it should be set equal to the data rate, while in 4FSK, it should
be set equal to 1.6 × symbol rate.
Rev. B | Page 51 of 64
ADF7021
Data Sheet
0
0
0
.
1
0
0
1
.
1
IR GAIN ADJUST UP/DN
0
1
GAIN
ATTENUATE
0
1
2
...
31
IFA6 IFA5 ...
0
0 ...
0
0 ...
0
0 ...
..
.. ...
0
1 ...
1
0 ...
1
0 ...
1
0 ...
1
. ...
1
1 ...
IF_CAL_COARSE
DB0
C1 (1)
DB3
C4 (0)
DB1
DB4
CC1
C2 (0)
DB5
IFD1
DB2
DB6
IFD2
C3 (1)
DB7
IFD3
CC1
CAL
0
1
NO CAL
DO CAL
IFD9 .
FILTER CLOCK
IFD6 IFD5 IFD4 IFD3 IFD2 IFD1 DIVIDE RATIO
0
0
.
.
.
.
1
0
0
.
.
.
.
1
.
.
.
.
.
.
.
0
0
.
.
.
.
1
0
0
.
.
.
.
1
0
0
.
.
.
.
1
0
1
.
.
.
.
1
1
0
.
.
.
.
1
1
2
.
.
.
.
511
IF FILTER
IFA2 IFA1 ADJUST
0
0
1
..
1
0
0
1
.
1
0
1
0
..
1
0
1
0
.
1
0
+1
+2
...
+31
0
–1
–2
...
–31
05876-035
ADJUST I CH
ADJUST Q CH
0
1
0
.
1
DB8
DB20
PM1
0
0
0
.
1
IFD4
DB21
PM2
IR GAIN
GM2 GM1 ADJUST
DB9
DB22
PM3
GM3
ADDRESS
BITS
IFD5
DB23
PM4
GM5 GM4
0
1
2
...
15
IFD7 DB11
DB24
PD1
IR PHASE ADJUST I/Q
ADJUST I CH
ADJUST Q CH
0
1
0
.
1
IFD8 DB12
DB25
GM1
0
0
1
.
1
IF_FILTER_DIVIDER
IFD9 DB13
DB26
GM2
0
0
0
.
1
IFA1 DB14
DB27
GM3
0
0
0
.
1
IFA2 DB15
DB28
GM4
IR PHASE
PM1 PM1 ADJUST
0
1
GA1
IFA3 DB16
DB29
GM5
PM2
GQ1 IR GAIN ADJUST I/Q
0
1
IFA4 DB17
DB30
GQ1
PM3
PD1
0
0
0
.
1
IFA6 DB19
DB31
IF_FILTER_ADJUST
IFD6 DB10
IR_PHASE_
ADJUST_MAG
GA1
IR_GAIN_
ADJUST_MAG
IFA5 DB18
IR PHASE
ADJUST DIRECTION
IR GAIN
ADJUST UP/DN
IR GAIN
ADJUST I/Q
REGISTER 5—IF FILTER SETUP REGISTER
Figure 67. Register 5—IF Filter Setup Register Map
A coarse IF filter calibration is performed when the
IF_CAL_COARSE bit (DB4) is set. If the IF_FINE_CAL bit
(R6_DB4) has been previously set, a fine IF filter calibration is
automatically performed after the coarse calibration.
Set IF_FILTER_DIVIDER such that
XTAL
= 50 kHz
IF _ FILTER _ DIVIDER
IF_FILTER_ADJUST allows the IF fine filter calibration result
to be programmed directly on subsequent receiver power-ups,
thereby saving on the need to redo a fine filter calibration in
some instances. Refer to the Filter Bandwidth Calibration
Readback section for information about using the
IF_FILTER_ADJUST bits.
DB[20:31] are used for image rejection calibration. Refer to the
Image Rejection Calibration section for details on how to
program these parameters.
Rev. B | Page 52 of 64
Data Sheet
ADF7021
IF_FINE_
CAL
DB12
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
LT6
LT5
LT4
LT3
LT2
LT1
FC1
C4 (0)
C3 (1)
C2 (1)
C1 (0)
DB17
UT5
LT7
DB18
UT6
LT8
DB19
UT7
DB13
DB20
UT8
UT1
DB21
CD1
ADDRESS
BITS
DB14
DB22
CD2
DB15
DB23
CD3
UT3
DB24
CD4
IF_CAL_LOWER_TONE_DIVIDE
UT2
DB25
CD5
DB16
DB26
UT4
DB27
IF_CAL_UPPER_TONE_DIVIDE
CD6
IF_CAL_DWELL_TIME
CD7
DB29 IR_CAL_
SOURCE_
IRC1 DB28 DRIVE_LEVEL
IRC2
IRD1 DB30 IR_CAL_
SOURCE ÷2
REGISTER 6—IF FINE CAL SETUP REGISTER
IRD1 IR CAL SOURCE ÷2
0
1
SOURCE ÷2 OFF
SOURCE ÷2 ON
UT8 UT7 ...
0
0
0
.
.
0
IR CAL SOURCE
IRC2 IRC1 DRIVE LEVEL
0
0
OFF
1
0
LOW
0
1
MED
1
1
HIGH
0
0
0
.
.
1
...
...
...
...
...
...
UT3
UT2
UT1
IF CAL UPPER
TONE DIVIDE
0
0
0
.
.
1
0
1
1
.
.
1
1
0
1
.
.
1
1
2
3
.
.
127
LT8 LT7 ... LT3
CD7
...
IF CAL
CD3 CD2 CD1 DWELL TIME
0
0
0
.
.
1
...
...
...
...
...
...
0
0
0
.
.
1
1
0
1
.
.
1
1
2
3
.
.
127
0
0
0
.
.
1
...
...
...
...
...
...
0
0
0
.
.
1
IF FINE CAL
0
1
DISABLED
ENABLED
LT2
LT1
IF CAL LOWER
TONE DIVIDE
0
1
1
.
.
1
1
0
1
.
.
1
1
2
3
.
.
255
05876-036
0
1
1
.
.
1
0
0
0
.
.
1
FC1
Figure 68. Register 6—IF Fine Cal Setup Register Map
A fine IF filter calibration is set by enabling the IF_FINE_CAL
Bit (R6_DB4). A fine calibration is then carried out only when
Register 5 is written to and R5_DB4 is set.
Set the IF upper and lower tones used during fine filter
calibration as follows:
For best practice, is recommended to have the IF tone
calibration time be at least 500 µs.
IF Tone Calibration Time =
IF _ CAL _ DWELL _ TIME
SEQ CLK
The total time for a fine IF filter calibration is
XTAL
= 65.8 kHz
IF _ CAL _ LOWER _ TONE _ DIVIDE × 2
IF Tone Calibration Time × 10.
XTAL
= 131.5 kHz
IF _ CAL _ UPPER _ TONE _ DIVIDE × 2
The IF tone calibration time is the amount of time that is spent
at an IF calibration tone. It is dependent on the sequencer clock.
DB[28:30] control the internal source for the image rejection
(IR) calibration. The IR_CAL_SOURCE_DRIVE_LEVEL bits
(DB[28:29]) set the drive strength of the source, whereas the
IR_CAL_SOURCE_÷2 bit (DB30) allows the frequency of the
internal signal source to be divided by 2.
Rev. B | Page 53 of 64
ADF7021
Data Sheet
REGISTER 7—READBACK SETUP REGISTER
READBACK
SELECT
ADC
MODE
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
RB3
RB2
RB1
AD2
AD1
C4 (0)
C3 (1)
C2 (1)
C1 (1)
RB3 READBACK
0
1
CONTROL
BITS
AD2 AD1 ADC MODE
DISABLED
ENABLED
RB2 RB1 READBACK MODE
0
1
0
1
0
1
0
1
MEASURE RSSI
BATTERY VOLTAGE
TEMP SENSOR
TO EXTERNAL PIN
AFC WORD
ADC OUTPUT
FILTER CAL
SILICON REV
05876-037
0
0
1
1
0
0
1
1
Figure 69. Register 7—Readback Setup Register Map
Readback of the measured RSSI value is valid only in Rx mode.
Readback of the battery voltage, temperature sensor, or voltage
at the external pin is not valid in Rx mode.
For AFC readback, use the following equations (see the
Readback Format section):
To read back the battery voltage, the temperature sensor, or the
voltage at the external pin in Tx mode, users should first power
up the ADC using R8_DB8 because it is turned off by default in
Tx mode to save power.
Rev. B | Page 54 of 64
FREQ RB [Hz] = (AFC_READBACK × DEMOD CLK)/218
VBATTERY = BATTERY_VOLTAGE_READBACK/21.1
VADCIN = ADCIN_VOLTAGE_READBACK/42.1
Temperature [°C] = −40 + (68.4 − TEMP_READBACK) × 9.32
Data Sheet
ADF7021
PD6
SYNTH_
ENABLE
LE1
RESERVED
DB10
SW1
LNA/MIXER_
ENABLE
DB11
PD7
FILTER_
ENABLE
DB12
ADC_
ENABLE
DB13
DEMOD_
ENABLE
DB14
CR1
LOG_AMP_
ENABLE
DB15
Tx/Rx_SWITCH_
ENABLE
Rx_RESET
PA_ENABLE_
Rx_MODE
COUNTER_
RESET
REGISTER 8—POWER-DOWN TEST REGISTER
DB9
DB8
DB7
DB6
DB5
DB4
PD5
PD4
PD3
PD1
CR1 COUNTER RESET
0
1
NORMAL
RESET
CONTROL
BITS
DB3
DB2
DB1
DB0
C4 (1) C3 (0) C2 (0) C1 (0)
PD1
SYNTH STATUS
0
1
SYNTH OFF
SYNTH ON
CDR
RESET
DEMOD
RESET
PD7
PA (Rx MODE)
0
1
PA OFF
PA ON
SW1 Tx/Rx SWITCH
DEFAULT (ON)
OFF
LE1
LOG AMP ENABLE
0
1
LOG AMP OFF
LOG AMP ON
PD6
DEMOD ENABLE
0
1
DEMOD OFF
DEMOD ON
LNA/MIXER ENABLE
0
1
LNA/MIXER OFF
LNA/MIXER ON
PD4
FILTER ENABLE
0
1
FILTER OFF
FILTER ON
PD5
ADC ENABLE
0
1
ADC OFF
ADC ON
05876-038
0
1
PD3
Figure 70. Register 8—Power-Down Test Register Map
It is not necessary to write to this register under normal
operating conditions.
For a combined LNA/PA matching network, DB11 should
always be set to 0, which enables the internal Tx/Rx switch. This
is the power-up default condition.
Rev. B | Page 55 of 64
ADF7021
Data Sheet
0
1
FILTER CURRENT
0
1
LOW
HIGH
0
1
0
1
DB1
DB0
C2 (0)
C1 (1)
DB9
GL6
DB2
DB10
GL7
DB3
DB12
DB11
DB13
GH3
GH1
0
0
0
0
.
.
.
1
1
1
0
0
0
0
.
.
.
1
1
1
0
0
0
1
.
.
.
1
1
1
0
0
0
0
.
.
.
1
1
1
1
0
1
0
.
.
.
1
0
1
0
1
1
0
.
.
.
0
1
1
1
2
3
4
.
.
.
61
62
63
AGC HIGH
GH7 GH6 GH5 GH4 GH3 GH2 GH1 THRESHOLD
0
0
0
0
.
.
.
1
1
1
FG2 FG1 FILTER GAIN
0
0
1
1
GH2
DB15
DB14
GH5
0
0
0
0
.
.
.
1
1
1
DEFAULT
REDUCED GAIN
FI1
C3 (0)
DB16
GH6
AUTO AGC
MANUAL AGC
FREEZE AGC
RESERVED
LG1 LNA MODE
0
1
C4 (1)
DB17
GH7
800µA (DEFAULT)
DB4
DB18
GM1
LNA BIAS
0
GL1
DB19
GM2
LI1
0
DB6
DB20
LG1
LI2
DB5
DB21
LG2
0
1
2
3
GL3
DB22
FG1
GH4
AGC LOW
GL7 GL6 GL5 GL4 GL3 GL2 GL1 THRESHOLD
AGC MODE
DEFAULT
HIGH
ADDRESS
BITS
AGC_LOW_THRESHOLD
GL2
DB23
FG2
ML1 MIXER LINEARITY
AGC_HIGH_THRESHOLD
DB7
DB24
FI1
AGC_
MODE
GL4
DB25
LG1
LNA_
CURRENT
LNA_
GAIN
DB8
FILTER_
CURRENT
DB26
LI1
FILTER_
GAIN
GL5
LNA_MODE
DB27
LI2
MIXER_
LINEARITY
DB28
ML1
REGISTER 9—AGC REGISTER
8
24
72
INVALID
0
0
0
0
.
.
.
0
0
0
0
0
0
0
.
.
.
0
0
1
0
0
0
0
.
.
.
1
1
0
0
0
0
1
.
.
.
1
1
0
0
1
1
0
.
.
.
1
1
0
1
0
1
0
.
.
.
0
1
0
1
2
3
4
.
.
.
78
79
80
LG2 LG1 LNA GAIN
0
1
0
1
3
10
30
INVALID
05876-039
0
0
1
1
Figure 71. Register 9—AGC Register Map
In receive mode, AGC is set to automatic AGC by default on
power-up. The default thresholds are AGC_LOW_THRESHOLD =
30 and AGC_HIGH_THRESHOLD = 70. See the RSSI/AGC
section for details. It is only necessary to program this register if
AGC settings, other than the defaults, are required.
AGC high and low settings must be more than 30 apart to
ensure correct operation.
An LNA gain of 30 is available only if LNA_MODE (DB25) is
set to 0.
Rev. B | Page 56 of 64
Data Sheet
ADF7021
DB6
DB5
M2
M1
KP1 KP
KI4
KI3
KI2
KI1
KI
AE1
AFC ENABLE
0
0
.
1
0
1
.
1
0
0
.
1
0
0
.
1
0
0
.
1
0
1
.
1
2^0
2^1
...
2^15
0
1
OFF
AFC ON
0
0
.
1
2^0
2^1
...
2^7
...
MAX AFC
MA3 MA2 MA1 RANGE
M12
...
M3
M2
M1
AFC SCALING
FACTOR
0
0
0
0
.
.
.
1
1
1
...
...
...
...
...
...
...
...
...
...
0
0
0
1
.
.
.
1
1
1
0
0
0
0
.
.
.
1
1
1
...
...
...
...
...
...
...
...
...
...
0
0
0
1
.
.
.
1
1
1
0
1
1
0
.
.
.
0
1
1
1
0
1
0
.
.
.
1
0
1
1
2
3
4
.
.
.
4093
4094
4095
1
0
1
0
.
.
.
1
0
1
1
2
3
4
.
.
.
253
254
255
05876-040
DB7
DB0
DB8
M4
M3
C1 (0)
DB9
M5
DB1
DB10
M6
DB2
DB11
M7
C2 (1)
DB12
M8
C3 (0)
DB13
M9
DB3
DB14
M10
C4 (1)
DB15
M11
DB4
DB16
M12
KP3 KP2
MA8
0
1
1
0
.
.
.
0
1
1
ADDRESS
BITS
AE1
DB17
KI1
DB18
DB20
KI4
DB19
DB21
AFC SCALING_FACTOR
KI2
DB22
KP2
KP1
KI3
DB23
DB26
MA3
KP3
DB27
MA4
DB24
DB28
MA5
DB25
DB29
MA6
KI
MA1
DB30
MA7
KP
MA2
DB31
MA8
MAX_AFC_RANGE
AFC_EN
REGISTER 10—AFC REGISTER
Figure 72. Register 10—AFC Register Map
To tradeoff between AFC settling time and AFC accuracy, the
KI and KP parameters can be adjusted from the recommended
settings (staying within the allowable range) such that
The AFC_SCALING_FACTOR can be expressed as
 2 24 × 500 

AFC _ SCALING _ FACTOR = Round 
 XTAL 
AFC Correction Range = MAX_AFC_RANGE × 500 Hz
The settings for KI and KP affect the AFC settling time and
AFC accuracy. The allowable range of each parameter is
KI > 6 and KP < 7
The recommended settings to give optimal AFC performance
are KI = 11 and KP = 4.
When the RF_DIVIDE_BY_2 (R1_DB18) is enabled, the
programmed AFC correction range is halved. The user
accounts for this halving by doubling the programmed
MAX_AFC_RANGE value.
Signals that are within the AFC pull-in range but outside the IF
filter bandwidth are attenuated by the IF filter. As a result, the
signal can be below the sensitivity point of the receiver and,
therefore, not detectable by the AFC.
Rev. B | Page 57 of 64
ADF7021
Data Sheet
DB0
C1 (1)
DB6
MT1
DB1
DB7
MT2
C2 (1)
DB8
SB1
DB2
DB9
SB2
DB3
DB10
SB3
C3 (0)
DB11
SB4
C4 (1)
DB12
SB5
DB4
DB13
SB6
CONTROL
BITS
PL1
DB14
SB7
PL2
DB15
SB8
DB5
MATCHING_
TOLERANCE
DB16
SB9
SB10 DB17
SB11 DB18
SB12 DB19
SB13 DB20
SB14 DB21
SB15 DB22
SB16 DB23
SB17 DB24
SB18 DB25
SB19 DB26
SB20 DB27
SB21 DB28
SB22 DB29
SB23 DB30
SYNC_BYTE_SEQUENCE
SB24 DB31
SYNC_BYTE_
LENGTH
REGISTER 11—SYNC WORD DETECT REGISTER
PL2
PL1
SYNC BYTE
LENGTH
0
0
1
1
0
1
0
1
12 BITS
16 BITS
20 BITS
24 BITS
0
0
1
1
0
1
0
1
ACCEPT 0 ERRORS
ACCEPT 1 ERROR
ACCEPT 2 ERRORS
ACCEPT 3 ERRORS
05876-041
MATCHING
MT2 MT1 TOLERANCE
Figure 73. Register 11—Sync Word Detect Register Map
SWD_MODE
LOCK_
THRESHOLD_
MODE
REGISTER 12—SWD/THRESHOLD SETUP REGISTER
DB15
DB14
DB13
DB12
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
DP7
DP6
DP5
DP4
DP3
DP2
DP1
IL2
IL1
LM2
LM1
C4 (1)
C3 (1)
C2 (0)
C1 (0)
CONTROL
BITS
DP8
DATA_PACKET_LENGTH
DATA PACKET LENGTH
0
1
...
255
INVALID
1 BYTE
...
255 BYTES
SWD MODE
0
1
2
3
SWD PIN LOW
SWD PIN HIGH AFTER NEXT SYNCWORD
SWD PIN HIGH AFTER NEXT SYNCWORD
FOR DATA PACKET LENGTH NUMBER OF BYTES
INTERRUPT PIN HIGH
3
THRESHOLD FREE RUNNING
LOCK THRESHOLD AFTER NEXT SYNCWORD
LOCK THRESHOLD AFTER NEXT SYNCWORD
FOR DATA PACKET LENGTH NUMBER OF BYTES
LOCK THRESHOLD
05876-042
LOCK THRESHOLD MODE
0
1
2
Figure 74. Register 12—SWD/Threshold Setup Register Map
Lock threshold locks the threshold of the envelope detector. This has the effect of locking the slicer in linear demodulation and locking
the AFC and AGC loops when using linear or correlator demodulation.
Rev. B | Page 58 of 64
Data Sheet
ADF7021
REGISTER 13—3FSK/4FSK DEMOD REGISTER
0
0
1
1
0
1
0
1
DB3
DB1
DB0
C2 (0)
C1 (1)
DB4
ST1
DB2
DB5
C3 (1)
DB6
ST3
ST2
C4 (1)
DB7
CONTROL
BITS
ST4
DB9
ST6
DB8
DB10
ST7
ST5
DB11
3FSK/4FSK_
SLICER_THRESHOLD
VD1
DB12
PC1
PHASE_
CORRECTION
3FSK_VITERBI_
DETECTOR
DB13
VM1
VT4
DB19
VT5
VM2 VM1
DB14
OFF
1
2
3
.
.
127
DB15
0
1
0
1
.
.
1
VT1
0
0
1
1
.
.
1
VM2
0
0
0
0
.
.
1
DB17
...
...
...
...
...
...
...
DB16
0
0
0
0
.
.
1
VT2
3FSK CDR
THRESHOLD
VT3
VT1
DB18
VT2
DB20
VT3
VT6
...
DB21
VT7
VT7
PTV1 DB22
3FSK_CDR_THRESHOLD
PTV2 DB23
PTV3 DB24
PTV4 DB25
3FSK_PREAMBLE_
TIME_VALIDATE
VITERBI_
PATH_
MEMORY
Refer to the Receiver Setup section for information about programming these settings.
3FSK VITERBI
VD1 DETECTOR
0
DISABLED
1
ENABLED
PHASE
PC1 CORRECTION
0
DISABLED
1
ENABLED
VITERBI PATH
MEMORY
4 BITS
6 BITS
8 BITS
32 BITS
ST7
...
ST3
ST2
ST1
SLICER
THRESHOLD
0
0
0
0
.
.
1
...
...
...
...
...
...
...
0
0
0
0
.
.
1
0
0
1
1
.
.
1
0
1
0
1
.
.
1
OFF
1
2
3
.
.
127
3FSK PREMABLE
PTV4 PTV3 PTV2 PTV1 TIME VALIDATE
0
0
0
0
.
.
1
0
0
1
1
.
.
1
0
1
0
1
.
.
1
0
1
2
3
.
.
15
05876-043
0
0
0
0
.
.
1
Figure 75. Register 13—3FSK/4FSK Demod Register Map
Rev. B | Page 59 of 64
ADF7021
Data Sheet
0
1
2
3
4
5
6
7
ED PEAK RESPONSE
TEST DAC GAIN
0
1
2
3
0
1
...
15
FULL RESPONSE TO PEAK
0.5 RESPONSE TO PEAK
0.25 RESPONSE TO PEAK
0.125 RESPONSE TO PEAK
TEST
TDAC EN
DB6
DB5
DB4
DB3
DB2
DB1
DB0
TO2
TE1
C4 (1)
C3 (1)
C2 (1)
C1 (0)
DB7
TO3
ADDRESS
BITS
TO1
DB9
DB8
TO6
TO4
DB10
TO7
TO5
DB12
DB11
TO8
DB13
TO9
TO11 DB15
TO10 DB14
TO12 DB16
TO14 DB18
DB21
TG1
TO13 DB17
DB22
TG2
TO15 DB19
DB23
TEST DAC OFFSET
TO16 DB20
DB24
DB25
ER1
LEAKAGE =
2^–8
2^–9
2^–10
2^–11
2^–12
2^–13
2^–14
2^–15
TG3
DB26
ED LEAK FACTOR
TEST_DAC_GAIN
TG4
PULSE_
EXTENSION
DB27
EF1
ER2
DB29
EF3
DB28
DB30
PE1
EF2
DB31
PE2
ED_LEAK_
FACTOR
ED_PEAK_
RESPONSE
REGISTER 14—TEST DAC REGISTER
NO GAIN
× 2^1
...
× 2^15
PULSE EXTENSION
NO PULSE EXTENSION
EXTENDED BY 1
EXTENDED BY 2
EXTENDED BY 3
05876-044
0
1
2
3
Figure 76. Register 14—Test DAC Register Map
The demodulator tuning parameters, PULSE_EXTENSION,
ED_LEAK_FACTOR, and ED_PEAK_RESPONSE, can only be
enabled by setting R15_DB[4:7] to 0x9.
Using the Test DAC to Implement Analog FM DEMOD
and Measuring SNR
The test DAC allows the post demodulator filter out for both
linear and correlator demodulators to be viewed externally. The
test DAC also takes the 16-bit filter output and converts it to a
high frequency, single-bit output using a second-order, error
feedback Σ-Δ converter. The output can be viewed on the SWD
pin. This signal, when filtered appropriately, can then be used to
do the following:
•
Monitor the signals at the FSK post demodulator filter
output. This allows the demodulator output SNR to be
measured. Eye diagrams of the received bit stream can also
be constructed to measure the received signal quality.
•
Provide analog FM demodulation.
While the correlators and filters are clocked by DEMOD CLK,
CDR CLK clocks the test DAC. Note that although the test
DAC functions in regular user mode, the best performance is
achieved when the CDR_CLK is increased to or above the
frequency of DEMOD CLK. The CDR block does not function
when this condition exists.
Programming Register 14 enables the test DAC. Both the
linear and correlator/demodulator outputs can be multiplexed
into the DAC.
Register 14 allows a fixed offset term to be removed from the
signal (to remove the IF component in the ddt case). It also has
a signal gain term to allow the usage of the maximum dynamic
range of the DAC.
Rev. B | Page 60 of 64
Data Sheet
ADF7021
DB0
C1 (1)
DB5
RT2
DB1
DB6
RT3
DB2
DB7
RT4
C2 (1)
DB8
TM1
C3 (1)
DB9
TM2
DB3
DB10
ADDRESS
BITS
DB4
DB11
Rx_TEST_
MODES
SD1
Tx_TEST_
MODES
TM3
Σ-Δ_TEST_
MODES
RT1
DB12
CLK_-MUX
C4 (1)
DB13
SD2
DB19
SD3
DB20
PM1
CM3
DB14
DB21
PM2
PC1
DB22
PM3
DB15 PFD/CP_TEST_
MODES
DB23
PM4
PC2
DB24
AM1
DB16
DB25
AM2
DB17
DB26
AM3
PC3
DB27
AM4
DB18
DB28
FH1
PLL_TEST_
MODES
CM1
DB29
RD1
ANALOG_TEST_
MODES
CM2
REG 1_PD
FORCE_LD
HIGH
DB30
CO1
CAL_
OVERRIDE
DB31
CO2
REGISTER 15—TEST MODE REGISTER
CAL OVERRIDE
AUTO CAL
OVERRIDE GAIN
OVERRIDE BW
OVERRIDE BW AND GAIN
PFD/CP TEST MODES
0
1
2
3
4
5
6
7
REG1 PD
0
1
NORMAL
PWR DWN
DEFAULT, NO BLEED
(+VE) CONSTANT BLEED
(–VE) CONSTANT BLEED
(–VE) PULSED BLEED
(–VE) PULSE BLD, DELAY UP?
CP PUMP UP
CP TRI-STATE
CP PUMP DN
FORCE LD HIGH
0
1
Σ-Δ TEST MODES
NORMAL
FORCE
0
1
2
3
4
5
6
7
ANALOG TEST MODES
0
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
BAND GAP VOLTGE
40µA CURRENT FROM REG4
FILTER I CHANNEL: STAGE 1
FILTER I CHANNEL: STAGE 2
FILTER I CHANNEL: STAGE 1
FILTER Q CHANNEL: STAGE 1
FILTER Q CHANNEL: STAGE 2
FILTER Q CHANNEL: STAGE 1
ADC REFERENCE VOLTAGE
BIAS CURRENT FROM RSSI 5µA
FILTER COARSE CAL OSCILLATOR O/P
ANALOG RSSI I CHANNEL
OSET LOOP +VE FBACK V (I CH)
SUMMED O/P OF RSSI RECTIFIER+
SUMMED O/P OF RSSI RECTIFIER–
BIAS CURRENT FROM BB FILTER
DEFAULT, 3RD ORDER SD, NO DITHER
1ST ORDER SD
2ND ORDER SD
DITHER TO FIRST STAGE
DITHER TO SECOND STAGE
DITHER TO THIRD STAGE
DITHER × 8
DITHER × 32
Tx TEST MODES
0
1
2
3
4
5
6
NORMAL OPERATION
Tx CARRIER ONLY
Tx +VE TONE ONLY
Tx –VE TONE ONLY
Tx "1010" PATTERN
Tx PN9 DATA, AT PROGRAMED RATE
Tx SYNC BYTE REPEATEDLY
Rx TEST MODES
0
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
PLL TEST MODES
0
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
NORMAL OPERATION
R DIV
N DIV
RCNTR/2 ON MUXOUT
NCNTR/2 ON MUXOUT
ACNTR TO MUXOUT
PFD PUMP UP TO MUXOUT
PFD PUMP DN TO MUXOUT
SDATA TO MUXOUT (OR SREAD?)
ANALOG LOCK DETECT ON MUXOUT
END OF COARSE CAL ON MUXOUT
END OF FINE CAL ON MUXOUT
FORCE NEW PRESCALER CONFIG.
FOR ALL N
TEST MUX SELECTS DATA
LOCK DETECT PERCISION
RESERVED
NORMAL
SCLK, SDATA -> I, Q
REVERSE I,Q
I,Q TO TxRxCLK, TxRxDATA
3FSK SLICER ON TxRxDATA
CORRELATOR SLICER ON TxRxDATA
LINEAR SLICER ON RXDATA
SDATA TO CDR
ADDITIONAL FILTERING ON I, Q
ENABLE REG 14 DEMOD PARAMETERS
POWER DOWN DDT AND ED IN T/4 MODE
ENVELOPE DETECTOR WATCHDOG DISABLED
RESERVED
PROHIBIT CALACTIVE
FORCE CALACTIVE
ENABLE DEMOD DURING CAL
CLK MUXES on CLKOUT pin
0
1
2
3
4
5
6
7
NORMAL, NO OUTPUT
DEMOD CLK
CDR CLK
SEQ CLK
BB OFFSET CLK
SIGMA DELTA CLK
ADC CLK
TxRxCLK
Figure 77. Register 15—Test Mode Register Map
Rev. B | Page 61 of 64
05876-045
0
1
2
3
ADF7021
Data Sheet
OUTLINE DIMENSIONS
0.30
0.23
0.18
PIN 1
INDICATOR
48
37
36
1
0.50
BSC
TOP VIEW
0.80
0.75
0.70
0.45
0.40
0.35
4.25
4.10 SQ
3.95
EXPOSED
PAD
12
25
24
13
BOTTOM VIEW
0.05 MAX
0.02 NOM
COPLANARITY
0.08
0.20 REF
SEATING
PLANE
PIN 1
INDICATOR
0.20 MIN
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
COMPLIANT TO JEDEC STANDARDS MO-220-WKKD.
08-16-2010-B
7.00
BSC SQ
Figure 78. 48-Lead Lead Frame Chip Scale Package [LFCSP_WQ]
7 mm × 7 mm Body, Very Very Thin Quad
(CP-48-5)
Dimensions shown in millimeters
ORDERING GUIDE
Model1
ADF7021BCPZ
ADF7021BCPZ-RL
ADF7021BCPZ-RL7
EVAL-ADF70XXMBZ
EVAL-ADF70XXMBZ2
EVAL-ADF7021DBJZ
EVAL-ADF7021DBZ2
EVAL-ADF7021DBZ3
EVAL-ADF7021DBZ5
EVAL-ADF7021DBZ6
EVAL-ADF7021DB9Z
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
48-Lead Lead Frame Chip Scale Package [LFCSP_WQ]
48-Lead Lead Frame Chip Scale Package [LFCSP_WQ]
48-Lead Lead Frame Chip Scale Package [LFCSP_WQ]
Control Mother Board
Evaluation Platform
426 MHz to 429 MHz Daughter Board
860 MHz to 870 MHz Daughter Board
431 MHz to 470 MHz Daughter Board
80 MHz to 650 MHz Daughter Board
608 MHz to 614 MHz Daughter Board
169 MHz Daughter Board
Z = RoHS Compliant Part.
Rev. B | Page 62 of 64
Package Option
CP-48-5
CP-48-5
CP-48-5
Data Sheet
ADF7021
NOTES
Rev. B | Page 63 of 64
ADF7021
Data Sheet
NOTES
©2007–2013 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D05876-0-4/13(B)
Rev. B | Page 64 of 64
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