Intersil ISL6730BEVAL1Z Power factor correction controller Datasheet

Power Factor Correction Controllers
ISL6730A, ISL6730B, ISL6730C, ISL6730D
The ISL6730A, ISL6730B, ISL6730C, ISL6730D are active
Features
Power Factor Correction (PFC) controller ICs that use a boost
topology. The controllers are suitable for AC/DC power
systems, up to 2kW and over the universal line input.
The ISL6730A, ISL6730B, ISL6730C, ISL6730D operate in
Continuous Conduction Mode (CCM). Accurate input current
shaping is achieved with a current error amplifier. A patent
pending breakthrough negative capacitance technology
minimizes zero crossing distortion and reduces the magnetic
components size. The small external components result in a
low cost design without sacrificing performance.
The internally clamped 12.5V gate driver delivers 1.5A peak
current to the external power MOSFET. The ISL6730A,
ISL6730B, ISL6730C, ISL6730D provide a highly reliable
system that is fully protected. Protection features include
cycle-by-cycle overcurrent, over power limit, over-temperature,
input brownout, output overvoltage and undervoltage
protection.
The ISL6730A, ISL6730B provide excellent power efficiency
and transitions into a power saving skip mode during light load
conditions, thus improving efficiency automatically. The
ISL6730A, ISL6730B, ISL6730C, ISL6730D can be shut down
by pulling the FB pin below 0.5V or grounding the BO pin. The
ISL6730C, ISL6730D have no skip mode.
Two switching frequency options are provided. The ISL6730B,
ISL6730D switch at 62kHz, and the ISL6730A, ISL6730C
switch at 124kHz.
• Reduce component size requirements
- Enables smaller, thinner AC/DC adapters
- Choke and cap size can be reduced
- Lower cost of materials
• Excellent power factor over line and load regulation
- Internal current compensation
- CCM Mode with Patent pending IP for smaller EMI filter
• Better light load efficiency
- Automatic pulse skipping
- Programmable or automatic shutdown
• High reliable design
- Cycle-by-cycle current limit
- Input average power limit
- OVP and OTP protection
- Input brownout protection
• Small 10 Ld MSOP package
Applications
• Desktop computer AC/DC adaptor
• Laptop computer AC/DC adaptor
• TV AC/DC power supply
• AC/DC brick converters
100
VI
+
VLINE
VOUT
95
EFFICIENCY (%)
90
VCC
ISEN
GATE
ICOMP
GND
ISL6730
VIN
FB
ISL6730A, SKIP
80
ISL6730C
75
70
COMP
BO
85
65
VREG
60
0
20
FIGURE 1. TYPICAL APPLICATION
40
60
OUTPUT POWER (W)
80
100
FIGURE 2. PFC EFFICIENCY
TABLE 1. KEY DIFFERENCES IN FAMILY OF ISL6730
August 8, 2013
FN8258.1
VERSION
ISL6730A
ISL6730B
ISL6730C
ISL6730D
Switching Frequency
124kHz
62kHz
124kHz
62kHz
Skip Mode
Yes-Fixed
Yes-Fixed
No
No
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas LLC 2013. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
ISL6730A, ISL6730B, ISL6730C, ISL6730D
Pin Configuration
ISL6730A, ISL6730B, ISL6730C, ISL6730D
(10 LD MSOP)
TOP VIEW
GND
1
10 GATE
ISEN
2
9 VCC
ICOMP
3
8 VREG
VIN
4
7 FB
BO
5
6 COMP
Pin Descriptions
PIN # I/O SYMBOL
DESCRIPTION
1
-
GND
Ground pin. All voltage levels refer to this pin.
2
I
ISEN
Current sense pin. The current through this pin is proportional to the inductor current.
3
I/O
ICOMP
4
I
VIN
Input voltage sense. This pin provides the reference voltage to shape inductor current. Connect this pin to a resistor divider from
the rectified input voltage. The resistor divider ratio is used to adjust the phase lag between input voltage and the input current.
The phase lag is required to compensate the phase lead generated by the EMI filter.
5
I/O
BO
This pin should be decoupled to GND with a minimum 0.1µF ceramic capacitor. The BO pin is a voltage follower, which will follow
the DC voltage of the VIN pin. The BO pin is internally tied to GND through a resistor RIS. The decoupling capacitor provides ripple
filtering. When the voltage at the BO pin (VBO) drops below brownout voltage threshold, the controller enters shutdown mode
and the gate drive is disabled. The BO pin will be disabled when the FB pin drops below the enabling threshold.
6
I/O
COMP
Output of the error amplifier. The voltage of the COMP pin sets the input power. During start-up, a small charge current will slowly
ramp up the voltage of the COMP pin.
7
I
FB
Voltage feed back pin. Connect this pin to a resistor divider from the output. The resistor divider sets the output voltage. When
the FB pin voltage exceeds 104% of the reference voltage, overvoltage-protection is triggered and gate drive is disabled. When
the FB pin is below 10%, the device is put into shutdown mode. There is an internal pull-down current source for open loop
protection.
8
-
VREG
Output of internal regulator. The voltage having a ±2% tolerance over line, load and operating temperature. Bypass to GND with
a 47nF low ESR capacitor. VREG can source up to 10mA. This pin is not recommended for usage other than bypass.
9
I
VCC
Power supply pin. The VCC pin should be decoupled to GND with a minimum 0.1µF ceramic capacitor.
10
O
GATE
Push-pull gate drive for the external MOSFET. Output voltage is clamped at 12.5V. This pin provides typically 2A sink and 1.5A
source capability.
Current error amplifier output pin.
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
PART
MARKING
TEMP.
RANGE (°C)
PACKAGE
(Pb-Free)
PKG.
DWG. #
ISL6730AFUZ
6730A
-40 to +125
10 Ld MSOP
M10.118
ISL6730BFUZ
6730B
-40 to +125
10 Ld MSOP
M10.118
ISL6730CFUZ
6730C
-40 to +125
10 Ld MSOP
M10.118
ISL6730DFUZ
6730D
-40 to +125
10 Ld MSOP
M10.118
ISL6730AEVAL1Z
Evaluation Board
ISL6730BEVAL1Z
Evaluation Board
NOTES:
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page. For more information on MSL please see techbrief TB363.
2
FN8258.1
August 8, 2013
Block Diagram
VI
EMI CHOKE
D
VOUT
L
VLINE
CF3
CF2
COUT
Q1
CF1
Lm
RCS
3
CREG
VCC
VREG
LINEAR
REGULATOR
CURRENT
MIRROR
VCC
UVLO
GATE
2:1
I
OC
I
> -------------CS
2
RSEN
ISEN
CONTROL
LOGIC
I CS
GND
OTP
PWM
COMP
SKIP
CEQ GEN.
ICOMP
V
IREF
Gmi
CS
R ×I
IS
ISEN
= ----------------------------------2
RIS
OSCILLATOR
0.25 × VIN
-----------------------------COMPB
2
BO
SOFT-START
ENABLE
2.5V
RIN2
VIN
COMPB
SKIP
CLAMP
COMP-1V
FB
RFB2
OVER POWER
LIMIT
Gmv
20µA
RIN1
IFB
SKIP
COMP
BO
CBO
RFB1
ISL6730A, ISL6730B, ISL6730C, ISL6730D
DF1
DF2
FN8258.1
August 8, 2013
Application Schematics
Typical 300W Application Schematic
D1
1N5406
L1
0u
+
R4
51k
R28 0.22
R27 0.22
R5 0.22
C6
2.2n
C11
R11
470k
C7
1u
R9
3k
ISEN
330p
C10
1nF
ICOMP
3
TP6
2
TP5
0.1
DZ1
3.3V
3.3M
1u
VCC
R10
GATE
ICOMP
GND
10
1
ISEN
DNP
R20
10k
3
R21
25k
DNP
1
DNP P8
DNP P9
VCC
C17
1n
DNP
VCC
3.3M
C OM P
VIN
6
TP4
2N7002
Q2
DNP
GND
GATE
DNP
TP2
R18
62k
C15
C18
150n
2.2u
7
ISL6730B/D
COMP
BO
C14
470n
FB
P6
TP7
R26
49.9
5
220p
GND
R6
TP8
U1
C13
2
R13
7.15k
C26
DNP
C9
VREG
BO
TP3
C12
DNP
0.1
P3
VIN
4
C19
2
C21
9
DNP
R8
470k
47n
C20
R14
8.2k
C1 390V
270u
450V
TP10
C8
220n
D8
S1M
3
R2
2.2
680n
D7
S1M
1
2
C4
DNP
C3
470n
C22
8
PE
P2
VC C
4
DB1
GBU808
AC2
C5
2.2n
P5
R3
2M
TP9
SPP20N60C3
Q1
TP12
GATE1
1
R EG
P4
R1
2M
5mH
L3
DC+
C3D04060
-
100n
C2
1
UNIVERSAL INPUT
85~265Vac
2
4
F2 8A
3
AC1
L2
1.5mH
1
TP1
C16
1n
FB
R17
0
R19
42.2k
P7
ISL6730A, ISL6730B, ISL6730C, ISL6730D
P1
VOUT
D2 2
FN8258.1
August 8, 2013
Application Schematics
(Continued)
Typical 85W Application Schematic
F1 3.15A
D3
TP9
1
2
C1
56u
450V
D6
C6
470p
R4
51k
0.22
R5
C8
DZ1
3.3V
220n
D8
S1M
P3
TP10
S3KB-TP
S3KB-TP
3.3M
R6
R9
2.1k
DNP
C11
R8
470k
470p
C10
6.8n
R11
470k
3
TP6
ISEN
2
TP5
1u
C24
47n
U1
VCC
VCC
GATE
ICOMP
GND
10
1
R10
3.3M
VIN
DNP
R20
10k
3
R21
25k
DNP
1
DNP P8
DNP P9
VCC
5
220p
6
BO
C13
2N7002
Q2
DNP
C17
1n
DNP
2
R13
7.15k
GND
GATE
ISEN
COMP
4
FB
TP4
TP2
VIN
COMP
C14
470n
R18
68k
C15
C18
100n
2.2u
7
ISL6730A/C
P6
TP7
VIN
TP3
GND
C9
R14
5.36k
ICOMP
C12
DNP
C26
DNP
DNP
9
D7
S1M
3
D5
390V
2
R2
2.2
R3
2M
VCC
P5
IPP60R600C6
Q1
1
C4
DNP
C3
330n
8
4
C5
470p
DC+
P2
AC2
PE
L2 2.2m
REG
P4
VOUT
1
D2
C3D04060E
7.5mH
L3
1
UNIVERSAL INPUT
85~265Vac
100n
C2
L1 0uH
D4
R1
2M
3
5
2
AC1
3
TP1
C16
1n
FB
R17
1.5k
R19
40.2k
P7
ISL6730A, ISL6730B, ISL6730C, ISL6730D
P1
2
S3KB-TP
D1
S3KB-TP
S3KB-TP
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
Absolute Maximum Ratings
Thermal Information
VCC to GND. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +28V
GATE to GND. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +18V
VIN, BO, ISEN, FB and COMP to GND. . . . . . . . . . . . . . . . . . . . . . . .-0.3V to +6.3V
ESD Rating
Human Body Model (Tested per JESD22-A114) . . . . . . . . . . . . . . .2.5kV
Machine Model (Tested per JESD22-A115). . . . . . . . . . . . . . . . . . . . 200V
Charged Device Model (Tested per JESD-C101E. . . . . . . . . . . . . . . . . 1kV
Latch Up (Tested per JESD-78B; Class 2, Level A) . . . . . . . . . . . . . . 100mA
Thermal Resistance (Typical)
θJA (°C/W) θJC (°C/W)
MSOP Package (Notes 4, 5) . . . . . . . . . . . .
136.9
39.4
Maximum Junction Temperature (Plastic Package) . . . . . . . . . . . .+150°C
Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C
Junction Temperature Range . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
VCC to GND. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15V to + 20V
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
5. For θJC, the "case temp" location is taken at the package top center.
Electrical Specifications
-40°C to +125°C.
PARAMETER
Operating Conditions: VCC = 15V, TA = +25°C. Boldface limits apply over the operating temperature range,
SYMBOL
TEST CONDITIONS
MIN
(Note 8)
TYP
MAX
(Note 8)
UNITS
VCC SUPPLY CURRENT
Start Up Current
ISTART
VFB = 1V, VCC < VCC(ON)
73
106
139
µA
Standby Current
ISTDN
VFB = GND, VCC > VCC(ON)
179
237
295
µA
VFB = 2.5V, COMP = SKIP*0.25 +1V
580
690
800
µA
GATE is floating
3.0
3.7
4.2
mA
Skip Mode Current
ICCSKIP
Operating Current (Note 6)
ICC
VCC UVLO
UVLO Rising Threshold
VCC(ON)
9
10
11
V
UVLO Falling Threshold
VCC(OFF)
6.7
7.5
8.3
V
UVLO Threshold Hysteresis
VCC(HYS)
-
2.5
-
V
5.1
5.4
5.6
V
30
50
70
mA
fSW = 124kHz for ISL6730A/C and
fSW = 62kHz for ISL6730B/D
94.8
96.5
-
%
Free Running Frequency, ISL6730A/C
TA = -40°C to +125°C, VIN = 0.6V
98
107
116
kHz
Free Running Frequency, ISL6730A/C
TA = -40°C to +125°C, VIN = 2.5V
114
125
136
kHz
Free Running Frequency, ISL6730B/D
TA = -40°C to +125°C, VIN = 0.6V
47
54
61
kHz
Free Running Frequency, ISL6730B/D
TA = -40°C to +125°C, VIN = 2.5V
57
64
71
kHz
1.33
1.46
1.59
V
-
2.33
4.46
Ω
0.15
0.3
0.45
V
REGULATOR VOLTAGE VREG
Overall Accuracy
VREG
IREG = 0 to -10mA, VCC = 15V, Load Capacitor = 47nF
Current Limit
PWM CONVERTERS
Maximum Duty Cycle
OSCILLATOR
PWM Ramp Amplitude
Vm
GATE DRIVER
Gate Drive Pull-Down Resistance
VCC = 15V, IGATE = 15mA
Gate Drive Pull-Up Voltage Drop
VCC = 9V, IGATE = 15mA
6
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
Electrical Specifications
-40°C to +125°C. (Continued)
PARAMETER
Operating Conditions: VCC = 15V, TA = +25°C. Boldface limits apply over the operating temperature range,
SYMBOL
TEST CONDITIONS
Gate Drive Max. Sourcing/Sinking
Current
MIN
(Note 8)
TYP
MAX
(Note 8)
UNITS
-
1.5
-
A
Rise Time
CO = 2.2nF, VCC = 15V, Gate Voltage Rise Time from 10%
to 90% of VGC
-
34
62
ns
Fall Time
CO = 2.2nF, VCC = 15V, Gate Voltage Fall Time from 10%
to 90% of VGC
-
34
57
ns
VGC
10.5
12
13.5
V
VREF
2.48
2.5
2.52
V
IFB
-
65
-
nA
Rising Threshold to Enable Converter
FB_EN
280
300
320
mV
Falling Threshold to Disable Converter
FB_DIS
190
202
214
mV
Enable Hysteresis
FB_Hys
-
100
-
mV
Gmv
50
77
104
µA/V
-
13
-
µA
Gate Clamp Voltage
VOLTAGE REFERENCE
Reference Voltage
Feedback Pin Pull-Down Current
VOLTAGE ERROR AMPLIFIER
Error Amp Transconductance
ISource/Sink
COMP Offset Voltage
VCOMP_OFF
0.95
1.01
1.07
V
COMP Soft-Start Enable Voltage
VCOMP_EN
0.58
0.64
0.75
V
-
9
-
nA
0.196
0.25
0.296
V/V
INPUT VOLTAGE SENSING
VIN Leakage Current
MULTIPLIER GAIN
GMUL
COMP = 2.5V, VIN = 1.0V, BO = 1.0V, ISEN = 50µA
CURRENT ERROR AMPLIFIER
Current DC Gain
AiDC
ΔIICOMP/ΔIISEN
1.6
1.9
2.2
A/A
Error Amp Transconductance
Gmi
IICOMP = ±20µA
205
268
331
µA/V
ICOMP Source/Sink Current (Note 7)
-
60
-
µA
Current Sensing Input Offset
-3
2
7
mV
1.32
1.36
1.4
V
LIGHT LOAD EFFICIENCY ENHANCEMENT AND OVERPOWER PROTECTION
Skip Mode COMP Threshold
VSCMT
Applied for ISL6730A/B
COMP Upper Limit
VCUL
3.53
3.85
4.17
V
COMP Valid Range
VCUL-1V
2.5
2.83
3.16
V
FB Exit Threshold Voltage
VFB_EXIT
Fraction of the set point (VREF), IISEN = 0µA, Applied for
ISL6730A/B
87
88
89
%
ISEN Exit Threshold Current
ISEN_EXIT
VFB = 2.5V, Applied for ISL6730A/B
-38
-29
-20
µA
BROWNOUT DETECTION
Brownout Rising Threshold
VBO_R
478
494
510
mV
Brownout Falling Threshold
VBO_F
387
401
415
mV
102.9
104.1
105.3
%
OVERVOLTAGE PROTECTION
Overvoltage Protection
VOVP
7
Fraction of the set point (VREF); ~1µs noise filter
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
Electrical Specifications
-40°C to +125°C. (Continued)
Operating Conditions: VCC = 15V, TA = +25°C. Boldface limits apply over the operating temperature range,
MIN
(Note 8)
TYP
MAX
(Note 8)
UNITS
-197
-177
-159
µA
Shutdown Temperature (Note 7)
-
160
-
°C
Thermal Shutdown Hysteresis (Note 7)
-
25
-
°C
PARAMETER
SYMBOL
TEST CONDITIONS
OVERCURRENT PROTECTION
Overcurrent Threshold
IOC
THERMAL SHUTDOWN
NOTES:
6. This is the VCC current consumed when the device is active but not switching. Does not include gate drive current.
7. Limits should be considered typical and are not production tested.
8. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified.
Typical Performance Curves
101.0
FSW NORMALIZED (%)
VFB NORMALIZED (%)
100.50
100.25
100.00
99.75
100.5
VIN = 2.5V
100.0
99.5
VIN = 0.6V
99.50
-40
-20
0
20
40
60
80
TEMPERATURE (°C)
100
120
99.0
-40
140
FIGURE 3. FEEDBACK ACCURACY
-20
0
20
40
60
80
TEMPERATURE (°C)
100
120
140
FIGURE 4. FSW vs TEMPERATURE, VCC = 15V
105
101
FSW NORMALIZED (%)
AIDC NORMALIZED (%)
100
100
99
98
95
90
85
80
97
75
-40
-20
0
20
40
60
80
TEMPERATURE (°C)
FIGURE 5. AIDC vs TEMPERATURE
8
100
120
140
0
0.5
1.0
1.5
2.0
2.5
3.0
VIN (V)
FIGURE 6. FSW vs VIN, TA = +25°C
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
Typical Performance Curves
(Continued)
VCC CURRENT NORMALIZED (%)
101
HYSTERSIS
100
UP
THRESHOLD
99
DOWN
THRESHOLD
98
-40
-20
0
20
40
60
80
100
120
102
101
100
ICC
(GATE FLOATING)
99
98
-40
140
ISTART
-20
0
20
40
60
80
100
120
140
TEMPERATURE (°C)
TEMPERATURE (°C)
FIGURE 8. VCC SUPPLY CURRENT vs TEMPERATURE
FIGURE 7. UVLO THRESHOLDS vs TEMPERATURE
112
DRIVER TIME NORMALIZED (%)
UVLO THRESHOLD NORMALIZED (%)
102
110
108
FALL TIME
106
104
102
RISE TIME
100
98
96
-40
-20
0
20
40
60
80
100
120
140
TEMPERATURE (°C)
FIGURE 9. GATE DRIVER ABILITY vs TEMPERATURE (LOAD = 2.2nF)
9
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
Functional Description
EMI CHOKE
VCC Undervoltage Lockout (UVLO)
VLINE
The ISL6730A, ISL6730B, ISL6730C, ISL6730D start
automatically once the voltage at VCC exceeds the UVLO
threshold.
CF3
CF2
Lm
DF1
Shutdown
DF2
When the VFB pin is below 0.2V, the controller is disabled and
the PWM output driver is tri-stated. When disabled, the IC power
will be reduced. During shutdown, the COMP pin is discharged to
GND and the controller is disabled. The Over-Temperature
Protection (OTP) is still alive to prevent the controller from
starting up in a high temperature ambient condition.
RIN2
VIN
BO
RIN1
CBO
In the event that the FB pin is disconnected from the feedback
resistors, the FB pin is pulled to ground by an internal current
source IFB. When the FB pin voltage drops below 0.2V, the gate
driver is disabled. The ISL6730A, ISL6730B, ISL6730C,
ISL6730D enters shutdown mode.
The BO pin also utilizes the VIN resistor divider for voltage
sensing. Set the resistor divider ratio to satisfy the brownout
requirement.
Soft-Start
First, calculate the resistor divider ratio, KBO.
The COMP pin is released once the soft-start operation begins. A
13µA current sources out to the RC network connected from the
COMP pin until the FB pin voltage reaches 90% of the reference
voltage.
V BORMAX
K BO = ------------------------------------------V RMSmin – 2V F
Switching is inhibited when the COMP pin voltage is below 1V.
When the COMP pin reaches 1V, the current error amplifier and
the gate driver are activated and the converter starts switching.
During UVLO, Brownout and Shutdown, the COMP is pulled to the
ground.
Input Voltage Sensing
The VIN pin is needed to sense the rectified input voltage. The
sensed semi-sinusoidal waveform is needed to shape inductor
current, which helps achieves unity power factor. At the same
time, the voltage on the VIN pin is used to generate the negative
capacitive element at the input. This will cancel the input filter
capacitor, CF. Canceling the effect of CF will increase the
displacement power factor and alleviate the zero crossing
distortion, which is related to the distortion power factor.
FIGURE 10. INPUT VOLTAGE SENSING SCHEMATIC
(EQ. 1)
Where VF is the forward voltage drop of the bridge rectifier and
the voltage drop of DF1; DF2.
Then, select the RIN2 based on the highest reasonable resistance
value. Then select the RIN1 based upon the desirable minimum
RMS value of the line voltage for the PFC operation.
K BO
R IN1 = --------------------- ⋅ R IN2
1 – K BO
(EQ. 2)
Inductor Current Sensing
The current sensing of the converter has two purposes. One is to
force the inductor current to track the input semi-sinusoidal
waveform. The other purpose is for overcurrent protection. Refer to
Figure 11 for the current sensing scheme. The sensed current ICS
is in proportion to the inductor current, IL as described in
Equation 3.
1 R CS
I CS = --- ⋅ ---------------- ⋅ I L
2 R SEN
(EQ. 3)
where:
RCS is the current sensing resistor with low value in the return
path to the bridge rectifier.
RSEN is the current scaling resistor connected between ISEN to
the RCS.
10
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
VI
EMI CHOKE
VOUT
BRIDGE RECTIFIER
LF
L
Q1
COUT
VLINE
CF3
CF1
CF2
CF1
Lm
RCS
FIGURE 12. TYPICAL PFC INPUT FILTER CIRCUIT
CURRENT
MIRROR
2:1
RSEN
BRIDGE RECTIFIER
EMI CHOKE
I
LF
CS
VLINE
CF3
ISEN
CF1
CF2
Lm
I
CS
> 0.5 I
OC
FIGURE 11. INDUCTOR CURRENT SENSING SCHEME
A high value RCS renders more accurate current sensing. It is
recommended to use the RCS to render 120mV peak voltage at
the maximum line voltage during full load condition.
120mV ⋅ V RMSMAX ⋅ η
R CS > ------------------------------------------------------------2 ⋅ P Omax
FIGURE 13. LOW COST PFC INPUT FILTER CIRCUIT
For applications where the output power is above 500W, the
negative capacitance helps to improve the power factor
dramatically. Please refer to Table 2 for the recommended
filtering capacitor to be placed after the bridge rectifier, CF1.
TABLE 2.
(EQ. 4)
Where η is the efficiency of the converter at the maximum line
input with full load.
Since the RCS sees the average input current, high value RCS
generates high power dissipation on the RCS. Use a reasonable
RCS according to the resistor power rating. The worst-case power
dissipation occurs at the input low line when input current is at
its maximum. Power dissipation by the resistor is:
(EQ. 5)
P RCS = ( I RMSMAX ) 2 ⋅ R CS
where:
IRMSMAX is the maximum input RMS current at the minimum
input line voltage, VRMSmin.
Select the RSEN according to the peak current limit requirement.
The resistor is sized for an overload current 25% more than the
peak inductor peak current.
Negative Input Capacitor Generation (Patent
Pending)
The patent pending negative capacitor generation capability of
the ISL6730A, ISL6730B, ISL6730C, ISL6730D allows the
capacitor CF2 to be moved from before the bridge rectifier
(Figure 12) to after the bridge rectifier (Figure 13). Thus, a
smaller lower cost CF2 can be used. The change in topology
reduces the size of the EMI filter. Furthermore, CF1 can be
increased thus decreasing the size of LF (Figure 13).
CF1
Po < 100W
Typical
C(µF)/100W
0.68
100W < Po < 500W Po > 500W
0.33
0.22
Additional CF1 may be used to accommodate the use of small
boost inductor or to eliminate the differential mode filter inductor
as long as the equipment meets the power factor or goal.
The equivalent negative capacitor is a function of the input
voltage divider ratio, KBO, the current sensing gain and current
compensation error integration gain.
Adjusting the negative Ceq can be achieved by adjusting the
current compensation network.
Frequency Modulation
The ISL6730A, ISL6730B, ISL6730C, ISL6730D can further
reduce EMI filter size by lowering the differential noise power
density. The reduction is achieved by switching frequency
modulation.
The frequency varies with the VIN pin. The switching frequency
reaches the peak value when the VIN pin voltage is 2V as shown
in Figure 6. The peak value of ISL6730A/C is 124kHz, and the
ISL6730B/D is 62kHz.
Output Voltage Regulation
The output voltage is sensed through a resistor divider. The
middle point of the resistor divider is fed to the FB pin. The
resistor divider ratio sets the output voltage. The
transconductance error amplifier generates a current in
proportion to the difference between the FB pin and the 2.5V
internal reference. The PFC is stabilized by the compensation
network that is connected from the COMP pin to the ground.
The voltage of the COMP sets the input average power by
determining the amplitude of the current reference. To keep the
11
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
harmonic distortion minimum, it is desirable to set the control
bandwidth much lower than twice of the line frequency. The
recommended voltage loop bandwidth is 10Hz.
During start-up, the compensation capacitors and the charging
current from the error amplifier sets the input power increase
rate. Thus, soft-start is achieved.
The COMP is discharged during shutdown and fault conditions.
Light Load Efficiency Enhancement
For PC, adaptor and TV applications, it is desirable to achieve
high efficiency at light load conditions and low standby current.
The ISL6730A, ISL6730B can enter light load efficiency mode
automatically.
The voltage error amplifier output, COMP, is an indicator of the
average input power level. The controller compares the V(COMP)
and V(SKIP). If V(COMP)-1V is less than V(SKIP)*0.25, the PFC
controller stops gate switching and the COMP pin voltage is
clamped to V(SKIP)+0.6V. ISL6730A/B use a fixed V(SKIP), which
is 1.4V; for ISL6730C/D, the SKIP function are disabled.
The controller exits skip mode when VFB drops to 88% (typical) of
the reference voltage or when the sensed returned current
exceeds 29µA.
Protection Circuits
Input Brownout, BO Protection
Brownout occurs when there is a drop in the line voltage. The BO
pin is a dual function pin. The BO pin detects the brownout
condition and shuts down the gate driver and controller. During
normal operation, the BO pin is used to compensate the effect of
the input line voltage change on the voltage loop. To keep the
harmonic distortion low, the corner frequency formed by the RBO
and CBO should be lower than 6Hz.
The BO pin is the output of the average voltage of the rectified
voltage. The PFC controller is turned off when the BO pin drops
below 0.4V. This protects the PFC power stage to enable
operation at or below brownout condition for long periods of
time. The controller resumes operation when the BO pin returns
to 0.5V.
The BO pin is usually connected to GND through a capacitor, CBO.
To avoid distortion on the VIN pin, select CBO so that:
Overvoltage Protection
If the voltage on the FB pin exceeds the reference voltage by about
4%, the gate driver is turned off. The controller resumes normal
operation after the FB pin drops below reference voltage.
Over-Temperature Protection
The ISL6730A, ISL6730B, ISL6730C, ISL6730D is protected
against over-temperature conditions. When the junction
temperature exceeds +160°C, the PWM shuts down. Normal
operation is resumed when the junction temperature decreases
below +135°C.
Application Guidelines
Layout Considerations
As in any high frequency switching converter, layout is very
important. Switching current from one power device to another
can generate voltage transients across the impedances of the
interconnecting bond wires and circuit traces. These
interconnecting impedances should be minimized by using wide,
short printed circuit traces. The critical components should be
located as close together as possible using ground plane
construction or single point grounding.
Figure 14 shows the critical power components; Q1, D and COUT.
To minimize the voltage overshoot, the interconnecting wires
indicated by heavy lines should be part of the ground or the
power plane in a printed circuit board. The components shown in
Figure 14 should be located as close together as possible. Please
note that the capacitors CVCC and CO each represent numerous
physical capacitors. Locate the ISL6730A, ISL6730B, ISL6730C,
ISL6730D within 2 inches of the MOSFET, Q1. The circuit traces
for the MOSFETs’ gate and source connections from the
ISL6730A, ISL6730B, ISL6730C, ISL6730D must be sized to
handle up to 1.5A peak current.
D
L
Q1
COUT
GATE
(EQ. 6)
C BO » 0.22μF
VCC
Overcurrent Protection
The peak current limiting function prevents the inductor from
saturation. The gate driver turns off when the current goes above
the current limit.
Overpower Protection
The overpower protection is implemented by limiting the COMP
pin voltage higher than 3.85V (typical).
12
CVCC
FIGURE 14. CRITICAL CURRENT POWER COMPONENTS
Component Selection Guidelines
A 300W, universal input, PFC converter design is provided for
demonstration. The design method is for a continuous current
mode power factor correction boost converter with the
ISL6730B/D. The switching frequency is 62kHz.
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
Table 3 shows the design parameters.
MIN
TYP
MAX UNIT
Select the bridge diode using Equation 15 and sufficient reverse
breakdown voltage. Assuming the forward voltage, VF,BR, is 1V
across each rectifier diode. The power loss of the rectifier bridge
can be calculated:
VLINE
85
115
265
VAC
P BR = 2 • V F, BR • I INAVE ( MAX )
(EQ. 15)
FLINE
47
63
Hz
300
W
P BR = 2 • 1V • 3.5A = 7W
(EQ. 16)
ms
INPUT CAPACITOR SELECTION
%
Refer to Table 2 for the recommended input filter capacitor value.
TABLE 3. CONVERTER DESIGN PARAMETERS
PARAMETER
CONDITIONS
POMAX
Maximum Output Power
THOLD
Hold Up Time
Efficiency
VLINE = 115VAC
20
92
0.33
C F1 = 300W • ----------- = 0.99μF
100
BOOST INDUCTOR SELECTION
First, calculate the maximum input RMS current, IINMAX.
P OMAX
I INMAX = ----------------------------------η • V RMSmin
(EQ. 7
Where η is the converter efficiency at VRMSmin. PF is the power
factor at VRMSmin.
300W
I INMAX = ---------------------------- = 3.84A
0.92 • 85V
(EQ. 8)
Assuming the current is sinusoidal and the peak to peak ripple at
line is 40%.
The boost inductor, LBST, is given by the following equation:
2 • V RMSmin⎞
2V RMSmin
⎛
L BST ≥ ---------------------------------------------------------------- • ⎜ 1 – ---------------------------------------⎟
V OUT
0.4 • F sw • 2 • I INMAX ⎝
⎠
85V
2 • 85V
L BST ≥ ------------------------------------------------------ • ⎛ 1 – ------------------------⎞ = 617μH
0.4 • 62kHz • 3.84 A ⎝
390V ⎠
(EQ. 9)
(EQ. 17)
This is the recommended capacitor used after the diode bridge.
For better power factor, less capacitance can be used. To lower
the input filter inductor size, more capacitance can be used.
Two 0.47µF capacitors in parallel are used for CF1.
BOOST DIODE SELECTION
The boost diode loss is determined by the diode forward voltage
drop, VF and the output average current. The maximum output
current is:
P OMAX
I OUT ( max ) = -------------------V OUT
(EQ. 18)
300W
I OUT ( max ) = ---------------- = 0.77A
390V
(EQ. 19)
The forward power loss on the diode is:
P FD = I OUT ( max ) • V F
(EQ. 20)
P FD = 0.77A • 1.85V = 1.42W
(EQ. 21)
(EQ. 10)
The peak current of the inductor is the sum of the average peak
inductor current and half of the peak to peak ripple current.
Select and design the boost inductor as given by Equation 11.
The ISL6730A, ISL6730B, ISL6730C, ISL6730D provides peak
current limit function that can prevent the boost inductor
saturation. Assuming 25% margin is given to the OCP threshold,
select and design the boost inductor with saturation current
given by Equation 11 with 25% more.
I LPeak =
0.4
2 • I INMAX • ⎛ 1 + --------⎞
⎝
2 ⎠
(EQ. 11)
I LPeak =
0.4
2 • 3.88A • ⎛ 1 + --------⎞ = 6.5A
⎝
2 ⎠
(EQ. 12)
The IDD03E60 part is selected.
The reverse recovery loss on the diode can be calculated. The
QRR is found from the diode datasheet. QRR = 220nC when
IF = 3.5A.
The reverse recover loss on the diode can be estimated:
1
P RRD = --- • Q
• V OUT • F
4
RR
sw
(EQ. 22)
1
P RRD = --- • 220nC • 390V • 62kHz = 1.33W
4
(EQ. 23)
The total power loss on the diode is:
P D = P FD + P RRD = ( 1.42 + 1.35 )W = 2.75W
INPUT RECTIFIER
The maximum average input current is calculated:
(EQ. 24)
MOSFET POWER DISSIPATION
2 • 2 • I INMAX
I INAVE ( max ) = -----------------------------------------π
(EQ. 13)
2 • 2 • 3.88A
I INAVE ( max ) = -------------------------------------- = 3.5A
π
(EQ. 14)
The power dissipation on the MOSFET is from two different types
of losses; the condition loss and the switching loss.
For the MOSFET, the worst case is at minimum line input voltage.
First, the drain to source RMS current is calculated:
8 2 V RMSmin
I DS ( max ) = I INMAX 1 – ----------- • -------------------------V
3π
(EQ. 25)
8 2 85V
I DS ( max ) = 3.88A 1 – ----------- • -------------- = 3.3A
3π 390V
(EQ. 26)
OUT
13
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
The MOSFET, SPP20N60C3 is selected.
(EQ. 27)
2
P COND = I DS ( max ) • R DS ( on )
2
P COND = 3.3A • 0.3Ω = 3.27W
When RG = 3.6Ω, ID = 6A, EON = 0.015mJ, EOFF = 0.007mJ.
The switching loss due to transition is calculated:
P SW = ( E ON + E OFF ) • F
sw
(EQ. 29)
P SW = ( 0.015mJ + 0.007mJ ) • 62kHz = 1.36W
(EQ. 30)
The diode reverse recovery incurs additional power loss on the
MOSFET. This loss can be estimated as:
(EQ. 31)
This loss is also related the di/dt during the MOSFET turn-on. The
di/dt can be found out from the MOSFET datasheet. At
RG = 3.6Ω, the turn-on di/dt is 4000A/µs. From the Typical
Reverse Recovery Charge curve at TJ = +125°C, the
QRR = 220nC when IF = 3.5A.
(EQ. 32)
RR
= 3.27W + 1.36W + 5.32W = 9.95W
CURRENT SENSING RESISTORS
Please refer to Equation 4 for calculation of the current sensing
resistor RCS.
120mV ⋅ 265V ⋅ 0.92
R CS ≥ ------------------------------------------------------- = 0.069Ω
2 ⋅ 300W
(EQ. 40)
While a large RCS renders better current sensing accuracy, larger
RCS also incurs higher power dissipation. Select RCS from
available standard value resistors to determine the sense
resistor.
(EQ. 41)
R CS = 0.068Ω
The maximum power dissipation on the RCS occurs at low line
and full load condition. The maximum power dissipation is
calculated:
P RCSMAX = 3.88A • 0.068Ω = 1.023W
(EQ. 42)
(EQ. 43)
The resistor, RSEN sets the overcurrent protection limit. From
Equation 3, RSEN should be greater than:
R CS • I LPeak • ( 1 + 0.25 )
R SEN ≥ -------------------------------------------------------------------2 • 0.5 I OC
(EQ. 44)
Where |x| stands for the ABS(x) function.
(EQ. 34)
0.068Ω • 6.6A • 1.25
R SEN ≥ -------------------------------------------------------- = 3.117kΩ
2 • 90μA
(EQ. 35)
Calculate the ripple RMS current through the capacitor:
(EQ. 45)
Select RSEN from available standard value resistors, the selected
RSEN is 3.16kΩ.
CURRENT LOOP COMPENSATION
V OUT
8 2
I CORMS ( max ) = I OUT ( max ) ----------- • -------------------------- – 1
3π V RMSmin
(EQ. 36)
8 2 390V
I CORMS ( max ) = 0.77A ----------- • -------------- – 1 = 1.635A
3π
85V
(EQ. 37)
Select the proper capacitor according to the hold time and ripple
RMS current requirement. The actual capacitance is 270µF.
It is important to make sure the output peak-to-peak ripple is
less than the minimum OVP threshold as specified in the
“Electrical Specifications” table on page 6. The ESR at 2 times of
the line frequency of the capacitor is found in the capacitor
datasheet. The ESR of the output capacitor is 770mΩ at 100Hz.
14
The minimum OVP threshold is 103% of the nominal output
value. The maximum output peak to peak ripple should be less
than 6% of the nominal value, which is 23.4VP-P.
2
(EQ. 33)
The output capacitor, COUT, is required to hold the output above
300V during one line cycle. For capacitors with 20% tolerance,
the tolerance should be taken into consideration. Thus, the
output capacitance should be greater than:
2 ⋅ 20ms ⋅ 300W
C OUT ≥ ---------------------------------------------- ⋅ 1.25 = 242μF
2
( 390 ) – ( 300V ) 2
(EQ. 39)
2
OUTPUT CAPACITOR SELECTION
2 ⋅ T HOLD ⋅ P
1
OMAX
C OUT ≥ ---------------------------------------------------- ⋅ ----------------2
2
1 – 0.2
V OUT
– V HOLD
( 4π ⋅ 50Hz ⋅ 270μF ⋅ 0.77Ω ) + 1
V Opp = 0.77A ⋅ -------------------------------------------------------------------------------------------- = 6.6V
( 4π ⋅ 50Hz ) ⋅ 270μF ⋅ 0.8
P RCSMAX = I INMAX • R CS
THE TOTAL LOSS ON THE MOSFET
P COND + P SW + P
(EQ. 38)
OUT
2
From the MOSFET datasheet, the typical switching losses curves
are provided.
P RR = 220nC • 390V • 62 kHz = 5.32W
line
(EQ. 28)
The switching loss of the MOSFET consists of three parts: the
turn-on loss, the turn-off loss and the diode reverse recovery loss.
P RR = Q RR • V OUT • F
sw
2
( 4πf line ⋅ C OUT ⋅ ESR ) + 1
V Opp = I OUT ( max ) ⋅ ------------------------------------------------------------------------------( 4πf
)⋅C
⋅ 0.8
The input current shaping is achieved by comparing the sensed
current signal to the sensed input voltage signal. The current
error amplifier (Gmi), together with the current compensation
network, adjusts the duty cycle so that the inductor current
traces the sensed rectified voltage. Thus, unity power factor is
achieved.
The compensation network consists of the Trans-Conductance
error amplifier (Gmi) and the impedance network (ZICOMP). The
goal of the compensation network is to provide a closed loop
transfer function with the sufficient 0dB crossing frequency
(f0dB) and adequate phase margin. Phase margin is the
difference between the open loop phase at f0dB and 180°. The
following equations relate the compensation network’s poles,
zeros and gain to the components (Ric, Cic and Cip) in Figure 15.
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
VI
The cross over frequency of the current loop should be set
between 2kHz to 100kHz. At cross over frequency, the transfer
function from duty cycle to inductor current is well approximated
by Equation 48:
VOUT
L
Q1
COUT
CF1
V OUT
G id ( s ) = ---------------------L BST ⋅ s
RCS
CURRENT
MIRROR
I
2:1
RSEN
CS
ICOMP
IREF
RIS
Gmi
Ric
2
⎛⎛
A iDC R CS ⎞
V OUT
1 + ( fc ⁄ fz ) ⎞
C ip + C ic = ⎜ ⎜ --------------------------------------- ⋅ ------------- ⋅ ----------------⎟ ⋅ ------------------------------⎟
⎜⎝
2 V
2⎟
R SEN⎠
+
(
⁄
f
)
1
f
m
L
⋅
(
2πf
)
c p ⎠
⎝ BST
c
FROM DUTY TO
INDUCTOR CURRENT
GAIN (dB)
COMPENSATION
GAIN
0
FP
FZ
-20
C ip + C ic = ( 19.8 )nF
(EQ. 52)
fz
C ip = ( C ip + C iC ) ---f
(EQ. 53)
The value of the noise filtering capacitor is:
-60
(EQ. 54)
2.12kHz
C ip = 14.9nF ⋅ ----------------------- = 1.35nF
31kHz
CURRENT GAIN
MODULATOR GAIN
100
1k
10k
The value of Cic is:
100k
(EQ. 55)
C ic = 19.8nF – 1.35nF = 18.4nF
FREQUENCY (Hz)
FIGURE 16. ASYMPTOTIC BODE PLOT OF CURRENT LOOP GAIN
1
F Z = -----------------------------------2π • R ic • C
(EQ. 51)
p
OPEN LOOP
GAIN
-40
(EQ. 50)
The total compensation capacitance is calculated:
40
-100
Where FC = FS/6 = 10.3kHz, ΦM is the phase margin, which is
60°. FP = FS/2 = 31kHz.
( 62KHz ) ⁄ 6
F Z = ------------------------------------------------------------- = 2.12kHz
2
⎛
tan atan ⎛ ---⎞ + 60deg⎞
⎝
⎝ 6⎠
⎠
80
-80
(EQ. 49)
Thus, the current loop compensation zero is:
Cip
FIGURE 15. INDUCTOR CURRENT SENSING SCHEME
20
It is recommended to set the cross over frequency from 1/10 to
1/6 of the switching frequency with phase margin of 60°. A high
frequency pole is set at 1/2 of the switching frequency for ripple
filtering. In this example, we set the cross over, FC at 1/6 of the
switching frequency.
FC
F Z = -------------------------------------------------------⎛
⎞
⎛ F C⎞
tan ⎜ atan ⎜ -------⎟ + Φ M⎟
⎝
⎠
⎝ F P⎠
ISEN
Cic
(EQ. 48)
(EQ. 46)
ic
The value of Ric is:
1
R ic = ----------------------------------------------------------- = 4.11kΩ
2π ⋅ 2.12kHz ⋅ 18.4nF
(EQ. 56)
Select the RC value from the standard value, we have:
1
F P = --------------------------------------------------C ip • C
ic
2π • R ic • -----------------------C ip + C ic
(EQ. 47)
Ric = 4.02kΩ, Cic = 18nF, Cip = 1.2nF. Figure 17 shows the actual
bode plot of current loop gain.
Use the following guidelines for locating the poles and zeros of
the compensation network.
15
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
80
For example, CF2 = 0.68µF, CF1 = 0.94µF, using the low cost EMI
filter shown in Figure 13. When VLINE = 230VAC, fLINE = 50Hz,
PO = 60W.
10.5kHz
GAIN (dB)
60
Assuming 95% efficiency under the above test condition, the
resistive component, which is in phase to voltage:
40
20
Po
I a = --------------------------------- = 0.275A
V LINE ⋅ 0.95
0
-20
The reactive current through the input capacitors:
180
PHASE (°)
(EQ. 62)
10.5kHz
(EQ. 63)
I c = V LINE • ( 2π ⋅ f LINE ) • ( C F1 + C F2 ) = 0.117A
135
Thus, the displacement power factor is:
90
60
45
Ia
PF DIS = ----------------------------------- = 0.92
2
2
( Ia ) + ( Ic )
1x103
The reactive current generated by the equivalent negative
capacitor is:
45
0
10
1x103
100
1x103
FREQUENCY (Hz)
FIGURE 17. BODE PLOT OF THE ACTUAL CURRENT LOOP GAIN
(EQ. 64)
(EQ. 65)
I cneg = V LINE • ( 2π ⋅ f LINE ) • ( C NEG ) = 0.045A
INPUT VOLTAGE SETTING
First, set the BO resistor divider gain, KBO according to
Equations 1 and 2.
With the equivalent negative capacitor, the total reactive current
reduces to:
Assuming the converter starts at VLINE = 80VRMS, then the BO
resistor divider gain, KBO should be:
I c – I cneg = 0.072A
(EQ. 57)
0.5V
K BO = ------------------------ = 0.00641
80V – 2V
In this design, two 3.3MΩ resistors in series are used for RIN2.
So, RIN1 is calculated:
0.00641
R IN1 = ------------------------------- ⋅ ( 6.6MΩ ) = 42.6kΩ
1 – 0.00641
(EQ. 58)
(EQ. 66)
The displacement power factor increases to:
Ia
PF DIS = -------------------------------------------------------- = 0.967
2
2
( I a ) + ( I c – I cneg )
(EQ. 67)
VOLTAGE LOOP COMPENSATION
The average diode forward current can be approximated by:
Using resistor from the standard value, RIN1 = 43kΩ, the actual
KBO is calculated:
P in
I D ( ave ) = ---------------V OUT
R IN1
K BO = --------------------------------- = 0.00647
R IN1 + R IN2
Assuming the input current traces the input voltage perfectly. The
input power is in proportion to (VCOMP - 1V).
(EQ. 59)
NEGATIVE INPUT CAPACITOR GENERATION
The ISL6730A, ISL6730B, ISL6730C, ISL6730D generates an
equivalent negative capacitance at the input to cancel the input
filter capacitance. Thus, more input capacitors can be used
without reducing the power factor.
The input equivalent negative capacitance is a function of the
current sensing gain, BO resistor divider gain and the
compensation components.
V m ⎞ R SEN
⎛
C NEG = ⎜ K BO ⋅ 0.8 – ----------------⎟ -------------------------- ( C ic + C ip )
V OUT⎠ R CS A iDC
⎝
(EQ. 68)
⎛
⎞
R SEN
1
0.25
I D ( ave ) = --------------------------------------- • ---------------- • ⎜ ------------------------------------------------⎟ • Δ COMP
⎜
⎟
2
R CS ⋅ 0.5 ⋅ R IS V OUT
⎝ ( ( 2 2 ) ⁄ π ) ⋅ K BO⎠
(EQ. 69)
Where ΔCOMP is the VCOMP - 1V. 1V is the offset voltage.
RIS is the internal current scaling resistor. RIS = 14.2kΩ.
A
I D ( ave ) = 0.598 ---- • Δ COMP
V
(EQ. 70)
(EQ. 60)
1.5
3.16k
C NEG = ⎛ 0.00647 ⋅ 0.8 – ----------⎞ --------------------------- ( 18nF + 1.2nF ) = 0.62μF
⎝
390⎠ 0.068 ⋅ 1.9
(EQ. 61)
This equivalent negative capacitor cancels the input filter
capacitor required for EMI filtering. Therefore, the displacement
power factor significantly improves.
16
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
VOUT
2.5V
The zero, FZv is calculated:
F CV
F Zv = -----------------------------------------------------------------------------tan ( Φ m + atan ( F CV ⁄ ( F Pv ) ) )
(EQ. 76)
8Hz
F Zv = ------------------------------------------------------------------------------------------------ = 1.15Hz
tan ( 60deg + atan ( ( 8Hz ) ⁄ ( 20Hz ) ) )
(EQ. 77)
RFB2
FB
Gmv
RFB1
IFB
Then the total capacitance used for compensation is calculated:
2
G PS ( i • ( 2πF CV ) ) • G DIV • Gmv
( F CV ⁄ F ZV ) + 1
C vc + C vp = ------------------------------------------------------------------------------------------- • ------------------------------------------2
( 2πF CV )
( F CV ⁄ F PV ) + 1
(EQ. 78)
COMP
Rvc
Thus, the total compensation capacitance is:
Cvp
Cvc
FIGURE 18. OUTPUT VOLTAGE SENSING AND COMPENSATION
Thus, the transfer function from VCOMP to VOUT is:
F ZV
C vp = 1829nF • ----------- = 105nF
F PV
(EQ. 81)
(EQ. 71)
⎛ I D ( ave )
1 ⎞
0.598
G PS ( s ) = ⎜ ------------------- ⋅ --------------------⎟ = ---------------CO ⋅ s
⎝ C O ⋅ s Δ COMP⎠
(EQ. 72)
As shown in Figure 18, the voltage loop gain is:
1
R vc = ------------------------------------------- = 81.2kΩ
2 ⋅ π ⋅ F ZV ⋅ C VC
(EQ. 82)
Choose components from the standard values. We have
CVP = 100nF, CVC = 1500nF, RVC = 82.5kΩ. The actual bode plot
is shown in Figure 20.
(EQ. 73)
G VLOOP ( s ) = G PS ( s ) • G DIV • gmv • Z COMP ( s )
60
V REF
G DIV = ---------------V OUT
(EQ. 74)
GAIN (dB)
40
The output feedback resistor divider gain, GDIV is:
R vc • C vc • s + 1
1
Z COMP ( s ) = --------------------------------------- • ------------------------------------------------------------( C vc + C vp ) ⋅ s R vc • C vc • C vp
------------------------------------------ • s + 1
C vc + C vp
20
0
0
-20
The compensation gain uses external impedance networks as
shown in Figure 18, ZCOMP(s) is given by:
-40
90
(EQ. 75)
PHASE (deg)
75
The targeted cross over frequency, FCV is 8Hz. The high frequency
pole, FPv is required in order to reject the 2 time line frequency
component. FPv = 20Hz. The targeted phase margin is 60°.
60
45
30
15
0
FZv
1
10
100
1x103
FREQUENCY (Hz)
80
FCV
FPv
FIGURE 20. BODE PLOT OF THE ACTUAL VOLTAGE LOOP GAIN
60
GAIN (dB)
(EQ. 80)
C vc = 1829nF – 105nF = 1724nF
I D ( ave )
V OUT ( s )
1
G PS ( s ) = ------------------------ = ---------------- ⋅ -------------------Δ COMP
C O ⋅ s Δ COMP
100
(EQ. 79)
C vc + C vp = 1829nF
Gmv*ZCOMP(s)
40
GPS(s)
20
0
GVLOOP(s)
-20
-40
-60
GDIV
1
10
100
1k
FREQUENCY (Hz)
FIGURE 19. ASYMPTOTIC BODE PLOT OF CURRENT LOOP GAIN
17
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to the web to make sure that
you have the latest revision.
DATE
REVISION
CHANGE
August 8, 2013
FN8258.1
Added electronic specifications to parts ISL6730B/D and made necessary changes throughout document.
February 26, 2013
FN8258.0
Initial Release.
About Intersil
Intersil Corporation is a leader in the design and manufacture of high-performance analog, mixed-signal and power management
semiconductors. The company's products address some of the largest markets within the industrial and infrastructure, personal
computing and high-end consumer markets. For more information about Intersil, visit our website at www.intersil.com.
For the most updated datasheet, application notes, related documentation and related parts, please see the respective product
information page found at www.intersil.com. You may report errors or suggestions for improving this datasheet by visiting
www.intersil.com/en/support/ask-an-expert.html. Reliability reports are also available from our website at
http://www.intersil.com/en/support/qualandreliability.html#reliability
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18
FN8258.1
August 8, 2013
ISL6730A, ISL6730B, ISL6730C, ISL6730D
Package Outline Drawing
M10.118
10 LEAD MINI SMALL OUTLINE PLASTIC PACKAGE
Rev 1, 4/12
5
3.0±0.05
A
DETAIL "X"
D
10
1.10 MAX
SIDE VIEW 2
0.09 - 0.20
4.9±0.15
3.0±0.05
5
0.95 REF
PIN# 1 ID
1
2
0.50 BSC
B
GAUGE
PLANE
TOP VIEW
0.55 ± 0.15
0.25
3°±3°
0.85±010
H
DETAIL "X"
C
SEATING PLANE
0.18 - 0.27
0.08 M C A-B D
0.10 ± 0.05
0.10 C
SIDE VIEW 1
(5.80)
NOTES:
(4.40)
(3.00)
1. Dimensions are in millimeters.
2. Dimensioning and tolerancing conform to JEDEC MO-187-BA
and AMSEY14.5m-1994.
3. Plastic or metal protrusions of 0.15mm max per side are not
included.
4. Plastic interlead protrusions of 0.15mm max per side are not
included.
(0.50)
(0.29)
(1.40)
5. Dimensions are measured at Datum Plane "H".
6. Dimensions in ( ) are for reference only.
TYPICAL RECOMMENDED LAND PATTERN
19
FN8258.1
August 8, 2013
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