STMicroelectronics AN4026 Using the l6563h and l6699 Datasheet

AN4026
Application note
19 V - 90 W adapter with PFC for laptop computers
using the L6563H and L6699
Introduction
This application note describes the performance of a 90 W, wide-range mains, power-factor
corrected, AC-DC adapter demonstration board. Its electrical specifications are tailored to a
typical hi-end portable computer power adapter.
The architecture is based on a two-stage approach; a front-end PFC pre-regulator based on
the L6563H TM PFC controller and a downstream LLC resonant half bridge converter using
the new L6699 resonant controller. Thanks to the chipset used, the main features of this
design are very high efficiency, compliant with ENERGY STAR® eligibility criteria (EPA rev.
2.0 EPS) and very good efficiency also at light load, and compliance to the new ErP Lot 6
Tier2 requirements. No load input power consumption is very low too, well within the
international regulation limits.
The controller of the LLC stage is the L6699, integrating innovative functions such as selfadjusting adaptive deadtime, anti-capacitive mode protection and proprietary “safe-start”
procedure preventing hard switching at startup.
Figure 1.
July 2012
EVL6699-90WADP: 90 W adapter demonstration board
Doc ID 022603 Rev 1
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www.st.com
Contents
AN4026
Contents
1
2
Main characteristics and circuit description . . . . . . . . . . . . . . . . . . . . . 6
1.1
Startup sequence . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.2
Brownout protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.3
Fast voltage feed-forward . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
1.4
Resonant power stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
1.5
Output voltage feedback loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
1.6
L6699 overload and short-circuit protection . . . . . . . . . . . . . . . . . . . . . . . . 8
1.7
Overvoltage and open loop protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
1.8
Light load operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Efficiency measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
2.1
Light load operation efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
3
Harmonic content measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
4
Functional check . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
4.1
Burst mode operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
4.2
Startup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
4.3
Overcurrent and short-circuit protection . . . . . . . . . . . . . . . . . . . . . . . . . . 21
4.4
Anti-capacitive mode protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
5
Thermal map . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
6
Conducted emission pre-compliance test . . . . . . . . . . . . . . . . . . . . . . 27
7
Bill of material . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
8
PFC coil specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
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8.1
General description and characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . 35
8.2
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
8.3
Electrical diagram and winding characteristics . . . . . . . . . . . . . . . . . . . . . 35
8.4
Mechanical aspect and pin numbering . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
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10
Contents
Transformer specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
9.1
General description and characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . 37
9.2
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
9.3
Electrical diagram and winding characteristics . . . . . . . . . . . . . . . . . . . . . 37
9.4
Mechanical aspect and pin numbering . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
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List of tables
AN4026
List of tables
Table 1.
Table 2.
Table 3.
Table 4.
Table 5.
Table 6.
Table 7.
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Overall efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Light load efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Thermal map reference points . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Bill of material . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
PFC coil winding data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Transformer winding data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
Document revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
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AN4026
List of figures
List of figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.
Figure 8.
Figure 9.
Figure 10.
Figure 11.
Figure 12.
Figure 13.
Figure 14.
Figure 15.
Figure 16.
Figure 17.
Figure 18.
Figure 19.
Figure 20.
Figure 21.
Figure 22.
Figure 23.
Figure 24.
Figure 25.
Figure 26.
Figure 27.
Figure 28.
Figure 29.
Figure 30.
Figure 31.
Figure 32.
Figure 33.
EVL6699-90WADP: 90 W adapter demonstration board . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Electrical diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Efficiency vs. output power diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Light load efficiency diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Compliance to EN61000-3-2 at 230 Vac - 50 Hz, full load . . . . . . . . . . . . . . . . . . . . . . . . . 15
Compliance to JEITA-MITI at 100 Vac - 60 Hz, full load. . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Mains voltage and current waveforms at 230 V - 50 Hz - full load . . . . . . . . . . . . . . . . . . . 16
Mains voltage and current waveforms at 100 V - 60 Hz - full load . . . . . . . . . . . . . . . . . . . 16
Resonant stage waveforms at 115 V - 60 Hz - full load . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Rectifier waveforms at 115 V - 50 Hz - full load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
HB transition at full load - rising edge . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
HB transition at full load - falling edge . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
HB transition at 0.25 A - rising edge . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
HB transition at 0.25 A - falling edge . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
L6699 pin signals-1. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
L6699 pin signals-2. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Pout = 250 mW operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Pout = 250 mW operation - detail . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Startup at 90 Vac - full load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Startup at 265 Vac - no load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Startup at full load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Startup at full load - detail . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Short-circuit at full load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Short-circuit at full load - detail . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Short-circuit - hiccup mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Thermal map at 115 Vac - 60 Hz - full load. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Thermal map at 230 Vac - 50 Hz - full load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
CE peak measurement at 115 Vac and full load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
CE average measurement at 230 Vac and full load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
PFC coil electrical diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
PFC coil mechanical aspect . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Transformer electrical diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
Transformer overall drawing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
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Main characteristics and circuit description
1
AN4026
Main characteristics and circuit description
The main features of the SMPS are listed here below:
●
Universal input mains range: from 90 to 264 Vac - frequency from 45 to 65 Hz
●
Output voltage: 19 V at 4.75 A continuous operation
●
Mains harmonics: meets EN61000-3-2 Class-D and JEITA-MITI Class-D
●
No load mains consumption: according to ENERGY STAR 2.0 for external power
supplies
●
Average efficiency: according to ENERGY STAR 2.0 for external power supplies
●
Light load efficiency: according to ErP Lot 6 Tier2 requirements
●
EMI: within EN55022-Class-B limits
●
Safety: meets EN60950
●
Dimensions: 65 x 151 mm, 25 mm components maximum height
●
PCB: double side, 70 µm, FR-4, mixed PTH/SMT.
The circuit is made up of two stages: a front-end PFC using the L6563H and an LLC
resonant converter featuring the L6699.
The PFC stage works as pre-regulator and powers the resonant stage with a constant
voltage of 400 V. The downstream converter operates only if the PFC is working and
regulating its output voltage. In this way, the resonant stage can be optimized for a narrow
input voltage range improving the efficiency of the primary side power components.
1.1
Startup sequence
As previously indicated, the PFC acts as master and the resonant stage can operate only if
the PFC output is delivering its rated output voltage. Therefore the circuit is designed so that
at startup the PFC starts first, the downstream converter then turns on by means of the
LINE pin (#7). At the beginning, the L6563H is supplied by the integrated high-voltage
startup circuit; once the PFC starts switching, a charging pump connected to the PFC
inductor supplies both PFC and resonant controllers. Once both stages are working, the
controllers are supplied also by the auxiliary winding of the resonant transformer, assuring
correct supply voltage even during standby operation.
After reaching the turn-on threshold on the VCC pin the L6563H integrated HV startup
circuit is turned off and it is therefore not dissipative during normal operation, significantly
contributing to the reduction of input power consumption once the power supply operates at
light load, and meeting the standby worldwide efficiency standards that are currently
required.
1.2
Brownout protection
Brownout protection prevents the circuit from working with abnormal mains levels. It is easily
achieved using the pin RUN (#12) of the L6563H: this pin is connected through a resistor
divider to the pin VFF (#5) which provides a DC voltage the same as the peak of the MULT
pin (#3) signal which is a partition of the rectified mains input voltage. An L6563H internal
comparator allows IC operations only if the mains level is correct, within the nominal limits,
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Main characteristics and circuit description
therefore at startup, if the input voltage is below 90 Vac (typ.), the circuit operation is
inhibited.
The L6699 has a similar protection monitoring the LLC input voltage on the LINE pin (#7). It
is used to prevent the resonant converter from working with too low input voltage that may
cause incorrect capacitive mode operation and the relevant protection intervention.
Therefore, if the bulk voltage (PFC output) is below 380 V (typ.), the resonant stage startup
is prevented. The L6699 LINE pin internal comparator has a current hysteresis, allowing to
set independently the turn-on and turn-off voltage. Turn-off threshold has been set to 300 V
(typ.) in order to avoid capacitive mode operation but allowing the resonant stage to operate
even in the case of a whole 20 ms mains cycle sag. Even with the consequent PFC output
drop the LLC converter is able to keep the output voltage regulated at the rated load.
1.3
Fast voltage feed-forward
Voltage on the L6563H VFF pin (#5) has the same value as the peak value of the voltage on
the MULT pin (#3) and it is generated by the RC network (R15 + R26, C12) connected to
VFF, completing an internal peak-holding circuit. This signal is necessary to derive
information from the RMS input voltage to compensate the loop gain that is mains voltage
dependent.
In general, if the VFF time constant is too small, the voltage generated is affected by a
considerable amount of ripple at twice the mains frequency. Because the VFF signal is fed
into the multiplier, the excessive ripple causes distortion of the current reference resulting in
high THD and poor PF. On the other hand, if the time constant is set too large, there is a
considerable delay in setting the right amount of feed-forward, resulting in excessive
overshoot or undershoot of the pre-regulator's output voltage in response to large line
voltage changes.
To overcome this issue, the L6563H implements the new fast voltage feed-forward function.
As soon as the voltage on the VFF pin decreases by a set threshold (40 mV typically), a
mains dip is assumed and an internal switch rapidly discharges the VFF capacitor via a 10
kΩ resistor. Thanks to this feature it is possible to set an RC circuit with a long time constant,
assuring a low THD, but obtaining a fast response to mains voltage variations by the PFC.
1.4
Resonant power stage
The downstream converter implements the L6699, a double-ended controller specific to the
series-resonant half bridge topology, supporting both LLC and LCC configurations. It
provides 50% complementary duty cycle: the high-side switch and the low-side switch are
driven ON/OFF 180° out-of-phase for exactly the same time. Output voltage regulation is
obtained by modulating the operating frequency. The deadtime inserted between the turnoff of one switch and the turn-on of the other is automatically adjusted to best fit the
transition times of the half bridge midpoint, allowing the improvement of efficiency thanks to
the transformer magnetizing inductance maximization and ensuring zero-voltage switching
in all LLC input voltage and output load conditions.
To drive the high-side switch with the bootstrap approach, the L6699 incorporates a highvoltage floating structure able to withstand more than 600 V with an internal synchronousdriven high-voltage DMOS replacing the external fast-recovery bootstrap diode to charge
the bootstrap capacitor powering the floating driver of the high-side MOSFET.
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Main characteristics and circuit description
AN4026
The L6699 enables the user to set the operating frequency range of the converter by means
of a high-accuracy externally programmable oscillator.
At startup, in addition to the traditional frequency-shift soft-start (the switching frequency
starts from a preset maximum value and then decays as far as the steady-state value
determined by the control loop), a proprietary circuit controls the half bridge to prevent hardswitching from occurring in the initial cycles because of the imbalance of the VOS applied to
the transformer. At light load, the L6699 is forced to enter a controlled burst mode operation
that keeps the converter input consumption as low as possible.
IC protection functions include a current sense input for OCP with frequency shift and
delayed shutdown with automatic restart. Fast shutdown with automatic restart occurs if this
first-level protection cannot control the primary current.
Additionally, the IC prevents the converter from working in, or too close to, capacitive mode,
to guarantee soft-switching. A latched disable input (DIS) is used to implement the OVP.
Other functions include a not-latched active-low disable input with current hysteresis, useful
for power sequencing or for brownout protection, and an interface with the PFC controller
that enables the pre-regulator to be switched off during fault conditions or during burst mode
operation.
The transformer uses the integrated magnetic approach, incorporating the resonant series
inductance. Therefore no external additional coil is needed for the resonance.
The transformer secondary winding configuration is centre tap and makes use of a couple of
power Schottky rectifiers p/n STPS30H60CFP. A small LC filter has been added on the
output to reduce the high frequency ripple and noise.
D15, R56, R62, R65, R66, Q5 and Q6 implement an output voltage “fast discharge” circuit
discharging quickly the output capacitors when the converter is turned off. It has been
implemented to quickly decrease the residual output voltage once the converter is turned off
at no load.
1.5
Output voltage feedback loop
The feedback loop is implemented by means of a typical circuit using a TL431 modulating
the current in the optocoupler diode.
On the primary side, R34 - connecting pin RFMIN (#4) to the optocoupler's phototransistor closes the feedback loop and its value sets the maximum switching frequency at about 105
kHz. This value has been chosen to limit the switching losses at light load operation. R31,
connecting the same pin to ground, sets the minimum switching frequency. The R-C series
R44 and C18 sets both soft-start maximum frequency and duration.
1.6
L6699 overload and short-circuit protection
The current into the primary winding is sensed by the lossless circuit R41, C27, D11, D10,
R39, and C25 and it is fed into the ISEN pin (#6). In case of overcurrent, the voltage on the
pin surpasses an internal comparator threshold (0.8 V), triggering a protection sequence.
The capacitor (C45) connected to the DELAY pin (#2) is charged by an internal 150 µA
current generator and is slowly discharged by the external resistor (R24). If the voltage on
the pin reaches 2 V, the soft-start capacitor is completely discharged so that the switching
frequency is pushed to its maximum value. As the voltage on the pin exceeds 3.5 V, the IC
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AN4026
Main characteristics and circuit description
stops switching and the internal generator is turned off, so that the voltage on the pin decays
because of the external resistor. The IC is soft-restarted as the voltage drops below 0.3 V. In
this way, under short-circuit conditions, the converter works intermittently with very low input
average power.
Please note that some preliminary demonstration boards use the PCB of the EVL6599A90WADP reworked for L6699. On these boards the silk screens report the original reference
designators for D10 and D11 but have mounted on those positions two resistors in place of
the diodes. The name of the boards reported on the PCB silkscreen top side is “90 W
adapter with L6563H and L6599A Rev. 1.0”.
More recent demonstration boards have updated the reference designators and the
silkscreen of D10 and D11 has been updated to R63 and R64 respectively. The name of
these more recent boards reported on the PCB top side is “90 W adapter with L6563H and
L6699 Rev. 1.0”. There is no circuit difference between the two boards.
1.7
Overvoltage and open loop protection
Both circuit stages, PFC and resonant, are equipped with their own overvoltage protection.
The L6563H controller monitors the PFC output voltage via the resistor divider connected to
a dedicated pin (PFC_OK, #7) protecting the circuit in the case of loop failures or
disconnection of the feedback loop divider connected to the INV pin (#1). If a fault condition
is detected, the PFC_OK circuitry latches the L6563H operations and, by means of the
PWM_LATCH pin (#8), it latches the L6699 too, via its DIS pin (#8). The converter is kept
latched by the L6563H HV circuit, which supplies the IC by charging the VCC capacitor
periodically. To resume converter operation, a mains restart is necessary.
The LLC open loop protection is guarantee by the Zener D8 sensing the voltage from the
transformer T1 auxiliary winding. In case of open loop the Zener stops the operation by
triggering the DIS pin (#8) of the L6699.
1.8
Light load operation
The board implements a burst mode function allowing a significant power saving during light
or no-load operation.
The L6699 STBY pin (#5) senses the optocoupler's collector voltage that is related to the
feedback control and is proportional to the output load. This signal is compared to an
internal reference (1.24 V); if the load decreases and the voltage on the STBY pin becomes
lower than the reference, the IC enters an idle state and its quiescent current is reduced.
Once the voltage exceeds the reference by 30 mV, the controller restarts switching. Burst
mode operation load threshold is programmed by properly choosing the resistor connecting
the optocoupler to pin RFmin (R34).
As already mentioned, the deadtime inserted between the two gate driver signals is
automatically adjusted to best fit the transition times of the half bridge midpoint, improving
the efficiency thanks to the transformer magnetizing inductance maximization. In detail,
increasing the transformer magnetizing inductance minimizes the magnetizing current,
providing a conduction loss decrease because of the lower total RMS current flowing into
the transformer primary side and half bridge MOSFETs. Because of the low current value at
MOSFET turn-off, a longer transition time of the half bridge is observed, for this reason the
deadtime takes longer in order to ensure the correct zero voltage switching operation by the
MOSFETs.
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Main characteristics and circuit description
AN4026
The L6563H implements its own burst mode function. If the COMP voltage falls below 2.5 V,
the IC stops switching causing an output voltage decrease, as a consequence the COMP
voltage rises again and the IC restarts switching.
In order to achieve a better load transient response, the PFC burst mode operation is
partially forced by the resonant converter: once the L6699 stops switching due to load
drops, its PFC_STOP pin pulls down the L6563H's PFC_OK pin, disabling PFC switching.
Thanks to this solution, the PFC is forced into idle state when the resonant stage is not
switching and rapidly wakes up when the downstream converter restarts switching. This
solution prevents a significant drop of the bulk voltage in the case of abrupt load rising.
10/40
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AN4026
Main characteristics and circuit description
Electrical diagram
":6 "
!-V
11/40
Efficiency measurements
2
AN4026
Efficiency measurements
Table 1 shows the no load consumption and the overall efficiency, measured at the nominal
mains voltages. At 115 Vac the average efficiency is 89.8%, while at 230 Vac it is 91.5%.
Both values are significantly higher than the minimum required by EPA rev2.0 external
power supply limits (87%).
Measurements are also reported in Figure 3 for reference.
Even at no load the board performance is superior: maximum no load consumption at
nominal mains voltage is less than 200 mW; also this value is significantly lower than the
limit imposed by the ENERGY STAR program, which is 500 mW, and it can meet more
stringent requirements.
Table 1.
Overall efficiency
230 V-50 Hz
Test
115 V-60 Hz
Vout
Iout
Pout
Pin
Eff.
Vout
Iout
Pout
Pin
Eff.
[V]
[A]
[W]
[W]
[%]
[V]
[A]
[W]
[W]
[%]
100% load eff. 18.75
4.74 88.88 96.70
91.9%
18.75
4.74
88.88
98.90
89.9%
75% load eff.
18.76
3.60 67.54 73.38
92.0%
18.76
3.60
67.54
74.82
90.3%
50% load eff.
18.77
2.40 45.05 49.32 91.34% 18.77
2.40
45.05
49.95
90.2%
25% load eff.
18.79
1.20 22.55 24.86 90.70% 18.78
1.20
22.54
25.39
88.8%
No load
18.98
0.00
0
0
0.162
0.00
0.186
Average eff.
Figure 3.
18.78
91.5%
89.8%
Efficiency vs. output power diagram
%FFICIENCY
-EAS %FF
6
-EAS %FF
6
!VG EFF REQUIRED
0OUT ;7=
!-V
12/40
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AN4026
2.1
Efficiency measurements
Light load operation efficiency
Measurement results are reported inTable 2. As seen, efficiency is better than 50% even for
very light loads such as 500 mW, as requested by the ErP Lot 6 Tier2 regulation for
computers. Such a high light load efficiency makes the board able to meet also the
regulation ENERGY STAR version 5.0 program for computers.
Measurement procedure:
Table 2.
1.
Because the current flowing through the circuit under measurement is relatively small,
the current measurement circuit is connected to the demonstration board side and the
voltage measurement circuit is connected to the AC source side. In this way the current
absorbed by the voltage circuit is not considered in the measured consumption
amount.
2.
During any efficiency measurement, remove any oscilloscope probe from the board.
3.
For any measurement load, apply a warm-up time of 20 minutes by each different load.
Loads have been applied increasing the output power from minimum to maximum.
4.
Because of the input current shape during light load condition, the input power
measurement may be critical or unreliable using a power meter in the usual way. To
overcome the issue, all light measurements have been done by measuring the active
energy consumption of the demonstration board under test and then calculating the
power as the energy divided by the integration time. The integration time has been set
at 36 seconds, as a compromise between a reliable measurement and a reasonable
time measurement time. The energy is measured in mWh, the result in mW is then
simply calculated by dividing the instrument reading (in mWh) by 100. The instrument
used was the Yokogawa WT210 power meter.
Light load efficiency
230 V-50 Hz
Test
115 V-60 Hz
Vout [V]
Iout
[mA]
Pout [W]
Pin [W]
Eff.
[%]
Vout [V]
Iout
[mA]
Pout
[W]
Pin [W]
Eff. [%]
0.125 W
18.79
7.30
0.137
0.342
40.1%
18.79
7.30
0.137
0.335
40.9%
0.25 W
18.79
13.30
0.250
0.47
53.1%
18.79
13.30
0.250
0.470
53.2%
0.5 W
18.79
27.00
0.507
0.760
66.8%
18.79
26.80
0.504
0.776
64.9%
1.0 W
18.79
54.00
1.015
1.335
76.0%
18.79
54.00
1.015
1.378
73.6%
1.5 W
18.79
79.50
1.494
1.876
79.6%
18.79
79.50
1.494
1.954
76.4%
2.0 W
18.79
106.50
2.001
2.446
81.8%
18.79
106.50
2.001
2.556
78.3%
2.5 W
18.79
133.30
2.505
3.030
82.7%
18.79
133.30
2.505
3.148
79.6%
3.0 W
18.79
106.50
3.016
3.598
83.8%
18.79
106.50
3.016
3.734
80.8%
3.5 W
18.79
186.00
3.495
4.138
84.5%
18.79
186.00
3.495
4.283
81.6%
4.0 W
18.79
213.00
4.002
4.708
85.0%
18.79
213.00
4.002
4.860
82.4%
4.5 W
18.79
240.00
4.510
5.269
85.6%
18.79
240.00
4.510
5.430
83.0%
5.0 W
18.79
267.10
5.019
5.832
86.1%
18.79
267.10
5.019
6.007
83.5%
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13/40
Efficiency measurements
Figure 4.
AN4026
Light load efficiency diagram
%FFICIENCY ;=
6 (Z
6 (Z
0OUT ;7=
!-V
The measurements are reported as a graph in Figure 4. Note that the efficiency is better
than 50% in all conditions and it becomes higher than 70% at 1 Watt output power, meeting
all worldwide light load consumption regulations such as the ErP Lot 6 Tier2 or the
ENERGY STAR program version 5.0 for computers.
14/40
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AN4026
3
Harmonic content measurement
Harmonic content measurement
The board has been tested against the European standard EN61000-3-2 Class-D and
Japanese standard JEITA-MITI Class-D compliance, at both the nominal input voltage
mains. As reported in the following figures, the circuit is able to reduce the harmonics well
below the limits of both regulations.
On the bottom side of the diagrams the total harmonic distortion and power factor have been
measured too. The values in all conditions give a clear idea about the correct functionality of
the PFC.
Figure 5.
Compliance to EN61000-3-2 at 230
Vac - 50 Hz, full load
THD = 17.2%
PF = 0.975
Figure 6.
Compliance to JEITA-MITI at 100
Vac - 60 Hz, full load
THD = 4.78%
PF = 0.992
Figure 7 and Figure 8 show the input mains current at both nominal mains input voltages,
European and Japanese. At European mains the waveforms show a slightly higher THD
value because in order to increase the efficiency, the PFC switching frequency is limited at a
value around 130 kHz. However, all harmonics are within the limits specified by both
regulations.
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Harmonic content measurement
Figure 7.
Mains voltage and current
waveforms at 230 V - 50 Hz - full
load
CH1: mains voltage
16/40
AN4026
CH2: mains current
Figure 8.
Mains voltage and current
waveforms at 100 V - 60 Hz - full
load
CH1: mains voltage
Doc ID 022603 Rev 1
CH2: mains current
AN4026
4
Functional check
Functional check
The following figures show the waveforms relevant to the resonant stage during steady-state
operation. The working frequency during steady-state operation at rated load is about 100
kHz, in order to have a good trade-off between transformer losses and dimensions. The
converter operates above the resonance frequency. Figure 9 shows the HB voltage and
resonant tank current with both the half bridge driving signals.
In Figure 10 the rectifiers reverse working voltages are captured: the good margin can be
noted compared with the BV rectifiers, ensuring reliable, long term operation.
Figure 9.
Resonant stage waveforms at 115 V Figure 10. Rectifier waveforms at 115 V - 50 Hz
- 60 Hz - full load
- full load
CH1: HB voltage
CH3: HVG
CH2: LVG
CH4: res. tank current
CH1: HB voltage
CH3: LV FET gate
CH2: HV FET gate
CH4: res. tank current
A peculiarity of the L6699 is the self-adaptive deadtime, modulated by the internal logic
according to the half bridge node transition time. Figure 11 and Figure 12 show the
waveforms during full load operation. It is possible to note the measurement of the edges
and the relevant deadtime. It can be seen that both MOSFETs are turned on when resonant
current is flowing through their body diodes and drain-source voltage is zero, therefore
achieving the MOSFETs zero voltage switching (ZVS) operation at turn-on.
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Functional check
AN4026
Figure 11. HB transition at full load - rising
edge
CH1: HB voltage
CH3: HVG
CH2: LVG
CH4: res. tank current
Figure 12. HB transition at full load - falling
edge
CH1: HB voltage
CH3: HVG
CH2: LVG
CH4: res. tank current
In Figure 13 and Figure 14 the same images are captured during light load operation: note
that the half bridge transition is slower because of the lower switched current. In this case
the L6699 increases the deadtime maintaining the correct zero voltage switching operation
of the circuit. This feature allows the maximization of the transformer magnetizing
inductance, therefore obtaining good light load efficiency and also keeping correct operation
by the HB.
Figure 13. HB transition at 0.25 A - rising edge Figure 14. HB transition at 0.25 A - falling edge
CH1: HB voltage
CH3: HVG
CH2: LVG
CH4: res. tank current
CH1: HB voltage
CH3: HVG
CH2: LVG
CH4: res. tank current
In Figure 15 some signals at the L6699 pins are measured. Note that the signal on pin ISEN
(#6) matches the instantaneous current flowing in the transformer primary side. Contrary to
the former resonant controllers such as the L6599A and others, requiring an integration of
current signal, the L6699 integrates the anti-capacitive mode protection it needs by sensing
the instantaneous value of the current, in order to check the phase between the voltage and
current. For the same reason, the time constant of the typical RC filter placed between the
sensing resistor and the IC pin must be maximum in the range of 200/250 ns. A significant
higher value would affect the correct anti-capacitive mode protection because of the phase
lag of current signal fed into the L6699.
18/40
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Functional check
The LINE pin (#7) has been dimensioned to start up the L6699 once the PFC output voltage
has reached the rated value, in order to have correct converter sequencing, with PFC
starting first and LLC starting later in order to optimize the design of the LLC converter and
prevent capacitive mode operation that may occur because of operation at too low input
voltage.
Figure 15. L6699 pin signals-1
CH1: DIS
CH3: DELAY
Figure 16. L6699 pin signals-2
CH1: RFmin
CH3: CSS
CH2: LINE
CH4: ISEN
CH2: STBY
CH4: CF
The DELAY pin (#2) is zero, as it must be, during normal operation because it works during
the overcurrent protection operation. The DIS pin (#8) is used for open loop protection and
therefore, even in this case, its voltage is at ground level.
In Figure 16 the pin voltages relevant to the control part of the L6699 are reported: the
RFmin pin (#4) is the 2 V (typ.) reference voltage of the oscillator, the switching frequency is
proportional to the current flowing out from the pin. The CSS pin (#1) voltage is the same
value as pin #4 because it is connected to the latter via a resistor (R44), determining the
soft-start frequency. A capacitor (C18) is also connected between the CSS pin and ground
to set the soft-start time. At the beginning of L6699 operation the voltage on the CSS pin is
at ground level because C18 is discharged, the CSS pin (#1) voltage then increases
according to the time constant till the RFmin voltage level is reached. The STBY pin (#5)
senses the optocoupler voltage, once the voltage decreases to 1.25 V both gate drivers stop
switching and the circuit works in burst mode. The CF pin (#3) is the controller oscillator; its
ramp speed is proportional to the current flowing out from the RFmin pin (#4). The CF signal
must be clean and undistorted to obtain correct symmetry by the half bridge current,
therefore, care must be taken in the layout of the PCB.
4.1
Burst mode operation
In Figure 17 some burst mode pulses are captured during a 250 mW load operation. Note
that the burst pulses are very narrow and their period is quite long, therefore the resulting
equivalent switching frequency is very low, ensuring high efficiency. The resulting output
voltage ripple during burst mode operation is about 58 mV peak-to-peak.
In Figure 18 the detail of the burst is reported: note that the first initial pulse is shorter than
the following ones, avoiding the typical high current peak at half bridge operation restart due
to the recharging of the resonant capacitor. It is also possible to note that the maximum
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Functional check
AN4026
operating frequency of the half bridge, set by the resistor R34 in series to the optocoupler, is
103 kHz.
Figure 17. Pout = 250 mW operation
CH1: HB voltage
CH3: STBY
4.2
Figure 18. Pout = 250 mW operation - detail
CH2: LVG
CH4: V.out (AC coupl.)
CH1: HB voltage
CH3: STBY
CH2: LVG
CH4: res. tank current
Startup
The waveforms relevant to the adapter startup at 90 Vac and full load have been captured in
Figure 19. The output voltage reaches the nominal value 600 ms after plug-in. The L6563H,
HV PFC controller, has an embedded high-voltage startup charging the VCC capacitor by a
constant current, ensuring a constant wake-up time. Comparing Figure 19 with Figure 20
relevant to a startup at 265 Vac and no load, the output voltage rises at the nominal level in
the same time. In both conditions the output voltage has no overshoot and the rise is
monotonic.
Figure 19. Startup at 90 Vac - full load
CH1: HB voltage
CH3: +19 Vout
Figure 20. Startup at 265 Vac - no load
CH2: GD L6563H
CH4: VCC L6563H
CH1: HB voltage
CH3: +19 Vout
CH2: GD L6563H
CH4: VCC L6563H
In Figure 21 the salient waveforms in the resonant tank during startup of the LLC are
reported. In Figure 22 the detail of waveforms at the beginning of operation shows that the
resonant circuit is working correctly in zero voltage switching operation from the initial
20/40
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AN4026
Functional check
cycles. In the L6699, a new startup procedure, called “safe-start”, has been implemented to
prevent loss of soft-switching during the initial switching cycles, which is typically not
guaranteed by the usual soft-start procedure. At startup, the voltage across Cr is often quite
different from Vin/2, as during the normal steady-state operation, so it takes some time for
its DC component to reach the steady-state value Vin/2. During this transient, the
transformer is not driven symmetrically and, then, there is a significant V·s imbalance in two
consecutive half-cycles. If this imbalance is large, there is a significant difference in the up
and down slopes of the tank current and, in a typical controller working with fixed 50% duty
cycle, with the duration of the two half-cycles being the same, the current may not reverse in
a switching half-cycle. Therefore, one MOSFET can be turned on while the body diode of
the other is conducting and this may happen for a few cycles. To prevent this, the L6699 is
provided with a proprietary circuit that modifies the normal operation of the oscillator during
the initial switching cycles, so that the initial V·s unbalance is nearly eliminated. Its operation
is such that current reversal in every switching half-cycle and, then, soft-switching is
ensured. In Figure 22 it is possible to note that at the beginning of operation the duty cycle
of the half bridge is initially considerably less than 50%, the tank current has lower peak
values and changes sign every half-cycle, while the DC voltage across the resonant
capacitor reaches the steady-state. The device goes to normal operation after
approximately 50 µs from the first switching cycle. This transition is nearly seamless and just
a small perturbation of the tank current can be observed.
Figure 21. Startup at full load
CH1: HB voltage
CH3: CSS
4.3
CH2: LVG
CH4: ISEN
Figure 22. Startup at full load - detail
CH1: HB voltage
CH3: CSS
CH2: LVG
CH4: ISEN
Overcurrent and short-circuit protection
The L6699 is equipped with a current sensing input (pin #6, ISEN) and a dedicated
overcurrent management system. The current flowing in the resonant tank is detected and
the signal is fed into the ISEN pin. It is internally connected to a first comparator, referenced
to 0.8 V, and to a second comparator referenced to 1.5 V. If the voltage externally applied to
the pin exceeds 0.8 V, the first comparator is tripped causing an internal switch to be turned
on and to discharge the soft-start capacitor CSS.
Under output short-circuit, this operation results in a nearly constant peak primary current.
With the L6699, the user can program externally the maximum time that the converter is
allowed to run overloaded or under short-circuit conditions. Overloads or short-circuits
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Functional check
AN4026
lasting less than the set time do not cause any other action, therefore providing the system
with immunity to short duration phenomena. If, instead, overload condition keeps going, a
protection procedure is activated that shuts down the L6699 and, in the case of continuous
overload/short-circuit, results in continuous intermittent operation with a user defined duty
cycle. This function is realized with the pin DELAY (#2), by means of a capacitor C45 and
the parallel resistor R24 connected to ground. As the voltage on the ISEN pin exceeds 0.8 V,
the first OCP comparator, in addition to discharging CSS, turns on an internal 150 µA
current generator that, via the DELAY pin, charges C45. As the voltage on C45 is 3.5 V, the
L6699 stops switching and the PFC_STOP pin is pulled low. Also the internal generator is
turned off so that C45 is now slowly discharged by R24. The IC restarts when the voltage on
C45 is less than 0.3 V. Additionally, if the voltage on the ISEN pin reaches 1.5 V for any
reason (e.g. transformer saturation), the second comparator is triggered, the L6699 shuts
down and the operation is resumed once the voltage on C45 drops below 0.3 V.
In Figure 23 a dead short-circuit event has been captured. In this case the overcurrent
protection is triggered by the second comparator referenced at 1.5 V, it immediately stops
switching by the L6699 and discharging of the soft-start capacitor; at the same time the
capacitor connected to the DELAY pin (#2) begins charging up to 3.5 V (typ.). Once the
voltage on the DELAY pin reaches 3.5 V, the L6699 stops charging the delay capacitor
(C45), the L6699 operation is resumed once the DELAY pin (#2) voltage decays to 0.3 V
(typ.) by the parallel resistor (R24), via a soft-start cycle. If the short-circuit condition is
removed, the converter again starts operation, otherwise, if the short is still there, it results
in an intermittent operation (hiccup mode) with a narrow operating duty cycle of the
converter, in order to prevent the overheating of power components, as can be noted in
Figure 25.
In Figure 24 details of peak current at short-circuit occurring is captured. Note the ZVS
correct operation by the half bridge MOSFETs.
Figure 23. Short-circuit at full load
CH1: HB voltage
CH3: CSS
22/40
Figure 24. Short-circuit at full load - detail
CH2: LVG
CH4: DELAY
CH1: HB voltage
CH3: CSS
Doc ID 022603 Rev 1
CH2: LVG
CH4: DELAY
AN4026
Functional check
Figure 25. Short-circuit - hiccup mode
CH1: HB voltage
CH3: CSS
4.4
CH2: LVG
CH4: DELAY
Anti-capacitive mode protection
The EVL6699-90WADP demonstration board has been designed in such a way that the
system does not work in capacitive mode during normal operation or failure conditions, as
seen in Figure 24, even in dead short condition the LLC operates correctly in the inductive
region, the same correct operation occurs during load and input voltage transients.
Normally, the resonant half bridge converter operates with the resonant tank current lagging
behind the square-wave voltage applied by the half bridge leg, like a circuit having a
reactance of an inductive nature. In this way the applied voltage and the resonant current
have the same sign at every transition of the half bridge, which is a necessary condition in
order for soft-switching to occur (zero-voltage switching, ZVS at turn-on for both MOSFETs).
Therefore, should the phase relationship reverse, i.e. the resonant tank current leading the
applied voltage, like in circuits having a capacitive reactance, soft-switching would be lost.
This is termed capacitive-mode operation and must be avoided because of its significant
drawbacks.
Both MOSFETs feature hard-switching at turn-on, like in conventional PWM-controlled
converters (see Figure 14). The associated capacitive losses may be considerably higher
than the total power normally dissipated under “soft-switching” conditions and this may
easily lead to their overheating, since heatsinking is not usually sized to handle this
abnormal condition.
The body diode of the MOSFET just switched off conducts current during the deadtime and
its voltage is abruptly reversed by the other MOSFET turned on (Figure 14). Therefore, the
conducting body diode (which does not generally have great reverse recovery
characteristics) keeps its low impedance until it recovers, therefore originating a condition
equivalent to a shoot-through of the half bridge leg. This is a potentially destructive condition
and causes additional power dissipation due to the current and voltage of the conducting
body diode simultaneously high during part of its recovery.
There is an extremely high reverse dv/dt (many tens of V/ns!) experienced by the conducting
body diode at the end of its recovery with the other MOSFET turned on. This dv/dt may
exceed the rating of the MOSFET and lead to an immediate failure because of the second
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Functional check
AN4026
breakdown of the parasitic BJT intrinsic in its structure. If a MOSFET is hot, the turn-on
threshold of its parasitic BJT is lower, and this dv/dt-induced failure is much more likely.
When either MOSFET is turned on, the other one can be parasitically turned on too, if the
current injected through its Cgd and flowing through the gate driver's pull-down is large
enough to raise the gate voltage close to the turn-on threshold. This would be a lethal shootthrough condition for the half bridge leg.
The recovery of the body diodes generates large and energetic negative voltage spikes
because of the unavoidable parasitic inductance of the PCB subject to its di/dt. These are
coupled to the OUT pin and may damage the L6699.
There is a large common-mode EMI generation that adversely affects EMC.
Resonant converters work in capacitive mode when their switching frequency falls below a
critical value that depends on the loading conditions and the input-to-output voltage ratio.
They are especially prone to run into capacitive-mode when the input voltage is lower than
the minimum specified and/or the output is overloaded or short-circuited. Designing a
converter so that it never works in capacitive-mode, even under abnormal operating
conditions, is definitely possible but this may pose unacceptable design constraints in some
cases.
To prevent the severe drawbacks of capacitive-mode operation, while enabling a design that
needs to ensure inductive-mode operation only in the specified operating range, neglecting
abnormal operating conditions, the L6699 provides the capacitive-mode detection function.
The IC monitors the phase relationship between the tank current circuit sensed on the ISEN
pin and the voltage applied to the tank circuit by the half bridge, checking that the former
lags behind the latter (inductive-mode operation). If the phase-shift approaches zero, which
is indicative of impending capacitive-mode operation, the monitoring circuit activates the
OCP procedure so that the resulting frequency rise keeps the converter away from that
dangerous condition. Also in this case the DELAY pin is activated, so that the OLP function,
if used, is eventually tripped after a time TSH causing intermittent operation and reducing
thermal stress.
If the phase relationship reverses abruptly (which may happen in the case of dead short at
the converter's output), the L6699 is stopped immediately, the soft-start capacitor CSS is
totally discharged and a new soft-start cycle is initiated after 50 µs idle time. During this idle
period the PFC_STOP pin is pulled low to stop the PFC stage as well.
24/40
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AN4026
5
Thermal map
Thermal map
In order to check the design reliability, a thermal mapping by means of an IR camera was
carried out. In Figure 26 and 27 the thermal measurements of the board, component side, at
nominal input voltage are shown. Some pointers visible on the images have been placed
across key components or showing high temperature. The ambient temperature during both
measurements was 27 °C. All components are working within their operating temperature
range with margin.
Figure 26. Thermal map at 115 Vac - 60 Hz - full load.
Figure 27. Thermal map at 230 Vac - 50 Hz - full load
Table 3.
Thermal map reference points
Point
Reference
Description
A
L1
EMI filtering inductor
B
D1
Bridge rectifier
C
L2
PFC inductor – hottest point
D
D4
PFC output diode
E
Q1
PFC MOSFET
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Thermal map
AN4026
Table 3.
26/40
Thermal map reference points (continued)
Point
Reference
F-G
Q3 and Q4
H-I-J
T1
J-K
D23 and D24
Description
Resonant HB MOSFETs
Resonant power transformer
Output rectifiers
Doc ID 022603 Rev 1
AN4026
6
Conducted emission pre-compliance test
Conducted emission pre-compliance test
Figure 28 and 29 show the measurements of the conducted noise in average detection, at
full load and nominal mains voltages. The limits shown in the diagrams are the EN55022
Class-B ones, which is the most popular rule for domestic equipment and has more severe
limits compared to Class-A, dedicated to IT technology equipment. As seen in the diagrams,
in all test conditions the measurements are well below the limits.
Figure 28. CE peak measurement at 115 Vac and full load
Doc ID 022603 Rev 1
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Conducted emission pre-compliance test
Figure 29. CE average measurement at 230 Vac and full load
28/40
Doc ID 022603 Rev 1
AN4026
AN4026
Bill of material
7
Bill of material
Table 4.
Bill of material
Part type/
Case style
part value
/package
C1
470 nF
9.0 × 18.0 p.15 mm
C2
2n2F
DWG
Y1 CAP. CD12-E2GA222MYGSA
TDK-EPC
C3
2n2F
DWG
Y1 CAP. CD12-E2GA222MYGSA
TDK-EPC
C4
470 nF
9.0 × 18.0 p.15 mm
C5
470 nF
7.0 x 16.0 p. 22.5
mm
C6
N.M.
0805
50 V CERCAP - general purpose
C7
100 nF
PTH
100 V CERCAP - general purpose
C8
47 μF
Dia. 6.3x11 mm
50 V aluminium ELCAP - YXF series 105 °C
C9
68 μF
Dia. 18x32 mm
450 V aluminium ELCAP - KXG series 105 °C
C10
1 nF
0805
50 V CERCAP - general purpose
AVX
C11
2n2F
0805
50 V CERCAP - general purpose
AVX
C12
1 μF
0805
25 V CERCAP - general purpose
AVX
C13
680 nF
1206
25 V CERCAP - general purpose
AVX
C14
68 nF
0805
50 V CERCAP - general purpose
AVX
C15
47 μF
Dia. 6.3x11 mm
C16
2n2F
1206
50 V CERCAP - general purpose
AVX
C17
220 pF
0805
50 V - 5% - C0G - CERCAP
AVX
C18
2.2 μF
1206
6.3 V CERCAP - general purpose
AVX
C19
100 nF
1206
50 V CERCAP - general purpose
AVX
C20
2n2F
DWG
Y2 CAP. CS11-E2GA222MYGSA
TDK-EPC
C21
2n2F
DWG
Y2 CAP. CS11-E2GA222MYGSA
TDK-EPC
C22
220 pF
0805
50 V CERCAP - general purpose
AVX
C23
10 nF
0805
50 V CERCAP - general purpose
AVX
C24
100 μF
Dia.10x12.5 mm
C25
470 pF
0805
C26
10 μF
Dia.6.3x11 mm
C27
220 pF
5x3 mm
Des.
Description
X2 - MKP FILM CAP - B32922C3474K
Supplier
EPCOS
X2 - MKP FILM CAP - B32922C3474K
EPCOS
400 V - FILM CAP - B32673Z5474
EPCOS
50 V aluminium ELCAP - YXF series 105 °C
50 V aluminium ELCAP - YXF series 105 °C
50 V CERCAP - general purpose
50 V aluminium ELCAP - YXF series 105 °C
500 V CERCAP - 5MQ221KAAAA
Doc ID 022603 Rev 1
AVX
Rubycon
UNITED
CHEMICON
Rubycon
Rubycon
AVX
Rubycon
AVX
29/40
Bill of material
Table 4.
AN4026
Bill of material (continued)
Part type/
Case style
part value
/package
C28
15 nF
5x18 p.15 mm
1000 V - MKP film cap B32652A0153K000
EPCOS
C29
470 μF
Dia.10x20 mm
35 V aluminium ELCAP - ZL series 105 °C
Rubycon
C30
470 μF
Dia.10x20 mm
35 V aluminium ELCAP - ZL series 105 °C
Rubycon
C31
100 μF
Dia. 8x11 mm
35 V aluminium ELCAP - YXF series 105 °C
Rubycon
C32
100 nF
0805
50 V CERCAP - general purpose
AVX
C33
1N0
0805
50 V - 5% - C0G - CERCAP
AVX
C34
470 nF
0805
25 V CERCAP - general purpose
AVX
C36
N.M.
Dia. 6.3x11 mm
C37
100 nF
0805
50 V CERCAP - general purpose
AVX
C39
100 nF
0805
50 V CERCAP - general purpose
AVX
C40
100 nF
1206
50 V CERCAP - general purpose
AVX
C43
4n7F
1206
50 V CERCAP - general purpose
AVX
C44
10 nF
1206
50 V CERCAP - general purpose
AVX
C45
220 nF
0805
25 V CERCAP - general purpose
AVX
C46
N.M.
0805
Not mounted
D1
GBU8J
STYLE GBU
D2
LL4148
D3
Des.
Description
Supplier
Not mounted
Single-phase bridge rectifier
Vishay
Mini Melf SOD-80
High speed signal diode
Vishay
1N4005
DO-41 DO - 41
General purpose rectifier
Vishay
D4
STTH2L06
DO-41
D5
LL4148
Mini Melf SOD-80
High speed signal diode
Vishay
D6
LL4148
Mini Melf SOD-80
High speed signal diode
Vishay
D7
BAT48Z
SOD-123
D8
BZV55-B24
Mini Melf SOD-80
D9
STPS1L60A
SMA
Power Schottky rectifier
D10
(R63)
470R
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
- [1]
Vishay
D11
(R64)
39R
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
- [1]
Vishay
D12
N.M.
Mini Melf SOD-80
Not mounted
D13
N.M.
Mini Melf SOD-80
Not mounted
D14
N.M.
Mini Melf SOD-80
Not mounted
D15
BZV55-C15
Mini Melf SOD-80
Zener diode
30/40
Ultrafast high-voltage rectifier
Small signal Schottky diode
Zener diode
Doc ID 022603 Rev 1
STMicroelectronics
STMicroelectronics
Vishay
STMicroelectronics
Vishay
AN4026
Table 4.
Bill of material
Bill of material (continued)
Part type/
Case style
part value
/package
D16
N.M.
Mini Melf SOD-80
High speed signal diode
D17
N.M.
Mini Melf SOD-80
Not mounted
D18
LL4148
Mini Melf SOD-80
High speed signal diode
Vishay
D19
LL4148
Mini Melf SOD-80
High speed signal diode
Vishay
D20
BZV55-C15
Mini Melf SOD-80
Zener diode
Vishay
D21
N.M.
Mini Melf SOD-80
Zener diode
D22
LL4148
Mini Melf SOD-80
Fast switching diode
D23
STPS30H60CFP
TO-220FP
Power Schottky rectifier
STMicroelectronics
D24
STPS30H60CFP
TO-220FP
Power Schottky rectifier
STMicroelectronics
F1
Fuse T4A
8.5x4 p.5.08 mm
Fuse 4A - time lag - 3921400
HS1
Heatsink
DWG
Heatsink for D1, Q1, Q3, Q4
HS2
Heatsink
DWG
Heatsink for D23, D24
J1
MKDS 1,5/ 3-5,08
DWG
PCB term. block, screw conn., pitch 5
mm - 3 W.
PHOENIX
CONTACT
J2
MKDS 1,5/ 2-5,08
DWG
PCB term. block, screw conn., pitch 5
mm - 2 W.
PHOENIX
CONTACT
JPX1
Jumper
Wire
Bare copper wire jumper
JPX2
Jumper
Wire
Bare copper wire jumper
JPX3
Jumper
Wire
Bare copper wire jumper
L1
2019.0002
L2
1974.0004
L3
Des.
Description
Supplier
Vishay
LITTLEFUSE
CM inductor 2x18mH 1.8A
MAGNETICA
DWG
PFC inductor - 0.52 mH
MAGNETICA
1071.0083
DWG
Inductor 1 µH - 5 A
MAGNETICA
Q1
STF12NM50N
TO-220FP
Q2
BC857C
SOT-23
Q3
STF8NM50N
TO-220FP
N-channel Power MOSFET
STMicroelectronics
Q4
STF8NM50N
TO-220FP
N-channel Power MOSFET
STMicroelectronics
Q5
BC847C
SOT-23
NPN small signal BJT
Vishay
Q6
BC847C
SOT-23
NPN small signal BJT
Vishay
Q7
N.M.
SOT-23
PNP small signal BJT - not used
Q9
BC847C
SOT-23
NPN small signal BJT
Q10
N.M.
SOT-23
NPN small signal BJT
R1
3M3
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
Vishay
R2
3M3
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
Vishay
R3
1Meg
1206
SMD film res. - 1/4 W - 1% - 100 ppm/°C
Vishay
R4
N.M.
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
N-channel Power MOSFET
PNP small signal BJT
Doc ID 022603 Rev 1
STMicroelectronics
Vishay
Vishay
31/40
Bill of material
Table 4.
AN4026
Bill of material (continued)
Part type/
Case style
part value
/package
R5
10R
R6
Des.
Description
Supplier
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
Vishay
NTC 2R5-S237
DWG
NTC resistor p/n B57237S0259M000
EPCOS
R7
1Meg
1206
SMD film res. - 1/4 W - 1% - 100 ppm/°C
Vishay
R8
1Meg
1206
SMD film res. - 1/4 W - 1% - 100 ppm/°C
Vishay
R9
62K
0805
SMD film res. - 1/8 W - 1% - 100 ppm/°C
Vishay
R10
27K
0805
SMD film res. - 1/8 W - 1% - 100 ppm/°C
Vishay
R11
2M2
1206
SMD film res. - 1/4 W - 1% - 100 ppm/°C
Vishay
R12
2M2
1206
SMD film res. - 1/4 W - 1% - 100 ppm/°C
Vishay
R13
8K2
1206
SMD film res. - 1/4 W - 1% - 100 ppm/°C
Vishay
R14
51K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R15
56K
1206
SMD film res. - 1/4 W - 1% - 100 ppm/°C
Vishay
R16
N.M.
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
R17
2M2
1206
SMD film res. - 1/4 W - 1% - 100 ppm/°C
Vishay
R18
82K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R19
56K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R20
10K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R21
39R
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R22
0R47
PTH
SFR25 axial stand. film res. - 0.4 W 5% - 250 ppm/°C
Vishay
R23
0R68
PTH
SFR25 axial stand. film res. - 0.4 W 5% - 250 ppm/°C
Vishay
R24
1Meg
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R25
56R
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R26
1Meg
0805
SMD film res. - 1/8 W - 1% - 100 ppm/°C
Vishay
R27
470R
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
Vishay
R28
33K
0805
SMD film res.- 1/8 W - 1% - 100 ppm/°C
Vishay
R29
1K0
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
Vishay
R30
10R
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R31
33K
0805
SMD film res. - 1/8 W - 1% - 100 ppm/°C
Vishay
R32
470R
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R33
N.M.
0805
Not mounted
R34
8K2
1206
SMD film res. - 1/4 W - 1% - 100 ppm/°C
Vishay
R35
180K
0805
SMD film res. - 1/8 W - 1% - 100 ppm/°C
Vishay
R36
N.M.
0805
Not mounted
R37
220K
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
32/40
Doc ID 022603 Rev 1
Vishay
AN4026
Table 4.
Bill of material
Bill of material (continued)
Part type/
Case style
part value
/package
R38
56R
R39
Des.
Description
Supplier
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
N.M.
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
R40
0R0
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
R41
Jumper
WIRE
Shorted by wire
R42
12K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R43
10R
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R44
15K
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
Vishay
R45
N.M.
0805
Not mounted
R46
100K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R47
1K5
0805
SMD film res.- 1/8 W - 5% - 250 ppm/°C
Vishay
R48
180K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R49
39K
0805
SMD film res. - 1/8 W - 1% - 100 ppm/°C
Vishay
R50
6K2
0805
SMD film res. - 1/8 W - 1% - 100 ppm/°C
Vishay
R51
120K
0805
SMD film res. - 1/8 W - 1% - 100 ppm/°C
Vishay
R52
12K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R53
2K2
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R54
0R0
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R55
2K7
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R56
18K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R57
47R
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R58
100K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R59
100K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R60
10K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R61
N.M.
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
R62
4K7
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R65
120K
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
Vishay
R66
2K2
1206
SMD film res. - 1/4 W - 5% - 250 ppm/°C
Vishay
R67
N.M.
0805
Not mounted
R68
N.M.
1206
Not mounted
R69
4K7
0805
SMD film res. - 1/8 W - 5% - 250 ppm/°C
T1
1860.0076
DWG - ETD34
U1
L6563H
SO-16
High-voltage startup TM PFC controller
STMicroelectronics
U2
L6699D
SO-16
Improved HV resonant controller
STMicroelectronics
U3
SFH617A-4
DIP-4 - 10.16 mm
Resonant power transformer
Optocoupler
Doc ID 022603 Rev 1
Vishay
Vishay
MAGNETICA
INFINEON
33/40
Bill of material
Table 4.
AN4026
Bill of material (continued)
Part type/
Case style
part value
/package
U4
TL431AIZ
TO-92
Z1
PCB REV. 1.0
Des.
Note:
34/40
Description
Supplier
Programmable shunt voltage reference
STMicroelectronics
Some preliminary demonstration boards use the PCB of the EVL6599A-90WADP reworked
for the L6699. On these boards the silkscreen reports the original reference designators of
these components as D10 and D11, like in the circuit diagram of the EVL6599A-90WADP.
The name of the boards reported on the PCB silkscreen top side is "90 W adapter with the
L6563H and L6599A Rev. 1.0". More recent demonstration boards have updated the
reference designators and the silkscreen of D10 and D11 has been updated to R63 and R64
respectively. The name of these more recent boards reported on the PCB top side is "90 W
adapter with the L6563H and L6699 Rev. 1.0". There isn't any circuit difference between the
two boards.
Doc ID 022603 Rev 1
AN4026
PFC coil specifications
8
PFC coil specifications
8.1
General description and characteristics
8.2
8.3
●
Application type: consumer, home appliance
●
Transformer type: open
●
Coil former: vertical type, 6+6 pins
●
Max. temperature rise: 45 ºC
●
Max. operating ambient temperature: 60 ºC
●
Mains insulation: n.a.
●
Unit finishing: varnished.
Electrical characteristics
●
Converter topology: boost, transition mode
●
Core type: PQ26/20-PC44 or equivalent
●
Min. operating frequency: 40 kHz
●
Typical operating frequency: 120 kHz
●
Primary inductance: 520 µH ± 15% at 1 kHz - 0.25 V(a).
Electrical diagram and winding characteristics
Figure 30. PFC coil electrical diagram
Table 5.
PFC coil winding data
Pins
Windings
DC resistance
Number of turns
Wire type
11-3
AUX
125 mΩ
5.5
φ 0.28 mm – G2
5-9
Primary
267 mΩ
57.5
30Xφ 0.1 mm- G1
a. Measured between pins #5 and #9.
Doc ID 022603 Rev 1
35/40
PFC coil specifications
8.4
AN4026
Mechanical aspect and pin numbering
●
Maximum height from PCB: 22 mm
●
Coil former type: vertical, 6+6 pins (pins #1, 2, 4, 6, 7, 10, 12 are removed)
●
Pin distance: 3.81 mm
●
Row distance: 25 mm
●
External copper shield: not insulated, wound around the ferrite core and including the
coil former. Height is 8 mm. Connected to pin #3 by a soldered solid wire.
Figure 31. PFC coil mechanical aspect
Manufacturer
36/40
●
MAGNETICA - Italy
●
Inductor p/n: 1974.0004.
Doc ID 022603 Rev 1
AN4026
Transformer specifications
9
Transformer specifications
9.1
General description and characteristics
9.2
9.3
●
Application type: consumer, home appliance
●
Transformer type: open
●
Coil former: horizontal type, 7+7 pins, two slots
●
Max. temperature rise: 45 ºC
●
Max. operating ambient temperature: 60 ºC
●
Mains insulation: acc. to EN60065.
Electrical characteristics
●
Converter topology: half bridge, resonant
●
Core type: ETD34-PC44 or equivalent
●
Min. operating frequency: 60 kHz
●
Typical operating frequency: 90 kHz
●
Primary inductance: 2.00 mH ± 10% at 1 kHz - 0.25 V(b)
●
Leakage inductance: 300 µH at 100 kHz - 0.25 V(c).
Electrical diagram and winding characteristics
Figure 32. Transformer electrical diagram
b. Measured between pins 2-4.
c. Measured between pins 2-4 with only one secondary winding shorted.
Doc ID 022603 Rev 1
37/40
Transformer specifications
Table 6.
AN4026
Transformer winding data
Pins
Winding
DC resistance
Number of turns
Wire type
2-4
Primary
422 mΩ
61
20x φ 0.1 mm – G1
8 mΩ
6
60x φ 0.1 mm – G1
13-12
SEC - A
(1)
10-9
SEC – B
8 mΩ
6
60x φ 0.1 mm – G1
5-6
AUX(2)
111 mΩ
4
φ 0.28 mm – G2
1. Secondary windings A and B are in parallel.
2. Aux winding is wound on top of primary winding.
9.4
Mechanical aspect and pin numbering
●
Maximum height from PCB: 30 mm
●
Coil former type: horizontal, 7+7 pins (pins #1 and 7 are removed)
●
Pin distance: 5.08 mm
●
Row distance: 25.4 mm
Figure 33. Transformer overall drawing
Manufacturer
38/40
●
MAGNETICA - Italy
●
Transformer p/n: 1860.0076.
Doc ID 022603 Rev 1
AN4026
10
Revision history
Revision history
Table 7.
Document revision history
Date
Revision
23-Jul-2012
1
Changes
Initial release.
Doc ID 022603 Rev 1
39/40
AN4026
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40/40
Doc ID 022603 Rev 1