Sample & Buy Product Folder Technical Documents Support & Community Tools & Software LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 LM46001 SIMPLE SWITCHER® 3.5 V to 60 V 1 A Synchronous Step-Down Voltage Converter 1 Features 3 Description • • • • • The LM46001 SIMPLE SWITCHER® regulator is an easy to use synchronous step-down DC-DC converter capable of driving up to 1 A of load current from an input voltage ranging from 3.5 V to 60 V. The LM46001 provides exceptional efficiency, output accuracy and drop-out voltage in a very small solution size. An extended family is available in 0.5 A and 2.0 A load current options in pin-to-pin compatible packages. Peak current mode control is employed to achieve simple control loop compensation and cycle-by-cycle current limiting. Optional features such as programmable switching frequency, synchronization, power-good flag, precision enable, internal soft-start, extendable softstart, and tracking provide a flexible and easy to use platform for a wide range of applications. Discontinuous conduction and automatic frequency reduction at light loads improve light load efficiency. The family requires few external components and pin arrangement allows simple, optimum PCB layout. Protection features include thermal shutdown, VCC under-voltage lockout, cycle-by-cycle current limit, and output short circuit protection. The LM46001 device is available in the HTSSOP / PWP 16 leaded package (5.1 mm x 6.6 mm x 1.2 mm) with 0.65 mm lead pitch. Pin to pin compatible with LM46000, LM46002, LM43603, LM43602, LM43601, and LM43600. 1 • • • • • • • • • • • 24 µA Quiescent Current in Regulation High Efficiency at Light Load (DCM and PFM) Meets EN55022/CISPR 22 EMI standards Integrated Synchronous Rectification Adjustable Switching Frequency: 200 kHz to 2.2 MHz (500 kHz default) Frequency Synchronization to External Clock Internal Compensation Stable with Many Combinations of Ceramic, Polymer, Tantalum, and Aluminum Capacitors Power-Good Flag Soft-Start into Pre-Biased Load Internal Soft-Start: 4.1 ms Extendable Soft-Start Time by External Capacitor Output Voltage Tracking Capability Precision Enable to Program System UVLO Output Short Circuit Protection with Hiccup Mode Over Temperature Thermal Shutdown Protection 2 Applications • • • • • • Industrial Power Supplies Telecommunications Systems Sub-AM Band 12 V and 24 V Automotive Commercial Vehicle Power Supplies General Purpose Wide VIN Regulation High Efficiency Point-Of-Load Regulation Device Information(1) PART NUMBER PACKAGE BODY SIZE (NOM) LM46001 HTSSOP (16) 5.1 mm x 6.6 mm (1) For all available packages, see the orderable addendum at the end of the datasheet. Simplified Schematic Radiated Emission Graph VOUT = 3.3 V, VIN = 24 V, FS= 500 kHz, IOUT = 1 A dBuV 80 L VIN VIN CIN SW LM46001 ENABLE CBOOT AGND Horizontal Polarization 50 EN 55022 Class B Limit 40 30 CBIAS SS/TRK SYNC Vertical Polarization 70 60 COUT CBOOT BIAS PGOOD RT VOUT 20 CFF RFBT 10 Evaluation Board Emissions 30 FB VCC CVCC 100 Frequency (MHz) 1000 RFBB PGND 1 An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications, intellectual property matters and other important disclaimers. PRODUCTION DATA. LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Table of Contents 1 2 3 4 5 6 7 Features .................................................................. Applications ........................................................... Description ............................................................. Revision History..................................................... Pin Configuration and Functions ......................... Specifications......................................................... 1 1 1 2 3 4 6.1 6.2 6.3 6.4 6.5 6.6 6.7 6.8 4 4 4 5 5 6 7 8 Absolute Maximum Ratings ...................................... Handling Ratings ...................................................... Recommended Operating Conditions....................... Thermal Information .................................................. Electrical Characteristics........................................... Timing Requirements ................................................ Switching Characteristics .......................................... Typical Characteristics .............................................. Detailed Description ............................................ 14 7.1 Overview ................................................................. 14 7.2 Functional Block Diagram ....................................... 14 7.3 Feature Description................................................. 15 7.4 Device Functional Modes........................................ 24 8 Applications and Implementation ...................... 25 8.1 Application Information............................................ 25 8.2 Typical Applications ................................................ 25 9 Power Supply Recommendations...................... 41 10 Layout................................................................... 41 10.1 Layout Guidelines ................................................. 41 10.2 Layout Example .................................................... 44 11 Device and Documentation Support ................. 45 11.1 Trademarks ........................................................... 45 11.2 Electrostatic Discharge Caution ............................ 45 11.3 Glossary ................................................................ 45 12 Mechanical, Packaging, and Orderable Information ........................................................... 45 4 Revision History Changes from Original (June 2014) to Revision A • 2 Page Changed device from Product Preview to Production Data .................................................................................................. 1 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 5 Pin Configuration and Functions 16-Pin HTSSOP (PWP) Top View SW 1 SW 2 16 15 PGND PGND CBOOT VCC 3 4 14 VIN 13 BIAS 5 VIN 12 SYNC EN 6 11 RT PGOOD SS/TRK 7 8 10 AGND PAD 9 FB Pin Functions PIN DESCRIPTION NAME NUMBER SW 1,2 CBOOT 3 Boot-strap capacitor connection for high-side driver. Connect a high quality 470 nF capacitor from CBOOT to SW. VCC 4 Internal bias supply output for bypassing. Connect bypass capacitor from this pin to AGND. Do not connect external load to this pin. Never short this pin to ground during operation. BIAS 5 Optional internal LDO supply input. To improve efficiency, it is recommended to tie to VOUT when 3.3 V ≤ VOUT ≤ 28 V, or tie to an external 3.3 V or 5 V rail if available. When used, place a bypass capacitor (1 to 10 µF) from this pin to ground. Tie to ground if not used (VOUT < 3.3 V). Do not float. SYNC 6 Clock input to synchronize switching action to an external clock. Use proper high speed termination to prevent ringing. Connect to ground if not used. Do not float. RT 7 Connect a resistor RT from this pin to AGND to program switching frequency. Leave floating for 500 kHz default switching frequency. PGOOD 8 Open drain output for power-good flag. Use a 10 kΩ to 100 kΩ pull-up resistor to logic rail or other DC voltage no higher than 12 V. FB 9 Feedback sense input pin. Connect to the midpoint of feedback divider to set VOUT. Do not short this pin to ground during operation. AGND 10 Analog ground pin. Ground reference for internal references and logic. Connect to system ground. SS/TRK 11 Soft-start control pin. Leave floating for internal soft-start. Connect to a capacitor to extend soft start time. Connect to external voltage ramp for tracking. EN 12 Enable input to the LM46001: High = ON and LOW = OFF. Connect to VIN, or to VIN through resistor divider, or to an external voltage or logic source. Do not float. VIN 13,14 Supply input pins to internal LDO and high side power FET. Connect to power supply and bypass capacitors CIN. Path from VIN pin to high frequency bypass CIN and PGND must be as short as possible. PGND 15,16 Power ground pins, connected internally to the low side power FET. Connect to system ground, PAD, AGND, ground pins of CIN and COUT. Path to CIN must be as short as possible. PAD 17 Low impedance connection to AGND. Connect to PGND on PCB. Major heat dissipation path of the die. Must be used for heat sinking to ground plane on PCB. Switching output of the regulator. Internally connected to both power MOSFETs. Connect to power inductor. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 3 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com 6 Specifications 6.1 Absolute Maximum Ratings (1) Over operating free-air temperature range (unless otherwise noted) Input Voltages Output Voltages (1) PARAMETER MIN MAX VIN to PGND -0.3 65 UNIT EN to PGND -0.3 VIN + 0.3 FB, RT, SS/TRK to AGND -0.3 3.6 PGOOD to AGND -0.3 15 SYNC to AGND -0.3 5.5 BIAS to AGND -0.3 30 AGND to PGND -0.3 0.3 SW to PGND -0.3 VIN + 0.3 SW to PGND less than 10ns Transients -3.5 65 CBOOT to SW -0.3 5.5 VCC to AGND -0.3 3.6 V V Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. 6.2 Handling Ratings Tstg Storage temperature range V(ESD) (1) (2) Electrostatic discharge MIN MAX UNIT -65 +150 °C Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins (1) 2000 Charged device model (CDM), per JEDEC specification JESD22-C101, all pins (2) 500 V JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. 6.3 Recommended Operating Conditions (1) Over operating free-air temperature range (unless otherwise noted) Input Voltages PARAMETER MIN MAX VIN to PGND 3.5 60 EN -0.3 VIN FB -0.3 1.1 PGOOD -0.3 12 BIAS input not used -0.3 0.3 BIAS input used 3.3 28 AGND to PGND -0.1 0.1 1.0 28 Output Voltage Adjustable VOUT Range Output Current IOUT Temperature Operating junction temperature range, TJ (1) 4 UNIT V V 0 1 A -40 +125 °C Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee specific performance limits. For guaranteed specifications, see Electrical Characteristics. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 6.4 Thermal Information THERMAL METRIC HTSSOP (16 PINS) (1) RθJA Junction-to-ambient thermal resistance RθJC(top) Junction-to-case (top) thermal resistance 26.9 RθJB Junction-to-board thermal resistance 21.7 ψJT Junction-to-top characterization parameter 0.8 ψJB Junction-to-board characterization parameter 21.5 RθJC(bot) Junction-to-case (bottom) thermal resistance 2.3 (1) (2) 39.9 UNIT (2) °C/W The package thermal impedance is calculated in accordance with JESD 51-7 standard with a 4-layer board and 1 W power dissipation. RθJA is highly related to PCB layout and heat sinking. Please refer to Figure 101 for measured RθJA vs PCB area from a 2-layer board and a 4-layer board. 6.5 Electrical Characteristics Limits apply over the recommended operating junction temperature (TJ) range of -40°C to +125°C, unless otherwise stated. Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated, the following conditions apply: VIN = 24 V, VOUT = 3.3 V, FS = 500 kHz. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT SUPPLY VOLTAGE (VIN PINS) VIN-MIN-ST Minimum input voltage for startup ISHDN Shutdown quiescent current VEN = 0 V IQ-NONSW Operating quiescent current (nonswitching) from VIN IBIAS-NONSW IQ-SW 3.8 V 2.3 5 µA VEN = 3.3 V VFB = 1.5 V VBIAS = 3.4 V external 7 13 µA Operating quiescent current (nonswitching) from external VBIAS VEN = 3.3 V VFB = 1.5 V VBIAS = 3.4 V external 85 140 µA Operating quiescent current (switching) VEN = 3.3 V IOUT = 0 A RT = open VBIAS = VOUT = 3.3 V RFBT = 1.0 Meg 24 µA ENABLE (EN PIN) VEN-VCC-H Voltage level to enable the internal LDO VEN-VCC-L Voltage level to disable the internal LDO VENABLE low level VENABLE high level VEN-VOUT-H Precision enable level for switching and regulator output VENABLE high level VEN-VOUT-HYS Hysteresis voltage between VOUT precision enable and disable thresholds VENABLE hysteresis ILKG-EN Enable input leakage current VEN = 3.3 V 1.2 2.00 V 2.1 0.4 V 2.42 V -305 0.8 mV 1.75 µA INTERNAL LDO (VCC PIN AND BIAS PIN) VCC Internal LDO output voltage VCC VCC-UVLO Under voltage lock out (UVLO) thresholds for VCC VBIAS-ON Internal LDO input change over threshold to BIAS VIN ≥ 3.8 V 3.3 V VCC rising threshold 3.14 V Hysteresis voltage between rising and falling thresholds -567 mV VBIAS rising threshold 2.96 Hysteresis voltage between rising and falling thresholds 3.2 -71 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 V mV 5 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Electrical Characteristics (continued) Limits apply over the recommended operating junction temperature (TJ) range of -40°C to +125°C, unless otherwise stated. Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated, the following conditions apply: VIN = 24 V, VOUT = 3.3 V, FS = 500 kHz. SYMBOL PARAMETER CONDITIONS MIN TYP MAX TJ = 25 ºC 1.009 1.016 1.023 TJ = -40 ºC to 85 ºC 0.999 1.016 1.031 TJ = -40 ºC to 125 ºC 0.999 UNIT VOLTAGE REFERENCE (FB PIN) VFB Feedback voltage ILKG-FB Input leakage current at FB pin V 1.016 1.039 FB = 1.016 V 0.2 65 Shutdown threshold 160 ºC Recovery threshold 150 ºC nA THERMAL SHUTDOWN TSD (1) Thermal shutdown CURRENT LIMIT AND HICCUP IHS-LIMIT Peak inductor current limit 2.07 2.45 2.71 A ILS-LIMIT Valley inductor current limit 0.94 1.1 1.25 A 1.17 2.2 2.85 µA SOFT START (SS/TRK PIN) ISSC Soft-start charge current RSSD Soft-start discharge resistance UVLO, TSD, OCP, or EN = 0 V 16 kΩ POWER GOOD (PGOOD PIN) VPGOOD-HIGH Power-good flag over voltage tripping threshold % of FB voltage VPGOOD-LOW Power-good flag under voltage tripping threshold % of FB voltage VPGOOD-HYS Power-good flag recovery hysteresis % of FB voltage RPGOOD PGOOD pin pull down resistance when power bad VEN = 3.3 V 40 125 VEN = 0 V 60 150 MOSFETS 110% 83% 113% 90% 6% Ω (2) RDS-ON-HS High-side MOSFET ON-resistance IOUT = 1 A VBIAS = VOUT = 3.3 V 419 mΩ RDS-ON-LS Low-side MOSFET ON-resistance IOUT = 1 A VBIAS = VOUT = 3.3 V 231 mΩ (1) (2) Guaranteed by design. Not production tested Measured at package pins 6.6 Timing Requirements Typical values represent the most likely parametric norm at TJ = 25°C. MIN TYP MAX UNIT CURRENT LIMIT AND HICCUP NOC Hiccup wait cycles when LS current limit tripped 32 Cycles TOC Hiccup retry delay time 5.5 ms 4.1 ms TPGOOD-RISE Power-good flag rising transition deglitch delay 220 µs TPGOOD-FALL Power-good flag falling transition deglitch delay 220 µs SOFT START (SS/TRK PIN) TSS Internal soft-start time when SS pin open circuit POWER GOOD (PGOOD PIN) 6 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 6.7 Switching Characteristics Limits apply over the recommended operating junction temperature (TJ) range of -40°C to +125°C, unless otherwise stated. Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated, the following conditions apply: VIN = 24 V, VOUT = 3.3 V, FS = 500 kHz. PARAMETER TEST CONDITIONS MIN TYP MAX UNIT SW (SW PIN) tON-MIN (1) Minimum high side MOSFET ON time 125 165 ns tOFF-MIN (1) Minimum high side MOSFET OFF time 200 250 ns 500 570 kHz OSCILLATOR (SW PINS AND SYNC PIN) FOSC- Oscillator default frequency RT pin open circuit 445 DEFAULT Minimum adjustable frequency FADJ Maximum adjustable frequency With 1% resistors at RT pin Frequency adjust accuracy 200 kHz 2200 kHz 10% VSYNC-HIGH Sync clock high level threshold 2 V VSYNC-LOW Sync clock low level threshold DSYNC-MAX Sync clock maximum duty cycle 90% DSYNC-MIN Sync clock minimum duty cycle 10% TSYNC-MIN Mininum sync clock ON and OFF time (1) 0.4 80 V ns Guaranteed by design. Not production tested. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 7 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com 6.8 Typical Characteristics 100 100 90 90 80 80 70 70 Efficiency (%) Efficiency (%) Unless otherwise specified, VIN = 24 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. Please refer to Application Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations. 60 50 VIN = 8V VIN = 12V VIN = 18V VIN = 24V VIN = 28V VIN = 36V 40 30 20 10 0 0.001 0.010 0.100 VOUT = 3.3 V 50 VIN = 18V 30 VIN = 24V 20 VIN = 28V 10 VIN = 36V VOUT = 5 V 90 90 80 80 70 70 60 50 40 30 10 0 0.001 VIN = 8V VIN = 24V VIN = 42V VIN = 12V VIN = 28V VIN = 48V 0.010 0.100 1.000 C003 FS = 500 kHz 60 50 40 30 VIN = 12V VIN = 18V VIN = 24V VIN = 28V 20 VIN = 18V VIN = 36V VIN = 60V 10 0 0.001 1.000 Load Current (A) VOUT = 5 V 0.100 Figure 2. Efficiency 100 Efficiency (%) Efficiency (%) Figure 1. Efficiency 100 20 VIN = 42V 0.010 Load Current (A) C002 FS = 500 kHz VIN = 12V 40 0 0.001 1.000 Load Current (A) 60 FS = 200 kHz 0.010 0.100 1.000 Load Current (A) C004 VOUT = 5 V Figure 3. Efficiency C005 FS = 1 MHz Figure 4. Efficiency 100 120 90 100 80 Efficiency (%) Efficiency (%) 70 60 50 VIN = 24V 40 VIN = 28V 30 VIN = 36V 20 VIN = 42V 10 VIN = 48V 0 0.001 0.100 Load Current (A) VOUT = 12 V FS = 500 kHz 60 VIN = 42V VIN = 48V VIN = 60V 1.000 0 0.001 0.010 0.100 Load Current (A) C007 VOUT = 24 V Figure 5. Efficiency 8 VIN = 36V 40 20 VIN = 60V 0.010 80 1.000 C008 FS = 500 kHz Figure 6. Efficiency Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Typical Characteristics (continued) Unless otherwise specified, VIN = 24 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. Please refer to Application Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations. 3.40 5.20 3.38 5.15 3.36 5.10 3.32 Vout (V) Vout (V) 3.34 3.30 3.28 3.24 3.22 VIN = 8V VIN = 12V 4.90 VIN = 18V VIN = 24V 4.85 VIN = 28V VIN = 36V 0.010 0.100 4.80 0.001 1.000 Load Current (A) VOUT = 3.3 V 5.00 4.95 3.26 3.20 0.001 5.05 VIN = 18V VIN = 24V VIN = 28V VIN = 36V VIN = 42V 0.010 0.100 1.000 Load Current (A) C012 FS = 500 kHz VOUT = 5 V Figure 7. VOUT Regulation C013 FS = 500 kHz Figure 8. VOUT Regulation 5.15 5.20 5.10 5.15 5.10 Vout (V) 5.05 Vout (V) VIN = 12V 5.00 4.95 5.05 5.00 4.95 4.90 4.85 4.80 0.001 VIN = 8V VIN = 12V VIN = 18V 4.90 VIN = 24V VIN = 28V VIN = 36V 4.85 VIN = 42V VIN = 48V VIN = 60V 0.010 0.100 Load Current (A) VOUT = 5 V 4.80 0.001 1.000 FS = 200 kHz VOUT = 5 V 0.100 12.4 24.8 12.3 24.6 12.2 24.4 12.1 24.2 12.0 11.9 11.8 11.6 VIN = 24V VIN = 28V VIN = 36V VIN = 42V VIN = 48V VIN = 60V 0.010 0.100 Load Current (A) FS = 500 kHz C015 FS = 1 MHz 24.0 23.8 23.6 11.7 1.000 Figure 10. VOUT Regulation 25.0 Vout (V) Vout (V) Figure 9. VOUT Regulation VOUT = 12 V 0.010 Load Current (A) C014 12.5 11.5 0.001 VIN = 12V VIN = 18V VIN = 24V VIN = 28V VIN = 36V VIN = 42V VIN = 48V VIN = 60V 23.4 23.2 1.000 23.0 0.001 0.010 0.100 Load Current (A) C017 VOUT = 24 V Figure 11. VOUT Regulation 1.000 C018 FS = 500 kHz Figure 12. VOUT Regulation Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 9 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Typical Characteristics (continued) Unless otherwise specified, VIN = 24 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. Please refer to Application Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations. 5.2 3.50 3.40 5.0 3.30 4.8 3.10 VOUT (V) VOUT (V) 3.20 3.00 2.90 Load = 0.25A 2.80 4.6 Load = 0.25A 4.4 Load = 0.5A Load = 0.5A 2.70 4.2 Load = 0.75A 2.60 2.50 3.5 4.0 4.0 4.5 5.0 VIN (V) VOUT = 3.3 V Load = 0.75A Load = 1A Load = 1A 5.0 6.0 6.5 VIN (V) FS = 500 kHz VOUT = 5 V Figure 13. Drop-Out Curve C023 FS = 500 kHz Figure 14. Drop-Out Curve 5.2 5.2 5.0 5.0 4.8 4.8 4.6 Load = 0.25A VOUT (V) VOUT (V) 5.5 C022 4.4 4.6 Load = 0.25A 4.4 Load = 0.5A Load = 0.5A Load = 0.75A 4.2 Load = 0.75A 4.2 Load = 1A Load = 1A 4.0 4.0 5.0 5.5 6.0 6.5 VIN (V) VOUT = 5 V 5.0 5.5 6.0 6.5 VIN (V) C024 FS = 200 kHz VOUT = 5 V Figure 15. Drop-Out Curve C025 FS = 1 MHz Figure 16. Drop-Out Curve 12.4 24.5 12.2 24.0 VOUT (V) VOUT (V) 12.0 11.8 11.6 Load = 0.25A 11.4 Load = 0.5A 11.2 Load = 0.75A 23.5 23.0 Load = 0.25A Load = 0.5A 22.5 Load = 0.75A Load = 1A Load = 1A 11.0 12.0 12.5 13.0 13.5 VIN (V) VOUT = 12 V FS = 500 kHz 14.0 22.0 24.0 VOUT = 24 V Figure 17. Drop-Out Curve 10 24.5 25.0 25.5 26.0 VIN (V) C027 26.5 27.0 C028 FS = 500 kHz Figure 18. Drop-Out Curve Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Typical Characteristics (continued) Unless otherwise specified, VIN = 24 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. Please refer to Application Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations. 1000000 Frequency (Hz) Frequency (Hz) 1000000 100000 Load = 0.01 A Load = 0.1 A Load = 0.5 A Load = 1 A 10000 3.4 3.6 3.8 4.0 4.2 4.4 4.6 4.8 VOUT = 3.3 V Load = 0.01 A Load = 0.1 A Load = 0.5 A Load = 1 A 10000 5.0 VIN (V) 100000 5.0 5.2 5.4 FS = 500 kHz VOUT = 5 V Figure 19. Switching Frequency vs VIN in Drop-Out Operation dBuV 80 Vertical Polarization 70 Horizontal Polarization 60 5.6 5.8 6.0 6.2 6.4 6.6 6.8 7.0 VIN (V) C001 C001 FS = 1 MHz Figure 20. Switching Frequency vs VIN in Drop-Out Operation dBuV 80 Vertical Polarization 70 Horizontal Polarization 60 50 50 EN 55022 Class B Limit 40 EN 55022 Class B Limit 40 30 30 20 20 10 10 Evaluation Board Emissions 30 100 Frequency (MHz) Evaluation Board Emissions 1000 VOUT = 3.3 V FS = 500 kHz IOUT = 1 A Measured on the LM46001PWPEVM with default BOM. No input filter used. 30 100 Frequency (MHz) 1000 VOUT = 5 V FS = 500 kHz IOUT = 1 A Measured on the LM46001PWPEVM with L = 27 µH, COUT = 66 µF, CFF = 33 pF. No input filter used. Figure 21. Radiated EMI Curve Figure 22. Radiated EMI Curve Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 11 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Typical Characteristics (continued) Unless otherwise specified, VIN = 24 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. Please refer to Application Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations. dBuV 100 dBuV 100 90 90 80 80 70 70 60 Quasi Peak Limit 60 Quasi Peak Limit 50 Average Limit Average Limit 40 50 40 30 30 20 20 10 10 Measured Peak Emissions 0.15 10 1 Frequency (MHz) 30 VOUT = 3.3 V FS = 500 kHz IOUT = 1 A Measured on the LM46001PWPEVM with default BOM. EVM input filter: Lin = 1 µH Cd = 47 µF CIN4 = 68 µF Measured Peak Emissions 0.15 10 1 Frequency (MHz) 30 VOUT = 5 V FS = 500 kHz IOUT = 1 A Measured on the LM46001PWPEVM with L = 27 µH, COUT = 66 µF, CFF = 33 pF. EVM input filter Lin = 1 µH Cd = 47 µF CIN4 = 68 µF Figure 23. Conducted EMI Curve Figure 24. Conducted EMI Curve 4 700 3.5 Shutdown Current (A) 800 Rdson (mohm) 600 500 400 300 200 HS 100 3 2.5 2 1.5 1 VIN = 12V 0.5 LS VIN = 24V 0 0 -50 0 50 100 Temperature (C) 150 -50 0 50 100 150 Temperature (C) C001 Figure 25. High-Side and Low-side On Resistance vs Junction Temperature C001 Figure 26. Shutdown Current vs Junction Temperature 2.5 1.4 Enable Threshold (V) 1.5 EN-VOUT Rising TH EN-VOUT Falling TH EN-VCC Rising TH EN-VCC Falling TH 1 0.5 EN Leakage Current (A) 1.2 2 1 0.8 0.6 0.4 0.2 0 VEN = 3.3V 0 -50 0 50 Temperature (C) 100 150 -50 C001 Figure 27. Enable Threshold vs Junction Temperature 12 Submit Documentation Feedback 0 50 100 Temperature (C) 150 C001 Figure 28. Enable Leakage Current vs Junction Temperature Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Typical Characteristics (continued) 120% 1.030 115% 1.025 110% 1.020 105% VFB (V) PGOOD Threshold / VOUT (%) Unless otherwise specified, VIN = 24 V, VOUT = 3.3 V, FS = 500 kHz, L = 18 µH, COUT = 100 µF, CFF = 33 pF. Please refer to Application Performance Curves for Bill of Materials (BOM) for other VOUT and FS combinations. 100% 95% 1.015 1.010 1.005 90% OVP Trip Level OVP Recover Level UVP Recover Level UVP Trip Level 85% 80% 75% -50 0 50 100 1.000 VIN = 12V 0.995 VIN = 24V 0.990 150 Temperature (C) -50 Figure 29. PGOOD Threshold vs Junction Temperature 50 100 150 Temperature (C) C001 Figure 30. Feedback Voltage vs Junction Temperature 3.0 70 2.5 60 50 IQ (A) 2.0 Current (A) 0 C001 1.5 1.0 40 30 20 0.5 10 IL Peak Limit IL Valley Limit 0.0 0 -50 0 50 Temperature (C) VIN = 24 V VOUT = 3.3 V 100 150 0 10 FS = 500 kHz Figure 31. Peak and Valley Current Limits vs Junction Temperature 20 30 40 50 60 VIN (V) C001 VOUT = 3.3 V FS = 500 kHz C001 IOUT = 0 A Figure 32. Operating IQ vs VIN with BIAS Connected to VOUT Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 13 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com 7 Detailed Description 7.1 Overview The LM46001 SIMPLE SWITCHER® regulator is an easy to use synchronous step-down DC-DC converter that operates from 3.5 V to 60 V supply voltage. It is capable of delivering up to 1 A DC load current with exceptional efficiency and thermal performance in a very small solution size. An extended family is available in 0.5 A and 2.0 A load options in pin-to-pin compatible packages. The LM46001 employs fixed frequency peak current mode control with Discontinuous Conduction Mode (DCM) and Pulse Frequency Modulation (PFM) mode at light load to achieve high efficiency across the load range. The device is internally compensated, which reduces design time, and requires fewer external components. The switching frequency is programmable from 200 kHz to 2.2 MHz by an external resistor, RT. It defaults at 500 kHz without RT. The LM46001 is also capable of synchronization to an external clock within the 200 kHz to 2.2 MHz frequency range. The wide switching frequency range allows the device to be optimized to fit small board space at higher frequency, or high efficient power conversion at lower frequency. Optional features are included for more comprehensive system requirements, including power-good (PGOOD) flag, precision enable, synchronization to external clock, extendable soft-start time, and output voltage tracking. These features provide a flexible and easy to use platform for a wide range of applications. Protection features include over temperature shutdown, VCC under-voltage lockout (UVLO), cycle-by-cycle current limit, and shortcircuit protection with hiccup mode. The family requires few external components and the pin arrangement was designed for simple, optimum PCB layout. The LM46001 device is available in the HTSSOP / PWP 16 pin leaded package (5.1 mm x 6.6 mm x 1.2 mm) with 0.65 mm lead pitch. 7.2 Functional Block Diagram ENABLE VCC Enable Internal SS ISSC BIAS VCC LDO VIN Precision Enable SS/TRK CBOOT HS I Sense + EA REF RC + ± +± TSD UVLO CC PGOOD AGND OV/UV Detector FB SW PWM CONTROL LOGIC PFM Detector PGood Slope Comp Freq Foldback Zero Cross HICCUP Detector Oscillator LS I Sense FB PGood SYNC 14 PGND RT Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 7.3 Feature Description 7.3.1 Fixed Frequency Peak Current Mode Controlled Step-Down Regulator The following operating description of the LM46001 will refer to the Functional Block Diagram and to the waveforms in Figure 33. The LM46001 is a step-down Buck regulator with both high-side (HS) switch and lowside (LS) switch (synchronous rectifier) integrated. The LM46001 supplies a regulated output voltage by turning on the HS and LS NMOS switches with controlled ON time. During the HS switch ON time, the SW pin voltage VSW swings up to approximately VIN, and the inductor current IL increases with a linear slope (VIN - VOUT) / L. When the HS switch is turned off by the control logic, the LS switch is turned on after a anti-shoot-through dead time. Inductor current discharges through the LS switch with a slope of -VOUT / L. The control parameter of Buck converters are defined as Duty Cycle D = tON / TSW, where tON is the HS switch ON time and TSW is the switching period. The regulator control loop maintains a constant output voltage by adjusting the duty cycle D. In an ideal Buck converter, where losses are ignored, D is proportional to the output voltage and inversely proportional to the input voltage: D = VOUT / VIN. VSW D = tON / TSW SW Voltage VIN tOFF tON 0 t -VD1 Inductor Current iL TSW ILPK IOUT ûiL t 0 Figure 33. SW Node and Inductor Current Waveforms in Continuous Conduction Mode (CCM) The LM46001 synchronous Buck converter employs peak current mode control topology. A voltage feedback loop is used to get accurate DC voltage regulation by adjusting the peak current command based on voltage offset. The peak inductor current is sensed from the HS switch and compared to the peak current to control the ON time of the HS switch. The voltage feedback loop is internally compensated, which allows for fewer external components, makes it easy to design, and provides stable operation with almost any combination of output capacitors. The regulator operates with fixed switching frequency in Continuous Conduction Mode (CCM) and Discontinuous Conduction Mode (DCM). At very light load, the LM46001 will operate in PFM to maintain high efficiency and the switching frequency will decrease with reduced load current. 7.3.2 Light Load Operation DCM operation is employed in the LM46001 when the inductor current valley reaches zero. The LM46001 will be in DCM when load current is less than half of the peak-to-peak inductor current ripple in CCM. In DCM, the LS switch is turned off when the inductor current reaches zero. Switching loss is reduced by turning off the LS FET at zero current and the conduction loss is lowered by not allowing negative current conduction. Power conversion efficiency is higher in DCM than CCM under the same conditions. In DCM, the HS switch ON time will reduce with lower load current. When either the minimum HS switch ON time (TON-MIN) or the minimum peak inductor current (IPEAK-MIN) is reached, the switching frequency will decrease to maintain regulation. At this point, the LM46001 operates in PFM. In PFM, switching frequency is decreased by the control loop when load current reduces to maintain output voltage regulation. Switching loss is further reduced in PFM operation due to less frequent switching actions. Figure 34 shows an example of switching frequency decreases with decreased load current. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 15 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Feature Description (continued) Frequency (Hz) 1000000 100000 VIN = 8 V VIN = 12 V VIN = 18 V VIN = 24 V 10000 0.001 VIN = 36 V 0.01 0.1 Load (A) 1 C001 Figure 34. Switching Frequency Decreases with Lower Load Current in PFM Operation VOUT = 5 V, FS = 1 MHz In PFM operation, a small positive DC offset is required at the output voltage to activate the PFM detector. The lower the frequency in PFM, the more DC offset is needed at VOUT. Please refer to the Typical Characteristics for typical DC offset at very light load. If the DC offset on VOUT is not acceptable for a given application, a static load at output is recommended to reduce or eliminate the offset. Lowering values of the feedback divider RFBT and RFBB can also serve as a static load. In conditions with low VIN and/or high frequency, the LM46001 may not enter PFM mode if the output voltage cannot be charged up to provide the trigger to activate the PFM detector. Once the LM46001 is operating in PFM mode at higher VIN, it will remain in PFM operation when VIN is reduced. 7.3.3 Adjustable Output Voltage The voltage regulation loop in the LM46001 regulates output voltage by maintaining the voltage on FB pin (VFB) to be the same as the internal REF voltage (VREF). A resistor divider pair is needed to program the ratio from output voltage VOUT to VFB. The resistor divider is connected from the VOUT of the LM46001 to ground with the mid-point connecting to the FB pin. VOUT RFBT FB RFBB Figure 35. Output Voltage Setting The voltage reference system produces a precise voltage reference over temperature. The internal REF voltage is 1.016 V typically. To program the output voltage of the LM46001 to be a certain value VOUT, RFBB can be calculated with a selected RFBT by VFB RFBB RFBT VOUT VFB (1) The choice of the RFBT depends on the application. RFBT in the range from 10 kΩ to 100 kΩ is recommended for most applications. A lower RFBT value can be used if static loading is desired to reduce VOUT offset in PFM operation. Lower RFBT will reduce efficiency at very light load. Less static current goes through a larger RFBT and might be more desirable when light load efficiency is critical. But RFBT larger than 1 MΩ is not recommended because it makes the feedback path more susceptible to noise. Larger RFBT value requires more carefully designed feedback path on the PCB. The tolerance and temperature variation of the resistor dividers affect the output voltage regulation. It is recommended to use divider resistors with 1% tolerance or better and temperature coefficient of 100 ppm or lower. 16 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Feature Description (continued) If the resistor divider is not connected properly, output voltage cannot be regulated since the feedback loop is broken. If the FB pin is shorted to ground, the output voltage will be driven close to VIN, since the regulator sees under voltage on the FB pin and increases the output voltage. The load connected to the output could be damaged under such a condition. Do not short FB pin to ground when the LM46001 is enabled. It is important to route the feedback trace away from the noisy area of the PCB. For more layout recommendations, please refer to the Layout section. 7.3.4 Enable (ENABLE) Voltage on the ENABLE pin (VEN) controls the ON or OFF functionality of the LM46001. Applying a voltage less than 0.4 V to the ENABLE input shuts down the operation of the LM46001. In shutdown mode the quiescent current drops to typically 2.3 µA at VIN = 24 V. The internal LDO output voltage VCC is turned on when VEN is higher than 1.2 V. The LM46001 switching action and output regulation are enabled when VEN is greater than 2.1 V (typical). The LM46001 supplies regulated output voltage when enabled and output current up to 1 A. The ENABLE pin is an input and cannot be left open circuited. The simplest way to enable the operation of the LM46001 is to connect the ENABLE pin to the VIN pins directly. This allows self-start-up of the LM46001 when VIN is within the operation range. Many applications will benefit from the employment of an enable divider RENT and RENB in Figure 36 to establish a precision system UVLO level for the stage. System UVLO can be used for supplies operating from utility power as well as battery power. It can be used for sequencing, ensuring reliable operation, or supply protection, such as a battery discharge voltage level. An external logic signal can also be used to drive EN input for system sequencing and protection. VIN RENT ENABLE RENB Figure 36. System UVLO By Enable Dividers 7.3.5 VCC, UVLO and BIAS The LM46001 integrates an internal LDO to generate VCC for control circuitry and MOSFET drivers. The nominal voltage for VCC is 3.3 V. The VCC pin is the output of the LDO and must be properly bypassed. A high quality ceramic capacitor with 2.2 µF to 10 µF capacitance and 6.3 V or higher rated voltage should be placed as close as possible to VCC and grounded to the exposed PAD and ground pins. The VCC output pin should not be loaded, left floating, connected to any other external supply, or shorted to ground during operation. Shorting VCC to ground during operation may cause damage to the LM46001. Under voltage lockout (UVLO) prevents the LM46001 from operating until the VCC voltage exceeds 3.14 V (typical). The VCC UVLO threshold has 567 mV of hysteresis (typically) to prevent undesired shuting down due to temperary VIN droops. The internal LDO has two inputs: primary from VIN and secondary from BIAS input. The BIAS input powers the LDO when VBIAS is higher than the change-over threshold. Power loss of an LDO is calculated by ILDO * (VIN-LDO VOUT-LDO). The higher the difference between the input and output voltages of the LDO, the more power loss occur to supply the same output current. The BIAS input is designed to reduce the difference of the input and output voltages of the LDO to reduce power loss and improve the LM46001 efficiency, especially at light load. It is recommended to tie the BIAS pin to VOUT when VOUT ≥ 3.3 V. The BIAS pin should be grounded in applications with VOUT less than 3.3 V. BIAS input can also come from an external voltage source, if available, to reduce power loss. When used, a 1µF to 10µF high quality ceramic capacitor is recommended to bypass the BIAS pin to ground. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 17 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Feature Description (continued) 7.3.6 Soft-Start and Voltage Tracking (SS/TRK) The LM46001 has a flexible and easy to use start up rate control pin: SS/TRK. The Soft-start feature is there to prevent inrush current impacting the LM46001 and its supply when power is first applied. Soft-start is achieved by slowly ramping up the target regulation voltage when the device is first enabled or powered up. The simplest way to use the device is to leave the SS/TRK pin open circuit. The LM46001 will employ the internal soft-start control ramp and start up to the regulated output voltage in 4.1 ms typically. In applications with a large amount of output capacitors, or higher VOUT, or other special requirements, the softstart time can be extended by connecting an external capacitor CSS from SS/TRK pin to AGND. Extended softstart time further reduces the supply current needed to charge up output capacitors and supply any output loading. An internal current source (ISSC = 2.2 µA) charges CSS and generates a ramp from 0 V to VFB to control the ramp-up rate of the output voltage. VFB value is typically 1V and therefore it is not mentioned in the below equation. For a desired soft start time tSS, the capacitance for CSS can be found by: CSS ISSC u t SS (2) The soft start capacitor CSS is discharged by an internal FET when VOUT is shutdown by hiccup protection due to excessive load, temperature shutdown due to overheating or ENABLE = logic low. A large CSS cap will take a long time to discharge when ENABLE is toggled low. If ENABLE is toggled high again before the CSS is completely discharged, then the next resulting soft-start ramp will follow the internal soft-start ramp. Only when the soft-start voltage reaches the left-over voltage on CSS, will the output follow the ramp programmed by CSS. This behavior will look as if there are two slopes at startup. If this is not acceptable by a certain application, a RC low-pass filter can be added to ENABLE to slow down the shutting down of VCC, which allows more time to discharge CSS. The LM46001 is capable of start up into prebiased output conditions. When the inductor current reaches zero, the LS switch will be turned off to avoid negative current conduction. This operation mode is also called diode emulation mode. It is built-in by the DCM operation at light loads. With a prebiased output voltage, the LM46001 will wait until the soft-start ramp allows regulation above the prebiased voltage. It will then follow the soft-start ramp to the regulation level. When an external voltage ramp is applied to the SS/TRK pin, the LM46001 FB voltage follows the external ramp if the ramp magnitude is lower than the internal soft-start ramp. A resistor divider pair can be used on the external control ramp to the SS/TRK pin to program the tracking rate of the output voltage. The final external ramp voltage applied at the SS/TRK pin should be above 1.2 V to avoid abnormal operation. EXT RAMP RTRT SS/TRK RTRB Figure 37. Soft Start Tracking External Ramp VOUT tracked to an external voltage ramp has the option of ramping up slower or faster than the internal voltage ramp. VFB always follows the lower potential of the internal voltage ramp and the voltage on the SS/TRK pin. Figure 38 shows the case when VOUT ramps slower than the internal ramp, while Figure 39 shows when VOUT ramps faster than the internal ramp. Faster start up time may result in inductor current tripping current protection during start-up. Use with special care. 18 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Feature Description (continued) Enable Internal SS Ramp Ext Tracking Signal to SS pin VOUT Figure 38. Tracking with Longer Start-up Time Than The Internal Ramp Enable Internal SS Ramp Ext Tracking Signal to SS pin VOUT Figure 39. Tracking with Shorter Start-up Time Than The Internal Ramp Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 19 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Feature Description (continued) 7.3.7 Switching Frequency (RT) and Synchronization (SYNC) The switching frequency of the LM46001 can be programmed by the resistor RT from the RT pin to ground. The frequency is inversely proportional to the RT resistance. The RT pin can be left floating and the LM46001 will operate at 500 kHz default switching frequency. The RT pin is not designed to be connected to ground or any other voltage. For a desired frequency, typical RT resistance can be found by Equation 3. RT(kΩ) = 40200 / Freq (kHz) - 0.6 (3) Figure 40 shows RT resistance vs switching frequency FS curve. 250 RT Resistance (k) 200 150 100 50 0 0 500 1000 1500 Switching Frequency (kHz) 2000 2500 C008 Figure 40. RT Resistance vs Switching Frequency Table 1 provides typical RT values for a given FS. Table 1. Typical Frequency Setting RT Resistance 20 FS (kHz) RT (kΩ) 200 200 350 115 500 80.6 750 53.6 1000 39.2 1500 26.1 2000 19.6 2200 17.8 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Feature Description (continued) The LM46001 switching action can also be synchronized to an external clock from 200 kHz to 2.2 MHz. Connect an external clock to the SYNC pin, with proper high speed termination, to avoid ringing. The SYNC pin should be grounded if not used. SYNC EXT CLOCK RTERM Figure 41. Frequency Synchronization The recommendations for the external clock include high level no lower than 2 V, low level no higher than 0.4 V, duty cycle between 10% and 90% and both positive and negative pulse width no shorter than 80 ns. When the external clock fails at logic high or low, the LM46001 will switch at the frequency programmed by the RT resistor after a time-out period. It is recommended to connect a resistor RT to the RT pin such that the internal oscillator frequency is the same as the target clock frequency when the LM46001 is synchronized to an external clock. This allows the regulator to continue operating at approximately the same switching frequency if the external clock fails. The choice of switching frequency is usually a compromise between conversion efficiency and the size of the application circuit. Lower switching frequency implies reduced switching losses (including gate charge losses, switch transition losses, etc.) and usually results in higher overall efficiency. However, higher switching frequency allows use of smaller LC output filters and hence a more compact design. Lower inductance also helps transient response (higher large signal slew rate of inductor current), and reduces the DCR loss. The optimal switching frequency is usually a trade-off in a given application and thus needs to be determined on a case-by-case basis. It is related to the input voltage, output voltage, most frequent load current level(s), external component choices, and circuit size requirement. The choice of switching frequency may also be limited if an operating condition triggers TON-MIN or TOFF-MIN. 7.3.8 Minimum ON-Time, Minimum OFF-Time and Frequency Foldback at Drop-Out Conditions Minimum ON-time, TON-MIN, is the smallest duration of time that the HS switch can be on. TON-MIN is typically 125 ns in the LM46001. Minimum OFF-time, TOFF-MIN, is the smallest duration that the HS switch can be off. TOFF-MIN is typically 200 ns in the LM46001. In CCM operation, TON-MIN and TOFF-MIN limits the voltage conversion range given a selected switching frequency. The minimum duty cycle allowed is DMIN = TON-MIN × FS (4) And the maximum duty cycle allowed is DMAX = 1 - TOFF-MIN × FS (5) Given fixed TON-MIN and TOFF-MIN, the higher the switching frequency the narrower the range of the allowed duty cycle. In the LM46001, a frequency foldback scheme is employed to extend the maximum duty cycle when TOFFMIN is reached. The switching frequency will decrease once longer duty cycle is needed under low VIN conditions. The switching frequency can be decreased to approximately 1/10 of the programmed frequency by RT or the synchronization clock. Such wide range of frequency foldback allows the LM46001 output voltage to stay in regulation with a much lower supply voltage VIN. This leads to a lower effective drop-out voltage. Please refer to Typical Characteristics for more details. Given an output voltage, the choice of the switching frequency affects the allowed input voltage range, solution size and efficiency. The maximum operatable supply voltage can be found by VIN-MAX = VOUT / (FS * TON-MIN ) (6) At lower supply voltage, the switching frequency will decrease once TOFF-MIN is tripped. The minimum VIN without frequency foldback can be approximated by: VIN-MIN = VOUT / (1 - FS * TOFF-MIN ) (7) Taking considerations of power losses in the system with heavy load operation, VIN-MIN is higher than the result calculated in Equation 7 . With frequency foldback, VIN-MIN is lowered by decreased FS. Figure 42 gives an example of how FS decreases with decreasing supply voltage VIN at drop-out operation. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 21 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Feature Description (continued) Frequency (Hz) 1000000 100000 Load = 0.01 A Load = 0.1 A Load = 0.5 A Load = 1 A 10000 5.0 5.2 5.4 5.6 5.8 6.0 6.2 6.4 6.6 VIN (V) 6.8 7.0 C001 Figure 42. Switching Frequency Decreases in Drop-Out Operation VOUT = 5 V, FS = 1 MHz 7.3.9 Internal Compensation and CFF The LM46001 is internally compensated with RC = 400 kΩ and CC = 50 pF as shown in Functional Block Diagram. The internal compensation is designed such that the loop response is stable over the entire operating frequency and output voltage range. Depending on the output voltage, the compensation loop phase margin can be low with all ceramic capacitors at the output. An external feed-forward cap CFF is recommended to be placed in parallel with the top resistor divider RFBT for optimum transient performance. VOUT RFBT CFF FB RFBB Figure 43. Feed-Forward Capacitor for Loop Compensation The feed-forward capacitor CFF in parallel with RFBT places an additional zero before the cross over frequency of the control loop to boost phase margin. The zero frequency can be found by: fZ-CFF = 1 / ( 2π × RFBT × CFF ). (8) An additional pole is also introduced with CFF at the frequency of fP-CFF = 1 / ( 2π × CFF × ( RFBT // RFBB )). (9) The CFF should be selected such that the bandwidth of the control loop without the CFF is centered between fZ-CFF and fP-CFF. The zero fZ-CFF adds phase boost at the crossover frequency and improves transient response. The pole fP-CFF helps maintaining proper gain margin at frequency higher than the crossover frequency. Designs with different combinations of output capacitors need different CFF. Different types of capacitors have different Equivalent Series Resistance (ESR). Ceramic capacitors have the smallest ESR and need the most CFF. Electrolytic capacitors have much larger ESR and the ESR zero frequency fZ-ESR = 1 / ( 2π × ESR × COUT) (10) would be low enough to boost the phase up around the crossover frequency. Designs using mostly electrolytic capacitors at the output may not need any CFF. The CFF creates a time constant with RFBT that couples in the attenuated output voltage ripple to the FB node. If the CFF value is too large, it can couple too much ripple to the FB and affect VOUT regulation. It could also couple too much transient voltage deviation and falsely trip PGOOD thresholds. Therefore, CFF should be calculated based on output capacitors used in the system. At cold temperatures, the value of CFF might change based on the tolerance of the chosen component. This may reduce its impedance and ease noise coupling on the FB node. To avoid this, more capacitance can be added to the output or the value of CFF can be reduced. Please refer to the Detailed Design Procedure for the calculation of CFF. 22 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Feature Description (continued) 7.3.10 Bootstrap Voltage (BOOT) The driver of the HS switch requires a bias voltage higher than VIN when the HS switch is ON. The capacitor connected between CBOOT and SW pins works as a charge pump to boost voltage on the CBOOT pin to (VSW + VCC). The boot diode is integrated on the LM46001 die to minimize the Bill-Of-Material (BOM). A synchronous switch is also integrated in parallel with the boot diode to reduce voltage drop on CBOOT. A high quality ceramic 0.47 µF 6.3 V or higher capacitor is recommended for CBOOT. 7.3.11 Power Good (PGOOD) The LM46001 has a built in power-good flag at the PGOOD pin to indicate whether the output voltage is within its regulation level. The PGOOD signal can be used for start-up sequencing of multiple rails or fault protection. The PGOOD pin is an open-drain output that requires a pull-up resistor to an appropriate DC voltage. Voltage seen by the PGOOD pin should never exceed 12 V. A resistor divider pair can be used to divide the voltage down from a higher potential. A typical range of pull-up resistor value is 10 kΩ to 100 kΩ. When the FB voltage is within the power-good band, +4% above and -4% below the internal reference VREF typically, the PGOOD switch will be turned off and the PGOOD voltage will be pulled up to the voltage level defined by the pull up resistor or divider. When the FB voltage is outside of the tolerance band, +10 % above or 10 % below VREF typically, the PGOOD switch will be turned on and the PGOOD pin voltage will be pulled low to indicate power bad. Both rising and falling edges of the power-good flag have a built-in 220 µs (typical) deglitch delay. 7.3.12 Over-Current and Short Circuit Protection The LM46001 is protected from over-current conditions by cycle-by-cycle current limiting on both peak and valley of the inductor current. Hiccup mode will be activated to prevent over heating if a fault condition persists. High-side MOSFET over-current protection is implemented by the nature of the Peak Current Mode control. The HS switch current is sensed when the HS is turned on after a set blanking time. The HS switch current is compared to the output of the Error Amplifier (EA) minus slope compensation every switching cycle. Please refer to Functional Block Diagram for more details. The peak current of the HS switch is limited by the maximum EA output voltage minus the slope compensation at every switching cycle. The slope compensation magnitude at the peak current is proportional to the duty cycle. When the LS switch is turned on, the current going through it is also sensed and monitored. The LS switch will not be turned OFF at the end of a switching cycle if its current is above the LS current limit ILS-LIMIT. The LS switch will be kept ON so that inductor current keeps ramping down, until the inductor current ramps below ILSLIMIT. Then the LS switch will be turned OFF and the HS switch will be turned on after a dead time. If the current of the LS switch is higher than the LS current limit for 32 consecutive cycles and the power-good flag is low, hiccup current protection mode will be activated. In hiccup mode, the regulator will be shutdown and kept off for 5.5 ms typically before the LM46001 tries to start again. If over-current or short-circuit fault condition still exist, hiccup will repeat until the fault condition is removed. Hiccup mode reduces power dissipation under severe overcurrent conditions, preventing over heating and potential damage to the device. Hiccup is only activated when the power-good flag is low. Under non-severe over-current conditions when VOUT has not fallen outside of the PGOOD tolerance band, the LM46001 will reduce the switching frequency and keep the inductor current valley clamped at the LS current limit level. This operation mode allows slight over current operation during load transients without tripping hiccup. If the power-good flag becomes low, hiccup operation will start after LS current limit is tripped 32 consecutive cycles. 7.3.13 Thermal Shutdown Thermal shutdown is a built-in self protection to limit junction temperature and prevent damages due to over heating. Thermal shutdown turns off the device when the junction temperature exceeds 160°C typically to prevent further power dissipation and temperature rise. The junction temperature will reduce after thermal shutdown. The LM46001 will attempt to restart when the junction temperature drops to 150°C. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 23 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com 7.4 Device Functional Modes 7.4.1 Shutdown Mode The EN pin provides electrical ON and OFF control for the LM46001. When VEN is below 0.4 V, the device is in shutdown mode. Both the internal LDO and the switching regulator are off. In shutdown mode the quiescent current drops to 2.3 µA typically with VIN = 24 V. The LM46001 also employs under voltage lock out protection. If the VCC voltage is below the UVLO level, the output of the regulator will be turned off. 7.4.2 Stand-by Mode The internal LDO has a lower enable threshold than the regulator. When ENABLE voltage is above 1.2 V and below the precision enable falling threshold (1.8 V typically), the internal LDO regulates the VCC voltage at 3.3 V. The precision enable circuitry is turned on once VCC is above the UVLO threshold. The switching action and voltage regulation are not enabled unless VEN rises above the precision enable threshold (2.1 V typically). 7.4.3 Active Mode The LM46001 is in Active Mode when VEN is above the precision enable threshold and VCC is above its UVLO level. The simplest way to enable the LM46001 is to connect the EN pin to VIN. This allows self start-up of the LM46001 when the input voltage is in the operation range: 3.5 V to 60 V. Please refer to Enable (ENABLE) and VCC, UVLO and BIAS for details on setting these operating levels. In Active Mode, depending on the load current, the LM46001 will be in one of four modes: 1. Continuous conduction mode (CCM) with fixed switching frequency when load current is above half of the peak-to-peak inductor current ripple; 2. Discontinuous conduction mode (DCM) with fixed switching frequency when load current is lower than half of the peak-to-peak inductor current ripple in CCM operation; 3. Pulse Frequency Modulation (PFM) when switching frequency is decreased at very light load; 4. Fold-back mode when switching frequency is decreased to maintain output regulation at lower supply voltage VIN. 7.4.4 CCM Mode Continuous Conduction Mode (CCM) operation is employed in the LM46001 when the load current is higher than half of the peak-to-peak inductor current. In CCM operation, the frequency of operation is fixed unless the minimum HS switch ON-time (TON_MIN), the minimum HS switch OFF-time (TOFF_MIN) or LS current limit is exceeded. The output voltage ripple will be at a minimum in this mode and the maximum output current of 1 A can be supplied by the LM46001 7.4.5 Light Load Operation When the load current is lower than half of the peak-to-peak inductor current in CCM, the LM46001 will operate in Discontinuous Conduction Mode (DCM), also known as Diode Emulation Mode (DEM). In DCM operation, the LS FET is turned off when the inductor current drops to 0 A to improve efficiency. Both switching losses and conduction losses are reduced in DCM, comparing to forced PWM operation at light load. At even lighter current loads, Pulse Frequency Mode (PFM) is activated to maintain high efficiency operation. When the HS switch ON-time reduces to TON_MIN or peak inductor current reduces to its minimum IPEAK-MIN, the switching frequency will reduce to maintain proper regulation. Efficiency is greatly improved by reducing switching and gate drive losses. 7.4.6 Self-Bias Mode For highest efficiency of operation, it is recommended that the BIAS pin be connected directly to VOUT when VOUT ≥ 3.3 V. In this Self-Bias Mode of operation, the difference between the input and output voltages of the internal LDO are reduced and therefore the total efficiency of the LM46001 is improved. These efficiency gains are more evident during light load operation. During this mode of operation, the LM46001 operates with a minimum quiescent current of 24 µA (typical). Please refer to VCC, UVLO and BIAS for more details. 24 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 8 Applications and Implementation 8.1 Application Information The LM46001 is a step down DC-to-DC regulator. It is typically used to convert a higher DC voltage to a lower DC voltage with a maximum output current of 1 A. The following design procedure can be used to select components for the LM46001. Alternately, the WEBENCH® software may be used to generate complete designs. When generating a design, the WEBENCH® software utilizes iterative design procedure and accesses comprehensive databases of components. Please go to www.ti.com for more details. This section presents a simplified discussion of the design process. 8.2 Typical Applications The LM46001 only requires a few external components to convert from a wide range of supply voltage to output voltage. Figure 44 shows a basic schematic when BIAS is connected to VOUT . This is recommended for VOUT ≥ 3.3 V. For VOUT < 3.3 V, BIAS should be connected to ground, as shown in Figure 45. L VIN VIN CIN VOUT SW LM46001 ENABLE CBOOT COUT CIN LM46001 CBOOT COUT CBOOT CBIAS RT VOUT SW ENABLE CFF SS/TRK AGND VIN CBOOT BIAS PGOOD SYNC L VIN PGOOD RFBT BIAS RT FB VCC CVCC AGND PGND Figure 44. LM46001 Basic Schematic for VOUT ≥ 3.3 V, tie BIAS to VOUT RFBT FB VCC SYNC RFBB CVCC CFF SS/TRK RFBB PGND Figure 45. LM46001 Basic Schematic for VOUT < 3.3 V, tie BIAS to Ground The LM46001 also integrates a full list of optional features to aid system design requirements, such as precision enable, VCC UVLO, programmable soft-start, output voltage tracking, programmable switching frequency, clock synchronization and power-good indication. Each application can select the features for a more comprehensive design. A schematic with all features utilized is shown in Figure 46. L VIN RENT VIN CIN LM46001 ENABLE RENB VOUT SW CBOOT COUT RFBT CBOOT FB VCC SS/TRK CSS CFF RFBB CVCC RT BIAS RT CBIAS SYNC RSYNC AGND PGOOD PGND Tie BIAS to PGND when VOUT < 3.3 V Figure 46. LM46001 Schematic with All Features Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 25 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Typical Applications (continued) The external components have to fulfill the needs of the application, but also the stability criteria of the device's control loop. The LM46001 is optimized to work within a range of external components. The LC output filter's inductance and capacitance have to be considered in conjunction, creating a double pole, responsible for the corner frequency of the converter. Table 2 can be used to simplify the output filter component selection. Table 2. L, COUT and CFF Typical Values FS (kHz) L (µH) (1) COUT (µF) (2) CFF (pF) (3) (4) RT (kΩ) RFBB (kΩ) (3) (4) VOUT = 1 V 200 18 500 none 200 100 500 6.8 330 none 80.6 or open 100 1000 3.3 180 none 39.2 100 2200 1.5 100 none 17.8 100 200 47 220 44 200 442 500 18 100 33 80.6 or open 442 1000 10 47 18 39.2 442 2200 4.7 27 12 17.8 442 200 56 150 66 200 249 VOUT = 3.3 V VOUT = 5 V 500 27 66 33 80.6 or open 249 1000 15 33 22 39.2 249 2200 6.8 22 18 17.8 249 200 100 33 200 93.1 500 47 22 47 80.6 or open 93.1 1000 22 15 33 39.2 93.1 200 180 22 see note (5) 200 44.2 500 82 15 see note (5) 80.6 or open 44.2 see note (5) 39.2 44.2 VOUT = 12 V see note (5) VOUT = 24 V 1000 (1) (2) (3) (4) (5) 26 47 10 Inductor values are calculated based on typical VIN = 24 V. For VOUT = 24 V, VIN = 48 V. All the COUT values are after derating. Add more when using ceramics RFBT = 0 Ω for VOUT = 1 V. RFBT = 1 MΩ for all other VOUT settings. For designs with RFBT other than 1 MΩ, please adjust CFF such that (CFF × RFBT) is unchanged and adjust RFBB such that (RFBT / RFBB) is unchanged. High ESR COUT will give enough phase boost and CFF might not be needed. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Typical Applications (continued) 8.2.1 Design Requirements A detailed design procedure is described based on a design example. For this design example, use the parameters listed in Table 3 as the input parameters. Table 3. Design Example Parameters DESIGN PARAMETER VALUE Input Voltage VIN 24 V typical, range from 3.8 V to 60 V Output Voltage VOUT 3.3 V Input Ripple Voltage 400 mV Output ripple voltage 30 mV Output Current Rating 1A Operating Frequency 500 kHz Soft-start time 10 ms 8.2.2 Detailed Design Procedure 8.2.2.1 Output Voltage Set-Point The output voltage of the LM46001 device is externally adjustable using a resistor divider network. The divider network is comprised of top feedback resistor RFBT and bottom feedback resistor RFBB. The following equation is used to determine the output voltage of the converter: VFB RFBB RFBT VOUT VFB (11) Choose the value of the RFBT to be 1 MΩ to minimize quiescent current to improve light load efficiency in this application. With the desired output voltage set to be 3.3 V and the VFB = 1.016 V, the RFBB value can then be calculated using Equation 11. The formula yields a value of 444.83 kΩ. Choose the closest available value of 442 kΩ for the RFBB. Please refer to Adjustable Output Voltage for more details. 8.2.2.2 Switching Frequency The default switching frequency of the LM46001 device is set at 500 kHz when RT pin is open circuit. The switching frequency is selected to be 500 kHz in this application for one less passive components. If other frequency is desired, use Equation 12 to calculate the required value for RT. RT(kΩ) = 40200 / Freq (kHz) - 0.6 (12) For 500 kHz, the calculated RT is 79.8 kΩ and standard value 80.6 kΩ can also be used to set the switching frequency at 500 kHz. 8.2.2.3 Input Capacitors The LM46001 device requires high frequency input decoupling capacitor(s) and a bulk input capacitor, depending on the application. The typical recommended value for the high frequency decoupling capacitor is 4.7 µF to 10 µF. A high-quality ceramic type X5R or X7R with sufficient voltage rating is recommended. The voltage rating must be greater than the maximum input voltage. To compensate the derating of ceramic capacitor, a voltage rating of twice the maximum input voltage is recommended. Additionally, some bulk capacitance may be required, especially if the LM46001 circuit is not located within approximately 5 cm from the input voltage source. The bulk input capacitor is used to provide damping to the voltage spiking due to the lead inductance of the cable or trace. The value for this capacitor is not critical but must be rated to handle the maximum input voltage including ripple. For this design, a 10 µF, X7R dielectric capacitor rated for 100 V is used for the input decoupling capacitor. The equivalent series resistance (ESR) is approximately 3 mΩ, and the current-rating is 3 A. Include a capacitor with a value of 0.1 µF for high-frequency filtering and place it as close as possible to the device pins. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 27 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com NOTE DC Bias effect: High capacitance ceramic capacitors have a DC Bias effect, which will have a strong influence on the final effective capacitance. Therefore the right capacitor value has to be chosen carefully. Package size and voltage rating in combination with dielectric material are responsible for differences between the rated capacitor value and the effective capacitance. 8.2.2.4 Inductor Selection The first criterion for selecting an output inductor is the inductance itself. In most Buck converters, this value is based on the desired peak-to-peak ripple current, ΔiL, that flows in the inductor along with the DC load current. As with switching frequency, the selection of the inductor is a tradeoff between size and cost. Higher inductance gives lower ripple current and hence lower output voltage ripple with the same output capacitors. Lower inductance could result in smaller, less expensive component. An inductance that gives a ripple current of 20% to 40% of the 1 A at the typical supply voltage is a good starting point. ΔiL = (1/5 to 2/5) x IOUT. The peak-to-peak inductor current ripple can be found by Equation 13 and the range of inductance can be found by Equation 14 with the typical input voltage used as VIN. 'iL (VIN VOUT ) u D L u FS (13) (VIN VOUT ) u D (V VOUT ) u D d L d IN 0.4 u FS u IL MAX 0.2 u FS u IL MAX (14) D is the duty cycle of the converter which in a Buck converter it can be approximated as D = VOUT / VIN, assuming no loss power conversion. By calculating in terms of amperes, volts, and megahertz, the inductance value will come out in micro Henries. The inductor ripple current ratio is defined by: 'iL r IOUT (15) The second criterion is the inductor saturation current rating. The inductor should be rated to handle the maximum load current plus the ripple current: IL-PEAK = ILOAD-MAX + Δ iL (16) The LM46001 has both valley current limit and peak current limit. During an instantaneous short, the peak inductor current can be high due to a momentary increase in duty cycle. The inductor current rating should be higher than the HS current limit. It is advised to select an inductor with a larger core saturation margin and preferably a softer roll off of the inductance value over load current. In general, it is preferable to choose lower inductance in switching power supplies, because it usually corresponds to faster transient response, smaller DCR, and reduced size for more compact designs. But too low of an inductance can generate too large of an inductor current ripple such that over-current protection at the full load could be falsely triggered. It also generates more conduction loss, since the RMS current is slightly higher relative that with lower current ripple at the same DC current. Larger inductor current ripple also implies larger output voltage ripple with the same output capacitors. With peak current mode control, it is not recommended to have too small of an inductor current ripple. Enough inductor current ripple improves signal-to-noise ratio on the current comparator and makes the control loop more immune to noise. Once the inductance is determined, the type of inductor must be selected. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates hard, which means that inductance collapses abruptly when the peak design current is exceeded. The ‘hard’ saturation results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! For the design example, a standard 18 μH inductor from Wurth, Coiltronics, or Vishay can be used for the 3.3 V output with plenty of current rating margin. 28 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 8.2.2.5 Output Capacitor Selection The device is designed to be used with a wide variety of LC filters. It is generally desired to use as little output capacitance as possible to keep cost and size down. The output capacitor (s), COUT, should be chosen with care since it directly affects the steady state output voltage ripple, loop stability and the voltage over/undershoot during load current transients. The output voltage ripple is essentially composed of two parts. One is caused by the inductor current ripple going through the Equivalent Series Resistance (ESR) of the output capacitors: ΔVOUT-ESR = ΔiL× ESR (17) The other is caused by the inductor current ripple charging and discharging the output capacitors: ΔVOUT-C = ΔiL/ ( 8 × FS × COUT ) (18) The two components in the voltage ripple are not in phase, so the actual peak-to-peak ripple is smaller than the sum of the two peaks. Output capacitance is usually limited by transient performance specifications if the system requires tight voltage regulation in the presence of large current steps and fast slew rates. When a fast large load transient happens, output capacitors provide the required charge before the inductor current can slew to the appropriate level. The initial output voltage step is equal to the load current step multiplied by the ESR. VOUT continues to droop until the control loop response increases or decreases the inductor current to supply the load. To maintain a small over-shoot or under-shoot during a transient, small ESR and large capacitance are desired. But these also come with higher cost and size. Thus, the motivation is to seek a fast control loop response to reduce the output voltage deviation. For a given input and output requirement, the following inequality gives an approximation for an absolute minimum output cap required: COUT ! ª§ r 2 · 1 u «¨ u (1 Dc) ¸ Dc u (1 r) ¨ ¸ (FS u r u 'VOUT / IOUT ) «¬© 12 ¹ º » »¼ (19) Along with this for the same requirement, the max ESR should be calculated as per the following inequality ESR Dc 1 u ( 0.5) FS u COUT r (20) where r = Ripple ratio of the inductor ripple current (ΔIL / IOUT) ΔVOUT = Target output voltage undershoot D’ = 1 – Duty cycle FS = Switching Frequency IOUT = Load Current A general guide line for COUT range is that COUT should be larger than the minimum required output capacitance calculated by Equation 19, and smaller than 10 times the minimum required output capacitance or 1 mF. In applications with VOUT less than 3.3 V, it is critical that low ESR output capacitors are selected. This will limit potential output voltage overshoots as the input voltage falls below the device normal operating range. To optimize the transient behavior a feed-forward capacitor could be added in parallel with the upper feedback resistor. For this design example, two 47 µF,10 V, X7R ceramic capacitors are used in parallel. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 29 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com 8.2.2.6 Feed-Forward Capacitor The LM46001 is internally compensated and the internal R-C values are 400 kΩ and 50 pF respectively. Depending on the VOUT and frequency FS, if the output capacitor COUT is dominated by low ESR (ceramic types) capacitors, it could result in low phase margin. To improve the phase boost an external feedforward capacitor CFF can be added in parallel with RFBT. CFF is chosen such that phase margin is boosted at the crossover frequency that occurs with CFF removed. A simple estimation for the crossover frequency without CFF (fx) is shown in Equation 21, assuming COUT has very small ESR. 2.73 fx VOUT u COUT (21) The following equation for CFF was tested: CFF 1 1 u 2Sfx RFBT u (RFBT / /RFBB ) (22) This equation indicates that the crossover frequency is geometrically centered on the zero and pole frequencies caused by the CFF capacitor. For designs with higher ESR, CFF is not neeed when COUT has very high ESR and CFF calculated from Equation 22 should be reduced with medium ESR. Table 2 can be used as a quick starting point. For the application in this design example, a 33 pF COG capacitor is selected. 8.2.2.7 Bootstrap Capacitors Every LM46001 design requires a bootstrap capacitor, CBOOT. The recommended bootstrap capacitor is 0.47 μF and rated at 6.3 V or higher. The bootstrap capacitor is located between the SW pin and the CBOOT pin. The bootstrap capacitor must be a high-quality ceramic type with X7R or X5R grade dielectric for temperature stability. 8.2.2.8 VCC Capacitor The VCC pin is the output of an internal LDO for LM46001. The input for this LDO comes from either VIN or BIAS (please refer to Functional Block Diagram for LM46001). To insure stability of the part, place a minimum of 2.2 µF, 10 V capacitor from this pin to ground. 8.2.2.9 BIAS Capacitors For an output voltage of 3.3 V and greater, the BIAS pin can be connected to the output in order to increase light load efficiency. This pin is an input for the VCC LDO. When BIAS is not connected, the input for the VCC LDO will be internally connected into VIN. Since this is an LDO, the voltage differences between the input and output will affect the efficiency of the LDO. If necessary, a capacitor with a value of 1 μF can be added close to the BIAS pin as an input capacitor for the LDO. 8.2.2.10 Soft-Start Capacitors The user can leave the SS/TRK pin floating and the LM46001 will implement a soft start time of 4.1 ms typically. In order to use an external soft start capacitor, the capacitor should be sized such that the soft start time will be longer than 4.1 ms. Use the following equation in order to calculate the soft start capacitor value: CSS ISSC u t SS (23) Where, CSS = Soft start capacitor value (µF) ISS = Soft start charging current (µA) tSS = Desired soft start time (s) For the desired soft start time of 10 ms and soft start charging current of 2.2 µA, the equation above yield a soft start capacitor value of 0.022 µF. 30 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 8.2.2.11 Under Voltage Lockout Set-Point The undervoltage lockout (UVLO) is adjusted using the external voltage divider network of RENT and RENB. RENT is connected between VIN and the EN pin of the LM46001 device. RENB is connected between the EN pin and the GND pin. The UVLO has two thresholds, one for power up when the input voltage is rising and one for power down or brown outs when the input voltage is falling. The following equation can be used to determine the rising VIN (UVLO) level: VIN-UVLO-RISING = VENH × (RENB + RENT) / RENB (24) The EN rising threshold for LM46001 is set to be 2.1 V. Choose the value of RENB to be 1 MΩ to minimize input current going into the converter. If the desired VIN (UVLO) level is at 5 V, then the value of RENT can be calculated using the equation below: RENT = (VIN-UVLO-RISING / VENH - 1) × RENB (25) The above equation yields a value of 1.37 MΩ. The resulting falling UVLO threshold can be calculated as follows: VIN-UVLO-FALLING = 1.8 × (RENB + RENT) / RENB (26) 8.2.2.12 PGOOD A typical pull-up resistor value is 10 kΩ to 100 kΩ from the PGOOD pin to a voltage no higher than 12 V. If it is desired to pull up the PGOOD pin to a voltage higher than 12 V, a resistor can be added from the PGOOD pin to ground to divide the voltage seen by the PGOOD pin to a value no higher than 12 V. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 31 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com 8.2.3 Application Performance Curves Please refer to Table 2 for Bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves were taken at TA = 25 °C. 100 90 VOUT = 3.3 V FS = 500 kHz 80 RT Efficiency (%) 70 L=18 µH LM46001 VOUT SW CBOOT CBOOT 0.47 µF COUT 100 µF CBIAS 1 µF CFF BIAS VCC CVCC 2.2 µF 33 pF FB RFBT 1 M 60 50 VIN = 8V VIN = 12V VIN = 18V VIN = 24V VIN = 28V VIN = 36V 40 30 20 RFBB 432 k 10 0 0.001 0.010 0.100 1.000 Load Current (A) VOUT = 3.3 V FS = 500 kHz VIN = 24 V VOUT = 3.3 V Figure 48. Efficiency 3.40 3.50 3.38 3.40 3.36 3.30 3.34 3.20 3.32 3.10 VOUT (V) Vout (V) Figure 47. BOM for VOUT = 3.3 V FS = 500 kHz 3.30 3.28 3.26 3.24 3.22 3.20 0.001 C002 FS = 500 kHz VIN = 8V VIN = 12V VIN = 18V VIN = 24V VIN = 28V VIN = 36V 3.00 2.90 Load = 0.25A 2.80 Load = 0.5A 2.70 Load = 0.75A 2.60 Load = 1A 2.50 0.010 0.100 1.000 Load Current (A) VOUT = 3.3 V 3.5 4.0 4.5 5.0 VIN (V) C012 FS = 500 kHz VOUT = 3.3 V Figure 49. Output Voltage Regulation C022 FS = 500 kHz Figure 50. Drop-Out Curve 1.2 VDROP_ON_0.75_LOAD (750 mV/DIV) 1 VOUT (200 mV/DIV) Current (A) 0.8 0.6 0.4 R,JA = 10 C/W IL (1 A/DIV) 0.2 R,JA = 20 C/W R,JA = 30 C/W 0 Time (200 µs/DIV) 50 60 70 80 90 100 Temperature (C) VOUT = 3.3 V FS = 500 kHz VIN = 24 V VOUT = 3.3 V Figure 51. Load Transient Between 0.1 A and 1 A 32 Submit Documentation Feedback FS = 500 kHz 110 120 C001 VIN = 24 V Figure 52. Derating Curve Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Please refer to Table 2 for Bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves were taken at TA = 25 °C. 100 90 VOUT = 5 V FS = 500 kHz 80 RT LM46001 Efficiency (%) 70 L=27 µH VOUT SW CBOOT COUT 66 µF CBOOT 0.47 µF BIAS CBIAS 1 µF VCC CVCC 2.2 µF CFF 33 pF FB RFBT 1 M RFBB 249 k 60 50 VIN = 12V 40 VIN = 18V 30 VIN = 24V 20 VIN = 28V 10 VIN = 36V VIN = 42V 0 0.001 0.010 0.100 1.000 Load Current (A) VOUT = 5 V FS = 500 kHz VIN = 24 V VOUT = 5 V Figure 53. BOM for VOUT = 5 V, FS = 500 kHz C003 FS = 500 kHz Figure 54. Efficiency 5.20 5.2 5.15 5.0 4.8 5.05 VOUT (V) Vout (V) 5.10 5.00 4.95 4.6 Load = 0.25A 4.4 4.90 4.85 4.80 0.001 VIN = 12V VIN = 18V VIN = 24V VIN = 28V VIN = 36V VIN = 42V Load = 0.5A 4.2 Load = 0.75A Load = 1A 4.0 0.010 0.100 1.000 Load Current (A) VOUT = 5 V 5.0 5.5 6.0 6.5 VIN (V) C013 FS = 500 kHz VOUT = 5 V Figure 55. Output Voltage Regulation C023 FS = 500 kHz Figure 56. Drop-Out Curve 1.2 VDROP_ON_0.75_LOAD (750 mV/DIV) 1 VOUT (200 mV/DIV) Current (A) 0.8 0.6 0.4 R,JA = 10 C/W IL (1 A/DIV) 0.2 R,JA = 20 C/W R,JA = 30 C/W 0 Time (200 µs/DIV) 50 60 70 80 90 100 110 120 Temperature (C) VOUT = 5 V FS = 500 kHz VIN = 24 V VOUT = 5 V Figure 57. Load Transient Between 0.1 A and 1 A FS = 500 kHz C001 VIN = 24 V Figure 58. Derating Curve Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 33 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Please refer to Table 2 for Bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves were taken at TA = 25 °C. 100 90 VOUT = 5 V FS = 200 kHz 80 RT RT 200 k Efficiency (%) 70 L=56 µH VOUT SW LM46001 CBOOT CBOOT 0.47 µF COUT 150 µF CBIAS 1 µF CFF BIAS VCC CVCC 2.2 µF 68 pF FB 60 50 40 30 RFBT 1 M 20 RFBB 249 k 10 0 0.001 VIN = 8V VIN = 24V VIN = 42V VIN = 12V VIN = 28V VIN = 48V 0.010 VIN = 18V VIN = 36V VIN = 60V 0.100 1.000 Load Current (A) VOUT = 5 V FS = 200 kHz VIN = 24 V VOUT = 5 V Figure 59. BOM for VOUT = 5 V, FS = 200 kHz C004 FS = 200 kHz Figure 60. Efficiency 5.15 5.2 5.10 5.0 5.05 VOUT (V) Vout (V) 4.8 5.00 4.95 4.90 4.85 4.80 0.001 4.6 Load = 0.25A 4.4 VIN = 8V VIN = 12V Load = 0.5A VIN = 18V VIN = 24V VIN = 28V VIN = 36V VIN = 42V VIN = 48V VIN = 60V Load = 0.75A 4.2 Load = 1A 4.0 0.010 0.100 1.000 Load Current (A) VOUT = 5 V 5.0 5.5 6.0 6.5 VIN (V) C014 FS = 200 kHz VOUT = 5 V Figure 61. Output Voltage Regulation C024 FS = 200 kHz Figure 62. Drop-Out Curve 1.2 VDROP_ON_0.75_LOAD (750 mV/DIV) 1 VOUT (200 mV/DIV) Current (A) 0.8 0.6 0.4 R,JA = 10 C/W IL (1 A/DIV) 0.2 R,JA = 20 C/W R,JA = 30 C/W 0 Time (200 µs/DIV) 50 60 70 80 90 100 Temperature (C) VOUT = 5 V FS = 200 kHz VIN = 24 V VOUT = 5 V Figure 63. Load Transient Between 0.1 A and 1 A 34 Submit Documentation Feedback FS = 200 kHz 110 120 C001 VIN = 24 V Figure 64. Derating Curve Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Please refer to Table 2 for Bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves were taken at TA = 25 °C. 100 90 VOUT = 5 V FS = 1 MHz 80 RT RT 39.2 k LM46001 Efficiency (%) 70 L=15 µH VOUT SW CBOOT CBOOT 0.47 µF COUT 33 µF CBIAS 1 µF CFF BIAS VCC CVCC 2.2 µF 22 pF FB 60 50 40 30 RFBT 1 M VIN = 12V VIN = 18V VIN = 24V VIN = 28V 20 RFBB 249 k 10 0 0.001 0.010 0.100 1.000 Load Current (A) VOUT = 5 V FS = 1 MHz VIN = 24 V VOUT = 5 V Figure 65. BOM for VOUT = 5 V, FS = 1 MHz C005 FS = 1 MHz VIN = 24 V Figure 66. Efficiency 5.20 5.2 5.15 5.0 4.8 5.05 VOUT (V) Vout (V) 5.10 5.00 4.95 4.6 Load = 0.25A 4.4 VIN = 12V VIN = 18V VIN = 24V VIN = 28V 4.90 4.85 4.80 0.001 0.010 0.100 Load = 0.75A 4.2 Load = 1A 4.0 1.000 Load Current (A) VOUT = 5 V Load = 0.5A 5.0 5.5 6.0 6.5 VIN (V) C015 FS = 1 MHz VOUT = 5 V Figure 67. Output Voltage Regulation C025 FS = 1 MHz Figure 68. Drop-Out Curve 1.2 VDROP_ON_0.75_LOAD (750 mV/DIV) 1 VOUT (200 mV/DIV) Current (A) 0.8 0.6 0.4 R,JA = 10 C/W IL (1 A/DIV) 0.2 R,JA = 20 C/W R,JA = 30 C/W 0 Time (200 µs/DIV) 50 60 70 80 90 100 110 120 Temperature (C) VOUT = 5 V FS = 1 MHz VIN = 24 V VOUT = 5 V Figure 69. Load Transient Between 0.1 A and 1 A FS = 1 MHz C001 VIN = 24 V Figure 70. Derating Curve Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 35 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Please refer to Table 2 for Bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves were taken at TA = 25 °C. 100 90 VOUT = 12 V FS = 500 kHz 80 RT LM46001 Efficiency (%) 70 L=47 µH VOUT SW CBOOT COUT 22 µF CBOOT 0.47 µF BIAS CBIAS 1 µF VCC CVCC 2.2 µF CFF 47 pF FB RFBT 1 M 60 50 RFBB 90.9 k VIN = 24V 40 VIN = 28V 30 VIN = 36V 20 VIN = 42V 10 VIN = 48V VIN = 60V 0 0.001 0.010 0.100 1.000 Load Current (A) VOUT = 12 V FS = 500 kHz VIN = 24 V VOUT = 12 V Figure 71. BOM for VOUT = 12 V, FS = 500 kHz C007 FS = 500 kHz Figure 72. Efficiency 12.5 12.4 12.4 12.2 12.3 12.0 12.1 VOUT (V) Vout (V) 12.2 12.0 11.9 11.8 11.7 11.6 11.5 0.001 VIN = 24V VIN = 28V VIN = 36V VIN = 42V VIN = 48V VIN = 60V 0.010 0.100 VOUT = 12 V 11.6 Load = 0.25A 11.4 Load = 0.5A 11.2 Load = 0.75A Load = 1A 1.000 Load Current (A) 11.8 11.0 12.0 12.5 13.0 13.5 14.0 VIN (V) C017 FS = 500 kHz VOUT = 12 V Figure 73. Output Voltage Regulation C027 FS = 500 kHz Figure 74. Drop-Out Curve 1.2 1 ILOAD (1 A/DIV) VOUT (500 mV/DIV) Current (A) 0.8 0.6 0.4 R,JA = 10 C/W IL (1 A/DIV) 0.2 R,JA = 20 C/W R,JA = 30 C/W 0 Time (200 µs/DIV) 50 60 70 80 90 100 Temperature (C) VOUT = 12 V FS = 500 kHz VIN = 24 V VOUT = 12 V Figure 75. Load Transient Between 0.1 A and 1 A 36 Submit Documentation Feedback FS = 500 kHz 110 120 C001 VIN = 24 V Figure 76. Derating Curve Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Please refer to Table 2 for Bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves were taken at TA = 25 °C. 120 VOUT = 24 V FS = 500 kHz 100 RT LM46001 CBOOT VOUT CBOOT 0.47 µF COUT 15 µF CBIAS 1 µF CFF BIAS VCC CVCC 2.2 µF 82 pF FB 80 Efficiency (%) L=82 µH SW 60 VIN = 36V 40 RFBT 1 M VIN = 42V 20 RFBB 43.2 k VIN = 48V VIN = 60V 0 0.001 0.010 0.100 1.000 Load Current (A) VOUT = 24 V FS = 500 kHz VIN = 48 V VOUT = 24 V C008 FS = 500 kHz Figure 77. BOM for VOUT = 24 V, FS = 500 kHz Figure 78. Efficiency 25.0 24.5 24.8 24.6 24.0 24.2 VOUT (V) Vout (V) 24.4 24.0 23.8 23.6 VIN = 36V VIN = 42V VIN = 48V VIN = 60V 23.4 23.2 23.0 0.001 0.010 0.100 VOUT = 24 V 23.0 Load = 0.25A Load = 0.5A 22.5 Load = 0.75A Load = 1A 1.000 Load Current (A) 23.5 22.0 24.0 24.5 25.0 25.5 26.0 26.5 27.0 VIN (V) C018 FS = 500 kHz VOUT = 24 V Figure 79. Output Voltage Regulation C028 FS = 500 kHz Figure 80. Drop-Out Curve 1.2 1 ILOAD (1 A/DIV) VOUT (500 mV/DIV) Current (A) 0.8 0.6 0.4 IL (1 A/DIV) R,JA = 10 C/W 0.2 R,JA = 20 C/W 0 Time (200 µs/DIV) 50 60 70 80 90 100 110 120 Temperature (C) VOUT = 24 V FS = 500 kHz VIN = 48 V VOUT = 24 V Figure 81. Load Transient Between 0.1 A and 1 A FS = 500 kHz C001 VIN = 48 V Figure 82. Derating Curve Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 37 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com 1.2 1.2 1 1 0.8 0.8 Current (A) Current (A) Please refer to Table 2 for Bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves were taken at TA = 25 °C. 0.6 0.4 0.6 0.4 Vin = 12V Vin = 12V Vin = 24V 0.2 Vin = 24V 0.2 Vin = 36V Vin = 36V 0 0 50 60 70 80 90 100 110 120 Temperature (C) VOUT = 3.3 V 50 70 FS = 500 kHz 80 90 100 110 120 Temperature (C) RθJA = 20 °C/W VOUT = 5 V Figure 83. Derating Curve with RθJA = 20 °C/W C001 FS = 500 kHz RθJA = 20 °C/W Figure 84. Derating Curve with RθJA = 20 °C/W 1.2 1.2 1 1 0.8 0.8 Current (A) Current (A) 60 C001 0.6 0.4 0.6 0.4 Vin = 12V Vin = 12V Vin = 24V 0.2 Vin = 24V 0.2 Vin = 36V Vin = 36V 0 0 50 60 70 80 90 100 110 120 Temperature (C) VOUT = 5 V 50 60 FS = 200 kHz 70 80 90 100 110 120 Temperature (C) C001 RθJA = 20 °C/W VOUT = 5 V Figure 85. Derating Curve with RθJA = 20 °C/W C001 FS = 1 MHz RθJA = 20 °C/W Figure 86. Derating Curve with RθJA = 20 °C/W 1000000 Frequency (Hz) Frequency (Hz) 1000000 100000 VIN = 5 V VIN = 8 V 10000 0.001 0.1 Load (A) VOUT = 3.3 V VIN = 12 V VIN = 18 V VIN = 18 V VIN = 24 V 1 10000 0.001 FS = 500 kHz VIN = 36 V 0.01 0.1 Load (A) C001 VOUT = 5 V Figure 87. Switching Frequency vs IOUT in PFM Operation 38 VIN = 8 V VIN = 12 V VIN = 24 V 0.01 100000 1 C001 FS = 1 MHz Figure 88. Switching Frequency vs IOUT in PFM Operation Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Please refer to Table 2 for Bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves were taken at TA = 25 °C. SW (10 V/DIV) SW (10 V/DIV) VOUT (10 mV/DIV) VOUT (10 mV/DIV) IL (1 A/DIV) IL (1 A/DIV) Time (2 µs/DIV) VOUT = 3.3 V FS = 500 kHz Time (2 µs/DIV) IOUT = 1 A Figure 89. Switching Waveform in CCM Operation VOUT = 3.3 V FS = 500 kHz IOUT = 90 mA Figure 90. Switching Waveform in DCM Operation SW (10 V/DIV) PGOOD (2 V/DIV) VOUT (2 V/DIV) VOUT (10 mV/DIV) IL (1 A/DIV) IL (1 A/DIV) Time (2 µs/DIV) VOUT = 3.3 V FS = 500 kHz Time (2 ms/DIV) IOUT = 10 mA Figure 91. Switching Waveform in PFM Operation VIN = 24 V VOUT = 3.3 V RLOAD = 3.3 Ω Figure 92. Startup Into Full Load with Internal Soft-Start Rate PGOOD (2 V/DIV) PGOOD (2 V/DIV) VOUT (2 V/DIV) VOUT (2 V/DIV) IL (500 mA/DIV) IL (200 mA/DIV) Time (2 ms/DIV) VIN = 24 V VOUT = 3.3 V Time (2 ms/DIV) RLOAD = 6.6 Ω Figure 93. Startup Into Half Load with Internal Soft-Start Rate VIN = 24 V VOUT = 3.3 V RLOAD = 33 Ω Figure 94. Startup Into 100 mA with Internal Soft-Start Rate Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 39 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Please refer to Table 2 for Bill of materials for each VOUT and FS combination. Unless otherwise stated, application performance curves were taken at TA = 25 °C. PGOOD (2 V/DIV) PGOOD (10 V/DIV) VOUT (1 V/DIV) VOUT (10 V/DIV) IL (200 mA/DIV) IL (1 A/DIV) Time (2 ms/DIV) VIN = 24 V VOUT = 3.3 V Time (5 ms/DIV) RLOAD = Open Figure 95. Startup Into 1.0 V Pre-biased Voltage VIN = 24 V VOUT = 12 V RLOAD = 6 Ω Figure 96. Startup with External Capacitor CSS = 33 nF VIN (20 V/DIV) VIN (20 V/DIV) VOUT (50 mV/DIV) VOUT (50 mV/DIV) IL (1 A/DIV) IL (500 mA/DIV) Time (2 ms/DIV) VOUT = 3.3 V Time (2 ms/DIV) FS = 500 kHz IOUT = 1 A Figure 97. Line Transient: VIN Transitions Between 12 V and 48 V VOUT = 3.3 V FS = 500 kHz IOUT = 0.5 A Figure 98. Line Transient: VIN Transitions Between 12 V and 48 V PGOOD (5 V/DIV) VOUT (5 V/DIV) IL (1 A/DIV) Time (10 ms/DIV) VOUT = 3.3 V FS = 500 kHz VIN = 24 V Figure 99. Short Circuit Protection and Recover 40 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 9 Power Supply Recommendations The LM46001 is designed to operate from an input voltage supply range between 3.5 V and 60 V. This input supply should be able to withstand the maximum input current and maintain a voltage above 3.5 V. The resistance of the input supply rail should be low enough that an input current transient does not cause a high enough drop at the LM46001 supply voltage that can cause a false UVLO fault triggering and system reset. If the input supply is located more than a few inches from the LM46001 additional bulk capacitance may be required in addition to the ceramic bypass capacitors. The amount of bulk capacitance is not critical, but a 47 µF or 100 µF electrolytic capacitor is a typical choice. 10 Layout The performance of any switching converter depends as much upon the layout of the PCB as the component selection. The following guidelines will help users design a PCB with the best power conversion performance, thermal performance, and minimized generation of unwanted EMI. 10.1 Layout Guidelines 1. Place ceramic high frequency bypass CIN as close as possible to the LM46001 VIN and PGND pins. Grounding for both the input and output capacitors should consist of localized top side planes that connect to the PGND pins and PAD. 2. Place bypass capacitors for VCC and BIAS close to the pins and ground the bypass capacitors to device ground. 3. Minimize trace length to the FB pin. Both feedback resistors, RFBT and RFBB should be located close to the FB pin. Place CFF directly in parallel with RFBT. If VOUT accuracy at the load is important, make sure VOUT sense is made at the load. Route VOUT sense path away from noisy nodes and preferably through a layer on the other side of a shielding layer. 4. Use ground plane in one of the middle layers as noise shielding and heat dissipation path. 5. Have a single point ground connection to the plane. The ground connections for the feedback, soft-start, and enable components should be routed to the ground plane. This prevents any switched or load currents from flowing in the analog ground traces. If not properly handled, poor grounding can result in degraded load regulation or erratic output voltage ripple behavior. 6. Make VIN, VOUT and ground bus connections as wide as possible. This reduces any voltage drops on the input or output paths of the converter and maximizes efficiency. 7. Provide adequate device heat-sinking. Use an array of heat-sinking vias to connect the exposed pad to the ground plane on the bottom PCB layer. If the PCB has multiple copper layers, these thermal vias can also be connected to inner layer heat-spreading ground planes. Ensure enough copper area is used for heat-sinking to keep the junction temperature below 125°C. 10.1.1 Compact Layout for EMI Reduction Radiated EMI is generated by the high di/dt components in pulsing currents in switching converters. The larger area covered by the path of a pulsing current, the more electromagnetic emission is generated. The key to minimize radiated EMI is to identify the pulsing current path and minimize the area of the path. In Buck converters, the pulsing current path is from the VIN side of the input capacitors to HS switch, to the LS switch, and then return to the ground of the input capacitors, as shown in Figure 100. BUCK CONVERTER VIN VIN SW L CIN VOUT COUT PGND High di/dt current PGND Figure 100. Buck Converter High di / dt Path Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 41 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com Layout Guidelines (continued) High frequency ceramic bypass capacitors at the input side provide primary path for the high di/dt components of the pulsing current. Placing ceramic bypass capacitor(s) as close as possible to the VIN and PGND pins is the key to EMI reduction. The SW pin connecting to the inductor should be as short as possible, and just wide enough to carry the load current without excessive heating. Short, thick traces or copper pours (shapes) should be used for high current conduction path to minimize parasitic resistance. The output capacitors should be place close to the VOUT end of the inductor and closely grounded to PGND pin and exposed PAD. The bypass capacitors on VCC and BIAS pins should be placed as close as possible to the pins respectively and closely grounded to PGND and the exposed PAD. 10.1.2 Ground Plane and Thermal Considerations It is recommended to use one of the middle layers as a solid ground plane. Ground plane provides shielding for sensitive circuits and traces. It also provides a quiet reference potential for the control circuitry. The AGND and PGND pins should be connected to the ground plane using vias right next to the bypass capacitors. PGND pins are connected to the source of the internal LS switch. They should be connected directly to the grounds of the input and output capacitors. The PGND net contains noise at the switching frequency and may bounce due to load variations. The PGND trace, as well as PVIN and SW traces, should be constrained to one side of the ground plane. The other side of the ground plane contains much less noise and should be used for sensitive routes. It is recommended to provide adequate device heat sinking by utilizing the PAD of the IC as the primary thermal path. Use a minimum 4 by 4 array of 10 mil thermal vias to connect the PAD to the system ground plane for heat sinking. The vias should be evenly distributed under the PAD. Use as much copper as possible for system ground plane on the top and bottom layers for the best heat dissipation. It is recommended to use a four-layer board with the copper thickness, for the four layers, starting from the top one, 2 oz / 1 oz / 1 oz / 2 oz. Four layer boards with enough copper thickness and proper layout provides low current conduction impedance, proper shielding and lower thermal resistance. The thermal characteristics of the LM46001 are specified using the parameter RθJA, which characterize the junction temperature of the silicon to the ambient temperature in a specific system. Although the value of RθJA is dependant on many variables, it still can be used to approximate the operating junction temperature of the device. To obtain an estimate of the device junction temperature, one may use the following relationship: TJ = PD× RθJA + TA (27) where TJ = Junction temperature in °C PD = VIN x IIN x (1 − Efficiency) − 1.1 x IOUT x DCR DCR = Inductor DC parasitic resistance in Ω RθJA = Junction-to-ambient thermal resistance of the device in °C/W TA = Ambient temperature in °C. The maximum operating junction temperature of the LM46001 is 125°C. RθJA is highly related to PCB size and layout, as well as enviromental factors such as heat sinking and air flow. Figure 101 shows measured results of RθJA with different copper area on a 2-layer board and a 4-layer board. 42 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 Layout Guidelines (continued) 50.0 1W @ 0fpm - 2 layer 2W @ 0fpm - 2 layer R,JA (C/W) 45.0 1W @ 0fpm - 4 layer 2W @ 0fpm - 4 layer 40.0 35.0 30.0 25.0 20.0 20mm x 20mm 30mm x 30mm 40mm x 40mm Copper Area 50mm x 50mm C030 Figure 101. Measured RθJA vs PCB Copper Area on a 2-layer Board and a 4-layer Board 10.1.3 Feedback Resistors To reduce noise sensitivity of the output voltage feedback path, it is important to place the resistor divider and CFF close to the FB pin, rather than close to the load. The FB pin is the input to the error amplifier, so it is a high impedance node and very sensitive to noise. Placing the resistor divider and CFF closer to the FB pin reduces the trace length of FB signal and reduces noise coupling. The output node is a low impedance node, so the trace from VOUT to the resistor divider can be long if short path is not available. If voltage accuracy at the load is important, make sure voltage sense is made at the load. Doing so will correct for voltage drops along the traces and provide the best output accuracy. The voltage sense trace from the load to the feedback resistor divider should be routed away from the SW node path, the inductor and VIN path to avoid contaminating the feedback signal with switch noise, while also minimizing the trace length. This is most important when high value resistors are used to set the output voltage. It is recommended to route the voltage sense trace on a different layer than the inductor, SW node and VIN path, such that there is a ground plane in between the feedback trace and inductor / SW node / VIN polygon. This provides further shielding for the voltage feedback path from switching noises. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 43 LM46001 SNVSA46A – JUNE 2014 – REVISED JULY 2014 www.ti.com 10.2 Layout Example TO LOAD + VOUT sense point is away from inductor and past COUT VOUT VOUT distribution point is away from inductor and past COUT COUT As much copper area as possible, for better thermal performance L GND CBOOT Place bypass caps close to terminals CVCC Ground bypass caps to DAP 1 2 CBOOT 3 VCC 4 BIAS 5 SYNC Thermal Vias under DAP PAD (17) 16 PGND 15 PGND 14 VIN 13 VIN 12 EN 6 11 SS/TRK RT 7 10 AGND PGOOD 8 9 CBIAS + SW SW CIN VIN Place ceramic bypass caps close to VIN and PGND terminals RFBB FB RFBT CFF Route VOUT sense trace away from SW and VIN nodes. Preferably shielded in an alternative layer GND Plane As much copper area as possible, for better thermal performance Figure 102. LM46001 PCB Layout Example 44 Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 LM46001 www.ti.com SNVSA46A – JUNE 2014 – REVISED JULY 2014 11 Device and Documentation Support 11.1 Trademarks SIMPLE SWITCHER is a registered trademark of Texas Instruments. All other trademarks are the property of their respective owners. 11.2 Electrostatic Discharge Caution These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 11.3 Glossary SLYZ022 — TI Glossary. This glossary lists and explains terms, acronyms, and definitions. 12 Mechanical, Packaging, and Orderable Information The following pages include mechanical, packaging, and orderable information. This information is the most current data available for the designated devices. This data is subject to change without notice and revision of this document. For browser-based versions of this data sheet, refer to the left-hand navigation. Submit Documentation Feedback Copyright © 2014, Texas Instruments Incorporated Product Folder Links: LM46001 45 PACKAGE OPTION ADDENDUM www.ti.com 18-Jul-2014 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish MSL Peak Temp (2) (6) (3) Op Temp (°C) Device Marking (4/5) LM46001PWP ACTIVE HTSSOP PWP 16 90 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR -40 to 125 LM46001 LM46001PWPR ACTIVE HTSSOP PWP 16 2000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR -40 to 125 LM46001 LM46001PWPT ACTIVE HTSSOP PWP 16 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR -40 to 125 LM46001 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device. (5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Device Marking for that device. (6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish value exceeds the maximum column width. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and Addendum-Page 1 Samples PACKAGE OPTION ADDENDUM www.ti.com 18-Jul-2014 continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. Addendum-Page 2 PACKAGE MATERIALS INFORMATION www.ti.com 23-Jun-2016 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) LM46001PWPR HTSSOP PWP 16 2000 330.0 12.4 LM46001PWPT HTSSOP PWP 16 250 180.0 12.4 Pack Materials-Page 1 B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant 6.9 5.6 1.6 8.0 12.0 Q1 6.9 5.6 1.6 8.0 12.0 Q1 PACKAGE MATERIALS INFORMATION www.ti.com 23-Jun-2016 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LM46001PWPR HTSSOP PWP 16 2000 367.0 367.0 35.0 LM46001PWPT HTSSOP PWP 16 250 210.0 185.0 35.0 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest issue. Buyers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. All semiconductor products (also referred to herein as “components”) are sold subject to TI’s terms and conditions of sale supplied at the time of order acknowledgment. TI warrants performance of its components to the specifications applicable at the time of sale, in accordance with the warranty in TI’s terms and conditions of sale of semiconductor products. Testing and other quality control techniques are used to the extent TI deems necessary to support this warranty. Except where mandated by applicable law, testing of all parameters of each component is not necessarily performed. TI assumes no liability for applications assistance or the design of Buyers’ products. Buyers are responsible for their products and applications using TI components. To minimize the risks associated with Buyers’ products and applications, Buyers should provide adequate design and operating safeguards. TI does not warrant or represent that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other intellectual property right relating to any combination, machine, or process in which TI components or services are used. Information published by TI regarding third-party products or services does not constitute a license to use such products or services or a warranty or endorsement thereof. Use of such information may require a license from a third party under the patents or other intellectual property of the third party, or a license from TI under the patents or other intellectual property of TI. Reproduction of significant portions of TI information in TI data books or data sheets is permissible only if reproduction is without alteration and is accompanied by all associated warranties, conditions, limitations, and notices. TI is not responsible or liable for such altered documentation. Information of third parties may be subject to additional restrictions. Resale of TI components or services with statements different from or beyond the parameters stated by TI for that component or service voids all express and any implied warranties for the associated TI component or service and is an unfair and deceptive business practice. TI is not responsible or liable for any such statements. Buyer acknowledges and agrees that it is solely responsible for compliance with all legal, regulatory and safety-related requirements concerning its products, and any use of TI components in its applications, notwithstanding any applications-related information or support that may be provided by TI. Buyer represents and agrees that it has all the necessary expertise to create and implement safeguards which anticipate dangerous consequences of failures, monitor failures and their consequences, lessen the likelihood of failures that might cause harm and take appropriate remedial actions. Buyer will fully indemnify TI and its representatives against any damages arising out of the use of any TI components in safety-critical applications. In some cases, TI components may be promoted specifically to facilitate safety-related applications. With such components, TI’s goal is to help enable customers to design and create their own end-product solutions that meet applicable functional safety standards and requirements. Nonetheless, such components are subject to these terms. No TI components are authorized for use in FDA Class III (or similar life-critical medical equipment) unless authorized officers of the parties have executed a special agreement specifically governing such use. Only those TI components which TI has specifically designated as military grade or “enhanced plastic” are designed and intended for use in military/aerospace applications or environments. Buyer acknowledges and agrees that any military or aerospace use of TI components which have not been so designated is solely at the Buyer's risk, and that Buyer is solely responsible for compliance with all legal and regulatory requirements in connection with such use. TI has specifically designated certain components as meeting ISO/TS16949 requirements, mainly for automotive use. In any case of use of non-designated products, TI will not be responsible for any failure to meet ISO/TS16949. Products Applications Audio www.ti.com/audio Automotive and Transportation www.ti.com/automotive Amplifiers amplifier.ti.com Communications and Telecom www.ti.com/communications Data Converters dataconverter.ti.com Computers and Peripherals www.ti.com/computers DLP® Products www.dlp.com Consumer Electronics www.ti.com/consumer-apps DSP dsp.ti.com Energy and Lighting www.ti.com/energy Clocks and Timers www.ti.com/clocks Industrial www.ti.com/industrial Interface interface.ti.com Medical www.ti.com/medical Logic logic.ti.com Security www.ti.com/security Power Mgmt power.ti.com Space, Avionics and Defense www.ti.com/space-avionics-defense Microcontrollers microcontroller.ti.com Video and Imaging www.ti.com/video RFID www.ti-rfid.com OMAP Applications Processors www.ti.com/omap TI E2E Community e2e.ti.com Wireless Connectivity www.ti.com/wirelessconnectivity Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265 Copyright © 2016, Texas Instruments Incorporated