AVAGO ACPL-C79B-000E Precision miniature isolation amplifier Datasheet

ACPL-C79B, ACPL-C79A, ACPL-C790
Precision Miniature Isolation Amplifiers
Data Sheet
Description
Features
The ACPL-C79B/C79A/C790 isolation amplifiers are
designed for current and voltage sensing in electronic
power converters in applications including motor drives
and renewable energy systems. In a typical motor drive
implementation, current flows through an external
resistor and the resulting analog voltage drop is sensed by
the isolation amplifier. A differential output voltage that is
proportional to the current is created on the other side of
the optical isolation barrier.
• ±0.5% High Gain Accuracy (ACPL-C79B)
For general applications, the ACPL-C79A (±1% gain
tolerance) and the ACPL-C790 (±3% gain tolerance)
are recommended. For high precision applications, the
ACPL-C79B (±0.5% gain tolerance) can be used. The
product operates from a single 5 V supply and provides
excellent linearity and dynamic performance of 60 dB SNR.
With 200 kHz bandwidth, 1.6 µs fast response time, the
product captures transients in short circuit and overload
conditions. The high common-mode transient immunity
(15 kV/µs) of the ACPL-C79B/C79A/C790 provides the
precision and stability needed to accurately monitor
motor current in high noise motor control environments,
providing for smoother control (less “torque ripple”) in
various types of motor control applications.
• –40°C to +105°C Operating Temperature Range
Combined with superior optical coupling technology, the
ACPL-C79B/C79A/C790 implements sigma-delta (∑-∆)
analog-to-digital converter, chopper stabilized amplifiers, and a fully differential circuit topology to provide
unequaled isolation-mode noise rejection, low offset,
high gain accuracy and stability. This performance is
delivered in a compact, auto-insertable Stretched SO-8
(SSO-8) package that meets worldwide regulatory safety
standards.
• Current/Voltage Sensing in AC and Servo Motor Drives
• –50 ppm/°C Low Gain Drift
• 0.6 mV Input Offset Voltage
• 0.05% Excellent Linearity
• 60 dB SNR
• 200 kHz Wide Bandwidth
• 3 V to 5.5 V Wide Supply Range for Output Side
• Advanced Sigma-Delta (∑-∆) A/D Converter Technology
• Fully Differential Isolation Amplifier
• 15 kV/µs Common-Mode Transient Immunity
• Compact, Auto-Insertable Stretched SO-8 Package
• Safety and Regulatory Approvals:
– IEC/EN/DIN EN 60747-5-5: 1414 Vpeak working
insulation voltage
– UL 1577: 5000 Vrms/1min double protection rating
– CSA: Component Acceptance Notice #5
Applications
• Solar Inverters, Wind Turbine Inverters
• Industrial Process Control
• Data acquisition systems
• Switching Power Supply Signal Isolation
• General Purpose Analog Signal Isolation
• Traditional Current Transducer Replacements
Functional Diagram
VDD1 1
IDD1
IDD2
8 VDD2
VIN+ 2
+
+
7 VOUT+
VIN- 3
-
-
6 VOUT-
GND1 4
SHIELD
5 GND2
NOTE:
A 0.1 μF bypass capacitor must be connected
between pins 1 and 4 and between pins 5 and 8.
Figure 1.
CAUTION: It is advised that normal static precautions be taken in handling and assembly
of this component to prevent damage and/or degradation which may be induced by ESD.
Table 1. Pin Description
Pin No.
Symbol
Description
1
VDD1
Supply voltage for input side (4.5 V to 5.5 V), relative to GND1
2
VIN+
Positive input (± 200 mV recommended)
3
VIN–
Negative input (normally connected to GND1)
4
GND1
Input side ground
5
GND2
Output side ground
6
VOUT–
Negative output
7
VOUT+
Positive output
8
VDD2
Supply voltage for output side (3 V to 5.5 V), relative to GND2
Table 2. Ordering Information
ACPL-C79B/C79A/C790 is UL recognized with 5000 Vrms/1 minute rating per UL 1577.
Part number
ACPL-C79B
ACPL-C79A
ACPL-C790
Option
(RoHS Compliant)
-000E
-500E
Package
Stretched
SO-8
Surface
Mount
Tape &
Reel
X
X
X
IEC/EN/DIN EN
60747-5-5
Quantity
X
80 per tube
X
1000 per reel
To order, choose a part number from the part number column and combine with the desired option from the option
column to form an order entry.
Example:
ACPL-C79B-500E to order product of Surface Mount package in Tape and Reel packaging with IEC/EN/DIN EN 60747-5-5
Safety Approval and RoHS compliance.
Option datasheets are available. Contact your Avago sales representative or authorized distributor for information.
2
Package Outline Drawings
Stretched SO-8 Package (SSO-8)
RECOMMENDED LAND PATTERN
5.850 ± 0.50
(0.230 ± 0.010)
PART NUMBER
8
7
6
5
C79B
YWW
RoHS-COMPLIANCE
INDICATOR
DATE CODE
12.650 (0.5)
6.807 ± 0.127
(0.268 ± 0.005)
1.905 (0.1)
1
2
3
4
7°
3.180 ± 0.127
(0.125 ± 0.005)
0.381 ± 0.130
(0.015 ± 0.005)
45°
0.450
(0.018)
0.750 ± 0.250
(0.0295 ± 0.010)
1.590 ± 0.127
(0.063 ± 0.005)
0.200 ± 0.100
(0.008 ± 0.004)
11.50 ± 0.250
(0.453 ± 0.010)
1.270
(0.050) BSG
Dimensions in millimeters and (inches).
Lead coplanarity = 0.1 mm (0.004 inches).
Figure 2. SSO-8 package.
Recommended Pb-Free IR Profile
Recommended reflow condition as per JEDEC Standard, J-STD-020 (latest revision). Non-Halide Flux should be used.
Regulatory Information
The ACPL-C79B/C79A/C790 is approved by the following organizations:
IEC/EN/DIN EN 60747-5-5
Approved with Maximum Working Insulation Voltage VIORM = 1414 Vpeak.
UL
Approval under UL 1577, component recognition program up to VISO = 5000 Vrms/1min. File E55361.
CSA
Approval under CSA Component Acceptance Notice #5, File CA 88324.
3
Table 3. Insulation and Safety Related Specifications
Parameter
Symbol
Value
Units
Conditions
Minimum External Air Gap
(External Clearance)
L(101)
8.0
mm
Measured from input terminals to output terminals, shortest
distance through air
Minimum External Tracking
(External Creepage)
L(102)
8.0
mm
Measured from input terminals to output terminals, shortest
distance path along body
0.5
mm
Through insulation distance, conductor to conductor,
usually the direct distance between the photoemitter and
photodetector inside the optocoupler cavity
>175
V
DIN IEC 112/VDE 0303 Part 1
Minimum Internal Plastic Gap
(Internal Clearance)
Tracking Resistance
(Comparative Tracking Index)
CTI
Isolation Group
IIIa
Material Group (DIN VDE 0110, 1/89, Table 1)
Table 4. IEC/EN/DIN EN 60747-5-5 Insulation Characteristics [1]
Description
Symbol
Value
Installation classification per DIN VDE 0110/1.89, Table 1
for rated mains voltage ≤ 150 Vrms
for rated mains voltage ≤ 300 Vrms
for rated mains voltage ≤ 450 V rms
for rated mains voltage ≤ 600 Vrms
for rated mains voltage ≤ 1000 Vrms
I-IV
I-IV
I-IV
I-IV
I-III
Climatic Classification
55/105/21
Pollution Degree (DIN VDE 0110/1.89)
2
Units
Maximum Working Insulation Voltage
VIORM
1414
Vpeak
Input to Output Test Voltage, Method b
VIORM x 1.875 = VPR, 100% Production Test with tm = 1 sec,
Partial Discharge < 5 pC
VPR
2652
Vpeak
Input to Output Test Voltage, Method a
VIORM x 1.6 = VPR, Type and Sample Test, tm = 10 sec,
Partial Discharge < 5 pC
VPR
2262
Vpeak
Highest Allowable Overvoltage (Transient Overvoltage, tini = 60 sec)
VIOTM
8000
Vpeak
Safety-limiting values
(Maximum values allowed in the event of a failure)
Case Temperature
Input Current [2]
Output Power [2]
TS
IS,INPUT
PS,OUTPUT
175
230
600
°C
mA
mW
Insulation Resistance at TS, VIO = 500 V
RS
≥ 109
Ω
Notes:
1. Insulation characteristics are guaranteed only within the safety maximum ratings, which must be ensured by protective circuits within the
application.
2. Safety-limiting parameters are dependent on ambient temperature. The Input Current, IS,INPUT, derates linearly above 25°C free-air temperature at
a rate of 2.53 mA/°C; the Output Power, PS,OUTPUT, derates linearly above 25°C free-air temperature at a rate of 4 mW/°C.
4
Table 5. Absolute Maximum Ratings
Parameter
Symbol
Min.
Max.
Units
Storage Temperature
TS
–55
+125
°C
Ambient Operating Temperature
TA
–40
+105
°C
Supply Voltages
VDD1, VDD2
–0.5
6.0
V
Steady-State Input Voltage [1, 3]
VIN+, VIN–
–2
VDD1 + 0.5
V
Two-Second Transient Input Voltage [2]
VIN+, VIN–
–6
VDD1 + 0.5
V
Output Voltages
VOUT+, VOUT–
–0.5
VDD2 + 0.5
V
Lead Solder Temperature
260°C for 10 sec., 1.6 mm below seating plane
Notes:
1. DC voltage of up to –2 V on the inputs does not cause latch-up or damage to the device; tested at typical operating conditions.
2. Transient voltage of 2 seconds up to –6 V on the inputs does not cause latch-up or damage to the device; tested at typical operating conditions.
3. Absolute maximum DC current on the inputs = 100 mA, no latch-up or device damage occurs.
Table 6. Recommended Operating Conditions
Parameter
Symbol
Min.
Max.
Units
Ambient Operating Temperature
TA
–40
+105
°C
VDD1 Supply Voltage
VDD1
4.5
5.5
V
VDD2 Supply Voltage
VDD2
3
5.5
V
Input Voltage Range [1]
VIN+, VIN–
–200
+200
mV
Notes:
1. ±200 mV is the nominal input range. Full scale input range (FSR) is ±300 mV. Functional input range is ±2 V.
5
Table 7. Electrical Specifications
Unless otherwise noted, TA = –40°C to +105°C, VDD1 = 4.5 V to 5.5 V, VDD2 = 3 V to 5.5 V, VIN+ = –200 mV to +200 mV, and
VIN– = 0 V (single-ended connection).
Symbol
Min.
Typ.[1]
Max.
Unit
Test Conditions/Notes
Fig.
Input Offset Voltage
VOS
–1
0.6
2
mV
TA = 25°C
3, 4
Magnitude of Input Offset
Change vs. Temperature
|dVOS/dTA|
3
10
µV/°C
TA = –40°C to +105°C;
absolute value
5
Gain (ACPL-C79B, ±0.5%)
G0
8.16
8.2
8.24
V/V
TA = 25°C; Note 2
6, 7
Gain (ACPL-C79A, ±1%)
G1
8.12
8.2
8.28
V/V
TA = 25°C; Note 2
6, 7
Gain (ACPL-C790, ±3%)
G3
7.95
8.2
8.44
V/V
TA = 25°C; Note 2
6, 7
Magnitude of Gain
Change vs. Temperature
dG/dTA
–0.00041
V/V/°C
TA = –40°C to +105°C;
Note 3
8
Nonlinearity over ±200 mV
Input Voltage
NL200
0.05
%
VIN+ = –200 mV to +200 mV,
TA = 25°C; Note 2
9, 10
Magnitude of NL200
Change vs. Temperature
dNL200/dTA
0.0003
%/°C
TA = –40°C to +85°C
11
%/°C
TA = +85°C to +105°C
11
Nonlinearity over ±100 mV
Input Voltage
NL100
0.02
%
VIN+ = –100 mV to +100 mV,
TA = 25°C; Note 2
9, 10, 11
FSR
±300
mV
VIN = VIN+ – VIN–; Note 4
12
VIN+ = 0 V, VIN– = 0 V; Note 5
13
14
Parameter
DC CHARACTERISTICS
0.13
0.004
0.06
INPUTS AND OUTPUTS
Full-Scale Differential
Voltage Input Range
Input Bias Current
IIN+
–0.1
µA
Magnitude of IIN+ Change vs.
Temperature
dIIN+/dTA
–0.05
nA/°C
Equivalent Input Impedance
RIN
22
kΩ
VIN+ or VIN–, single-ended
Output Common-Mode Voltage
VOCM
1.23
V
VOUT+ or VOUT–; Note 6
Output Voltage Range
OVR
0 to 2.5
V
VOUT+ or VOUT–; Note 4
Output Short-Circuit Current
|IOSC|
11
mA
VOUT+ or VOUT–, shorted to GND2 or
VDD2
Output Resistance
ROUT
21
Ω
VOUT+ or VOUT–
Input DC Common-Mode
Rejection Ratio
CMRRIN
76
dB
Note 2
Signal-to-Noise Ratio
SNR
60
dB
VIN+ = 300 mVpp 10 kHz sine wave;
Note 7
15, 16
Signal-to-(Noise + Distortion)
Ratio
SNDR
56
dB
VIN+ = 300 mVpp 10 kHz sine wave;
Note 8
15, 16
Small-Signal Bandwidth (-3 dB)
f–3 dB
Input to Output
10%-10%
Propagation Delay
50%-50%
tPD10
1.6
2.3
µs
200 mV/µs step input
19
tPD50
2
2.6
µs
200 mV/µs step input
19
90%-90%
tPD90
2.6
3.3
µs
200 mV/µs step input
19
Output Rise/Fall Time (10%-90%)
tR/F
1.7
µs
Step input
19
Common Mode Transient
Immunity
CMTI
15
kV/μs
VCM = 1 kV, TA = 25°C; Note 2
Power Supply Rejection
PSR
–78
dB
1 Vpp 1 kHz sine wave ripple on VDD1,
differential output; Note 9
Input Side Supply Current
IDD1
13
18.5
mA
VIN+ = 400 mV; see Note 10
20
Output Side Supply Current
IDD2
7
12
mA
5 V supply
20
6.8
11
mA
3.3 V supply
20
–1
12
AC CHARACTERISTICS
140
10
200
kHz
17, 18
POWER SUPPLIES
6
Notes:
1. All Typical values are under Typical Operating Conditions at TA = 25°C, VDD1 = 5 V, VDD2 = 3.3 V.
2. See Definitions section.
3. Gain temperature drift can be normalized and expressed as Temperature Coefficient of Gain (TCG) of –50 ppm/°C.
4. When FSR is exceeded, outputs saturate.
5. Because of the switched-capacitor nature of the input sigma-delta converter, time-averaged values are shown.
6. Under Typical Operating Conditions, part-to-part variation ±0.04 V.
7. Under Typical Operating Conditions, part-to-part variation ±1 dB.
8. Under Typical Operating Conditions, part-to-part variation ±1 dB.
9. Ripple voltage applied to VDD1 with a 0.1 µF bypass capacitor connected; differential amplitude of the ripple outputs measured. See Definitions
section.
10. The input supply current decreases as the differential input voltage (VIN+ – VIN–) decreases.
Table 8. Package Characteristics
Parameter
Symbol
Min.
Input-Output Momentary
Withstand Voltage
VISO
5000
Resistance (Input-Output)
RI-O
Capacitance (Input-Output)
CI-O
Typ.
Max.
Unit
Test Condition
Note
Vrms
RH < 50%, t = 1 min., TA = 25°C
1, 2
>1012
Ω
VI-O = 500 VDC
3
0.5
pF
f = 1 MHz
3
Notes:
1. In accordance with UL 1577, each optocoupler is proof tested by applying an insulation test voltage ≥ 6000 Vrms for 1 second (leakage detection
current limit, II-O ≤ 5 µA). This test is performed before the 100% production test for partial discharge (method b) shown in IEC/EN/DIN EN 607475-5 Insulation Characteristic Table.
2. The Input-Output Momentary Withstand Voltage is a dielectric voltage rating that should not be interpreted as an input-output continuous
voltage rating. For the continuous voltage rating, refer to the IEC/EN/DIN EN 60747-5-5 insulation characteristics table and your equipment level
safety specification.
3. This is a two-terminal measurement: pins 1–4 are shorted together and pins 5–8 are shorted together.
7
Typical Performance Plots
1.0
1.0
0.9
0.9
0.8
0.8
0.7
VDD2 = 3.3V
0.6
OFFSET - mV
OFFSET - mV
Unless otherwise noted, TA = 25°C, VDD1 = 5 V, VDD2 = 3.3 V.
VDD2 = 5V
0.5
0.4
0.4
4.75
5
5.25
VDD1 - SUPPLY VOLTAGE - V
0.3
5.5
Figure 3. Input offset vs. supply VDD1.
1.0
8.24
0.9
8.23
0.7
0.6
0.5
4
4.5
VDD2 - SUPPLY VOLTAGE - V
5
5.5
8.21
8.20
VDD2 = 3.3V
VDD2 = 5V
8.19
8.18
0.4
8.17
0.3
-40
-20
0
20
40
60
TA - TEMPERATURE - °C
80
100
8.16
4.5
120
Figure 5. Input offset vs. temperature.
8.23
8.26
8.22
8.24
8.21
8.22
G - GAIN - V/V
8.28
8.20
VDD1 = 5V
8.19
8.17
8.14
4
4.5
VDD2 - SUPPLY VOLTAGE - V
Figure 7. Gain vs. supply VDD2.
5.5
8.18
8.16
3.5
5
5.25
VDD1 - SUPPLY VOLTAGE - V
8.20
8.18
3
4.75
Figure 6. Gain vs. supply VDD1.
8.24
8.16
3.5
8.22
GAIN - V/V
OFFSET - mV
3
Figure 4. Input offset vs. supply VDD2.
0.8
GAIN - V/V
VDD1 = 5V
0.6
0.5
0.3
4.5
8
0.7
5
5.5
8.12
-40
-20
0
20
40
60
TA - TEMPERATURE - °C
Figure 8. Gain vs. temperature.
80
100
120
0.050
0.045
0.045
NL200, VDD2 = 5V
0.040
NL - NONLINEARITY - %
NL - NONLINEARITY - %
0.050
NL200, VDD2 = 3.3V
0.035
0.030
0.025
NL100, VDD2 = 5V
0.020
NL100, VDD2 = 3.3V
0.010
4.5
4.75
5
5.25
VDD1 - SUPPLY VOLTAGE - V
0.025
NL100
0.020
3
3.5
4
4.5
VDD2 - SUPPLY VOLTAGE - V
5
5.5
3.0
2.5
VOUT - OUTPUT VOLTAGE - V
NL - LINEARITY - %
NL200
NL100
-20
0
20
40
60
TA - TEMPERATURE - °C
80
100
VOUT–
2.0
VOUT+
1.5
1.0
0.5
0.0
-0.5
-0.4
120
-0.3
-0.2
-0.1
0
0.1
0.2
VIN+ - INPUT VOLTAGE - V
0.3
0.4
Figure 12. Output voltage vs. input voltage.
25
VIN+ = 0 to 300mV
RIN - INPUT IMPEDANCE - kohm
IIN+ - INPUT CURRENT - µA
0.030
Figure 10. Nonlinearity vs. supply VDD2.
Figure 11. Nonlinearity vs. temperature.
-0.4
-0.2
0
0.2
VIN+ - INPUT VOLTAGE - V
Figure 13. Input current vs. input voltage.
9
NL200
0.035
0.010
5.5
Figure 9. Nonlinearity vs. supply VDD1.
20
15
10
5
0
-5
-10
-15
-20
-25
-30
-0.6
0.040
0.015
0.015
0.20
0.18
0.16
0.14
0.12
0.10
0.08
0.06
0.04
0.02
0.00
-40
VDD1 = 5V
0.4
0.6
24
23
22
21
-40
-20
0
20
40
60
TA - TEMPERATURE - °C
Figure 14. Input impedance vs. temperature.
80
100
120
64
64
61
58
SNDR
55
52
49
43
-20
0
20
40
60
TA - TEMPERATURE - °C
80
100
5
0
-5
-10
-15
-20
-25
-30
-35
-40
-45
-50
100
150
200
250
300
VIN+ - INPUT VOLTAGE - m Vpp
350
400
Figure 16. SNR, SNDR vs. input voltage.
180
150
120
90
60
30
0
-30
-60
-90
-120
-150
-180
100
PHASE - DEGREES
NORMALIZED GAIN - dB
1,000
10,000
100,000
FREQUENCY - Hz
1,000,000
tPD90
tPD50
tR/F
tPD10
tPD10, tPD50, tPD90: 200mV/µs step input; tR/F: step input
-20
0
20
40
60
TA - TEMPERATURE - °C
1,000
10,000
FREQUENCY - Hz
100,000
1,000,000
Figure 18. Phase frequency response.
IDD - SUPPLY CURRENT - mA
PROPAGATION DELAY - µs
49
40
100
120
Figure 17. Gain frequency response.
80
100
Figure 19. Propagation delay, output rise/fall time vs. temperature.
10
52
43
VIN+ = 300mVpp 10 kHz sine wave
Figure 15. SNR, SNDR vs. temperature.
3.0
2.8
2.6
2.4
2.2
2.0
1.8
1.6
1.4
1.2
1.0
-40
SNDR
55
46
46
40
-40
SNR
58
SNR, SNDR - dB
SNR, SNDR - dB
61
SNR
120
15
14
13
12
11
10
9
8
7
6
5
-0.4
IDD1 (VDD1 = 5V)
IDD2 (VDD2 = 5V)
IDD2 (VDD2 = 3.3V)
-0.3
-0.2
-0.1
0
0.1
0.2
VIN+ - INPUT VOLTAGE - V
Figure 20. Supply current vs. input voltage.
0.3
0.4
Definitions
Gain
Gain is defined as the slope of the best-fit line of differential output voltage (VOUT+ – VOUT–) vs. differential input
voltage (VIN+ – VIN–) over the nominal input range, with
offset error adjusted out.
Nonlinearity
Nonlinearity is defined as half of the peak-to-peak output
deviation from the best-fit gain line, expressed as a percentage of the full-scale differential output voltage.
Input DC Common Mode Rejection Ratio, CMRRIN
CMRRIN is defined as the ratio of the differential signal
gain (signal applied differentially between pins VOUT+ and
VOUT–) to the input side common-mode gain (input pins
tied together and the signal applied to both inputs with
respect to pin GND1), expressed in dB.
PSR is the ratio of differential amplitude of the ripple
outputs over power supply ripple voltage, referred to the
input, expressed in dB.
C1
0.1
µF
R3
***
U1
78L05
IN OUT
R5
10
VDD1
C2
0.1
µF
10.0 K
VDD2 (+5 V)
1
8
2
7
C3
47 nF 3
U2
C4
0.1 µF
+15 V
2.00 K
R2
6
4
GND1
HV-
Figure 21. Typical application circuit for motor phase current sensing.
5
ACPL-C79B/
ACPL-C79A/
ACPL-C790
C6
47 pF
GND2
C8
0.1 µF
GND2
–
U3
+ TL032A
R1
2.00 K
+ –
RSENSE
***
11
Power Supply Rejection, PSR
C5
47 pF
GATE DRIVE
CIRCUIT
***
CMTI is tested by applying an exponentially rising/falling
voltage step on pin 4 (GND1) with respect to pin 5 (GND2).
The rise time of the test waveform is set to approximately
50 ns. The amplitude of the step is adjusted until the differential output (VOUT+ – VOUT–) exhibits more than a 200
mV deviation from the average output voltage for more
than 1μs. The ACPL-C79B/C79A/C790 will continue to
function if more than 10 kV/μs common mode slopes are
applied, as long as the breakdown voltage limitations are
observed.
POSITIVE
FLOATING
SUPPLY
HV+
MOTOR
Common Mode Transient Immunity, CMTI, also known as Common Mode Rejection
C7
R4
0.1 µF
10.0 K
-15 V
GND2
GND2
VOUT
Application Information
Application Circuit
The typical application circuit is shown in Figure 21. A
floating power supply (which in many applications could
be the same supply that is used to drive the high-side
power transistor) is regulated to 5 V using a simple threeterminal voltage regulator (U1). The voltage from the
current sensing resistor, or shunt (RSENSE), is applied to
the input of the ACPL-C79B/C79A/C790 through an RC
anti-aliasing filter (R5 and C3). And finally, the differential
output of the isolation amplifier is converted to a groundreferenced single-ended output voltage with a simple
differential amplifier circuit (U3 and associated components). Although the application circuit is relatively
simple, a few recommendations should be followed to
ensure optimal performance.
Power Supplies and Bypassing
As mentioned above, an inexpensive 78L05 three-terminal
regulator can be used to reduce the gate-drive power
supply voltage to 5 V. To help attenuate high frequency
power supply noise or ripple, a resistor or inductor can
be used in series with the input of the regulator to form a
low-pass filter with the regulator’s input bypass capacitor.
The power supply for the isolation amplifier is most
often obtained from the same supply used to power the
power transistor gate drive circuit. If a dedicated supply
is required, in many cases it is possible to add an additional winding on an existing transformer. Otherwise,
some sort of simple isolated supply can be used, such as
a line powered transformer or a high-frequency DC-DC
converter.
As shown in Figure 21, 0.1 µF bypass capacitors (C2, C4)
should be located as close as possible to the pins of the
isolation amplifier. The bypass capacitors are required
because of the high-speed digital nature of the signals
inside the isolation amplifier. A 47 nF bypass capacitor
(C3) is also recommended at the input pins due to the
switched-capacitor nature of the input circuit. The input
bypass capacitor also forms part of the anti-aliasing filter,
which is recommended to prevent high-frequency noise
from aliasing down to lower frequencies and interfering
with the input signal. The input filter also performs an
important reliability function – it reduces transient spikes
from ESD events flowing through the current sensing
resistor.
12
PC Board Layout
The design of the printed circuit board (PCB) should follow
good layout practices, such as keeping bypass capacitors
close to the supply pins, keeping output signals away from
input signals, the use of ground and power planes, etc. In
addition, the layout of the PCB can also affect the isolation
transient immunity (CMTI) of the ACPL-C79B/C79A/C790,
due primarily to stray capacitive coupling between the
input and the output circuits. To obtain optimal CMTI
performance, the layout of the PC board should minimize
any stray coupling by maintaining the maximum possible
distance between the input and output sides of the circuit
and ensuring that any ground or power plane on the PC
board does not pass directly below or extend much wider
than the body of the ACPL-C79B/C79A/C790. Figure 22
shows an example PCB layout.
TO GND1 TO VDD1
TO VDD2 TO GND2
C4
C2
U2
TO RSENSE+
R5
VOUT+
VOUT–
C3
TO RSENSE–
Note: Drawing not to scale
ACPL-C79B/C79A/C790
Figure 22. Example printed circuit board layout.
Shunt Resistor Selection
The current sensing resistor should have low resistance (to
minimize power dissipation), low inductance (to minimize
di/dt induced voltage spikes which could adversely
affect operation), and reasonable tolerance (to maintain
overall circuit accuracy). Choosing a particular value for
the resistor is usually a compromise between minimizing power dissipation and maximizing accuracy. Smaller
sense resistance decreases power dissipation, while larger
sense resistance can improve circuit accuracy by utilizing
the full input range of the ACPL-C79B/C79A/C790.
The first step in selecting a sense resistor is determining
how much current the resistor will be sensing. The graph in
Figure 23 shows the RMS current in each phase of a threephase induction motor as a function of average motor
output power (in horsepower, hp) and motor drive supply
voltage. The maximum value of the sense resistor is determined by the current being measured and the maximum
recommended input voltage of the isolation amplifier. The
maximum sense resistance can be calculated by taking
the maximum recommended input voltage and dividing
by the peak current that the sense resistor should see
during normal operation. For example, if a motor will have
a maximum RMS current of 10 A and can experience up
to 50% overloads during normal operation, then the peak
current is 21.1 A (=10 x 1.414 x 1.5). Assuming a maximum
input voltage of 200 mV, the maximum value of sense resistance in this case would be about 10 mΩ.
The maximum average power dissipation in the sense
resistor can also be easily calculated by multiplying the
sense resistance times the square of the maximum RMS
current, which is about 1 W in the previous example. If
the power dissipation in the sense resistor is too high, the
resistance can be decreased below the maximum value
to decrease power dissipation. The minimum value of the
sense resistor is limited by precision and accuracy requirements of the design. As the resistance value is reduced,
the output voltage across the resistor is also reduced,
which means that the offset and noise, which are fixed,
become a larger percentage of the signal amplitude. The
selected value of the sense resistor will fall somewhere
between the minimum and maximum values, depending
on the particular requirements of a specific design.
When sensing currents large enough to cause significant
heating of the sense resistor, the temperature coefficient
(tempco) of the resistor can introduce nonlinearity due to
the signal dependent temperature rise of the resistor. The
effect increases as the resistor-to-ambient thermal resisMOTOR OUTPUT POWER - HORSEPOWER
40
440 V
380 V
220 V
120 V
35
30
25
20
15
10
5
0
0
5
15
10
25
20
MOTOR PHASE CURRENT - A (rms)
30
35
Figure 23. Motor output horsepower vs. motor phase current and supply
voltage.
13
tance increases. This effect can be minimized by reducing
the thermal resistance of the current sensing resistor or
by using a resistor with a lower tempco. Lowering the
thermal resistance can be accomplished by repositioning the current sensing resistor on the PC board, by using
larger PC board traces to carry away more heat, or by
using a heat sink.
For a two-terminal current sensing resistor, as the value of
resistance decreases, the resistance of the leads become a
significant percentage of the total resistance. This has two
primary effects on resistor accuracy. First, the effective
resistance of the sense resistor can become dependent
on factors such as how long the leads are, how they are
bent, how far they are inserted into the board, and how far
solder wicks up the leads during assembly (these issues
will be discussed in more detail shortly). Second, the leads
are typically made from a material, such as copper, which
has a much higher tempco than the material from which
the resistive element itself is made, resulting in a higher
tempco overall.
Both of these effects are eliminated when a four-terminal
current sensing resistor is used. A four-terminal resistor has
two additional terminals that are Kelvin connected directly
across the resistive element itself; these two terminals are
used to monitor the voltage across the resistive element
while the other two terminals are used to carry the load
current. Because of the Kelvin connection, any voltage
drops across the leads carrying the load current should
have no impact on the measured voltage.
When laying out a PC board for the current sensing
resistors, a couple of points should be kept in mind. The
Kelvin connections to the resistor should be brought
together under the body of the resistor and then run very
close to each other to the input of the ACPL-C79B/C79A/
C790; this minimizes the loop area of the connection and
reduces the possibility of stray magnetic fields from interfering with the measured signal. If the sense resistor is not
located on the same PC board as the isolation amplifier
circuit, a tightly twisted pair of wires can accomplish the
same thing.
Also, multiple layers of the PC board can be used to
increase current carrying capacity. Numerous platedthrough vias should surround each non-Kelvin terminal of
the sense resistor to help distribute the current between
the layers of the PC board. The PC board should use 2 or
4 oz. copper for the layers, resulting in a current carrying
capacity in excess of 20 A. Making the current carrying
traces on the PC board fairly large can also improve the
sense resistor’s power dissipation capability by acting as a
heat sink. Liberal use of vias where the load current enters
and exits the PC board is also recommended.
Shunt Resistor Connections
The typical method for connecting the ACPL-C79B/C79A/
C790 to the current sensing resistor is shown in Figure 21.
VIN+ (pin 2) is connected to the positive terminal of the
sense resistor, while VIN– (pin 3) is shorted to GND1 (pin
4), with the power-supply return path functioning as the
sense line to the negative terminal of the current sense
resistor. This allows a single pair of wires or PC board traces
to connect the isolation amplifier circuit to the sense
resistor. By referencing the input circuit to the negative
side of the sense resistor, any load current induced noise
transients on the resistor are seen as a common-mode
signal and will not interfere with the current-sense signal.
This is important because the large load currents flowing
through the motor drive, along with the parasitic inductances inherent in the wiring of the circuit, can generate
both noise spikes and offsets that are relatively large
compared to the small voltages that are being measured
across the current sensing resistor.
If the same power supply is used both for the gate drive
circuit and for the current sensing circuit, it is very important
that the connection from GND1 of the ACPL-C79B/C79A/
C790 to the sense resistor be the only return path for supply
current to the gate drive power supply in order to eliminate
potential ground loop problems. The only direct connection between the ACPL-C79B/C79A/C790 circuit and the
gate drive circuit should be the positive power supply line.
Differential Input Connection
The differential analog inputs of the ACPL-C79B/C79A/
C790 are implemented with a fully-differential, switchedcapacitor circuit. In the typical application circuit (Figure
21), the isolation amplifier is connected in a single-ended
input mode. Given the fully differential input structure,
a differential input connection method (balanced input
mode as shown in Figure 24) is recommended to achieve
better performance. The input currents created by the
switching actions on both of the pins are balanced on
the filter resistors and cancelled out each other. Any noise
induced on one pin will be coupled to the other pin by the
capacitor C and creates only common mode noise which
is rejected by the device. Typical value for Ra and Rb is
10 Ω and 22 nF for C.
For product information and a complete list of distributors, please go to our web site:
5V
+Input
–Input
VDD1
Ra
Rb
VIN+
C
VIN–
ACPL-C79B/
ACPL-C79A/
ACPL-C790
GND1
Figure 24. Simplified differential input connection diagram.
Output Side
The op-amp used in the external post-amplifier circuit
should be of sufficiently high precision so that it does not
contribute a significant amount of offset or offset drift
relative to the contribution from the isolation amplifier.
Generally, op-amps with bipolar input stages exhibit
better offset performance than op-amps with JFET or
MOSFET input stages.
In addition, the op-amp should also have enough
bandwidth and slew rate so that it does not adversely
affect the response speed of the overall circuit. The postamplifier circuit includes a pair of capacitors (C5 and C6)
that form a single-pole low-pass filter; these capacitors
allow the bandwidth of the post-amp to be adjusted independently of the gain and are useful for reducing the
output noise from the isolation amplifier.
The gain-setting resistors in the post-amp should have a
tolerance of 1% or better to ensure adequate CMRR and
adequate gain tolerance for the overall circuit. Resistor
networks can be used that have much better ratio tolerances than can be achieved using discrete resistors. A
resistor network also reduces the total number of compo­
nents for the circuit as well as the required board space.
Voltage Sensing
The ACPL-C79B/C79A/C790 can also be used to isolate
signals with amplitudes larger than its recommended
input range with the use of a resistive voltage divider at
its input. The only restrictions are that the impedance of
the divider be relatively small (less than 1 kΩ) so that the
input resistance (22 kΩ) and input bias current (0.1 µA)
do not affect the accuracy of the measurement. An input
bypass capacitor is still required, although the 10 Ω series
damping resistor is not (the resistance of the voltage
divider provides the same function). The low-pass filter
formed by the divider resistance and the input bypass
capacitor may limit the achievable bandwidth.
www.avagotech.com
Avago, Avago Technologies, and the A logo are trademarks of Avago Technologies in the United States and other countries.
Data subject to change. Copyright © 2005-2013 Avago Technologies. All rights reserved.
AV02-2460EN - May 13, 2013
Similar pages