AD AD8031AR-REEL7 2.7 v, 800 ua, 80 mhz rail-to-rail i/o amplifier Datasheet

a
FEATURES
Low Power
Supply Current 800 ␮A/Amplifier
Fully Specified at +2.7 V, +5 V and ⴞ5 V Supplies
High Speed and Fast Settling on +5 V
80 MHz –3 dB Bandwidth (G = +1)
30 V/␮s Slew Rate
125 ns Settling Time to 0.1%
Rail-to-Rail Input and Output
No Phase Reversal with Input 0.5 V Beyond Supplies
Input CMVR Extends Beyond Rails by 200 mV
Output Swing to Within 20 mV of Either Rail
Low Distortion
–62 dB @ 1 MHz, VO = 2 V p-p
–86 dB @ 100 kHz, VO = 4.6 V p-p
Output Current: 15 mA
High Grade Option
VOS (max) = 1.5 mV
APPLICATIONS
High-Speed Battery-Operated Systems
High Component Density Systems
Portable Test Instruments
A/D Buffer
Active Filters
High-Speed Set-and-Demand Amplifier
2.7 V, 800 ␮A, 80 MHz
Rail-to-Rail I/O Amplifiers
AD8031/AD8032
CONNECTION DIAGRAMS
8-Lead Plastic DIP (N)
and SOIC (R) Packages
NC 1
AD8031
8-Lead Plastic DIP (N),
SOIC (R) and ␮SOIC (RM)
Packages
8 NC
OUT1 1
–IN 2
7 +VS
–IN1 2
7 OUT2
+IN 3
6 OUT
+IN1 3
6 –IN2
–VS 4
5 NC
–VS 4
8 +VS
AD8032
5 +IN2
NC = NO CONNECT
5-Lead Plastic Surface Mount Package
SOT-23-5 (RT-5)
VOUT 1
AD8031
5
+VS
4
–IN
–VS 2
+IN 3
(Not to Scale)
to high-speed systems where component density requires lower
power dissipation. The AD8031/AD8032 are available in 8-lead
plastic DIP and SOIC packages and will operate over the industrial temperature range of –40°C to +85°C. The AD8031A is also
available in the space-saving 5-lead SOT-23-5 package and the
AD8032A is available in AN 8-lead µSOIC package.
The products have true single supply capability with rail-to-rail
input and output characteristics and are specified for +2.7 V,
+5 V and ±5 V supplies. The input voltage range can extend to
500 mV beyond each rail. The output voltage swings to within
20 mV of each rail providing the maximum output dynamic range.
The AD8031/AD8032 also offer excellent signal quality for only
800 µA of supply current per amplifier; THD is –62 dBc with a
2 V p-p, 1 MHz output signal and –86 dBc for a 100 kHz, 4.6 V p-p
signal on +5 V supply. The low distortion and fast settling time
make them ideal as buffers to single supply, A-to-D converters.
1V/Div
The AD8031 (single) and AD8032 (dual) single supply voltage
feedback amplifiers feature high-speed performance with 80 MHz
of small signal bandwidth, 30 V/µs slew rate and 125 ns settling
time. This performance is possible while consuming less than
4.0 mW of power from a single +5 V supply. These features
increase the operation time of high speed battery-powered
systems without compromising dynamic performance.
1V/Div
GENERAL DESCRIPTION
2ms/Div
2ms/Div
Input VIN
Output VOUT
+5V
VOUT
VIN
1kV
1.7pF
+2.5V
Operating on supplies from +2.7 V to +12 V and dual supplies up to
±6 V, the AD8031/AD8032 are ideal for a wide range of applications,
from battery-operated systems with large bandwidth requirements
Circuit Diagram
Figure 1. Rail-to-Rail Performance at 100 kHz
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1999
AD8031/AD8032–SPECIFICATIONS
+2.7 V Supply
(@ T A = +25ⴗC, VS = +2.7 V, RL = 1 k⍀ to +1.35 V, RF = 2.5 k⍀ unless otherwise noted)
Parameter
Conditions
DYNAMIC PERFORMANCE
–3 dB Small Signal Bandwidth
Slew Rate
Settling Time to 0.1%
G = +1, VO < 0.4 V p-p
G = –1, VO = 2 V Step
G = –1, VO = 2 V Step, C L = 10 pF
AD8031A/AD8032A
Min
Typ
Max
AD8031B/AD8032B
Min
Typ
Max
54
25
54
25
DISTORTION/NOISE PERFORMANCE
Total Harmonic Distortion
fC = 1 MHz, VO = 2 V p-p, G = +2
fC = 100 kHz, VO = 2 V p-p, G = +2
Input Voltage Noise
f = 1 kHz
Input Current Noise
f = 100 kHz
f = 1 kHz
Crosstalk (AD8032 Only)
f = 5 MHz
80
30
125
–62
–86
15
2.4
5
–60
Units
80
30
125
MHz
V/µs
ns
–62
–86
15
2.4
5
–60
dBc
dBc
nV/√Hz
pA/√Hz
pA/√Hz
dB
DC PERFORMANCE
Input Offset Voltage
VCM =
V CC
2
; VOUT = 1.35 V
TMIN to T MAX
Offset Drift
VCM =
Input Bias Current
VCM =
V CC
2
V CC
2
±1
±6
± 0.5
± 1.5
mV
±6
± 10
± 1.6
± 2.5
mV
10
; VOUT = 1.35 V
0.45
2
0.45
2
µA
50
2.2
500
50
2.2
500
µA
nA
TMIN to T MAX
Input Offset Current
Open Loop Gain
VCM =
V CC
2
; VOUT = 0.35 V to 2.35 V
TMIN to T MAX
76
Input Common-Mode Voltage Range
80
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current per Amplifier
Power Supply Rejection Ratio
76
VCM = 0 V to 2.7 V
VCM = 0 V to 1.55 V
46
58
80
40
280
1.6
–0.5 to
+3.2
–0.2 to
+2.9
64
74
46
58
dB
40
280
1.6
–0.5 to
+3.2
–0.2 to
+2.9
64
74
3.4
RL = 10 kΩ
+0.05
+2.6
+0.15
+2.55
RL = 1 kΩ
Sourcing
Sinking
G = +2 (See Figure 41)
+0.02
+2.68
+0.08
+2.6
15
21
–34
15
+2.7
750
VS – = 0 V to –1 V or
VS + = +2.7 V to +3.7 V
75
dB
74
Differential Input Voltage
OUTPUT CHARACTERISTICS
Output Voltage Swing Low
Output Voltage Swing High
Output Voltage Swing Low
Output Voltage Swing High
Output Current
Short Circuit Current
10
74
INPUT CHARACTERISTICS
Common-Mode Input Resistance
Differential Input Resistance
Input Capacitance
Input Voltage Range
Common-Mode Rejection Ratio
µV/°C
; VOUT = 1.35 V
86
V
3.4
+0.05
+2.6
+0.15
+2.55
+12
1250
MΩ
kΩ
pF
+0.02
+2.68
+0.08
+2.6
15
21
–34
15
+2.7
750
75
86
V
dB
dB
V
V
V
V
V
mA
mA
mA
pF
+12
1250
V
µA
dB
Specifications subject to change without notice.
–2–
REV. B
AD8031/AD8032
SPECIFICATIONS
+5 V Supply
(@ TA = +25ⴗC, VS = +5 V, RL = 1 k⍀ to +2.5 V, RF = 2.5 k⍀ unless otherwise noted)
Parameter
Conditions
DYNAMIC PERFORMANCE
–3 dB Small Signal Bandwidth
Slew Rate
Settling Time to 0.1%
G = +1, VO < 0.4 V p-p
G = –1, VO = 2 V Step
G = –1, VO = 2 V Step, C L = 10 pF
AD8031A/AD8032A
Min
Typ
Max
AD8031B/AD8032B
Min
Typ
Max
54
27
54
27
DISTORTION/NOISE PERFORMANCE
Total Harmonic Distortion
fC = 1 MHz, VO = 2 V p-p, G = +2
fC = 100 kHz, VO = 2 V p-p, G = +2
Input Voltage Noise
f = 1 kHz
Input Current Noise
f = 100 kHz
f = 1 kHz
Differential Gain
RL = 1 kΩ
Differential Phase
RL = 1 kΩ
Crosstalk (AD8032 Only)
f = 5 MHz
80
32
125
–62
–86
15
2.4
5
0.17
0.11
–60
Units
80
32
125
MHz
V/µs
ns
–62
–86
15
2.4
5
0.17
0.11
–60
dBc
dBc
nV/√Hz
pA/√Hz
pA/√Hz
%
Degrees
dB
DC PERFORMANCE
Input Offset Voltage
VCM =
V CC
2
; VOUT = 2.5 V
TMIN to T MAX
Offset Drift
VCM =
Input Bias Current
VCM =
V CC
2
V CC
2
±1
±6
± 0.5
± 1.5
mV
±6
± 10
± 1.6
± 2.5
mV
5
; VOUT = 2.5 V
0.45
1.2
0.45
1.2
µA
50
2.0
350
50
2.0
250
µA
nA
TMIN to T MAX
Input Offset Current
Open Loop Gain
VCM =
V CC
2
; VOUT = 1.5 V to 3.5 V
TMIN to T MAX
76
Input Common-Mode Voltage Range
82
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current per Amplifier
Power Supply Rejection Ratio
VCM = 0 V to 5 V
VCM = 0 V to 3.8 V
56
66
82
RL = 10 kΩ
+0.05
+4.95
+0.2
+4.8
RL = 1 kΩ
Sourcing
Sinking
G = +2 (See Figure 41)
40
280
1.6
–0.5 to
+5.5
–0.2 to
+5.2
70
80
56
66
+0.02
+4.98
+0.1
+4.9
15
28
–46
15
+2.7
800
VS – = 0 V to –1 V or
VS + = +5 V to +6 V
75
–3–
dB
dB
40
280
1.6
–0.5 to
+5.5
–0.2 to
+5.2
70
80
3.4
Specifications subject to change without notice.
REV. B
76
74
Differential Input Voltage
OUTPUT CHARACTERISTICS
Output Voltage Swing Low
Output Voltage Swing High
Output Voltage Swing Low
Output Voltage Swing High
Output Current
Short Circuit Current
5
74
INPUT CHARACTERISTICS
Common-Mode Input Resistance
Differential Input Resistance
Input Capacitance
Input Voltage Range
Common-Mode Rejection Ratio
µV/°C
; VOUT = 2.5 V
86
V
3.4
+0.05
+4.95
+0.2
+4.8
+12
1400
MΩ
kΩ
pF
+0.02
+4.98
+0.1
+4.9
15
28
–46
15
+2.7
800
75
86
V
dB
dB
V
V
V
V
V
mA
mA
mA
pF
+12
1400
V
µA
dB
AD8031/AD8032–SPECIFICATIONS
ⴞ5 V Supply (@ T = +25ⴗC, V = ⴞ5 V, R = 1 k⍀ to 0 V, R
A
S
L
F
= 2.5 k⍀ unless otherwise noted)
Parameter
Conditions
DYNAMIC PERFORMANCE
–3 dB Small Signal Bandwidth
Slew Rate
Settling Time to 0.1%
G = +1, VO < 0.4 V p-p
G = –1, VO = 2 V Step
G = –1, VO = 2 V Step, C L = 10 pF
AD8031A/AD8032A
Min
Typ
Max
AD8031B/AD8032B
Min
Typ
Max
54
30
54
30
DISTORTION/NOISE PERFORMANCE
Total Harmonic Distortion
fC = 1 MHz, VO = 2 V p-p, G = +2
fC = 100 kHz, VO = 2 V p-p, G = +2
Input Voltage Noise
f = 1 kHz
Input Current Noise
f = 100 kHz
f = 1 kHz
Differential Gain
RL = 1 kΩ
Differential Phase
RL = 1 kΩ
Crosstalk (AD8032 Only)
f = 5 MHz
DC PERFORMANCE
Input Offset Voltage
Offset Drift
Input Bias Current
Input Offset Current
Open Loop Gain
–62
–86
15
2.4
5
0.15
0.15
–60
±1
±6
5
0.45
VCM = 0 V; V OUT = 0 V
TMIN to T MAX
VCM = 0 V; V OUT = 0 V
VCM = 0 V; V OUT = 0 V
TMIN to T MAX
VCM = 0 V; V OUT = ± 2 V
TMIN to T MAX
76
74
INPUT CHARACTERISTICS
Common-Mode Input Resistance
Differential Input Resistance
Input Capacitance
Input Voltage Range
Input Common-Mode Voltage Range
Common-Mode Rejection Ratio
80
35
125
VCM = –5 V to +5 V
VCM = –5 V to +3.5 V
60
66
50
80
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current per Amplifier
Power Supply Rejection Ratio
80
35
125
MHz
V/µs
ns
–62
–86
15
2.4
5
0.15
0.15
–60
dBc
dBc
nV/√Hz
pA/√Hz
pA/√Hz
%
Degrees
dB
± 0.5
± 1.6
5
0.45
1.2
2.0
350
76
74
40
280
1.6
–5.5 to
+5.5
–5.2 to
+5.2
80
90
Differential/Input Voltage
OUTPUT CHARACTERISTICS
Output Voltage Swing Low
Output Voltage Swing High
Output Voltage Swing Low
Output Voltage Swing High
Output Current
Short Circuit Current
±6
± 10
60
66
50
80
–4.94
+4.94
–4.7
+4.7
RL = 1 kΩ
Sourcing
Sinking
G = +2 (See Figure 41)
–4.98
+4.98
–4.85
+4.75
15
35
–50
15
± 1.35
900
VS – = –5 V to –6 V or
VS + = +5 V to +6 V
76
86
1.2
2.0
250
V
–4.98
+4.98
–4.85
+4.75
15
35
–50
15
± 1.35
900
76
86
mV
mV
µV/°C
µA
µA
nA
dB
dB
MΩ
kΩ
pF
3.4
–4.94
+4.94
–4.7
+4.7
±6
1600
± 1.5
± 2.5
40
280
1.6
–5.5 to
+5.5
–5.2 to
+5.2
80
90
3.4
RL = 10 kΩ
Units
V
dB
dB
V
V
V
V
V
mA
mA
mA
pF
±6
1600
V
µA
dB
Specifications subject to change without notice.
–4–
REV. B
AD8031/AD8032
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +12.6 V
Internal Power Dissipation2
Plastic DIP Package (N) . . . . . . . . . . . . . . . . . . . 1.3 Watts
Small Outline Package (R) . . . . . . . . . . . . . . . . . . 0.8 Watts
µSOIC (RM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.6 Watts
SOT-23-5 (RT) . . . . . . . . . . . . . . . . . . . . . . . . . . 0.5 Watts
Input Voltage (Common-Mode) . . . . . . . . . . . . . ±VS ± 0.5 V
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . ± 3.4 V
Output Short Circuit Duration
. . . . . . . . . . . . . . . . . . . . . . Observe Power Derating Curves
Storage Temperature Range (N, R, RM, RT)
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +125°C
Lead Temperature Range (Soldering 10 sec) . . . . . . . . +300°C
temperature of the plastic, approximately +150°C. Exceeding
this limit temporarily may cause a shift in parametric performance due to a change in the stresses exerted on the die by
the package. Exceeding a junction temperature of +175°C for
an extended period can result in device failure.
While the AD8031/AD8032 are internally short circuit protected, this may not be sufficient to guarantee that the maximum junction temperature (+150°C) is not exceeded under
all conditions. To ensure proper operation, it is necessary to
observe the maximum power derating curves shown in Figure 2.
2.0
MAXIMUM POWER DISSIPATION – Watts
ABSOLUTE MAXIMUM RATINGS 1
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
Specification is for the device in free air:
8-Lead Plastic DIP Package: θJA = 90°C/W.
8-Lead SOIC Package: θJA = 155°C/W.
8-Lead µSOIC Package: θ JA = 200°C/W.
5-Lead SOT-23-5 Package: θ JA = 240°C/W.
MAXIMUM POWER DISSIPATION
8-LEAD PLASTIC
DIP PACKAGE
TJ = +1508C
1.5
8-LEAD SOIC PACKAGE
1.0
8-LEAD mSOIC
SOT-23-5
0.5
0
–50 –40 –30 –20 –10 0 10 20 30 40 50 60
AMBIENT TEMPERATURE – 8C
The maximum power that can be safely dissipated by the
AD8031/AD8032 are limited by the associated rise in junction
temperature. The maximum safe junction temperature for plastic encapsulated devices is determined by the glass transition
70
80
90
Figure 2. Maximum Power Dissipation vs. Temperature
ORDERING GUIDE
Model
Temperature
Range
Package
Descriptions
Package
Options
Brand
Code
AD8031AN
AD8031AR
AD8031AR-REEL
AD8031AR-REEL7
AD8031ART-REEL
AD8031ART-REEL7
AD8031BN
AD8031BR
AD8031BR-REEL
AD8031BR-REEL7
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
8-Lead Plastic DIP
8-Lead SOIC
13" Tape and Reel
7" Tape and Reel
13" Tape and Reel
7" Tape and Reel
8-Lead Plastic DIP
8-Lead SOIC
13" Tape and Reel
7" Tape and Reel
N-8
SO-8
SO-8
SO-8
RT-5
RT-5
N-8
SO-8
SO-8
SO-8
H0A
H0A
AD8032AN
AD8032AR
AD8032AR-REEL
AD8032AR-REEL7
AD8032ARM
AD8032ARM-REEL
AD8032ARM-REEL7
AD8032BN
AD8032BR
AD8032BR-REEL
AD8032BR-REEL7
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
8-Lead Plastic DIP
8-Lead SOIC
13" Tape and Reel
7" Tape and Reel
8-Lead µSOIC
13" Tape and Reel
7" Tape and Reel
8-Lead Plastic DIP
8-Lead SOIC
13" Tape and Reel
7" Tape and Reel
N-8
SO-8
SO-8
SO-8
RM-8
RM-8
RM-8
N-8
SO-8
SO-8
SO-8
H9A
H9A
H9A
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD8031/AD8032 feature proprietary ESD protection circuitry, permanent damage
may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
REV. B
–5–
WARNING!
ESD SENSITIVE DEVICE
AD8031/AD8032–Typical Performance Characteristics
90
800
80
600
INPUT BIAS CURRENT – nA
NUMBER OF PARTS IN BIN
N = 250
70
60
50
40
30
20
400
200
VS = +2.7V
0
VS = +5V
VS = +10V
–200
–400
–600
10
0
–5
–4
–3
–2
–1
0
1
VOS – mV
2
3
4
–800
5
0
Figure 3. Typical VOS Distribution @ VS = 5 V
1
2
3
4
5
6
7
COMMON-MODE VOLTAGE – V
8
9
10
Figure 6. Input Bias Current vs. Common-Mode Voltage
0
2.5
VS = +5V
–0.1
OFFSET VOLTAGE – mV
OFFSET VOLTAGE – mV
2.3
2.1
VS = +5V
1.9
VS = 65V
1.7
1.5
–40 –30 –20 –10 0
–0.2
–0.3
–0.4
–0.5
–0.6
10 20 30 40 50
TEMPERATURE – 8C
60
70
80
0
90
Figure 4. Input Offset Voltage vs. Temperature
0.5
1
1.5
2
2.5
3
3.5
COMMON-MODE VOLTAGE – V
1
VS = +5V
SUPPLY CURRENT/AMPLIFIER – mA
INPUT BIAS – mA
5
1000
0.85
0.8
0.75
0.7
0.65
0.6
0.55
0.5
–40 –30 –20 –10 0
4.5
Figure 7. VOS vs. Common-Mode Voltage
0.95
0.9
4
10 20 30 40 50
TEMPERATURE – 8C
60
70
80
950
900
850
Figure 5. Input Bias Current vs. Temperature
+IS, VS = +5V
800
750
+IS, VS = +2.7V
700
650
600
–40 –30 –20 –10 0
90
6IS, VS = 65V
10 20 30 40 50
TEMPERATURE – 8C
60
70
80
90
Figure 8. Supply Current vs. Temperature
–6–
REV. B
AD8031/AD8032
1.2
VCC = +2.7V
DIFFERENCE FROM VEE – Volts
DIFFERENCE FROM VCC – Volts
0
–0.5
VCC = +5V
–1
VCC
–1.5
VCC = +10V
VOUT
VIN
RLOAD
VEE
–2
VCC
2
VCC
1
VCC = +10V
VOUT
RLOAD
VIN
0.8
VEE
0.6
VCC
2
VCC = +5V
0.4
0.2
VCC = +2.7V
–2.5
100
1k
RLOAD – Ohms
10k
Figure 9. +Output Saturation Voltage vs. RLOAD @ +85 °C
DIFFERENCE FROM VCC – Volts
–0.5
VCC = +5V
VCC
–1.5
VCC = +10V
VOUT
VIN
RLOAD
VEE
–2
–2.5
100
VCC
2
1k
RLOAD – Ohms
10k
DIFFERENCE FROM VEE – Volts
DIFFERENCE FROM VCC – Volts
VEE
0.6
VCC
2
VCC = +5V
0.4
1k
RLOAD – Ohms
10k
1.2
VCC = +5V
VCC
–1.5
VCC = +10V
VOUT
VIN
RLOAD
VEE
–2
1k
RLOAD – Ohms
VCC
2
VCC
1
VCC = +10V
VOUT
RLOAD
VIN
0.8
VEE
0.6
VCC
2
VCC = +5V
0.4
0.2
VCC = +2.7V
0
100
10k
Figure 11. +Output Saturation Voltage vs. RLOAD @ –40 °C
REV. B
VOUT
RLOAD
VIN
0.8
Figure 13. –Output Saturation Voltage vs. RLOAD @ +25 °C
–0.5
–2.5
100
VCC = +10V
VCC = +2.7V
0
100
VCC = +2.7V
–1
VCC
1
0.2
Figure 10. +Output Saturation Voltage vs. RLOAD @ +25 °C
0
10k
1.2
VCC = +2.7V
–1
1k
RLOAD – Ohms
Figure 12. –Output Saturation Voltage vs. RLOAD @ +85 °C
DIFFERENCE FROM VCC – Volts
0
0
100
1k
RLOAD – Ohms
10k
Figure 14. –Output Saturation Voltage vs. RLOAD @ –40 °C
–7–
AD8031/AD8032–Typical Performance Characteristics
110
VS = +5V
105
500mV
100
INPUT BIAS CURRENT – mA
–AOL
95
GAIN – dB
90
+AOL
85
80
75
70
1V
100
10
90
0
–10
VS = +5V
10
0%
500mV
65
–1.5
60
0
2k
4k
6k
RLOAD – Ohms
8k
10k
Figure 15. Open-Loop Gain (A OL) vs. RLOAD
0.05
DIFF GAIN – %
VS = +5V
RL = 1kV
84
–AOL
0.00
–0.05
–0.10
82
–0.15
+AOL
DIFF PHASE – Degrees
80
78
76
–40 –30 –20 –10 0
10 20 30 40 50
TEMPERATURE – 8C
60
70
80
90
Figure 16. Open-Loop Gain (A OL) vs. Temperature
2nd
3rd
4th
5th
6th
7th
8th
9th
10th 11th
1st
2nd
3rd
4th
5th
6th
7th
8th
9th
10th 11th
0.05
0.00
–0.05
–0.10
Figure 19. Differential Gain and Phase @ VS = ± 5 V;
RL = 1 k⍀
110
100
VS = +5V
VS = +5V
INPUT VOLTAGE NOISE – nV/ Hz
RLOAD = 10kV
100
90
RLOAD = 1kV
80
70
60
50
1st
0.10
0
0.5
1
1.5
2
2.5
3
VOUT – V
3.5
4
4.5
VOLTAGE NOISE
10
10
3
1
CURRENT NOISE
Figure 17. Open-Loop Gain (AOL) vs. VOUT
0.1
1
0.3
10
5
100
30
100
1k
10k
100k
FREQUENCY – Hz
1M
INPUT CURRENT NOISE – pA/ Hz
GAIN – dB
6.5
Figure 18. Differential Input Overvoltage I-V
Characteristics
86
AOL – dB
0.5
2.5
4.5
INPUT VOLTAGE – Volts
10M
Figure 20. Input Voltage Noise vs. Frequency
–8–
REV. B
AD8031/AD8032
5
VS = +5V
G = +1
RL = 1kV
40
30
GAIN
2
20
1
10
0
0
PHASE – Degree
NORMALIZED GAIN – dB
3
–1
–2
–3
–4
–5
0.1
1
10
FREQUENCY – MHz
Figure 21. Unity Gain , –3 dB Bandwidth
TOTAL HARMONIC DISTORTION – dBc
NORMALIZED GAIN – dB
+858C
1
0
–408C
+258C
–1
VS
–2
2kV
VOUT
VIN
50V
–4
0.1
1
10
FREQUENCY – MHz
CLOSED-LOOP GAIN – dB
TOTAL HARMONIC DISTORTION – dBc
VS = +5V
RL + CL
TO 2.5V
0
VS = 65V
–1
–2
–3
G = +1
CL = 5pF
RL = 1kV
–6
–7
1M
10M
FREQUENCY – Hz
100
–30
VCC
G = +1, RL = 2kV TO
2
–40
–50
2.5V p-p
VS = +2.7V
1.3V p-p
VS = +2.7V
–60
2V p-p
VS = +2.7V
–70
4.8V p-p
VS = +5V
–30
–40
10M
10k
100k
1M
FUNDAMENTAL FREQUENCY – Hz
G = +2
VS = +5V
VCC
RL = 1kV TO 2
–50
4.8V p-p
–60
1V p-p
–70
4.6V p-p
–80
–90
1k
100M
Figure 23. Closed-Loop Gain vs. Supply Voltage
REV. B
10
FREQUENCY – MHz
–20
VS = +2.7V
RL + CL TO 1.35V
1
–8
100k
1
Figure 25. Total Harmonic Distortion vs. Frequency; G = +1
2
–5
–225
–80
1k
100
Figure 22. Closed-Loop Gain vs. Temperature
–4
–180
–20
VS = +5V
VIN = –16dBm
2
–5
–20
Figure 24. Open-Loop Frequency Response
3
–3
PHASE
–135
0.3
100
–10
–90
OPEN-LOOP GAIN – dB
4
4V p-p
10k
100k
1M
FUNDAMENTAL FREQUENCY – Hz
10M
Figure 26. Total Harmonic Distortion vs. Frequency; G = +2
–9–
AD8031/AD8032–Typical Performance Characteristics
10
0
POWER SUPPLY REJECTION RATIO – dB
VS = 65V
OUTPUT – V p-p
8
6
VS = +5V
4
VS = +2.7V
2
0
1k
100k
FREQUENCY – Hz
10k
1M
10M
–20
VS = +5V
–40
–60
–80
–100
–120
1k
100
Figure 27. Large Signal Response
100
50
10k
100k
1M
FREQUENCY – Hz
10M
100M
Figure 30. PSRR vs. Frequency
RBT = 50V
VS = +5V
RL = 10kV TO 2.5V
VIN = 6V p-p
G = +1
5.5
10
1V / Div
ROUT – V
4.5
3.5
2.5
1.5
1
0.5
RBT
VOUT
0.1
RBT = 0
1
10
FREQUENCY – MHz
0.1
–0.5
10ms / Div
100 200
Figure 31. Output Voltage
Figure 28. ROUT vs. Frequency
VS = +5V
INPUT
–20
5.5
VS = +5V
G = +1
INPUT = 650mV
BEYOND RAILS
4.5
–40
1V / Div
COMMON-MODE REJECTION RATIO – dB
0
3.5
2.5
1.5
–60
0.5
–0.5
–80
–100
100
1k
10k
100k
FREQUENCY – Hz
1M
10ms / Div
10M
Figure 32. Output Voltage Phase Reversal Behavior
Figure 29. CMRR vs. Frequency
–10–
REV. B
AD8031/AD8032
RL TO
+2.5V
VS = +2.7V
RL = 1kV
G = –1
2.85
500mV/Div
500mV/Div
2.35
1.85
1.35
RL TO
1.35V
0.85
0.35
VS = +5V
RL = 1kV
G = –1
RL TO GND
RL TO GND
0
10ms / Div
10ms / Div
Figure 33. Output Swing
3.1
G = +2
RF = RG = 2.5kV
RL = 2kV
CL = 5pF
VS = +5V
G = +1
RF = 0
RL = 2kV TO 2.5V
CL = 5pF TO 2.5V
VS = +5V
2.56
2.54
2.7
20mV/Div
200mV/Div
2.9
Figure 35. Output Swing
2.5
2.3
2.52
2.50
2.48
2.1
2.46
1.9
2.44
50ns/Div
50ns / Div
Figure 34. 1 V Step Response
Figure 36. 100 mV Step Response
CROSSTALK – dB
–50
–60
–70
VS = 62.5V
VIN = +10dBm
–80
–90
–100
2.5kV 2.5kV
2.5kV 2.5kV
VOUT
VIN
50V
1kV
50V
TRANSMITTER
0.1
1
10
FREQUENCY – MHz
RECEIVER
100 200
Figure 37. Crosstalk vs. Frequency
REV. B
–11–
AD8031/AD8032
THEORY OF OPERATION
Switching to the NPN pair as the common-mode voltage is
driven beyond 1 V within the positive supply allows the amplifier to provide useful operation for signals at either end of the
supply voltage range and eliminates the possibility of phase
reversal for input signals up to 500 mV beyond either power
supply. Offset voltage will also change to reflect the offset of the
input pair in control. The transition region is small, on the order
of 180 mV. These sudden changes in the dc parameters of
the input stage can produce glitches that will adversely affect
distortion.
The AD8031/AD8032 are single and dual versions of high
speed, low power voltage feedback amplifiers featuring an innovative architecture that maximizes the dynamic range capability
on the inputs and outputs. Linear input common-mode range
exceeds either supply voltage by 200 mV, and the amplifiers
show no phase reversal up to 500 mV beyond supply. The output swings to within 20 mV of either supply when driving a light
load; 300 mV when driving up to 5 mA.
Fabricated on Analog Devices’ XFCB, a 4 GHz dielectrically
isolated fully complementary bipolar process, the amplifier
provides an impressive 80 MHz bandwidth when used as a
follower and 30 V/µs slew rate at only 800 µA supply current.
Careful design allows the amplifier to operate with a supply
voltage as low as 2.7 volts.
Overdriving the Input Stage
Sustained input differential voltages greater than 3.4 volts
should be avoided as the input transistors may be damaged.
Input clamp diodes are recommended if the possibility of this
condition exists.
Input Stage Operation
A simplified schematic of the input stage appears in Figure 38.
For common-mode voltages up to 1.1 volts within the positive
supply, (0 V to 3.9 V on a single 5 V supply) tail current I2
flows through the PNP differential pair, Q13 and Q17. Q5 is cut
off; no bias current is routed to the parallel NPN differential
pair Q2 and Q3. As the common-mode voltage is driven within
1.1 V of the positive supply, Q5 turns on and routes the tail
current away from the PNP pair and to the NPN pair. During
this transition region, the amplifier’s input current will change
magnitude and direction. Reusing the same tail current ensures
that the input stage has the same transconductance (which determines the amplifier’s gain and bandwidth) in both regions of
operation.
The voltages at the collectors of the input pairs are set to 200 mV
from the power supply rails. This allows the amplifier to remain
in linear operation for input voltages up to 500 mV beyond the
supply voltages. Driving the input common-mode voltage beyond that point will forward bias the collector junction of the
input transistor, resulting in phase reversal. Sustaining this
condition for any length of time should be avoided as it is easy
to exceed the maximum allowed input differential voltage when
the amplifier is in phase reversal.
VCC
R1
2kV
I2
90mA
Q9
I3
25mA
R2
2kV
1.1V
R5
50kV
Q3
VIN
R6
850V
Q5
R8
850V
VIP
Q13
Q2
R7
850V
R9
850V
1
Q6
Q10
1
Q8
Q7
4
Q17
OUTPUT STAGE,
COMMON-MODE
FEEDBACK
Q14
Q11
4
1
I1
5mA
VEE
Q18
4
I4
25mA
Q4
Q15
Q16
R3
2kV
4
1
R4
2kV
Figure 38. Simplified Schematic of AD8031 Input Stage
–12–
REV. B
AD8031/AD8032
Output Stage, Open-Loop Gain and Distortion vs. Clearance
from Power Supply
The AD8031 features a rail-to-rail output stage. The output
transistors operate as common emitter amplifiers, providing the
output drive current as well as a large portion of the amplifier’s
open-loop gain.
I1
25mA
Output overdrive of an amplifier occurs when the amplifier
attempts to drive the output voltage to a level outside its normal
range. After the overdrive condition is removed, the amplifier
must recover to normal operation in a reasonable amount of
time. As shown in Figure 40, the AD8031/AD8032 recover
within 100 ns from negative overdrive and within 80 ns from
positive overdrive.
I2
25mA
Q51
Q42
Output Overdrive Recovery
Q47
RF = RG = 2kV
DIFFERENTIAL
DRIVE
FROM
INPUT STAGE
Q37
Q38
Q68
R29
300V
Q20
RG
RF
VOUT
RL
VIN
C9
5pF
50V
Q27
Q21
Q43
VOUT
C5
1.5pF
Q48
Q49
I4
25mA
I5
25mA
Q50
Q44
1V
Figure 39. Output Stage Simplified Schematic
The output voltage limit depends on how much current the
output transistors are required to source or sink. For applications with very low drive requirements (a unity gain follower
driving another amplifier input, for instance), the AD8031 typically swings within 20 mV of either voltage supply. As the required current load increases, the saturation output voltage will
increase linearly as ILOAD × R C, where ILOAD is the required load
current and RC is the output transistor collector resistance. For
the AD8031, the collector resistances for both output transistors
are typically 25 Ω. As the current load exceeds the rated output
current of 15 mA, the amount of base drive current required to
drive the output transistor into saturation will reach its limit,
and the amplifier’s output swing will rapidly decrease.
The open-loop gain of the AD8031 decreases approximately
linearly with load resistance and also depends on the output
voltage. Open-loop gain stays constant to within 250 mV of the
positive power supply, 150 mV of the negative power supply and
then decreases as the output transistors are driven further into
saturation.
VS = 62.5V
VIN = 62.5V
RL = +1kV TO GND
100ns
Figure 40. Overdrive Recovery
Driving Capacitive Loads
Capacitive loads interact with an op amp’s output impedance to
create an extra delay in the feedback path. This reduces circuit
stability, and can cause unwanted ringing and oscillation. A
given value of capacitance causes much less ringing when the
amplifier is used with a higher noise gain.
The capacitive load drive of the AD8031/AD8032 can be increased by adding a low valued resistor in series with the capacitive load. Introducing a series resistor tends to isolate the
capacitive load from the feedback loop, thereby, diminishing its
influence. Figure 41 shows the effects of a series resistor on
capacitive drive for varying voltage gains. As the closed-loop
gain is increased, the larger phase margin allows for larger capacitive loads with less overshoot. Adding a series resistor at
lower closed-loop gains accomplishes the same effect. For large
capacitive loads, the frequency response of the amplifier will be
dominated by the roll-off of the series resistor and capacitive
load.
1000
The distortion performance of the AD8031/AD8032 amplifiers
differs from conventional amplifiers. Typically an amplifier’s
distortion performance degrades as the output voltage amplitude increases.
RS = 5V
CAPACITIVE LOAD – pF
VS = +5V
200mV STEP
WITH 30% OVERSHOOT
Used as a unity gain follower, the AD8031/AD8032 output will
exhibit more distortion in the peak output voltage region around
VCC –0.7 V. This unusual distortion characteristic is caused by
the input stage architecture and is discussed in detail in the
section covering “Input Stage Operation.”
RS = 0V
100
RS = 20V
RS = 20V
10
RG
RF
RS = 0V, 5V
RS
VOUT
CL
1
0
1
2
3
CLOSED-LOOP GAIN – V/V
4
5
Figure 41. Capacitive Load Drive vs. Closed-Loop Gain
REV. B
–13–
AD8031/AD8032
0
APPLICATIONS
A 2 MHz Single Supply Biquad Bandpass Filter
Figure 42 shows a circuit for a single supply biquad bandpass
filter with a center frequency of 2 MHz. A 2.5 V bias level is
easily created by connecting the noninverting inputs of all three
op amps to a resistor divider consisting of two 1 kΩ resistors
connected between +5 V and ground. This bias point is also
decoupled to ground with a 0.1 µF capacitor. The frequency
response of the filter is shown in Figure 43.
GAIN – dB
–10
In order to maintain an accurate center frequency, it is essential
that the op amp has sufficient loop gain at 2 MHz. This requires
the choice of an op amp with a significantly higher unity gain
crossover frequency. The unity gain crossover frequency of the
AD8031/AD8032 is 40 MHz. Multiplying the open-loop gain by
the feedback factors of the individual op amp circuits yields the
loop gain for each gain stage. From the feedback networks of the
individual op amp circuits, we can see that each op amp has a
loop gain of at least 21 dB. This level is high enough to ensure
that the center frequency of the filter is not affected by the op
amp’s bandwidth. If, for example, an op amp with a gain bandwidth product of 10 MHz was chosen in this application, the
resulting center frequency would shift by 20% to 1.6 MHz.
R6
1kV
–20
–30
–40
–50
10k
100k
1M
FREQUENCY – Hz
10M
100M
Figure 43. Frequency Response of 2 MHz Bandpass Filter
High Performance Single Supply Line Driver
Even though the AD8031/AD8032 swing close to both rails,
the AD8031 has optimum distortion performance when the
signal has a common-mode level half way between the supplies
and when there is about 500 mV of headroom to each rail. If
low distortion is required in single supply applications for signals that swing close to ground, an emitter follower circuit can
be used at the op amp output.
C1
50pF
+5V
R2
2kV
R4
2kV
+5V
0.1mF
VIN
10mF
+5V
0.1mF
R1
3kV
R3
2kV
1kV
0.1mF
C2
50pF
3
VIN
R5
2kV
6
49.9V
AD8031
1/2
AD8032
0.1mF
7
2
2N3904
4 AD8031
1/2
AD8032
2.49kV
1kV
2.49kV
49.9V
200V
VOUT
49.9V
VOUT
Figure 42. A 2 MHz Biquad Bandpass Filter Using AD8031/
AD8032
–14–
Figure 44. Low Distortion Line Driver for Single Supply
Ground Referenced Signals
REV. B
AD8031/AD8032
Figure 44 shows the AD8031 configured as a single supply gainof-2 line driver. With the output driving a back terminated 50 Ω
line, the overall gain from VIN to VOUT is unity. In addition to
minimizing reflections, the 50 Ω back termination resistor protects the transistor from damage if the cable is short circuited.
The emitter follower, which is inside the feedback loop, ensures
that the output voltage from the AD8031 stays about 700 mV
above ground. Using this circuit, very low distortion is attainable even when the output signal swings to within 50 mV of
ground. The circuit was tested at 500 kHz and 2 MHz. Figures
45 and 46 show the output signal swing and frequency spectrum
at 500 kHz. At this frequency, the output signal (at VOUT),
which has a peak-to-peak swing of 1.95 V (50 mV to 2 V), has a
THD of –68 dB (SFDR = –77 dB).
Figures 47 and 48 show the output signal swing and frequency
spectrum at 2 MHz. As expected, there is some degradation in
signal quality at the higher frequency. When the output signal
has a peak-to-peak swing of 1.45 V (swinging from 50 mV to
1.5 V), the THD is –55 dB (SFDR = –60 dB).
This circuit could also be used to drive the analog input of a
single supply high speed ADC whose input voltage range is
referenced to ground (e.g., 0 V to 2 V or 0 V to 4 V). In this
case, a back termination resistor is not necessary (assuming a
short physical distance from transistor to ADC), so the emitter of the external transistor would be connected directly to the
ADC input. The available output voltage swing of the circuit
would, therefore be doubled.
1.5V
100
100
90
90
10
0%
2V
10
0%
0.2V
50mV
0.5V
Figure 45. Output Signal Swing of Low Distortion Line
Driver at 500 kHz
Figure 47. Output Signal Swing of Low Distortion Line
Driver at 2 MHz
+7dBm
VERTICAL SCALE – 10dB/Div
VERTICAL SCALE – 10dB/Div
+9dBm
START 0Hz
START 0Hz
STOP 5MHz
STOP 20MHz
Figure 48. THD of Low Distortion Line Driver at 2 MHz
Figure 46. THD of Low Distortion Line Driver at 500 kHz
REV. B
200ns
50mV
1ms
–15–
AD8031/AD8032
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
0.1968 (5.00)
0.1890 (4.80)
0.39 (9.91)
MAX
8
5
0.25
(6.35)
0.31
(7.87)
1
0.1574 (4.00)
0.1497 (3.80)
4
0.035 ±0.01
(0.89 ±0.25)
PIN 1
0.165 ±0.01
(4.19 ±0.25)
0.30 (7.62)
REF
0.018 ±0.003 0.10 0.033
(0.46 ±0.08) (2.54) (0.84)
BSC NOM
SEATING
PLANE
15°
0°
0.011 ±0.003
(0.28 ±0.08)
SEATING
PLANE
4
0.2440 (6.20)
0.2284 (5.80)
0.0688 (1.75)
0.0532 (1.35)
0.0500 0.0192 (0.49)
(1.27) 0.0138 (0.35)
BSC
0.0196 (0.50)
x 45°
0.0099 (0.25)
0.0098 (0.25)
0.0075 (0.19)
8°
0°
0.0500 (1.27)
0.0160 (0.41)
5-Lead Plastic Surface Mount (SOT-23)
(RT-5)
0.122 (3.10)
0.114 (2.90)
0.1181 (3.00)
0.1102 (2.80)
5
0.0669 (1.70)
0.0590 (1.50)
0.199 (5.05)
0.187 (4.75)
1
5
1
PIN 1
8-Lead ␮SOIC
(RM-8)
8
8
0.0098 (0.25)
0.0040 (0.10)
0.18 ±0.03
(4.57 ±0.76)
0.125 (3.18)
MIN
0.122 (3.10)
0.114 (2.90)
C2152b–0–9/99
8-Lead Plastic SOIC
(SO-8)
8-Lead Plastic DIP
(N-8)
5
1
4
2
0.1181 (3.00)
0.1024 (2.60)
3
4
PIN 1
PIN 1
0.0374 (0.95) BSC
0.0256 (0.65) BSC
0.120 (3.05)
0.112 (2.84)
0.120 (3.05)
0.112 (2.84)
0.0512 (1.30)
0.0354 (0.90)
0.043 (1.09)
0.037 (0.94)
0.018 (0.46)
0.008 (0.20)
0.011 (0.28)
0.003 (0.08)
33°
27°
0.0059 (0.15)
0.0019 (0.05)
0.028 (0.71)
0.016 (0.41)
0.0079 (0.20)
0.0031 (0.08)
0.0571 (1.45)
0.0374 (0.95)
0.0197 (0.50)
0.0138 (0.35)
SEATING
PLANE
108
08
0.0217 (0.55)
0.0138 (0.35)
PRINTED IN U.S.A.
0.006 (0.15)
0.002 (0.05)
SEATING
PLANE
0.0748 (1.90)
BSC
–16–
REV. B
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