SUPERTEX HV9923

HV9921/HV9922/HV9923
Initial Release
3-Pin Switch-Mode LED Lamp Driver ICs
Features
General Description
Constant Output Current:
o HV9921 – 20mA
o HV9922 – 50mA
o HV9923 – 30mA
Universal 85-264VAC Operation
Fixed OFF-Time Buck Converter
Internal 500V Power MOSFET
The HV9921/22/23 are pulse width modulated (PWM)
high-efficiency LED driver control ICs. They allow efficient
operation of LED strings from voltage sources ranging up
to 400VDC. The HV9921/22/23 include an internal high
voltage switching MOSFET controlled with fixed off-time TOFF
of approximately 10µs. The LED string is driven at constant
current, thus providing constant light output and enhanced
reliability. The output current is internally fixed at 20mA for
HV9921, 50mA for HV9922, and 30mA for the HV9923. The
peak current control scheme provides good regulation of the
output current throughout the universal AC line voltage range
of 85 to 264VAC or DC input voltage of 20 to 400V.
Applications
Decorative Lighting
Low Power Lighting Fixtures
Typical Application Circuit
HV9921/22/23
NR092005
1
HV9921/HV9922/HV9923
Absolute Maximum Ratings
Ordering Information
Supply Voltage, VDD
-0.3 to +10V
Package Options
DEVICE
+5mA
Supply Current, IDD
TO-92
SOT-89
Operating Ambient Temperature Range
-40°C to +85°C
HV9921
HV9921N3-G
HV9921N8-G
Operating Junction Temperature Range
-40° to +125°C
HV9922
HV9922N3-G
HV9922N8-G
Storage Temperature Range
-65° to +150°C
HV9923
HV9923N3-G
HV9923N8-G
Power Dissipation @ 25°C, TO-92
Power Dissipation @ 25°C, SOT-89
-G indicates package is RoHS compliant (‘Green’)
740mW
1600mW†
† Mounted on FR4 board, 25mm x 25mm x 1.57mm. Stresses beyond those listed under
“Absolute Maximum Ratings” may cause permanent damage to the device. These are
stress ratings only, and functional operation of the device at these or any other conditions
beyond those indicated in the operational sections of the specifications is not implied.
Exposure to absolute maximum rating conditions for extended periods may affect device
reliability.
Electrical Characteristics
(The * denotes the specifications which apply over the full operating junction temperature range of
-40°C < TA < +85°C, otherwise the specifications are at TA=25°C, VDRAIN=50V, unless otherwise noted)
Regulator (VDD)
Symbol
VDD
Parameter
VDD Regulator Output
Min
7.1
Typ
7.5
Max
7.8
Units
V
VDRAIN
VDRAIN Supply Voltage
20
V
VUVLO
VDD Undervoltage Threshold
5.0
V
Conditions
∆VUVLO
VDD Undervoltage Lockout
Hysteresis
200
IDD
Operating Supply Current
200
350
µA
Typ
Max
Units
V
210
Ω
IDRAIN = 20mA (HV9921)
IDRAIN = 50mA (HV9922)
IDRAIN = 30mA (HV9923)
5
pF
VDRAIN = 400V
mV
VDD(EXT) = 8.5V, VDRAIN = 40V
Output (DRAIN)
Symbol
VBR
RON
CDRAIN
ISAT
Parameter
Breakdown Voltage
Min
500
ON Resistance – HV9921,22, 23
Output Capacitance
MOSFET Saturation Current
1
100
150
Conditions
mA
Current Sense Comparator
Symbol
ITH
ITH
ITH
Parameter
Threshold Current – HV9921
Threshold Current – HV9922
Threshold Current – HV9923
Min
20.5
52
30.8
Typ
Max
25.5
63
38.2
Units
mA *
mA *
mA *
TBLANK
Leading Edge Blanking Delay
200
300
400
ns
TON(MIN)
Minimum ON Time
650
ns
Max
13
Units
µs
Conditions
*
OFF-Time Generator
Symbol
TOFF
Parameter
OFF Time
Min
8
Typ
10.5
2
Conditions
NR092005
HV9921/HV9922/HV9923
Functional Block Diagram
GND
DRAIN
VDD
TOFF = 10µs
Regulator
7.0V
REF
S
Q
R
Q
+
R
TBLANK = 300ns
HV9921/22/23
Pin Configuration
1-Drain
2-GND
3-VDD
1
2
3
TO-243AA (SOT-89)
123
TO-92
Drain – This is a drain terminal of the output switching
MOSFET and a linear regulator input.
GND – This is a common connection for all circuits.
VDD –
This is a power supply pin for all control circuits.
Bypass this pin with a 0.1uF low impedance
capacitor.
3
NR092005
HV9921/HV9922/HV9923
Typical Performance Characteristics (T
J
= 25°C unless otherwise noted)
200
1.1
Normalized Threshold Current
180
ON Resistance, Ohm
1.05
1
0.95
0.9
0.85
160
140
120
100
80
60
40
20
0.8
-40
-15
10
35
60
85
0
-40
110
Junction Temperature, °C
-15
10
35
60
85
110
Junction Tem perature, °C
12
1000
DRAIN Capacitance (pF)
OFF Time, uS
10
8
6
4
2
0
-40
100
10
1
-15
10
35
60
85
110
0
10
20
30
40
30
40
DRAIN Voltage (V)
580
180
570
160
560
140
DRAIN Current, mA
DRAIN Breakdown Voltage, V
Junction Tem perature, °C
550
540
530
520
510
TJ = 25°C
TJ = 125°C
120
100
80
60
40
500
20
490
-40
-15
10
35
60
85
0
110
0
Junction Tem perature, °C
10
20
DRAIN Voltage, V
4
NR092005
HV9921/HV9922/HV9923
Functional Description
where ITH is the current sense comparator threshold.
The ripple current introduces a peak-to-average error in
the output current setting that needs to be accounted for.
Due to the constant off-time control technique used in the
HV9921/22/23, the ripple current is independent of the input
AC or DC line voltage variation. Therefore, the output current
will remain unaffected by the varying input voltage.
The HV9921/22/23 are PWM peak current controllers
for controlling a buck converter topology in continuous
conduction mode (CCM). The output current is internally
preset at 20mA (HV9921), 50mA (HV9922), or 30mA
(HV9923).
When the input voltage of 20 to 400V appears at the
DRAIN pin, the internal high-voltage linear regulator seeks
to maintain a voltage of 7VDC at the VDD pin. Until this
voltage exceeds the internally programmed under-voltage
threshold, the output switching MOSFET is non-conductive.
When the threshold is exceeded, the MOSFET turns on. The
input current begins to flow into the DRAIN pin. Hysteresis
is provided in the under-voltage comparator to prevent
oscillation.
Adding a filter capacitor across the LED string can reduce
the output current ripple even further, thus permitting a
reduced value of L1. However, one must keep in mind that
the peak-to-average current error is affected by the variation
of TOFF. Therefore, the initial output current accuracy might
be sacrificed at large ripple current in L1.
Another important aspect of designing an LED driver with the
HV9921/22/23 is related to certain parasitic elements of the
circuit, including distributed coil capacitance of L1, junction
capacitance and reverse recovery of the rectifier diode D1,
capacitance of the printed circuit board traces CPCB and output
capacitance CDRAIN of the controller itself. These parasitic
elements affect the efficiency of the switching converter and
could potentially cause false triggering of the current sense
comparator if not properly managed. Minimizing these
parasitics is essential for efficient and reliable operation of
the HV9921/22/23.
When the input current exceeds the internal preset level,
a current sense comparator resets an RS flip-flop, and the
MOSFET turns off. At the same time, a one-shot circuit
is activated that determines the duration of the off-state
(10.5µS typ.). As soon as this time is over, the flip-flop sets
again. The new switching cycle begins.
A “blanking” delay of 300nS is provided that prevents false
triggering of the current sense comparator due to the leading
edge spike caused by circuit parasitics.
Coil capacitance of inductors is typically provided in the
manufacturer’s data books either directly or in terms of the
self-resonant frequency (SRF).
Application Information
SRF = 1 /( 2π L ⋅ CL )
The HV9921/22/23 is a low-cost off-line buck converter IC
specifically designed for driving multi-LED strings. It can be
operated from either universal AC line range of 85 to 264VAC,
or 20 to 400VDC, and drives up to tens of high brightness
LEDs. All LEDs can be run in series, and the HV9921/22/23
regulates at constant current, yielding uniform illumination.
The HV9921/22/23 is compatible with triac dimmers. The
output current is internally fixed at 20mA for the HV9921,
50mA for the HV9922, and 30mA for HV9923. These parts
are available in space saving TO-92 and SOT-89 packages.
where L is the inductance value, and CL is the coil
capacitance.) Charging and discharging this capacitance
every switching cycle causes high-current spikes in the LED
string. Therefore, connecting a small capacitor CO (~10nF) is
recommended to bypass these spikes.
Using an ultra-fast rectifier diode for D1 is recommended
to achieve high efficiency and reduce the risk of false
triggering of the current sense comparator. Using diodes
with shorter reverse recovery time trr and lower junction
capacitance CJ achieves better performance. The reverse
voltage rating VR of the diode must be greater than the
maximum input voltage of the LED lamp.
Selecting L1 and D1
There is a certain trade-off to be considered between
optimal sizing of the output inductor L1 and the tolerated
output current ripple. The required value of L1 is inversely
proportional to the ripple current ∆IO in it.
V ⋅T
L1 = O OFF
∆ IO
The total parasitic capacitance present at the DRAIN pin of
the HV9921/22/23 can be calculated as:
(1)
CP = CDRAIN + CPCB + CL + C J
When the switching MOSFET turns on, the capacitance CP
is discharged into the DRAIN pin of the IC. The discharge
current is limited to about 150mA typically. However, it
may become lower at increased junction temperature. The
duration of the leading edge current spike can be estimated
VO is the forward voltage of the LED string. TOFF is the offtime of the HV9921/22/23. The output current in the LED
string (IO) is calculated then as:
I O = ITH −
1
⋅ ∆ IO
2
(3)
(2)
5
NR092005
HV9921/HV9922/HV9923
as:
TSPIKE
V ⋅C
= IN P + trr
I SAT
IDD is the internal linear regulator current.
When the LED driver is powered from the full-wave
rectified AC line input, the exact equation for calculating the
conduction loss is more cumbersome. However, it can be
estimated using the following equation:
(4)
PCOND = K C ⋅ I O 2 ⋅ RON + K d ⋅ I DD ⋅ VAC
In order to avoid false triggering of the current sense
comparator, CP must be minimized in accordance with the
following expression:
CP <
I SAT ⋅ (TBLANK ( MIN ) − trr )
(10)
where VAC is the input AC line voltage. The coefficients KC
and Kd can be determined from the minimum duty ratio of
the HV9921/22/23
(5)
VIN ( MAX )
0.7
where TBLANK(MIN) is the minimum blanking time of 200ns, and
VIN(MAX) is the maximum instantaneous input voltage.
0.6
Estimating Power Loss
0.5
Discharging the parasitic capacitance CP into the DRAIN
pin of the HV9921/22/23 is responsible for the bulk of the
switching power loss. It can be estimated using the following
equation:
⎛ V 2C
⎞
PSWITCH = ⎜ IN P + VIN I SAT ⋅ trr ⎟ ⋅ FS
⎝ 2
⎠
Kd (Dm)
Kc (Dm)
0.3
(6)
0.2
where Fs is the switching frequency, ISAT is the saturated
DRAIN current of the HV9921/22/23. The switching loss is
the greatest at the maximum input voltage.
0.1
−1
VIN − η ⋅ VO
VIN ⋅ TOFF
(7)
(
)
0.4
0.5
0.6
0.7
Design Example 1
Let us design an HV9921 LED lamp driver meeting the
following specifications:
The switching power loss associated with turn-off transitions
of the DRAIN pin can be disregarded. Due to the large amount
of parasitic capacitance connected to this switching node,
the turn-off transition occurs essentially at zero-voltage.
Input:
Universal AC, 85-264VAC
Output Current: 20mA
Load:
String of 10 LED (LW541C by OSRAM
VF = 4.1V max. each)
Conduction power loss in the HV9921/22/23 can be
calculated as:
PCOND = D ⋅ I O ⋅ RON + I DD ⋅ VIN ⋅ (1 − D )
0.3
As with all off-line converters, selecting an input filter is critical
to obtaining good EMI. A switching side capacitor, albeit of
small value, is necessary in order to ensure low impedance
to the high frequency switching currents of the converter. As
a rule of thumb, this capacitor should be approximately 0.10.2 µF/W of LED output power. A recommended input filter is
shown in Figure 2 for the following design example.
VAC is the input AC line voltage.
2
0.2
EMI Filter
When the HV9921/22/23 LED driver is powered from the
full-wave rectified AC input, the switching power loss can be
estimated as:
(8)
1
(VAC ⋅ CP + 2 ⋅ I SAT ⋅ trr ) VAC − η −1 ⋅ VO
2 ⋅ TOFF
0.1
Fig. 1. Conduction Loss Coefficients KC and Kd
where η is the efficiency of the power converter.
PSWITCH ≈
0
Dm
The switching frequency is given by the following:
FS =
0.4
Step 1. Calculating L1.
(9)
The output voltage VO = 10 ⋅ VF ≈ 41V (max.). Use equation
(1) assuming a 30% peak-to-peak ripple.
L1 =
where D = VO /ηVIN is the duty ratio, RON is the ON resistance,
6
41V ⋅ 10.5 µ s
= 72mH
0.3 ⋅ 20mA
NR092005
HV9921/HV9922/HV9923
Let us assume that the overall efficiency η = 0.7.
Switching power loss:
1
41V ⎞
PSWITCH ≈
(264V ⋅ 31pF + 2 ⋅ 100mA ⋅ 20ns )⎛⎜ 264V −
⎟
2 ⋅ 10.5 µ s
07
. ⎠
⎝
Select L1 68mH, I=30mA. Typical SRF=170KHz. Calculate
the coil capacitance.
CL =
1
1
=
≈ 13 pF
2
L1 ⋅ ( 2π ⋅ SRF )
68mH ⋅ ( 2π ⋅ 170KHz )2
PSWITCH ≈ 120mW
Minimum duty ratio:
Step 2. Selecting D1
Dm = 41V /( 07
. ⋅ 264V ⋅ 2 ) ≈ 0.16
Usually, the reverse recovery characteristics of ultrafast rectifiers at IF=20~50mA are not provided in the
manufacturer’s data books. The designer may want to
experiment with different diodes to achieve the best result.
Conduction power loss:
PCOND = 0.25 ⋅ (20mA ) ⋅ 210Ω + 0.63 ⋅ 200 µ A ⋅ 264V ≈ 55mW
2
Select D1 MUR160 with VR = 600V, trr ≈ 20ns (IF=20mA,
IRR=100mA) and CJ ≈ 8pF (VF>50V).
Step 3. Calculating total parasitic capacitance using (3)
Total power dissipation in HV9921:
CP = 5 pF + 5 pF + 13 pF + 8 pF = 31pF
PTOTAL = 120mW + 55mW = 175mW
Step 6. Selecting input capacitor CIN
Step 4. Calculating the leading edge spike duration using
(4), (5)
TSPIKE =
Output Power = 41V ⋅ 20mA = 820mW
264V ⋅ 2 ⋅ 31pF
+ 20ns ≈ 136ns < TBLANK ( MIN )
Select CIN ECQ-E4104KF by Panasonic (0.1µF, 400V,
100mA
Metalized Polyester Film).
Step 5. Estimating power dissipation in HV9921 at 264VAC
using (8) and (10)
Figure 2. Universal 85-264VAC LED Lamp Driver
D2
LIN
D3
CIN
CIN2
D4
LED1 -LED12
CO
D5
D1
U1
AC Line
85-264V
L1
VRD1
HV9921/22/23
F1
3
CDD
VDD
DRAIN 1
GND
2
7
NR092005
HV9921/HV9922/HV9923
Efficiency (%)
Figure 3. Typical Efficiency
Figure 4. Switch-Off Transition. Ch1: VDRAIN, Ch3: IDRAIN
82.00
80.00
78.00
76.00
74.00
72.00
70.00
68.00
66.00
64.00
62.00
ZERO VOLTAGE
TRANSITION
75
100 125 150 175 200 225 250 275
Input AC Line Voltage (VAC)
Figure 5. Typical Efficiency
Figure 6. Switch-Off Transition. Ch1: VDRAIN, Ch3: IDRAIN
LEADING EDGE SPIKE
SWITCH OFF
25mA
8
NR092005
HV9921/HV9922/HV9923
Thermal Considerations vs. Radiated EMI
The copper area where GND pin is connected acts not only
as a single point ground, but also as a heat sink. This area
should be maximized for good heat sinking, especially when
HV9921N8, HV9922N8, and HV9923N8 (SOT-89 package)
are used. The same applies to the cathode of the freewheeling diode D1. Both nodes are quiet and therefore, will
not cause radiated RF emission. The switching node copper
area connected to the DRAIN pin of the HV9921/22/23, the
anode of D1 and the inductor L1 needs to be minimized.
A large switching node area can increase high frequency
radiated EMI.
HV9921/22/23 Layout Considerations
See Figure 7 for a recommended circuit board layout for the
HV9921/22/23.
Single Point Grounding
Use a single point ground connection from the input filter
capacitor to the area of copper connected to the GND pin.
Bypass Capacitor (CDD)
The VDD pin bypass capacitor CDD should be located as near
as possible to the VDD and GND pins.
Input Filter Layout Considerations
Switching Loop Areas
The area of the switching loop connecting the input filter
capacitor CIN, the diode D1 and the HV9921/22/23 together
should be kept as small as possible.
The input circuits of the EMI filter must not be placed in the
direct proximity to the inductor L1 in order to avoid magnetic
coupling of its leakage fields. This consideration is especially
important when unshielded construction of L1 is used. When
an axial input EMI filter inductor LIN is selected, it must be
positioned orthogonal with respect to L1. The loop area
formed by CIN2, LIN and CIN should be minimized. The input
lead wires must be twisted together.
The switching loop area connecting the output filter capacitor
CO, the inductor L1 and the diode D1 together should be kept
as small as possible.
Figure 7. Recommended circuit board layout with the HV9921N3/HV9922N3/HV9923N3
COMPONENT SIDE VIEW
VRD1
CO
LIN
LED +
D1
F1
LED -
CIN
CIN2
AC Line
85-264VAC
L1
D2-5
CDD
9
U1
NR092005
HV9921/HV9922/HV9923
3 Lead TO-92 Plastic Package (N3)
0.190 ± 0.020
(4.826 ± 0.508)
D
0.190 ± 0.020
(4.826 ± 0.508)
A
5° NOM
Seating Plane
0.500
MIN
(12.700)
0.018
Typ
(0.457)
0.050
MAX
(1.270)
Uncontrolled
Lead Dia.
L
123
b
C
0.050 ± 0.005
(1.270 ± 0.127)
e1
0.100 ± 0.005
(2.540 ± 0.127)
e
0.015 ± 0.005
Typ
(0.381 ± 0.127)
Before Lead Finish
H
0.050 ± 0.005
(1.270 ± 0.127)
E
0.140 ± 0.005
(3.556 ± 0.127)
10° NOM
1
2 3
0.090
R NOM
(2.286)
3-Lead TO-243 (SOT-89) Surface Mount Package (N8)
0.177 ±0.004
(4.496 ± 0.102)
0.059 ±0.004
(1.499 ± 0.102)
D
0.068 ±0.004
(1.727 ± 0.102)
A
b3
0.039 ±0.002
(0.990 ± 0.060)
C
0.016 ±0.002
(0.394 ± 0.038)
K
0.161 ±0.006
(4.089 ± 0.152)
H
M
E
0.048 ±0.001
(1.225 ± 0.025)
0.096 ±0.006
(2.438 ± 0.152)
L
0.087 ±0.003
(2.210 ± 0.076)
b1
b2
0.041 ±0.006
(1.041 ± 0152)
e1
0.017 ±0.003
(0.419 ± 0.064)
0.020 ±0.003
(0.508 ± 0.076)
DSFP# HV9921/22/23
NR092005
e
Measurement =
0.118
BSC
(2.997)
Inches
(millimeters)
Note: Circle (eg. ©) indicates JEDEC reference.
0.059
BSC
(1.499)
10