May 8, 2008 IRS233(0,2)(D)(S & J)PbF 3-PHASE-BRIDGE DRIVER Features • • • • • • • • • • • • • Floating channel designed for bootstrap operation Fully operational to +600 V Tolerant to negative transient voltage – dV/dt immune Gate drive supply range from 10 V to 20 V Undervoltage lockout for all channels Over-current shutdown turns off all six drivers Independent half-bridge drivers Matched propagation delay for all channels 3.3 V logic compatible Outputs out of phase with inputs Cross-conduction prevention logic Integrated Operational Amplifier Integrated Bootstrap Diode function (IRS233(0,2)D) RoHS Compliant Description The IRS233(0,2)(D)(S & J) is a high voltage, high speed power MOSFET and IGBT driver with three independent high and low side referenced output channels. Proprietary HVIC technology enables ruggedized monolithic construction. Logic inputs are compatible with CMOS or LSTTL outputs, down to 3.3 V logic. A ground-referenced operational amplifier provides analog feedback of bridge current via an external current sense resistor. A current trip function which terminates all six outputs is also derived from this resistor. An open drain FAULT signal indicates if an over-current or undervoltage shutdown has occurred. The output drivers feature a high pulse current buffer stage designed for minimum driver cross-conduction. Propagation delays are matched to simplify use at high frequencies. The floating channel can be used to drive N-channel power MOSFET or IGBT in the high side configuration which operates up to 600 volts. Product Summary VOFFSET 600V max. IO+/- 200 mA / 420 mA VOUT 10 V – 20 V (233(0,2)(D)) ton/off (typ.) 500 ns Deadtime (typ.) 2.0 us (IRS2330(D)) 0.7 us (IRS2332(D)) Applications: *Motor Control *Air Conditioners/ Washing Machines *General Purpose Inverters *Micro/Mini Inverter Drives Packages 28-Lead SOIC 44-Lead PLCC w/o 12 Leads Typical Connection Absolute Maximum Ratings www.irf.com 1 IRS233(0,2)(D)(S&J)PbF † Qualification Information Qualification Level Industrial†† Comments: This family of ICs has passed JEDEC’s Industrial qualification. IR’s Consumer qualification level is granted by extension of the higher Industrial level. SOIC28W MSL3†††, 260°C (per IPC/JEDEC J-STD-020) PLCC44 MSL3†††, 245°C (per IPC/JEDEC J-STD-020) Moisture Sensitivity Level Human Body Model ESD Machine Model IC Latch-Up Test RoHS Compliant † †† ††† Class 2 (per JEDEC standard JESD22-A114) Class B (per EIA/JEDEC standard EIA/JESD22-A115) Class I, Level A (per JESD78) Yes Qualification standards can be found at International Rectifier’s web site http://www.irf.com/ Higher qualification ratings may be available should the user have such requirements. Please contact your International Rectifier sales representative for further information. Higher MSL ratings may be available for the specific package types listed here. Please contact your International Rectifier sales representative for further information. www.irf.com 2 IRS233(0,2)(D)(S&J)PbF Absolute Maximum Ratings Absolute Maximum Ratings indicate sustained limits beyond which damage to the device may occur. All voltage parameters are absolute voltages referenced to VSO. The thermal resistance and power dissipation ratings are measured under board mounted and still air conditions. Symbol Definition Min. Max. VB1,2,3 High Side Floating Supply Voltage -0.3 620 VS1,2,3 High Side Floating Offset Voltage VB1,2,3 - 20 VB1,2,3 + 0.3 VHO1,2,3 High Side Floating Output Voltage VS1,2,3 - 0.3 VB1,2,3 + 0.3 -0.3 20 VCC VSS VLO1,2,3 Low Side and Logic Fixed Supply Voltage Logic Ground Low Side Output Voltage VCC - 20 VCC + 0.3 -0.3 VCC + 0.3 Units V _______ ______ Logic Input Voltage ( HIN1,2,3, LIN1,2,3 & ITRIP) VSS -0.3 VFLT VCAO FAULT Output Voltage Operational Amplifier Output Voltage VSS -0.3 VSS -0.3 (VSS + 15) or (VCC + 0.3) Whichever is lower VCC +0.3 VCC +0.3 VCA- Operational Amplifier Inverting Input Voltage VSS -0.3 VCC +0.3 — 50 V/ns — — 1.6 2.0 W 78 63 150 VIN dVS/dt PD Allowable Offset Supply Voltage Transient Package Power Dissipation @ TA ≤ +25 °C (28 lead SOIC) (44 lead PLCC) (28 lead SOIC) (44 lead PLCC) TJ Junction Temperature — — — TS Storage Temperature -55 150 TL Lead Temperature (soldering, 10 seconds) — 300 RthJA Thermal Resistance, Junction to Ambient www.irf.com °C/W °C 3 IRS233(0,2)(D)(S&J)PbF Recommended Operating Conditions The Input/Output logic timing diagram is shown in figure 1. For proper operation the device should be used within the recommended conditions. All voltage parameters are absolute voltage referenced to VSO. The VS offset rating is tested with all supplies biased at 15 V differential. Symbol Definition Min. Max. VS1,2,3 +10 VS1,2,3 +20 VSO-8 (Note1) 600 -50 (Note2) VS1,2,3 600 VB1,2,3 VB1,2,3 High Side Floating Supply Voltage VS1,2,3 Static High side floating offset voltage VSt1,2,3 VHO1,2,3 Transient High side floating offset voltage VCC Low Side and Logic Fixed Supply Voltage 10 20 VSS Logic Ground -5 5 0 VSS VSS VCC VSS + 5 VCC VLO1,2,3 VIN VFLT High Side Floating Output Voltage Low Side Output Voltage Logic Input Voltage (HIN1,2,3, LIN1,2,3 & ITRIP) FAULT Output Voltage VCAO Operational Amplifier Output Voltage VSS VSS + 5 VCA- Operational Amplifier Inverting Input Voltage VSS VSS + 5 Ambient temperature -40 125 TA Units V °C Note 1: Logic operational for VS of (VSO -8 V) to (VSO +600 V). Logic state held for VS of (VSO -8 V) to (VSO – VBS). Note 2: Operational for transient negative VS of VSS - 50 V with a 50 ns pulse width. Guaranteed by design. Refer to the Application Information section of this datasheet for more details. Note 3: CAO input pin is internally clamped with a 5.2 V zener diode. Dynamic Electrical Characteristics VBIAS (VCC, VBS1,2,3) = 15 V, VSO1,2,3 = VSS , CL = 1000 pF, TA = 25 °C unless otherwise specified. Symbol Definition Min Typ Max Units Test Conditions ton Turn-on propagation delay 400 500 700 toff Turn-off propagation delay 400 500 700 tr Turn-on rise time — 80 125 tf Turn-off fall time — 35 55 titrip ITRIP to Output Shutdown Propagation Delay 400 660 920 tbl tflt ITRIP Blanking Time ITRIP to FAULT Indication Delay Input Filter Time (All Six Inputs) LIN1,2,3 to FAULT Clear Time (2330/2) — 350 — 400 550 325 — 870 — tflt, in tfltclr DT MDT Deadtime: (IRS2330(D)) (IRS2332(D)) Deadtime matching: : (IRS2330(D)) (IRS2332(D)) 5300 8500 13700 1300 2000 3100 500 700 1100 — — 400 — — 140 MT Delay matching time (t ON , t OFF) — — 50 PM Pulse width distortion — — 75 VS1,2,3 = 0 V to 600 V VS1,2,3 = 0 V ns VIN = 0 V & 5 V without external deadtime VIN = 0 V & 5 V without external deadtime larger than DT PM input 10 µs NOTE: For high side PWM, HIN pulse width must be > 1.5 usec www.irf.com 4 IRS233(0,2)(D)(S&J)PbF Dynamic Electrical Characteristics VBIAS (VCC, VBS1,2,3) = 15 V, VSO1,2,3 = VSS , CL = 1000 pF, TA = 25 °C unless otherwise specified. Symbol SR+ SR- www.irf.com Definition Operational Amplifier Slew Rate (+) Operational Amplifier Slew Rate (-) Min Typ Max Units Test Conditions 5 2.4 10 3.2 — — V/µs 1 V input step 5 IRS233(0,2)(D)(S&J)PbF Static Electrical Characteristics VBIAS (VCC, VBS1,2,3) = 15 V, VSO1,2,3 = VSS and TA = 25 °C unless otherwise specified. The VIN, VTH and IIN parameters are referenced to VSS and are applicable to all six logic input leads: HIN1,2,3 & LIN1,2,3. The VO and IO parameters are referenced to VSO1,2,3 and are applicable to the respective output leads: HO1,2,3 or LO1,2,3. Symbol Definition VIH Logic “0” input Voltage (OUT = LO) VIL Min Typ Max Units Test Conditions — — 2.2 VIT,TH+ Logic “1” input Voltage (OUT = HI) ITRIP Input Positive Going Threshold 0.8 400 — 490 — 580 VOH High Level Output Voltage, VBIAS - VO — — 1000 VOL Low Level Output Voltage, VO — — 400 ILK Offset Supply Leakage Current — — 50 IQBS Quiescent VBS Supply Current — 30 50 IQCC Quiescent VCC Supply Current — 4 6.2 IIN+ IIN- Logic “1” Input Bias Current (OUT =HI) Logic “0” Input Bias Current (OUT = LO) “High” ITRIP Bias Current “LOW” ITRIP Bias Current VBS Supply Undervoltage Positive Going Threshold VBS Supply Undervoltage Negative Going Threshold VCC Supply Undervoltage Positive going Threshold VCC Supply Undervoltage Negative Going Threshold IITRIP+ IITRIPVBSUV+ VBSUVVCCUV+ VCCUV- -400 -300 -100 -300 -220 -100 — 5 10 — — 30 7.5 8.35 9.2 7.1 7.95 8.8 8.3 9 9.7 8 8.7 9.4 VCCUVH Hysteresis — 0.3 — VBSUVH Hysteresis FAULT Low On-Resistance — 0.4 — — 55 75 IO+ Output High Short Circuit Pulsed Current — -250 -180 IO- Output Low Short Circuit Pulsed Current 420 500 — — — — 200 — — — 20 100 — 80 — Ron, FLT RBS VOS ICACMRR PSRR VOH,AMP VOL,AMP V mV VIN = 5 V, IO = 20 mA µA mA µA nA VB = VS = 600 V VIN = 0 V or 4 V VIN = 0 V VIN = 0 V VIN = 4 V ITRIP = 4 V ITRIP = 0 V V Ω mA Integrated Bootstrap Diode resistance Operational Amplifier Input Offset Voltage CA- Input Bias Current Operational Amplifier Common Mode Rejection Ratio Operational Amplifier Power Supply Rejection Ratio Operational Amplifier High Level Output Voltage Operational Amplifier Low Level Output Voltage VIN = 0 V, IO = 20 mA Ω mV nA dB VO = 0 V, VIN = 0 V PW ≤ 10 us VO = 15 V, VIN = 5 V PW ≤ 10 us VSO = 0.2 V VCA- = 1 V VSO = 0.1 V & 5 V VSO = 0.2 V VCC = 9.7 V & 20 V — 75 — 4.8 5.2 5.6 V VCA- = 0 V, VSO =1 V — — 40 mV VCA- = 1 V, VSO =0 V Note: The integrated bootstrap diode does not work well with the trapezoidal control. www.irf.com 6 IRS233(0,2)(D)(S&J)PbF Static Electrical Characteristics- Continued VBIAS (VCC, VBS1,2,3) = 15 V, VSO1,2,3 = VSS and TA = 25 °C unless otherwise specified. The VIN, VTH and IIN parameters are referenced to VSS and are applicable to all six logic input leads: HIN1,2,3 & LIN1,2,3. The VO and IO parameters are referenced to VSO1,2,3 and are applicable to the respective output leads: HO1,2,3 or LO1,2,3. Symbol Definition ISRC,AMP Operational Amplifier Output Source Current — -7 -4 ISNK,AMP Operational Amplifier Output Sink Current 1 2.1 — -30 -10 — — 4 — IO+,AMP IO-,AMP Min Typ Max Units Test Conditions Operational Amplifier Output High Short Circuit Current Operational Amplifier Output Low Short Circuit Current mA VCA- = 0 V, VSO =1 V VCAO = 4 V VCA- = 1 V, VSO =0 V VCAO = 2 V VCA- = 0 V, VSO =5 V VCAO = 0 V VCA- = 5 V, VSO =0 V VCAO = 5 V Functional Block Diagram IRS2330D/IRS2332D INPUT SIGNAL GENERATOR HIN1 HIN2 H1 L1 PULSE GENERATOR LEVEL SHIFTER SET RESET VB1 LATCH UV DETECTOR HIN3 DRIVER HO1 VS1 Integrated BS Diode LIN1 INPUT SIGNAL GENERATOR LIN2 H2 L2 LIN3 PULSE GENERATOR LEVEL SHIFTER SET VB2 LATCH UV RESET DETECTOR DRIVER HO2 VS2 Integrated BS Diode FAULT FAULT LOGIC CLEAR LOGIC C S INPUT SIGNAL GENERATOR H3 L3 PULSE GENERATOR LEVEL SHIFTER SET RESET LATCH UV DETECTOR VB3 DRIVER HO3 VS3 Integrated BS Diode VCC DRIVER ITRIP 0.5V CURRENT COMPARATOR LO1 UNDER VOLTAGE DETECTOR DRIVER LO2 DRIVER LO3 CAO CURRENT AMP CA- VSO VSS Note: IRS2330 & IRS2332 are without integrated bootstrap diode. www.irf.com 7 IRS233(0,2)(D)(S&J)PbF Lead Definitions Symbol HIN1,2,3 LIN1,2,3 FAULT VCC Description Logic input for high side gate driver outputs (HO1,2,3), out of phase Logic input for low side gate driver output (LO1,2,3), out of phase Indicates over-current or undervoltage lockout (low side) has occurred, negative logic Low side and logic fixed supply ITRIP Input for over-current shutdown CAO Output of current amplifier CA- Negative input of current amplifier VSS VB1,2,3 HO1,2,3 VS1,2,3 Logic Ground High side floating supply High side gate drive output High side floating supply return LO1,2,3 Low side gate drive output VSO Low side return and positive input of current amplifier Lead Assignments www.irf.com 8 IRS233(0,2)(D)(S&J)PbF Application Information and Additional Details Information regarding the following topics are included as subsections within this section of the datasheet. • • • • • • • • • • • • • • • • • • • • IGBT/MOSFET Gate Drive Switching and Timing Relationships Deadtime Matched Propagation Delays Input Logic Compatibility Undervoltage Lockout Protection Shoot-Through Protection Fault Reporting Over-Current Protection Over-Temperature Shutdown Protection Truth Table: Undervoltage lockout, ITRIP Advanced Input Filter Short-Pulse / Noise Rejection Integrated Bootstrap Functionality Bootstrap Power Supply Design Separate Logic and Power Grounds Negative VS Transient SOA DC- bus Current Sensing PCB Layout Tips Additional Documentation IGBT/MOSFET Gate Drive The IRS233(2,0)(D) HVICs are designed to drive up to six MOSFET or IGBT power devices. Figures 1 and 2 illustrate several parameters associated with the gate drive functionality of the HVIC. The output current of the HVIC, used to drive the gate of the power switch, is defined as IO. The voltage that drives the gate of the external power switch is defined as VHO for the highside power switch and VLO for the low-side power switch; this parameter is sometimes generically called VOUT and in this case does not differentiate between the high-side or low-side output voltage. Figure 1: HVIC sourcing current www.irf.com Figure 2: HVIC sinking current 9 IRS233(0,2)(D)(S&J)PbF Switching and Timing Relationships The relationship between the input and output signals of the IRS233(0,2)(D) are illustrated below in Figures 3. From these figures, we can see the definitions of several timing parameters (i.e., PWIN, PWOUT, tON, tOFF, tR, and tF) associated with this device. LINx (or HINx) 50% 50% PWIN tON LOx (or HOx) tR PWOUT 90% 10% tOFF tF 90% 10% Figure 3: Switching time waveforms The following two figures illustrate the timing relationships of some of the functionality of the IRS233(0,2)(D); this functionality is described in further detail later in this document. During interval A of Figure 4, the HVIC has received the command to turn-on both the high- and low-side switches at the same time; as a result, the shoot-through protection of the HVIC has prevented this condition and both the high- and low-side output are held in the off state. Interval B of Figures 4 shows that the signal on the ITRIP input pin has gone from a low to a high state; as a result, all of the gate drive outputs have been disabled (i.e., see that HOx has returned to the low state; LOx is also held low) and a fault is reported by the FAULT output transitioning to the low state. Once the ITRIP input has returned to the low state, the fault condition is latched until the all LINx become high. www.irf.com 10 IRS233(0,2)(D)(S&J)PbF Figure 4: Input/output timing diagram Deadtime This family of HVICs features integrated deadtime protection circuitry. The deadtime for these ICs is fixed; other ICs within IR’s HVIC portfolio feature programmable deadtime for greater design flexibility. The deadtime feature inserts a time period (a minimum deadtime) in which both the high- and low-side power switches are held off; this is done to ensure that the power switch being turned off has fully turned off before the second power switch is turned on. This minimum deadtime is automatically inserted whenever the external deadtime is shorter than DT; external deadtimes larger than DT are not modified by the gate driver. Figure 5 illustrates the deadtime period and the relationship between the output gate signals. The deadtime circuitry of the IRS233(0,2)(D) is matched with respect to the high- and low-side outputs of a given channel; additionally, the deadtimes of each of the three channels are matched. LINx HINx 50% LOx HOx DT 50% DT 50% 50% Figure 5: Illustration of deadtime www.irf.com 11 IRS233(0,2)(D)(S&J)PbF Matched Propagation Delays The IRS233(0,2)(D) family of HVICs is designed with propagation delay matching circuitry. With this feature, the IC’s response at the output to a signal at the input requires approximately the same time duration (i.e., tON, tOFF) for both the lowside channels and the high-side channels. Additionally, the propagation delay for each low-side channel is matched when compared to the other low-side channels and the propagation delays of the high-side channels are matched with each other. The propagation turn-on delay (tON) of the IRS233(0,2)(D) is matched to the propagation turn-on delay (tOFF). Input Logic Compatibility The inputs of this IC are compatible with standard CMOS and TTL outputs. The IRS233(0,2)(D) family has been designed to be compatible with 3.3 V and 5 V logic-level signals. The IRS233(0,2)(D) features an integrated 5.2 V Zener clamp on the HIN, LIN, and ITRIP pins. Figure 6 illustrates an input signal to the IRS233(0,2)(D), its input threshold values, and the logic state of the IC as a result of the input signal. Figure 6: HIN & LIN input thresholds Undervoltage Lockout Protection This family of ICs provides undervoltage lockout protection on both the VCC (logic and low-side circuitry) power supply and the VBS (high-side circuitry) power supply. Figure 7 is used to illustrate this concept; VCC (or VBS) is plotted over time and as the waveform crosses the UVLO threshold (VCCUV+/- or VBSUV+/-) the undervoltage protection is enabled or disabled. Upon power-up, should the VCC voltage fail to reach the VCCUV+ threshold, the IC will not turn-on. Additionally, if the VCC voltage decreases below the VCCUV- threshold during operation, the undervoltage lockout circuitry will recognize a fault condition and shutdown the high- and low-side gate drive outputs, and the FAULT pin will transition to the low state to inform the controller of the fault condition. Upon power-up, should the VBS voltage fail to reach the VBSUV threshold, the IC will not turn-on. Additionally, if the VBS voltage decreases below the VBSUV threshold during operation, the undervoltage lockout circuitry will recognize a fault condition, and shutdown the high-side gate drive outputs of the IC. The UVLO protection ensures that the IC drives the external power devices only when the gate supply voltage is sufficient to fully enhance the power devices. Without this feature, the gates of the external power switch could be driven with a low voltage, resulting in the power switch conducting current while the channel impedance is high; this could result in very high conduction losses within the power device and could lead to power device failure. www.irf.com 12 IRS233(0,2)(D)(S&J)PbF Figure 7: UVLO protection Shoot-Through Protection The IRS233(0,2)(D) family of high-voltage ICs is equipped with shoot-through protection circuitry (also known as crossconduction prevention circuitry). Figure 8 shows how this protection circuitry prevents both the high- and low-side switches from conducting at the same time. Table 1 illustrates the input/output relationship of the devices in the form of a truth table. Note that the IRS233(0,2)(D) has inverting inputs (the output is out-of-phase with its respective input). Shoot-through protection enabled HIN LIN HO LO Figure 8: Illustration of shoot-through protection circuitry IRS233(0,2)(D) HIN LIN HO LO 0 0 0 0 0 1 1 0 1 0 0 1 1 1 0 0 Table 1: Input/output truth table www.irf.com 13 IRS233(0,2)(D)(S&J)PbF Fault Reporting The IRS233(0,2)(D) family provides an integrated fault reporting output. There are two situations that would cause the HVIC to report a fault via the FAULT pin. The first is an undervoltage condition of VCC and the second is if the ITRIP pin recognizes a fault. Once the fault condition occurs, the FAULT pin is internally pulled to VSS and the fault condition is latched. The fault output stays in the low state until the fault condition has been removed by all LINx set to high state. Once the fault is removed, the voltage on the FAULT pin will return to VCC. Over-Current Protection The IRS233(0,2)(D) HVICs are equipped with an ITRIP input pin. This functionality can be used to detect over-current events in the DC- bus. Once the HVIC detects an over-current event through the ITRIP pin, the outputs are shutdown, a fault is reported through the FAULT pin. The level of current at which the over-current protection is initiated is determined by the resistor network (i.e., R0, R1, and R2) connected to ITRIP as shown in Figure 9, and the ITRIP threshold (VIT,TH+). The circuit designer will need to determine the maximum allowable level of current in the DC- bus and select R0, R1, and R2 such that the voltage at node VX reaches the over-current threshold (VIT,TH+) at that current level. VIT,TH+ = R0IDC-(R1/(R1+R2)) Vcc HIN(x3) VB ( x3) LIN(x3) HO( x3) FAULT VS (x3) LO(x3) ITRIP COM VSS R1 R2 R0 IDC- Figure 9: Programming the over-current protection For example, a typical value for resistor R0 could be 50 mΩ. The voltage of the ITRIP pin should not be allowed to exceed 5 V; if necessary, an external voltage clamp may be used. Over-Temperature Shutdown Protection The ITRIP input of the IRS233(0,2)(D) can also be used to detect over-temperature events in the system and initiate a shutdown of the HVIC (and power switches) at that time. In order to use this functionality, the circuit designer will need to design the resistor network as shown in Figure 10 and select the maximum allowable temperature. This network consists of a thermistor and two standard resistors R3 and R4. As the temperature changes, the resistance of the thermistor will change; this will result in a change of voltage at node VX. The resistor values should be selected such the voltage VX should reach the threshold voltage (VIT,TH+) of the ITRIP functionality by the time that the maximum allowable temperature is reached. The voltage of the ITRIP pin should not be allowed to exceed 5 V. When using both the over-current protection and over-temperature protection with the ITRIP input, OR-ing diodes (e.g., DL4148) can be used. This network is shown in Figure 11; the OR-ing diodes have been labeled D1 and D2. www.irf.com 14 IRS233(0,2)(D)(S&J)PbF Figure 10: Programming over-temperature protection Figure 11: Using over-current protection and over-temperature protection Truth Table: Undervoltage lockout and ITRIP Table 2 provides the truth table for the IRS233(0,2)(D). The first line shows that the UVLO for VCC has been tripped; the FAULT output has gone low and the gate drive outputs have been disabled. VCCUV is not latched in this case and when VCC is greater than VCCUV, the FAULT output returns to the high impedance state. The second case shows that the UVLO for VBS has been tripped and that the high-side gate drive outputs have been disabled. After VBS exceeds the VBSUV threshold, HO will stay low until the HVIC input receives a new falling transition of HIN. The third case shows the normal operation of the HVIC. The fourth case illustrates that the ITRIP trip threshold has been reached and that the gate drive outputs have been disabled and a fault has been reported through the fault pin. The fault output stays in the low state until the fault condition has been removed by all LINx set to high state. Once the fault is removed, the voltage on the FAULT pin will return to VCC. UVLO VCC UVLO VBS Normal operation ITRIP fault VCC <VCCUV 15 V 15 V 15 V VBS --<VBSUV 15 V 15 V ITRIP --0V 0V >VITRIP FAULT 0 High impedance High impedance 0 LO 0 LIN LIN 0 HO 0 0 HIN 0 Table 2: IRS233(0,2)(D) UVLO, ITRIP & FAULT truth table Advanced Input Filter The advanced input filter allows an improvement in the input/output pulse symmetry of the HVIC and helps to reject noise spikes and short pulses. This input filter has been applied to the HIN and LIN. The working principle of the new filter is shown in Figures 12 and 13. Figure 12 shows a typical input filter and the asymmetry of the input and output. The upper pair of waveforms (Example 1) show an input signal with a duration much longer then tFIL,IN; the resulting output is approximately the difference between the input signal and tFIL,IN. The lower pair of waveforms (Example 2) show an input signal with a duration slightly longer then tFIL,IN; the resulting output is approximately the difference between the input signal and tFIL,IN. Figure 13 shows the advanced input filter and the symmetry between the input and output. The upper pair of waveforms (Example 1) show an input signal with a duration much longer then tFIL,IN; the resulting output is approximately the same duration as the input signal. The lower pair of waveforms (Example 2) show an input signal with a duration slightly longer then tFIL,IN; the resulting output is approximately the same duration as the input signal. www.irf.com 15 IRS233(0,2)(D)(S&J)PbF Figure 12: Typical input filter Figure 13: Advanced input filter Short-Pulse / Noise Rejection Example 2 Example 1 This device’s input filter provides protection against short-pulses (e.g., noise) on the input lines. If the duration of the input signal is less than tFIL,IN, the output will not change states. Example 1 of Figure 14 shows the input and output in the low state with positive noise spikes of durations less than tFIL,IN; the output does not change states. Example 2 of Figure 19 shows the input and output in the high state with negative noise spikes of durations less than tFIL,IN; the output does not change states. Figure 14: Noise rejecting input filters Figures 15 and 16 present lab data that illustrates the characteristics of the input filters while receiving ON and OFF pulses. The input filter characteristic is shown in Figure 15; the left side illustrates the narrow pulse ON (short positive pulse) characteristic while the left shows the narrow pulse OFF (short negative pulse) characteristic. The x-axis of Figure 20 shows the duration of PWIN, while the y-axis shows the resulting PWOUT duration. It can be seen that for a PWIN duration less than tFIL,IN, that the resulting PWOUT duration is zero (e.g., the filter rejects the input signal/noise). We also see that once the PWIN duration exceed tFIL,IN, that the PWOUT durations mimic the PWIN durations very well over this interval with the symmetry improving as the duration increases. To ensure proper operation of the HVIC, it is suggested that the input pulse width for the high-side inputs be ≥ 500 ns. The difference between the PWOUT and PWIN signals of both the narrow ON and narrow OFF cases is shown in Figure 16; the careful reader will note the scale of the y-axis. The x-axis of Figure 21 shows the duration of PWIN, while the y-axis shows the resulting PWOUT–PWIN duration. This data illustrates the performance and near symmetry of this input filter. www.irf.com 16 IRS233(0,2)(D)(S&J)PbF Narrow Pulse OFF 1000 PWOUT PWIN Time (ns) 800 600 400 200 0 0 200 400 600 800 1000 Time (ns) Figure 15: IRS233(0,2)(D) input filter characteristic Figure 16: Difference between the input pulse and the output pulse Integrated Bootstrap Functionality The new IRS233(0,2)D family features integrated high-voltage bootstrap MOSFETs that eliminate the need of the external bootstrap diodes and resistors in many applications. There is one bootstrap MOSFET for each high-side output channel and it is connected between the VCC supply and its respective floating supply (i.e., VB1, VB2, VB3); see Figure 17 for an illustration of this internal connection. The integrated bootstrap MOSFET is turned on only during the time when LO is ‘high’, and it has a limited source current due to RBS. The VBS voltage will be charged each cycle depending on the on-time of LO and the value of the CBS capacitor, the drain-source (collector-emitter) drop of the external IGBT (or MOSFET), and the low-side free-wheeling diode drop. The bootstrap MOSFET of each channel follows the state of the respective low-side output stage (i.e., the bootstrap MOSFET is ON when LO is high, it is OFF when LO is low), unless the VB voltage is higher than approximately 110% of VCC. In that case, the bootstrap MOSFET is designed to remain off until VB returns below that threshold; this concept is illustrated in Figure 18. www.irf.com 17 IRS233(0,2)(D)(S&J)PbF Figure 17: Internal bootstrap MOSFET connection Figure 18: Bootstrap MOSFET state diagram A bootstrap MOSFET is suitable for most of the PWM modulation schemes and can be used either in parallel with the external bootstrap network (i.e., diode and resistor) or as a replacement of it. The use of the integrated bootstrap as a replacement of the external bootstrap network may have some limitations. An example of this limitation may arise when this functionality is used in non-complementary PWM schemes (typically 6-step modulations) and at very high PWM duty cycle. In these cases, superior performances can be achieved by using an external bootstrap diode in parallel with the internal bootstrap network. Bootstrap Power Supply Design For information related to the design of the bootstrap power supply while using the integrated bootstrap functionality of the IRS233(0,2)D family, please refer to Application Note 1123 (AN-1123) entitled “Bootstrap Network Analysis: Focusing on the Integrated Bootstrap Functionality.” This application note is available at www.irf.com. For information related to the design of a standard bootstrap power supply (i.e., using an external discrete diode) please refer to Design Tip 04-4 (DT04-4) entitled “Using Monolithic High Voltage Gate Drivers.” This design tip is available at www.irf.com. Separate Logic and Power Grounds The IRS233(0,2)(D) has separate logic and power ground pin (VSS and VSO respectively) to eliminate some of the noise problems that can occur in power conversion applications. Current sensing shunts are commonly used in many applications for power inverter protection (i.e., over-current protection), and in the case of motor drive applications, for motor current measurements. In these situations, it is often beneficial to separate the logic and power grounds. Figure 19 shows a HVIC with separate VSS and VSO pins and how these two grounds are used in the system. The VSS is used as the reference point for the logic and over-current circuitry; VX in the figure is the voltage between the ITRIP pin and the VSS pin. Alternatively, the VSO pin is the reference point for the low-side gate drive circuitry. The output voltage used to drive the low-side gate is VLO-VSO; the gate-emitter voltage (VGE) of the low-side switch is the output voltage of the driver minus the drop across RG,LO. www.irf.com 18 IRS233(0,2)(D)(S&J)PbF DC+ BUS DBS VB (x3) VCC HO (x3) HVIC ITRIP VSS CBS RG,HO VS (x3) LO (x3) COM VS1 VS2 VS3 RG,LO + + + VGE1 VGE2 VGE3 - - - R2 R0 + VX R1 - DC- BUS Figure 19: Separate VSS and VSO pins Negative VS Transient SOA A common problem in today’s high-power switching converters is the transient response of the switch node’s voltage as the power switches transition on and off quickly while carrying a large current. A typical 3-phase inverter circuit is shown in Figure 20; here we define the power switches and diodes of the inverter. If the high-side switch (e.g., the IGBT Q1 in Figures 21 and 22) switches off, while the U phase current is flowing to an inductive load, a current commutation occurs from high-side switch (Q1) to the diode (D2) in parallel with the low-side switch of the same inverter leg. At the same instance, the voltage node VS1, swings from the positive DC bus voltage to the negative DC bus voltage. Figure 20: Three phase inverter www.irf.com 19 IRS233(0,2)(D)(S&J)PbF DC+ BUS DC+ BUS Q1 ON Q1 OFF D1 IU VS1 Q2 OFF VS1 D2 DC- BUS Figure 21: Q1 conducting Q2 OFF IU D2 DC- BUS Figure 22: D2 conducting Also when the V phase current flows from the inductive load back to the inverter (see Figures 23 and 24), and Q4 IGBT switches on, the current commutation occurs from D3 to Q4. At the same instance, the voltage node, VS2, swings from the positive DC bus voltage to the negative DC bus voltage. DC+ BUS Q3 OFF D3 VS2 IV Q4 ON DC- BUS Figure 23: D3 conducting Figure 24: Q4 conducting However, in a real inverter circuit, the VS voltage swing does not stop at the level of the negative DC bus, rather it swings below the level of the negative DC bus. This undershoot voltage is called “negative VS transient”. The circuit shown in Figure 25 depicts one leg of the three phase inverter; Figures 26 and 27 show a simplified illustration of the commutation of the current between Q1 and D2. The parasitic inductances in the power circuit from the die bonding to the PCB tracks are lumped together in LC and LE for each IGBT. When the high-side switch is on, VS1 is below the DC+ voltage by the voltage drops associated with the power switch and the parasitic elements of the circuit. When the high-side power switch turns off, the load current momentarily flows in the low-side freewheeling diode due to the inductive load connected to VS1 (the load is not shown in these figures). This current flows from the DC- bus (which is connected to the VSO pin of the HVIC) to the load and a negative voltage between VS1 and the DC- Bus is induced (i.e., the VSO pin of the HVIC is at a higher potential than the VS pin). www.irf.com 20 IRS233(0,2)(D)(S&J)PbF Figure 25: Parasitic Elements Figure 26: VS positive Figure 27: VS negative In a typical motor drive system, dV/dt is typically designed to be in the range of 3-5 V/ns. The negative VS transient voltage can exceed this range during some events such as short circuit and over-current shutdown, when di/dt is greater than in normal operation. International Rectifier’s HVICs have been designed for the robustness required in many of today’s demanding applications. An indication of the IRS233(0,2)(D)’s robustness can be seen in Figure 28, where there is represented the IRS233(0,2)(D) Safe Operating Area at VBS=15V based on repetitive negative VS spikes. A negative VS transient voltage falling in the grey area (outside SOA) may lead to IC permanent damage; viceversa unwanted functional anomalies or permanent damage to the IC do not appear if negative Vs transients fall inside SOA. At VBS=15V in case of -VS transients greater than -16.5 V for a period of time greater than 50 ns; the HVIC will hold by design the high-side outputs in the off state for 4.5 μs. Figure 28: Negative VS transient SOA for IRS233(0,2)(D) Even though the IRS233(0,2)(D) has been shown able to handle these large negative VS transient conditions, it is highly recommended that the circuit designer always limit the negative VS transients as much as possible by careful PCB layout and component use. www.irf.com 21 IRS233(0,2)(D)(S&J)PbF DC- bus Current Sensing A ground referenced current signal amplifier has been included so that the current in the return leg of the DC bus may be monitored. A typical circuit configuration is provided in Fig.29. The signal coming from the shunt resistor is amplified by the ratio (R1+R2)/R2. Additional details can be found on Design Tip DT 92-6. This design tip is available at www.irf.com. Figure 29: Current amplifier typical configuration In the following Figures 30, 31, 32, 33 the configurations used to measure the operational amplifier characteristics are shown. 15V VCC VSO CAO CA- 0.2V VSS + 20K 1K VSO Figure 30: Operational Amplifier Slew rate measurement www.irf.com VSO 0.2V 21 Figure 31: Operational Amplifier Input Offset Voltage measurement 22 IRS233(0,2)(D)(S&J)PbF Figure 32: Operational Amplifier Common mode rejection measurement Figure 33: Operational Amplifier Power supply rejection measurement PCB Layout Tips Distance between high and low voltage components: It’s strongly recommended to place the components tied to the floating voltage pins (VB and VS) near the respective high voltage portions of the device. The IRS233(0,2)(D) in the PLCC44 package has had some unused pins removed in order to maximize the distance between the high voltage and low voltage pins. Please see the Case Outline PLCC44 information in this datasheet for the details. Ground Plane: In order to minimize noise coupling, the ground plane should not be placed under or near the high voltage floating side. Gate Drive Loops: Current loops behave like antennas and are able to receive and transmit EM noise (see Figure 34). In order to reduce the EM coupling and improve the power switch turn on/off performance, the gate drive loops must be reduced as much as possible. Moreover, current can be injected inside the gate drive loop via the IGBT collector-to-gate parasitic capacitance. The parasitic auto-inductance of the gate loop contributes to developing a voltage across the gate-emitter, thus increasing the possibility of a self turn-on effect. Figure 34: Antenna Loops www.irf.com 23 IRS233(0,2)(D)(S&J)PbF Supply Capacitor: It is recommended to place a bypass capacitor (CIN) between the VCC and VSS pins. This connection is shown in Figure 35. A ceramic 1 μF ceramic capacitor is suitable for most applications. This component should be placed as close as possible to the pins in order to reduce parasitic elements. Vcc HIN(x3) VB ( x3) LIN(x3) I RS233(0,2)(D) FAULT ITRIP HO( x3) VS (x3) LO(x3) COM VSS R1 R2 R0 IDC- Figure 35: Supply capacitor Routing and Placement: Power stage PCB parasitic elements can contribute to large negative voltage transients at the switch node; it is recommended to limit the phase voltage negative transients. In order to avoid such conditions, it is recommended to 1) minimize the high-side emitter to low-side collector distance, and 2) minimize the low-side emitter to negative bus rail stray inductance. However, where negative VS spikes remain excessive, further steps may be taken to reduce the spike. This includes placing a resistor (5 Ω or less) between the VS pin and the switch node (see Figure 36), and in some cases using a clamping diode between VSS and VS (see Figure 37). See DT04-4 at www.irf.com for more detailed information. Figure 36: VS resistor Figure 37: VS clamping diode Additional Documentation Several technical documents related to the use of HVICs are available at www.irf.com; use the Site Search function and the document number to quickly locate them. Below is a short list of some of these documents. DT97-3: Managing Transients in Control IC Driven Power Stages AN-1123: Bootstrap Network Analysis: Focusing on the Integrated Bootstrap Functionality DT04-4: Using Monolithic High Voltage Gate Drivers AN-978: HV Floating MOS-Gate Driver ICs www.irf.com 24 IRS233(0,2)(D)(S&J)PbF Parameter Temperature Trends Figures 38-76 provide information on the experimental performance of the IRS233(0,2)(D)(S&J) HVIC. The line plotted in each figure is generated from actual lab data. A small number of individual samples were tested at three temperatures (-40 ºC, 25 ºC, and 125 ºC) in order to generate the experimental (Exp.) curve. The line labeled Exp. consist of three data points (one data point at each of the tested temperatures) that have been connected together to illustrate the understood temperature trend. The individual data points on the curve were determined by calculating the averaged experimental value of the parameter (for a given temperature). 800 800 700 700 600 600 Exp. Exp. 500 tON (ns) tON (ns) 500 400 400 300 300 200 200 100 100 0 -50 -25 0 25 50 75 100 0 -50 125 -25 0 Fig. 38. Turn-on Propagation Delay vs. Temperature 800 700 700 tOFF (ns) tOFF (ns) 500 400 300 300 100 100 25 50 75 100 Temperature (o C) Fig. 40. Turn-off Propagation Delay vs. Temperature www.irf.com 125 400 200 0 100 500 200 -25 75 Exp. 600 Exp. 0 -50 50 Fig. 39. Turn-on Propagation Delay vs. Temperature 800 600 25 Temperature (o C) Temperature (o C) 125 0 -50 -25 0 25 50 75 100 Temperature (o C) Fig. 41. Turn-off Propagation Delay vs. Temperature 25 125 IRS233(0,2)(D)(S&J)PbF 200 60 180 50 160 40 120 tF (ns) tR (ns) 140 100 80 Exp. 60 30 Exp. 20 40 10 20 0 -50 -25 0 25 50 75 100 0 -50 125 -25 0 25 Temperature (oC) Fig. 42. Turn-on Rise Time vs. Temperature 1000 900 900 800 700 600 tFLT (ns) tITRIP (ns) 100 125 800 Exp. 500 400 500 400 300 200 200 100 100 -25 0 25 50 75 100 Exp. 600 300 0 -50 0 -50 125 -25 0 Temperature (oC) 50 75 100 125 Fig. 45. ITRIP to FAULT Indication Delay vs. Temperature 16000 1200 14000 1000 Exp. Exp. DLTon1 (ns) 12000 25 Temperature (oC) Fig. 44. ITRIP to Output Shutdown Propagation Delay vs. Temperature TFLTCLR (ns) 75 Fig.43. Turn-off Fall Time vs. Temperature 1000 700 50 Temperature (oC) 10000 8000 6000 800 600 400 4000 200 2000 0 -50 -25 0 25 50 75 100 Temperature (oC) Fig.46. FAULT Clear Time vs. Temperature www.irf.com 125 0 -50 -25 0 25 50 75 100 Temperature (oC) Fig. 47. Dead Time vs. Temperature 26 125 IRS233(0,2)(D)(S&J)PbF 60 6 50 5 SR-_Amp (V/uS) SR+_Amp (V/uS) Exp. 40 30 20 4 3 2 Exp. 10 1 0 -50 -25 0 25 50 75 100 0 -50 125 -25 0 25 Temperature (oC) Fig. 48. Operational Amplifier Slew Rate (+) vs. Temperature 2.0 LIN1_VTH- (V) LIN1_VTH+ (V) Exp. 1.0 0.5 Exp. 1.5 1.0 -25 0 25 50 75 100 0.0 -50 125 -25 0 Temperature (oC) 25 50 75 100 125 Temperature (oC) Fig. 50. Input Positive Going Threshold vs. Temperature Fig. 51. Input Negative Going Threshold vs. Temperature 800 800 700 700 600 600 VIT,TH- (mV) VIT,TH+ (mV) 125 0.5 0.0 -50 EXP. 400 300 500 400 Exp. 300 200 200 100 100 0 -50 100 2.5 1.5 500 75 Fig. 49. Operational Amplifier Slew Rate (-) vs. Temperature 2.5 2.0 50 Temperature (oC) 0 -25 0 25 50 75 100 Temperature (oC) Fig. 52. ITRIP Input Positive Going Threshold vs. Temperature www.irf.com 125 -50 -25 0 25 50 75 100 125 Temperature (oC) Fig. 53. ITRIP Input Negative Going Threshold vs. Temperature 27 IRS233(0,2)(D)(S&J)PbF 60 450 400 50 ileak1_VCCMAX (µA) VOL_LO1 (mV) 350 300 250 200 150 Exp. 100 40 30 20 10 Exp. 50 0 0 -50 -25 0 25 50 75 100 125 -50 -25 0 Temperature (oC) 25 50 75 100 125 Temperature (oC) Fig. 54. Low Level Output Voltage vs. Temperature Fig. 55. Offset Supply Leakage Current vs. Temperature 7 12 6 10 Exp. 5 Exp. IQCC0 (mA) IQCC1 (mA) 8 6 4 4 3 2 2 1 0 -50 -25 0 25 50 75 100 0 -50 125 -25 0 Temperature (oC) 80 80 70 70 60 60 50 50 Exp. 30 10 10 0 25 50 75 100 Temperature (oC) Fig. 58. Quiescent VBS Supply Current vs. Temperature www.irf.com 100 125 30 20 -25 75 125 Exp. 40 20 0 -50 50 Fig. 57. Quiescent VCC Supply Current vs. Temperature IQBS11 (μA) IQBS10 (μA) Fig. 56. Quiescent VCC Supply Current vs. Temperature 40 25 Temperature (oC) 0 -50 -25 0 25 50 75 100 Temperature (oC) Fig. 59. Quiescent VBS Supply Current vs. Temperature 28 125 IRS233(0,2)(D)(S&J)PbF 9.6 9.8 9.4 9.6 9.2 9.4 VCCUV+ (V) VCCUV- (V) 9.0 8.8 8.6 Exp. 8.4 9.2 9.0 Exp. 8.8 8.6 8.2 8.4 8.0 7.8 -50 -25 0 25 50 75 100 8.2 -50 125 -25 0 25 Temperature (oC) Fig. 60. VCC Supply Undervoltage Negative Going Threshold vs. Temperature 100 125 9.5 9.0 8.5 8.5 8.0 Exp. Exp. VBSUV+ (V) VBSUV- (V) 75 Fig. 61. VCC Supply Undervoltage Positive Going Threshold vs. Temperature 9.0 7.5 7.0 8.0 7.5 7.0 6.5 6.5 6.0 -50 -25 0 25 50 75 100 6.0 -50 125 -25 0 25 50 75 100 Fig. 62. VBS Supply Undervoltage Negative Going Threshold vs. Temperature Fig. 63. VBS Supply Undervoltage Positive Going Threshold vs. Temperature 0 -50 -50 90 80 70 -25 0 25 50 75 100 -100 60 -150 50 I O+ (mA) Exp. 40 30 -200 -250 Exp. -300 20 -350 10 0 -50 -400 -25 0 25 50 75 100 Temperature (oC) Fig. 64. FAULT Low On-Resistance vs. Temperature www.irf.com 125 Temperature (oC) Temperature (o C) RON,FLT (Ω) 50 Temperature (oC) 125 -450 Temperature (oC) Fig. 65. Output High Short Circuit Pulsed Current vs. Temperature 29 125 IRS233(0,2)(D)(S&J)PbF 20 706 Exp. 606 15 10 VOS_AMP (mV) IO- (mA) 506 406 306 206 5 0 -50 -5 Exp. -25 0 25 50 75 100 125 -10 106 -15 6 -50 -25 0 25 50 75 100 125 -20 Temperature (oC) Fig. 67. Offset Opamp vs. Temperature 200 200 180 180 160 160 140 140 CMRR_AMP (dB) PSRR_AMP (dB) Fig. 66. Output Low Short Circuit Pulsed Current vs. Temperature 120 100 Exp. 80 60 100 Exp. 80 60 40 20 20 -25 0 25 50 Temperature (oC) 75 100 0 -50 125 Fig. 68. Operational Amplifier Power Supply Rejection Ratio vs. Temperature 5.6 35 5.5 30 VOH_AMP (mV) 5.3 5.2 5.1 Exp. 5.0 0 25 50 Temperature (oC) 75 100 125 25 20 Exp. 15 10 5 4.9 4.8 -50 -25 Fig. 69. Operational Amplifier Common Mode Rejection Ratio vs. Temperature 5.4 VOH_AMP (V) 120 40 0 -50 Temperature (oC) -25 0 25 50 75 100 125 Temperature (oC) Fig. 70. Operational Amplifier High Level Output Voltage vs. Temperature www.irf.com 0 -50 -25 0 25 50 75 100 Temperature (oC) Fig. 71. Operational Amplifier Low Level Output Voltage vs. Temperature 30 125 IRS233(0,2)(D)(S&J)PbF 16 6 14 5 Exp. 4 Io-_AMP (mA) Isnk_AMP (mA) 12 Exp. 3 2 10 8 6 4 1 2 0 -50 -25 0 25 50 75 100 0 -50 125 -25 0 Temperature (oC) Fig. 72. Operational Amplifier Output Sink Current vs. Temperature 0 -50 -2 -25 0 25 50 75 100 0 -50 125 75 100 -25 0 25 50 75 100 -5 -10 Io+_AMP (mA) Isrc_AMP (mA) 50 -6 -8 Exp. -15 Exp. -20 -25 -12 -30 -14 -16 -35 Temperature (oC) Temperature (oC) Fig. 74. Operational Amplifier Output Source Current vs. Temperature 0 -50 -25 0 Fig. 75. Operational Amplifier Output High Short Circuit Current vs. Temperature 25 50 75 100 125 Vs1_RST_domin (V) -2 -4 -6 -8 -10 Exp. -12 -14 Temperature (o C) Fig. 76. Max –Vs vs. Temperature www.irf.com 125 Fig. 73. Operational Amplifier Output Low Short Circuit Current vs. Temperature -4 -10 25 Temperature (oC) 31 125 IRS233(0,2)(D)(S&J)PbF Case Outlines www.irf.com 32 IRS233(0,2)(D)(S&J)PbF Case Outlines www.irf.com 33 IRS233(0,2)(D)(S&J)PbF Tape and Reel Details: SOIC28W LOADED TAPE FEED DIRECTION A B H D F C NOTE : CONTROLLING DIM ENSION IN M M E G CARRIER TAPE DIMENSION FOR Metric Code Min Max A 11.90 12.10 B 3.90 4.10 C 23.70 24.30 D 11.40 11.60 E 10.80 11.00 F 18.20 18.40 G 1.50 n/a H 1.50 1.60 28SOICW Imperial Min Max 0.468 0.476 0.153 0.161 0.933 0.956 0.448 0.456 0.425 0.433 0.716 0.724 0.059 n/a 0.059 0.062 F D C B A E G H REEL DIMENSIONS FOR 28SOICW Metric Imperial Code Min Max Min Max A 329.60 330.25 12.976 13.001 B 20.95 21.45 0.824 0.844 C 12.80 13.20 0.503 0.519 D 1.95 2.45 0.767 0.096 E 98.00 102.00 3.858 4.015 F n/a 30.40 n/a 1.196 G 26.50 29.10 1.04 1.145 H 24.40 26.40 0.96 1.039 www.irf.com 34 IRS233(0,2)(D)(S&J)PbF Tape and Reel Details: PLCC44 LOADED TAPE FEED DIRECTION A B H D F C NOTE : CONTROLLING DIM ENSION IN M M E G CARRIER TAPE DIMENSION FOR Metric Code Min Max A 23.90 24.10 B 3.90 4.10 C 31.70 32.30 D 14.10 14.30 E 17.90 18.10 F 17.90 18.10 G 2.00 n/a H 1.50 1.60 44PLCC Imperial Min Max 0.94 0.948 0.153 0.161 1.248 1.271 0.555 0.562 0.704 0.712 0.704 0.712 0.078 n/a 0.059 0.062 F D C B A E G H REEL DIMENSIONS FOR 44PLCC Metric Code Min Max A 329.60 330.25 B 20.95 21.45 C 12.80 13.20 D 1.95 2.45 E 98.00 102.00 F n/a 38.4 G 34.7 35.8 H 32.6 33.1 www.irf.com Imperial Min Max 12.976 13.001 0.824 0.844 0.503 0.519 0.767 0.096 3.858 4.015 n/a 1.511 1.366 1.409 1.283 1.303 35 IRS233(0,2)(D)(S&J)PbF Ordering Information Base Part Number Package Type SOIC28W IRS233(0,2)(D) PLCC44 Standard Pack Form Quantity Tube/Bulk 25 Tape and Reel 1000 Tube/Bulk 27 Tape and Reel 500 Complete Part Number IRS233(0,2)(D)SPbF IRS233(0,2)(D)STRPbF IRS233(0,2)(D)JPbF IRS233(0,2)(D)JTRPbF The information provided in this document is believed to be accurate and reliable. However, International Rectifier assumes no responsibility for the consequences of the use of this information. International Rectifier assumes no responsibility for any infringement of patents or of other rights of third parties which may result from the use of this information. No license is granted by implication or otherwise under any patent or patent rights of International Rectifier. The specifications mentioned in this document are subject to change without notice. This document supersedes and replaces all information previously supplied. For technical support, please contact IR’s Technical Assistance Center http://www.irf.com/technical-info/ WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245 Tel: (310) 252-7105 www.irf.com 36