19-3758; Rev 0; 8/05 KIT ATION EVALU E L B AVAILA High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor PART TEMP RANGE -40°C to +85°C 56 Thin QFN-EP MAX19586ETN+ -40°C to +85°C 56 Thin QFN-EP MAX19586ETN T5688-2 DVDD DGND DGND D0 D1 D5 D6 D7 D8 DVDD DVDD TOP VIEW 42 41 40 39 38 37 36 35 34 33 32 31 30 29 D9 43 28 AGND D10 44 27 REFIN Multistandard Receivers D11 45 26 REFOUT D12 46 25 AVDD E911 Location Receivers D13 47 24 AVDD D14 48 23 AVDD High-Performance Instrumentation D15 49 22 AGND MAX19586 DAV 50 Antenna Array Processing T5688-2 Pin Configuration Wireless Local Loop (WLL) Multicarrier Receivers PKG CODE +Denotes lead-free package. Applications Cellular Base-Station Transceiver Systems (BTS) PIN-PACKAGE D2 The MAX19586 features parallel, low-voltage CMOScompatible outputs in two’s-complement output format. The MAX19586 is manufactured in an 8mm x 8mm, 56-pin thin QFN package with exposed paddle (EP) for low thermal resistance, and is specified for the extended industrial (-40°C to +85°C) temperature range. Ordering Information D3 The MAX19586 operates from a 3.3V analog supply voltage and a 1.8V digital voltage, features a 2.56VP-P full-scale input range, and allows for a guaranteed sampling speed of up to 80Msps. The input track-and-hold stage operates with a 600MHz full-scale, full-power bandwidth. ♦ 80Msps Minimum Sampling Rate ♦ -82dBFS Noise Floor ♦ Excellent Dynamic Performance 80dB/79.2dB SNR at fIN = 10MHz/70MHz and -2dBFS 96dBc/102dBc Single-Tone SFDR1/ SFDR2 at fIN = 10MHz 84.3dBc/100dBc Single-Tone SFDR1/ SFDR2 at fIN = 70MHz ♦ Less than 0.1ps Sampling Jitter ♦ 1.1W Power Dissipation ♦ 2.56VP-P Fully Differential Analog Input Voltage Range ♦ CMOS-Compatible Two’s-Complement Data Output ♦ Separate Data Valid Clock and Over-Range Outputs ♦ Flexible Input Clock Buffer ♦ 3.3V Analog Power Supply; 1.8V Digital Output Supply ♦ Small 8mm x 8mm x 0.8mm 56-Pin Thin QFN Package ♦ EV Kit Available for MAX19586 (Order MAX19586EVKIT) D4 The MAX19586 is a 3.3V, high-speed, high-performance analog-to-digital converter (ADC) featuring a fully differential wideband track-and-hold (T/H) and a 16-bit converter core. The MAX19586 is optimized for multichannel, multimode receivers, which require the ADC to meet very stringent dynamic performance requirements. With a -82dBFS noise floor, the MAX19586 allows for the design of receivers with superior sensitivity requirements. At 80Msps, the MAX19586 achieves a 79.2dB signal-tonoise ratio (SNR) and an 84.3dBc/100dBc single-tone spurious-free dynamic range (SFDR) performance (SFDR1/SFDR2) at fIN = 70MHz. The MAX19586 is not only optimized for excellent dynamic performance in the 2nd Nyquist region, but also for high-IF input frequencies. For instance, at 130MHz, the MAX19586 achieves an 82.5dBc SFDR and its SNR performance stays flat (within 2.5dB) throughout the 4th Nyquist region. This level of performance makes the part ideal for high-performance digital receivers. Features 21 AGND DVDD 51 20 AGND DGND 52 19 AVDD DOR 53 18 AVDD N.C. 54 17 AVDD AVDD 55 16 N.C. 15 N.C. 10 11 12 13 14 AGND CLKN 9 AGND CLKP 8 AGND AGND 7 INN AVDD 6 INP 5 AGND 4 AGND 3 AGND 2 AGND 1 AVDD AVDD 56 THIN QFN 8mm x 8mm ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX19586 General Description MAX19586 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor ABSOLUTE MAXIMUM RATINGS AVDD to AGND ..................................................... -0.3V to +3.6V DVDD to DGND..................................................... -0.3V to +2.4V AGND to DGND.................................................... -0.3V to +0.3V INP, INN, CLKP, CLKN, REFP, REFN, REFIN, REFOUT to AGND....................-0.3V to (AVDD + 0.3V) D0–D15, DAV, DOR, DAV to GND...........-0.3V to (DVDD + 0.3V) Continuous Power Dissipation (TA = +70°C) 56-Pin Thin QFN (derate 47.6mW/°C above +70°C) .........................3809.5mW Operating Temperature Range ..........................-40°C to +85°C Thermal Resistance θJA ..................................................21°C/W Thermal Resistance θJC .................................................0.6°C/W Junction Temperature ......................................................+150°C Storage Temperature Range .............................-60°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (AVDD = 3.3V, DVDD = 1.8V, AGND = DGND = 0, INP and INN driven differentially, internal reference CLKP and CLKN driven differentially, CL = 5pF at digital outputs, fCLK = 80MHz, TA = TMIN to TMAX, unless otherwise noted. Typical values are at TA = +25°C, unless otherwise noted.) (Note 1) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS DC ACCURACY Resolution N Offset Error VOS 0 16 Gain Error GE -3.5 10 Bits 20 mV +3.5 %FS ANALOG INPUTS (INP, INN) Input Voltage Range VDIFF Fully differential input, VIN = VINP - VINN 2.56 VP-P Common-Mode Voltage VCM Internally self-biased 2.2 V Differential Input Resistance RIN 10 ±20% kΩ Differential Input Capacitance CIN 7 pF Full-Power Analog Bandwidth BW-3dB 600 MHz 1.28 ±10% V 1.28 V AIN < -35dBFS -82 dBFS fIN = 10MHz, AIN = -2dBFS 80 -3dB rolloff for FS Input REFERENCE INPUT/OUTPUT (REFIN, REFOUT) Reference Input Voltage Range REFIN Reference Output Voltage REFOUT DYNAMIC SPECIFICATIONS (fCLK = 80Msps) Thermal Plus Quantization Noise Floor Signal-to-Noise Ratio (First 4 Harmonics Excluded) (Notes 2, 3) NF fIN = 70MHz, AIN = -2dBFS SNR 77.5 78.5 fIN = 130MHz, AIN = -2dBFS 77.9 fIN = 168MHz, AIN = -2dBFS 77.2 fIN = 10MHz, AIN = -2dBFS Signal-to-Noise Plus Distortion (Notes 2, 3) 2 fIN = 70MHz, AIN = -2dBFS SINAD 79.2 fIN = 100MHz, AIN = -2dBFS dB 79.6 75 77.6 fIN = 100MHz, AIN = -2dBFS 77.4 fIN = 130MHz, AIN = -2dBFS 76.4 fIN = 168MHz, AIN = -2dBFS 72.7 _______________________________________________________________________________________ dB High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor (AVDD = 3.3V, DVDD = 1.8V, AGND = DGND = 0, INP and INN driven differentially, internal reference CLKP and CLKN driven differentially, CL = 5pF at digital outputs, fCLK = 80MHz, TA = TMIN to TMAX, unless otherwise noted. Typical values are at TA = +25°C, unless otherwise noted.) (Note 1) PARAMETER SYMBOL CONDITIONS MIN fIN = 10MHz, AIN = -2dBFS Spurious-Free Dynamic Range (Worst Harmonic, 2nd and 3rd) Spurious-Free Dynamic Range (Worst Harmonic, 4th and Higher) (Note 3) Second-Order Harmonic Distortion Third-Order Harmonic Distortion Two-Tone Intermodulation Distortion Two-Tone SFDR fIN = 70MHz, AIN = -2dBFS SFDR1 HD2 HD3 TTIMD TTSFDR MAX UNITS 96 80 84.3 fIN = 100MHz, AIN = -2dBFS 84 fIN = 130MHz, AIN = -2dBFS 82.5 fIN = 168MHz, AIN = -2dBFS 78 fIN = 10MHz, AIN = -2dBFS 102 fIN = 70MHz, AIN = -2dBFS SFDR2 TYP 90 100 fIN = 100MHz, AIN = -2dBFS 92 fIN = 130MHz, AIN = -2dBFS 94 fIN = 168MHz, AIN = -2dBFS 90 fIN = 10MHz, AIN = -2dBFS -100 fIN = 70MHz, AIN = -2dBFS -95 fIN = 100MHz, AIN = -2dBFS -94 fIN = 130MHz, AIN = -2dBFS -88.8 fIN = 168MHz, AIN = -2dBFS -78 fIN = 10MHz, AIN = -2dBFS -96 fIN = 70MHz, AIN = -2dBFS -84.3 fIN = 100MHz, AIN = -2dBFS -84 fIN = 130MHz, AIN = -2dBFS -82.5 fIN = 168MHz, AIN = -2dBFS -78 fIN1 = 65.1MHz, AIN = -8dBFS fIN2 = 70.1MHz, AIN = -8dBFS fIN1 = 65.1MHz, fIN2 = 70.1MHz -100dBFS < AIN < -10dBFS dBc dBc -84 dBc -80 dBc -85.2 dBc 99 dBFS CONVERSION RATE Maximum Conversion Rate fCLKMAX Minimum Conversion Rate fCLKMIN Aperture Jitter 80 MHz 20 tJ MHz 0.094 psRMS 1.0 to 5.0 VP-P CLOCK INPUTS (CLKP, CLKN) Differential Input Swing VDIFFCLK Fully differential inputs Common-Mode Voltage VCMCLK Self-biased 1.6 V Differential Input Resistance RINCLK 10 kΩ Differential Input Capacitance CINCLK 3 pF CMOS-COMPATIBLE DIGITAL OUTPUTS (D0–D15, DOR, DAV) Digital Output High Voltage VOH ISOURCE = 200µA Digital Output Low Voltage VOL ISINK = 200µA DVDD 0.2 V 0.2 V _______________________________________________________________________________________ 3 MAX19586 ELECTRICAL CHARACTERISTICS (continued) MAX19586 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor ELECTRICAL CHARACTERISTICS (continued) (AVDD = 3.3V, DVDD = 1.8V, AGND = DGND = 0, INP and INN driven differentially, internal reference CLKP and CLKN driven differentially, CL = 5pF at digital outputs, fCLK = 80MHz, TA = TMIN to TMAX, unless otherwise noted. Typical values are at TA = +25°C, unless otherwise noted.) (Note 1) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS TIMING SPECIFICATION (Figures 4, 5), CL = 5pF (D0–D15, DOR); CL = 15pF (DAV) CLKP - CLKN High tCLKP (Note 2) 5 CLKP - CLKN Low tCLKN (Note 2) 5 ns ns Effective Aperture Delay tAD -300 ps Output Data Delay tDAT 3.3 ns Data Valid Delay tDAV Pipeline Latency (Note 2) 2.8 tP 3.8 5.0 7 CLKP Rising Edge to DATA Not Valid tDNV (Note 2) CLKP Rising Edge to DATA Guaranteed Valid tDGV (Note 2) ns Clock Cycles 1.2 ns 6.5 ns DATA Setup Time Before Rising DAV tS Clock duty cycle = 50% (Note 2) 3 ns DATA Hold Time After Rising DAV tH Clock duty cycle = 50% (Note 2) 3 ns POWER SUPPLIES Analog Power-Supply Voltage AVDD 3.13 3.3 3.46 V Digital Output Power-Supply Voltage DVDD 1.7 1.8 1.9 V Analog Power-Supply Current IAVDD 320 382 mA Digital Output Power-Supply Current IDVDD 28 35 mA Power Dissipation PDISS 1105 1325 mW Note 1: ≥ +25°C guaranteed by production test, < +25°C guaranteed by design and characterization. Typical values are at TA = +25°C. Note 2: Parameter guaranteed by design and characterization. Note 3: AC parameter measured in a 32,768-point FFT record, where the first 2 bins of the FFT and 2 bins on either side of the carrier are excluded. 4 _______________________________________________________________________________________ High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor -80 -100 -120 5 10 15 20 25 30 35 40 -40 -60 -80 MAX19586 toc03 MAX19586 toc02 -100 2 2 -120 -140 0 5 10 15 20 25 30 35 40 0 5 10 15 20 25 30 35 40 ANALOG INPUT FREQUENCY (MHz) ANALOG INPUT FREQUENCY (MHz) ANALOG INPUT FREQUENCY (MHz) SNR/SINAD vs. ANALOG INPUT FREQUENCY (fCLK = 80MHz, AIN = -2dBFS) SFDR1/SFDR2 vs. ANALOG INPUT FREQUENCY (fCLK = 80MHz, AIN = -2dBFS) HD2/HD3 vs. ANALOG INPUT FREQUENCY (fCLK = 80MHz, AIN = -2dBFS) SINAD 74 72 70 100 -75 -80 SFDR2 HD2/HD3 (dBc) 78 95 90 85 20 40 60 -90 -95 80 -100 75 -105 80 100 120 140 160 180 HD2 -110 0 20 40 60 80 100 120 140 160 180 0 20 40 60 80 100 120 140 160 180 fIN (MHz) fIN (MHz) fIN (MHz) SNR vs. ANALOG INPUT AMPLITUDE (fCLK = 80MHz, fIN = 10.10011MHz) SFDR1 vs. ANALOG INPUT AMPLITUDE (fCLK = 80MHz, fIN = 10.10011MHz) SFDR2 vs. ANALOG INPUT AMPLITUDE (fCLK = 80MHz, fIN = 10.10011MHz) SFDR1 (dBc, dBFS) 70 65 60 SNR (dB) SFDR1 (dBFS) 110 100 90 SFDR1 (dBc) 80 50 SFDR = 90dB REFERENCE LINE 70 45 40 -30 -25 -20 -15 -10 -5 ANALOG INPUT AMPLITUDE (dBFS) 0 SFDR2 (dBFS) 110 100 SFDR2 (dBc) 90 80 SFDR = 90dB REFERENCE LINE 70 60 -35 120 MAX19586toc09 SNR (dBFS) 75 55 120 SFDR2 (dBc, dBFS) 80 MAX19586toc07 85 -40 HD3 -85 SFDR1 70 0 MAX19586toc06 105 SFDR1/SFDR2 (dBc) SNR -70 MAX19586toc05 80 76 110 MAX19586toc04 82 SNR/SINAD (dB) 3 -120 0 SNR (dB, dBFS) -80 -100 3 2 -60 fCLK = 80.00012288MHz fIN = 130.00050486MHz AIN = -1.82dBFS SNR = 77.7dB SINAD = 76.4dB SFDR1 = 83.1dBc SFDR2 = 91.2dBc HD2 = -89.4dBc HD3 = -83.1dBc 3 -20 AMPLITUDE (dBFS) -60 -40 0 MAX19586toc08 AMPLITUDE (dBFS) -40 fCLK = 80.00012288MHz fIN = 70.16368199MHz AIN = -2.06dBFS SNR = 79.3dB SINAD = 77.7dB SFDR1 = 83.3dBc SFDR2 = 98.2dBc HD2 = -93.5dBc HD3 = -83.3dBc -20 AMPLITUDE (dBFS) fCLK = 80.00012288MHz fIN = 10.10011317MHz AIN = -2.02dBFS SNR = 80dB SINAD = 79.8dB SFDR1 = 96.2dBc SFDR2 = 101dBc HD2 = -99.6dBc HD3 = -96.2dBc -20 0 MAX19586 toc01 0 60 -40 -35 -30 -25 -20 -15 -10 -5 ANALOG INPUT AMPLITUDE (dBFS) 0 -40 -35 -30 -25 -20 -15 -10 -5 0 ANALOG INPUT AMPLITUDE (dBFS) _______________________________________________________________________________________ 5 MAX19586 Typical Operating Characteristics (AVDD = 3.3V, DVDD = 1.8V, INP and INN driven differentially, internal reference, CLKP and CLKN driven differentially, CL = 5pF at digital outputs, fCLK = 80MHz, TA = +25°C. Unless otherwise noted, all AC data based on 32k-point FFT records and under coherent sampling conditions.) FFT PLOT FFT PLOT FFT PLOT (32,768-POINT RECORD) (32,768-POINT RECORD) (261,244-POINT DATA RECORD) Typical Operating Characteristics (continued) (AVDD = 3.3V, DVDD = 1.8V, INP and INN driven differentially, internal reference, CLKP and CLKN driven differentially, CL = 5pF at digital outputs, fCLK = 80MHz, TA = +25°C. Unless otherwise noted, all AC data based on 32k-point FFT records and under coherent sampling conditions.) SNR vs. ANALOG INPUT AMPLITUDE SFDR1 vs. ANALOG INPUT AMPLITUDE SFDR2 vs. ANALOG INPUT AMPLITUDE (fCLK = 80MHz, fIN = 70.163683MHz) (fCLK = 80MHz, fIN = 70.163683MHz) (fCLK = 80MHz, fIN = 70.163683MHz) SFDR1 (dBc, dBFS) SNR (dB) 65 60 55 45 SFDR1 (dBc) 80 70 SFDR = 90dB REFERENCE LINE 35 -35 -30 -25 -20 -15 -10 -5 0 MAX19586toc12 SFDR2 (dBc) 80 70 SFDR = 90dB REFERENCE LINE 50 40 -40 90 60 50 40 40 -40 -35 -30 -25 -20 -15 -10 -5 -40 0 -35 -30 -25 -20 -15 -10 -5 ANALOG INPUT AMPLITUDE (dBFS) ANALOG INPUT AMPLITUDE (dBFS) ANALOG INPUT AMPLITUDE (dBFS) SNR/SINAD vs. SAMPLING FREQUENCY (fIN = 9.9757395MHz, AIN = -2dBFS) SFDR/SFDR2 vs. SAMPLING FREQUENCY (fIN = 10.10011MHz, AIN = -2dBFS) HD2/HD3 vs. SAMPLING FREQUENCY (fIN = 10.10011MHz, AIN = -2dBFS) 79 110 SFDR1/SFDR2 (dBc) SNR SINAD 77 -70 SFDR2 105 100 95 SFDR1 90 -80 HD2/HD3 (dBc) 81 115 85 75 0 MAX19586toc15 120 MAX19586toc13 83 SNR/SINAD (dB) 100 90 60 50 SFDR2 (dBFS) 110 MAX19586toc14 SNR (dB, dBFS) 75 70 SFDR1 (dBFS) 100 SFDR2 (dBc, dBFS) 80 120 MAX19586toc11 SNR (dBFS) 85 110 MAX19586toc10 90 -90 HD2 -100 HD3 -110 80 73 75 20 30 40 50 60 70 80 30 40 50 60 70 80 20 90 100 110 30 40 50 60 70 80 90 100 110 fCLK (MHz) fCLK (MHz) SNR/SINAD vs. SAMPLING FREQUENCY (fIN = 70.163683MHz, AIN = -2dBFS) SFDR1/SFDR2 vs. SAMPLING FREQUENCY (fIN = 70.163683MHz, AIN = -2dBFS) HD2/HD3 vs. SAMPLING FREQUENCY (fIN = 70.163683MHz, AIN = -2dBFS) 76 SINAD HD3 -80 95 90 85 MAX19586toc18 -75 HD2/HD3 (dBc) 78 SFDR2 -70 MAX19586toc17 100 SFDR/SFDR2 (dBc) SNR 105 MAX19586toc16 80 SFDR1 -85 -90 HD2 -95 -100 74 80 72 -105 75 20 30 40 50 60 70 fCLK (MHz) 6 -120 20 90 100 110 fCLK (MHz) 82 SNR/SINAD (dB) MAX19586 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor 80 90 100 110 -110 20 30 40 50 60 70 fCLK (MHz) 80 90 100 110 20 30 40 50 60 70 fCLK (MHz) _______________________________________________________________________________________ 80 90 100 110 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor 115 -85 110 SINAD 79 78 -90 SFDR2 105 HD2/HD3 (dBc) SFDR1/SFDR2 (dBc) SNR 80 100 SFDR1 95 80 -40 -15 10 35 60 HD2 -115 -40 -15 10 35 60 -40 85 -15 10 35 60 85 TEMPERATURE (°C) TEMPERATURE (°C) TEMPERATURE (°C) SNR/SINAD vs. TEMPERATURE (fCLK = 80MHz, fIN = 70.163683MHz, AIN = -2dBFS) SFDR1/SFDR2 vs. TEMPERATURE (fCLK = 80MHz, fIN = 70.163683MHz, AIN = -2dBFS) HD2/HD3 vs. TEMPERATURE (fCLK = 80MHz, fIN = 70.163683MHz, AIN = -2dBFS) SNR 110 -75 SINAD 77 HD2/HD3 (dBc) SFDR1/SFDR2 (dBc) 78 100 90 SFDR1 HD3 -80 SFDR2 79 MAX19586toc24 80 -70 MAX19586toc23 120 MAX19586toc22 81 SNR/SINAD (dB) 85 HD3 -100 -110 85 76 -95 -105 90 77 MAX19586toc21 -80 MAX19586toc20 81 SNR/SINAD (dB) 120 MAX19586toc19 82 80 -85 -90 HD2 -95 -100 -105 70 75 60 10 35 60 -115 -40 -15 10 35 60 85 -40 -15 10 60 TEMPERATURE (°C) TEMPERATURE (°C) POWER DISSIPATION vs. TEMPERTURE REFERENCE VOLTAGE vs. TEMPERTURE POWER DISSIPATION vs. ANALOG SUPPLY VOLTAGE 1080 1040 fCLK = 80.00012288MHz fIN = 70.163683MHz AIN = -2dBFS 1.295 1300 MAX19586toc26 1.300 1.290 1200 1100 IAVCC, PDISS (mA, mW) 1120 1.285 1.280 1.275 1.270 fCLK = 80.00012288MHz fIN = 70.163683MHz AIN = -2dBFS 1000 10 35 TEMPERATURE (°C) 60 85 PDISS 900 800 700 600 500 IAVCC 300 200 1.260 -15 85 1000 400 1.265 -40 35 TEMPERATURE (°C) fCLK = 80.00012288MHz fIN = 70.163683MHz AIN = -2dBFS 1160 85 REFERENCE VOLTAGE (V) POWER DISSIPATION (mW) 1200 -15 MAX19586toc25 -40 -110 MAX19586toc27 76 -40 -15 10 35 TEMPERATURE (°C) 60 85 3.15 3.20 3.25 3.30 3.35 3.40 3.45 ANALOG SUPPLY VOLTAGE (V) _______________________________________________________________________________________ 7 MAX19586 Typical Operating Characteristics (continued) (AVDD = 3.3V, DVDD = 1.8V, INP and INN driven differentially, internal reference, CLKP and CLKN driven differentially, CL = 5pF at digital outputs, fCLK = 80MHz, TA = +25°C. Unless otherwise noted, all AC data based on 32k-point FFT records and under coherent sampling conditions.) SNR/SINAD vs. TEMPERATURE SFDR1/SFDR2 vs. TEMPERATURE HD2/HD3 vs. TEMPERATURE (fCLK = 80MHz, fIN = 10.10011MHz, (fCLK = 80MHz, fIN = 10.10011MHz, (fCLK = 80MHz, fIN = 10.10011MHz, AIN = -2dBFS) AIN = -2dBFS) AIN = -2dBFS) Typical Operating Characteristics (continued) (AVDD = 3.3V, DVDD = 1.8V, INP and INN driven differentially, internal reference, CLKP and CLKN driven differentially, CL = 5pF at digital outputs, fCLK = 80MHz, TA = +25°C. Unless otherwise noted, all AC data based on 32k-point FFT records and under coherent sampling conditions.) SFDR1/SFDR2 vs. ANALOG SUPPLY VOLTAGE 80 SNR/SINAD (dB) 1.283 1.280 1.278 78 77 SINAD 76 95 90 85 75 80 1.273 74 75 73 3.25 3.30 3.35 3.40 3.45 fCLK = 80.00012288MHz fIN = 70.163683MHz AIN = -2dBFS 3.20 3.25 3.30 3.35 3.40 3.45 3.15 3.20 3.25 3.30 3.35 3.40 ANALOG SUPPLY VOLTAGE (V) ANALOG SUPPLY VOLTAGE (V) ANALOG SUPPLY VOLTAGE (V) HD2/HD3 vs. ANALOG SUPPLY VOLTAGE TWO-TONE SFDR PLOT (32,768-POINT DATA RECORD) TWO-TONE SFDR PLOT (32,768-POINT DATA RECORD) 0 -20 AMPLITUDE (dBFS) -80 HD3 -85 -90 HD2 fIN1 -40 fIN2 fCLK = 80MHz fIN1 = 10.1001MHz fIN2 = 14.871MHz AIN1 = -8.04dBFS AIN2 = -8.00dBFS TTSFDR = 99.6dBFS 0 -20 AMPLITUDE (dBFS) fCLK = 80.00012288MHz fIN = 70.163683MHz AIN = -2dBFS -95 SFDR2 70 3.15 MAX19586 toc32 -60 -80 fIN2 fIN1 -40 3.45 fCLK = 80MHz fIN1 = 65.1002MHz fIN2 = 70.1MHz AIN1 = -8.03dBFS AIN2 = -8.00dBFS TTSFDR = 93.2dBFS -60 2fIN1 - fIN2 -80 fIN1 + fIN2 -100 -100 -105 -110 -100 -120 3.15 3.20 3.25 3.30 3.35 3.40 3.45 -120 0 ANALOG SUPPLY VOLTAGE (V) 5 10 15 20 25 30 35 MAX19586toc34 SFDR (dBFS) SFDR (dBc) SFDR = 90dB REFERENCE LINE -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 0 ANALOG INPUT AMPLITUDE (dBFS) 5 10 15 20 25 120 110 100 90 80 70 60 50 40 30 20 10 0 30 35 ANALOG INPUT FREQUENCY (MHz) TWO-TONE SFDR vs. ANALOG INPUT AMPLITUDE (fCLK = 80MHz, fIN1 = 65.1MHz, fIN2 = 70.1MHz) TTSFDR (dBc, dBFS) TTSFDR (dBc, dBFS) 0 ANALOG INPUT FREQUENCY (MHz) TWO-TONE SFDR vs. ANALOG INPUT AMPLITUDE (fCLK = 80MHz, fIN1 = 10.1MHz, fIN2 = 14.87MHz) 120 110 100 90 80 70 60 50 40 30 20 10 0 40 SFDR (dBFS) SFDR (dBc) SFDR = 90dB REFERENCE LINE -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 0 ANALOG INPUT AMPLITUDE (dBFS) _______________________________________________________________________________________ MAX19586toc35 -70 -75 3.20 MAX19586to31 3.15 SFDR1 100 1.275 1.270 8 105 SFDR1/SFDR2 (dBc) 79 1.285 110 MAX19586toc30 fCLK = 80.00012288MHz fIN = 70.163683MHz SNR AIN = -2dBFS MAX19586 toc33 fCLK = 80.00012288MHz fIN = 70.163683MHz AIN = -2dBFS REFERENCE VOLTAGE (V) 1.288 81 MAX19586toc28 1.290 SNR/SINAD vs. ANALOG SUPPLY VOLTAGE MAX19586toc29 REFERENCE VOLTAGE vs. ANALOG SUPPLY VOLTAGE HD2/HD3 (dBc) MAX19586 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor 40 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor PIN NAME FUNCTION 1, 2, 17, 18, 19, 23, 24, 25, 55, 56 AVDD Analog Supply Voltage. Provide local bypassing to ground with 0.01µF and 0.1µF capacitors. 3, 6–9, 12, 13, 14, 20, 21, 22, 28 AGND Converter Ground. Analog, digital, and output-driver grounds are internally connected to the same potential. Connect the converter’s exposed paddle (EP) to GND. 4 CLKP Differential Clock, Positive Input Terminal 5 CLKN Differential Clock, Negative Input Terminal 10 INP Differential Analog Input, Positive Terminal 11 INN Differential Analog Input, Negative/Complementary Terminal 15, 16, 54 N.C. No Connection. Do not connect to this pin. 26 REFOUT 27 REFIN Reference Voltage Input 29, 41, 42, 51 DVDD Digital Supply Voltage. Provide local bypassing to ground with 0.01µF and 0.1µF capacitors. 30, 31, 52 DGND 32 D0 Digital CMOS Output Bit 0 (LSB) 33 D1 Digital CMOS Output Bit 1 34 D2 Digital CMOS Output Bit 2 35 D3 Digital CMOS Output Bit 3 36 D4 Digital CMOS Output Bit 4 37 D5 Digital CMOS Output Bit 5 38 D6 Digital CMOS Output Bit 6 39 D7 Digital CMOS Output Bit 7 40 D8 Digital CMOS Output Bit 8 43 D9 Digital CMOS Output Bit 9 44 D10 Digital CMOS Output Bit 10 45 D11 Digital CMOS Output Bit 11 46 D12 Digital CMOS Output Bit 12 47 D13 Digital CMOS Output Bit 13 48 D14 Digital CMOS Output Bit 14 49 D15 Digital CMOS Output Bit 15 (MSB) 50 DAV Data Valid Output. This output can be used as a clock control line to drive an external buffer or dataacquisition system. The typical delay time between the falling edge of the converter clock and the rising edge of DAV is 3.8ns. 53 DOR Data Over-Range Bit. This control line flags an over-/under-range condition in the ADC. If DOR transitions high, an over-/under-range condition was detected. If DOR remains low, the ADC operates within the allowable full-scale range. — EP Internal Bandgap Reference Output Converter Ground. Digital output-driver ground. Exposed Paddle. Must be connected to AGND. _______________________________________________________________________________________ 9 MAX19586 Pin Description MAX19586 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor ential signal inputs to the MAX19586 should be AC-coupled and carefully balanced to achieve the best dynamic performance (see Differential, AC-Coupled Analog Inputs in the Applications Information section for more details). AC-coupling of the input signal is required because the MAX19586 inputs are self-biasing as shown in Figure 2. Although the track-and-hold inputs are high impedance, the actual differential input impedance is nominally 10kΩ because of the two 5kΩ resistors connected to the common-mode bias circuitry. Avoid injecting any DC leakage currents into these analog inputs. Exceeding a DC leakage current of 10µA shifts the self-biased common-mode level, adversely affecting the converter’s performance. Detailed Description Figure 1 provides an overview of the MAX19586 architecture. The MAX19586 employs an input track-andhold (T/H) amplifier, which has been optimized for low thermal noise and low distortion. The high-impedance differential inputs to the T/H amplifier (INP and INN) are self-biased at 2.2V, and support a full-scale 2.56VP-P differential input voltage. The output of the T/H amplifier is applied to a multistage pipelined ADC core, which is designed to achieve a very low thermal noise floor and low distortion. A clock buffer receives a differential input clock waveform and generates a low-jitter clock signal for the input T/H. The signal at the analog inputs is sampled at the rising edge of the differential clock waveform. The differential clock inputs (CLKP and CLKN) are highimpedance inputs, are self-biased at 1.6V, and support differential clock waveforms from 1VP-P to 5VP-P. The outputs from the multistage pipelined ADC core are delivered to error correction and formatting logic, which deliver the 16-bit output code in two’s-complement format to digital output drivers. The output drivers provide 1.8V CMOS-compatible outputs. On-Chip Reference Circuit The MAX19586 incorporates an on-chip 1.28V, low-drift bandgap reference. This reference potential establishes the full-scale range for the converter, which is nominally 2.56V P-P differential (Figure 3). The internal reference voltage can be monitored by REFOUT. To use the internal reference voltage the reference input (REFIN) must be connected to REFOUT through a 10kΩ resistor. Bypass both pins with separate 1µF capacitors to AGND. The MAX19586 also allows an external reference source to be connected to REFIN, enabling the user to overdrive the internal bandgap reference. REFIN accepts a 1.28V ±10% input voltage range. Analog Inputs (INP, INN) The signal inputs to the MAX19586 (INP and INN) are balanced differential inputs. This differential configuration provides immunity to common-mode noise coupling and rejection of even-order harmonic terms. The differ- CLKP CLKN AVDD CLOCK BUFFER AGND DVDD DAV CMOS DRIVER INP T/H INN PIPELINE ADC CMOS OUTPUT DRIVERS DOR D0–D15 DGND MAX19586 REFERENCE REFOUT REFIN Figure 1. Block Diagram 10 ______________________________________________________________________________________ High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor System Timing Requirements Figure 4 depicts the general timing relationships for the signal input, clock input, data output, and DAV output. Figure 5 shows the detailed timing specifications and signal relationships, as defined in the Electrical Characteristics table. The MAX19586 samples the input signal on the rising edge of the input clock. Output data is valid on the rising edge of the DAV signal, with a 7 clock-cycle data latency. Note that the clock duty cycle should typically be 50% ±10% for proper operation. T/H AMPLIFIER TO FIRST QUANTIZER STAGE INP 5kΩ OTA 5kΩ Digital Outputs (D0–D15, DAV, DOR) Although designed for low-voltage 1.8V logic systems, the logic-high level of the low-voltage CMOS-compatible digital outputs (D0–D15, DAV, and DOR) offer some flexibility, as it allows the user to select the digital voltage within the 1.7V to 1.9V range. For best performance, the capacitive loading on the digital outputs of the MAX19586 should be kept as low as possible (< 10pF). Due to the current-limited dataoutput driver of the MAX19586, large capacitive loads increase the rise and fall time of the data and can make it more difficult to register the data into the next IC. The loading capacitance can be kept low by keeping the output traces short and by driving a single CMOS buffer or latch input (as opposed to multiple CMOS inputs). The output data is in two’s-complement format, as illustrated in Table 1. Data is valid at the rising edge of DAV (Figures 4, 5). DAV may be used as a clock signal to latch the output data. Note that the DAV output driver is not current limited, hence it allows for higher capacitive loading. TO FIRST QUANTIZER STAGE INN T/H AMPLIFIER Figure 2. Simplified Analog Input Architecture +640mV 2.56VP-P DIFFERENTIAL FSR INP INN COMMON-MODE VOLTAGE (2.2V) -640mV Figure 3. Full-Scale Voltage Range ______________________________________________________________________________________ 11 MAX19586 Clock Inputs (CLKP, CLKN) The differential clock buffer for the MAX19586 has been designed to accept an AC-coupled clock waveform. Like the signal inputs, the clock inputs are self-biasing. In this case, the self-biased potential is 1.6V and each input is connected to the reference potential with a 5kΩ resistor. Consequently, the differential input resistance associated with the clock inputs is 10kΩ. While differential clock signals as low as 0.5VP-P can be used to drive the clock inputs, best dynamic performance is achieved with 1VP-P to 5VP-P clock input voltage levels. Jitter on the clock signal translates directly to jitter (noise) on the sampled signal. Therefore, the clock source must be a very low-jitter (low-phase-noise) source. Additionally, extremely low phase-noise oscillators and bandpass filters should be used to obtain the true AC performance of this converter. See the Differential, AC-Coupled Clock Inputs and Testing the MAX19586 topics in the Applications Information section for additional details on the subject of driving the clock inputs. MAX19586 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor 7 CLOCK-CYCLE LATENCY N+1 N+2 N N+3 N+7 N+4 ANALOG INPUT N+6 N+5 CLOCK INPUT D0–D15 N-5 N-6 N-7 N-4 N-3 N-2 N-1 DAV Figure 4. General System and Output Timing Diagram INP INN tCLKP tA tCLKN CLKN N N+1 N+2 N+3 CLKP tDAT tDNV tDGV D0–D15 N-7 N-6 N-5 N-4 DOR tS tDAV tH DAV ENCODE AT CLKP - CLKN > 0 (RISING EDGE) tCLKP CLKP - CLKN > 0 tCLKN CLKP - CLKN < 0 EFFECTIVE APERTURE DELAY tAD tDAT DELAY FROM CLKP TO OUTPUT DATA TRANSITION tDAV tDNV tDGV tS tH DELAY FROM CLKN TO DATA VALID CLOCK DAV CLKP RISING EDGE TO DATA NOT VALID CLKP RISING EDGE TO DATA GUARANTEED VALID DATA SETUP TIME BEFORE RISING DAV DATA HOLD TIME AFTER RISING DAV Figure 5. Detailed Timing Information for Clock Operation 12 ______________________________________________________________________________________ N High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor MAX19586 Table 1. MAX19586 Digital Output Coding INP ANALOG VOLTAGE LEVEL INN ANALOG VOLTAGE LEVEL D15–D0 TWO’S-COMPLEMENT CODE VCM + 0.64V VCM - 0.64V 0111111111111111 (positive full-scale) VCM VCM 0000000000000000 (midscale + δ) 1111111111111111 (midscale - δ) VCM - 0.64V VCM + 0.64V 1000000000000000 (negative full-scale) The converter’s DOR output signal is used to identify over- and under-range conditions. If the input signal exceeds the positive or negative full-scale range for the MAX19586 then DOR will be asserted high. The timing for DOR is identical to the timing for the data outputs, and DOR therefore provides an over-range indication on a sample-by-sample basis. Applications Information Differential, AC-Coupled Clock Inputs The clock inputs to the MAX19586 are driven with an AC-coupled differential signal, and best performance is achieved under these conditions. However, it is often the case that the available clock source is single-ended. Figure 6 demonstrates one method for converting a single-ended clock signal into a differential signal with a transformer. In this example, the transformer turns ratio from the primary to secondary side is 1:1.414. The impedance ratio from primary to secondary is the square of the turns ratio, or 1:2. So terminating the secondary side with a 100Ω differential resistance results in a 50Ω load looking into the primary side of the transformer. The termination resistor in this example is comprised of the series combination of two 50Ω resistors with their common node AC-coupled to ground. Figure 6 illustrates the secondary side of the transformer to be coupled directly to the clock inputs. Since the clock inputs are self-biasing, the center tap of the transformer must be AC-coupled to ground or left floating. If the center tap of the transformer’s secondary side is DC-coupled to ground, it is necessary to add blocking capacitors in series with the clock inputs. Clock jitter is generally improved if the clock signal has a high slew rate at the time of its zero-crossing. Therefore, if a sinusoidal source is used to drive the clock inputs the clock amplitude should be as large as possible to maximize the zero-crossing slew rate. The back-to-back Schottky diodes shown in Figure 6 are not required as long as the input signal is held to a differential voltage potential of 3VP-P or less. If a larger amplitude signal is provided (to maximize the zero-crossing slew rate), then the diodes serve to limit the differential signal swing at the clock inputs. Note that all AC specifications for the MAX19586 are measured within this configuration and with an input clock amplitude of approximately 12dBm. Any differential mode noise coupled to the clock inputs translates to clock jitter and degrades the SNR performance of the MAX19586. Any differential mode coupling of the analog input signal into the clock inputs results in harmonic distortion. Consequently, it is important that the clock lines be well isolated from the analog signal input and from the digital outputs. See the Signal Routing section for more discussion on the subject of noise coupling. Differential, AC-Coupled Analog Inputs The analog inputs INP and INN are driven with a differential AC-coupled signal. It is important that these inputs be accurately balanced. Any common-mode signal applied to these inputs degrades even-order distortion terms. Therefore, any attempt at driving these inputs in a single-ended fashion will result in significant even-order distortion terms. Figure 7 presents one method for converting a singleended signal to a balanced differential signal using a transformer. The primary-to-secondary turns ratio in this example is 1:1.414. The impedance ratio is the square of the turns ratio, so in this example the impedance ratio is 1:2. To achieve a 50Ω input impedance at the primary side of the transformer, the secondary side is terminated with a 100Ω differential load. This load, in shunt with the differential input resistance of the MAX19586, results in a 100Ω differential load on the secondary side. It is rea- ______________________________________________________________________________________ 13 MAX19586 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor sonable to use a larger transformer turns ratio to achieve a larger signal step-up, and this may be desirable to relax the drive requirements for the circuitry driving the MAX19586. However, the larger the turns ratio, the larger the effect of the differential input impedance of the MAX19586 on the primary-referred input impedance. As stated previously, the signal inputs to the MAX19586 must be accurately balanced to achieve the best evenorder distortion performance. AVDD DVDD INP D0–D15 MAX19586 16 INN 0.1µF BACK-TO-BACK DIODE CLKP CLKN AGND DGND T2-1T-KK81 49.9Ω 49.9Ω 0.1µF Figure 6. Transformer-Coupled Clock Input Configuration AVDD DVDD POSITIVE TERMINAL 0.1µF ADT2-1T T1-1T-KK81 INP 49.9Ω D0–D15 MAX19586 16 49.9Ω INN 0.1µF CLKP CLKN AGND DGND Figure 7. Transformer-Coupled Analog Input Configuration with Primary-Side Balun Transformer 14 ______________________________________________________________________________________ High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor Testing the MAX19586 The MAX19586 has a very low thermal noise floor (-82dBFS) and very low jitter (< 100fs). As a consequence, test system limitations can easily obscure the SIGNAL PATH AGILENT 8644B BANDPASS FILTER 10dB 3dB PAD BOTH SIGNAL GENERATORS ARE PHASE-LOCKED MAX19586 CLOCK PATH AGILENT 8644B BANDPASS FILTER Figure 8a. Standard High-Speed ADC Test Setup (Simplified Block Diagram) performance of the ADC. Figure 8a is a block diagram of a conventional high-speed ADC test system. The input signal and the clock source are generated by lowphase-noise synthesizers (e.g., Agilent 8644B). Bandpass filters in both the signal and the clock paths then attenuate noise and harmonic components. Figure 8b shows the resulting power spectrum, which results from this setup for a 70MHz input tone and an 80Msps clock. Note the substantial lift in the noise floor near the carrier. The bandwidth of this particular noisefloor lift near the carrier corresponds to the bandwidth of the filter in the input signal path. Figure 8c illustrates the impact on the spectrum if the input frequency is shifted away from the center frequency of the input signal filter. Note that the fundamental tone has moved, but the noise-floor lift remains in the same location. This is evidence of the validity of the claim that the lift in the noise floor is due to the test system and not the ADC. In this figure, the magnitude of the lift in the noise floor increased relative to the previous figure because the signal is located on the skirt of the filter and the signal amplitude had to be increased to obtain a signal near full scale. To truly reveal the performance of the MAX19586, the test system performance must be improved substantially. Figure 8d depicts such an improved test system. In this system, the synthesizers provide reference inputs to two dedicated low-noise phase-locked loops (PLLs), one centered at approximately 80MHz (for the clock path) and the other centered at 70MHz (for the signal path). The oscillators in these PLLs are very low-noise oscillators, and the FFT PLOT (32,768-POINT DATA RECORD) FFT PLOT (32,768-POINT DATA RECORD) 0 0 fCLK = 80.00012288MHz fIN = 70.163683MHz AIN = -2dBFS -20 -40 POWER (dBFS) POWER (dBFS) -20 fCLK = 80.00012288MHz fIN = 68MHz AIN = -2dBFS -60 CARRIER WAS INTENTIONALLY LOWERED BY 2MHz TO SHOW THE STATIONARY BEHAVIOR OF THE NOISE -40 -60 -80 -80 3 3 2 2 -100 -100 -120 -120 0 5 10 15 20 25 30 35 40 ANALOG INPUT FREQUENCY (MHz) Figure 8b. 70MHz FFT with Standard High-Speed ADC Test Setup 0 5 10 15 20 25 30 35 40 FREQUENCY (MHz) Figure 8c. 68MHz FFT with Standard High-Speed ADC Test Setup ______________________________________________________________________________________ 15 MAX19586 One note of caution in relation to transformers is important. Any DC current passed through the primary or secondary windings of a transformer may magnetically bias the transformer core. When this happens the transformer is no longer accurately balanced and a degradation in the distortion of the MAX19586 may be observed. The core must be demagnetized to return to balanced operation. MAX19586 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor AGILENT 8644B SIGNAL PATH VARIABLE BANDPASS ATTENUATOR FILTER REF TUNE PLL SIGNAL 10dB 3dB PAD VCXO MAX19586 LOW-NOISE PLL AGILENT 8644B BOTH SIGNAL GENERATORS ARE PHASE-LOCKED CLOCK PATH BANDPASS FILTER REF TUNE PLL SIGNAL 10dB VCXO LOW-NOISE PLL Figure 8d. Improved Test System Employing Narrowband PLLs (Simplified Block Diagram) FFT PLOT (32,768-POINT DATA RECORD) 110 0 fCLK = 80.00012288MHz fIN = 70.163683MHz AIN = -2dBFS -20 105 INPUT FREQUENCY = 70MHz 100 95 -40 90 SNR (dB) ANALOG POWER (dBFS) SNR vs. RMS JITTER PERFORMANCE -60 -80 80 75 3 2 -100 85 70 INPUT FREQUENCY = 140MHz 65 60 -120 0 5 10 15 20 25 30 35 40 ANALOG INPUT FREQUENCY (MHz) Figure 8e. 70MHz FFT with Improved High-Speed ADC Test Setup PLLs act as extremely narrow bandwidth filters (on the order of 20Hz) to attenuate the noise of the synthesizers. The system provides a total system jitter on the order of 20fs. Note that while the low-noise oscillators could be used by themselves without being locked to their respective signal sources, this would result in FFTs that are not coherent and which would require windowing. 16 10 100 1000 RMS JITTER (fs) Figure 8f. SNR vs. System Jitter Performance Graph Figure 8e is an FFT plot of the spectrum obtained when the improved test system is employed. The noise-floor lift in the vicinity of the carrier is now almost completely eliminated. The SNR associated with this FFT is about 79.1dB, whereas the SNR obtained using the standard test system is on the order of 77.6dB. ______________________________________________________________________________________ High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor Layer Assignments The MAX19586 EV kit is a 6-layer board, and the assignment of layers is discussed in this context. It is recommended that the ground plane be on a layer between the signal routing layer and the supply routing layer(s). This prevents coupling from the supply lines into the signal lines. The MAX19586 EV kit PC board places the signal lines on the top (component) layer and the ground plane on layer 2. Any region on the top layer not devoted to signal routing is filled with the ground plane with vias to layer 2. Layers 3 and 4 are devoted to supply routing, layer 5 is another ground plane, and layer 6 is used for the placement of additional components and for additional signal routing. A four-layer implementation is also feasible using layer 1 for signal lines, layer 2 as a ground plane, layer 3 for supply routing, and layer 4 for additional signal routing. However, care must be taken to ensure that the clock and signal lines are isolated from each other and from the supply lines. Signal Routing To preserve good even-order distortion, the signal lines (those traces feeding the INP and INN inputs) must be carefully balanced. To accomplish this, the signal traces should be made as symmetric as possible, meaning that each of the two signal traces should be the same length and should see the same parasitic environment. As mentioned previously, the signal lines must be isolated from the supply lines to prevent coupling from the supplies to the inputs. This is accomplished by making the necessary layer assignments as described in the previous section. Additionally, it is crucial that the clock lines be isolated from the signal lines. On the MAX19586 EV kit this is done by routing the clock lines on the bottom layer (layer 6). The clock lines then connect to the ADC through vias placed in close proximity to the device. The clock lines are isolated from the supply lines as well by virtue of the ground plane on layer 5. As with all high-speed designs, digital output traces should be kept as short as possible to minimize capacitive loading. The ground plane on layer 2 beneath these traces should not be removed so that the digital ground return currents have an uninterrupted path back to the bypass capacitors. Grounding The practice of providing a split ground plane in an attempt to confine digital ground-return currents has often been recommended in ADC application literature. However, for converters such as the MAX19586 it is strongly recommended to employ a single, uninterrupted ground plane. The MAX19586 EV kit achieves excellent dynamic performance with such a ground plane. The exposed paddle of the MAX19586 should be soldered directly to a ground pad on layer 1 with vias to the ground plane on layer 2. This provides excellent electrical and thermal connections to the PC board. Supply Bypassing The MAX19586 EV kit uses 220µF capacitors (and smaller values such as 47µF and 2µF) on power-supply lines AV DD and DV DD to provide low-frequency bypassing. The loss (series resistance) associated with these capacitors is beneficial in eliminating high-Q supply resonances. Ferrite beads are also used on each of the power-supply lines to enhance supply bypassing (Figure 9). Combinations of small value (0.01µF and 0.1µF), lowinductance surface-mount capacitors should be placed at each supply pin or each grouping of supply pins to attenuate high-frequency supply noise. Place these capacitors on the top side of the board and as close to the converter as possible with short connections to the ground plane. Parameter Definitions Offset Error Offset error is a figure of merit that indicates how well the actual transfer function matches the ideal transfer function at a single point. Ideally, the midscale MAX19586 transition occurs at 0.5 LSB above midscale. The offset error is the amount of deviation between the measured midscale transition point and the ideal midscale transition point. ______________________________________________________________________________________ 17 MAX19586 Figure 8f demonstrates the impact of test system jitter on measured SNR. The figure plots SNR due to test system jitter only, neglecting all other sources of noise, for two different input frequencies. For example, note that for a 70MHz input frequency a test system jitter number of 100fs results in an SNR (due to the test system alone) of about 87.1dB. In the case of the MAX19586, which has a -82dBFS noise floor, this is not an inconsequential amount of additional noise. In conclusion, careful attention must be paid to both the input signal source and the clock signal source, if the true performance of the MAX19586 is to be properly characterized. Dedicated PLLs with low-noise VCOs, such as those used in Figure 8d, are capable of providing signals with the required low jitter performance. MAX19586 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor BYPASSING—ADC LEVEL BYPASSING—BOARD LEVEL DVDD AVDD 0.01µF AVDD 0.1µF 0.1µF FERRITE BEAD 0.01µF 47µF 2µF 220µF ANALOG POWERSUPPLY SOURCE 220µF DIGITAL POWERSUPPLY SOURCE AGND DGND D0–D15 DVDD FERRITE BEAD MAX19586 16 2µF AGND 47µF DGND Figure 9. Grounding, Bypassing, and Decoupling Recommendations for the MAX19586 Gain Error Signal-to-Noise Ratio (SNR) Gain error is a figure of merit that indicates how well the slope of the actual transfer function matches the slope of the ideal transfer function. The slope of the actual transfer function is measured between two data points: positive full scale and negative full scale. Ideally, the positive full-scale MAX19586 transition occurs at 1.5 LSBs below positive full scale, and the negative fullscale transition occurs at 0.5 LSB above negative full scale. The gain error is the difference of the measured transition points minus the difference of the ideal transition points. For a waveform perfectly reconstructed from digital samples, the theoretical maximum SNR is the ratio of the full-scale analog input (RMS value) to the RMS quantization error (residual error). The ideal, theoretical minimum analog-to-digital noise is caused by quantization error only and results directly from the ADC’s resolution (N bits): SNR[max] = 6.02 x N + 1.76 In reality, there are other noise sources besides quantization noise: thermal noise, reference noise, clock jitter, etc. SNR is computed by taking the ratio of the RMS signal to the RMS noise. RMS noise includes all spectral components to the Nyquist frequency excluding the fundamental, the first four harmonics (HD2 through HD5), and the DC offset. SNR = 20 x log (SIGNALRMS / NOISERMS) Small-Signal Noise Floor (SSNF) Small-signal noise floor is the integrated noise and distortion power in the Nyquist band for small-signal inputs. The DC offset is excluded from this noise calculation. For this converter, a small signal is defined as a single tone with an amplitude of less than -35dBFS. This parameter captures the thermal and quantization noise characteristics of the data converter and can be used to help calculate the overall noise figure of a digital receiver signal path. 18 Signal-to-Noise Plus Distortion (SINAD) SINAD is computed by taking the ratio of the RMS signal to the RMS noise plus distortion. RMS noise plus distortion includes all spectral components to the Nyquist frequency excluding the fundamental and the DC offset. ______________________________________________________________________________________ High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor SFDR is the ratio expressed in decibels of the RMS amplitude of the fundamental (maximum signal component) to the RMS value of the next largest spurious component, excluding DC offset. SFDR1 reflects the MAX19586 spurious performance based on worst 2ndor 3rd-order harmonic distortion. SFDR2 is defined by the worst spurious component excluding 2nd- and 3rdorder harmonic spurs and DC offset. Two-Tone Spurious-Free Dynamic Range (TTSFDR) Two-tone SFDR is the ratio of the full scale of the converter to the RMS value of the peak spurious component. The peak spurious component can be related to the intermodulation distortion components, but does not have to be. Two-tone SFDR for the MAX19586 is expressed in dBFS. Two-Tone Intermodulation Distortion (TTIMD) IMD is the total power of the IM2 to IM5 intermodulation products to the Nyquist frequency relative to the total input power of the two input tones fIN1 and fIN2. The individual input tone levels are at -8dBFS. The intermodulation products are as follows: Second-Order Intermodulation Products (IM2): fIN1 + fIN2, fIN2 - fIN1 Third-Order Intermodulation Products (IM3): 2 x fIN1 - fIN2, 2 x fIN2 - fIN1, 2 x fIN1 + fIN2, 2 x fIN2 + fIN1 Fourth-Order Intermodulation Products (IM4): 3 x fIN1 - fIN2, 3 x fIN2 - fIN1, 3 x fIN1 + fIN2, 3 x fIN2 + fIN1, 2 x fIN1 - 2 x fIN2 Fifth-Order Intermodulation Products (IM5): 3 x fIN1 - 2 x fIN2, 3 x fIN2 - 2 x fIN1, 3 x fIN1 + 2 x fIN2, 3 x fIN2 + 2 x fIN1, 4 x fIN1 - fIN2 Note that the two-tone intermodulation distortion is measured with respect to a single-carrier amplitude and not the peak-to-average input power of both input tones. Aperture Jitter Aperture jitter (tAJ) represents the sample-to-sample variation in the aperture delay specification. Aperture Delay Aperture delay (tAD) is the time defined between the rising edge of the sampling clock and the instant when an actual sample is taken (Figure 5). ______________________________________________________________________________________ 19 MAX19586 Spurious-Free Dynamic Range (SFDR1 and SFDR2) Package Information (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information go to www.maxim-ic.com/packages.) 56L THIN QFN.EPS MAX19586 High-Dynamic-Range, 16-Bit, 80Msps ADC with -82dBFS Noise Floor Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2005 Maxim Integrated Products Freed Printed USA is a registered trademark of Maxim Integrated Products, Products. Inc.