A4403 Datasheet

A4403
Valley Current Mode Control Buck Converter
Features and Benefits
Description
▪ Extremely fast load-transient response with minimal
output voltage delta
▪ Achieves high step-down ratios with on-times < 50 ns
▪ User-configurable on-time, achieving switching
frequencies up to 2 MHz
▪ Minimal external components required
▪ Optimized for low value filter capacitors and inductors
▪ Wide input voltage range: 9 to 46 V
▪ Output Current: 3 A
▪ Low standby current <100 μA
▪ Supplied in thermally-enhanced QFN package
The A4403 is a buck converter that uses valley current-mode
control. This control scheme allows very short switch on-times
to be achieved, making it ideal for applications that require
high switching frequencies combined with high input voltages
and low output voltages.
Applications:
▪ Printers, scanners
▪ Cable, DSL modems/routers
▪ Network and telecom
▪ Industrial control
▪ Distributed power systems
▪ Set top box
▪ High power LED supply
▪ Battery chargers
▪ GPS / Infotainment
Package 16-contact QFN (suffix EU):
Low cost is accomplished through high switching frequencies
of up to 2 MHz, allowing smaller and lower value inductors
and capacitors. In addition, minimal external components are
required through high levels of integration. Optimal drive
circuits are utilized to minimize switching losses.
The switching frequency is maintained constant, as the on-time
is modulated by the input voltage. This feed-forward control
ensures excellent line correction. The on-time is set by an
external resistor pulled-up to the input supply.
When power is initially applied and the device is enabled, a
user-configurable soft-start function occurs to minimize inrush
current and to prevent output overshoot. Internal housekeeping
and bootstrap supplies are provided which only require the
addition of one small ceramic capacitor. A top-off charge pump
is also provide to ensure correct operation at light loads.
Internal diagnostics provide comprehensive protection against
overcurrents, input undervoltages, and overtemperatures.
The device package is a 16-contact, 4 mm × 4 mm, 0.75 mm
nominal overall height QFN, with exposed pad for enhanced
thermal dissipation. It is lead (Pb) free, with 100% matte tin
leadframe plating.
4 mm × 4 mm × 0.75 mm
Typical Application Diagram
R1
100 kΩ
BOOT
VIN
C1
2.2 μF
100 V
A 4403
TON
C2
22 nF
VOUT
5.0 V
3A
L1
6.3 μH
LX
C3
10 μF
6.3 V
ISEN
R3
50 mΩ
DIS
C4
10 μF
6.3 V
R5
3.92 kΩ
C5
47 nF
3.3 V
80.00
75.00
70.00
65.00
FB
GND
5V
85.00
SGND
SS
Efficiency versus Output Current
VIN = 42 V
90.00
D1
Efficiency (%)
VIN
9 to 46 V
R6
750 Ω
60.00
0
1
2
Output Current (A)
All capacitors are X5R or X7R ceramic
Resistors R3 and R4 should be surface mount, low inductance type, rated at 250 mW at 70°C
4403-DS, Rev. 2
3
A4403
Valley Current Mode Control Buck Converter
Selection Guide
Part Number
A4403GEU-T
A4403GEUTR-T
Packing
Package
92 pieces per tube
1500 pieces per 7-in. reel
16-contact 4 mm × 4 mm QFN with exposed thermal pad
Absolute Maximum Ratings (reference to GND)
Characteristic
Symbol
Notes
Rating
Units
VIN Pin Supply Voltage
VIN
–0.3 to 50
V
LX Pin Switching Node Voltage
VLX
–1 to 50
V
ISEN Pin Current Sense Voltage
VISEN
–1.0 to 0.5
V
DIS Pin Disable Voltage
VDIS
–0.3 to 7
V
TON Pin On-Time Voltage
VTON
–0.3 to 50
V
–40 to 105
ºC
TJ(max)
150
ºC
Tstg
–55 to 150
ºC
Operating Ambient Temperature
TA
Maximum Junction Temperature
Storage Temperature
Range G
Recommended Operating Conditions
Characteristic
Symbol
Conditions
Min.
Typ.
Max.
Units
Supply Voltage
VIN
9
–
46
V
Switching Node
VLX
–0.7
–
46
V
Switching Frequency Range
fSW
0.45
–
2
MHz
Operating Ambient Temperature
TA
–40
–
105
ºC
Junction Temperature
TJ
–40
–
125
ºC
Continuous conduction mode
Thermal Characteristics may require derating at maximum conditions, see application information
Characteristic
Symbol
Test Conditions*
Value
Units
Package Thermal Resistance, Junction to Ambient
RθJA
On 4-layer PCB based on JEDEC standard
36
ºC/W
Package Thermal Resistance, Junction to Pad
RθJP
On 4-layer PCB based on JEDEC standard
2
ºC/W
*Additional thermal information available on the Allegro website.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
2
A4403
Valley Current Mode Control Buck Converter
Functional Block Diagram
VIN 9 to 44.1 V
C2
22 nF
C1
2.2 μF
BOOT
VIN
Top-off
Charge
Pump
Linear
Regulator
R2
Not
Fitted
R1
68 kΩ
VIN
LX
Sleep
Circuit
VOUT
3.3 V
3A
L1
4.7 μH
C3
10 μF
Driver
C4
10 μF
D1
TON
On
Timer
Off
Timer
Control
Logic
ISEN
Blank
DIS
R3
100 mΩ
Overvoltage Reg Ref
Comparator +80 mV
Switch
Closed = On
R4
100 mΩ
C6
10 nF
R5
2.37 kΩ
SGND
+
–
+
–
Linear OK
Fault
FB
+
–
VIN UVLO
Regulator
Comparator
Amplifier
TSD
Soft-Start
R6
750 Ω
Reg Ref
SS
NC
C5
47 nF
GND
Switching Frequency = 1 MHz
All capacitors are X5R or X7R ceramic
Resistors R3 and R4 should be surface mount, low inductance type, rated at 250 mW at 70°C
C6 is an optional speed-up capacitor, to improve the transient response
Terminal List Table
VIN
1
13 NC
14 NC
15 NC
16 NC
Pin-out Diagram
12 LX
Number
Name
1
VIN
Input supply
Function
2, 7, 13, 14, 15, 16
NC
No connection; tie to GND
3
TON
4
SS
Terminal for soft-start setting with external capacitor
5
FB
Feedback terminal
Terminal for on-time setting with external resistor
NC
2
6
GND
Ground terminal
TON
3
10 DIS
8
ISEN
Current sense input
SS
4
9
9
SGND
Current sense ground reference
10
DIS
11 BOOT
SGND
8
ISEN
6
7
NC
FB
GND
5
PAD
(Top View)
11
BOOT
12
LX
–
PAD
Disable logic input; active high
Bootstrap supply node
Switch node
Exposed thermal pad; connect to ground plane
(GND) by through-hole vias
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
3
A4403
Valley Current Mode Control Buck Converter
ELECTRICAL CHARACTERISTICS1 valid at TJ = 25°C, VIN = 9 to 46 V, unless otherwise noted
Characteristic
Symbol
Conditions
Min.
Typ.
Max.
Units
–
–
100
μA
General
VIN Quiescent Current
Feedback Voltage
Feedback Input Bias Current
Output Voltage Tolerance2
On-Time Tolerance
IVINOFF
IVINON
VFB
DIS = high, VIN = 46 V
DIS = low, VIN = 46 V, ILOAD= 1 mA
TJ = 25°C
IBIAS
–
4.3
5.5
mA
0.792
0.8
0.808
V
–400
–100
100
nA
∆VOUT
ILOAD = 1 mA to 3 A
–2.5
–
2.5
%
∆TON
Based on selected value
–15
–
15
%
Minimum On-Time Period
Ton(min)
–
50
60
ns
Minimum Off-Time Period
Toff(min)
–
–
350
ns
Buck Switch On-Resistance
RDS(on)
TJ = 25°C, ILOAD = 3 A
–
350
–
mΩ
TJ = 125°C, ILOAD = 3 A
–
550
–
mΩ
3.0
3.6
4.2
A
5
10
15
μA
–
–
1
V
Current Limit Threshold
ILIM
Soft Start Current Source
ISS
Valley current in external sense resistors = 50 mΩ
Input
DIS Input Voltage Threshold
DIS Open-Circuit Voltage
DIS Input Current
VDIS
VDISOC
IIN
Device enabled
2
–
7
V
DIS = 0 V
Device disabled
–10
–
–1
μA
–
0.88
–
V
Voltage rising
6.4
–
7.5
V
Protection
FB Overvoltage Shutdown
VFBOV
VIN Undervoltage Shutdown Threshold
VINUV
VIN Undervoltage Shutdown Hysteresis
VINUV(hys)
Overtemperature Shutdown Threshold
TJTSD
Overtemperature Shutdown Hysteresis
TJTSD(hys)
0.7
–
1.1
V
Temperature rising
–
165
–
°C
Recovery = TJTSD – TJTSD(hys)
–
15
–
°C
1Specifications
over the junction temperature range of –40°C to 125°C are assured by design and characterization.
2Average value of V
OUT relative to target voltage. Note that the tolerance effects of the feedback resistors are not taken into account. This figure does
include the feedback voltage tolerance.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
4
A4403
Valley Current Mode Control Buck Converter
Functional Description
Basic Operation The A4403 is a buck converter that utilizes
valley current-mode control. The on-time is set by the amount
of current that flows into the TON pin. This is determined by the
value of the TON resistors chosen (R1 and R2 in the Functional
Block diagram) and the magnitude of the input voltage, VIN.
Under a specific set of conditions, an on-time can be set that
then dictates the switching frequency. This switching frequency
remains reasonably constant throughout load and line conditions
as the on-time varies inversely with the input voltage. The Switch
On-Time and Switching Frequency section provides more details
on this subject.
At the beginning of the switching cycle, the buck switch is turned
on for a fixed period that is determined by the current flowing
into TON. Once the current comparator trips, a one-shot monostable, the On Timer, is reset, turning off the switch. The current
through the inductor then decays. This current is sensed through
the external sense resistors (R3 and R4), and then compared
against the current-demand signal. The current-demand signal is
generated by comparing the output voltage against an accurate
bandgap reference. After the current through the sense resistors
decreases to the valley of the current-demand signal, the On
Timer is set to turn the buck switch back on again and the cycle is
repeated.
Figure 1 illustrates how the current is limited during an overload
condition. The current decay (period with switch off) is proportional to the output voltage. As the overload is increased, the output voltage tends to decrease and the switching period increases.
Output Voltage Selection The output voltage of the converter
is set by selecting the appropriate feedback resistors, using the
following formula:
⎛V
⎞
R5 = R6 ⎜⎜ OUT – 1⎟⎟ ,
(1)
V
⎝ FB
⎠
where (refering to the Functional Block diagram):
R6 has a value between 750 Ω and 12 kΩ (R6 connected between
the GND and FB pins),
R5 is the dependent value (R5 connected between the output rail
and the FB pin),
VOUT is the user-configured output regulator voltage, and
VFB is the reference voltage.
The tolerance of the feedback resistors influences the voltage setpoint. It is therefore important to consider the tolerance selection
when targeting an overall regulation figure.
Inductor current operating at maximum load
Under light load conditions, the converter automatically operates
in pulse frequency modulation (PFM) mode to maintain regulation. This mode of operation ensures optimum efficiency as
switching losses are reduced.
During an overload condition, the switch is turned on for the
period determined by the constant on-time circuitry. The switch
off-time is extended until the current decays to the current limit
value of 3.6 A typical (which corresponds to a sense voltage of
180 mV). The switch is then turned on again.
Because no slope compensation is required in this control
scheme, the current limit is maintained at a reasonably constant
level across the input voltage range.
Current
Maximum load
Constant On-Time
Constant period
Time
Inductor current operating in a “soft” overload
Overload
Current Limit level
Current
Overcurrent Protection The converter utilizes pulse-by-pulse
valley current limiting, which operates when the current through
the sense resistors, R3 and R4 (set for 50 mΩ by two 100 mΩ
resistors in parallel), increases above 3.6 A typical at the valley
point. The corresponding sense voltage (at the ISEN pin) that creates a current limiting condition is 180 mV typical. It is possible,
by careful selection of the sense resistors, to reduce the current
limit for systems with maximum loads of less than 3 A.
Current Limit level
Constant On-Time
Extended period
Time
Figure 1. Current limiting during overload
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
5
A4403
Valley Current Mode Control Buck Converter
In general, the feedback resistors should have the lowest resistance possible, to minimize any noise pick-up effects and to
minimize voltage offsets on the output caused by the bias current,
IBIAS, flowing out of the FB node into R6. Reducing the feedback
resistances does introduce another loading effect on the output,
which has an effect on the standby current.
For example, if you limited IVIN to 250 mA, and assumed VOUT =
It should be noted that a minimum load of 1 mA is required (see
the Light Load Operation section). This may be provided by the
feedback resistors. For example, if R6 = 750 Ω, this guarantees a
1 mA load current.
This means a soft-start duration greater than 400 μs should be
5 V and COUT = 20 μF, the soft start time could be determined as:
tCHARGE =
20 μF × 5 V
= 400 μs
0.25 A
selected to ensure the inrush current is less than 250 mA.
Shutdown The converter is disabled in the event of either an
overtemperature event, or an undervoltage on VIN (VINUVR) or
Disable The converter is enabled by pulling the DIS pin low.
Once enabled, the output converter is started-up under the control
of the soft-start routine.
on an internal housekeeping supply.
To disable the converter, the DIS pin can simply be disconnected
(open circuit).
assuming DIS = 0, the output voltage, VOUT , is brought-up under
Soft Start A soft-start routine is initiated when: DIS = 0, no
thermal shutdown exists, and VIN and the internal housekeeping
supplies are above the minimum values. Note that an overcurrent
event does not initiate a soft start, unless the converter is recovering from a thermal shutdown condition.
As soon as any of the above faults have been removed and
the control of the soft-start routine.
Output Overvoltage Protection In the event of an overvoltage condition appearing on the output rail, the FB terminal
will also experience the overvoltage, scaled by the feedback resistors. If the FB terminal voltage rises above the nominal voltage
The soft-start routine controls the rate of rise of the reference
voltage, which in turn controls the output voltage. This function
minimizes the amount of inrush current drawn from VIN and
potential voltage overshoot on the output rail, VOUT.
by 10% (typical), the on-time of the buck switch will terminate
The soft-start period, TSS , is set by an internal current source
that charges the external capacitor (C5) connected to the SS pin.
Control by the soft-start routine is completed when the SS pin
reaches 0.8 V. The duration of TSS is set by selecting the appropriate capacitance, according to the formula:
Switch On-Time and Switching Frequency The switch
TSS =
C5 × 0.8
10 ×10–6
(2)
.
Note: If the soft start function is not required for the application,
a 220 kΩ resistor should be connected between the SS pin and
GND. Without soft start, or with a soft start period that is too
rapid, coupled with a high load that is present during start-up, the
converter may operate in current limit, placing maximum stress
on the input circuit.
Assuming no load is drawn until the start-up process is complete,
the current drawn from the input supply is determined by how
quickly the output capacitors (C3 and C4) are charged. The output capacitors are charged according to the following formula:
tCHARGE =
COUT × VOUT
IVIN
where IVIN is the input supply current.
,
(3)
and the switch will remain off until the FB voltage reduces to the
correct VFB range.
on-time effectively determines the operating frequency of the
converter. The selection of the operating frequency is generally
a trade-off between the size of the external passive components
(inductor, and input and output capacitors) and switching losses.
Another consideration in selecting the switching frequency is to
ensure that none of the on- or off-time limits are reached under
extreme conditions.
The minimum on-time occurs at maximum input voltage and
minimum load. Consider the following example.
Given:
VIN (max) = 46 V, VOUT = 5 V, fSW = 1 MHz, and:
⎛V
+ Vf
Ton(min) = ⎜⎜ OUT
V + Vf
⎝ IN
⎞
1
⎟×
⎟ fSW
⎠
,
(4)
where Vf is the voltage drop of the recirculation diode (D1) and
sense resistors (R3 and R4).
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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6
A4403
Valley Current Mode Control Buck Converter
Then, the minimum on-time is:
⎞
1
⎟×
⎟ 1× 106 = 118 ns
⎠
The specified minimum on-time, Ton(min) , is 60 ns maximum, so
there is reasonable margin in this case.
The specified minimum off-time, Toff(min) , 350 ns maximum,
also has to be considered. The minimum off-time occurs at
minimum input voltage and maximum load. As was shown in the
minimum on-time calculation (equation 4), you have to examine the extreme operating conditions to ensure adequate margin
exists.
The switch on-time, Ton, is set by the current flowing into the
TON pin. The current is determined by the input voltage, VIN,
and the resistor R1. The on-time can be found as:
⎛
R1
Ton = ⎜⎜
V × 2.05 ×1010
⎝ IN
⎞
⎟ + 10 × 10–9
⎟
⎠
.
(5)
The switching frequency may be slightly modulated by load
changes. The on-time is always constant for a given input voltage
and across the load range. To compensate for any losses in the
circuitry (for example, in the series switch and inductor, or in the
voltage drop across the recirculation diode), the off-time, hence
the switching frequency, has to be adjusted. This effect is most
noticeable at low input voltages and high output currents.
To calculate the actual switching frequency, the Ton of equation 5 can be used in conjunction with the transfer function of the
converter:
⎛ VOUT + Vf
fSW = ⎜⎜
V + Vf
⎝ IN
⎞
1
⎟×
⎟ T
on
⎠
.
(6)
An alternative approach to selecting the TON resistor (R1), to
accomplish an approximate switching frequency is found in the
following formula:
R1 =
VOUT × 2.05 × 1010
fSW
.
(7)
Figure 2 illustrates a range of switching frequencies that can be
achieved with various TON resistances and output voltages.
Light Load Operation To avoid the output voltage peak charging due to leakage effects from the buck switch and the charge
pump recirculation current, a minimum load of 1 mA must be
applied to the output.
The output feedback resistor network provides some loading.
Depending on the values selected, this network may provide all,
or at least some, of the minimum loading requirement.
Control Loop The process of closing the control loop for the
A4403 has been greatly simplified through the integration of
the compensation components into the device. The control loop
bandwidth has been optimized for operation across the full input
and output voltage range and for switching frequencies between
450 kHz and 2 MHz. Loop optimization is achieved with a 20 μF
ceramic capacitor placed across the output (VOUT to GND) and
a power inductor that achieves a peak to peak ripple current of
around 720 mA. For example, for a 3.3 V output operating at a
frequency of 1 MHz, the power inductor = 4.7 μH.
Larger output capacitors can be used; however, this tends to
decrease the bandwidth of the control loop. Note that the output
capacitance should not exceed 1000 μF or be less than 10 μF, as
this may cause a loop instability to occur.
2000
1800
Switching Frequency (kHz)
⎛ 5 + 0.5
Ton(min) = ⎜⎜
46 + 0.5
⎝
Top-Off Charge Pump During light load operation, when
operating in PFM mode, the top-off charge pump provides
enough charge to drive the buck switch.
VOUT
1600
1400
1200
1000
0.8 V
1.5 V
3.3 V
5V
12 V
800
600
400
10
100
Resistor R1 (kΩ)
1000
Figure 2. Switching frequencies versus TON resistor values, at various
levels of VOUT
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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7
A4403
Valley Current Mode Control Buck Converter
When the output voltage is set for 0.8 V, the typical bandwidth
is 90 kHz with a phase margin of 45° at full load. As the load is
reduced, the bandwidth remains largely constant; however, the
phase margin tends to reduce slightly because the output power
pole is shifted down in frequency, introducing the phase lag
sooner. At light loads, before pulse frequency modulation occurs,
the phase margin reduces to approximately 40°, which is reasonable given that it is the worst-case condition. Note that when
pulse frequency modulation occurs, the system no longer operates
as a linear system, therefore, the control laws do not apply.
When the output voltage is set for higher voltages, the DC gain
is reduced by the resistor feedback network from the output. This
effectively reduces the bandwidth of the control loop. An optional
speed-up capacitor (C6) can be used in parallel with the feedback
resistor (R5) to compensate for this effect. The addition of this
capacitor introduces an additional zero which increases the gain
and extends the bandwidth to maintain it in the region of 90 kHz.
The position of the zero depends on the values of R5 and C6.
The following time constants should be used for various output
voltages:
Output Voltage
(V)
5
Time Constant
(τ)
3.6 ×
2.4 × 10–5
2.5
1.8 × 10–5
1.5
1.1 × 10–5
0.8
Not required
For example, assume a target output voltage of 5 V, and an R5
of 3.92 kΩ to achieve that voltage. Then C6 = 9.18 × 10–9. The
nearest commonly available value is 10 nF.
For applications that require output voltages (VOUT) other than
what is defined above, the following formula should be used to
calculate the time constant:
τ = VOUT × 7.2 × 10–6 ,
The maximum peak to peak ripple current, IRIPP , occurs at the
maximum input voltage. Therefore the duty cycle, D, should be
found under these conditions:
D (min) =
VOUT+Vf
VIN(max)+Vf
,
(9)
where Vf is the forward voltage drop of the recirculation diode
and the sense resistor.
The required inductance can be found:
L (min) =
VIN – VOUT
× D (min) ×
IRIPP
1
fSW(min)
. (10)
Note that the manufacturers inductance tolerance should also be
taken into account. This value may be as high as ±20%.
In addition, because the control is dependant on the valley signal,
it is important to consider the minimum peak to peak valley
voltage that is developed across the sense resistor. The minimum
peak to peak ripple current occurs at minimum input voltage. The
peak to peak voltage is simply the peak to peak current multiplied
by the sense resistor value. It is recommended that the peak to
peak sense voltage should be greater than 25 mV.
10–5
3.3
A good starting point in selecting the inductance for a given
application is to specify a maximum peak-to-peak ripple current
of about 25% of the maximum load. The equates to a ripple current of approximately 750 mA for a maximum load of 3 A. This
often gives a good compromise between size, cost, and performance.
(8)
Inductor The main factor in selecting the inductance value is
the ripple current. The ripple current affects the output voltage
ripple and also has an effect on the current limit. Because slope
compensation is not used, the ripple current is not constrained by
this factor.
It is recommended that gapped ferrite solutions be used as
opposed to powdered iron solutions. The latter exhibit relatively
high core losses that can have a large impact on long term reliability.
Inductors are typically specified at two current levels:
• RMS current. It is important to understand how the RMS current level is specified, in terms of ambient temperature. Some
manufacturers quote an ambient only, whilst others quote a temperature that includes a self-temperature rise. For example, if an
inductor is rated for 85°C and includes a self-temperature rise of
25°C at maximum load, then the inductor cannot be safely operated beyond an ambient temperature of 60°C at full load. The
RMS current can be assumed to be simply the maximum load
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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8
A4403
Valley Current Mode Control Buck Converter
current, with perhaps some margin to allow for overloads, and
so forth.
to reduce the bandwidth and therefore compromise the transient
response performance.
• saturation current. The worst case maximum peak current
should not exceed the saturation current and indeed some margin should be allowed. The maximum peak current can be found
to ensure the saturation current level of the chosen inductor is
not exceeded:
In general the output capacitance should not exceed 1000 μF or
be less than 10 μF, as this may cause a loop instability to occur.
Isat = ILOAD +
IRIPPLE
2
.
(11)
It is important to ensure that, under worst-case conditions (minimum input voltage, maximum load current, minimum inductance,
and minimum switching frequency), that the minimum current
limit is not exceeded and in fact has some margin. The current
limit is measured at the valley level. The maximum current at the
valley is found from:
Ivalley
IRIPPLE
= ILOAD –
2
.
(12)
The minimum current limit threshold should be at least 20%
above this level.
Recommended inductor manufacturers and ranges are:
• Tayo Yuden: NR6045 series
• Sumida: CDR7D43MN series
Output Capacitor In the interests of size, cost, and performance, this control architecture has been designed for ceramic
capacitors. It is imperative that ceramic X5R or X7R capacitors
are used. On no account should Y5V, Y5U, Z5U, or similar types
be used.
When using ceramic capacitors, another important consideration is the E-field effects on the actual value of the capacitor.
To minimize the effects of the capacitance being reduced with
output voltage, it is recommended that the working voltage of the
capacitor be considerably more than the set output voltage. Check
with the vendor to obtain this information.
The output capacitor determines the output voltage ripple and is
used to close the control loop. As outlined in the Control Loop
section, the bandwidth has been optimized for an output capacitance of 20 μF.
If a particular application requires an extremely low output voltage, the output capacitor can be increased. Any increase will tend
The output ripple is largely determined by the output capacitance,
and the effects of ESR and ESL can largely be ignored assuming good layout practice is observed. To help reduce the effects
of ESL it is a good idea to split the 20 μF capacitance into two
separate 10 μF components.
The output voltage ripple can be approximated to:
VRIPPLE ≈
IRIPPLE
8 × fSW × COUT
,
(13)
where IRIPPLE is as found in the Inductor section.
When using ceramic capacitors, due to the negligible heating
effects of the ESR, there is generally no need to consider the current carrying capability. Also, the RMS current flowing into the
output capacitor is extremely low.
Input Capacitor It is recommended that ceramic X5R or X7R
capacitors be used, or at least that they be used in conjunction
with some other capacitor technology; for example, aluminum
electrolytic. Note that the self-resonance of electrolytics tend to
occur in the 100s of kHz, therefore the effects of ESL become
apparent at switching frequencies in the region of 1 MHz.
The value of the input capacitance determines the amount of
ripple voltage that appears at the source terminals. If a system is
designed correctly, the input capacitor should supply the switching current minus the input average current during the on-time of
the power switch. During the off-time of the power switch, the
input capacitor is charged-up.
The RMS current that flows in the input capacitor can be found
from:
Irms =
IOUT × VOUT
VIN
1/2
⎛ VIN
⎞
× ⎜⎜
– 1 ⎟⎟
V
⎝ OUT
⎠
,
(14)
The amount of ripple voltage that appears across the input terminals depends on: the amount of charge removed during the switch
on-time and the actual capacitor value. If a capacitor technology
such as an electrolytic is used, then the effects of ESR also have
to be considered.
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9
A4403
Valley Current Mode Control Buck Converter
The amount of capacitance required for a given ripple voltage can
be found:
CIN =
Irms × Ton
VRIPPLE
.
(15)
As mentioned in the previous section, E-field biasing effects
can reduce the actual capacitance and this should be taken into
account when making the selection.
Again, there is generally no need to consider the heating effects
of the RMS current flowing through the ESR of a ceramic
capacitor. If an electrolytic device is used, then the ripple current
rating should be considered. Note that most manufacturers only
consider the RMS current rating at 100 kHz.
Recirculation Diode This diode (D1) conducts during the
switch off-time. A Schottky diode is recommended to minimize
both the forward drop and switching losses. The worst-case
dissipation occurs at maximum VIN , when the duty cycle is at a
minimum.
The average current through the diode can be found:
IDIODE(av) = ILOAD × (1 – D (min)) .
(16)
The forward voltage drop, Vf , can be found from the diode
characteristics by using the actual load current (not the average
current).
The static power dissipation can be found:
PSTAT = IDIODE(av) × Vf .
The sense resistor value is selected depending on the maximum
output load current. The typical sense voltage that causes a current limit is 180 mV. So, for example, a 50 mΩ value would be
appropriate for a maximum load of 3 A, as it allows for margin
between maximum load and the current limit. A tolerance of up to
±5% is acceptable.
The power rating of the resistor has to be considered. The current
flowing in the resistor is essentially the same as the current flowing through the recirculation diode, although the power dissipation is worked out using the RMS current.
To a first approximation, the sense resistor dissipation can be
worked out as:
PSENSE = ILOAD2 × (1 – D (min)) × RSENSE .
(18)
For a converter working with a load of 3 A, a very narrow duty
cycle, and a sense resistor of 50 mΩ, the power dissipation would
be 450 mW.
The optimal solution from a cost perspective is to use two
100 mΩ, 1206-style resistors connected in parallel. Each resistor
is generally rated at 250 mW at 70°C ambient. Check the vendor
datasheet to verify the maximum ambient at full power.
When laying out the PCB, it is essential that the sense resistor
connections, carrying the power current (see figure 3), are as
short and wide as possible to minimize the effects of leakage
inductance noise. In addition, the Kelvin sense circuit connections should be as close to the sense resistor pads as possible.
(17)
It is also important to take into account the thermal rating of
the package, RθJA , and the ambient temperature, to ensure that
enough heatsinking is provided to maintain the diode junction
temperature within the safe operating area for the device.
To minimize the heating effects from the A4403 on the diode and
vice-versa, it is recommended that the diode be mounted on the
reverse side of the printed circuit board.
Sense Resistor The sense resistor should be a surface mount
package, with low inductance. On no account should a wirewound or through hole package be used. To prevent potential
mistriggering problems from occurring in noisy systems, it is
recommended that an R-C filter be applied across the sense resistor, as shown in figure 3.
Kelvin
connection
ISEN
RFILTER
47 Ω
A4403
RSENSE
CFILTER
1 nF
Power
current
SGND
Kelvin
connection
Figure 3. R-C filter added to the current sense circuit
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A4403
Valley Current Mode Control Buck Converter
RFILTER and CFILTER (R7 and C7 in the Typical Application
diagram) should be placed close to the A4403 pins. The ground
sense should connect directly to the SGND and not to the power
ground.
Estimate the RDS(on) of the buck switch at the given junction
temperature:
⎛
TJ – 25 ⎞
⎟ .
RDS(on)TJ = RDS(on)25C ⎜⎜ 1+
(20)
170 ⎟
⎝
⎠
Support Components The bootstrap capacitor (C2) and softstart capacitor (C5) should be ceramic X5R or X7R.
The static loss for each switch can be determined:
Thermal Considerations To ensure the A4403 operates in
the safe operating area, which effectively means restricting the
junction temperature to less than 150°C, several checks should be
made. The general approach is to work out what thermal impedance, RJA , is required to maintain the junction temperature at
a given level, for a particular power dissipation. (Another factor
worth considering is that other power dissipating components
on the system PCB may influence the thermal performance of
the A4403. For example, the power loss contribution from the
recirculation diode and the sense resistor may cause the junction
temperature of the A4403 to be higher than expected.) It should
be noted that this process is usually an iterative one to achieve the
optimum solution.
The following steps can be used as a guideline for determining
the RJA for a suitable thermal solution. :
1. Estimate the maximum ambient temperature, TA(max) , of the
application.
2. Define the maximum junction temperature, TJ(max). Note that
the absolute maximum is 150°C.
3. Determine the worst case power dissipation, PD(max). This will
occur at maximum load and minimum VIN. Contributors are:
(a) Switch static losses
VOUT + Vf
VIN (min) + Vf
(21)
where ILOAD is the load.
(b) Switch dynamic losses
Both the turn-on and the turn-off losses can be estimated:
PDYN = VIN(min) ×
ILOAD
× 5 × 10 –9 × fSW× 1.6
2
,
(22)
where fSW is the switching frequency.
(c) Diode capacitance turn-on loss
At turn-on, an additional current spike flows into the switch, causing a loss as follows:
PDIODECAP =
CDIODE × VIN2 × fSW
2
,
(23)
where CDIODE is the body capacitance of the Schottky diode (D1).
(d) Control losses
The control losses can be estimated as follows:
PCTRL = IVINON × VIN ,
(24)
where IVINON is the quiescent current with the converter enabled.
(e) Gate charge losses
Estimate the maximum duty cycle:
D (max) =
PSTAT = ILOAD2 × D (max) × RDS(on)TJ ,
,
Estimate the charge losses as follows:
(19)
where Vf is the forward voltage drop of the Schottky diode (D1)
and sense resistor (R2, R3) under the given load current.
PGATE = Q × fSW × VIN ,
(25)
where Q = 5 nC and is the charge that is required to turn on the
buck switch.
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A4403
Valley Current Mode Control Buck Converter
(c) Diode capacitance turn-on loss (equation 23):
(f) The total losses can now be estimated:
PTOTAL = PSTAT + PDYN + PDIODECAP + PCTRL + PGATE . (26)
4. The thermal impedance required for the solution can now be
determined:
RQJA =
T J – TA
PTOTAL
.
(27)
Note that if a four-layer high thermal efficiency board is used, a
thermal impedance of around 30°C/W can be achieved.
Example
Given selected parameters:
VIN(min) = 42 V,
VOUT = 3.3 V at 3 A,
fSW = 1 MHz,
TA = 70°C,
Target junction temperature, TJ = 115°C,
Vf = 0.55 V, and
CDIODE = 150 pF, then:
(a) Switch static losses
Maximum duty cycle (equation 19):
3.3 + 0.55
D (max) =
= 0.09
42 + 0.55
RDS(on) of the buck switch (equation 20):
⎛
115 – 25 ⎞⎟
= 0.535 Ω
RDS(on)TJ = 350 × 10 –3 ⎜⎜ 1+
170 ⎟
⎝
⎠
Static loss for each switch (equation 21):
PSTAT = 32 × 0.09 × 0.535 = 0.433 W
(b) Switch dynamic losses (equation 22):
PDYN = 42 ×
3
× 5 × 10 –9 × 1000 × 103 ×1.6 = 0.504 W
2
PDIODECAP =
150 × 10 –12 × 422 ×1 × 10 6
= 0.132 W
2
(d) Control losses (equation 24):
PCTRL = 0.004 × 42 = 0.168 W
(e) Gate charge losses (equation 25):
PGATE = 5 × 10 –9 × 1 × 106 × 42 = 0.21 W
(f) Total losses (equation 26):
PTOTAL = 0.433 + 0.504 + 0.132 + 0.168 + 0.21 = 1.447 W
Thermal impedance (equation 27):
RQJA =
115 – 70
= 31°C/W
1.447
For this particular solution, a PCB with high thermal efficency is
required to ensure the junction temperature is kept below 115°C.
For maximum effectiveness, the PCB area underneath the thermal
pad of the A4403 should be flooded with copper. Several thermal
vias (say between 4 and 8) should be used to connect the thermal
pad to the internal ground plane. If possible, a further thermal
copper plane should be applied to the bottom side of the PCB and
connected to the thermal pad of the A4403 through the vias.
This calculation assumes no thermal influence from other components. If possible, it is advisable to mount the recirculation diode
(D1) on the reverse side of the printed circuit board. Ensure low
impedance electrical connections are implemented between board
layers.
PCB Layout Guidelines The ground plane is largely dictated by the thermal requirements of the previous section. The
ground-referenced power components should be referenced to a
star ground, located away from the A4403 to minimize ground
bounce issues.
A small, local, relatively quiet ground plane near the A4403
should be used for the ground-referenced support components,
to minimize interference effects of ground noise from the power
circuitry. Figure 4 illustrates the recommended grounding architecture.
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A4403
Valley Current Mode Control Buck Converter
To avoid ground offset issues in the output voltage, it is highly
recommended that the ground-referenced feedback resistor R6
should be connected directly to the GND connection of the
A4403. In other words, the R6 ground return should avoid the use
of the internal ground plane.
(ISEN and SGND). Note that it is imperative that the PCB traces
between the sense resistor pads and the sense connections are as
short as possible to minimize the effects of leakage inductance.
In noisy systems, it is highly recommended that an R-C filter be
used to filter the signal produced across the ISEN pin. See the
Sense Resistor section and the Typical Application schematic.
All ground-referenced support components (C5 and the DIS
switch) should also be located as close to the GND connection as
possible. A “local quiet” ground plane around these components
can be implemented; however, this ground plane should have a
high impedance connection to the star ground connection of the
power stages, as referenced below.
If an internal ground plane is used, it is recommended that it
does not overlap the switching node, LX, to avoid the possibility
of noise pick up. To minimize the possibility of noise injection
issues, it is recommended to isolate the ground plane around the
high impedance nodes, such as FB and SS.
The sense resistor connections should be connected in a Kelvin
circuit (see figure 3) to the corresponding pins on the A4403
A4403 Support
Components
Power Circuitry
Switch
Cin
R6
A4403
C5
Local ‘quiet’
Ground Plane
D
Cout
R
GND
SGND
Star Connection
Thermal Vias
Internal Ground Plane
Figure 4. Ground plane configurations
VIN
Input
Voltage
L
LX
Q
VOUT
D
CIN
VIN
Input
Voltage
VOUT
D
CIN
COUT
L
LX
Q
COUT
R
R
RLOAD
RLOAD
Star Connection
Figure 5. FET on-cycle current conduction paths
Star Connection
Figure 6. FET off-cycle current conduction paths
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13
A4403
Valley Current Mode Control Buck Converter
In terms of grounding the power components, a star connection
should be made to minimize the ground loop impedances. Note
that, although a ground plane may be required to meet the thermal characteristics of the solution, it is still imperative to implement a star ground connection for the power components.
Figures 5 and 6 illustrate the importance of keeping the ground
connections as short as possible and forming good star connections.
Figure 6 shows the current conduction path during the off-cycle
of the switching FET. The following points should be noted:
• The boostrap capacitor, C2, and the soft start capacitor, C5,
should be located as close as possible to their respective terminal connections. The ground reference of the soft start capacitor
should be connected as close to the GND terminal as possible.
• The capacitor CIN should be placed as close as possible to the
VIN terminal.
R1
100 kΩ
• Good separation should exist between the LX connection and
any adjacent components or traces.
• The diode D should be placed as close as possible to both the
switching FET and to the inductor. The resistor R should be
placed as close as possible to the diode D.
Figure 5 also illustrates the current conduction paths during the
on-cycle of the switching FET. The following points should be
noted:
VIN
9 to 46 V
• The inductor L should placed as close as possible to the LX
terminal and to the output capacitors COUT.
BOOT
VIN
C1
2.2 μF
100 V
C2
22 nF
VOUT
5.0 V
3A
L1
6.3 μH
LX
D1
TON
A 4403
ISEN
R2
47 Ω
C7
1 nF
DIS
R3
100 mΩ
SGND
SS
C3
10 μF
6.3 V
C4
10 μF
6.3 V
R4
100 mΩ
R5
3.92 kΩ
C6
10 nF
C5
47 nF
FB
NC
GND
R6
750 Ω
Switching Frequency = 1 MHz
All capacitors are X5R or X7R ceramic
Resistors R3 and R4 should be surface mount, low inductance type, rated at 250 mW at 70°C
Figure 7. Typical application
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14
A4403
Valley Current Mode Control Buck Converter
Package EU, 16-Contact QFN
0.35
4.00 ±0.15
1
0.65
16
16
0.95
A
1
2
2
4.00 ±0.15
2.70
4.10
2.70
4.10
17X
D
SEATING
PLANE
0.08 C
0.30 ±0.05
0.75 ±0.05
0.65
C
C
PCB Layout Reference View
For Reference Only
(reference JEDEC MO-220WGGC)
Dimensions in millimeters
Exact case and lead configuration at supplier discretion within limits shown
A Terminal #1 mark area
B Exposed thermal pad (reference only, terminal #1
identifier appearance at supplier discretion)
0.40 ±0.10
B
2.70
2
1
C Reference land pattern layout (reference IPC7351
QFN65P400X400X80-17W2M)
All pads a minimum of 0.20 mm from all adjacent pads; adjust as necessary
to meet application process requirements and PCB layout tolerances; when
mounting on a multilayer PCB, thermal vias at the exposed thermal pad land
can improve thermal dissipation (reference EIA/JEDEC Standard JESD51-5)
D Coplanarity includes exposed thermal pad and terminals
16
2.70
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15
A4403
Valley Current Mode Control Buck Converter
Revision History
Revision
Revision Date
Rev. 2
June 17, 2013
Description of Revision
Update Functional Description
Copyright ©2008-2013, Allegro MicroSystems, LLC
Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to
permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that
the information being relied upon is current.
Allegro’s products are not to be used in life support devices or systems, if a failure of an Allegro product can reasonably be expected to cause the
failure of that life support device or system, or to affect the safety or effectiveness of that device or system.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its
use; nor for any infringement of patents or other rights of third parties which may result from its use.
For the latest version of this document, visit our website:
www.allegromicro.com
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