INTERSIL ISL8121IRZ

ISL8121
Features
The ISL8121 is a two-phase buck PWM controller
featuring an input voltage range of 3V to 20V and
integrated MOSFET drivers. Precision voltage regulation
is provided for point-of-load and other high-current
applications that require efficient and compact
implementation. Multiphase buck converter architecture
uses interleaved timing to multiply channel ripple
frequency allowing smaller filter inductors and reducing
input and output ripple currents due to the ripple
cancellation effect. Lower ripple results in fewer input
and output capacitors. Smaller and low cost transistors
can be used resulting from the higher efficiency and
reduced power dissipation.
• Integrated Two-Phase Power Conversion
The ISL8121 offers an internal 0.6V reference with a
system regulation accuracy of ±0.8% (Industrial
Temperature range), an optional external reference
input, and user-adjustable switching frequency. Unity
gain differential amplifier targeted at remote voltage
sensing capability enhances the regulation. A power good
signal (PGD) is issued when the output voltage is within
the regulated window. An internal shunt regulator with
optional external connection capability extends the
operational input voltage range. For applications
requiring voltage tracking or sequencing, the ISL8121
offers a host of possibilities, including coincidental,
ratiometric, or offset tracking, as well as sequential startups, user adjustable for a wide range of applications.
• Integrated High Capable 4A Drivers
Additional features include overvoltage and overcurrent
protection. Overcurrent protection can be tailored to
various applications with no need for additional parts.
The ISL8121 uses cost and space-saving rDS(ON) sensing
for channel current balance, dynamic voltage positioning,
and overcurrent protection. Channel current balancing is
automatic and accurate with the integrated currentbalance control system.
• Precision Output Voltage Regulation
- ±0.8% System Accuracy Over-Temperature
(Industrial)
- Differential Remote Voltage Sensing for Increased
Voltage Sensing Accuracy
• Shunt Regulator for Wide Input Power Conversion
- 5V and Higher Bias
- Up to 20V Power Down-Conversion
• Precision Channel Current Sharing
- Loss-Less rDS(ON) Current Sampling
• 0.6V Internal Reference
• Full Spectrum Voltage Tracking
- Coincidental, Ratiometric, or Offset
• Sequential Start-up Control
• Selectable Switching Frequency up to 2MHz Per
Phase
• Fast Transient Recovery Time
• Overcurrent Protection
• Overvoltage Protection
• Capable of Start-up in a Pre-Biased Load
• QFN Packages:
- QFN - Compliant to JEDEC PUB95 MO-220
- QFN - Quad Flat No Leads - Package Outline
- Near Chip Scale Package footprint, which improves
PCB efficiency and has a thinner profile
• Pb-Free Available (RoHS Compliant)
Applications
• General Purpose High Current DC/DC Converters
• High Current, Low Voltage FPGA/ASIC DC/DC
Converters
• Telecom System
October 27, 2009
FN6352.2
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2006, 2009. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL8121
3V to 20V, Two-Phase Buck PWM Controller with
Integrated 4A MOSFET Drivers
ISL8121
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
PART
MARKING
ISL8121IRZ
TEMP. RANGE
(°C)
81 21IRZ
PACKAGE
(Pb-Free)
-40 to +85
24 Ld 4x4 QFN
PKG.
DWG. #
L24.4x4C
NOTES:
1. Add “-T” suffix for tape and reel. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach
materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both
SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that
meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL8121. For more information on MSL please see
techbrief TB363.
Pin Configuration
REFTRK
SS
FS
PGD
BOOT1
UG1
ISL8121
(24 LD QFN)
TOP VIEW
24
23
22
21
20
19
18 PHASE1
VSEN- 1
17 ISEN1
VSEN+ 2
VMON 3
16 LG1
25
GND
VDIFF 4
15 PVCC
FB 5
14 LG2
13 ISEN2
2
7
8
9
10
11
12
VREG
VCC
EN
BOOT2
UG2
PHASE2
COMP 6
FN6352.2
October 27, 2009
Block Diagram
FS
PGD
VMON
VCC
EN
PVCC
BOOT1
20μA
300mV +
110%
-
90%
3
OSCILLATOR
VCC
10µA
UG1
POWER-ON
RESET (POR)
10µA
GATE
CONTROL
OVP
COMP
LG1
120%
SOFT-START
AND
FAULT LOGIC
REFERENCE
PWM1
GND
CONTROL
LOGIC
EA
FB
BOOT2
+
OC
-
VDIFF
PWM2
Σ
VCC
UG2
Σ
VSEN-
20μA
X1
VSEN+
VREG
VCC
CURRENT
CORRECTION
20µA/1mA
FN6352.2
October 27, 2009
ISEN2
PHASE2
LG2
SHUNT LINEAR
REGULATOR
ISEN1
GATE
CONTROL
SS
ISL8121
REFTRK
PHASE1
ISL8121
Simplified Power System Diagram
VIN
Q1
EN
CHANNEL1
Q2
PGD
VOUT
Q3
CHANNEL2
Q4
ISL8121
Typical Application
VIN
LIN
CHFIN1
RSHUNT
CBIN1
CF2
CF1
VCC
VREG
PVCC
BOOT1
REFTRK
CBOOT1
UG1
FS
Q1
PHASE1
LOUT1
RFS
ISEN1
SS
RISEN1
CSS
Q2
VMON
LG1
RPG
BOOT2
PGD
CBOOT2
ISL8121
R4
VOUT
CHFIN2
EN
UG2
CBIN2
CHFOUT
CBOUT
Q3
R5
PHASE2
ISEN2
C1
R2
COMP
LG2
C2
GND
RISEN2
RS
LOUT2
Q4
RP
FB
VDIFF
VSEN+ VSEN-
R1
4
FN6352.2
October 27, 2009
ISL8121
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC, PVCC . . . . . . . . . . . . . -0.3V to +6.5V
Shunt Regulator Voltage, VVREG . . . . . . . . . . -0.3V to +6.5V
Boot Voltage, VBOOT . . . . . . . . . PGND - 0.3V to PGND + 27V
Phase Voltage, VPHASE . . . . . . .VBOOT - 7V to VBOOT + 0.3V
Upper Gate Voltage, VUG . . . .VPHASE - 0.3V to VBOOT + 0.3V
Lower Gate Voltage, VLG . . . . . . PGND - 0.3V to VCC + 0.3V
Input, Output, or I/O Voltage. . . . GND - 0.3V to VCC + 0.3V
Thermal Resistance
θJA (°C/W) θJC (°C/W)
QFN Package (Notes 4, 5) . . . . . . .
43
7
Maximum Junction Temperature (Note 6). . . . . . . . . +150°C
Maximum Storage Temperature Range . . . -65°C to +150°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . +4.9V to +5.5V
Ambient Temperature . . . . . . . . . . . . . . . . . -40°C to +85°C
Junction Temperature Range . . . . . . . . . . . -40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact
product reliability and result in failures not covered by warranty.
NOTES:
4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach”
features. See Tech Brief TB379.
5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
6. Operation with die temperatures between +125°C and +150°C can be tolerated for short periods of time, however, in order
to maximize the operating life of the IC, it is strongly recommended that the effective continuous operating junction
temperature of the die should not exceed +125°C.
Electrical Specifications
Operating Conditions: VCC = 5V, TJ = -40°C to +85°C, unless otherwise specified. Boldface
limits apply over the operating temperature range, -40°C to +85°C.
PARAMETER
TEST CONDITIONS
MIN
MAX
(Note 9) TYP (Note 9) UNITS
BIAS SUPPLY AND INTERNAL OSCILLATOR
Input Bias Supply Current
IVCC; EN > 0.7V; LG, UG open
7.6
9
mA
Rising VCC POR (Power-On Reset) Threshold
4.30
4.40
4.50
V
VCC POR Hysteresis
0.46
0.51
0.58
V
Rising PVCC POR Threshold
3.60
3.67
3.75
V
Shunt Regulation
VVCC; IVREG = 0mA to 120mA
4.90
5.10
5.35
V
Maximum Shunt Current
IVREG_MAX
120
Switching Frequency (per Channel; Note 8)
FSW
150
-10
mA
2000
kHz
10
%
Frequency Tolerance
FSW
Oscillator Peak-to-Peak Ramp Amplitude
VOSC
1.4
V
Maximum Duty Cycle
dMAX
66
%
CONTROL THRESHOLDS
EN Threshold
EN Hysteresis Current
VMON Power-Good Enable Threshold
VVMON_TH
290
VMON Hysteresis Current
0.65
V
20
µA
305
320
mV
10
µA
22
µA
SOFT-START
SS Current
ISS
SS Ramp Amplitude
0.55
SS Threshold for Output Gates Turn-Off
0.40
3.60
V
V
REFERENCE AND DAC
System Accuracy (Industrial Temp. Range)
-0.8
Internal Reference
VREF
External Reference DC Amplitude Range
VREFTRK (DC)
5
0.8
0.6
0.1
%
V
2.3
V
FN6352.2
October 27, 2009
ISL8121
Electrical Specifications
Operating Conditions: VCC = 5V, TJ = -40°C to +85°C, unless otherwise specified. Boldface
limits apply over the operating temperature range, -40°C to +85°C. (Continued)
PARAMETER
MIN
MAX
(Note 9) TYP (Note 9) UNITS
TEST CONDITIONS
External Reference DC Offset Range
VREFTRK (DC) offset
-4.5
4.5
mV
ERROR AMPLIFIER AND REMOTE SENSING
RL = 10k to ground
80
Gain-Bandwidth Product (Note 7)
CL = 10pF
95
MHz
Slew Rate (Note 7)
CL = 10pF
30
V/µs
Maximum Output Voltage
No load
Minimum Output Voltage
No load
DC Gain (Note 7)
dB
4.0
V
0.7
VSEN+, VSEN- Input Resistance
140
V
225
kΩ
PGD Rising Lower Threshold
92
%
PGD Rising Upper Threshold
112
%
PGD Threshold Hysteresis
2.5
%
POWER GOOD
PROTECTION
Overcurrent Trip Level
80
103
120
µA
Overvoltage Threshold
VDIFF Rising
122
%
Overvoltage Hysteresis
VDIFF Falling
5.5
%
SWITCHING TIME
UG Rise Time (Note 7)
tRUG; VVCC = 5V, 3nF Load
8
ns
LG Rise Time (Note 7)
tRLG; VVCC = 5V, 3nF Load
8
ns
UG Fall Time (Note 7)
tFUG; VVCC = 5V, 3nF Load
8
ns
LG Fall Time (Note 7)
tFLG; VVCC = 5V, 3nF Load
4
ns
UG Turn-On Non-overlap (Note 7)
tPDHUG; VVCC = 5V, 3nF Load
8
ns
LG Turn-On Non-overlap (Note 7)
tPDHLG; VVCC = 5V, 3nF Load
8
ns
OUTPUT
Upper Drive Source Resistance
100mA Source Current
1.0
2.5
Ω
Upper Drive Sink Resistance
100mA Sink Current
1.0
2.5
Ω
Lower Drive Source Resistance
100mA Source Current
1.0
2.5
Ω
Lower Drive Sink Resistance
100mA Sink Current
0.4
1.0
Ω
NOTES:
7. Parameter magnitude guaranteed by design.
8. Not a tested parameter; range provided for reference only.
9. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established
by characterization and are not production tested.
Timing Diagram
tPDHUG
tRUG
tFUG
UG
LG
tFLG
tRLG
tPDHLG
6
FN6352.2
October 27, 2009
ISL8121
Functional Pin Descriptions
VCC (Pin 8)
Bias supply for the IC’s small-signal circuitry. Connect
this pin to a 5V supply and locally decouple using a
quality 0.1µF ceramic capacitor. This pin is monitored for
POR purposes. VCC bias may be applied in the absence
of PVCC bias.
PVCC (Pin 15)
Power supply pin for the MOSFET drives. Connect this pin
to a 5V supply and locally decouple using a quality 1µF
ceramic capacitor. This pin is monitored for POR
purposes. PVCC bias should not be applied in the
absence of VCC bias.
VREG (Pin 7)
This pin is the output of the internal shunt regulator. The
internal shunt regulator monitors and regulates the
voltage at the VCC pin. In applications where the chip
bias, including that necessary to drive the external
MOSFETs, is below the current rating of this pin, connect
it to VCC and PVCC, then connect this node to the input
supply via a properly sized resistor. Should the input
voltage vary over a wide range and/or the bias current
required exceed the intrinsic capability of the on-board
regulator, use this pin in conjunction with an external
NPN transistor and a couple of resistors to create a more
flexible bias supply for the ISL8121. In any configuration,
pay particular attention to the chip’s limitations in terms
of both current sinking capability of the shunt regulator,
as well as the internal power dissipation.
For more information, see “Bias Supply Considerations”
on page 16.
GND (Pin 25)
Connect this pad to the circuit ground using the shortest
possible path (one to four vias to the internal ground
plane, placed on the soldering pad are recommended).
All internal small-signal circuitry, as well as the lower
gates’ return paths are referenced to this pin.
REFTRK (Pin 24)
This pin represents an optional reference input, as well
as a clamp voltage for the internal reference. If utilizing
the ISL8121’s internal 0.6V reference, and desire no
special tracking features enabled, electrically connect
this pin to the VCC pin, or leave it open. Internal or
external reference operation mode is dictated by the
VMON pin.
While operating in internal reference mode, this pin
represents an internal reference clamp that can be used
for implementation of various tracking features. In this
operating mode, a small internal current is sourced on
this pin, pulling it high if left open.
If utilizing the ISL8121 in conjunction with an external
reference, connect the desired stimulus to this pin; the
sensed output of the ISL8121 converter follows this
input.
7
While operating with an external reference, the powergood and overvoltage protection functions are disabled
while the VMON pin voltage is below its threshold
(typically 300mV).
VMON (Pin 3)
The status of this pin is checked every time the chip is
enabled or POR is released; should its potential be lower
than 3.5V (typical), the REFTRK potential is assumed to
be an externally-provided reference and the ISL8121
proceeds to regulate the sensed output voltage to this
external reference. When operating using the internal
reference voltage, connect this pin to VCC (to bypass
the mechanism described above).
While operating with an externally-provided reference,
connect this pin to a properly-sized resistor divider off
the voltage to be monitored. PGD and OVP functions are
enabled when this pin exceeds its monitored threshold
(typically 300mV).
This pin is normally floating (high impedance input) until
it exceeds its detect threshold. Once the threshold is
exceeded, a small current is sourced on this pin; this
current, along with a properly sized resistor network,
allows the user to adjust the threshold hysteresis.
For more information, refer to “External Reference
Operation” on page 13.
EN (Pin 9)
This pin is a precision-threshold (approximately 0.6V)
enable pin. Pulled above the threshold, the pin enables
the controller for operation, initiating a soft-start.
Normally a high impedance input, once it is pulled
above its threshold, a small current is sourced on this
pin; this current, along with a properly sized resistor
network, allows the user to adjust the threshold
hysteresis. Pulled below the falling threshold, this pin
disables controller operation, by ramping down the SS
voltage and discharging the output.
VSEN+, VSEN-, and VDIFF (Pins 2, 1, and 4)
The inputs and output of the on-board unity-gain
operational amplifier intended for differential output
sensing. Connect VSEN- and VSEN+ to the output load’s
local GND and VOUT, respectively; VDIFF will reflect the
load voltage referenced to the chip’s local ground.
Connect the feedback network to the voltage thus
reflected at the VDIFF pin. Should the circuit not allow
implementation of remote sensing, connect the VSEN+
and VSEN- pins to the physical place where voltage is to
be regulated.
Connect the resistor divider setting the output voltage at
the input of the differential amplifier. To minimize the
error introduced by the resistance of differential
amplifier’s inputs, select resistor divider values smaller
than 1kΩ. VDIFF is monitored for overvoltage events and
for PGD reporting purposes.
FN6352.2
October 27, 2009
ISL8121
FB and COMP (Pins 5 and 6)
The internal error amplifier’s inverting input and output
respectively. These pins are connected to the external
network used to compensate the regulator’s feedback
loop.
ISEN1, ISEN2 (Pins 17, 13)
These pins are used to close the current feedback loop
and set the overcurrent protection threshold. A resistor
connected between each of these pins and their
corresponding PHASE pins determine a certain current
flow magnitude during the lower MOSFET’s conduction
interval. The resulting currents established through these
resistors are used for channel current balancing and
overcurrent protection.
Use Equation 1 to select the proper RISEN resistor:
r DS ( ON ) × I OUT
R ISEN = -----------------------------------------50μA
(EQ. 1)
where:
rDS(ON) = lower MOSFET drain-source ON resistance (Ω)
IOUT = channel maximum output current (A)
during start-up. Connect this pin to a capacitor
referenced to ground. Small internal current sources
linearly charge and discharge this capacitor, leading to
similar variation in the ramp up/down of the output
voltage. While below 0.3V, all output drives are turned
off. As this pin ramps up, the drives are not enabled but
only after the first UG pulse emerges (avoid draining the
output, if pre-charged). If no UG pulse are generated
until the SS exceeds the top of the oscillator ramp, at
that time all gate operation is enabled, allowing
immediate draining of the output, as necessary.
SS voltage has a ~0.7V offset above the reference
clamp, meaning the reference clamp rises from 0V with
unity gain correspondence as the SS pin exceeds 0.7V.
For more information, please refer to “Soft-Start” on
page 11.
FS (Pin 22)
This pin is used to set the switching frequency. Connect
a resistor, RFS, from this pin to ground and size it
according to the graph in Figure 1 or Equation 2.
R FS = 10
Read “Current Loop” on page 9, “Current Sensing” on
page 11, “Channel-Current Balance” on page 11, and
“Overcurrent Protection” on page 11 for more
information.
( 10.61 – ( 1.035 ⋅ log ( F SW ) ) )
(EQ. 2)
200k
Connect these pins to the upper MOSFETs’ gates. These
pins are used to control the upper MOSFETs and are
monitored for shoot-through prevention purposes.
Minimize the impedance of these connections. Maximum
individual channel duty cycle is limited to 66%.
BOOT1, BOOT2 (Pins 20, 10)
These pins provide the bias voltage for the upper
MOSFETs’ drives. Connect these pins to appropriatelychosen external bootstrap capacitors. Internal bootstrap
diodes connected to the PVCC pins provide the necessary
bootstrap charge. Minimize the impedance of these
connections.
PHASE1, PHASE2 (Pins 18, 12)
Connect these pins to the sources of the upper MOSFETs.
These pins are the return path for the upper MOSFETs’
drives. Minimize the impedance of these connections.
LG1, LG2 (Pins 16, 14)
These pins are used to control the lower MOSFETs and
are monitored for shoot-through prevention purposes.
Connect these pins to the lower MOSFETs’ gates.
Minimize the impedance of these connections.
SS (Pin 23)
This pin allows adjustment of the output voltage
soft-start ramp rate, as well as the hiccup interval
following an overcurrent event. The potential at this pin
is used as a clamp voltage for the internal error
amplifier’s non-inverting input, regulating its rate of rise
8
RFS VALUE (Ω)
UG1, UG2 (Pins 19, 11)
100k
50k
20k
10k
100k
200k
500k
1M
2M
SWITCHING FREQUENCY (Hz)
FIGURE 1. SWITCHING FREQUENCY VS. RFS VALUE
PGD (Pin 21)
This pin represents the output of the on-board
power-good monitor. Thus, the FB pin is monitored and
compared against a window centered around the
available reference; an FB voltage within the window
disables the open-collector output, allowing the external
resistor to pull-up PGD high. Approximate pull-down
device impedance is 65Ω.
While operating with an external reference, the
power-good function is enabled once the VMON pin
amplitude exceeds its monitored threshold (typically
300mV).
FN6352.2
October 27, 2009
ISL8121
ISL8121
VIN
OSCILLATOR
REFERENCE
UG1
PWM
CIRCUIT
Σ
HALF-BRIDGE
DRIVE
PHASE1
LG1
COMP
L1
R2
PWM
CIRCUIT
Σ
C1
HALF-BRIDGE
DRIVE
VOUT
RISEN1
ISEN1
FB
Σ
R1
COUT
VIN
CURRENT
SENSE
L2
UG2
AVERAGE
PHASE2
Σ
VDIFF
CURRENT
SENSE
LG2
ISEN2
RISEN2
VSEN+
VSEN-
FIGURE 2. SIMPLIFIED BLOCK DIAGRAM OF THE ISL8121 VOLTAGE AND CURRENT CONTROL LOOPS
Operation
Figure 2 shows a simplified diagram of the voltage
regulation and current loops. The voltage loop is used to
precisely regulate the output voltage, while the current
feedback is used to balance the output currents, IL1 and
IL2, of the two power channels.
VOLTAGE LOOP
Feedback from the output voltage is fed via the on-board
differential amplifier and applied via resistor R1 to the
inverting input of the Error Amplifier. The signal
generated by the error amplifier is summed up with the
current correction error signal and applied to the positive
inputs of the PWM circuit comparators. Out-of-phase
sawtooth signals are applied to the two PWM
comparators inverting inputs. Increasing error amplifier
voltage results in increased duty cycle. This increased
duty cycle signal is passed to the output drivers with no
phase reversal to drive the external MOSFETs. Increased
duty cycle, translating to increased ON-time for the
upper MOSFET transistor, results in increased output
voltage, compensating for the low output voltage which
lead to the increase in the error signal in the first place.
this control is the voltage that is developed across
rDS(ON) of each lower MOSFET, while they are
conducting. A resistor converts and scales the voltage
across each MOSFET to a current that is applied to the
current sensing circuits within the ISL8121. Output from
these sensing circuits is averaged and used to compute a
current error signal. Each PWM channel receives a
current signal proportional to the difference between the
average sensed current and the individual channel
current. When a PWM channel’s current is greater than
the average current, the signal applied via the summing
correction circuit to the PWM comparator reduces the
output pulse width (duty cycle) of the comparator to
compensate for the detected above average current in
the respective channel.
MULTI-PHASE POWER CONVERSION
Multi-phase power conversion provides a cost-effective
power solution when load currents are no longer easily
supported by single-phase converters. Although its
greater complexity presents additional technical
challenges, the multi-phase approach offers advantages
with improved response time, superior ripple
cancellation, and thermal distribution.
CURRENT LOOP
INTERLEAVING
The current control loop is only used to finely adjust the
individual channel duty cycle, in order to balance the
current carried by each phase. The information used for
The switching of each channel in a ISL8121-based
converter is timed to be symmetrically out of phase with
the other channel. As a result, the two-phase converter
9
FN6352.2
October 27, 2009
ISL8121
has a combined ripple frequency twice the frequency of
one of its phases. In addition, the peak-to-peak
amplitude of the combined inductor currents is
proportionately reduced. Increased ripple frequency and
lower ripple amplitude generally translate to lower perchannel inductance and/or lower total output capacitance
for any given set of performance specification.
CIN CURRENT
Q1 D-S CURRENT
IL1 + IL2
Q3 D-S CURRENT
IL2
PWM2
FIGURE 4. INPUT CAPACITOR CURRENT AND
INDIVIDUAL CHANNEL CURRENTS IN A
2-PHASE CONVERTER
IL1
PWM1
FIGURE 3. PWM AND INDUCTOR-CURRENT
WAVEFORMS FOR 2-PHASE CONVERTER
Figure 3 illustrates the additive effect on output ripple
frequency. The two channel currents (IL1 and IL2),
combine to form the AC ripple current and the DC load
current. The ripple component has two times the ripple
frequency of each individual channel current.
To understand the reduction of ripple current amplitude
in the multi-phase circuit, examine Equation 3, which
represents an individual channel’s peak-to-peak inductor
current.
( V IN – V OUT ) ⋅ V OUT
I L, PP = --------------------------------------------------------L ⋅ fS ⋅ V
(EQ. 3)
IN
VIN and VOUT are the input and output voltages,
respectively, L is the single-channel inductor value, and
fS is the switching frequency.
The output capacitors conduct the ripple component of
the inductor current. In the case of multi-phase
converters, the capacitor current is the sum of the ripple
currents from each of the individual channels.
( V IN – N ⋅ V OUT ) ⋅ V OUT
I PP = -------------------------------------------------------------------L ⋅ fS ⋅ V
(EQ. 4)
IN
Peak-to-peak ripple current decreases by an amount
proportional to the number of channels. Output-voltage
ripple is a function of capacitance, capacitor equivalent
series resistance (ESR), and inductor ripple current.
Reducing the inductor ripple current allows the designer
to use fewer or less costly output capacitors (should
output high-frequency ripple be an important design
parameter).
10
Another benefit of interleaving is the reduction of input
ripple current. Input capacitance is determined in a large
part by the maximum input ripple current. Multi-phase
topologies can improve overall system cost and size by
lowering input ripple current and allowing the designer to
reduce the cost of input capacitance. The example in
Figure 4 illustrates input currents from a two-phase
converter combining to reduce the total input ripple
current.
Figure 28, part of “Input Capacitor Selection” on
page 24, can be used to determine the input-capacitor
RMS current based on load current and duty cycle. The
figure is provided as an aid in determining the optimal
input capacitor solution.
PWM OPERATION
One switching cycle for the ISL8121 is defined as the
time between consecutive PWM pulse terminations (turnoff of the upper MOSFET on a channel). Each cycle begins
when a switching clock signal commands the upper
MOSFET to go off. The other channel’s upper MOSFET
conduction is terminated 1/2 of a cycle later.
Once a channel’s upper MOSFET is turned off, the lower
MOSFET remains on for a minimum of 1/3 cycle. This
forced off time is required to assure an accurate current
sample. Following the 1/3-cycle forced off time, the
controller enables the upper MOSFET output. Once
enabled, the upper MOSFET output transitions high when
the sawtooth signal crosses the adjusted error-amplifier
output signal, as illustrated in Figure 2. Just prior to the
upper drive turning the MOSFET on, the lower MOSFET
drive turns the freewheeling element off. The upper
MOSFET is kept on until the clock signals the beginning of
the next switching cycle and the PWM pulse is
terminated.
FN6352.2
October 27, 2009
ISL8121
CURRENT SENSING
SOFT-START
ISL8121 senses current by sampling the voltage across
the lower MOSFET during its conduction interval.
MOSFET rDS(ON) sensing is a no-added-cost method to
sense current for load line regulation, channel current
balance, module current sharing, and overcurrent
protection.
The soft-start function allows the converter to bring up
the output voltage in a controlled fashion, resulting in a
linear ramp-up. As soon as the controller is fully enabled
for operation, the SS pin starts to output a small current
which charges the external capacitor, CSS, connected to
this pin. An internal reference clamp controlled by the
potential at the SS pin releases the reference to the input
of the error amplifier with a 1:1 correspondence for SS
potential exceeding 0.7V (typically). Figure 5 details a
normal soft-start start-up. Equation 6 helps determine
the approximate time period during which the controlled
output voltage is ramped from 0V to the desired DC-set
level.
The ISEN pins are used as current inputs for each
channel. Internally, a virtual ground is created at the
ISEN pins. The RISEN resistors are used to size the
current flow through the ISEN pins, proportional to the
lower MOSFETs’ rDS(ON) voltage, during their conduction
periods. The current thus developed through the ISEN
pins is internally averaged, then the current error signals
resulting from comparing the average to the individual
current signals are used for channel current balancing.
C SS ⋅ V REF
t SS = -------------------------------I SS
Select the value for the RISEN resistors based on the
room temperature rDS(ON) of the lower MOSFETs and the
full-load total converter output current, IFL.
r DS ( ON ) I FL
- ⋅ -------R ISEN = ---------------------50 ×10 – 6 2
(EQ. 6)
VOUT (0.5V/DIV)
(EQ. 5)
As this current sense path is also used for OC detection,
ensure that at maximum power train temperature rise
and maximum output current loading the OC protection
is not inadvertently tripped. OC protection current level
through the ISEN pins is listed in the “Electrical
Specifications” table on page 6.
CHANNEL-CURRENT BALANCE
Another benefit of multi-phase operation is the thermal
advantage gained by distributing the dissipated heat over
multiple devices and greater area. By doing this, the
designer avoids the complexity of driving multiple
parallel MOSFETs and the expense of using expensive
heat sinks and exotic magnetic materials.
All things being equal, in order to fully realize the thermal
advantage, it is important that each channel in a
multi-phase converter be controlled to deliver about the
same current at any load level. Intersil’s ISL8121 ensure
current balance by comparing each channel’s current to
the average current delivered by both channels and
making appropriate adjustments to each channel’s pulse
width based on the resultant error. The error signal
modifies the pulse width to correct any unbalance and
force the error toward zero.
Conversely, should a channel-to-channel imbalance be
desired, such imbalance can be created by adjusting the
individual channel’s RISEN resistor. Asymmetrical layouts,
where one phase of the converter is naturally carrying
more current than the other, or where one of the two
phases is subject to a more stringent thermal
environment limiting its current-carrying capability, are
instances where this adjustment is particularly useful,
helping to cancel out the design-intrinsic thermal or
current imbalances.
11
VintREF (0.5V/DIV)
GND>
VSS (1V/DIV)
EN (5V/DIV)
GND>
FIGURE 5. NORMAL SOFT-START WAVEFORMS FOR
ISL8121-BASED MULTI-PHASE CONVERTER
Whenever the ISL8121’s power-on reset falling threshold
is tripped, or it is disabled via the EN pin, the SS
capacitor is quickly discharged via an internal pull-down
device (represented as the 1mA, typical, current source).
As the SS pin’s positive excursion is internally clamped to
about 3.5V, insure that any external pull-up device does
not force more than 3mA into this pin.
Should OC protection be tripped while the ISL8121 is
operating in internal-reference mode and the SS pin not
be allowed to fully discharge the SS capacitor, the
ISL8121 cannot continue the normal SS cycling.
OVERCURRENT PROTECTION
The individual channel currents, as sensed via the PHASE
pins and scaled via the ISEN resistors, as well as their
combined average are continuously monitored and
compared with an internal overcurrent (OC) reference. If
the combined channel current average exceeds the
reference current, the overcurrent comparator triggers
an overcurrent event. Similarly, an OC event is also
triggered if either channel’s current exceeds the OC
reference for seven consecutive switching cycles.
FN6352.2
October 27, 2009
ISL8121
SETTING THE OUTPUT VOLTAGE
OUTPUT CURRENT
EN
GND>
SS
The ISL8121 uses a precision internal reference voltage
to set the output voltage. Based on this internal
reference, the output voltage can thus be set from 0.6V
up to a level determined by the input voltage, the
maximum duty cycle, and the conversion efficiency of
the circuit. Equation 7 estimates this maximum
amplitude the output voltage can be regulated to.
(EQ. 7)
V OUTMAX = d MAX ⋅ V IN ⋅ Efficiency
GND>
OUTPUT VOLTAGE
FIGURE 6. OVERCURRENT BEHAVIOR WHILE IN
INTERNAL REFERENCE MODE
As a result of an OC event, output drives turn off both
upper and lower MOSFETs, and the SS capacitor is
discharged via a 20µA current source. The behavior
following this standard response varies depending
whether the controller is operating in internal (using
internal reference; VMON > 3.5V) or external reference
mode (using external reference; VMON < 3.5V). As
shown in Figure 6, the soft-start capacitor discharge
prompted by the OC event is followed by two SS cycles,
during which the ISL8121 stays off. Following the
dormant SS cycles, the controller attempts to
re-establish the output. Should the OC condition been
removed, the output voltage is ramped up and operation
resumes as normal. Should the OC condition still be
present and result in another OC event, the entire
behavior repeats until the OC condition is removed or the
IC is disabled. Figure 7 details the OC behavior while in
external reference mode. Following the OC event, the
output drives are turned off, the ISL8121 latches off, and
the SS capacitor is discharged to ground. Resetting the
OC latch involves removal of bias power or cycling of the
EN pin (pictured in Figure 7). Should the OC event been
removed, the controller initiates a new SS cycle and
restores the output voltage.
OUTPUT CURRENT
EN
GND>
GND>
Obviously, ensure that the input voltage and all the
voltages sampled by the ISL8121 do not exceed the
controller’s absolute maximum limits, or any other limits
specified in this document.
ISL8121
EXTERNAL CIRCUIT
VREF
+
FB
ERROR
AMPLIFIER
R1
VDIFF
RS
VSEN+
+
+
X1
DIFFERENTIAL
AMPLIFIER
-
VSEN-
RP
TO VOUT
-
FIGURE 8. SETTING THE OUTPUT VOLTAGE AT THE
INPUT OF THE DIFFERENTIAL AMPLIFIER
The output voltage can be set via a simple resistor
divider, as shown in Figure 8. It is recommended this
resistor divider is connected at the input of the
differential amplifier (as this amplifier is powered from
the IC’s 5V bias and has limited input range).
V OUT
R P = R ⋅ ------------------------------------V OUT – V REF
V OUT
R S = R ⋅ ---------------V REF
(EQ. 8)
To avoid degradation of DC regulation tolerance due to
the differential amplifier’s input resistance, a size
requirement is placed on the combined value of RP and
RS. Consider R to be the parallel combination of these
two resistors, and use a value of 2kΩ or less for R; use
the following equations to determine the value of RP and
RS, based on the desired out-put voltage, the reference
voltage, and the chosen value of R.
SS
GND>
OUTPUT VOLTAGE
FIGURE 7. OVERCURRENT BEHAVIOR WHILE IN
EXTERNAL REFERENCE MODE
12
FN6352.2
October 27, 2009
ISL8121
ISL8121
EXTERNAL CIRCUIT
VREF
+
RS(R1)
FB
ERROR
AMPLIFIER
-
+
TO VOUT
RP
VDIFF
+
VSEN+
-
VSEN-
X1
DIFFERENTIAL
AMPLIFIER
FIGURE 9. SETTING THE OUTPUT VOLTAGE AT THE
FB PIN
The differential amplifier can be used even if remote
output sensing is not desired or not feasible, simply
connect VSEN- to the local ground and connect VSEN+ to
the output voltage being monitored. Should one desire to
bypass the differential amplifier, the circuit in Figure 9 is
recommended as the proper implementation. Since its
output is monitored for OVP and PGD purposes, the
differential amplifier needs to be connected to the
feedback circuit at all times, hence its input connections
to FB and local ground. However, its output, VDIFF, can
be left open. The resistor divider setting the output
voltage is calculated in a manner identical to that already
revealed.
ISL8121
EXTERNAL
CIRCUIT
VDIFF
ZIN
~2µA
-
VSEN-
+
DIFFERENTIAL
AMPLIFIER
VSEN+
ZIN > 240kΩ
OVER PROCESS, TEMPERATURE, AND
0 < (VSEN+)-(VSEN-) < 2.5V
FIGURE 10. SETTING THE OUTPUT VOLTAGE AT THE
INPUT OF THE DIFFERENTIAL AMPLIFIER
DIFFERENTIAL AMPLIFIER’S UNITY GAIN
NETWORK
The differential amplifier on the ISL8121 utilizes a typical
resistive network along with active compensating
circuitry to set its unity gain. This resistive network can
13
affect the DC regulation setpoint in proportion to its
relative magnitude compared to the external output
voltage setting resistor divider. Figure 10 details the
internal resistive network. For minimal impact on the
output voltage setting, follow the guidelines presented in
“Setting the Output Voltage” on page 12.
EXTERNAL REFERENCE OPERATION
The ISL8121 is capable of accepting an external voltage
and using it as a reference for its output regulation. To
enable this mode of operation, the VMON pin potential
has to be below 3.5V and the reference voltage has to be
connected to the REFTRK pin. The internal or external
reference mode of operation is latched in every time the
POR is released or the ISL8121 is enabled. The highest
magnitude external reference fed to the REFTRK pin that
the ISL8121 can follow is limited to 2.3V. The ISL8121
utilizes a small initial negative offset (typically about
50mV) in the voltage loop at the beginning of it softstart, to counteract any positive offsets that may have
undesirable effects. As this initial offset is phased out as
the reference is ramped up to around 200mV, in order to
avoid an error in the output regulation level, it is
recommended the external reference has an amplitude
(final, DC level) exceeding 300mV.
ISL8121
IH
(10µA)
EXTERNAL CIRCUIT
VDIFF
VCC
RS
VMON
+
VMON
COMPARATOR
-
RP
+
VREF
(300mV) -
FIGURE 11. SETTING THE MONITORING THRESHOLD
AND HYSTERESIS
While in external reference (ER) mode, the threshold
sensitive VMON pin can be used to control when the
ISL8121 starts to monitor the output for PGD and OVP
protection purposes. As shown in Figure 11, connect the
VMON pin to the voltage to be monitored via a resistor
divider. An internal current source helps set a
user-adjustable monitor threshold hysteresis. Choose
resistor values according to desired hysteresis voltage,
VH, and desired rising threshold, VT.
VH
R S = -------IH
R S ⋅ V REF
R P = ---------------------------V T – V REF
(EQ. 9)
Make note that, in Equation 9, VREF refers to the
reference of the VMON comparator (300mV).
Intersil recommends the VMON threshold is set to be
tripped when the external reference voltage reaches at
least 90% of the final DC value. Since PGD and OVP
monitoring are relative to the external reference
FN6352.2
October 27, 2009
ISL8121
magnitude, it is important to understand that the PGD
and OVP thresholds will move in proportion to the
moving reference (externally soft-started reference).
When operating in internal-reference mode, the value of
the reference is known to the ISL8121, so the VMON pin
function can be bypassed by tying it to VCC potential.
ISL8121 POWER-GOOD OPERATION
VOLTAGE TRACKING AND SEQUENCING
The open-collector PGD output reports on the quality of
the regulated output voltage. Once the ISL8121 is
enabled and the VMON pin is above its threshold, PGD
goes open circuit when the output enters the power-good
window (see “Electrical Specifications” table on page 6),
and stays open for as long as the output remains within
the specified window. The PGD is immediately pulled low
if the ISL8121 is disabled by removal of bias, toggling of
the EN pin, or upon encountering of an overcurrent
event; PGD is allowed to report the status of the output
as soon as operation is resumed following any of these
events.
By making creative use of the reference clamps at the SS
and REFTRK pins, and/or the available external reference
input, as well as the functionality of the EN pin, the
ISL8121 can accommodate the full spectrum of tracking
and sequencing options. Figures 13 and 14 offer some
implementation suggestions for a few typical situations.
VTARGET
VOUT
VTARGET
ISL8121
REFTRK
VOUT
+
EXT CIRCUIT
VREF
To VTARGET R
S
RP
+
E/A
FB
-
ISL8121
REFTRK
+
EXT CIRCUIT
R1
VREF
VDIFF
TO VTARGET R
S
+
RP
E/A
VSEN+
FB
-
+
RP
X1
R1
-
VDIFF
RP
V REF
--------------- = --------------------V OUT
RP + RS
VSEN+
VSEN-
RS
-
+
TO VOUT
FIGURE 13. COINCIDENTAL VOLTAGE TRACKING
+
X1
-
RP
V OUT
-------------------------- = --------------------V TARGET
RP + RS
VSEN-
+
TO VOUT
FIGURE 12. RATIOMETRIC VOLTAGE TRACKING
The VMON pin is used in external-reference
configurations, where the reference is controlled by a
circuit external to the ISL8121. As such, the ISL8121 has
no way of ‘knowing’ when the external reference has
stabilized to its full value, or is within a certain
percentage of its final value. Thus, the VMON pin’s
functionality can be used to indicate when a desired
threshold has been reached (either by monitoring the
reference itself, or the output voltage controlled by the
ISL8121). By default, when operating in external
reference mode and desiring PGD monitoring as shown in
this datasheet, it is recommended the VMON is set to trip
its threshold when the output voltage (or reference)
reaches 92% of the final set value, choosing the resistor
divider as to achieve a 2% hysteresis.
14
Simple ratiometric external voltage tracking, such as that
required by the termination voltage regulator for double
data rate (DDR) memory can be implemented by feeding
a reference voltage equal to 0.5 of the memory core
voltage (VDDQ) to the reference input of the ISL8121, as
shown in Figure 12. The resistor divider at the REFTRK
pin sets the VOUT level. Select a suitable SS capacitor,
such that the SS clamp does not interfere with the
desired ramp-up time or slope of VOUT.
Coincidental tracking using the internal reference results
in a behavior similar to that presented in Figure 13. The
resistor divider at the input of the differential amplifier
sets the output voltage, VOUT, to the desired regulation
level. The same resistor divider used at the REFTRK pin
divides down the voltage to be tracked, effectively
scaling it to the magnitude of the internal reference. As a
result, the output voltage ramps up at the same rate as
the target voltage, its ramp-up leveling off at the
programmed regulation level established by the RS/RP
resistor divider.
FN6352.2
October 27, 2009
ISL8121
OVP has the highest priority, bypassing all other control
mechanisms and acting directly onto the lower MOSFETs,
as described. Disabling the IC via the EN pin does not
turn off OVP protection.
VTARGET
VOUT
START-UP INTO A PRE-CHARGED OUTPUT
VOFS
ISL8121
REFTRK
EXT CIRCUIT
VREF
-
+
+
VOFS
The ISL8121 also has the ability to start up into a
pre-charged output, without causing any unnecessary
disturbance. The FB pin is monitored during soft-start,
and should it be higher than the equivalent internal
ramping reference voltage, the output drives hold both
MOSFETs off. Once the internal ramping reference
exceeds the FB pin potential, the output drives are
enabled, allowing the output to ramp from the
pre-charged level to the final level dictated by the circuit
setting.
TO VTARGET
RS
+
RP
E/A
FB
R1
VOFS
+
-
+
+
OUTPUT PRE-CHARGED:
ABOVE INTERNAL REFERENCE
ABOVE EXTERNAL REFERENCE
BELOW REFERENCE
VDIFF
VSEN+
+
-
-
RP
V REF
--------------- = --------------------V OUT
RP + RS
VOUT (1.0V/DIV)
RP
X1
VSEN-
RS
-
SS (1V/DIV)
+
TO VOUT
GND>
OUTPUT INITIALLY
DISCHARGED
FIGURE 14. OFFSET VOLTAGE TRACKING
Offset tracking can be accomplished via a circuit similar
to that used for coincidental tracking (see Figure 14). The
desired offset can be implemented via a voltage source
inserted in line with the resistor divider present at the
REFTRK pin. Since most offset tracking requirements are
subject to fairly broad tolerances, simple voltage drop
sources can be used. Figure 14 exemplifies the use of
various counts of forward-biased diodes or that of a
schottky, although other options are available.
Sequential start-up control is easily implemented via the
EN pin, using either a logic control signal or the
ISL8121’s own EN threshold as a power-good detector
for the tracked, or sequence-triggering, voltage. See
Figure 15 for details of control using the EN pin.
OVERVOLTAGE PROTECTION
Although the normal feedback loop operation naturally
counters overvoltage (OV) events the ISL8121 benefits
from a secondary, fixed threshold overvoltage protection.
Should the output voltage exceed 120% of the reference,
the lower MOSFETs are turned on. Once turned on, the
lower MOSFETs are only turned off when the sensed
output voltage drops below the 110% falling threshold of
the OV comparator. The OVP behavior repeats for as long
as the ISL8121 is biased, should the sensed output
voltage rise back above the designated threshold. The
occurrence of an OVP event does not latch the controller;
should the phenomenon be transitory, the controller
resumes normal operation following such an event.
EN (5V/DIV)
GND>
FIGURE 15. SOFT-START WAVEFORMS INTO A
PRE-CHARGED OUTPUT CAPACITOR BANK
As shown in Figure 15, while operating in internal
reference mode, should the output be pre-charged to a
level exceeding the circuit’s output voltage setting, the
output drives are enabled at the conclusion of the
internal reference ramp, leading to an abrupt correction
in the output voltage down to the set level.
When operating in external reference mode, should the
output voltage be pre-charged above the regulation level
driven by the external reference, the output drives are
fully enabled when the SS pin levels out at the top of its
range.
CONTROL OF ISL8121 OPERATION
The internal power-on reset circuit (POR) prevents the
ISL8121 from starting before the bias voltage at VCC and
PVCC reach the rising POR thresholds, as defined in
“Electrical Specifications” on page 5. The POR levels are
sufficiently high to guarantee that all parts of the
ISL8121 can perform their functions properly once bias is
applied to the part. While bias is below the rising POR
thresholds, the controlled MOSFETs are kept in an off
state.
When operating in external-reference mode, the OVP
monitoring is enabled when the VMON pin exceeds its
rising threshold. For as long as the ISL8121 is biased,
15
FN6352.2
October 27, 2009
ISL8121
ISL8121
ISL8121
EXTERNAL CIRCUIT
+5V
POR
CIRCUIT
EXTERNAL CIRCUIT
PVCC
POR
CIRCUIT
VIN
VCC
VIN
VCC
VREF
(0.61V)
RBIAS
RS
E/A
-
+
EN
COMP
Q1
EN
RP
+
OFF
VREF
ON
IH
(20µA)
VREG
SHUNT REG
VCC
FIGURE 16. START-UP COORDINATION USING THE EN
PIN
A secondary disablement feature is available via the
threshold-sensitive enable input, the EN. This optional
feature prevents the ISL8121 from operating until a
certain chosen voltage rail is available and above some
selectable threshold. One example would be the input
voltage: it may be desirable the ISL8121-based
converter does not start up until the input voltage is
sufficiently high. The schematic in Figure 16
demonstrates coordination of the ISL8121 start-up with
such a rail. The internal current source, IH, provides the
means to a user-adjustable hysteresis. The resistor value
selection process follows the same equations as those
presented in “External Reference Operation” on page 13.
Additionally, an open-drain or open-collector device (Q1)
can be used to wire-AND a second (or multiple) control
signal. To defeat the threshold-sensitive enable, connect
EN to VCC directly or via a pull-up resistor.
In summary, for the ISL8121 to operate, VCC and PVCC
must be greater than their respective POR thresholds
and the voltage at EN must be greater than the
comparator’s reference (see typical threshold in
“Electrical Specifications” on page 5). Once these
conditions are met, the controller immediately initiates a
soft-start sequence.
General Application Design
Guide
This design guide is intended to provide a high-level
explanation of the steps necessary to create a multiphase power converter. It is assumed that the reader is
familiar with many of the basic skills and techniques in
the following sections. In addition to this guide, Intersil
provides complete reference designs that include
schematics, bills of materials, and example board
layouts for typical applications.
16
FIGURE 17. INTERNAL SHUNT REGULATOR USE WITH
EXTERNAL RESISTOR (PASSIVE
CONFIGURATION)
BIAS SUPPLY CONSIDERATIONS
The ISL8121 features an on-board shunt regulator
capable of sinking up to 100mA (minimally). This
integrated regulator can be used to produce the
necessary bias voltage for the controller and the
MOSFETs. The integrated regulator can be utilized
directly, via a properly sized resistor, as shown in
Figure 17, or via an external NPN transistor and
additional resistors when either the current needed or
the power being dissipated becomes too large to be
handled inside the ISL8121 in the given operating
environment.
A first step in determining the feasibility and selecting the
proper bias regulator configuration consists in
determining the maximum bias current required by the
circuit. While the bias current required by the ISL8121 is
listed in “Electrical Specifications” table on page 5, the
bias current required by the controlled MOSFETs needs
be calculated. The following equation helps determine
this bias current function of the sum of the gate charge
of all the controlled MOSFETs at 5V VGS, QGTOTAL, and
the switching frequency, FSW:
I B ≅ Q GTOTAL ⋅ F SW
(EQ. 10)
I BIAS = I VCC + I B
Total required bias current, IBIAS, sums up the ISL8121’s
bias current, IVCC, to that required by the MOSFETs, IB.
FN6352.2
October 27, 2009
ISL8121
7
90
70
6
5
60
50
40
30
DVIN = 5V
DVIN = 6V
DVIN = 7V
DVIN = 8V
20
10
RBIASmin (kΩ•mA)
IBIAS_MAX (% OF IVREG_MAX)
80
DVIN = 1V
DVIN = 2V
DVIN = 3V
DVIN = 4V
4
3
2
1
0
6
7
8
9
VINmin (V)
10
11
0
6
FIGURE 18. NORMALIZED MAXIMUM BIAS CURRENT
OBTAINABLE IN PASSIVE
CONFIGURATION vs INPUT VOLTAGE
RANGE CHARACTERISTIC; VVCC = 5V
VCC
To exemplify the use, for an input voltage ranging from
10V to 14V, find the intersection of the ΔVIN = 4V curve
with the VINmin = 10V mark and project the result on the
Y axis to find the maximum bias current obtainable
(approximately 56% of the maximum current obtainable
via the integrated shunt regulator, IVREG_MAX).
Once the maximum obtainable bias current, IBIAS_MAX,
is determined, and providing it is greater than the bias
current, IBIAS, required by the circuit, RSHUNT can be
determined based on the lowest input voltage, VINMIN:
V INMIN – V VCC
R BIAS = -----------------------------------------I BIAS
(EQ. 12)
Figure 19 helps with a quick resistor selection based on
the previous guidelines presented. Divide the value thus
obtained by the maximum desired bias current, IBIAS, to
obtain the actual resistor value to be used.
Figure 20 details the normalized maximum power
dissipation RBIAS will be subject to in the given
application. To use the graph provided, find the power
dissipation level corresponding to the minimum input
voltage and the input voltage range and multiply it by the
maximum desired bias current to obtain the maximum
power RBIAS will dissipate.
17
10
150
11
ΔVIN = 1V
ΔVIN = 2V
ΔVIN = 3V
ΔVIN = 4V
ΔVIN = 5V
ΔVIN = 6V
ΔVIN = 7V
ΔVIN = 8V
135
120
PMAX_RBIAS (mΩ/mA)
INMAX
(EQ. 11)
8
9
VINmin (V)
FIGURE 19. NORMALIZED RESISTOR VALUE IN
PASSIVE CONFIGURATION; VVCC = 5V
The maximum bias current, IBIAS, that can be obtained
via the internal shunt regulator and a simple external
resistor is characterized in Figure 18 and can also be
determined using Equation 11:
V INMIN – V VCC
I BIASMAX = I VREGMAX ⋅ -------------------------------------------V
–V
7
105
90
75
60
45
30
15
0
6
7
8
9
VINmin (V)
10
11
FIGURE 20. NORMALIZED RESISTOR POWER
DISSIPATION (AS SELECTED IN FIGURE 17)
vs MINIMUM INPUT VOLTAGE; VVCC = 5V
Alternately, the maximum power dissipation inside RBIAS
can be calculated using Equation 13:
P MAXRBIAS = ( V INMAX – V VCC ) ⋅ I BIAS
(EQ. 13)
Maximum power dissipation in the bias resistor will take
place at the upper end of the input voltage range. Select
a resistor with a power dissipation rating above the
calculated value and pay attention to design aspects
related to the power dissipation level of this component.
Although Figures 18 through 20 assume a VCC voltage of
5V, the design aid curves can be translated to a different
VCC voltage by translating them in the amount of the
voltage differential, to the left for a lower VCC voltage, or
to the right for a higher VCC voltage,
FN6352.2
October 27, 2009
ISL8121
Should the simple series bias resistor configuration fall
short of providing the necessary bias current, the internal
shunt regulator can be used in conjunction with an
external BJT transistor to increase the shunt regulator
current. Figure 21 details such an implementation
utilizing a PNP transistor. Selection of R1 can be based on
the graphs provided for the passive regulator
configuration. Maximum power dissipation inside Q1 will
take place when maximum voltage is applied to the
circuit and the ISL8121 is disabled; determine IVREGMAX
by reverse-use of the graph in Figure 18 and use the
obtained number to calculate Q1 power dissipation.
ISL8121
VCC
E/A
VIN
Q1
POR
CIRCUIT
R1
(OPTIONAL)
+
VREF
ISL8121
EXTERNAL CIRCUIT
PVCC
VREG
EXTERNAL CIRCUIT
R2
PVCC
VIN
POR
CIRCUIT
R1
VCC
Q1
E/A
-
R2
(OPTIONAL)
+
VREF
VREG
SHUNT REGULATOR
SHUNT REGULATOR
FIGURE 22. INTERNAL SHUNT REGULATOR USE WITH
EXTERNAL NPN TRANSISTOR (ACTIVE
CONFIGURATION)
FREQUENCY COMPENSATION
The ISL8121 multi-phase converter behaves in a similar
manner to a voltage-mode controller. This section
highlights the design consideration for a voltage-mode
controller requiring external compensation. To address a
broad range of applications, a type-3 feedback network is
recommended (see Figure 23).
C2
FIGURE 21. INTERNAL SHUNT REGULATOR USE WITH
EXTERNAL PNP TRANSISTOR (ACTIVE
CONFIGURATION)
An NPN transistor can also be used to increase the
maximum available bias current, as shown in Figure 22.
Used as a series pass element, Q1 will dissipate the most
power when the circuit is enabled and operational, and
the input voltage, VIN, is at its highest level.
With the series pass element configuration shown in
Figure 22, the difference between the input and the
regulation level at the VCC pin has to be higher than the
lowest acceptable VCE of Q1 (may choose to run Q1 into
saturation, but must consider the reduced gain). Thus,
R2 has to be chosen such that it will provide appropriate
base current at lowest VCE of Q1. Next, ensure the
ISL8121’s IVREGMAX is not exceeded when the input
voltage swings to its highest extreme (assume base
current goes to 0 when the IC is disabled). R1 is an
optional circuit element: it can be added to offset some
of the power dissipation in Q1, but it also reduces the
available VCE for Q1. If utilizing such a series resistor,
check that it does not impede on the proper operation at
the lowest input voltage and choose a power rating
corresponding to the highest bias current that the
ISL8121 may require to drive the switching MOSFETs.
18
R2
C1
COMP
FB
R1
C3
ISL8121
R3
VDIFF (VOUT)
FIGURE 23. COMPENSATION CONFIGURATION FOR
ISL8121 CIRCUIT
Figure 24 highlights the voltage-mode control loop for a
synchronous-rectified buck converter, applicable, with a
small number of adjustments, to the multi-phase
ISL8121 circuit. The output voltage (VOUT) is regulated
to the reference voltage, VREF, level. The error amplifier
output (COMP pin voltage) is compared with the
oscillator (OSC) modified saw-tooth wave to provide a
pulse-width modulated wave with an amplitude of VIN at
the PHASE node. The PWM wave is smoothed by the
output filter (L and C). The output filter capacitor bank’s
equivalent series resistance is represented by the series
resistor E.
FN6352.2
October 27, 2009
ISL8121
1. Select a value for R1 (1kΩ to 5kΩ, typically).
Calculate value for R2 for desired converter
bandwidth (F0). If setting the output voltage via an
offset resistor connected to the FB pin, Ro in
Figure 24, the design procedure can be followed as
presented. However, when setting the output
voltage via a resistor divider placed at the input of
the differential amplifier, in order to compensate for
the attenuation introduced by the resistor divider,
the obtained R2 value needs be multiplied by a
factor of (RP+RS)/RP. The remainder of the
calculations remain unchanged, as long as the
compensated R2 value is used.
C2
COMP
R2
C3
R3
C1
-
R1
FB
E/A
+
Ro
VREF
VDIFF
-
V OSC ⋅ R1 ⋅ F 0
R2 = --------------------------------------------d MAX ⋅ V IN ⋅ F LC
VSEN-
+
VSEN+
VOUT
OSCILLATOR
VIN
PWM
CIRCUIT
VOSC
UG
HALF-BRIDGE
DRIVE
L
LG
ISL8121
C
E
EXTERNAL CIRCUIT
FIGURE 24. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN
The modulator transfer function is the small-signal
transfer function of VOUT /VCOMP. This function is
dominated by a DC gain, given by dMAXVIN /VOSC , and
shaped by the output filter, with a double pole break
frequency at FLC and a zero at FCE . For the purpose of
this analysis, L and D represent the individual channel
inductance and its DCR divided by 2 (equivalent parallel
value of the two output inductors), while C and E
represents the total output capacitance and its
equivalent series resistance.
1
F LC = --------------------------2π ⋅ L ⋅ C
1
F CE = -----------------------2π ⋅ C ⋅ E
(EQ. 14)
The compensation network consists of the error amplifier
(internal to the ISL8121) and the external R1-R3, C1-C3
components. The goal of the compensation network is to
provide a closed loop transfer function with high 0dB
crossing frequency (F0; typically 0.1 to 0.3 of FSW) and
adequate phase margin (better than 45 degrees). Phase
margin is the difference between the closed loop phase
at F0dB and 180°. The equations that follow relate the
compensation network’s poles, zeros and gain to the
components (R1, R2, R3, C1, C2, and C3) in Figure 23.
Use the following guidelines for locating the poles and
zeros of the compensation network:
19
2. Calculate C1 such that FZ1 is placed at a fraction of the
FLC, at 0.1 to 0.75 of FLC (to adjust, change the 0.5
factor to desired number). The higher the quality
factor of the output filter and/or the higher the ratio
FCE/FLC, the lower the FZ1 frequency (to maximize
phase boost at FLC).
1
C1 = -----------------------------------------------2π ⋅ R2 ⋅ 0.5 ⋅ F LC
D
PHASE
(EQ. 15)
(EQ. 16)
3. Calculate C2 such that FP1 is placed at FCE.
C1
C2 = --------------------------------------------------------2π ⋅ R2 ⋅ C1 ⋅ F CE – 1
(EQ. 17)
4. Calculate R3 such that FZ2 is placed at FLC. Calculate
C3 such that FP2 is placed below FSW (typically, 0.5
to 1.0 times FSW). FSW represents the per-channel
switching frequency. Change the numerical factor to
reflect desired placement of this pole. Placement of
FP2 lower in frequency helps reduce the gain of the
compensation network at high frequency, in turn
reducing the HF ripple component at the COMP pin
and minimizing resultant duty cycle jitter.
R1
R3 = ---------------------F SW
------------ – 1
F LC
1
C3 = ------------------------------------------------2π ⋅ R3 ⋅ 0.7 ⋅ F SW
(EQ. 18)
It is recommended a mathematical model is used to plot
the loop response. Check the loop gain against the error
amplifier’s open-loop gain. Verify phase margin results
and adjust as necessary. Equation 19 describes the
frequency response of the modulator (GMOD), feedback
compensation (GFB) and closed-loop response (GCL):
d MAX ⋅ V IN
1 + s(f) ⋅ E ⋅ C
G MOD ( f ) = ------------------------------ ⋅ ---------------------------------------------------------------------------------------2
V OSC
1 + s(f) ⋅ (E + D) ⋅ C + s (f) ⋅ L ⋅ C
1 + s ( f ) ⋅ R2 ⋅ C1
G FB ( f ) = ------------------------------------------------------ ⋅
s ( f ) ⋅ R1 ⋅ ( C1 + C2 )
(EQ. 19)
1 + s ( f ) ⋅ ( R1 + R3 ) ⋅ C3
⋅ ----------------------------------------------------------------------------------------------------------------------------C1 ⋅ C2
( 1 + s ( f ) ⋅ R3 ⋅ C3 ) ⋅ ⎛ 1 + s ( f ) ⋅ R2 ⋅ ⎛ ----------------------⎞ ⎞
⎝ C1 + C2⎠ ⎠
⎝
G CL ( f ) = G MOD ( f ) ⋅ G FB ( f )
where, s ( f ) = 2π ⋅ f ⋅ j
FN6352.2
October 27, 2009
ISL8121
COMPENSATION BREAK FREQUENCY EQUATIONS
1
F Z1 = -------------------------------2π ⋅ R2 ⋅ C1
1
F P1 = ----------------------------------------------C1 ⋅ C2
2π ⋅ R2 ⋅ ---------------------C1 + C2
1
F Z2 = --------------------------------------------------2π ⋅ ( R1 + R3 ) ⋅ C3
1
F P2 = -------------------------------2π ⋅ R3 ⋅ C3
(EQ. 20)
Figure 25 shows an asymptotic plot of the DC/DC
converter’s gain vs frequency. The actual Modulator Gain
has a high gain peak dependent on the quality factor (Q)
of the output filter, which is not shown. Using the
previously mentioned guidelines should yield a
compensation gain similar to the curve plotted. The open
loop error amplifier gain bounds the compensation gain.
Check the compensation gain at FP2 against the
capabilities of the error amplifier. The closed loop gain,
GCL, is constructed on the log-log graph of Figure 25 by
adding the modulator gain, GMOD (in dB), to the
feedback compensation gain, GFB (in dB). This is
equivalent to multiplying the modulator transfer function
and the compensation transfer function and then plotting
the resulting gain.
GAIN
FZ1FZ2
FP1
FP2
MODULATOR GAIN
COMPENSATION GAIN
CLOSED LOOP GAIN
OPEN LOOP E/A GAIN
system transient response leaving the output capacitor
bank to supply the load current or sink the inductor
currents, all while the current in the output inductors
increases or decreases to meet the load demand.
In high-speed converters, the output capacitor bank is
amongst the costlier (and often the physically largest)
parts of the circuit. Output filter design begins with
consideration of the critical load parameters: maximum
size of the load step, ΔI, the load-current slew rate, di/dt,
and the maximum allowable output voltage deviation
under transient loading, ΔVMAX. Capacitors are
characterized according to their capacitance, ESR, and
ESL (equivalent series inductance).
At the beginning of the load transient, the output
capacitors supply all of the transient current. The output
voltage will initially deviate by an amount approximated
by the voltage drop across the ESL. As the load current
increases, the voltage drop across the ESR increases
linearly until the load current reaches its final value. The
capacitors selected must have sufficiently low ESL and
ESR so that the total output-voltage deviation is less
than the allowable maximum. Neglecting the contribution
of inductor current and regulator response, the output
voltage initially deviates according to Equation 21:
di
ΔV ≈ ( ESL ) ----- + ( ESR ) ΔI
dt
R2
20 log ⎛ --------⎞
⎝ R1⎠
d MAX ⋅ V
IN
20 log --------------------------------V OSC
0
The filter capacitor must have sufficiently low ESL and
ESR so that ΔV < ΔVMAX.
GFB
LOG
GCL
LOG
GMOD
FLC
FCE
F0
FREQUENCY
FIGURE 25. ASYMPTOTIC BODE PLOT OF CONVERTER
GAIN
A stable control loop has a gain crossing with close to a
-20dB/decade slope and a phase margin greater than
45°. Include worst case component variations when
determining phase margin. The mathematical model
presented makes a number of approximations and is
generally not accurate at frequencies approaching or
exceeding half the switching frequency. When designing
compensation networks, select target crossover
frequencies in the range of 10% to 30% of the perchannel switching frequency, FSW.
OUTPUT FILTER DESIGN
The output inductors and the output capacitor bank
together form a low-pass filter responsible for smoothing
the square wave voltage at the phase nodes.
Additionally, the output capacitors must also provide the
energy required by a fast transient load during the short
interval of time required by the controller and power train
to respond. Because it has a low bandwidth compared to
the switching frequency, the output filter limits the
20
(EQ. 21)
Most capacitor solutions rely on a mixture of highfrequency capacitors with relatively low capacitance in
combination with bulk capacitors having high capacitance
but limited high-frequency performance. Minimizing the
ESL of the high-frequency capacitors allows them to
support the output voltage as the current increases.
Minimizing the ESR of the bulk capacitors allows them to
supply the increased current with less output voltage
deviation.
The ESR of the bulk capacitors is also responsible for the
majority of the output-voltage ripple. As the bulk
capacitors sink and source the inductor AC ripple current,
a voltage develops across the bulk-capacitor ESR equal
to IPP. Thus, once the output capacitors are selected and
a maximum allowable ripple voltage, VPP(MAX), is
determined from an analysis of the available output
voltage budget, Equation 22 can be used to determine a
lower limit on the output inductance.
( V IN – 2 ⋅ V OUT ) ⋅ V OUT
L ≥ ESR ⋅ ----------------------------------------------------------------f S ⋅ V IN ⋅ V PP ( MAX )
(EQ. 22)
Since the capacitors are supplying a decreasing portion
of the load current while the regulator recovers from the
transient, the capacitor voltage becomes slightly
depleted. The output inductors must be capable of
assuming the entire load current before the output
voltage decreases more than ΔVMAX. This places an
upper limit on inductance.
FN6352.2
October 27, 2009
ISL8121
4 ⋅ C ⋅ V OUT
L ≤ -------------------------------- ⋅ ( ΔV MAX – ΔI ⋅ ESR )
2
( ΔI )
(EQ. 23)
While the previous equation addresses the leading edge,
the following equation gives the upper limit on L for
cases where the trailing edge of the current transient
causes a greater output voltage deviation than the
leading edge.
2.5 ⋅ C
L ≤ ----------------- ⋅ ( ΔV MAX – ΔI ⋅ ESR ) ⋅ ( V IN – V O )
2
( ΔI )
(EQ. 24)
Normally, the trailing edge dictates the selection of L, if
the duty cycle is less than 50%. Nevertheless, both
inequalities should be evaluated, and L should be
selected based on the lower of the two results. In all
equations in this paragraph, L is the per-channel
inductance and C is the total output bulk capacitance.
LAYOUT CONSIDERATIONS
MOSFETs switch very fast and efficiently. The speed with
which the current transitions from one device to another
causes voltage spikes across the interconnecting
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the
circuit and lead to device overvoltage stress. Careful
component layout and printed circuit design minimizes
the voltage spikes in the converter. Consider, as an
example, the turnoff transition of the upper PWM
MOSFET. Prior to turnoff, the upper MOSFET was carrying
channel current. During the turnoff, current stops flowing
in the upper MOSFET and is picked up by the lower
MOSFET. Any inductance in the switched current path
generates a large voltage spike during the switching
interval. Careful component selection, tight layout of the
critical components, and short, wide circuit traces
minimize the magnitude of voltage spikes.
There are two sets of critical components in a DC/DC
converter using a ISL8121 controller. The power
components are the most critical because they switch
large amounts of energy. Next are small signal
components that connect to sensitive nodes or supply
critical bypassing current and signal coupling.
Although the ISL8121 allows for external adjustment of
the channel-to-channel current balancing (via the RISEN
resistors), it is desirable to have a symmetrical layout,
preferably with the controller equidistantly located from
the two power trains it controls. Equally important are
the gate drive lines (UG, LG, PHASE): since they drive
the power train MOSFETs using short, high current
pulses, it is important to size them accordingly and
reduce their overall impedance. Equidistant placement of
the controller to the two power trains also helps keeping
21
these traces equally long (equal impedances, resulting in
similar driving of both sets of MOSFETs).
The power components should be placed first. Locate the
input capacitors close to the power switches. Minimize
the length of the connections between the input
capacitors, CIN, and the power switches. Locate the
output inductors and output capacitors between the
MOSFETs and the load. Locate all the high-frequency
decoupling capacitors (ceramic) as close as practicable to
their decoupling target, making use of the shortest
connection paths to any internal planes, such as vias to
GND immediately next, or even onto the capacitor’s
grounded solder pad.
The critical small components include the bypass
capacitors for VCC and PVCC. Locate the bypass
capacitors, CBP, close to the device. It is especially
important to locate the components associated with the
feedback circuit close to their respective controller pins,
since they belong to a high-impedance circuit loop,
sensitive to EMI pick-up. It is important to place the
RISEN resistors close to the respective terminals of the
ISL8121.
A multi-layer printed circuit board is recommended.
Figure 26 shows the connections of the critical
components for one output channel of the converter.
Note that capacitors CxxIN and CxxOUT could each
represent numerous physical capacitors. Dedicate one
solid layer, usually the one underneath the component
side of the board, for a ground plane and make all critical
component ground connections with vias to this layer.
Dedicate another solid layer as a power plane and break
this plane into smaller islands of common voltage levels.
Keep the metal runs from the PHASE terminal to inductor
LOUT short. The power plane should support the input
power and output power nodes. Use copper filled
polygons on the top and bottom circuit layers for the
phase nodes. Use the remaining printed circuit layers for
small signal wiring.
Size the trace interconnects commensurate with the
signals they are carrying. Use narrow (0.005” to
0.008”) and short traces for the high-impedance, smallsignal connections, such as the feedback,
compensation, soft-start, frequency set, enable,
reference track, etc. The wiring traces from the IC to
the MOSFETs’ gates and sources should be wide (0.02”
to 0.05”) and short, encircling the smallest area
possible.
FN6352.2
October 27, 2009
ISL8121
+12VIN
LIN
CBIN1
+5VIN
(CHFIN1)
(CF2)
(CF1)
VCC
PVCC
BOOT1
REFTRK
UG1
FS
RFS
CBOOT1
Q1
SS
PHASE1
CSS
ISEN1 R
ISEN1
VMON
LOCATE NEAR LOAD
(MINIMIZE CONNECTION PATH)
LOUT1
Q2
LG1
BOOT2
RPG
REN
VOUT
CBOOT2
ISL8121
(CHFOUT)
PGD
CBIN2
UG2
Q3
EN
C2
CBOUT
(CHFIN2)
PHASE2
COMP
ISEN2 R
ISEN2
C1
LG2
R2
LOUT2
RS
Q4
PGND
RP
FB
R1
GND
VDIFF VSEN+ VSEN-
LOCATE NEAR SWITCHING TRANSISTORS
(MINIMIZE CONNECTION PATH)
KEY
LOCATE CLOSE TO IC
(MINIMIZE CONNECTION PATH)
HEAVY TRACE ON CIRCUIT PLANE LAYER
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT PLANE LAYER
VIA CONNECTION TO GROUND PLANE
FIGURE 26. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
Component Selection Guidelines
MOSFETS
The selection of MOSFETs revolves closely around the
current each MOSFET is required to conduct, the
switching characteristics, the capability of the devices to
dissipate heat, as well as the characteristics of available
heat sinking. Since the ISL8121 drives the MOSFETs
with a 5V signal, the selection of appropriate MOSFETs
should be done by comparing and evaluating their
characteristics at this specific VGS bias voltage. The
following paragraphs addressing MOSFET power
dissipation ignore the driving losses associated with the
gate resistance.
22
The aggressive design of the shoot-through protection
circuits, part of the ISL8121 output drivers, is geared
toward reducing the deadtime between the conduction
of the upper and the lower MOSFET/s. The short
deadtimes, coupled with strong pull-up and pull-down
output devices driving the UG and LG pins make the
ISL8121 best suited to driving low gate resistance (RG),
late-generation MOSFETs. If employing MOSFETs with a
nominal gate resistance in excess of 1Ω to 2Ω, check for
the circuit’s proper operation. A few examples (non
exclusive list) of suitable MOSFETs to be used in
ISL8121 applications include the D8 (and later)
generation from Renesas and the OptiMOS®2 line from
Infineon. Along the same line, the use of gate resistors
is discouraged, as they may interfere with the circuits
just mentioned.
FN6352.2
October 27, 2009
ISL8121
LOWER MOSFET POWER CALCULATION
Since virtually all of the heat loss in the lower MOSFET is
conduction loss (due to current conducted through the
channel resistance, rDS(ON)), a quick approximation for
heat dissipated in the lower MOSFET can be found in
Equation 25:
P LMOS1 = r DS ( ON )
2
I L ,PP ( 1 – D )
⎛ I OUT⎞ 2
⎜
⎟ ( 1 – D ) + --------------------------------
------------⎝ 2 ⎠
(EQ. 25)
12
where: IM is the maximum continuous output current,
IL,PP is the peak-to-peak inductor current, and D is the
duty cycle (approximately VOUT/VIN).
An additional term can be added to the lower-MOSFET
loss equation to account for additional loss accrued
during the dead time when inductor current is flowing
through the lower-MOSFET body diode. This term is
dependent on the diode forward voltage at IM, VD(ON);
the switching frequency, fS; and the length of dead
times, td1 and td2, at the beginning and the end of the
lower-MOSFET conduction interval, respectively.
⎛ I OUT I PP⎞
⎛ I OUT I ⎞
PP-⎟ t
P LMOS 2 = V D ( ON ) f S ⎜ ------------ + ---------⎟ t d1 + ⎜ ------------ – -------2
2
2 ⎠ d2
⎝
⎠
⎝ 2
(EQ. 26)
Equation 26 assumes the current through the lower
MOSFET is always positive; if so, the total power
dissipated in each lower MOSFET is approximated by the
summation of PLMOS1 and PLMOS2.
UPPER MOSFET POWER CALCULATION
In addition to rDS(ON) losses, a large portion of the
upper-MOSFET losses are switching losses, due to
currents conducted through the device while the input
voltage is present as VDS. Upper MOSFET losses can be
divided into separate components, separating the
upper-MOSFET switching losses, the lower-MOSFET body
diode reverse recovery charge loss, and the upper
MOSFET rDS(ON) conduction loss.
In most typical circuits, when the upper MOSFET turns
off, it continues to conduct a decreasing fraction of the
output inductor current as the voltage at the phase node
falls below ground. Once the lower MOSFET begins
conducting (via its body diode or enhancement channel),
the current in the upper MOSFET decreases to zero. In
Equation 27, the required time for this commutation is
t1and the associated power loss is PUMOS,1.
⎛ I OUT I L ,PP⎞ ⎛ t 1 ⎞
P UMOS ,1 ≈ V IN ⎜ ------------+ -------------⎟ ⎜ ---- ⎟ f S
2 ⎠ ⎝ 2⎠
⎝ N
(EQ. 27)
Similarly, the upper MOSFET begins conducting as soon
as it begins turning on. Assuming the inductor current is
in the positive domain, the upper MOSFET sees
approximately the input voltage applied across its drain
and source terminals, while it turns on and starts
conducting the inductor current. This transition occurs
23
over a time t2, and the approximate the power loss is
PUMOS,2.
⎛ I OUT I L ,PP⎞ ⎛ t 2 ⎞
P UMOS , 2 ≈ V IN ⎜ ------------– -------------⎟ ⎜ ---- ⎟ f S
2 ⎠⎝ 2⎠
⎝ N
(EQ. 28)
A third component involves the lower MOSFET’s reverserecovery charge, QRR. Since the lower MOSFET’s body
diode conducts the full inductor current before it has fully
switched to the upper MOSFET, the upper MOSFET has to
provide the charge required to turn off the lower
MOSFET’s body diode. This charge is conducted through
the upper MOSFET across VIN, the power dissipated as a
result, PUMOS,3 can be approximated as:
P UMOS ,3 = V IN Q rr f S
(EQ. 29)
Lastly, the conduction loss part of the upper MOSFET’s
power dissipation, PUMOS,4, can be calculated using
Equation 30:
2
I PP2
⎛ I OUT⎞
P UMOS ,4 = r DS ( ON ) ⎜ ------------⎟ d + ---------12
⎝ N ⎠
(EQ. 30)
In this case, of course, rDS(ON) is the ON-resistance of
the upper MOSFET.
The total power dissipated by the upper MOSFET at full
load can be approximated as the summation of these
results. Since the power equations depend on MOSFET
parameters, choosing the correct MOSFETs can be an
iterative process that involves repetitively solving the
loss equations for different MOSFETs and different
switching frequencies until converging upon the best
solution.
OUTPUT CAPACITOR SELECTION
The output capacitor is selected to meet both the
dynamic load requirements and the voltage ripple
requirements. The load transient a microprocessor
impresses is characterized by high slew rate (di/dt)
current demands. In general, multiple high quality
capacitors of different size and dielectric are paralleled to
meet the design constraints.
Should the load be characterized by high slew rates,
attention should be particularly paid to the selection and
placement of high-frequency decoupling capacitors
(MLCCs, typically multi-layer ceramic capacitors). High
frequency capacitors supply the initially transient
current and slow the load rate-of-change seen by the
bulk capacitors. The bulk filter capacitor values are
generally determined by the ESR (effective series
resistance) and capacitance requirements.
High frequency decoupling capacitors should be placed
as close to the power pins of the load, or for that reason,
to any decoupling target they are meant for, as physically
possible. Attention should be paid as not to add
inductance in the circuit board wiring that could cancel
the usefulness of these low inductance components.
FN6352.2
October 27, 2009
ISL8121
Consult with the manufacturer of the load on specific
decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors.
The bulk capacitor’s ESR determines the output ripple
voltage and the initial voltage drop following a high slewrate transient’s edge. In most cases, multiple capacitors
of small case size perform better than a single large case
capacitor.
Bulk capacitor choices include aluminum electrolytic,
OS-Con, Tantalum and even ceramic dielectrics. An
aluminum electrolytic capacitor’s ESR value is related to
the case size with lower ESR available in larger case
sizes. However, the equivalent series inductance (ESL)
of these capacitors increases with case size and can
reduce the usefulness of the capacitor to high slew-rate
transient loading. Unfortunately, ESL is not a specified
parameter. Consult the capacitor manufacturer and/or
measure the capacitor’s impedance with frequency to
help select a suitable component.
OUTPUT INDUCTOR SELECTION
One of the parameters limiting the converter’s response
to a load transient is the time required to change the
inductor current. In a multi-phase converter, small
inductors reduce the response time with less impact to
the total output ripple current (as compared to
single-phase converters).
V OUT
K NORM = ------------------L ⋅ F SW
(EQ. 32)
where L is the channel inductor value.
Find the intersection of the active channel curve and duty
cycle for your particular application. The resulting ripple
current multiplier from the y-axis is then multiplied by
the normalization factor, KNORM, to determine the total
output ripple current for the given application.
(EQ. 33)
ΔI TOTAL = K NORM ⋅ K CM
INPUT CAPACITOR SELECTION
The important parameters for the bulk input
capacitors are the voltage rating and the RMS current
rating. For reliable operation, select bulk input
capacitors with voltage and current ratings above the
maximum input voltage and largest RMS current
required by the circuit. The capacitor voltage rating
should be at least 1.25 times greater than the
maximum input voltage. The input RMS current
required for a multi-phase converter can be
approximated with the aid of Figure 28. For a more
exact calculation of the input RMS current use
Equation 34:
I IN ( RMS ) =
0.8
2
2
2
D
I O ⋅ ( D – D ) + I L, PP ⋅ -----12
(EQ. 34)
0.3
0.6
0.4
0.2
0
0
0.1
0.2
0.3
0.4
0.5
DUTY CYCLE (VO/VIN)
FIGURE 27. RIPPLE CURRENT vs DUTY CYCLE
The output inductor of each power channel controls the
ripple current. The control IC is stable for channel ripple
current (peak-to-peak) up to twice the average current.
A single channel’s ripple current is approximated by
Equation 31:
V IN – V OUT V OUT
I L, PP = -------------------------------- × ---------------V IN
F SW ⋅ L
(EQ. 31)
The current from multiple channels tend to cancel each
other and reduce the total ripple current. The total output
ripple current can be determined using the curve in
24
INPUT CAPACITOR CURRENT (IIN(RMS) / IO)
CURRENT MULTIPLIER, KCM
1.0
Figure 27; it provides the total ripple current as a
function of duty cycle and number of active channels,
normalized to the parameter KNORM at zero duty cycle.
0.2
0.1
IL,PP = 0
IL,PP = 0.5 x IO
IL,PP = 0.75 x IO
0
0
0.1
0.2
0.3
0.4
0.5
DUTY CYCLE (VO /VIN)
FIGURE 28. NORMALIZED INPUT RMS CURRENT vs
DUTY CYCLE FOR A 2-PHASE CONVERTER
As the input capacitors are responsible for sourcing the
AC component of the input current flowing into the upper
MOSFETs, their RMS current capacity must be sufficient
to handle the AC component of the current drawn by the
upper MOSFETs. Figure 28 can be used to determine the
input-capacitor RMS current function of duty cycle,
maximum sustained output current (IO), and the ratio of
FN6352.2
October 27, 2009
ISL8121
Use a mix of input bypass capacitors to control the
input voltage ripple. Use ceramic capacitance for
the high frequency decoupling and bulk capacitors
to supply the RMS current. Minimize the connection
path inductance of the high frequency decoupling
ceramic capacitors (from drain of upper MOSFET to
source of lower MOSFET).
For bulk capacitance, several electrolytic or highcapacity MLC capacitors may be needed. For surface
mount designs, solid tantalum capacitors can be used,
but caution must be exercised with regard to the
capacitor surge current rating. These capacitors must
be capable of handling the surge-current at power-up.
APPLICATION SYSTEM DC TOLERANCE
Although the ISL8121 features a tight voltage
reference, the overall system DC tolerance can be
affected by the tolerance of the other components
employed. The resistive divider used to set the output
voltage will directly influence the system DC voltage
tolerance. Figure 29 details the absolute worst case
tolerance stack-up for 1% and 0.1% feedback
resistors, and assuming the ISL8121 is regulating at
0.8% above its nominal reference. Other component
tolerance stack-ups may be investigated using the
following equation, where REFTM, RPTM, and RSTM are
the tolerance multipliers corresponding to VREF, RS,
and RP, respectively.
( k – 1 ) ⋅ R STM + R PTM
REF TM ⋅ ------------------------------------------------------------- – 1
k ⋅ R PTM
TOL = -----------------------------------------------------------------------------------------------100
(EQ. 35)
[%]
2.8
RSTM = 1.01
2.6 R
PTM = 0.99
REFTM = 1.008
2.4
TOLERANCE (%)
the peak-to-peak inductor current (IL,PP) to the
maximum sustained load current, IO.
2.2
2.0
1.8
1.6
1.4
RSTM = 1.001
RPTM = 0.999
REFTM = 1.008
1.2
1.0
0.8
1
2
3
4
5
6
7
8
9
10
k = VOUT/VREF
FIGURE 29. WORST CASE SYSTEM DC REGULATION
TOLERANCE (VREF AT 0.8% ABOVE
NOMINAL)
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in the quality certifications found at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications
at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by
Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any
infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any
patent or patent rights of Intersil or its subsidiaries.
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25
FN6352.2
October 27, 2009
ISL8121
Package Outline Drawing
L24.4x4C
24 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 2, 10/06
4X 2.5
4.00
A
20X 0.50
B
PIN 1
INDEX AREA
PIN #1 CORNER
(C 0 . 25)
24
19
1
4.00
18
2 . 50 ± 0 . 15
13
0.15
(4X)
12
7
0.10 M C A B
0 . 07
24X 0 . 23 +- 0
. 05 4
24X 0 . 4 ± 0 . 1
TOP VIEW
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
C
0 . 90 ± 0 . 1
BASE PLANE
( 3 . 8 TYP )
SEATING PLANE
0.08 C
SIDE VIEW
(
2 . 50 )
( 20X 0 . 5 )
C
0 . 2 REF
5
( 24X 0 . 25 )
0 . 00 MIN.
0 . 05 MAX.
( 24X 0 . 6 )
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
26
FN6352.2
October 27, 2009