HIP6601, HIP6603 CT CT RODU T PRODU P E T LE MEN OBSO EPLACE 03B R Data Sheet 6 ED IP 6 MEND P6601B, H M O HI REC Synchronous-Rectified Buck MOSFET Drivers August 2004 Features • Drives Two N-Channel MOSFETs The HIP6601 and HIP6603 are high frequency, dual MOSFET drivers specifically designed to drive two power N-Channel MOSFETs in a synchronous-rectified buck converter topology. These drivers combined with a HIP630x Multi-Phase Buck PWM controller and Intersil UltraFETs™ form a complete core-voltage regulator solution for advanced microprocessors. The HIP6601 drives the lower gate in a synchronous-rectifier bridge to 12V, while the upper gate can be independently driven over a range from 5V to 12V. The HIP6603 drives both upper and lower gates over a range of 5V to 12V. This drive-voltage flexibility provides the advantage of optimizing applications involving trade-offs between switching losses and conduction losses. The output drivers in the HIP6601 and HIP6603 have the capacity to efficiently switch power MOSFETs at frequencies up to 2MHz. Each driver is capable of driving a 3000pF load with a 30ns propagation delay and 50ns transition time. Both products implement bootstrapping on the upper gate with only an external capacitor required. This reduces implementation complexity and allows the use of higher performance, cost effective, N-Channel MOSFETs. Adaptive shoot-through protection is integrated to prevent both MOSFETs from conducting simultaneously. • Adaptive Shoot-Through Protection • Internal Bootstrap Device • Supports High Switching Frequency - Fast Output Rise Time - Propagation Delay 30ns • Small 8 Lead SOIC Package • Dual Gate-Drive Voltages for Optimal Efficiency • Three-State Input for Bridge Shutdown • Supply Under Voltage Protection Applications • Core Voltage Supplies for Intel Pentium® III, AMD® Athlon™ Microprocessors • High Frequency Low Profile DC-DC Converters • High Current Low Voltage DC-DC Converters Pinout HIP6601CB/HIP6603CB (SOIC) TOP VIEW Ordering Information PART NUMBER TEMP. RANGE (oC) PACKAGE PKG. NO. HIP6601CB 0 to 85 8 Ld SOIC M8.15 HIP6603CB 0 to 85 8 Ld SOIC M8.15 FN4819.1 UGATE 1 8 PHASE BOOT 2 7 PVCC PWM 3 6 VCC GND 4 5 LGATE Block Diagram PVCC BOOT UGATE VCC +5V PHASE SHOOTTHROUGH PROTECTION 10K † VCC FOR HIP6601 PVCC FOR HIP6603 † PWM CONTROL LOGIC LGATE 10K GND 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2000, 2004. All Rights Reserved All other trademarks mentioned are the property of their respective owners. HIP6601, HIP6603 Typical Application +12V +5V BOOT VCC PWM PVCC UGATE DRIVE PHASE HIP6601 LGATE +12V +5V +5V +VCORE BOOT VFB COMP PWM1 VSEN PVCC PWM PWM2 PWM3 PGOOD UGATE VCC VCC DRIVE PHASE HIP6601 LGATE MAIN CONTROL HIP6301 VID ISEN1 ISEN2 FS +12V ISEN3 GND +5V BOOT PVCC UGATE VCC PWM DRIVE PHASE HIP6601 LGATE 2 HIP6601, HIP6603 Absolute Maximum Ratings Thermal Information Supply Voltage (VCC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .15V Supply Voltage (PVCC) . . . . . . . . . . . . . . . . . . . . . . . . . VCC + 0.3V BOOT Voltage (VBOOT - VPHASE) . . . . . . . . . . . . . . . . . . . . . . .15V Input Voltage (VPWM) . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 7V UGATE. . . . . . .VPHASE - 5V(<400ns pulse width) to VBOOT + 0.3V . . . . . . . . . . . VPHASE - 3.0V(>400ns pulse width) to VBOOT + 0.3V LGATE . . . . . . . . . GND - 5V(<400ns pulse width) to VPVCC + 0.3V . . . . . . . . . . . . . . GND - 3.0V(>400ns pulse width) to VPVCC + 0.3V PHASE. . . . . . . . . . . . . . . . . .GND - 5V(<400ns pulse width) to 15V . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V(>400ns pulse width) to 15V ESD Rating Human Body Model (Per MIL-STD-883 Method 3015.7) . . . . .3kV Machine Model (Per EIAJ ED-4701 Method C-111) . . . . . . .200V Thermal Resistance θJA (oC/W) SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113 Maximum Junction Temperature (Plastic Package) . . . . . . . .150oC Maximum Storage Temperature Range . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . .300oC (SOIC - Lead Tips Only) Operating Conditions Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . 0oC to 85oC Maximum Operating Junction Temperature . . . . . . . . . . . . . 125oC Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12V ±10% Supply Voltage Range, PVCC . . . . . . . . . . . . . . . . . . . . . 5V to 12V CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS HIP6601, fPWM = 1MHz, VPVCC = 12V - 4.4 6.2 mA HIP6603, fPWM = 1MHz, VPVCC = 12V - 2.5 3.6 mA HIP6601, fPWM = 1MHz, VPVCC = 12V - 200 430 µA HIP6603, fPWM = 1MHz, VPVCC = 12V - 1.8 3.3 mA VCC Rising Threshold 9.7 9.9 10.0 V VCC Falling Threshold 9.0 9.1 9.2 V - 500 - µA VCC SUPPLY CURRENT Bias Supply Current IVCC Power Supply Current IPVCC POWER-ON RESET PWM INPUT Input Current IPWM VPWM = 0 or 5V (See Block Diagram) PWM Rising Threshold 3.6 3.7 - V PWM Falling Threshold - 1.3 1.4 V UGATE Rise Time TRUGATE VPVCC = VVCC = 12V, 3nF load - 20 - ns LGATE Rise Time TRLGATE VPVCC = VVCC = 12V, 3nF load - 50 - ns UGATE Fall Time TFUGATE VPVCC = VVCC = 12V, 3nF load - 20 - ns TFLGATE LGATE Fall Time VPVCC = VVCC = 12V, 3nF load - 20 - ns UGATE Turn-Off Propagation Delay TPDLUGATE VVCC = VPVCC = 12V, 3nF load - 30 - ns LGATE Turn-Off Propagation Delay TPDLLGATE VVCC = VPVCC = 12V, 3nF load - 20 - ns 1.5 - 3.6 V - 230 - ns VVCC = 12V, VPVCC = 5V - 2.5 3.0 Ω VVCC = VPVCC = 12V - 7.0 7.5 Ω Shutdown Window Shutdown Holdoff Time OUTPUT Upper Drive Source Impedance RUGATE Upper Drive Sink Impedance RUGATE Lower Drive Source Impedance RLGATE Lower Drive Sink Impedance RLGATE 3 VVCC = 12V, VPVCC = 5V - 2.3 2.8 Ω VVCC = 12V, VPVCC = 12V - 1.0 1.3 Ω VVCC = 12V, VPVCC = 5V - 4.5 5.0 Ω VVCC = 12V, VPVCC = 12V - 9.0 9.5 Ω VVCC = VPVCC = 12V - 1.5 2.9 Ω HIP6601, HIP6603 Functional Pin Description PVCC (Pin 7) UGATE (Pin 1) For the HIP6601, this pin supplies the upper gate drive bias. Connect this pin from +12V down to +5V. Upper gate drive output. Connect to gate of high-side power N-Channel MOSFET. BOOT (Pin 2) Floating bootstrap supply pin for the upper gate drive. Connect the bootstrap capacitor between this pin and the PHASE pin. The bootstrap capacitor provides the charge to turn on the upper MOSFET. A resistor in series with boot capacitor is required in certain applications to reduce ringing on the BOOT pin. See the Internal Bootstrap Device section under DESCRIPTION for guidance in choosing the appropriate capacitor and resistor values. PWM (Pin 3) The PWM signal is the control input for the driver. The PWM signal can enter three distinct states during operation, see the three-state PWM Input section under DESCRIPTION for further details. Connect this pin to the PWM output of the controller. GND (Pin 4) Bias and reference ground. All signals are referenced to this node. LGATE (Pin 5) Lower gate drive output. Connect to gate of the low-side power N-Channel MOSFET. VCC (Pin 6) Connect this pin to a +12V bias supply. Place a high quality bypass capacitor from this pin to GND. For the HIP6603, this pin supplies both the upper and lower gate drive bias. Connect this pin to either +12V or +5V. PHASE (Pin 8) Connect this pin to the source of the upper MOSFET and the drain of the lower MOSFET. The PHASE voltage is monitored for adaptive shoot-through protection. This pin also provides a return path for the upper gate drive. Description Operation Designed for versatility and speed, the HIP6601 and HIP6603 dual MOSFET drivers control both high-side and low-side NChannel FETs from one externally provided PWM signal. The upper and lower gates are held low until the driver is initialized. Once the VCC voltage surpasses the VCC Rising Threshold (See Electrical Specifications), the PWM signal takes control of gate transitions. A rising edge on PWM initiates the turn-off of the lower MOSFET (see Timing Diagram). After a short propagation delay [TPDLLGATE], the lower gate begins to fall. Typical fall times [TFLGATE] are provided in the Electrical Specifications section. Adaptive shoot-through circuitry monitors the LGATE voltage and determines the upper gate delay time [TPDHUGATE] based on how quickly the LGATE voltage drops below 1.0V. This prevents both the lower and upper MOSFETs from conducting simultaneously or shoot-through. Once this delay period is complete the upper gate drive begins to rise [TRUGATE] and the upper MOSFET turns on. Timing Diagram PWM TPDHUGATE TPDLUGATE TRUGATE TFUGATE UGATE LGATE TRLGATE TFLGATE TPDLLGATE TPDHLGATE 4 HIP6601, HIP6603 A falling transition on PWM indicates the turn-off of the upper MOSFET and the turn-on of the lower MOSFET. A short propagation delay [TPDLUGATE] is encountered before the upper gate begins to fall [TFUGATE]. Again, the adaptive shoot-through circuitry determines the lower gate delay time, TPDHLGATE. The PHASE voltage is monitored and the lower gate is allowed to rise after PHASE drops below 0.5V. The lower gate then rises [TRLGATE], turning on the lower MOSFET. Three-State PWM Input A unique feature of the HIP660X drivers is the addition of a shutdown window to the PWM input. If the PWM signal enters and remains within the shutdown window for a set holdoff time, the output drivers are disabled and both MOSFET gates are pulled and held low. The shutdown state is removed when the PWM signal moves outside the shutdown window. Otherwise, the PWM rising and falling thresholds outlined in the ELECTRICAL SPECIFICATIONS determine when the lower and upper gates are enabled. Adaptive Shoot-Through Protection Both drivers incorporate adaptive shoot-through protection to prevent upper and lower MOSFETs from conducting simultaneously and shorting the input supply. This is accomplished by ensuring the falling gate has turned off one MOSFET before the other is allowed to rise. During turn-off of the lower MOSFET, the LGATE voltage is monitored until it reaches a 1.0V threshold, at which time the UGATE is released to rise. Adaptive shoot-through circuitry monitors the PHASE voltage during UGATE turn-off. Once PHASE has dropped below a threshold of 0.5V, the LGATE is allowed to rise. PHASE continues to be monitored during the lower gate rise time. If the PHASE voltage exceeds the 0.5V threshold during this period and remains high for longer than 2µs, the LGATE transitions low. Both upper and lower gates are then held low until the next rising edge of the PWM signal. Power-On Reset (POR) Function During initial startup, the VCC voltage rise is monitored and gate drives are held low until a typical VCC rising threshold of 9.9V is reached. Once the rising VCC threshold is exceeded, the PWM input signal takes control of the gate drives. If VCC drops below a typical VCC falling threshold of 9.1V during operation, then both gate drives are again held low. This condition persists until the VCC voltage exceeds the VCC rising threshold. Internal Bootstrap Device The HIP6601 and HIP6603 drivers feature an internal bootstrap device. Simply adding an external capacitor across the BOOT and PHASE pins completes the bootstrap circuit. 5 The bootstrap capacitor must have a maximum voltage rating above VCC + 5V. The bootstrap capacitor can be chosen from the following equation: Q GATE C BOOT ≥ -----------------------∆V BOOT Where QGATE is the amount of gate charge required to fully charge the gate of the upper MOSFET. The ∆VBOOT term is defined as the allowable droop in the rail of the upper drive. As an example, suppose a HUF76139 is chosen as the upper MOSFET. The gate charge, QGATE , from the data sheet is 65nC for a 10V upper gate drive. We will assume a 200mV droop in drive voltage over the PWM cycle. We find that a bootstrap capacitance of at least 0.325µF is required. The next larger standard value capacitance is 0.33µF. In applications which require down conversion from +12V or higher and PVCC is connected to a +12V source, a boot resistor in series with the boot capacitor is required. The increased power density of these designs tend to lead to increased ringing on the BOOT and PHASE nodes, due to faster switching of larger currents across given circuit parasitic elements. The addition of the boot resistor allows for tuning of the circuit until the peak ringing on BOOT is below 29V from BOOT to GND and 17V from BOOT to VCC. A boot resistor value of 5Ω typically meets this criteria. In some applications, a well tuned boot resistor reduces the ringing on the BOOT pin, but the PHASE to GND peak ringing exceeds 17V. A gate resistor placed in the UGATE trace between the controller and upper MOSGET gate is recommended to reduce the ringing on the PHASE node by slowing down the upper MOSFET turn-on. A gate resistor value between 2Ω to 10Ω typically reduces the PHASE to GND peak ringing below 17V. Gate Drive Voltage Versatility The HIP6601 and HIP6603 provide the user total flexibility in choosing the gate drive voltage. The HIP6601 lower gate drive is fixed to VCC [+12V], but the upper drive rail can range from 12V down to 5V depending on what voltage is applied to PVCC. The HIP6603 ties the upper and lower drive rails together. Simply applying a voltage from 5V up to 12V on PVCC will set both driver rail voltages. Power Dissipation Package power dissipation is mainly a function of the switching frequency and total gate charge of the selected MOSFETs. Calculating the power dissipation in the driver for a desired application is critical to ensuring safe operation. Exceeding the maximum allowable power dissipation level will push the IC beyond the maximum recommended operating junction temperature of 125oC. The maximum allowable IC power dissipation for the SO8 package is approximately 800mW. When designing the driver into an application, it is recommended that the following calculation HIP6601, HIP6603 be performed to ensure safe operation at the desired frequency for the selected MOSFETs. The power dissipated by the driver is approximated as: 3 P = 1.05f sw --- V U Q + V L Q + I DDQ V 2 L CC U Test Circuit +5V OR +12V +12V where fsw is the switching frequency of the PWM signal. VU and VL represent the upper and lower gate rail voltage. QU and QL is the upper and lower gate charge determined by MOSFET selection and any external capacitance added to the gate pins. The IDDQ VCC product is the quiescent power of the driver and is typically 30mW. UGATE Figure 2 shows the dissipation in the driver with 3nF loading on both gates and each individually. Note the higher upper gate power dissipation which is due to the bootstrap device refresh cycle. Again PVCC and VCC are tied together and to a +12V supply. 6 HIP660X VCC 0.15µF CU PHASE LGATE PWM 100kΩ 2N7002 CL GND 1000 PVCC = VCC = 12V POWER (mW) 800 CU = CL = 3nF 600 400 CU = CL = 1nF CU = CL = 2nF 200 CU = CL = 4nF CU = CL = 5nF 0 500 1000 1500 2000 FREQUENCY (kHz) where QLOSS is the total charge removed from the bootstrap capacitor and provided to the upper gate load. FIGURE 1. POWER DISSIPATION vs FREQUENCY 1000 PVCC = VCC = 12V 800 CU = CL = 3nF POWER (mW) In Figure 1, CU and CL values are the same and frequency is varied from 50kHz to 2MHz. PVCC and VCC are tied together to a +12V supply. Curves do exceed the 800mW cutoff, but continuous operation above this point is not recommended. 2N7002 0.15µF 1 1 V = --- f SW Q V P REFRESH = --- f SW Q LOSS PVCC U U 2 2 The 1.05 factor is a correction factor derived from the following characterization. The base circuit for characterizing the drivers for different loading profiles and frequencies is provided. CU and CL are the upper and lower gate load capacitors. Decoupling capacitors [0.15µF] are added to the PVCC and VCC pins. The bootstrap capacitor value is 0.01µF. BOOT PVCC The power dissipation approximation is a result of power transferred to and from the upper and lower gates. But, the internal bootstrap device also dissipates power on-chip during the refresh cycle. Expressing this power in terms of the upper MOSFET total gate charge is explained below. The bootstrap device conducts when the lower MOSFET or it’s body diode conducts and pulls the PHASE node toward GND. While the bootstrap device conducts, a current path is formed that refreshes the bootstrap capacitor. Since the upper gate is driving a MOSFET, the charge removed from the bootstrap capacitor is equivalent to the total gate charge of the MOSFET. Therefore, the refresh power required by the bootstrap capacitor is equivalent to the power used to charge the gate capacitance of the MOSFET. 0.01µF 600 CU = 3nF CL = 3nF 400 200 0 500 1000 1500 2000 FREQUENCY (kHz) FIGURE 2. 3nF LOADING PROFILE The impact of loading on power dissipation is shown in Figure 3. Frequency is held constant while the gate capacitors are varied from 1nF to 5nF. VCC and PVCC are tied together and to a +12V supply. Figures 4 through 6 show the same characterization for the HIP6603 with a +5V supply on PVCC and VCC tied to a +12V supply. HIP6601, HIP6603 Since both upper and lower gate capacitance can vary, Figure 7 shows dissipation curves versus lower gate capacitance with upper gate capacitance held constant at three different values. These curves apply only to the HIP6601 due to power supply configuration. 600 400 PVCC = VCC = 12V FREQUENCY = 800kHz PVCC = 5V VCC = 12V 320 POWER (mW) POWER (mW) 500 400 FREQUENCY = 500kHz 300 CU = CL = 5nF CU = CL = 4nF 240 CU = CL = 3nF 160 200 CU = CL = 2nF CU = CL = 1nF 80 FREQUENCY = 200kHz 100 1.0 2.0 3.0 4.0 0 5.0 500 GATE CAPACITANCE (CU = CL), (nF) 1000 FIGURE 3. POWER DISSIPATION vs LOADING 250 PVCC = 5V VCC = 12V PVCC = 5V VCC = 12V POWER (mW) CU = 3nF 120 FREQUENCY = 800kHz 150 100 FREQUENCY = 500kHz 50 CL = 1nF 60 0 500 1000 FREQUENCY = 200kHz 1500 0 1.0 2000 FIGURE 5. 3nF LOADING PROFILE (HIP6603) 400 2.0 3.0 FIGURE 6. VARIABLE LOADING PROFILE (HIP6603) CU = 5nF PVCC = 5V VCC = 12V 350 300 CU = 3nF 200 CU = 1nF 150 100 1.0 2.0 3.0 4.0 5.0 FREQUENCY (kHz) FIGURE 7. POWER DISSIPATION vs LOADING (HIP6601) 7 4.0 GATE CAPACITANCE (CU = CL), (nF) FREQUENCY (kHz) POWER (mW) POWER (mW) 200 CU = CL = 3nF 180 2000 FIGURE 4. POWER DISSIPATION vs FREQUENCY (HIP6603) 300 240 1500 FREQUENCY (kHz) 5.0 HIP6601, HIP6603 Small Outline Plastic Packages (SOIC) M8.15 (JEDEC MS-012-AA ISSUE C) N INDEX AREA 0.25(0.010) M H 8 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE B M E INCHES -B- 1 2 SYMBOL 3 L SEATING PLANE -A- h x 45o A D -C- e µα A1 B 0.25(0.010) M C C A M B S 1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95. MILLIMETERS MIN MAX NOTES A 0.0532 0.0688 1.35 1.75 - 0.0040 0.0098 0.10 0.25 - B 0.013 0.020 0.33 0.51 9 C 0.0075 0.0098 0.19 0.25 - D 0.1890 0.1968 4.80 5.00 3 E 0.1497 0.1574 3.80 4.00 4 0.050 BSC 1.27 BSC - H 0.2284 0.2440 5.80 6.20 - h 0.0099 0.0196 0.25 0.50 5 L 0.016 0.050 0.40 1.27 6 8o 0o N NOTES: MAX A1 e 0.10(0.004) MIN α 8 0o 8 7 8o Rev. 0 12/93 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension “D” does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. “L” is the length of terminal for soldering to a substrate. 7. “N” is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch). 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification. Intersil semiconductor products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see web site www.intersil.com 8