NS ES I G D W NT R NE D F O LA C EM E E D MEN ED REP b. 2004) COM D E R Data ENSheet le Fe NOT ECOMM (Availab R 232 I S L6 IPM6220 TM Advanced Triple PWM and Dual Linear Power Controller for Portable Applications The IPM6220 provides a highly integrated power control and protection solution for five output voltages required in highperformance notebook PC applications. The IC integrates three fixed frequency pulse-width-modulation (PWM) controllers and two linear regulators along with monitoring and protection circuitry into a single 24 lead SSOP package. The two PWM controllers that regulate the system main 5V and 3.3V voltages are implemented with synchronousrectified buck converters. Synchronous rectification and hysteretic operation at light loads contribute to high efficiency over a wide range of input voltage and load variation. Efficiency is further enhanced by using the lower MOSFET’s rDS(ON) as the current sense element. Input voltage feedforward ramp modulation, current-mode control, and internal feed-back compensation provide fast and stable handling of input voltage load transients encountered in advanced portable computer chip sets. December 2000 FN4903.1 Features • Provides Five Regulated Voltages - +5V ALWAYS - +3.3V ALWAYS - +5V Main - +3.3V Main - +12V • High Efficiency Over Wide Line and Load Range - Synchronous Buck Converters on Main Outputs - Hysteretic Operation at Light Load • No Current-Sense Resistor Required - Uses MOSFET’s rDS(ON) - Optional Current-Sense Resistor for More Precision • Operates Directly From Battery 5.6 to 22V Input • Input Undervoltage Lock-Out (UVLO) • Excellent Dynamic Response - Voltage Feed-Forward and Current-Mode Control • Monitors Output Voltages The third PWM controller is a boost converter that regulates a resistor selectable output voltage of nominally 12V. Two internal linear regulators provide +5V ALWAYS and +3.3V ALWAYS low current outputs required by the notebook system controller. TEMP. RANGE (oC) IPM6220CA 0 to 70 IPM6220EVAL1 • Separate Shut-Down Pins for Advanced Configuration and Power Interface (ACPI) Compatibility • 300kHz Fixed Switching Frequency on Main Outputs • Thermal Shut-Down Protection Ordering Information PART NUMBER • Synchronous Converters Operate Out of Phase PKG. NO. PACKAGE 24 Ld SSOP M24.15 Applications • Mobile PCs • Hand-Held Portable Instruments Evaluation Board Related Literature Pinout • Application Note AN9915 IPM6220 (SSOP) TOP VIEW 24 BOOT1 VBATT 1 3.3V ALWAYS 2 23 UGATE1 BOOT2 3 22 PHASE1 UGATE2 4 21 ISEN1 PHASE2 5 20 LGATE1 5V ALWAYS 6 19 PGND1 LGATE2 7 18 VSEN1 PGND2 8 17 SDWN1 ISEN2 9 16 GATE3 VSEN2 10 15 VSEN3 SDWN2 11 14 GND PGOOD 12 13 SDWNALL 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Intersil and Design is a trademark of Intersil Corporation. | Copyright © Intersil Corporation 2000 Block Diagram VSEN3 VBATT GATE3 BOOT1 GND UVFLT GATE LOGIC 1 HGDR1 UGATE1 HI BOOST CONTROLLER CLK1 CLK 200ns RAMP 2 CLK2 CLK1 PHASE1 SHUTOFF POWER-ON RESET (POR) RAMP 1 PWMMD1 POR DEADTIME 2 REF VCC PWM/HYST BOOT2 LGDR1 PWM ON UGATE2 HYST ON GATE LOGIC 2 HGDR2 OVP1 HI PHASE2 SHUTOFF - + - VOLTSECOND CLAMP + PWMMD1 R1 = 20K + + - VBATT - + D Q R > Q REF VOLTSECOND CLAMP ∑ - POR SDWN1 LDO1 REFERENCE VCC + UVFLT LGATE2 LGATE1 LGATE1 SDWN LGATE2 VSEN1 + + PWM MODE 2 PWMMD2 R1 = 20K ISEN1 - VCC PWM LATCH 2 OC LOGIC2 + - + - PWM MODE 1 OC COMP2 2.8V EA2 ISEN2 ∑ + - CLK2 REF HYST COMP2 VSEN2 IPM6220 HYST ON EA1 + LO OVP2 - OC LOGIC1 Q D R Q < PWM ON CLK1 + LGDR2 VSEN1 + PWM VCC LATCH 1 DEADTIME PWM/HYST VCC PGND2 PGND1 HYST COMP1 PWMMD2 OC COMP1 LGATE2 LGATE1 LO OVP1 OVP2 SDWN OUTPUT VOLTAGE MONITOR REF 3.3V-ALWAYS PGOOD FIGURE 1. AND SDWN2 SOFT START LDO2 5V-ALWAYS 2.5V SDWNALL IPM6220 Simplified Power System Diagram VBATT VBATT Q1 3.3V ALWAYS 5V ALWAYS LINEAR CONTROLLER 5V MAIN PWM1 CONTROLLER Q2 LINEAR CONTROLLER IPM6220 VBATT VOLTAGE, CURRENT MONITORS PGOOD 12V BOOST Q1 3.3V MAIN PWM2 CONTROLLER PWM3 CONTROLLER Q2 FIGURE 2. Typical Application +VBATT PROCESSOR 5V MAIN SDWN1 VCORE µP CORE 3.3V MAIN SDWN2 IPM6220 VI/O 5V ALWAYS 3.3V ALWAYS C8051 12V VID CODE SDWN PCM CIA VCLOCK ON/OFF FIGURE 3. I/O CLOCK PGOOD PGOOD ENABLE SDWNALL 3 IPM6210 RESET IPM6220 I Absolute Maximum Ratings Thermal Information Input Voltage, VBATT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +29.0V Phase, ISEN and SDWNALL Pins . . . . . . . . . . . GND -0.3V to +29.0V Boot and UGATE Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +29.0V BOOT1, 2 with Respect to PHASE1, 2 . . . . . . . . . . . . . . . . . . . +6.5V All Other Pins. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +6.5V Thermal Resistance (Typical, Note 1) θJA (oC/W) θJC (oC/W) SSOP Package . . . . . . . . . . . . . . . . . . 88 28.5 Maximum Junction Temperature (Plastic Package) . . . . . . . .150oC Maximum Storage Temperature Range . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . .300oC (SSOP - Lead Tips Only) Operating Conditions Input Voltage, VBATT . . . . . . . . . . . . . . . . . . . . . . . . +5.6V to +22.0V Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . . 0oC to 70oC Junction Temperature Range . . . . . . . . . . . . . . . . . . . 0oC to 125oC CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTE: 1. θJA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief TB379 for details. Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System Diagrams, and Typical Application Schematic PARAMETER SYMBOL Input Quiescent Current TEST CONDITIONS MIN TYP MAX UNITS 2.0 mA ICC SDWN1 = SDWN2 = 5V, SDWNALL = VIN, Outputs open circuited - 1.4 Stand-by Current ICCSB SDWN1 = SDWN2 = 0V, SDWNALL = VIN, Outputs open circuited - 300 µA Shut-down Current ICCSN SDWNALL = 0V - <1.0 µA Input Under-voltage Lock Out UVLO Rising VBATT 4.3 4.7 Input Under-voltage Lock Out UVLO VBATT, Hysteresis 5.1 150 V mV OSCILLATOR PWM1,2 Oscillator Frequency Fc1,2 255 300 345 kHz VREF - 2.472 - V -1.0 - +1.0 % - 5 - µA REFERENCE AND SOFT START Internal Reference Voltage Reference Voltage Accuracy SDWN1, SDWN2 Output Current During Start-up ISS PWM1 CONVERTER, 5V Main Output Voltage VOUT1 Line and Load Regulation 5.0 V 0.0 < IVOUT1 < 5.0A; 5.6V < VBATT < 22.0V -2 0.5 +2 % Under-Voltage Shut-Down Level VUV1 2µs delay, % Feedback Voltage at VSNS1 pin 70 75 80 % Current Limit Threshold IOC2 Current from ISNS1 Pin Through RSNS1 90 135 180 µA Over-Voltage Threshold VOVP1 2µs delay, % Feedback Voltage at VSNS1 pin 110 115 120 % Maximum Duty Cycle DCMAX SDWN1 > 4.0V 94 % 3.3 V PWM2 CONVERTER, 3.3V Main Output Voltage VOUT2 Line and Load Regulation 0.0 < IVOUT2 < 5.0A; 5.6V < VBATT < 24.0V -2 0.5 +2 % Under-Voltage Shut-Down Level VUV2 2µs delay, % Feedback Voltage at VSNS2 pin 70 75 80 % Current Limit Threshold IOC2 Current from ISNS2 Pin Through RSNS2 90 135 180 µA Over-Voltage Threshold VOVP2 2µs delay, % Feedback Voltage at VSNS2 pin 110 115 120 % Maximum Duty Cycle DCMAX SDWN2 > 4.0V 4 94 % IPM6220 Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted. Refer to Block and Simplified Power System Diagrams, and Typical Application Schematic (Continued) PARAMETER SYMBOL Internal Resistance to GND on VSNS2 Pin TEST CONDITIONS MIN RVSNS2 TYP MAX UNITS Ω 66K PWM1 and PWM2 CONTROLLER GATE DRIVERS Upper Drive Pull-Up Resistance R2UGPUP - 7 12 Ω Upper Drive Pull-Down Resistance R2UGPDN - 4 10 Ω Lower Drive Pull-Up Resistance R2LGPUP - 6 9 Ω Lower Drive Pull-Down Resistance R2LGPDN - 5 8 Ω PWM 3 CONVERTER 12V Feedback Regulation Voltage VSEN3 2.472 12V Feedback Regulation Voltage Input Current IVSEN3 0.1 Line and Load Regulation Under-Voltage Shut-Down Level Over-Voltage Threshold PWM3 Oscillator Frequency 0.0 < IVOUT3 < 120mA, 4.9V< 5VMain <5.1V -2 VUV3 2µs delay, % Feedback Voltage at VSNS3 pin 70 VOVP3 2µs delay, % Feedback Voltage at VSNS3 pin Fc3 85 Maximum Duty Cycle V 1.0 µA +2 % 75 80 % 115 120 % 100 115 kHz 33 % PWM 3 CONTROLLER GATE DRIVERS Pull-Up Resistance R3GPUP 6 12 Ω Pull-Down Resistance R3GPDN 6 12 Ω 5V and 3.3V ALWAYS Linear Regulator Accuracy PWM1, 5V Output OFF (SDWN1 = 0V); 5.6V < VBATT < 22V; 0 < ILOAD < 50mA -2.0 0.5 +2.0 % 5V ALWAYS Output Voltage Regulation PWM1, 5V Output ON (SDWN1 = 5V); 0 < ILOAD < 50mA -3.3 1.0 +2.0 % Maximum Output Current Combined 5V ALWAYS and 3.3V ALWAYS 50 Current Limit Combined 5V ALWAYS and 3.3V ALWAYS 100 5V ALWAYS Under-Voltage Shut-Down Bypass Switch rDS(ON) PWM1, 5V Output ON (SDWN1 = 5V) mA 180 mA 75 % 1.3 Ω POWER GOOD AND CONTROL FUNCTIONS Power Good Threshold for PWM1 and PWM2 Output Voltages PGOOD Leakage Current IPGLKG PGOOD Voltage Low VPGOOD VPULLUP = 5.0V IPGOOD = -4mA -14 -12 -10 % - - 1.0 µA 0.2 0.5 V 10 µs SDWN1, 2, - Low (Off) 0.8 V SDWN1, 2, - High (On) 4.3 V SDWNALL - High (On) 2.4 V 40 mV Over-Temperature Shutdown 150 oC Over-Temperature Hysteresis 25 oC PGOOD Minimum Pulse Width TPGmin SDWNALL - Low (Off) SDWNALL, Hysteresis 5 IPM6220 Functional Pin Descriptions VBATT (Pin 1) Supplies all the power necessary to operate the chip. The IC starts to operate when the voltage on this pin exceeds 4.7V and stops operating when the voltage on this pin drops below approximately 4.5V. Also provides battery voltage to the oscillator for feed-forward rejection to input voltage variations. 3.3V ALWAYS (Pin 2) Output of 3.3V ALWAYS linear regulator. 5V ALWAYS (Pin 6) Output of 5V ALWAYS linear regulator or the +5V Main output. If the +5V Main output is enabled, it is switched internally from the VSEN1 pin to the 5V ALWAYS output. This improves efficiency and reduces the power dissipation in the controller. BOOT1, BOOT2 (Pins 24 and 3) controllers. The PGOOD, overvoltage protection (OVP) and undervoltage shutdown circuits use these signals to determine output-voltage status and/or to initiate undervoltage shut down. The VSEN1 input is also switched internally to the 5V ALWAYS output if the +5V Main output is enabled. SDWNALL (Pin 13) This pin provides enable/disable function for all outputs. The chip is completely disabled when this pin is pulled to ground. When this pin is pulled high, the 5V ALWAYS and 3.3 ALWAYS outputs are on and the other outputs are enabled. The state of 5V Main and 3.3V Main outputs depend on the voltage on SDWN1 and SDWN2 respectively. See Table 1. SDWN1 (Pin 17) This pin provides enable/disable function and soft-start for the PWM1, 5V Main, output. The output is enabled when this pin is high and SDWNALL is also high. The 5V output is held off when the pin is pulled to the ground. SDWN2 (Pin 11) Power is supplied to the upper MOSFET drivers of PWM1 and PWM2 converters via the BOOT pins. Connect these pins to the respective junctions of bootstrap capacitors with the cathodes of the bootstrap diodes. Anodes of the bootstrap diodes are connected to pin 6, 5V ALWAYS. This pin provides enable/disable function and soft-start for PWM2, 3.3V Main, output. The output is enabled when this pin is high and SDWNALL is also high. The 3.3V output is held off when the pin is pulled to the ground. UGATE1, UGATE2 (Pins 23 and 4) VSEN3 (Pin 15) These pins provide the gate drive for the upper MOSFETs. Connect UGATE pins to the respective PWM converter’s upper MOSFET gate. This input pin is the voltage feedback signal for PWM3, the boost controller. The boost controller regulates this point to a voltage divided level of 2.472 VDC. The PGOOD, overvoltage protection (OVP) and undervoltage shutdown circuits use this signal to determine output-voltage status and/or to initiate undervoltage shut down. PHASE1, PHASE2 (Pins 22 and 5) The phase nodes are the junctions of the upper MOSFET sources, output filter inductors, and lower MOSFET drains. Connect the PHASE pins directly to the respective PWM converter’s lower MOSFET drain. ISEN1, ISEN2 (Pins 21 and 9) These pins are used to monitor the voltage drop across the lower MOSFETs for current feedback and current-limit protection. For more precise current detection, these inputs can be connected to optional current sense resistors placed in series with the sources of the lower MOSFETs. This pin can also be used to independently disable the PWM3 controller. Connect this pin to 5V ALWAYS if the boost converter is not populated in your design. GATE3 (Pin 16) This pin drives the gate of the boost MOSFET. PGOOD (Pin 12) These pins provide the gate drive for the lower MOSFETs. Connect the lower MOSFET gate of each converter to the corresponding pin. PGOOD is an open drain output used to indicate the status of the PWM converters’ output voltages. This pin is pulled low when any of the outputs except PWM3 (12V) is not within -10% of respective nominal voltages, or when PWM3 (12V) is not within its undervoltage and overvoltage thresholds. PGND1, PGND2 (Pins 19 and 8) GND (Pin 14) These are the lower MOSFET gate drive return connection for PWM1 and PWM2 converters, respectively. Tie each lower MOSFET source directly to the corresponding pin. Signal ground for the IC. All voltage levels are measured with respect to this pin. LGATE1, LGATE 2 (Pins 20 and 7) VSEN1, VSEN2 (Pins 18, 10) These pins are connected to the main outputs and provide the voltage feedback signal for the respective PWM 6 General Description The IPM6220 addresses the system electronics power needs of modern notebook and sub-notebook PCs. The IC integrates control circuits for two synchronous buck IPM6220 converters for 5V Main and 3.3V Main buses, two linear regulators for 3.3V ALWAYS and 5V ALWAYS, and a 12V boost converter. The two synchronous converters operate out of phase to substantially reduce the input-current ripple, minimizing input filter requirements, minimizing battery heating and prolonging battery life. The 12V boost controller uses a 100kHz clock derived from the main clock. This controller uses leading edge modulation with the maximum duty cycle limited to 33%. Rcs Vo RSNS = ------------------ Iocdc + ----------------------------------------- – 100 135µA L × 2 × 300kHz VIN = 10.8V IL3.3V (2A/DIV.) 5A 3.3V PHASE (10V/DIV.) 0 A, V 5A The chip has three input control lines SDWN1, SDWN2 and SDWNALL. These are provided for Advanced Configuration and Power Interface (ACPI) compatibility. They turn on and off all outputs, as well as provide independent control of the 3.3V Main and +5V Main outputs. To maximize efficiency for the 5V Main and 3.3V Main outputs, the current-sense technique is based on the lower MOSFET rDS(ON). Light-load efficiency is further enhanced by a hysteretic mode of operation which is automatically engaged at light loads when the inductor current becomes discontinuous. 3.3V Main and 5V Main Architecture These main outputs are generated from the unregulated battery input by two independent synchronous buck converters. The IC integrates all the components required for output voltage setpoint and feedback compensation, significantly reducing the number of external components, saving board space and parts cost. The buck PWM controllers employ a 300kHz fixed frequency current-mode control scheme with input voltage feedforward ramp programming for better rejection of input voltage variations. Figure 4 shows the out-of-phase operation for the 3.3V Main and 5V Main outputs. The phase node is the junction of the upper MOSFET, lower MOSFET and the output inductor. The phase node is high when the upper MOSFET is conducting and the inductor current rises accordingly. When the phase node is low, the lower MOSFET is conducting and the inductor current is ramping down as shown. Current Sensing and Current Limit Protection Both PWM converters use the lower MOSFET on-state resistance, rDS(ON), as the current-sensing element. This technique eliminates the need for a current sense resistor and the associated power losses. If more accurate current protection is desired, current sense resistors may be used in series with the lower MOSFETs’ source. To set the current limit, place a resistor, RSNS, between the ISEN inputs and the drain of the lower MOSFET (or optional current sense resistor). The required value of the RSNS resistor is determined from the following equation: 7 IL5V (2A/DIV.) 0 A, V 5V PHASE (10V/DIV.) 1µs/DIV. FIGURE 4. OUT OF PHASE OPERATION where IOCDC is the desired DC overcurrent limit; RCS is either the rDS(ON) of the lower MOSFET, or the value of the optional current-sense resistor, Vo is the output voltage and L is the output inductor. Also, the value of RCS should be specified for the expected maximum operating temperature. The sensed voltage, and the resulting current out of the ISEN pin through RSNS, is used for current feedback and current limit protection. This is compared with an internal current limit threshold. When a sampled value of the output current is determined to be above the current limit threshold, the PWM drive is terminated and a counter is initiated. This limits the inductor current build-up and essentially switches the converter into current-limit mode. If an overcurrent is detected between 26µs to 53µs later, an overcurrent shutdown is initiated. If during the 26µs to 53µs period, an overcurrent is not detected, the counter is reset and sampling continues as normal. This current limit scheme has proven to be very robust in applications like portable computers where fast inductor current build-up is common due to a large difference between input and output voltages and a low value of the inductor. Light-Load (Hysteretic) Operation In the light-load (hysteretic) mode the output voltage is regulated by the hysteretic comparator which regulates the output voltage by maintaining the output voltage ripple as shown in Figure 5. In Hysteretic mode, the inductor current flows only when the output voltage reaches the lower limit of the hysteretic comparator and turns off at the upper limit. Hysteretic mode saves converter energy at light loads by supplying energy only at the time when the output voltage requires it. This mode conserves energy by reducing the power dissipation associated with continuous switching. IPM6220 During the time between inductor current pulses, both the upper and lower MOSFETs are turned off. This is referred to as ‘diode emulation mode’ because the lower MOSFET performs the function of a diode. This diode emulation mode prevents the output capacitor from discharging through the lower MOSFET when the upper MOSFET is not conducting. The gate drive is synchronized to the main clock, so the outof-phase timing is maintained in hysteretic mode. Such a scheme insures a seamless transition between the operational modes. VOUT t IL t PHASE COMP HYSTERETIC PWM The other mechanism for changing from hysteretic to PWM is due to a sudden increase in the output current. This step load causes an instantaneous decrease in the output voltage due to the voltage drop on the output capacitor ESR. If the decrease causes the output voltage to drop below the hysteretic regulation level, the mode is changed to PWM on the next clock cycle. This insures the full power required by the increase in output current. t 1 2 3 4 5 6 7 8 MODE OF OPERATION The mode change from hysteretic to PWM can be caused by one of two events. One event is the same mechanism that causes a PWM to hysteretic transition. But instead of looking for eight consecutive positive occurrences on the phase node, it is looking for eight consecutive negative occurrences on the phase node. The operation mode will be changed from hysteretic to PWM when these eight consecutive pulses occur. This transition technique prevents jitter of the operation mode at load levels close to boundary. IL t t FIGURE 5. REGULATION IN HYSTERETIC MODE Operation-Mode Control The mode-control circuit changes the converter’s mode of operation based on the voltage polarity of the phase node when the lower MOSFET is conducting and just before the upper MOSFET turns on. For continuous inductor current, the phase node is negative when the lower MOSFET is conducting and the converters operate in fixed-frequency PWM mode as shown in Figure 6. When the load current decreases to the point where the inductor current flow through the lower MOSFET in the ‘reverse’ direction, the phase node becomes positive, and the mode is changed to hysteretic. A phase comparator handles the timing of the phase node voltage sensing. A low level on the phase comparator output indicates a negative phase voltage during the conduction time of the lower MOSFET. A high level on the phase comparator output indicates a positive phase voltage. When the phase node is positive (phase comparator high), at the end of the lower MOSFET conduction time, for eight consecutive clock cycles, the mode is changed to hysteretic as shown in Figure 6. The dashed lines indicate when the phase node goes positive and the phase comparator output goes high. The solid vertical lines at 1,2,...8 indicate the sampling time, of the phase comparator, to determine the polarity (sign) of the phase node. At the transition between PWM and hysteretic mode both the upper and lower MOSFETs are turned off. The phase node will ‘ring’ based on the output inductor and the parasitic capacitance on the phase node and settle out at the value of the output voltage. 8 PHASE NODE t 1 2 3 4 5 6 7 8 PHASE COMP t MODE OF OPERATION PWM HYSTERETIC t FIGURE 6. MODE CONTROL WAVEFORMS Gate Control Logic The gate control logic translates generated PWM control signals into the MOSFET gate drive signals providing necessary amplification, level shifting and shoot-through protection. Also, it has functions that help optimize the IC performance over a wide range of operational conditions. Since MOSFET switching time can vary dramatically from type to type and with the input voltage, the gate control logic provides adaptive dead time by monitoring the gate-tosource voltages of both upper and lower MOSFETs. The lower MOSFET is not turned on until the gate-to-source voltage of the upper MOSFET has decreased to less than approximately 1 volt. Similarly, the upper MOSFET is not turned on until the gate-to-source voltage of the lower MOSFET has decreased to less than approximately 1 volt. This allows a wide variety of upper and lower MOSFETs to be used without a concern for simultaneous conduction, or shoot-through. IPM6220 3.3V Main and 5V Main Soft Start, Sequencing and Stand-by See Table 1 for the output voltage control algorithm. The 5V Main and 3.3V Main converters are enabled if SDWN1 and SDWN2 are high and SDWNALL is also high. The stand-by mode is defined as a condition when SDWN1 and SDWN2 are low and the PWM converters are disabled but SDWNALL is high (3.3V ALWAYS and 5V ALWAYS outputs are enabled). In this power saving mode, only the low power micro-controller and keyboard may be powered. TABLE 1. OUTPUT VOLTAGE CONTROL SDWN2 3V AND 5V ALWAYS 5V MAIN 3V MAIN 0 X X OFF OFF OFF 1 0 0 ON OFF OFF 1 1 0 ON ON OFF 1 0 1 ON OFF ON 1 1 1 ON ON ON Soft start of the 3.3V Main and 5V Main converters is accomplished by means of capacitors connected from pins SDWN1 and SDWN2 to ground. In conjunction with 5µA internal current sources, they provide a controlled rise of the 3.3V Main and 5V Main output voltages. The value of the soft-start capacitors can be calculated from the following expression. 5µA × Tss Css = ---------------------------3.5V 12V Converter Architecture The 12V boost converter generates its output voltage from the 5V Main output. An external MOSFET, inductor, diode and capacitor are required to complete the circuit. The output signal is fed back to the controller via an external resistive divider. The boost controller can be disabled by connecting the VSEN3 pin to 5V ALWAYS. The control circuit for the 12V converter consists of a 3:1 frequency divider which drives a ramp generator and resets a PWM latch as shown in Figure 8. The width of the CLK/3 pulses is equal to the period of the main clock, limiting the duty cycle to 33%. The output of a non-inverting error amplifier is compared with the rising ramp voltage. When the ramp voltage becomes higher than the error signal, the PWM comparator sets the latch and the output of the gate driver is pulled high providing leading edge, voltage mode PWM. The falling edge of the CLK/3 pulses resets the latch and pulls the output of the gate driver low. VSEN3 By varying the values of the soft-start capacitors, it is possible to provide sequencing of the main outputs at start-up. Figure 7 shows the soft-start initiated by the SDWNALL pin being pulled high with the Vbatt input at 10.8V and the resulting 3.3V Main and 5V Main outputs. + Where Tss is the desired soft-start time. - REF EA3 GATE3 PWM PWM COMPARATOR LATCH 3 S Q RAMP R Q + SDWNALL SDWN1 and 5V ALWAYS outputs. With the 3.3V Main and 5V Main outputs enabled, at T1, the internal 5µA current sources start charging the soft start capacitors on the SDWN1 and SDWN2 pins. At T2 the outputs begin to rise and because they both have the same value of soft-start capacitors, 0.022µF, they both reach regulation at the same time, T3. The soft-start capacitors continue to charge and are completely charged at T4. CLK/3 CLK DIVIDER 3:1 CLK/3 RAMP GENERATOR CLK t VIN = 10.8V CLK/3 SDWNALL,10V/DIV. t RAMP VEA3 SDWN2, 2V/DIV. t 3.3VOUT , 2V/DIV. GATE3 t 0V FIGURE 8. 12V BOOST OPERATION SDWN1, 2V/DIV. The 33% maximum duty cycle of the converter guarantees discontinuous inductor current and unconditional stability over all operating conditions. 5VOUT , 2V/DIV. 0V 4ms/DIV. T0 T1 T2 T3 T4 FIGURE 7. SOFT START ON 3.3V AND 5V OUTPUTS While the SDWNALL pin is held low, prior to T0, all outputs are off. Pulling SDWNALL high enables the 3.3V ALWAYS 9 The boost converter with the limited duty cycle and discontinuous inductor current can deliver to the load a limited amount of power before the output voltage starts to drop. When the duty cycle has reached DMAX, the control IPM6220 loop is operating open circuit and the output voltage varies with the output load resistance, Ro, as given by: Ro Vo = Vin × Dmax ------------------- 2 ( LxF ) Where Vin is the 5V Main voltage, Dmax = 0.33, L is the value of the boost inductor, L3, and F = 100kHz. This provides automatic output current limiting. When the maximum duty cycle has been reached and for a given inductor, a further reduction in Ro by one-half will pull the output voltage down to 0.707 of nominal and cause an under-voltage condition. The 12V converter starts to operate at the same time as the 5V Main converter. The rising voltage on the 5V Main output and the 33% duty cycle limit provides a similar soft-start, as the 5V Main, for the 12V output. MOSFET is turned off and the lower MOSFET is turned on. This ‘soft-crowbar’ condition will be maintained until the output voltage returns to the regulation window and then normal operation will continue. This ‘soft-crowbar’ and monitoring of the output, prevents the output voltage from ringing negative as the inductor current flows in the ‘reverse’ direction through the lower MOSFET and output capacitors. Over-Temperature Protection The IC incorporates an over-temperature protection circuit that shuts all the outputs down when the die temperature exceeds 150oC. Normal operation is automatically restored when the die temperature cools to 125oC. Component Selection Guidelines Output Capacitor Selection 3V ALWAYS, 5V ALWAYS Linear Regulators The 3.3V ALWAYS and 5V ALWAYS outputs are derived from the battery voltage and are the first voltages available in the notebook when power on is initiated. The 5V ALWAYS output is generated directly from the battery voltage by a linear regulator. It is used to power the system microcontroller and to internally power the chip and the gate drivers. The 3.3V ALWAYS output is generated from the 5V ALWAYS output and may be used to power the keyboard controller or other peripherals. The combined current capability of these outputs is 50mA. When the 5V Main output is greater than it’s undervoltage level, it is switched to the 5V ALWAYS output via an internal 1.3Ω MOSFET switch. Simultaneously, the 5V ALWAYS linear regulator is disabled to prevent excessive power dissipation. The rise time of the 5V ALWAYS is determined by the value of the output capacitance on the 5V and 3.3V ALWAYS outputs. The internal regulator is current limited to about 180mA, so the start up time is approximately: 5V t = C OUT × ------------------180mA Where COUT is the sum of the capacitances on the 5V and 3.3V ALWAYS outputs. Power Good Status The IPM6220 monitors all the output voltages except for the 3.3V ALWAYS. A single power-good signal, PGOOD, is issued when soft-start is completed and all monitored outputs are within 10% of their respective set points. After the soft-start sequence is completed, undervoltage protection latches the chip off when any of the monitored outputs drop below 75% of its set point. A ‘soft-crowbar’ function is implemented for an overvoltage on the 3.3V Main or 5V Main outputs. If the output voltage goes above 115% of their nominal output level, the upper 10 The output capacitors for each output have unique requirements. In general, the output capacitors should be selected to meet the dynamic regulation requirements including ripple voltage and load transients. 3.3V Main and 5V Main PWM Output Capacitors Selection of the output capacitors is also dependent on the output inductor so some inductor analysis is required to select the output capacitors. One of the parameters limiting the converter’s response to a load transient is the time required for the inductor current to slew to it’s new level. Given a sufficiently fast control loop design, the IPM6220 will provide either 0% or 94% duty cycle in response to a load transient. The response time is the time interval required to slew the inductor current from an initial current value to the load current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor(s). Minimizing the response time can minimize the output capacitance required. Also, if the load transient rise time is slower than the inductor response time, as in a hard drive or CD drive, this reduces the requirement on the output capacitor. The maximum capacitor value required to provide the full, rising step, transient load current during the response time of the inductor is: I TRAN L O × I TRAN C OUT = ---------------------------------------------- × -------------------( V IN – V OUT ) × 2 DV OUT Where: COUT is the output capacitor(s) required, LO is the output inductor, ITRAN is the transient load current step, VIN is the input voltage, VOUT is output voltage, and ∆VOUT is the drop in output voltage allowed during the load transient. High frequency capacitors initially supply the transient current and slow the load rate-of-change seen by the bulk capacitors. The bulk filter capacitor values are generally IPM6220 determined by the ESR (Equivalent Series Resistance) and voltage rating requirements as well as actual capacitance requirements. The output voltage ripple is due to the inductor ripple current and the ESR of the output capacitors as defined by: expression is given in the capacitor selection section and the ripple current is approximated by the following equation: V IN – V OUT V OUT ∆I L = -------------------------------- × ---------------V IN FS × L Input Capacitor Selection where, ∆IL is calculated in the Inductor Selection section. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load circuitry for specific decoupling requirements. Use only specialized low-ESR capacitors intended for switching-regulator applications, at 300kHz, for the bulk capacitors. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. The stability requirement on the selection of the output capacitor is that the ‘ESR zero’, fZ, be between 1.2kHz and 30kHz. This range is set by an internal, single compensation zero at 6kHz. The ESR zero can be a factor of five on either side of the internal zero and still contribute to increased phase margin of the control loop. Therefore: 1 C OUT = ------------------------------------------2 × π × ESR × f Z In conclusion, the output capacitors must meet three criteria: By varying the values of the soft-start capacitors, it is possible to provide sequencing of the main outputs at start-up. 1. They must have sufficient bulk capacitance to sustain the output voltage during a load transient while the output inductor current is slewing to the value of the load transient 2. The ESR must be sufficiently low to meet the desired output voltage ripple due to the output inductor current, and 3. The ESR zero should be placed, in a rather large range, to provide additional phase margin. 3.3V ALWAYS and 5V ALWAYS Output Capacitors The output capacitors for the linear regulators insure stability and provide dynamic load current. The 3.3V ALWAYS and the 5V ALWAYS linear regulators should have, as a minimum, 10µF capacitors on their outputs. 3.3V Main and 5V Main PWM Output Inductor Selection The PWM converters require output inductors. The output inductor is selected to meet the output voltage ripple requirements. The inductor value determines the converter’s ripple current and the ripple voltage is a function of the ripple current and output capacitor(s) ESR. The ripple voltage 11 The important parameters for the bulk input capacitor(s) are the voltage rating and the RMS current rating. For reliable operation, select bulk input capacitors with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and 1.5 times is a conservative guideline. The AC RMS input current varies with load as shown in Figure 9. Depending on the specifics of the input power and it’s impedance, most (or all) of this current is supplied by the input capacitor(s). Figure 9 also shows the advantage of having the PWM converters operating out of phase. If the converters were operating in phase, the combined RMS current would be the algebraic sum, which is a much larger value as shown. The combined out-of-phase current is the square root of the sum of the square of the individual reflected currents and is significantly less than the combined in-phase current. INPUT RMS CURRENT V RIPPLE = ∆I L × ESR 5 4.5 4 3.5 3 2.5 2 1.5 1 0.5 0 IN PHASE OUT OF PHASE 5V 3.3V 0 1 2 3 3.3V AND 5V LOAD CURRENT 4 5 FIGURE 9. INPUT RMS CURRENT vs LOAD Use a mix of input bypass capacitors to control the voltage ripple across the MOSFETs. Use ceramic capacitors for the high frequency decoupling and bulk capacitors to supply the RMS current. Small ceramic capacitors can be placed very close to the upper MOSFET to suppress the voltage induced in the parasitic circuit impedances. For board designs that allow through-hole components, the Sanyo OS-CON® series offer low ESR and good temperature performance. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. The TPS series available from AVX is surge current tested. OS-CON® is a registered trademark of Sanyo Electric Company, Ltd. (Japan) IPM6220 +12V Boost Converter Inductor Selection The inductor value is chosen to provide the required output power to the load. Vinmin 2 × Dmax 2 × Ro Lmax = ---------------------------------------------------------------2 × Vo 2 × F where, Vinmin is the minimum input voltage, 4.9V; Dmax = 1/3, the maximum duty cycle; Ro is the minimum load resistance; Vo is the nominal output voltage and F is the switching frequency, 100kHz. +12V Boost Converter Output Capacitor Selection The total capacitance on the 12V output should be chosen appropriately, so that the output voltage will be higher than the undervoltage limit (9V) when the 5V Main soft-start time has elapsed. This will avoid triggering of the 12V undervoltage protection. The maximum value of the boost capacitor, Comax that will charge to 9V in the soft start time, Tss , is shown below, where L is the value of the boost inductor. Tss Comax = ---------- × 0.115µF L The output capacitor ESR and the boost inductor ripple current determines the output voltage ripple. The ripple voltage is given by: V RIPPLE = ∆I L × ESR and the maximum ripple current, ∆IL, is given by: 5V ∆I L = ------- × 3.3 µ L where L is the boost inductor calculated above, 5V is the boost input voltage and 3.3µ is the maximum on time for the boost MOSFET. MOSFET Considerations The logic level MOSFETs are chosen for optimum efficiency given the potentially wide input voltage range and output power requirements. Two N-channel MOSFETs are used in each of the synchronous-rectified buck converters for the PWM1 and PWM2 outputs. These MOSFETs should be selected based upon rDS(ON) , gate supply requirements, and thermal management considerations. The power dissipation includes two loss components; conduction loss and switching loss. These losses are distributed between the upper and lower MOSFETs according to duty cycle (see the following equations). The conduction losses are the main component of power dissipation for the lower MOSFETs. Only the upper MOSFET has significant switching losses, since the lower device turns on and off into near zero voltage. The equations assume linear voltage-current transitions and do not model power loss due to the reverse-recovery of the lower MOSFET’s body diode. The gate-charge losses are 12 2 I O × r DS ( ON ) × V OUT I O × V IN × t SW × F S P UPPER = ----------------------------------------------------------- + ---------------------------------------------------V IN 2 2 I O × r DS ( ON ) × ( V IN – V OUT ) P LOWER = -------------------------------------------------------------------------------V IN dissipated by the IPM6220 and do not heat the MOSFETs. However, a large gate-charge increases the switching time, tSW which increases the upper MOSFET switching losses. Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermal-resistance specifications. Layout Considerations MOSFETs switch very fast and efficiently. The speed with which the current transitions from one device to another causes voltage spikes across the interconnecting impedances and parasitic circuit elements. The voltage spikes can degrade efficiency, radiate noise into the circuit, and lead to device overvoltage stress. Careful component layout and printed circuit design minimizes the voltage spikes in the converter. Consider, as an example, the turn-off transition of one of the upper PWM MOSFETs. Prior to turn-off, the upper MOSFET is carrying the full load current. During the turn-off, current stops flowing in the upper MOSFET and is picked up by the lower MOSFET. Any inductance in the switched current path generates a voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide circuit traces minimize the magnitude of voltage spikes. See the Application Note AN9915 for the evaluation board component placement and the printed circuit board layout details. There are two sets of critical components in a DC-DC converter using an IPM6220 controller. The switching power components are the most critical because they switch large amounts of energy, and as such, they tend to generate equally large amounts of noise. The critical small signal components are those connected to sensitive nodes or those supplying critical bias currents. Power Components Layout Considerations The power components and the controller IC should be placed first. Locate the input capacitors, especially the highfrequency ceramic decoupling capacitors, close to the power MOSFETs. Locate the output inductor and output capacitors between the MOSFETs and the load. Locate the PWM controller close to the MOSFETs. Insure the current paths from the input capacitors to the MOSFETs, to the output inductors and output capacitors are as short as possible with maximum allowable trace widths. A multi-layer printed circuit board is recommended. Dedicate one solid layer for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. The IPM6220 power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes, but do not unnecessarily oversize these particular islands. Since the phase nodes are subjected to very high dV/dt voltages, the stray capacitor formed between these islands and the surrounding circuitry will tend to couple switching noise. Use the remaining printed circuit layers for small signal wiring. The wiring traces from the control IC to the MOSFET gate and source should be sized to carry 2A peak currents. This filter configuration may be helpful on both the 3.3V and 5V Main outputs. RSNS = R19 + R9 ISEN1 R9 200 FROM PHASE NODE C12 100pF FIGURE 10. NOISE FILTER FOR ISEN1 INPUT 6. The bypass capacitors for VBATT and the soft-start capacitors, CSS1 and CSS2 should be located close to their connecting pins on the control IC. Minimize any leakage current paths from SDWN1 and SDWN2 nodes, since the internal current source is only 5µA. Small Components Signal Layout Considerations 4. The VSNS1 and VSNS2 inputs should be bypassed with a 1.0µF capacitor close to their respective IC pins. 5. A ‘T’ filter consisting of a ‘split’ RSNS and a small, 100pF, capacitor as shown in Figure 10, may be helpful in reducing noise coupling into the ISEN input. For example, if the calculated value of RSNS1 is 2.2kΩ, dividing it as shown with a 100pF capacitor provides filtering without changing the current limit set point. For any calculated value of RSNS, keep the value of the R9 portion to approximately 200Ω, and the remainder of the resistance in the R19 position. The 200Ω resistor and 100pF capacitor provide effective filtering for noise above 8MHz. +5.6-22VIN R19 2K 7. Refer to the Application Note AN9915 for a recommended component placement and interconnections. Figure 11 shows an application circuit of a power supply for a notebook PC microprocessor system. The power supply provides +5V ALWAYS, +3.3V ALWAYS, +5.0V, +3.3V, and 12V from +5.6-22VDC battery voltage. For detailed information on the circuit, including a Bill of Materials and circuit board description, see Application Note AN9915. Also see Intersil’s web site (www.intersil.com) for the latest information. C4 56µF C3, 6, 10 3x1µF D2 BAT54WT1 GND VBATT 1 +3.3V ALWAYS (50mA) 3.3V ALWAYS + +5V ALWAYS (50mA) + 5V ALWAYS C1 100µF Q2 HUF76112SK8 UGATE2 L1 R10, 11 ISEN2 C22 330µF 22 20 19 4 Q4 HUF76112SK8 LGATE2 PGND2 VSEN2 SDWN2 C16 0.022µF PGOOD L4 2.7µH ISEN1 R9, 19 2.2K LGATE1 PGND1 +5V (5A) L2 8.2µH + Q5 HUF76112SK8 L3 6.8µH + C21, 32 2x330µF C36 22µF 5 18 9 VSEN1 16 7 GATE3 +12V (120mA) D3 Q5 HUF76112SK8 + R14 C24, 33 2x47µF 97.6K 8 R13 24.9K VSEN3 10 15 11 17 SDWN1 C17 0.022µF 12 14 13 SDWNALL GND FIGURE 11. APPLICATIONS CIRCUIT 13 C9 0.15µF PHASE1 IPM6220 3 Q3 HUF76112SK8 UGATE1 2.2K 8.2µH + 6 21 PHASE2 +3.3V (5A) 23 D1 BAT54WT1 BOOT2 C7 0.15µF BOOT1 24 2 C2 10µF IPM6220 Shrink Small Outline Plastic Packages (SSOP) M24.15 N INDEX AREA H 0.25(0.010) M 24 LEAD THIN SHRINK NARROW BODY SMALL OUTLINE PLASTIC PACKAGE B M E 1 2 INCHES GAUGE PLANE -B3 L 0.25 0.010 SEATING PLANE -A- h x 45o A D -C- α e B 0.17(0.007) M A2 A1 C 0.10(0.004) C A M B S NOTES: 1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95. 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension “D” does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. “L” is the length of terminal for soldering to a substrate. 7. “N” is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. Dimension “B” does not include dambar protrusion. Allowable dambar protrusion shall be 0.10mm (0.004 inch) total in excess of “B” dimension at maximum material condition. 10. Controlling dimension: INCHES. Converted millimeter dimensions are not necessarily exact. 14 MILLIMETERS SYMBOL MIN MAX MIN MAX NOTES A 0.053 0.069 1.35 1.75 - A1 0.004 0.010 0.10 0.25 - A2 - 0.061 - 1.54 - B 0.008 0.012 0.20 0.30 9 C 0.007 0.010 0.18 0.25 - D 0.337 0.344 8.55 8.74 3 E 0.150 0.157 3.81 3.98 4 e 0.025 BSC 0.635 BSC - H 0.228 0.244 5.80 6.19 - h 0.0099 0.0196 0.26 0.49 5 L 0.016 0.050 0.41 1.27 6 N α 24 0o 24 8o 0o 7 8o Rev. 0 12/00