LT1302/LT1302-5 Micropower High Output Current Step-Up Adjustable and Fixed 5V DC/DC Converters U DESCRIPTIO FEATURES ■ ■ ■ ■ ■ ■ ■ ■ 5V at 600mA or 12V at 120mA from 2-Cell Supply 200µA Quiescent Current Logic Controlled Shutdown to 15µA Low VCESAT Switch: 310mV at 2A Typical Burst ModeTM Operation at Light Load Current Mode Operation for Excellent Line and Load Transient Response Available in 8-Lead SO or PDIP Operates with Supply Voltage as Low as 2V UO APPLICATI ■ ■ ■ ■ The internal low loss NPN power switch can handle current in excess of 2A and switch at frequencies up to 400kHz. Quiescent current is just 200µA and can be further reduced to 15µA in shutdown. Available in 8-pin PDIP or 8-pin SO packaging, the LT1302/ LT1302-5 have the highest switch current rating of any similarly packaged switching regulators presently on the market. Notebook and Palmtop Computers Portable Instruments Personal Digital Assistants Cellular Telephones Flash Memory , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. UO ■ S The LT®1302/LT1302-5 are micropower step-up DC/DC converters that maintain high efficiency over a wide range of output current. They operate from a supply voltage as low as 2V and feature automatic shifting between Burst Mode operation at light load, and current mode operation at heavy load. TYPICAL APPLICATI 6 7 2 CELLS + C1 100µF C3 0.1µF D1 VIN SW SHDN PGND GND 1 + C2 100µF IT LT1302-5 8 2-Cell to 5V Converter Efficiency NC 5 SENSE VC 90 3 88 SHUTDOWN 2 RC 20k CC 0.01µF VIN = 3V 86 4 EFFICIENCY (%) L1 10µH 84 VIN = 2.5V 82 VIN = 2V 80 78 76 74 72 OUTPUT 5V 600mA LT1302 • F01 C1 = C2 = SANYO OS-CON L1 = COILTRONICS CTX10-3 COILCRAFT DO3316-103 D1 = MOTOROLA MBRS130LT3 70 1 10 100 LOAD CURRENT (mA) 1000 LT1302 • TA02 Figure 1. 2-Cell to 5V/600mA DC/DC Converter 1 LT1302/LT1302-5 U U RATI GS W W W W AXI U U ABSOLUTE PACKAGE/ORDER I FOR ATIO VIN Voltage ............................................................. 10V SW Voltage ............................................................. 25V FB Voltage .............................................................. 10V SHDN Voltage ......................................................... 10V VC Voltage ................................................................ 4V IT Voltage .................................................................. 4V Maximum Power Dissipation ............................ 700mW Operating Temperature Range .................... 0°C to 70°C Storage Temperature Range ............... – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C TOP VIEW GND 1 8 PGND VC 2 7 SW SHDN 3 6 VIN (SENSE*)FB 4 5 IT N8 PACKAGE S8 PACKAGE 8-LEAD PDIP 8-LEAD PLASTIC SO *FIXED VERSION PINS 1 AND 8 ARE INTERNALLY CONNECTED IN SOIC PACKAGE ORDER PART NUMBER LT1302CN8 LT1302CS8 LT1302CN8-5 LT1302CS8-5 S8 PART MARKING 1302 13025 TJMAX = 125°C, θJA = 100°C/W (N8) TJMAX = 125°C, θJA = 80°C/W (S8) Consult factory for Industrial and Military grade parts. DC ELECTRICAL CHARACTERISTICS SYMBOL IQ PARAMETER Quiescent Current VIN Input Voltage Range TA = 25°C, VIN = 2.5V, unless otherwise noted. CONDITIONS VSHDN = 0.5V, VFB = 1.3V VSHDN = 1.8V MIN ● ● ● 2.0 2.2 1.22 ● 4.85 ● 175 160 75 ● VFB VOS DC tON tOFF VCESAT Feedback Voltage (LT1302) Feedback Pin Bias Current (LT1302) Output Sense Voltage (LT1302-5) Output Ripple Voltage (LT1302-5) Sense Pin Resistance to Ground (LT1302-5) Offset Voltage Comparator Hysteresis Oscillator Frequency Maximum Duty Cycle Switch On Time Switch Off Time Output Line Regulation Switch Saturation Voltage VC = 0.4V VFB = 1V VC = 0.4V VC = 0.4V See Block Diagram (Note 1) Current Limit Not Asserted (Note 2) Current Limit Not Asserted 2 < VIN < 8V ISW = 2A ● TYP 200 15 1.24 100 5.05 50 420 15 5 220 86 3.9 0.7 0.06 310 ● Switch Leakage Current Switch Current Limit VSHDNH VSHDNL ISHDN Error Amplifier Voltage Gain Shutdown Pin High Shutdown Pin Low Shutdown Pin Bias Current VSW = 5V, Switch Off VC = 0.4V (Burst Mode Operation) VC = 1.25V (Full Power) (Note 3) 0.9V ≤ VC ≤ 1.2V, ∆VC/∆VFB ● ● 2.0 50 1.8 0.1 1 2.8 75 ● VSHDN = 5V VSHDN = 2V VSHDN = 0V I T Pin Resistance to Ground The ● denotes specifications which apply over the 0°C to 70°C temperature range. Note 1: Hysteresis is specified at DC. Output ripple depends on capacitor size and ESR. 2 ● ● ● 8 3 0.1 3.9 MAX 300 25 8 1.26 5.25 265 310 95 0.15 400 475 10 3.9 0.5 20 1 UNITS µA µA V V V nA V mV kΩ mV mV kHz kHZ % µs µs %/V mV mV µA A A V/ V V V µA µA µA kΩ Note 2: The LT1302 operates in a variable frequency mode. Switching frequency depends on load inductance and operating conditions and may be above specified limits. Note 3: Minimum switch current 100% tested. Maximum switch current guaranteed by design. LT1302/LT1302-5 U W TYPICAL PERFORMANCE CHARACTERISTICS No-Load Quiescent Current Circuit of Figure 1 Switch Saturation Voltage 400 TA = 25°C TA = 25°C 450 ISW = 2A 400 350 SATURATION VOLTAGE (mV) 500 400 VCESAT (V) QUIESCENT CURRENT (µA) Switch Saturation Voltage 600 500 300 250 200 300 200 150 100 100 350 300 250 200 150 50 0 2.0 0 2.5 3.0 3.5 4.0 SUPPLY VOLTAGE (V) 4.5 5.0 0 1 2 3 SWITCH CURRENT (A) 100 –50 4 –25 0 25 50 TEMPERATURE (°C) 75 100 1302 G02 1302 G03 1302 G01 LT1302 Feedback Voltage LT1302-5 Sense Pin Resistance 1.250 Quiescent Current 600 300 500 250 VIN = 2.5V SWITCH OFF SENSE RESISTANCE (kΩ) FEEDBACK VOLTAGE (V) 1.240 1.235 1.230 1.225 1.220 1.215 1.210 QUIESCENT CURRENT (µA) 1.245 400 300 200 200 150 100 50 100 1.205 1.200 –50 –25 0 25 50 TEMPERATURE (°C) 75 0 –50 100 –25 0 25 50 TEMPERATURE (°C) 1302 G04 Error Amplifier Offset Voltage LT1302-5 Output Voltage 10 5 0 25 50 TEMPERATURE (°C) 75 100 1302 G07 100 4.5 5.050 5.025 5.000 4.975 4.0 3.5 3.0 4.950 2.5 4.925 –25 75 Maximum On-Time ON-TIME (µs) OUTPUT VOLTAGE (V) 15 0 25 50 TEMPERATURE (°C) 5.0 5.075 20 –25 1302 G06 5.100 25 OFFSET VOLTAGE (mV) 0 –50 100 1302 G05 30 0 –50 75 4.900 –50 –25 0 25 50 TEMPERATURE (°C) 75 100 1302 G08 2.0 –50 –25 0 25 50 TEMPERATURE (°C) 75 100 1302 G09 3 LT1302/LT1302-5 U W TYPICAL PERFORMANCE CHARACTERISTICS Shutdown Pin Bias Current Oscillator Frequency Maximum Duty Cycle 100 300 20 TA = 25°C 18 FREQUENCY (kHz) DUTY CYCLE (%) 90 80 70 60 SHUTDOWN CURRENT (µA) 275 250 225 200 175 16 14 12 10 8 6 4 2 50 –50 –25 0 25 50 TEMPERATURE (°C) 75 100 150 –50 –25 75 0 25 50 TEMPERATURE (°C) 1302 G10 0 100 0 1 6 4 2 3 5 SHUTDOWN VOLTAGE (V) 1302 G11 LT1302-5 Output Voltage vs Load Current 7 8 1302 G12 Maximum Output Power* Boost Mode 5.20 20 16 5.10 OUTPUT POWER (W) OUTPUT VOLTAGE (V) 5.15 5.05 5.00 VIN = 4V 4.95 VIN = 2.2V VIN = 3V 12 8 4.90 4 4.85 4.80 0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 LOAD CURRENT (A) 1302 G13 0 2 6 4 INPUT VOLTAGE (V) * APPROXIMATE 8 10 1302 G14 U U U PI FU CTIO S GND (Pin 1): Signal Ground. Feedback resistor and 0.1µF ceramic bypass capacitor from VIN should be connected directly to this pin. VC (Pin 2): Frequency Compensation Pin. Connect series RC to GND. Keep trace short. SHDN (Pin 3): Shutdown. Pull high to effect shutdown; tie to ground for normal operation. FB/Sense (Pin 4): Feedback/Sense. On the LT1302 this pin connects to CMP1 input. On the LT1302-5 this pin connects to the output resistor string. 4 IT (Pin 5): Normally left floating. Addition of a 3.3k resistor to GND forces the LT1302 into current mode at light loads. Efficiency drops at light load but increases at medium loads. See Applications Information section. VIN (Pin 6): Supply Pin. Must be bypassed with: (1) a 0.1µF ceramic to GND, and (2) a large value electrolytic to PGND. When VIN is greater than 5V, a low value resistor (2Ω to 10Ω) is recommended to isolate the VIN pin from input supply noise. LT1302/LT1302-5 U U U PI FU CTIO S SW (Pin 7): Switch Pin. Connect inductor and diode here. Keep layout short and direct. and 8 are thermally connected to the die. One square inch of PCB copper provides an adequate heat sink for the device. PGND (Pin 8): Power Ground. Pins 8 and 1 should be connected under the package. In the SO package, pins 1 W BLOCK DIAGRA SM D1 L1 VIN C1 + 6 C3 7 VIN 36mV R5 730Ω A2 – 1.24V REFERENCE + VOS 15mV R2 220kHz OSCILLATOR Q5 SHUTDOWN Q4 160X DRIVER VIN Q1 A1 SHDN Q3 VIN – 3 A3 HYSTERETIC COMPARATOR 2µA FB 4 C5 100pF OFF ENABLE – R1 SW R4 1.75Ω + CMP1 VOUT + C2 0.1µF Q2 BIAS + ERROR AMPLIFIER 300Ω 1 GND 2 VC R3 22k C4 0.01µF 5 IT 3.6k 8 PGND 1302 F02 Figure 2. LT1302 Block Diagram 5 LT1302/LT1302-5 W BLOCK DIAGRA SM SENSE VIN SW 4 6 7 36mV R4 1.75Ω + R5 730Ω A2 R1 315k – 1.24V REFERENCE CMP1 + OFF ENABLE – 3 Q5 – SHUTDOWN Q3 Q4 160X DRIVER VIN VIN Q1 A1 SHDN A3 HYSTERETIC COMPARATOR 2µA VOS 15mV R2 105k 220kHz OSCILLATOR Q2 BIAS + ERROR AMPLIFIER 300Ω 3.6k 1 2 5 8 GND VC IT PGND 1302 F03 Figure 3. LT1302-5 Block Diagram OPERATIO U The LT1302’s operation can best be understood by examining the block diagram in Figure 2. The LT1302 operates in one of two modes, depending on load. With light loads, comparator CMP1 controls the output; with heavy loads, control is passed to error amplifier A1. Burst Mode operation consists of monitoring the FB pin voltage with hysteretic comparator CMP1. When the FB voltage, related to the output voltage by external attenuator R1 and R2, falls below the 1.24V reference voltage, the oscillator is enabled. Switch Q4 alternately turns on, causing current buildup in inductor L1, then turns off, allowing the built-up current to flow into output capacitor C3 via D1. As the output voltage increases, so does the FB voltage; when it exceeds the reference plus 6 CMP1’s hysteresis (about 5mV) CMP1 turns the oscillator off. In this mode, peak switch current is limited to approximately 1A by A2, Q2, and Q3. Q2’s current, set at 34µA, flows through R5, causing A2’s negative input to be 25mV lower than VIN. This node must fall more than 36mV below VIN for A2 to trip and turn off the oscillator. The remaining 11mV is generated by Q3’s current flowing through R4. Emitter-area scaling sets Q3’s collector current to 0.625% of switch Q4’s current. When Q4’s current is 1A, Q3’s current is 6.25mA, creating an 11mV drop across R4 which, added to R5’s 25mV drop, is enough to trip A2. When the output load is increased to the point where the 1A peak current cannot support the output voltage, LT1302/LT1302-5 OPERATIO U CMP1 stays on and the peak switch current is regulated by the voltage on the VC pin (A1’s output). VC drives the base of Q1. As the VC voltage rises, Q2 conducts less current, resulting in less drop across R5. Q4’s peak current must then increase in order for A2 to trip. This current mode control results in good stability and immunity to input voltage variations. Because this is a linear, closed-loop system, frequency compensation is required. A series RC from VC to ground provides the necessary pole-zero combination. The LT1302-5 incorporates feedback resistors R1 and R2 into the device. Output voltage is set at 5.05V in Burst Mode, dropping to 4.97V in current mode. U W U U APPLICATIONS INFORMATION Inductor Selection Inductors used with the LT1302 must fulfill two requirements. First, the inductor must be able to handle current of 2.5A to 3A without runaway saturation. Rod or drum core units usually saturate gradually and it is acceptable to exceed manufacturers’ published saturation currents by 20% or so. Second, it should have low DCR, under 0.05Ω so that copper loss is kept low. Inductance value is not critical. Generally, for low voltage inputs down to 2V, a 10µH inductor is recommended (such as Coilcraft DO3316103). For inputs above 4V to 5V use a 22µH unit (such as Coilcraft DO3316-223). Switching frequency can reach up to 400kHz so the core material should be able to handle high frequency without loss. Ferrite or molypermalloy cores are a better choice than powdered iron. If EMI is a concern a toroidal inductor is suggested, such as Coiltronics CTX20-4. For a boost converter, duty cycle can be calculated by the following formula: V DC = 1– IN VOUT A special situation exists where the VOUT/VIN differential is high, such as a 2V-to-12V converter. The required duty cycle is higher than the LT1302 can provide, so the converter must be designed for discontinuous operation. This means that inductor current goes to zero during the switch off-time. In the 2V-to-12V case, inductance must be low enough so that current in the inductor can reach 2A in a single cycle. Inductor value can be defined by: L≤ (V IN − VSW )× t ON 2A With the 2V input a value of 3.3µH is acceptable. Since the inductance is so low, usually a smaller core size can be used. Efficiency will not be as high as for the continuous case since peak currents will necessarily be higher. Table 1 lists inductor suppliers along with appropriate part numbers. Table 1. Recommended Inductors VENDOR Coilcraft Coiltronics Dale Sumida PART NO. DO3316-103 DO3316-153 DO3316-223 CTX10-2 CTX20-4 LPT4545-100LA LPT4545-200LA CD105-100 CD105-150 CDR125-220 VALUE(µH) 10 15 22 10 20 10 20 10 15 22 PHONE NO. (708) 639-6400 (407) 241-7876 (605) 665-9301 (708) 956-0666 Capacitor Selection The output capacitor should have low ESR for proper performance. A high ESR capacitor can result in “modehopping” between current mode and Burst Mode at high load currents because the output voltage will increase by ISW × ESR when the inductor current is flowing into the diode. Figure 4 shows output voltage of an LT1302-5 boost converter with two 220µF AVX TPS capacitors at the output. Ripple voltage at a 510mA load is about 30mVP-P 7 LT1302/LT1302-5 U W U U APPLICATIONS INFORMATION and there is no low frequency component. The total ESR is under 0.03Ω. If a single 100µF aluminum electrolytic capacitor is used instead, the converter mode-hops between current mode and Burst Mode due to high ESR, causing the voltage comparator to trip as shown in Figure 5. The ripple voltage is now over 500mVP-P and contains a low frequency component. Maximum allowable output capacitor ESR can be calculated by the following formula: ESRMAX = VOS × VOUT VREF × 1A where, VOS = 15mV VREF = 1.24V Input Capacitor The input supply should be decoupled with a good quality electrolytic capacitor close to the LT1302 to provide a stable input supply. Long leads or traces from power source to the switcher can have considerable impedance at the LT1302’s switching frequency. The input capacitor provides a low impedance at high frequency. A 0.1µF ceramic capacitor is required right at the VIN pin. When the input voltage can be above 5V, a 10Ω/1µF decoupling network for VIN is recommended as detailed in Figure 6. This network is also recommended when driving a transformer. VIN > 5V 10Ω VIN + VOUT 50mV/DIV AC COUPLED 47µF TO 100µF SW + 1µF LT1302 GND • • • PGND 1302 F06 ILOAD 510mA 10mA Figure 6. A 10Ω/1µF Decoupling Network at VIN Is Recommended When Input Voltage Is Above 5V 500µs/DIV 1302 F04 Figure 4. Low ESR Output Capacitor Results in Stable Operation. Ripple Voltage is Under 30mVP-P SERIES TPS OS-CON 595D TYPE Surface Mount Through Hole Surface Mount PHONE NO. (803) 448-9411 (619) 661-6835 (603) 224-1961 Diode Selection 510mA 10mA 500µs/DIV 1302 F05 Figure 5. Inexpensive Electrolytic Capacitor Has High ESR, Resulting in Mode-Hop, Ripple Voltage Amplitude Is Over 500mVP-P and Includes Low Frequency Component 8 Table 2. Recommended Capacitors VENDOR AVX Sanyo Sprague VOUT 200mV/DIV AC COUPLED ILOAD Table 2 lists capacitor vendors along with device types. A 2A Schottky diode such as Motorola MBRS130LT3 has been found to be the best available. Other choices include 1N5821 or MBRS130T3. Do not use “general purpose” diodes such as 1N4001. They are much too slow for use in switching regulator applications. LT1302/LT1302-5 U U W U APPLICATIONS INFORMATION Frequency Compensation Obtaining proper RC values for the frequency compensation network is largely an empirical procedure, since variations in input and output voltage, topology, capacitor ESR and inductance make a simple formula elusive. As an example, consider the case of a 2.5V to 5V boost converter supplying 500mA. To determine optimum compensation, the circuit is built and a transient load is applied to the circuit. Figure 7 shows the setup. In Figure 7a, the VC pin is simply left floating. Although output voltage is maintained and transient response is good, switch current rises instantaneously to the internal current limit upon application of load. This is an undesirable situation as it places maximum stress on the switch and the other power components. Additionally, efficiency is well down from its optimal value. Next, a 0.1µF capacitor is connected with no resistor. Figure 7b details response. Although the circuit eventually stabilizes, the loop is quite underdamped. Initial output “sag” exceeds 5%. Aberrant VIN 2.5V VIN D1 0.1µF C2 220µF + The VC pin is sensitive to high frequency noise. Some layouts may inject enough noise to modulate peak switch current at 1/2 the switching frequency. A small capacitor connected from VC to ground will eliminate this. Do not exceed 1/10 of the compensation capacitor value. C3 220µF IT LT1302-5 PGND GND + Finally, a 0.01µF/24k series network results in the response shown in Figure 7f. This has optimal damping, undershoot less than 100mV and settles in less than 1ms. SHDN SW C1 330µF In Figure 7c, the 0.1µF capacitor has been replaced by a 0.01µF unit. Undershoot is less but the response is still underdamped. Figure 7d shows the results of the 0.1µF capacitor and a 10k resistor in series. Now some amount of damping is observed, and behavior is more controlled. Figure 7e details response with a 0.01µF/10k series network. Undershoot is down to around 100mV, or 2%. A slight underdamping is still noticeable. NC L1 10µH + behavior in the 4th graticule is the result of the LT1302’s Burst Mode comparator turning off all switching as output voltage rises above its threshold. SENSE VC 500Ω 10Ω 2W R PULSE GENERATOR C MTP3055EL C1, C2, C3 = AVX TPS SERIES D1 = MOTOROLA MBRS130LT3 L1 = COILCRAFT DO3316-103K 50Ω 1302 F07 Figure 7. Boost Converter with Simulated Load VOUT 100mV/DIV AC COUPLED VOUT 100mV/DIV AC COUPLED 510mA 510mA ILOAD ILOAD 10mA 2ms/DIV 1302 F07a Figure 7a. VC Pin Left Unconnected. Output Shows Low Frequency Components Under Load 10mA 2ms/DIV 1302 F07b Figure 7b. 0.1µF from VC to Ground. Better, but More Improvement Needed 9 LT1302/LT1302-5 U U W U APPLICATIONS INFORMATION IT Pin VOUT 100mV/DIV AC COUPLED ILOAD The IT pin is used to disable Burst Mode, forcing the LT1302 to operate in current mode even at light load. To disable Burst Mode, 3.3k resistor R1 is connected from IT to gound. More conservative frequency compensation must be used when in this mode. A 0.1µF capacitor and 4.7k resistor from VC to ground has been found to be adequate. Low frequency Burst Mode ripple can be reduced or eliminated using this technique in many applications. 510mA 10mA 2ms/DIV 1302 F07c Figure 7c. 0.01µF from VC to Ground. Underdamped Response Requires Series R To illustrate, the transient load response of Figure 8’s circuit is pictured without and with R1. Figure 8a shows output voltage and inductor current without the resistor. Note the 6kHz burst rate when the converter is delivering 25mA. By adding the 3.3k resistor, the low frequency bursting is eliminated, as shown in Figure 8b. This feature is useful in systems that contain audio circuitry. At very light or zero load, switching frequency drops and eventu- VOUT 100mV/DIV AC COUPLED 510mA ILOAD 10mA 2ms/DIV 1302 F07d Figure 7d. 0.1µF with 10k Series RC. Classic Overdamped Response VIN 2.5V 10µH + VOUT 100mV/DIV AC COUPLED VIN C1 330µF MBRS130LT3 + 220µF 10V 0.1µF + LT1302-5 PGND GND 220µF 10V 510mA ILOAD SENSE SW VC IT 4.7k R1 3.3k 0.1µF 10mA 1302 F08 2ms/DIV 1302 F07e Figure 7e. 0.01µF, 10k Series RC Shows Good Transient Response. Slight Underdamping Still Noticeable VOUT 100mV/DIV AC COUPLED ILOAD Figure 8. Addition of R1 Eliminates Low Frequency Output Ripple in This 2.5V to 5V Boost Converter VOUT 100mV/DIV AC COUPLED INDUCTOR CURRENT 1A/DIV 510mA 10mA ILOAD 525mA 25mA 2ms/DIV 1302 F07f Figure 7f. 0.01µF, 24k Series RC Results in Optimum Response 10 VOUT 5V 600mA 1ms/DIV 1302 F08a Figure 8a. IT Pin Floating. Note 6kHz Burst Rate at ILOAD = 25mA. 0.1µF/4.7k Compensation Network Causes 220mV Undershoot LT1302/LT1302-5 U U W U APPLICATIONS INFORMATION ally reaches audio frequencies, but at a much lighter load than without the IT feature. At some input voltage/load current combinations, some residual bursting may occur at frequencies out of the audio band. Figure 8c details efficiency with and without the addition of R1. Burst Mode operation keeps efficiency high at light load with IT floating. Efficiency falls off at light load with R1 added because the LT1302 cannot transition into Burst Mode. VOUT 100mV/DIV AC COUPLED INDUCTOR CURRENT 1A/DIV Layout 525mA ILOAD 25mA 1ms/DIV 1302 F08b Figure 8b. 3.3k Resistor from IT Pin to Ground Forces LT1302 into Current Mode Regardless of Load. Audio Frequency Component Eliminated 90 IT FLOATING 80 EFFICIENCY (%) The IT pin cannot be used as a soft-start. Large capacitors connected to the pin will cause erratic operation. If operating the device in Burst Mode, let the pin float. Keep high dV/dt signals away from the pin. 70 3.3k IT TO GND 60 50 40 30 1 10 100 OUTPUT CURRENT (mA) 1000 1302 F08c Figure 8c. 3.3k Resistor for IT to Ground Increases Efficiency at Moderate Load, Decreases at Light Load The high speed, high current switching associated with the LT1302 mandates careful attention to layout. Follow the suggested component placement in Figure 9 for proper operation. High current functions are separated by the package from sensitive control functions. Feedback resistors R1 and R2 should be close to the feedback pin (pin4). Noise can easily be coupled into this pin if care is not taken. A small capacitor (100pF to 200pF) from FB to ground provides a high frequency bypass. If the LT1302 is operated off a three-cell or higher input, R3 (2Ω to 10Ω) in series with VIN is recommended. This isolates the device from noise spikes on the input supply. Do not put in R3 if the device must operate from a 2V input, as input current will cause the voltage at the LT1302’s VIN pin to go below 2V. The 0.1µF ceramic bypass capacitor C3 (use X7R, not Z5U) should be mounted as close as possible to the package. When R3 is used, C3 should be a 1µF tantalum unit. Grounding should be segregated as illustrated. C3’s ground trace should not carry switch current. Run a VIN R2 C3 R3 2Ω L1 + 4 3 6 LT1302 C1 D1 5 7 2 8 1 R1 200pF SHUTDOWN RC CC + C2 VOUT GND (BATTERY AND LOAD RETURN) 1302 F09 Figure 9. Suggested Component Placement for LT1302 11 LT1302/LT1302-5 U W U U APPLICATIONS INFORMATION separate ground trace up under the package as shown. The battery and load return should go to the power side of the ground copper. Thermal Considerations The LT1302 contains a thermal shutdown feature which protects against excessive internal (junction) temperature. If the junction temperature of the device exceeds the protection threshold, the device will begin cycling between normal operation and an off state. The cycling is not harmful to the part. The thermal cycling occurs at a slow rate, typically 10ms to several seconds, which depends on the power dissipation and the thermal time constants of the package and heat sinking. Raising the ambient temperature until the device begins thermal shutdown gives a good indication of how much margin there is in the thermal design. For surface mount devices heat sinking is accomplished by using the heat spreading capabilities of the PC board and its copper traces. Experiments have shown that the heat spreading copper layer does not need to be electrically connected to the tab of the device. The PCB material can be very effective at transmitting heat between the pad area attached to pins 1 and 8 of the device, and a ground or power plane layer either inside or on the opposite side of the board. Although the actual thermal resistance of the PCB material is high, the length/area ratio of the thermal resistance between the layer is small. Copper board stiffeners and plated through holes can also be used to spread the heat generated by the device. Table 3 lists thermal resistance for the SO package. Measured values of thermal resistance for several different board sizes and copper areas are listed for each surface mount package. All measurements were taken in still air on 3/32" FR-4 board with 1oz copper. This data can be used as a rough guideline in estimating thermal resistance. The thermal resistance for each application will be affected by thermal interactions with other components as well as board size and shape. 12 Table 3. S8 Package, 8-Lead Plastic SO COPPER AREA TOPSIDE* BACKSIDE BOARD AREA THERMAL RESISTANCE (JUNCTION-TO-AMBIENT) 2500 sq. mm 2500 sq. mm 2500 sq. mm 60°C/W 1000 sq. mm 2500 sq. mm 2500 sq. mm 62°C/W 225 sq. mm 2500 sq. mm 2500 sq. mm 65°C/W 100 sq. mm 2500 sq. mm 2500 sq. mm 69°C/W 100 sq. mm 1000 sq. mm 2500 sq. mm 73°C/W 100 sq. mm 225 sq. mm 2500 sq. mm 80°C/W 100 sq. mm 100 sq. mm 2500 sq. mm 83°C/W * Pins 1 and 8 attached to topside copper N8 Package, 8-Lead DIP: Thermal Resistance (Junction-to-Ambient) = 100°C/W Calculating Temperature Rise Power dissipation internal to the LT1302 in a boost regulator configuration is approximately equal to: 2 V V 2 OUT + VD OUT + VD − PD = IOUT R IOUT VOUT R V − IOUT VOUT R VIN − IN VIN VIN + ( ) IOUT VOUT + VD − VIN 27 The first term in this equation is due to switch “onresistance.” The second term is from the switch driver. R is switch resistance, typically 0.15Ω. VD is the diode forward drop. The temperature rise can be calculated from: ∆T = PD × θJA where: ∆T = Temperature Rise PD = Device Power Dissipation θJA = Thermal Resistance (Junction-to-Ambient) LT1302/LT1302-5 U W U U APPLICATIONS INFORMATION As an example, consider a boost converter with the following specifications: VIN = 3V VOUT = 6V IOUT = 700mA Total power loss in the LT1302, assuming R = 0.15Ω and VD = 0.45V, is: ( PD = 700mA 2 0.7 6 + 0.45 − 3 6 + 0.45 6 + 0.45 0.15Ω − + × × . . . . 0 7 6 0 15 0 7 6 0 15 27 × × 3 − 3− 3 3 )( 2 ) ( )( ) = 223mW + 89mW = 312mW Using the CS8 package with 100 sq. mm topside and backside heat sinking: ∆T = (312mW)(84°C/W) = 25.9°C rise With the N8 package: ∆T = 31.2°C At a 70°C ambient, die temperature would be 101.2°C. 13 LT1302/LT1302-5 U TYPICAL APPLICATIONS Single Cell to 5V/150mA Converter 5V/150mA OUTPUT L1 3.3µH D1 220Ω 10Ω R1 301k 1% 2N3906 (169k FOR 3.3V) 100k 1.5V CELL 100k IL SET 100k VIN SW VIN SW1 LT1073 FB GND SHDN FB LT1302 AO IT PGND SW2 56.2k 1% VC GND 100pF 4.99k 1% 20k + C1 47µF L1 = COILCRAFT DO3316-332 D1 = MOTOROLA MBRS130LT3 C2 220µF + 0.1µF 0.01µF C1 = AVX TPSD476M016R0150 C2 = AVX TPSE227M010R0100 COILCRAFT (708) 639-2361 36.5k 1% 1302 TA03 2V to 12V/120mA Converter L1 3.3µH 6 7 2 CELLS + C1 100µF C3 0.1µF D1 VIN SW 33µF LT1302 8 PGND GND + C2 FB 3 SHUTDOWN 4 VC 2 RC 20k R1 100k 1% 100pF 33µF CC 0.02µF C1 = AVX TPSD107M010R0100 C2 = AVX TPSD336M025R0200 D1 = MOTOROLA MBRS130LT3 L1 = COILCRAFT DO3316-332 OUTPUT 12V 120mA 14 IT SHDN 1 + C2 NC 5 LT1302 • TA04 R2 866k 1% LT1302/LT1302-5 U TYPICAL APPLICATIONS 3 Cell to 3.3V Buck-Boost Converter with Auxiliary 12V Regulated Output VIN 2.5V-8V 10Ω SHUTDOWN SHDN FB 169k 1% 200pF VC GND 6 T1D T1E 4 5 + C3 47µF 16V VIN SW D2 LT1302 100k 1% 7 13V 0.1µF IT PGND 2 + C1 100µF 16V D1 24k 4700pF IN 1 T1C T1A 12V 120mA OUT + 22µF 9 25V 330k 1% LT1121 SHDN 3 8 T1B + ADJ GND + C2 330µF 6.3V 3.3µF 150k 1% 10 1302 TA05 3.3V OUTPUT 400mA T1 = D1, D2 = C1 = C2 = C3 = DALE LPE-6562-A069, 1:3:1:1:1 TURNS RATIO, 10µH PRIMARY. DALE (605) 665-9301 MOTOROLA MBRS130LT3 AVX TPSE107016R0100 AVX TPSE337006R0100 AVX TPSD476016R0150 2 Li-Ion Cell to 5.8V/600mA DC/DC Converter C2 220µF 10V L1 22µH VIN 4V TO 9V + 10Ω + C1 100µF 16V SW FB VIN IT + 1µF 365k 1% L2 22µH MBRS130LT3 VOUT 5.8V 600mA LT1302 SHDN GND SHUTDOWN L1, L2=COILCRAFT DO3316-223 C1=AVX TPSE107016R0100 C2, C3=AVX TPSE227010R0100 VC PGND + 20k 100k 1% C3 220µF 10V 10nF DUTY CYCLE = VOUT VIN + VOUT 1302 TA07 PEAK SWITCH VOLTAGE = VIN + VOUT Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT1302/LT1302-5 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead Plastic DIP 0.400* (10.160) MAX 8 7 6 5 1 2 3 4 0.255 ± 0.015* (6.477 ± 0.381) 0.300 – 0.325 (7.620 – 8.255) 0.065 (1.651) TYP 0.009 – 0.015 (0.229 – 0.381) ( +0.025 0.325 –0.015 8.255 +0.635 –0.381 0.130 ± 0.005 (3.302 ± 0.127) 0.045 – 0.065 (1.143 – 1.651) 0.125 (3.175) MIN 0.045 ± 0.015 (1.143 ± 0.381) ) 0.018 ± 0.003 (0.457 ± 0.076) 0.100 ± 0.010 (2.540 ± 0.254) 0.015 (0.380) MIN N8 0694 *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTURSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm). S8 Package 8-Lead Plastic SOIC 0.189 – 0.197* (4.801 – 5.004) 8 7 6 5 0.150 – 0.157* (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) 1 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.053 – 0.069 (1.346 – 1.752) 0°– 8° TYP 0.016 – 0.050 0.406 – 1.270 0.014 – 0.019 (0.355 – 0.483) 2 3 4 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) BSC SO8 0294 *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006 INCH (0.15mm). 16 Linear Technology Corporation LT/GP 0295 10K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7487 (408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977 LINEAR TECHNOLOGY CORPORATION 1995