MIC2169B DATA SHEET (11/05/2015) DOWNLOAD

MIC2169B
500kHz PWM Synchronous Buck
Control IC
General Description
Features
The MIC2169B is a high-efficiency, simple to use 500kHz
PWM synchronous buck control IC housed in small
MSOP-10 and MSOP-10 ePad packages. The MIC2169B
allows compact DC/DC solutions with a minimal external
component count and cost. The device features highoutput driver capability to drive loads up to 30A.
The MIC2169B operates from a 3V to 14.5V input, without
the need of any additional bias voltage. The output voltage
can be precisely regulated down to 0.8V. The adaptive all
N-Channel MOSFET drive scheme allows efficiencies over
95% across a wide load range within the smallest possible
printed circuit board space area.
The MIC2169B senses current across the high-side NChannel MOSFET, eliminating the need for an expensive
and lossy current-sense resistor. Current-limit accuracy is
maintained by a positive temperature coefficient that tracks
the increasing RDS(ON) of the external MOSFET. Further
cost and space are saved by the internal in-rush currentlimiting digital soft-start. The MIC2169B is identical to the
MIC2169A with the exception that the MIC2169B supports
pre-bias loads and has a lower impedance gate-drive
circuit. Internal pre-bias circuit prevents output voltage
drooping and excessive reverse inductor current when
powering up with a pre-bias voltage at the output.
The MIC2169B is available in a 10-pin MSOP and a
thermally-capable 10-pin ePad MSOP package, with a
wide junction operating range of -40°C to +125°C.
All support documentation can be found on Micrel’s web
site at www.micrel.com.
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3V to 14.5V input voltage range
Adjustable output voltage down to 0.8V
500kHz PWM operation
Up to 95% efficiency
Output Pre-biased Protection
Build-in 2.2Ω drivers to drive two n-channel MOSFETs
Adaptive gate drive increases efficiency
Simple, externally-compensated voltage-mode PWM
control
Short minimum ON time of 30ns allowing very-low duty
cycle
Fast transient response
Adjustable current limit senses high-side N-Channel
MOSFET current
Hiccup mode short-circuit protection
No external current-sense resistor
Internal soft-start current source
Dual function COMP and EN pin allows low-power
shutdown
Available in small-size 10-pin MSOP and 10-pin MSOP
ePad packages
Applications
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Point-of-load DC/DC conversion
High-Current Power Supplies
Telecom/Datacom and Networking Power Supplies
Servers and Workstations
Graphic cards and other PC Peripherals
Set-top boxes
LCD power supplies
___________________________________________________________________________________________________________
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
April 2010
M9999-041210-B
Micrel, Inc.
MIC2169B
Typical Application
VIN = 5V to 12V
SD103BWS
100µF
10µF
0.1µF
10µF
VIN
BST
CS
COMP/EN
150pF
100nF
IRF7821
HSD
MIC2169B
VSW
LSD
1µF
GND
EP
1.0µH
3.3V
IRF7821
1000pF
330µF x 2
FB
EFFICIENCY (%)
VDD
0.1µF
MIC2169B Efficienc
100
95
90
85
80
75
70
65
60
55
50
VIN = 5V
VOUT = 3.3V
0
2
4
6 8 10 12 14 16
ILOAD (A)
MIC2169B Adjustable Output 500kHz Converter
April 2010
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M9999-041210-B
Micrel, Inc.
MIC2169B
Ordering Information
Part Number
Frequency
Junction Temperature Range(1)
Package
Lead Finish
MIC2169BYMME
500kHz
-40° to +125°C
10-Lead ePad MSOP
Pb-Free
MIC2169BYMM
500kHz
-40° to +125°C
10-Lead MSOP
Pb-Free
Pin Configuration
10-Pin ePad MSOP (MME)
10-Pin MSOP
Pin Description
Pin Number
April 2010
Pin Name
Pin Function
1
VIN
Supply Voltage (Input): +3V to +14.5V.
2
VDD
5V Internal Linear Regulator (Output): VDD is the external MOSFET gate-drive
supply voltage and an internal supply bus for the IC. When VIN is <5V, short
VDD to the input supply through a 10Ω resistor.
3
CS
Current Sense (Input): Current-limit comparator noninverting input. The current
limit is sensed across the MOSFET during the ON time. The current can be set
by the resistor in series with the CS pin.
4
COMP/EN
Compensation / Enable (Input): Dual function pin. Pin for external compensation.
If this pin is pulled below 0.25V, with the reference fully up the device shuts
down (50μA typical current draw).
5
FB
6
GND
Ground (Return).
7
LSD
Low-Side Drive (Output): High-current driver output for external synchronous
MOSFET.
8
VSW
Switch (Return): High-side MOSFET driver return.
9
HSD
High-Side Drive (Output): High-current output-driver for the high-side MOSFET.
When VIN is between 3.0V to 5V, 2.5V threshold MOSFETs should be used. At
VIN > 5V, 4.5V threshold MOSFETs should be used.
10
BST
Boost (Input): Provides the drive voltage for the high-side MOSFET driver. The
gate-drive voltage is higher than the source voltage by VDD minus a diode drop.
ePad
EP
Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V.
Connect to Ground.
3
M9999-041210-B
Micrel, Inc.
MIC2169B
Absolute Maximum Ratings(1)
Operating Ratings(2)
Supply Voltage (VIN) ...................................... -0.3V to 15.5V
Booststrapped Voltage (VBST) .................... -0.3V to VIN +6V
VSW .............................................................. -0.3V to 15.5V
CS ............................................................................15.25V
LSD,FB............................................................... -0.3V to 6V
Storage Temperature (TS)..........................-65°C to +150°C
Peak Reflow Temperature (10 to 20 sec) ................ +260°C
ESD (HBM) (3) ................................................................. 2kV
ESD (MM).....................................................................200V
Supply Voltage (VIN)...................................... +3V to +14.5V
Ambient Temperature (TA) ...........................-40°C to +85°C
Junction Temperature (TJ) ..........................-40°C to+125°C
Junction Thermal Resistance
ePad MSOP (θJA)............................................76.7°C/W
ePad MSOP (θJC) .............................................9.6°C/W
MSOP (θJA) ......................................................130°C/W
MSOP (θJC).....................................................42.6°C/W
Output Voltage Range............................. 0.8V to VIN × DMAX
Electrical Characteristics(4)
TJ = 25°C, VIN = 5V; Bold values indicate –40°C ≤ TJ ≤ +125°C; unless otherwise specified.
Min
Typ
Max
Units
(±1%)
0.792
0.8
0.808
V
(±2% over temp)
0.784
0.8
0.816
V
Feedback Bias Current
150
350
nA
Output Voltage Line
Regulation
0.03
%/V
Output Voltage Load
Regulation
0.5
%
Parameter
Condition
Feedback Voltage Reference
Feedback Voltage Reference
Output Voltage Total
Regulation
3V ≤ VIN ≤ 14.5V; 1A ≤ IOUT ≤ 10A; (VOUT = 2.5V)(4)
0.6
1.5
%
500
550
kHz
Oscillator Section
Oscillator Frequency
450
Maximum Duty Cycle
92
Minimum On-Time
(5)
%
30
60
ns
Input and VDD Supply
PWM Mode Supply Current
VCS = VIN –0.25V; VFB = 0.7V (output switching but
excluding external MOSFET gate current.)
1.5
3
mA
Shutdown Quiescent Current
VCOMP/EN = 0V
50
150
µA
0.25
0.35
V
0.1
VCOMP Shutdown Threshold
VCOMP Shutdown Blanking
Period
CCOMP = 100nF
Digital Supply Voltage (VDD)
VIN ≥ 6V
April 2010
675
4.7
4
5
μs
5.3
V
M9999-041210-B
Micrel, Inc.
MIC2169B
Electrical Characteristics(4) (continued)
TJ = 25°C, VIN = 5V; Bold values indicate –40°C ≤ TJ ≤ +125°C; unless otherwise specified.
Parameter
Min
Condition
Typ
Max
Units
Error Amplifier
DC Gain(5)
70
dB
Transconductance
1.1
mΩ–1
Soft-Start
Soft-Start Current
After time out of internal timer. VCOMP = 0.8V
4
8.5
13
µA
160
200
240
µA
Current Sense
CS Over Current Trip Point
VCS = VIN –0.25V
Temperature Coefficient
1800
ppm/°C
ns
Gate Drivers
Rise/Fall Time
Into 3000pF at VIN > 5V
15
Output Driver Impedance
Source, VIN = 4.5V
2.2
3
Ω
Sink, VIN = 4.5V
1.3
3
Ω
Source, VIN = 3V
2.7
4
Ω
Sink, VIN = 3V
1.7
4
Ω
Driver Non-Overlap Time
(5)
50
ns
Notes:
1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating
the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(max),
the junction-to-ambient thermal resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive
die temperature.
2. The device is not guaranteed to function outside its operating rating.
3. Devices are ESD sensitive, handling precautions required.
4. Specification for packaged product only.
5.
Guaranteed by design.
April 2010
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M9999-041210-B
Micrel, Inc.
MIC2169B
Typical Characteristics
0.8005
0.5
5
10
SUPPLY VOLTAGE (V)
15
VDD Line Regulation
3
2
1
0.794
0
0.792
-60 -30 0 30 60 90 120 150
TEMPERATURE (°C)
0
0.0
-60 -30 0 30 60 90 120 150
TEMPERATURE (°C)
15
Oscillator Frequency
vs. Temperature
540
530
520
510
500
490
480
470
460
450
-60
-30
0
30
60
90
TEMPERATURE (°C)
120
150
10
15
5.00
4.98
4.96
4.94
4.92
4.90
0
5
10 15 20 25
LOAD CURRENT (mA)
30
Oscillator Frequency
vs. Suppl Voltage
1.5
550
FREQUENCY (kHz)
1.5
1.0
0.5
10
5
VDD Load Regulation
5.02
VIN (V)
VDD Line Regulation
vs. Temperature
3.5
3.0
2.5
2.0
5
0
VIN (V)
VDD REGULATOR VOLTAGE (V)
VDD (V)
VFB (V)
0
4
0.796
VDD LINE REGULATION (%)
0.7980
6
0.802
April 2010
VFB (V)
0.7985
5
0.798
0.7995
0.7990
1.0
0.804
5.0
4.5
4.0
0.8000
1.5
VFB vs. Temperature
0.800
VFB Line Regulation
0.8010
FREQUENCY VARIATION (%)
0.806
PWM Mode Supply Current
vs. Suppl Voltage
2.0
QUIESCENT CURRENT (mA)
IDD (mA)
PWM Mode Supply Current
vs. Temperature
2.9
2.7
2.5
2.3
2.1
1.9
1.7
1.5
1.3
1.1
0.9
0.7
0.5
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
1.0
0.5
0
-0.5
-1.0
-1.5
0
5
10
15
VIN (V)
6
M9999-041210-B
Micrel, Inc.
MIC2169B
Typical Characteristics (continued)
260
Overcurrent Trip Point
vs. Temperature
240
ICS ( μA)
220
200
180
160
140
120
100
-60 -30 0 30 60 90 120 150
TEMPERATURE (°C)
Functional Diagram
MIC2169B Block Diagram
April 2010
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M9999-041210-B
Micrel, Inc.
MIC2169B
power up, and the chip’s internal oscillator starts
switching. At this point in time, the COMP pin current
source increases to 40µA and an internal 12-bit counter
starts counting which takes approximately 2ms to
complete.
During counting, the COMP voltage is
clamped at 0.65V. After this counting cycle the COMP
current source is reduced to 8.5µA and the COMP pin
voltage rises from 0.65V to 0.95V, the bottom edge of
the saw-tooth oscillator. This is the beginning of 0% duty
cycle and it increases slowly causing the output voltage
to rise slowly. The MIC2169B has one hysteretic
comparator whose output is asserted high when VOUT is
within -3% of steady state. When the output voltage
reaches 97% of programmed output voltage then the gm
error amplifier is enabled along with the hysteretic
comparator output is asserted high. This point onwards,
the voltage control loop (gm error amplifier) is fully in
control and will regulate the output voltage.
Soft-start time can be calculated approximately by
adding the following four time frames:
Functional Description
The MIC2169B is a voltage-mode, synchronous stepdown switching regulator controller designed for high
power. Current limit is implemented without the use of an
external sense resistor. It includes an internal soft-start
function which reduces the power supply input surge
current at start-up by controlling the output voltage rise
time, a PWM generator, a reference voltage, two
MOSFET drivers, and short-circuit current limiting
circuitry to form a complete 500kHz switching regulator.
MIC2169B is identical to the MIC2169A except it
supports pre-bias loads and has a lower impedance
gate-drive circuit.
Theory of Operation
The MIC2169B is a voltage mode step-down regulator.
The figure above illustrates the block diagram for the
voltage control loop. The output voltage variation due to
load or line changes will be sensed by the inverting input
of the transconductance error amplifier via the feedback
resistors R3, and R2 and compared to a reference
voltage at the non-inverting input. This will cause a small
change in the DC voltage level at the output of the error
amplifier which is the input to the PWM comparator. The
other input to the comparator is a 0.95V to 1.45V
triangular waveform. The comparator generates a
rectangular waveform whose width tON is equal to the
time from the start of the clock cycle t0 until t1, the time
the triangle crosses the output waveform of the error
amplifier. To illustrate the control loop, let us assume the
output voltage drops due to sudden load turn-on, this
would cause the inverting input of the error amplifier,
which is divided down version of VOUT, to be slightly less
than the reference voltage, causing the output voltage of
the error amplifier to go high. This will cause the PWM
comparator to increase tON time of the top side
MOSFET, causing the output voltage to go up and
bringing VOUT back in regulation.
t1 = Cap_COMP × 0.25V/8.5µA
t2 = 12 bit counter, approx 2ms
t3 = Cap_COMP × 0.3V/8.5µA
⎛V
t 4 = ⎜⎜ OUT
⎝ VIN
Soft-Start Time(Cap_COMP=100nF) = t1 + t2
+ t3 + t4 = 2.9ms + 2ms + 3.5ms + 1.6ms =
10ms
Current Limit
The MIC2169B uses the RDS(ON) of the top power
MOSFET to measure output current. Since it uses the
drain to source resistance of the power MOSFET, it is
not very accurate. This scheme is adequate to protect
the power supply and external components during a fault
condition by cutting back the time the top MOSFET is on
if the feedback voltage is greater than 0.67V. In case of
a hard short when feedback voltage is less than 0.67V,
the MIC2169B discharges the COMP capacitor to 0.65V,
resets the digital counter and automatically shuts off the
top gate drive, the gm error amplifier is completely
disabled, the –3% hysteretic comparators is asserted
low, and the soft-start cycles restart from t2 to t4. This
mode of operation is called the “hiccup mode” and its
purpose is to protect the down stream load in case of a
hard short. The circuit in Figure 1 illustrates the
MIC2169B current limiting circuit.
Soft-Start
The COMP/EN pin on the MIC2169B is used for the
following three functions:
1. Disables the part by grounding this pin
2. External compensation to stabilize the voltage
control loop
3. Soft-start
For better understanding of the soft-start feature,
assume VIN = 12V, and the MIC2169B is allowed to
power-up by un-grounding the COMP/EN pin. The
COMP pin has an internal 8.5µA current source that
charges the external compensation capacitor. As soon
as this voltage rises to 250mV (t = Cap_COMP ×
0.25V/8.5µA) and VIN crosses the 2.6V UVLO threshold,
the MIC2169B allows the internal VDD linear regulator to
April 2010
⎞
Cap _ COMP
⎟⎟ × 0.5 ×
8.5µA
⎠
8
M9999-041210-B
Micrel, Inc.
MIC2169B
VIN
C2
CIN
HSD
MOSFET Gate Drive
The MIC2169B high-side drive circuit is designed to
switch an N-Channel MOSFET. The Functional Block
Diagram shows a bootstrap circuit, consisting of D1 and
CBST, supplies energy to the high-side drive circuit.
Capacitor CBST is charged while the low-side MOSFET is
on and the voltage on the VSW pin is approximately 0V.
When the high-side MOSFET driver is turned on, energy
from CBST is used to turn the MOSFET on. As the
MOSFET turns on, the voltage on the VSW pin
increases to approximately VIN. Diode D1 is reversed
biased and CBST floats high while continuing to keep the
high-side MOSFET on. When the low-side switch is
turned back on, CBST is recharged through D1. The drive
voltage is derived from the internal 5V VDD bias supply.
The nominal low-side gate drive voltage is 5V and the
nominal high-side gate drive voltage is approximately
4.5V due the voltage drop across D1. An approximate
50ns delay between the high-side and low-side driver
transitions is used to prevent current from
simultaneously flowing unimpeded through both
MOSFETs (shoot-through).
Adaptive gate drive is implemented on the high-side (off)
to low-side (on) driver transition to reduce losses in the
flywheel diode and to prevent shoot-through. This is
operated by detecting the VSW pin; once this pin is
detected to reach 1.5V, the high-side MOSFET can be
assumed to be off and the low side driver is enabled.
Q1
MOSFET N
0.1µF
L1 Inductor
VOUT
RCS
VSW
LSD
CS
Q2
MOSFET N
1000pF
C1
COUT
200 A
Figure 1. The MIC2169B Current Limiting Circuit
The current limiting resistor RCS is calculated by the
following equation:
RCS =
RDS(ON)Q1 × IL
200µA
where:
IL = ILOAD +
Inductor Ripple Current
2
Inductor Ripple Current = VOUT ×
(VIN − VOUT )
VIN × FS × L
FS = 500kHz
200µA is the internal sink current to program the
MIC2169B current limit.
The MOSFET RDS(ON) varies 30% to 40% with
temperature; therefore, it is recommended to add a 50%
margin to the load current (ILOAD) in the above equation
to avoid false current limiting due to increased MOSFET
junction temperature rise. It is also recommended to
connect RCS resistor directly to the drain of the top
MOSFET Q1, and the RSW resistor to the source of Q1 to
accurately sense the MOSFETs RDS(ON). To make the
MIC2169B insensitive to board layout and noise
generated by the switch node, a 1.4Ω resistor and a
1000pF capacitor is recommended between the switch
node and GND.
Total Power Dissipation and Thermal Considerations
Total power dissipation in the MIC2169B equals the
power dissipation caused by driving the external
MOSFETs plus the quiescent supply current:
PdissTOTAL = PdissSUPPLY + PdissDRIVE
where:
PdissSUPPLY = VDD × IDD
IDD is shown in the “PWM Mode Supply Current” graph in
the Typical Characteristics section of the specification.
PdissDRIVE calculations are shown in the Applications
section of the specification.
The die temperature may be calculated once the total
power dissipation is known:
Internal VDD Supply
The MIC2169B controller internally generates VDD for
self biasing and to provide power to the gate drives. This
VDD supply is generated through a low-dropout regulator
and generates 5V from VIN supply greater than 5V. For
supply voltage less than 5V, the VDD linear regulator is
approximately 200mV in dropout. Therefore, it is
recommended to short the VDD supply to the input supply
through a 10Ω resistor for input supplies between 3.0V
to 5V.
April 2010
TJ = TA + PdissTOTAL × θJA
where:
TA is the maximum ambient temperature (°C)
TJ is the junction temperature (°C)
PdissTOTAL is the power dissipation of the
MIC2169B (W)
θJC is the thermal resistance from junction-toambient air (°C/W)
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M9999-041210-B
Micrel, Inc.
MIC2169B
The following graphs are used to determine the
maximum gate charge that can be driven with respect to
supply voltage and ambient temperature. Figure 2 shows
the power dissipation in the driver for different values of
gate charge.
CASE TEMPERATURE RISE (°C)
Figures 4 and 5 show the increase in junction and case
temperature for a given power dissipation.
POWER DISSIPATION (W)
0.70
0.60
0.50
12Vin
0.40
0.30
50
40
MSOP
30
20
ePAD
MSOP
10
0
0.0
0.20
5Vin
0.2
0.4
0.6
0.8
1.0
1.2
POWER DISSIPATION (W)
0.10
0.00
0
20
40
60
80
GATE CHARGE (nC)
Figure 4. Case Temperature Rise vs.
Power Dissipation
100
JUNCTION TEMPERATURE RISE (°C)
Figure 2. Power Dissipation vs. Total Gate Charge
Figure 3 shows the maximum allowable power
dissipation vs ambient temperature. For a given total
gate charge, the maximum operating ambient
temperature can be found by using the two graphs.
MAXIMUM AMBIENT TEMPERATURE (°C)
60
120
100
MSOP
80
ePAD
MSOP
60
40
20
0
140
0.0
120
0.2
0.4
0.6
0.8
1.0
1.2
POWER DISSIPATION (W)
100
80
Figure 5. Junction Temperature Rise vs.
Power Dissipation
ePAD
MSOP
60
MSOP
40
20
0
0.0
0.2
0.4
0.6
0.8
1.0
1.2
POWER DISSIPATION (W)
Figure 3. Maximum Ambient Temperature vs.
Power Dissipation
April 2010
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M9999-041210-B
Micrel, Inc.
MIC2169B
A convenient figure of merit for switching MOSFETs is
the on resistance times the total gate charge RDS(ON)×QG.
Lower numbers translate into higher efficiency. Low
gate-charge logic-level MOSFETs are a good choice for
use with the MIC2169B.
Parameters that are important to MOSFET switch
selection are:
Application Information
MOSFET Selection
The MIC2169B controller works from input voltages of
3V to 14.5V and has an internal 5V regulator to provide
power to turn the external N-Channel power MOSFETs
for high- and low-side switches. For applications where
VIN < 5V, the internal VDD regulator operates in dropout
mode, and it is necessary that the power MOSFETs
used are sub-logic level and are in full conduction mode
for VGS of 2.5V. For applications when VIN > 5V; logiclevel MOSFETs, whose operation is specified at VGS =
4.5V must be used. For the lower (<5V) applications, the
VDD supply can be connected directly to VIN to help
increase the driver voltage to the MOSFET.
It is important to note the on-resistance of a MOSFET
increases with increasing temperature. A 75°C rise in
junction temperature will increase the channel resistance
of the MOSFET by 50% to 75% of the resistance
specified at 25°C. This change in resistance must be
accounted for when calculating MOSFET power
dissipation and in calculating the value of current-sense
(CS) resistor. Total gate charge is the charge required to
turn the MOSFET on and off under specified operating
conditions (VDS and VGS). The gate charge is supplied by
the MIC2169B gate-drive circuit. At 500kHz switching
frequency and above, the gate charge can be a
significant source of power dissipation in the MIC2169B.
At low output load, this power dissipation is noticeable
as a reduction in efficiency. The average current
required to drive the high-side MOSFET is:
• Voltage rating
• On-resistance
• Total gate charge
The voltage ratings for the top and bottom MOSFET are
essentially equal to the input voltage. A safety factor of
20% should be added to the VDS(max) of the MOSFETs
to account for voltage spikes due to circuit parasitics.
The power dissipated in the switching transistor is the
sum of the conduction losses during the on-time
(PCONDUCTION) and the switching losses that occur during
the period of time when the MOSFETs turn on and off
(PAC).
PSW = PCONDUCTION + PAC
where:
PCONDUCTION = ISW (rms )2 × RSW
PAC = PAC(off ) + PAC(on)
RSW = on-resistance of the MOSFET switch
⎛V ⎞
D = duty cycle = ⎜⎜ O ⎟⎟
⎝ VIN ⎠
IG[high −side](avg ) = Q G × f S
Making the assumption the turn-on and turn-off transition
times are equal; the transition times can be
approximated by:
where:
IG[high-side](avg) = average high-side MOSFET gate
current.
QG = total gate charge for the high-side MOSFET
taken from manufacturer’s data sheet for VGS = 5V.
The low-side MOSFET is turned on and off at VDS = 0
because the freewheeling diode is conducting during this
time. The switching loss for the low-side MOSFET is
usually negligible. Also, the gate-drive current for the
low-side MOSFET is more accurately calculated using
CISS at VDS = 0 instead of gate charge.
For the low-side MOSFET:
tT =
where:
CISS and COSS are measured at VDS = 0
IG = gate-drive current (1.4A for the MIC2169B)
The total high-side MOSFET switching loss is:
PAC = (VIN + VD ) × IPK × t T × fS
where:
tT = switching transition time (typically 20ns to
50ns)
VD = freewheeling diode drop, typically 0.5V
fS it the switching frequency, nominally 500kHz
The low-side MOSFET switching losses are negligible
and can be ignored for these calculations.
IG[low − side](avg ) = CISS × VGS × fS
Since the current from the gate drive comes from the
input voltage, the power dissipated in the MIC2169B due
to gate drive is:
(
PGATEDRIVE = VIN × IG[high −side](avg) + IG[low −side](avg)
April 2010
CISS × VGS + COSS × VIN
IG
)
11
M9999-041210-B
Micrel, Inc.
MIC2169B
currents, the core losses can be a significant contributor.
Core loss information is usually available from the
magnetics vendor. Copper loss in the inductor is
calculated by the equation below:
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by the equation below.
L=
(
VOUT × VIN(max) − VOUT
PINDUCTORCU
The resistance of the copper wire, RWINDING, increases
with temperature. The value of the winding resistance
used should be at the operating temperature.
R WINDING (hot ) = R WINDING ( 20°C) × (1 + 0.0042 × (THOT − T20°C )
where:
THOT = temperature of the wire under operating
load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
)
VIN(max) × f S × 0.2 × IOUT(max)
Output Capacitor Selection
The output capacitor values are usually determined by
the capacitors ESR (equivalent series resistance).
Voltage and RMS current capability are two other
important factors selecting the output capacitor.
Recommended capacitors are tantalum, low-ESR
aluminum electrolytics, and POSCAPS. The output
capacitor’s ESR is usually the main cause of output
ripple. The output capacitor ESR also affects the overall
voltage feedback loop from stability point of view. See
“Feedback Loop Compensation” section for more
information. The maximum value of ESR is calculated:
where:
fS = switching frequency, 500kHz
0.2 = ratio of AC ripple current to DC output
current
VIN(max) = maximum input voltage
The peak-to-peak inductor current (AC ripple current) is:
IPP =
(
VOUT × VIN(max) − VOUT
VIN(max) × fS × L
)
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor ripple
current.
RESR ≤
IPK = IOUT(max) + 0.5 × IPP
ΔVOUT
IPP
where:
VOUT = peak-to-peak output voltage ripple
IPP = peak-to-peak inductor ripple current
The total output ripple is a combination of the ripple due
to the output capacitors’ ESR and the ripple due to the
output capacitor. The total ripple is calculated below:
The RMS inductor current is used to calculate the I2 × R
losses in the inductor.
I 2
IINDUCTOR = (IOUT _ MAX ) 2 + PP
12
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC2169B requires the
use of ferrite materials for all but the most cost sensitive
applications.
Lower cost iron powder cores may be used but the
increase in core loss will reduce the efficiency of the
power supply. This is especially noticeable at low output
power. The winding resistance decreases efficiency at
the higher output current levels. The winding resistance
must be minimized although this usually comes at the
expense of a larger inductor. The power dissipated in the
inductor is equal to the sum of the core and copper
losses. At higher output loads, the core losses are
usually insignificant and can be ignored. At lower output
April 2010
2
= IINDUCTOR(rms) × R WINDING
2
ΔVOUT
⎛ I × (1 − D) ⎞
⎟⎟ + (IPP × RESR )2
= ⎜⎜ PP
⎝ COUT × fS ⎠
where:
D = duty cycle
COUT = output capacitance value
fS = switching frequency
12
M9999-041210-B
Micrel, Inc.
MIC2169B
The output voltage is determined by the equation:
The voltage rating of capacitor should be twice the
voltage for a tantalum and 20% greater for aluminum
electrolytic.
The output capacitor RMS current is calculated below:
IC OUT ( rms ) =
R1 ⎞
⎛
VO = VREF × ⎜1 +
⎟
R
2⎠
⎝
where
VREF for the MIC2169B is typically 0.8V
IPP
12
A typical value of R1 can be between 3kΩ and 10kΩ. If
R1 is too large, it may allow noise to be introduced into
the voltage feedback loop. If R1 is too small, in value, it
will decrease the efficiency of the power supply,
especially at light loads. Once R1 is selected, R2 can be
calculated using:
The power dissipated in the output capacitor is:
(
PDISS(C OUT ) = IC OUT ( rms )
)2 × RESR(C
OUT
)
Input Capacitor Selection
The input capacitor should be selected for ripple current
rating and voltage rating. Tantalum input capacitors may
fail when subjected to high inrush currents, caused by
turning the input supply on. A tantalum input capacitor’s
voltage rating should be at least 2 times the maximum
input voltage to maximize reliability. Aluminum
electrolytic, OS-CON, and multilayer polymer film
capacitors can handle the higher inrush currents without
voltage derating. The input voltage ripple will primarily
depend on the input capacitor’s ESR. The peak input
current is equal to the peak inductor current, so:
R2 =
External Schottky Diode
An external freewheeling diode is used to keep the
inductor current flow continuous while both MOSFETs
are turned off. This dead time prevents current from
flowing unimpeded through both MOSFETs and is
typically 50ns. The diode conducts twice during each
switching cycle. Although the average current through
this diode is small, the diode must be able to handle the
peak current.
ΔVIN = IINDUCTOR(peak ) × RESR(CIN )
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor ripple current is low:
ID(avg) = IOUT × 2 × 50ns × fS
The reverse voltage requirement of the diode is:
VDIODE(rrm) = VIN
The power dissipated by the Schottky diode is:
PDIODE = ID(avg) × VF
where:
ICIN( rms ) ≈ IOUT(max) × D × (1 − D)
The power dissipated in the input capacitor is:
(
PDISS(CIN ) = ICIN(rms )
)2 × RESR(C
IN
VF = forward voltage at the peak diode current
The external Schottky diode, D1, is not necessary for
circuit operation since the low-side MOSFET contains a
parasitic body diode. The external diode will improve
efficiency and decrease high frequency noise. If the
MOSFET body diode is used, it must be rated to handle
the peak and average current. The body diode has a
relatively slow reverse recovery time and a relatively
high forward voltage drop. The power lost in the diode is
proportional to the forward voltage drop of the diode. As
the high-side MOSFET starts to turn on, the body diode
becomes a short circuit for the reverse recovery period,
dissipating additional power. The diode recovery and the
circuit inductance will cause ringing during the high-side
MOSFET turn-on. An external Schottky diode conducts
at a lower forward voltage preventing the body diode in
the MOSFET from turning on. The lower forward voltage
drop dissipates less power than the body diode. The lack
of a reverse recovery mechanism in a Schottky diode
causes less ringing and less power loss.
)
Voltage Setting Components
The MIC2169B requires two resistors to set the output
voltage as shown in Figure 6.
R1
Error
Amp
FB
5
R2
VREF
0.8V
MIC2169B
Figure 6. Voltage-Divider Configuration
April 2010
VREF × R1
VO − VREF
13
M9999-041210-B
Micrel, Inc.
MIC2169B
Depending on the circuit components and operating
conditions, an external Schottky diode will give a ½% to
1% improvement in efficiency.
Feedback Loop Compensation
The MIC2169B controller comes with an internal
transconductance error amplifier used for compensating
the voltage feedback loop by placing a capacitor (C1) in
series with a resistor (R1) and another capacitor C2 in
parallel from the COMP pin to ground. See “Functional
Block Diagram.”
Power Stage
The power stage of a voltage mode controller has an
inductor, L1, with its winding resistance (DCR)
connected to the output capacitor, COUT, with its
electrical series resistance (ESR) as shown in Figure 7.
The transfer function G(s), for such a system is:
L
Figure 9. Phase Curve for G(s)
It can be seen from the transfer function G(s) and the
gain curve that the output inductor and capacitor create
a two pole system with a break frequency at:
DCR
VO
fLC =
ESR
1
2 × π L × COUT
Therefore, fLC = 6.2kHz
By looking at the phase curve, it can be seen that the
output capacitor ESR (0.025Ω) cancels one of the two
poles (LCOUT) system by introducing a zero at:
COUT
Figure 7. The Output LC Filter in a Voltage-Mode Buck
Converter
fZERO =
⎛
⎞
(1 + ESR × s × C)
⎟⎟
G(s) = ⎜⎜
2
⎝ DCR × s × C + s × L × C + 1 + ESR × s × C ⎠
1
2 × π × ESR × COUT
Therefore, FZERO = 9.6kHz.
From the point of view of compensating the voltage loop,
it is recommended to use higher ESR output capacitors
since they provide a 90° phase gain in the power path.
For comparison purposes, Figure 10, shows the same
phase curve with an ESR value of 0.002Ω.
Plotting this transfer function with the following assumed
values
(L=1μH,
DCR=0.009Ω,
COUT=660μF,
ESR=0.025Ω) gives lot of insight as to why one needs to
compensate the loop by adding resistor and capacitors
on the COMP pin. Figures 8 and 9 show the gain curve
and phase curve for the above transfer function.
Figure 10. The Phase Curve with ESR = 0.002Ω
Figure 8. The Gain Curve for G(s)
April 2010
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M9999-041210-B
Micrel, Inc.
MIC2169B
It can be seen from Figure 9 that at 50kHz, the phase is
approximately –90° versus Figure 10 where the number
is –150°. This means that the transconductance error
amplifier has to provide a phase boost of about 45° to
achieve a closed loop phase margin of 45° at a
crossover frequency of 50kHz for Figure 9, versus 105°
for Figure 10. The simple RC and C2 compensation
scheme allows a maximum error amplifier phase boost
of about 90°. Therefore, it is easier to stabilize the
MIC2169B voltage control loop by using high-ESR value
output capacitors.
gm Error Amplifier
It is undesirable to have high error amplifier gain at high
frequencies because high frequency noise spikes would
be picked up and transmitted at large amplitude to the
output, thus, gain should be permitted to fall off at high
frequencies. At low frequency, it is desired to have high
open-loop gain to attenuate the power line ripple. Thus,
the error amplifier gain should be allowed to increase
rapidly at low frequencies.
The transfer function with R1, C1, and C2 for the internal
gm error amplifier can be approximated by the following
equation:
Figure 11. Error Amplifier Gain Curve
⎡
⎤
⎢
⎥
1 + s × R1 × C1
Error Amplifier (z) = gm × ⎢
⎥
⎢ s × (C1 + C2 ) × ⎛⎜ 1 + s × R1 × C1 × C2 ⎞⎟ ⎥
⎥
C1 + C2 ⎠ ⎦
⎝
⎣⎢
The above equation can be simplified by assuming
C2<<C1,
Figure 12. Error Amplifier Phase Curve
⎤
⎡
1 + s × R1 × C1
Error Amplifier (z) = gm × ⎢
⎥
⎣ s × C1 × (1 + s × R1 × C2) ⎦
Total Open-Loop Response
The open-loop response for the MIC2169B controller is
easily obtained by adding the power path and the error
amplifier gains together, since they already are in Log
scale. It is desirable to have the gain curve intersect zero
dB at tens of kilohertz, this is commonly called crossover
frequency; the phase margin at crossover frequency
should be at least 45°. Phase margins of 30° or less
cause the power supply to have substantial ringing when
subjected to transients, and have little tolerance for
component or environmental variations.
Figures 13 and 14 show the open-loop gain and phase
margin for the 5V input and 1.8V output application, and
it can be seen from Figure 13 that the gain curve
intersects the 0dB at approximately 50kHz, and from
Figure 14 that at 50kHz, the phase shows approximately
74° of margin.
From the above transfer function, one can see that R1
and C1 introduce a zero and R1 and C2 a pole at the
following frequencies:
FZERO= 1/2 π × R1 × C1
FPOLE = 1/2 π × C2 × R1
FPOLE@origin = 1/2 π × C1
Figures 11 and 12 show the gain and phase curves for
the above transfer function with R1 = 4.02k, C1 = 100nF,
C2 = 150pF, and gm = 1.1mΩ–1.
April 2010
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M9999-041210-B
Micrel, Inc.
MIC2169B
Figure 13. Open-Loop Gain Margin
Figure 15. MIC2169A Startup without Pre-Bias Support
Figure 15 shows MIC2169B startup with a pre-bias of 1V
on the output, in which the pre-existing output voltage
discharges during soft start.
Figure 14. Open-Loop Phase Margin
Pre-Biased Loads
The MIC2169B supports pre-biased loads. Some
applications have a pre-existing voltage on the output.
This pre-existing or pre-biased load is generated by an
external supply (other than the MIC2169B). During
startup without pre-bias support, MIC2169A will pull the
output voltage to ground through the inductor and low
side FET (see Figure 15).
The MIC2169B prevents the current sinking of any preexisting voltage source at the output (see Figure 16). It
does this by keeping the low-side FET off during the soft
start period. In some applications this pre-bias current
sink is not a problem, and the MIC2169A may be used.
In some applications the pre-bias current sink may
cause a problem, and the MIC2169B should be used.
The MIC2169B can support up to 90% of a pre-bias
condition (up to 90% of the final regulated output
voltage) see Figure 17.
April 2010
Figure 16. MIC2169B Startup with Pre-Bias Support
Figure 16 shows MIC2169B startup with a pre-bias of 1V
on the output, in which the pre-existing output voltage
has no discharge.
16
M9999-041210-B
Micrel, Inc.
MIC2169B
Figure 17. MIC2169B Startup with Pre-Bias Support,
Pre-Bias At 90% of VOUT_FINAL
•
Keep both the VIN and power GND connections
short.
•
Place several vias to the ground plane close to the
VIN input capacitor ground terminal.
•
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
•
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
•
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
•
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is
suddenly applied.
•
An additional Tantalum or Electrolytic bypass input
capacitor of 22uF or higher is required at the input
power connection.
•
Use a 5Ω resistor from the input supply to the VDD
pin on the MIC2169B. Also, place a 1µF ceramic
capacitor from this pin to GND, preferably not
through a via. The capacitor must be located right at
the IC. The Vdd terminal is very noise sensitive and
placement of the capacitor is very critical.
Connections must be made with wide trace.
Figure 17 shows MIC2169B startup with a pre-bias of
2.2V on the output (90% of VOUT) without the pre-existing
output voltage discharge.
Design and PCB Layout Guideline
WARNING!!! TO MINIMIZE EMI AND OUTPUT NOISE,
FOLLOW THESE LAYOUT RECOMMENDATIONS:
Inductor
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC2169B converter.
•
Keep the inductor connection to the switch node
(SW) short.
•
Do not route any digital lines underneath or close to
the inductor.
•
Keep the switch node (SW) away from the feedback
(FB) pin.
•
To minimize noise, place a ground plane underneath
the inductor.
IC
•
Place the IC and MOSFETs close to the point of
load (POL).
•
Use fat traces to route the input and output power
lines.
•
Signal and power grounds should be kept separate
and connected at only one location.
Output Capacitor
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high-current
load trace can degrade the DC load regulation.
Input Capacitor
•
Place the VIN input capacitor next.
•
Place the VIN input capacitors on the same side of
the board and as close to the MOSFETs as
possible.
April 2010
17
M9999-041210-B
Micrel, Inc.
MIC2169B
MOSFETs
Others
•
•
Connect the current limiting (R2) resistor directly to
the drain of top MOSFET Q3.
•
The feedback resistors R3 and R4/R5/R6 should be
placed close to the FB pin. The top side of R3
should connect directly to the output node. Run this
trace away from the switch node (junction of Q3, Q2,
and L1). The bottom side of R3 should connect to
the GND pin on the MIC2169B.
•
The compensation resistor and capacitors should be
placed right next to the COMP pin and the other side
should connect directly to the GND pin on the
MIC2169B rather than going to the plane.
•
Add a place holder for a gate resistor on the top
MOSFET gate drive. Do not use a resistor in series
with the low-side MOSFET gate.
Low gate charge MOSFETs should be used to
maximize efficiency, such as Si4800, Si4804BDY,
IRF7821, IRF8910, FDS6680A and FDS6912A, etc.
RC Snubber
•
Add a RC snubber of 1.4Ω resistor and a 1000pF
capacitor from the switch node to ground pin. Place
the snubber on the same side of the board and as
close to the MOSFETs as possible. See page 8,
Current Limiting section for more detail.
Schottky Diode (Optional)
•
Place the Schottky diode on the same side of the
board as the MOSFETs and VIN input capacitor.
•
The connection from the Schottky diode’s Anode to
the input capacitors ground terminal must be as
short as possible.
•
The diode’s Cathode connection to the switch node
(SW) must be keep as short as possible.
Evaluation Board Schematics
MIC2169B Evaluation Board Schematic
April 2010
18
M9999-041210-B
Micrel, Inc.
MIC2169B
Bill of Materials
Item
Part Number
Manufacturer
Description
U1
MIC2169B-YMME
Micrel, Inc.
Buck Controller
1
Q1, Q2
IRF7821-TR
SI4174DY
IR
Vishay
30V, N-Channel HEXFET, Power MOSFET
2
0
Q3
2N7002E
On Semiconductor
60V, N-Channel MOSFET
0
D1
SD103BWS
Vishay
30V, Schottky Diode
1
D2
1N5819HW
SL04
CMMSH1-40
Diodes, Inc.
Vishay
Central Semi
40V, Schottky Diode
1
0
0
L1
CDRH127LDNP-1R0NC
HC5-1R0
SER1360-1R0
Sumida
Cooper Electronic
Coilcraft
1.0μH, 10A Inductor
1
0
0
C1
C3225X7R1C106M
TDK
10μF/16V, X7R Ceramic Capacitor
1
C2, C3
TPSD686M020R0070
594D686X0020D2T
AVX
Vishay/Sprague
68μF, 20V Tantalum
2
0
C4
C2012X5R0J106M
TDK
10μF/6.3V, 0805 Ceramic Capacitor
1
C5, C10, C12,
C16
VJ1206Y104KXXAT
Vishay Victramon
0.1μF/25V Ceramic Capacitor
4
C6, C7
TPSD337M006R0045
AVX
330μF/6.3V, Tantalum
2
Open
0
C8, C11, C17
Qty.
C13
C2012X7R1C105K
GRM21BR71C105KA01B
VJ1206S105KXJAT
TDK
muRata
Vishay Victramon
1μF/16V, 0805 Ceramic Capacitor
1
0
0
C15
VJ0603A102KXXAT
Vishay Victramon
1000pF/25V, 0603, NPO
1
R2
CRCW06034700JRT1
Vishay
470Ω, 0603, 1/16W, 5%
1
R3
CRCW08051002FRT1
Vishay
10kΩ, 0805, 1/10W, 1%
1
R4
CRCW08053161FRT1
Vishay
3.16kΩ, 0805, 1/10W, 1%
1
R5
CRCW08054641FRT1
Vishay
4.64kΩ, 0805, 1/10W, 1%
1
R6
CRCW08051132FRT1
Vishay
11.3kΩ, 0805, 1/10W, 1%
1
R7
CRCW08051003FRT1
Vishay
100kΩ, 0805, 1/10W, 1%
1
C9
VJ0603A151KXAAT
Vishay
150pF/50V, 0603, NPO
1
Notes:
1.
Micrel.Inc
408-944-0800
2.
Vishay corp
206-452-5664
3.
Diodes. Inc
805-446-4800
4.
Sumida
408-321-9660
5.
TDK
847-803-6100
6.
muRata
800-831-9172
7.
AVX
843-448-9411
8.
International Rectifier
847-803-6100
9.
Fairchild Semiconductor
207-775-8100
10. Cooper Electronic
561-752-5000
11. Coilcraft
1-800-322-2645
12. Central Semi
631-435-1110
April 2010
19
M9999-041210-B
Micrel, Inc.
MIC2169B
Bill of Materials (continued)
Item
Part Number
Manufacturer
Description
R8
CRCW06034021FRT1
Vishay
4.02kΩ, 0603, 1/16W, 1%
1
R9
CRCW120610R0FRT1
Vishay
10Ω, 1/8W, 1206, 1%
1
R10
CRCW12062R00FRT1
Vishay
2Ω, 1/8W, 1206, 1%
1
R12
CRCW12061R40FRT1
Vishay
1.4Ω, 1/8W, 1206, 1%
R14
J1, J3, J4, J5
2551-2-00-01-00-00-07-0
MillMax
Qty.
Open
0
Turrent Pins
4
Notes:
13. Micrel.Inc
408-944-0800
14. Vishay corp
206-452-5664
15. Diodes. Inc
805-446-4800
16. Sumida
408-321-9660
17. TDK
847-803-6100
18. muRata
800-831-9172
19. AVX
843-448-9411
20. International Rectifier
847-803-6100
21. Fairchild Semiconductor
207-775-8100
22. Cooper Electronic
561-752-5000
23. Coilcraft
1-800-322-2645
24. Central Semi
631-435-1110
April 2010
20
M9999-041210-B
Micrel, Inc.
MIC2169B
MIC2169B PCB Layout
MIC2169B Top Layer
April 2010
MIC2169B Bottom Layer
21
M9999-041210-B
Micrel, Inc.
MIC2169B
Package Information
10-Pin ePad MSOP (MME)
April 2010
22
M9999-041210-B
Micrel, Inc.
MIC2169B
Package Information (continued)
10-Pin MSOP (MM)
April 2010
23
M9999-041210-B
Micrel, Inc.
MIC2169B
MIC2169B Land Patterns
Recommended Land Pattern for 10-Pin MSOP
April 2010
24
M9999-041210-B
Micrel, Inc.
MIC2169B
MIC2169B Land Patterns (continued)
Recommended Land Pattern for ePad 10-Pin MSOP
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2006 Micrel, Incorporated.
April 2010
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M9999-041210-B