2-, 3-, 4-Phase Controller for CPU Applications

NCP5392T
2/3/4-- Phase Controller with
Light Load Power Saving
Enhancement for CPU
Applications
The NCP5392T provides up to a four-- phase buck solution which
combines differential voltage sensing, differential phase current
sensing, and adaptive voltage positioning to provide accurately
regulated power for Intel processors. It also receives power saving
command (PSI) from CPU, and operates in a single phase emulation
diode mode to obtain a high efficiency at light load. Dual-- edge
pulse-- width modulation (PWM) combined with precise inductor
current sensing provides the fastest initial response to dynamic load
events both in power saving and normal modes. Dual-- edge
multiphase modulation reduces the total bulk and ceramic output
capacitance required therefore reducing the system cost to meet
transient regulation specifications.
A high performance operational error amplifier is provided to
simplify compensation of the system. Dynamic Reference Injection
further simplifies loop compensation by eliminating the need to
compromise between closed-- loop transient response and Dynamic
VID performance. An enhancement of normal mode and PSI mode
operation has been achieved in NCP5392T both under heavy load
and light load condition or the load changing.
Features
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MARKING
DIAGRAM
1
1 40
NCP5392T
AWLYYWWG
40 PIN QFN, 6x6
MN SUFFIX
CASE 488AR
NCP5392T = Specific Device Code
A
= Assembly Location
WL
= Wafer Lot
YY
= Year
WW
= Work Week
G
= Pb--Free Package
*Pin 41 is the thermal pad on the bottom of the device.
ORDERING INFORMATION
Device
Package
Shipping†
Meets Intel’s VR11.1 Specifications
NCP5392TMNR2G* QFN--40 2500/Tape & Reel
Enhanced Power Saving Operation (PSI)
(Pb--Free)
Dual-- edge PWM for Fastest Initial Response to Transient Loading
*Temperature Range: 0C to 85C
High Performance Operational Error Amplifier
†For information on tape and reel specifications,
including part orientation and tape sizes, please
Internal Soft Start
refer to our Tape and Reel Packaging Specification
Brochure, BRD8011/D.
Dynamic Reference Injection
DAC Range from 0.375 V to 1.6 V
DAC Feed Forward Function
0.5% DAC Voltage Accuracy from 1.0 V to 1.6 V
True Differential Remote Voltage Sensing Amplifier
Phase-- to-- Phase Current Balancing
“Lossless” Differential Inductor Current Sensing
 Threshold Sensitive Enable Pin for VTT Sensing
Accurate Current Monitoring (IMON)
 Power Good Output with Internal Delays
Differential Current Sense Amplifiers for each Phase
 Thermally Compensated Current Monitoring
Adaptive Voltage Positioning (AVP)
 This is a Pb-- Free Device
Oscillator Frequency Range of 100 kHz – 1 MHz
Applications
Latched Over Voltage Protection (OVP)
 Desktop Processors
Guaranteed Startup into Pre-- Charged Loads
 Semiconductor Components Industries, LLC, 2010
November, 2010 - Rev. 1
1
Publication Order Number:
NCP5392T/D
NCP5392T
31
G2
33
34
32
G3
VCC
12VMON
35
36
DAC
37
PSI
38
NTC
39
VR_RDY
G4
CS3
NCP5392T
VID4
CS3N
2/3/4--Phase Buck Controller
(QFN40)
VID5
CS2
COMP
CS1N
30
29
28
27
26
25
24
23
22
21
20
19
18
17
11
ILIM
ROSC
CSSUM
CS1
VDFB
VID7
VDRP
CS2N
VFB
VID6
16
9
10
VID3
DIFFOUT
8
CS4N
VSN
7
VID2
15
6
CS4
14
5
VID1
VSP
4
DRVON
13
3
G1
VID0
IMON
2
EN
12
1
VR_HOT
40
PIN CONNECTIONS
Figure 1. NCP5392T QFN40 Pin Connections (Top View)
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2
NCP5392T
VID0
VID1
VID2
VID3
VID4
VID5
VID6
VID7
Flexible DAC
Overvoltage
Protection
--
DAC
+
VSN
--
VSP
+
+
G1
--
Diff Amp
DIFFOUT
+
VFB
Error Amp
+
--
1.3 V
+
VDRP
+
--
Droop Amp
VDFB
--2/3
CSSUM
CS1P
CS1N
CS2P
CS2N
CS3P
CS3N
CS4P
CS4N
G2
--
COMP
+
+
-+
-+
-+
--
+
+
G3
--
Gain = 6
+
Gain = 6
+
Gain = 6
+
+
G4
--
+
Gain = 6
Oscillator
IMON
ROSC
DRVON
+
--
ILIM
ILimit
EN
+
VCC
4.25 V
+
-UVLO
GND (FLAG)
Figure 2. NCP5392T Block Diagram
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3
Control,
Fault Logic
and
Monitor
Circuits
PSI
NTC
VR_HOT
12VMON
VR_RDY
NCP5392T
12V_FILTER
12V_FILTER
+5V
12V_FILTER
D1
VTT
C4
C3
PSI
5
6
7
8
9
1
39
40
PSI
37
NTC
VID1
G1
VID6
VID7
CS1P
CS1N
EN
VR_RDY
G2
VR_HOT
CS2P
CS2N
G3
CS3P
CS3N
16
17
CH
18
RDRP
19
RISO1
36
CS4P
COMP
CS4N
RS1
PGND
30
CS1
22
21
12V_FILTER
12V_FILTER
31
24
23
32
BST
26
VCC
DRH
NCP5359
SW
OD
25
33
DRL
28
IN
27
PGND
VFB
DRVON
VDRP
29
VDFB
12V_FILTER
12V_FILTER
CSSUM
+
DAC
GND
CDFB
R6 20
CDNI
RNOR
R2
RT2 RISO2
41
11
ROSC
RF
G4
DIFFOUT
ILIM
CF
Q2
C2
VID5
RFB
15
IN
L1
VID4
VSN NCP5392T
13 VSP
RFB1
DRL
RT1
IMON
VID3
14
CFB1
Q1
DRH
NCP5359
SW
OD
38
IMON 12
VID2
BST
VCC
10
CPU GND
4
35
VID0
VCC
3
12VMON
34
2
C1
RNTC1
BST
RDNP
VCC
RLIM1
DRH
NCP5359
SW
OD
RLIM2
IN
DRL
PGND
12V_FILTER
12V_FILTER
BST
VCC
DRH
NCP5359
SW
OD
IN
DRL
PGND
VCCP
VSSN
Figure 3. Application Schematic for Four Phases
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4
NCP5392T
12V_FILTER
12V_FILTER
+5V
12V_FILTER
D1
VTT
C4
C3
PSI
5
6
7
8
9
1
39
40
PSI
VID0
37
NTC
VID1
VID5
G1
VID6
VID7
CS1P
CS1N
EN
VR_RDY
G2
VR_HOT
CS2P
CS2N
G3
CS3P
RFB
CS3N
16
17
CH
18
RDRP
19
RISO1
20
36
CS4P
COMP
IN
Q2
CS4N
R2
RS1
C2
CS1
PGND
30
22
21
12V_FILTER
12V_FILTER
31
24
23
32
BST
26
VCC
DRH
NCP5359
SW
OD
25
33
DRL
28
IN
27
PGND
VFB
VDRP
DRVON
29
VDFB
12V_FILTER
12V_FILTER
CSSUM
+
DAC
GND
CDFB
R6
CDNI
RNOR
L1
RT2 RISO2
41
11
ROSC
RF
G4
DIFFOUT
ILIM
CF
15
Q1
VID4
VSN NCP5392T
13 VSP
RFB1
DRL
RT1
IMON
VID3
14
CFB1
BST
VCC
DRH
NCP5359
SW
OD
38
IMON 12
VID2
C1
10
BST
RDNP
VCC
RLIM1
DRH
NCP5359
SW
OD
RLIM2
IN
DRL
PGND
VCCP
VSSN
Figure 4. Application Schematic for Three Phases
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5
CPU GND
4
35
VCC
3
12VMON
34
2
RNTC1
NCP5392T
12V_FILTER
12V_FILTER
+5V
12V_FILTER
D1
VTT
C4
C3
PSI
5
6
7
8
9
1
39
40
PSI
VID0
37
NTC
VID1
VID5
G1
VID6
VID7
CS1P
CS1N
EN
VR_RDY
G2
VR_HOT
CS2P
CS2N
G3
CS3P
RFB
CS3N
16
17
CH
18
RDRP
19
RISO1
20
36
CS4P
COMP
IN
Q2
CS4N
R2
RS1
C2
CS1
PGND
30
22
21
31
24
23
32
26
25
33
28
27
VFB
VDRP
DRVON
29
VDFB
12V_FILTER
12V_FILTER
CSSUM
+
DAC
GND
CDFB
R6
CDNI
RNOR
L1
RT2 RISO2
41
11
ROSC
RF
G4
DIFFOUT
ILIM
CF
15
Q1
VID4
VSN NCP5392T
13 VSP
RFB1
DRL
RT1
IMON
VID3
14
CFB1
BST
VCC
DRH
NCP5359
SW
OD
38
IMON 12
VID2
C1
10
BST
RDNP
VCC
RLIM1
DRH
NCP5359
SW
OD
RLIM2
IN
DRL
PGND
VCCP
VSSN
Figure 5. Application Schematic for Two Phases
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6
CPU GND
4
35
VCC
3
12VMON
34
2
RNTC1
NCP5392T
PIN DESCRIPTIONS
Pin No.
Symbol
1
EN
Description
2
VID0
Voltage ID DAC input
3
VID1
Voltage ID DAC input
4
VID2
Voltage ID DAC input
5
VID3
Voltage ID DAC input
6
VID4
Voltage ID DAC input
7
VID5
Voltage ID DAC input
8
VID6
Voltage ID DAC input
9
VID7
Voltage ID DAC input
10
ROSC
11
ILIM
12
IMON
13
VSP
Non--inverting input to the internal differential remote sense amplifier
14
VSN
Inverting input to the internal differential remote sense amplifier
15
DIFFOUT
16
COMP
Threshold sensitive input. High = startup, Low = shutdown.
A resistance from this pin to ground programs the oscillator frequency according to fSW. This pin supplies a
trimmed output voltage of 2 V.
Overcurrent shutdown threshold setting. Connect this pin to the ROSC pin via a resistor divider as shown in
the Application Schematics. To disable the overcurrent feature, connect this pin directly to the ROSC pin. To
guarantee correct operation, this pin should only be connected to the voltage generated by the ROSC pin; do
not connect this pin to any externally generated voltages.
0 mV to 900 mV analog signal proportional to the output load current. VSN referenced
Output of the differential remote sense amplifier
Output of the error amplifier
17
VFB
18
VDRP
Compensation Amplifier Voltage feedback
Voltage output signal proportional to current used for current limit and output voltage droop
19
VDFB
Droop Amplifier Voltage Feedback
20
CSSUM
21
CS1N
Inverted Sum of the Differential Current Sense inputs. Av=CSSUM/CSx = --4
Inverting input to current sense amplifier #1
22
CS1
23
CS2N
Non--inverting input to current sense amplifier #1
24
CS2
25
CS3N
26
CS3
27
CS4N
28
CS4
29
DRVON
30
G1
PWM output pulse to gate driver. 3--level output: Low = LSFET Enabled, Mid = Diode Emulation Enabled,
High = HSFET Enabled
31
G2
PWM output pulse to gate driver. 3--level output (see G1)
32
G3
PWM output pulse to gate driver. 3--level output (see G1)
33
G4
PWM output pulse to gate driver. 3--level output (see G1)
34
12VMON
35
VCC
Power for the internal control circuits.
36
DAC
DAC Feed Forward Output
37
PSI
Power Saving Control. Low = single phase operation, High = normal operation.
38
NTC
Threshold sensitive input for thermal monitoring
39
VR_RDY
Open collector output. High indicates that the output is regulating
40
VR_HOT
Open collector output indicates the state of the thermal monitoring input. Low impedance output indicating a
normal status when the voltage of NTC pin is above the specified threshold. This pin will transition to high
impedance when the voltage of NTC pin decrease (temperature increase) below the specified threshold.
This pin requires an external pullup resistor
FLAG
GND
Inverting input to current sense amplifier #2
Non--inverting input to current sense amplifier #2
Inverting input to current sense amplifier #3
Non--inverting input to current sense amplifier #3
Inverting input to current sense amplifier #4
Non--inverting input to current sense amplifier #4
Bidirectional Gate Drive Enable
Monitor a 12 V input through a resistor divider.
Power supply return (QFN Flag)
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NCP5392T
PIN CONNECTIONS VS. PHASE COUNT
Number of Phases
G4
G3
G2
G1
CS4-- CS4N
CS3-- CS3N
CS2-- CS2N
CS1-- CS1N
4
Phase 4
Out
Phase 3
Out
Phase 2
Out
Phase 1
Out
Phase 4 CS
input
Phase 3 CS
input
Phase 2 CS
input
Phase 1 CS
input
3
Tie to
GND
Phase 3
Out
Phase 2
Out
Phase 1
Out
Tie to
VCCP
Phase 3 CS
input
Phase 2 CS
input
Phase 1 CS
input
2
Tie to
GND
Phase 2
Out
Tie to
GND
Phase 1
Out
Tie to
VCCP
Phase 2 CS
input
Tie to
VCCP
Phase 1 CS
input
MAXIMUM RATINGS
ELECTRICAL INFORMATION
Pin Symbol
VMAX
VMIN
ISOURCE
ISINK
COMP
5.5 V
--0.3 V
10 mA
10 mA
VDRP
5.5 V
--0.3 V
5 mA
5 mA
V+
5.5 V
GND – 300 mV
1 mA
1 mA
V–
GND + 300 mV
GND – 300 mV
1 mA
1 mA
DIFFOUT
5.5 V
--0.3 V
20 mA
20 mA
VR_RDY
5.5 V
--0.3 V
N/A
20 mA
VCC
7.0 V
--0.3 V
N/A
10 mA
--0.3 V
1 mA
N/A
ROSC
5.5 V
IMON Output
1.1 V
All Other Pins
5.5 V
--0.3 V
*All signals referenced to AGND unless otherwise noted.
THERMAL INFORMATION
Rating
Thermal Characteristic, QFN Package (Note 1)
Symbol
Value
Unit
RθJA
34
C/W
Operating Junction Temperature Range (Note 2)
TJ
0 to 125
C
Operating Ambient Temperature Range
TA
0 to +85
C
Maximum Storage Temperature Range
TSTG
--55 to +150
C
Moisture Sensitivity Level, QFN Package
MSL
1
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
*The maximum package power dissipation must be observed.
1. JESD 51--5 (1S2P Direct--Attach Method) with 0 LFM.
2. JESD 51--7 (1S2P Direct--Attach Method) with 0 LFM.
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8
NCP5392T
ELECTRICAL CHARACTERISTICS
(Unless otherwise stated: 0C < TA < 85C; 4.75 V < VCC < 5.25 V; All DAC Codes; CVCC = 0.1 mF)
Parameter
Test Conditions
Min
Typ
Max
Unit
ERROR AMPLIFIER
--200
Input Bias Current (Note 3)
Noninverting Voltage Range (Note 3)
0
200
nA
1.3
3
V
1.0
mV
Input Offset Voltage (Note 3)
V+ = V-- = 1.1 V
--1.0
--
Open Loop DC Gain
CL = 60 pF to GND,
RL = 10 KΩ to GND
--
100
Open Loop Unity Gain Bandwidth
CL = 60 pF to GND,
RL = 10 KΩ to GND
--
10
--
MHz
Open Loop Phase Margin
CL = 60 pF to GND,
RL = 10 KΩ to GND
--
80
--

Slew Rate
ΔVin = 100 mV, G = -- 10 V/V,
ΔVout = 1.5 V – 2.5 V,
CL = 60 pF to GND,
DC Load = 125 mA to GND
--
5
--
V/ms
Maximum Output Voltage
ISOURCE = 2.0 mA
3.5
--
--
V
Minimum Output Voltage
ISINK = 0.2 mA
--
--
50
mV
Output source current (Note 3)
Vout = 3.5 V
2
--
--
mA
Output sink current (Note 3)
Vout = 1.0 V
2
--
--
mA
dB
DIFFERENTIAL SUMMING AMPLIFIER
VSN Input Bias Current
VSN Voltage = 0 V
30
mA
VSP Input Resistance
DRVON = Low
DRVON = High
1.5
17
kΩ
VSP Input Bias Voltage
DRVON = Low
DRVON = High
0.09
0.66
V
Input Voltage Range (Note 3)
--0.3
--
3.0
V
--
10
--
MHz
1.025
V/V
--3 dB Bandwidth
CL = 80 pF to GND,
RL = 10 KΩ to GND
Closed Loop DC Gain VS to Diffout
VS+ to VS-- = 0.5 to 1.6 V
0.98
1.0
Maximum Output Voltage
ISOURCE = 2 mA
3.0
--
--
V
Minimum Output Voltage
ISINK = 2 mA
--
--
0.5
V
Output source current (Note 3)
Vout = 3 V
2.0
--
--
mA
Output sink current (Note 3)
Vout = 0.5 V
2.0
--
--
mA
--
1.30
--
V
INTERNAL OFFSET VOLTAGE
Offset Voltage to the (+) Pin of the
Error Amp and the VDRP pin
VDROOP AMPLIFIER
--200
Input Bias Current (Note 3)
Non--inverting Voltage Range (Note 3)
0
200
nA
1.3
3
V
4.0
mV
Input Offset Voltage (Note 3)
V+ = V-- = 1.1 V
--4.0
--
Open Loop DC Gain
CL = 20 pF to GND including
ESD, RL = 1 kΩ to GND
--
100
Open Loop Unity Gain Bandwidth
CL = 20 pF to GND including
ESD, RL = 1 kΩ to GND
--
10
--
MHz
Slew Rate
CL = 20 pF to GND including
ESD, RL = 1 kΩ to GND
--
5
--
V/ms
Maximum Output Voltage
ISOURCE = 4.0 mA
3
--
--
V
Minimum Output Voltage
ISINK = 1.0 mA
--
--
1
V
Output source current (Note 3)
Vout = 3.0 V
4
--
--
mA
Output sink current (Note 3)
Vout = 1.0 V
1
--
--
mA
3. Guaranteed by design, not tested in production.
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9
dB
NCP5392T
ELECTRICAL CHARACTERISTICS
(Unless otherwise stated: 0C < TA < 85C; 4.75 V < VCC < 5.25 V; All DAC Codes; CVCC = 0.1 mF)
Parameter
Test Conditions
Min
Typ
Max
Unit
CSSUM AMPLIFIER
Current Sense Input to CSSUM Gain
--60 mV < CS < 60 mV
--4.00
--3.88
--3.76
V/V
Current Sense Input to CSSUM --3 dB
Bandwidth
CL = 10 pF to GND,
RL = 10 kΩ to GND
--
4
--
MHz
Current Sense Input to CSSUM
Output Slew Rate
ΔVin = 25 mV, CL = 10 pF to
GND, Load = 1 k to 1.3 V
--
4
--
V/s
Current Summing Amp Output Offset
Voltage
CSx – CSNx = 0, CSx = 1.1 V
--15
--
+15
mV
Maximum CSSUM Output Voltage
CSx – CSxN = --0.15 V
(All Phases) ISOURCE = 1 mA
3.0
--
--
V
Minimum CSSUM Output Voltage
CSx – CSxN = 0.066 V
(All Phases) ISINK = 1 mA
--
--
0.3
V
Output source current (Note 3)
Vout = 3.0 V
1
--
--
mA
Output sink current (Note 3)
Vout = 0.3 V
1
--
--
mA
Enable High Input Leakage Current
External 1 K Pullup to 3.3 V
--
--
1.0
mA
Upper Threshold
VUPPER
--
650
770
mV
Lower Threshold
VLOWER
450
550
--
mV
Hysteresis
VUPPER -- VLOWER
--
100
--
mV
3.0
--
--
V
mA
PSI (Power Saving Control, Active Low)
DRVON
Output High Voltage
Sourcing 500 mA
Sourcing Current for Output High
VCC = 5 V
--
2.5
4.0
Output Low Voltage
Sinking 500 mA
--
--
0.7
V
2.5
--
--
mA
Sinking Current for Output Low
Delay Time
Propagation Delay from EN Low
to DRVON
--
10
--
ns
Rise Time
CL (PCB) = 20 pF, ΔVo = 10% to
90%
--
130
--
ns
Fall Time
CL (PCB) = 20 pF, ΔVo = 10% to
90%
--
10
--
ns
35
70
140
kΩ
--
--
2.0
V
Internal Pulldown Resistance
VCC Voltage when DRVON
Output Valid
CURRENT SENSE AMPLIFIERS
--
0
--
nA
Common Mode Input Voltage Range
(Note 3)
--0.3
--
2.0
V
Differential Mode Input Voltage Range
(Note 3)
--120
--
120
mV
Input Bias Current (Note 3)
CSx = CSxN = 1.4 V
Input Offset Voltage
CSx = CSxN = 1.1 V,
--1.0
--
1.0
mV
Current Sense Input to PWM Gain
(Note 3)
0 V < CSx -- CSxN < 0.1 V,
5.7
6.0
6.3
V/V
Current Sharing Offset CS1 to CSx
All VID codes
--2.5
--
2.5
mV
VDRP to IMON Gain
1.325 V< VDRP < 1.8 V
1.98
2
2.02
V/V
VDRP to IMON --3 dB Bandwidth
CL = 30 pF to GND,
RL = 100 kΩ to GND
--
4
Output Referred Offset Voltage
VDRP = 1.6 V, ISOURCE = 0 mA
8
23
38
mV
Minimum Output Voltage
VDRP = 1.2 V, ISINK = 100 mA
--
--
0.11
V
IMON
3. Guaranteed by design, not tested in production.
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10
MHz
NCP5392T
ELECTRICAL CHARACTERISTICS
(Unless otherwise stated: 0C < TA < 85C; 4.75 V < VCC < 5.25 V; All DAC Codes; CVCC = 0.1 mF)
Parameter
Test Conditions
Min
Typ
Max
Unit
IMON
Output source current (Note 3)
Vout = 1 V
300
--
--
mA
Output sink current (Note 3)
Vout = 0.3 V
300
--
--
mA
Maximum Clamp Voltage
VDRP Voltage = 2 V,
RLOAD = 100 k
--
--
1.15
V
100
--
1000
kHz
ROSC = 49.9 kΩ
200
--
224
kHz
ROSC = 24.9 kΩ
374
--
414
ROSC = 10 kΩ
800
--
978
ROSC = 49.9 kΩ
191
--
234
ROSC = 24.9 kΩ
354
--
434
ROSC = 10 kΩ
755
--
1000
1.95
2.01
2.065
V
OSCILLATOR
Switching Frequency Range (Note 3)
Switching Frequency Accuracy 2-- or
4--Phase
Switching Frequency Accuracy
3--Phase
ROSC Output Voltage
kHz
MODULATORS (PWM Comparators)
Minimum Pulse Width
FSW = 800 KHz
--
30
--
ns
Propagation Delay
20 mV of Overdrive
--
10
--
ns
0% Duty Cycle
COMP Voltage when the PWM
Outputs Remain LO
--
1.3
--
V
100% Duty Cycle
COMP Voltage when the PWM
Outputs Remain HI
--
2.3
--
V
PWM Ramp Duty Cycle Matching
Between Any Two Phases
--
90
--
%
PWM Phase Angle Error (Note 3)
Between Adjacent Phases
15
--
15

VR_RDY (POWER GOOD) OUTPUT
VR_RDY Output Saturation Voltage
IPGD = 10 mA,
--
--
0.4
V
VR_RDY Rise Time (Note 3)
External Pullup of 1 kΩ to 1.25
V, CTOT = 45 pF, ΔVo = 10% to
90%
--
100
150
ns
VR_RDY Output Voltage at Powerup
(Note 3)
VR_RDY Pulled up to 5 V via
2 kΩ, tR(VCC)  3 x tR(5V)
100 ms  tR(VCC)  20 ms
--
--
1.0
V
VR_RDY High – Output Leakage
Current (Note 3)
VR_RDY = 5.5 V via 1 K
--
--
0.2
mA
VR_RDY Upper Threshold Voltage
VCore Increasing, DAC = 1.3 V
--
310
270
mV
Below
DAC
VR_RDY Lower Threshold Voltage
VCore Decreasing
DAC = 1.3 V
410
370
VR_RDY Rising Delay
VCore Increasing
--
500
--
ms
VR_RDY Falling Delay
VCore Decreasing
--
5
--
ms
3.0
--
--
V
V
mV
Below
DAC
PWM OUTPUTS
Output High Voltage
Sourcing 500 mA
Mid Output Voltage
1.4
1.5
1.6
Output Low Voltage
Sinking 500 mA
--
--
0.7
V
Delay + Fall Time (Note 3)
CL (PCB) = 50 pF,
ΔVo = VCC to GND
--
10
15
ns
Delay + Rise Time (Note 3)
CL (PCB) = 50 pF,
ΔVo = GND to VCC
--
10
15
ns
3. Guaranteed by design, not tested in production.
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11
NCP5392T
ELECTRICAL CHARACTERISTICS
(Unless otherwise stated: 0C < TA < 85C; 4.75 V < VCC < 5.25 V; All DAC Codes; CVCC = 0.1 mF)
Parameter
Test Conditions
Min
Typ
Max
Unit
Resistance to VCC (HI) or GND
(LO)
--
75
--
Ω
Gate Pin Source Current
60
80
150
mA
Gate Pin Threshold Voltage
210
240
265
mV
Phase Detect Timer
15
20
27
ms
1.0
--
1.5
ms
400
500
600
ms
PWM OUTPUTS
Output Impedance – HI or LO State
2/3/4-- PhASE DETECTION
DIGITAL SOFT-- START
Soft--Start Ramp Time
DAC = 0 to DAC = 1.1 V
VR11 Vboot time
VID7/VR11 INPUT
VID Upper Threshold
VUPPER
--
650
800
mV
VID Lower Threshold
VLOWER
300
550
--
mV
VID Hysteresis
VUPPER -- VLOWER
--
100
VR11 Input Bias Current (Note 3)
Delay before Latching VID Change
(VID De--Skewing) (Note 3)
Measured from the edge of the
1st VID change
200
--
VID7 Valid Range
--
mV
200
nA
300
ns
3.33
V
ENABLE INPUT
Enable High Input Leakage Current
(Note 3)
Pullup to 1.3 V
VR11 Rising Threshold
VR11 Falling Threshold
--
--
200
nA
--
650
770
mV
450
550
--
mV
--
100
--
mV
5.0
ms
VR11 Total Hysteresis
Rising-- Falling Threshold
Enable Delay Time
Measure Time from Enable
Transitioning HI to when Output
Begins
2.5
ILIM to VDRP Gain
Between VDRP -- VDFB = 450 mV
and VDRP -- VDFB = 650 mV
0.95
1.0
1.05
V/V
ILIM to VDRP Gain in PSI 4 phase
Between VDRP -- VDFB = 450 mV
and VDRP -- VDFB = 650 mV
--
0.25
--
V/V
ILIM to VDRP Gain in PSI 3 phase
Between VDRP -- VDFB = 450 mV
and VDRP -- VDFB = 650 mV
--
0.33
--
V/V
ILIM to VDRP Gain in PSI 2 phase
Between VDRP -- VDFB = 450 mV
and VDRP -- VDFB = 650 mV
--
0.5
--
V/V
ILIM Offset
VDRP -- VDFB = 520 mV
--50
0
50
mV
--
100
--
ns
DAC +150
DAC +185
DAC +200
mV
(1.6 V DAC)
+200
mV
CURRENT LIMIT
Delay
OVERVOLTAGE PROTECTION
VR11 Overvoltage Threshold
VR11 PSI Overvoltage Threshold
(Note 3)
(1.6 V DAC)
+150
Delay
100
ns
UNDERVOLTAGE PROTECTION
VCC UVLO Start Threshold
4
4.25
4.5
VCC UVLO Stop Threshold
3.8
4.05
4.3
VCC UVLO Hysteresis
200
3. Guaranteed by design, not tested in production.
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12
V
V
mV
NCP5392T
ELECTRICAL CHARACTERISTICS
(Unless otherwise stated: 0C < TA < 85C; 4.75 V < VCC < 5.25 V; All DAC Codes; CVCC = 0.1 mF)
Parameter
Test Conditions
Min
Typ
Max
Unit
VR_HOT Upper Voltage Threshold
19.6 kΩ P.U. to VCC, 68 kΩNTC,
β = 3740
0.257
0.268
0.280
VCC
VR_HOT Lower Voltage Threshold
19.6 kΩ P.U. to VCC, 68 KΩNTC,
β = 3740
0.316
0.329
0.343
VCC
VR_HOT Output Voltages at
Power--up (Note 3)
External Pull--up resistor of 2 KΩ
to 5 V, tR_VCC  3 x tR_5 V,
100 ms  tR_VCC  20 ms
--
--
1.0
V
VR_HOT Saturation Output Voltage
ISINK = 4 mA
--
--
0.3
V
VR_HOT Output Leakage Current
--
--
1
mA
NTC Pin Bias Current
--
--
1
mA
VR_HOT
12VMON UVLO
12VMON (High Threshold)
VCC Valid
--
0.77
0.82
V
12VMON (Low Threshold)
VCC Valid
0.66
0.68
--
V
Output Source Current
VOUT = 3 V
0.25
mA
Output Sink Current
VOUT = 0.3 V
1.5
mA
Max Output Voltage (Note 3)
Isource = 2 mA
3
V
Min Output Voltage (Note 3)
Isink = 2 mA
DAC (FEED FORWARD FUNCTION)
0.5
V
VRM 11 DAC
11
--
16.5
mV/ms
1.0 V < DAC < 1.6 V
0.8 V < DAC < 1.0 V
0.5 V < DAC < 0.8 V
----
----
0.5
5
8
%
mV
mV
EN Low, No PWM
--
15
30
mA
Positive DAC Slew Rate
System Voltage Accuracy
(DAC Value has a 19 mV Offset Over
the Output Value)
VCC
VCC Operating Current
3. Guaranteed by design, not tested in production.
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13
NCP5392T
Table 1. VRM11 VID Codes
VID7
800 mV
VID6
400 mV
VID5
200 mV
VID4
100 mV
VID3
50 mV
VID2
25 mV
VID1
12.5 mV
VID0
6.25 mV
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
0
0
0
0
0
0
1
0
1.60000
02
0
0
0
0
0
0
1
1
1.59375
03
0
0
0
0
0
1
0
0
1.58750
04
0
0
0
0
0
1
0
1
1.58125
05
0
0
0
0
0
1
1
0
1.57500
06
0
0
0
0
0
1
1
1
1.56875
07
0
0
0
0
1
0
0
0
1.56250
08
0
0
0
0
1
0
0
1
1.55625
09
0
0
0
0
1
0
1
0
1.55000
0A
0
0
0
0
1
0
1
1
1.54375
0B
0
0
0
0
1
1
0
0
1.53750
0C
0
0
0
0
1
1
0
1
1.53125
0D
0
0
0
0
1
1
1
0
1.52500
0E
0
0
0
0
1
1
1
1
1.51875
0F
0
0
0
1
0
0
0
0
1.51250
10
0
0
0
1
0
0
0
1
1.50625
11
0
0
0
1
0
0
1
0
1.50000
12
0
0
0
1
0
0
1
1
1.49375
13
0
0
0
1
0
1
0
0
1.48750
14
0
0
0
1
0
1
0
1
1.48125
15
0
0
0
1
0
1
1
0
1.47500
16
0
0
0
1
0
1
1
1
1.46875
17
0
0
0
1
1
0
0
0
1.46250
18
0
0
0
1
1
0
0
1
1.45625
19
0
0
0
1
1
0
1
0
1.45000
1A
0
0
0
1
1
0
1
1
1.44375
1B
0
0
0
1
1
1
0
0
1.43750
1C
0
0
0
1
1
1
0
1
1.43125
1D
0
0
0
1
1
1
1
0
1.42500
1E
0
0
0
1
1
1
1
1
1.41875
1F
0
0
1
0
0
0
0
0
1.41250
20
0
0
1
0
0
0
0
1
1.40625
21
0
0
1
0
0
0
1
0
1.40000
22
0
0
1
0
0
0
1
1
1.39375
23
0
0
1
0
0
1
0
0
1.38750
24
0
0
1
0
0
1
0
1
1.38125
25
0
0
1
0
0
1
1
0
1.37500
26
0
0
1
0
0
1
1
1
1.36875
27
0
0
1
0
1
0
0
0
1.36250
28
0
0
1
0
1
0
0
1
1.35625
29
0
0
1
0
1
0
1
0
1.35000
2A
0
0
1
0
1
0
1
1
1.34375
2B
0
0
1
0
1
1
0
0
1.33750
2C
0
0
1
0
1
1
0
1
1.33125
2D
0
0
1
0
1
1
1
0
1.32500
2E
0
0
1
0
1
1
1
1
1.31875
2F
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14
Voltage
(V)
HEX
00
01
NCP5392T
Table 1. VRM11 VID Codes
VID7
800 mV
VID6
400 mV
VID5
200 mV
VID4
100 mV
VID3
50 mV
VID2
25 mV
VID1
12.5 mV
VID0
6.25 mV
Voltage
(V)
HEX
0
0
1
1
0
0
0
0
1.31250
30
0
0
1
1
0
0
0
1
1.30625
31
0
0
1
1
0
0
1
0
1.30000
32
0
0
1
1
0
0
1
1
1.29375
33
0
0
1
1
0
1
0
0
1.28750
34
0
0
1
1
0
1
0
1
1.28125
35
0
0
1
1
0
1
1
0
1.27500
36
0
0
1
1
0
1
1
1
1.26875
37
0
0
1
1
1
0
0
0
1.26250
38
0
0
1
1
1
0
0
1
1.25625
39
0
0
1
1
1
0
1
0
1.25000
3A
0
0
1
1
1
0
1
1
1.24375
3B
0
0
1
1
1
1
0
0
1.23750
3C
0
0
1
1
1
1
0
1
1.23125
3D
0
0
1
1
1
1
1
0
1.22500
3E
0
0
1
1
1
1
1
1
1.21875
3F
0
1
0
0
0
0
0
0
1.21250
40
0
1
0
0
0
0
0
1
1.20625
41
0
1
0
0
0
0
1
0
1.20000
42
0
1
0
0
0
0
1
1
1.19375
43
0
1
0
0
0
1
0
0
1.18750
44
0
1
0
0
0
1
0
1
1.18125
45
0
1
0
0
0
1
1
0
1.17500
46
0
1
0
0
0
1
1
1
1.16875
47
0
1
0
0
1
0
0
0
1.16250
48
0
1
0
0
1
0
0
1
1.15625
49
0
1
0
0
1
0
1
0
1.15000
4A
0
1
0
0
1
0
1
1
1.14375
4B
0
1
0
0
1
1
0
0
1.13750
4C
0
1
0
0
1
1
0
1
1.13125
4D
0
1
0
0
1
1
1
0
1.12500
4E
0
1
0
0
1
1
1
1
1.11875
4F
0
1
0
1
0
0
0
0
1.11250
50
0
1
0
1
0
0
0
1
1.10625
51
0
1
0
1
0
0
1
0
1.10000
52
0
1
0
1
0
0
1
1
1.09375
53
0
1
0
1
0
1
0
0
1.08750
54
0
1
0
1
0
1
0
1
1.08125
55
0
1
0
1
0
1
1
0
1.07500
56
0
1
0
1
0
1
1
1
1.06875
57
0
1
0
1
1
0
0
0
1.06250
58
0
1
0
1
1
0
0
1
1.05625
59
0
1
0
1
1
0
1
0
1.05000
5A
0
1
0
1
1
0
1
1
1.04375
5B
0
1
0
1
1
1
0
0
1.03750
5C
0
1
0
1
1
1
0
1
1.03125
5D
0
1
0
1
1
1
1
0
1.02500
5E
0
1
0
1
1
1
1
1
1.01875
5F
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15
NCP5392T
Table 1. VRM11 VID Codes
VID7
800 mV
VID6
400 mV
VID5
200 mV
VID4
100 mV
VID3
50 mV
VID2
25 mV
VID1
12.5 mV
VID0
6.25 mV
Voltage
(V)
HEX
0
1
1
0
0
0
0
0
1.01250
60
0
1
1
0
0
0
0
1
1.00625
61
0
1
1
0
0
0
1
0
1.00000
62
0
1
1
0
0
0
1
1
0.99375
63
0
1
1
0
0
1
0
0
0.98750
64
0
1
1
0
0
1
0
1
0.98125
65
0
1
1
0
0
1
1
0
0.97500
66
0
1
1
0
0
1
1
1
0.96875
67
0
1
1
0
1
0
0
0
0.96250
68
0
1
1
0
1
0
0
1
0.95625
69
0
1
1
0
1
0
1
0
0.95000
6A
0
1
1
0
1
0
1
1
0.94375
6B
0
1
1
0
1
1
0
0
0.93750
6C
0
1
1
0
1
1
0
1
0.93125
6D
0
1
1
0
1
1
1
0
0.92500
6E
0
1
1
0
1
1
1
1
0.91875
6F
0
1
1
1
0
0
0
0
0.91250
70
0
1
1
1
0
0
0
1
0.90625
71
0
1
1
1
0
0
1
0
0.90000
72
0
1
1
1
0
0
1
1
0.89375
73
0
1
1
1
0
1
0
0
0.88750
74
0
1
1
1
0
1
0
1
0.88125
75
0
1
1
1
0
1
1
0
0.87500
76
0
1
1
1
0
1
1
1
0.86875
77
0
1
1
1
1
0
0
0
0.86250
78
0
1
1
1
1
0
0
1
0.85625
79
0
1
1
1
1
0
1
0
0.85000
7A
0
1
1
1
1
0
1
1
0.84375
7B
0
1
1
1
1
1
0
0
0.83750
7C
0
1
1
1
1
1
0
1
0.83125
7D
0
1
1
1
1
1
1
0
0.82500
7E
0
1
1
1
1
1
1
1
0.81875
7F
1
0
0
0
0
0
0
0
0.81250
80
1
0
0
0
0
0
0
1
0.80625
81
1
0
0
0
0
0
1
0
0.80000
82
1
0
0
0
0
0
1
1
0.79375
83
1
0
0
0
0
1
0
0
0.78750
84
1
0
0
0
0
1
0
1
0.78125
85
1
0
0
0
0
1
1
0
0.77500
86
1
0
0
0
0
1
1
1
0.76875
87
1
0
0
0
1
0
0
0
0.76250
88
1
0
0
0
1
0
0
1
0.75625
89
1
0
0
0
1
0
1
0
0.75000
8A
1
0
0
0
1
0
1
1
0.74375
8B
1
0
0
0
1
1
0
0
0.73750
8C
1
0
0
0
1
1
0
1
0.73125
8D
1
0
0
0
1
1
1
0
0.72500
8E
1
0
0
0
1
1
1
1
0.71875
8F
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16
NCP5392T
Table 1. VRM11 VID Codes
VID7
800 mV
VID6
400 mV
VID5
200 mV
VID4
100 mV
VID3
50 mV
VID2
25 mV
VID1
12.5 mV
VID0
6.25 mV
Voltage
(V)
HEX
1
0
0
1
0
0
0
0
0.71250
90
1
0
0
1
0
0
0
1
0.70625
91
1
0
0
1
0
0
1
0
0.70000
92
1
0
0
1
0
0
1
1
0.69375
93
1
0
0
1
0
1
0
0
0.68750
94
1
0
0
1
0
1
0
1
0.68125
95
1
0
0
1
0
1
1
0
0.67500
96
1
0
0
1
0
1
1
1
0.66875
97
1
0
0
1
1
0
0
0
0.66250
98
1
0
0
1
1
0
0
1
0.65625
99
1
0
0
1
1
0
1
0
0.65000
9A
1
0
0
1
1
0
1
1
0.64375
9B
1
0
0
1
1
1
0
0
0.63750
9C
1
0
0
1
1
1
0
1
0.63125
9D
1
0
0
1
1
1
1
0
0.62500
9E
1
0
0
1
1
1
1
1
0.61875
9F
1
0
1
0
0
0
0
0
0.61250
A0
1
0
1
0
0
0
0
1
0.60625
A1
1
0
1
0
0
0
1
0
0.60000
A2
1
0
1
0
0
0
1
1
0.59375
A3
1
0
1
0
0
1
0
0
0.58750
A4
1
0
1
0
0
1
0
1
0.58125
A5
1
0
1
0
0
1
1
0
0.57500
A6
1
0
1
0
0
1
1
1
0.56875
A7
1
0
1
0
1
0
0
0
0.56250
A8
1
0
1
0
1
0
0
1
0.55625
A9
1
0
1
0
1
0
1
0
0.55000
AA
1
0
1
0
1
0
1
1
0.54375
AB
1
0
1
0
1
1
0
0
0.53750
AC
1
0
1
0
1
1
0
1
0.53125
AD
1
0
1
0
1
1
1
0
0.52500
AE
1
0
1
0
1
1
1
1
0.51875
AF
1
0
1
1
0
0
0
0
0.51250
B0
1
0
1
1
0
0
0
1
0.50625
B1
1
0
1
1
0
0
1
0
0.50000
B2
1
1
1
1
1
1
1
0
OFF
FE
1
1
1
1
1
1
1
1
OFF
FF
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NCP5392T
FUNCTIONAL DESCRIPTION
General
corresponding gate output (G1, G2, G3, or G4). If a phase
is unused, the differential inputs to that phase’s current
sense amplifier must be shorted together and connected to
the output as shown in the 2-- and 3-- phase Application
Schematics.
The current signals sensed from inductor DCR are fed into
a summing amplifier to have a summed--up output (CSSUM).
Signal of CSSUM combines information of total current of all
phases in operation.
The outputs of current sense amplifiers control three
functions. First, the summing current signal (CCSUM) of
all phases will go through DROOP amplifier and join the
voltage feedback loop for output voltage positioning.
Second, the output signal from DROOP amplifier also goes
to ILIM amplifier to monitor the output current limit.
Finally, the individual phase current contributes to the
current balance of all phases by offsetting their ramp
signals of PWM comparators.
The NCP5392T provides up to four-- phase buck solution
which combines differential voltage sensing, differential
phase current sensing, and adaptive voltage positioning to
provide accurately regulated power necessary for both
Intel VR11.1 CPU power system. NCP5392T has been
designed to work with the NCP5359 driver.
Remote Output Sensing Amplifier(RSA)
A true differential amplifier allows the NCP5392T to
measure Vcore voltage feedback with respect to the Vcore
ground reference point by connecting the Vcore reference
point to VSP, and the Vcore ground reference point to VSN.
This configuration keeps ground potential differences
between the local controller ground and the Vcore ground
reference point from affecting regulation of Vcore between
Vcore and Vcore ground reference points. The RSA also
subtracts the DAC (minus VID offset) voltage, thereby
producing an unamplified output error voltage at the
DIFFOUT pin. This output also has a 1.3 V bias voltage as
the floating ground to allow both positive and negative
error voltages.
Thermal Compensation Amplifier with VDRP and VDFB
Pins
Thermal compensation amplifier is an internal amplifier
in the path of droop current feedback for additional
adjustment of the gain of summing current and temperature
compensation. The way thermal compensation is
implemented separately ensures minimum interference to
the voltage loop compensation network.
Precision Programmable DAC
A precision programmable DAC is provided and system
trimmed. This DAC has 0.5% accuracy over the entire
operating temperature range of the part. The DAC can be
programmed to support Intel VR11 VID code
specifications.
Oscillator and Triangle Wave Generator
A programmable precision oscillator is provided. The
oscillator’s frequency is programmed by the resistance
connected from the ROSC pin to ground. The user will
usually form this resistance from two resistors in order to
create a voltage divider that uses the ROSC output voltage
as the reference for creating the current limit setpoint
voltage. The oscillator frequency range is 100 kHz per
phase to 1.0 MHz per phase. The oscillator generates up to
4 symmetrical triangle waveforms with amplitude between
1.3 V and 2.3 V. The triangle waves have a phase delay
between them such that for 2-- , 3-- and 4-- phase operation
the PWM outputs are separated by 180, 120, and 90 angular
degrees, respectively.
High Performance Voltage Error Amplifier
The error amplifier is designed to provide high slew rate
and bandwidth. Although not required when operating as the
controller of a voltage regulator, a capacitor from COMP to
VFB is required for stable unity gain test configurations.
Gate Driver Outputs and 2/3/4 Phase Operation
The part can be configured to run in 2-- , 3-- , or 4-- phase
mode. In 2-- phase mode, phases 1 and 3 should be used to
drive the external gate drivers as shown in the 2-- phase
Applications Schematic, G2 and G4 must be grounded. In
3-- phase mode, gate output G4 must be grounded as shown
in the 3-- phase Applications Schematic. In 4-- phase mode
all 4 gate outputs are used as shown in the 4-- phase
Applications Schematic. The Current Sense inputs of
unused channels should be connected to VCCP shown in
the Application Schematics. Please refer to table “PIN
CONNECTIONS vs. PHASE COUNTS” for details.
PWM Comparators with Hysteresis
Four PWM comparators receive an error signal at their
noninverting input. Each comparator receives one of the
triangle waves at its inverting output. The output of each
comparator generates the PWM outputs G1, G2, G3, and G4.
During steady state operation, the duty cycle will center
on the valley of the triangle waveform, with steady state
duty cycle calculated by Vout/Vin. During a transient event,
both high and low comparator output transitions shift phase
to the points where the error signal intersects the down and
up ramp of the triangle wave.
Differential Current Sense Amplifiers and Summing
Amplifier
Four differential amplifiers are provided to sense the
output current of each phase. The inputs of each current
sense amplifier must be connected across the current
sensing element of the phase controlled by the
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NCP5392T
Power Saving Mode
reads the VID pins to determine the DAC setting. Then
ramps Vcore to the final DAC setting at the Dynamic VID
slew rate of up to 12.5 mV/mS. Typical VR11 soft-- start
sequences are shown in the following graphs (Figure 9 and
10).
Upon receiving PSI low command, the NCP5392T
enters power saving mode with PWM signals varying
between high and mid level to allow diode emulation. The
device is also forced into RPM mode.
PROTECTION FEATURES
APPLICATION INFORMATION
The NCP5392T demo board for the NCP5392T is
available by request. It is configured as a four phase
solution with decoupling designed to provide a 1 mΩ load
line under a 100 A step load.
Undervoltage Lockout (VCC) and 12VMON
An undervoltage lockout (UVLO) senses the VCC input
directly. 12 V UVLO senses the 12 V power supply by
connecting it to the 12VMON pin through an appropriate
resistor divider. During power-- up, both the VCC input and
12VMON are monitored, and the PWM outputs and the
soft-- start circuit are disabled until both input voltages
exceed the threshold voltages of their individual UVLO
comparators. The UVLO comparators both incorporate
hysteresis to avoid chattering. The second function of
12VMON pin is to provide a feed-- forward input voltage
information when the device works in RPM mode.
Startup Procedure
Start by installing the test tool software. It is best to
power the test tool from a separate ATX power supply. The
test tool should be set to a valid VID code of 0.5 V or above
in order for the controller to start. Consult the VTT help
manual for more detailed instruction.
Step Load Testing
The VTT tool is used to generate the di/dt step load.
Select the dynamic loading option in the VTT test tool
software. Set the desired step load size, frequency, duty,
and slew rate. See Figure 6.
Overcurrent Shutdown
A programmable overcurrent function is incorporated
within the IC. A comparator and latch make up this
function. The inverting input of the comparator is
connected to the ILIM pin. The voltage at this pin sets the
maximum output current the converter can produce. The
ROSC pin provides a convenient and accurate reference
voltage from which a resistor divider can create the
overcurrent setpoint voltage. Although not actually
disabled, tying the ILIM pin directly to the ROSC pin sets
the limit above useful levels - effectively disabling
overcurrent shutdown. The comparator noninverting input
is the summed current information from the VDRP minus
offset voltage. The overcurrent latch is set when the current
information exceeds the voltage at the ILIM pin. The
outputs are pulled low, and the soft-- start is pulled low. The
outputs will remain disabled until the VCC voltage is
removed and re-- applied, or the ENABLE input is brought
low and then high.
Output Overvoltage and Undervoltage Protection and
Power Good Monitor
Figure 6. Typical Load Step Response
(full load, 35 A -- 100 A)
An output voltage monitor is incorporated. During
normal operation, if the output voltage is 180 mV (typical)
over the DAC voltage, the VR_RDY goes low, the DRVON
signal remains high, the PWM outputs are set low. The
outputs will remain disabled until the VCC voltage is
removed and reapplied. During normal operation, if the
output voltage falls more than 350 mV below the DAC
setting, the VR_RDY pin will be set low until the output
voltage rises.
Dynamic VID Testing
The VTT tool provides for VID stepping based on the
Intel Requirements. Select the Dynamic VID option.
Before enabling the test set the lowest VID to 0.5 V or
greater and set the highest VID to a value that is greater than
the lowest VID selection, then enable the test. See Figures
7 and 8.
Soft--Start
The VR11 mode ramps Vcore to 1.1 V boot voltage at a
fixed rate of 0.8 mV/mS, pauses at 1.1 V for around 500 mS,
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NCP5392T
Figure 7. 1.6 V to 0.5 V Dynamic VID response
Figure 9. VR11.1 Startup
Figure 8. Dynamic VID Settling Time Rising
(CH1: VID1, CH2: DAC, CH3:VCCP)
Figure 10. VR11.1 Biased Startup
Programming the Current Limit and the Oscillator
Frequency
DESIGN METHODOLOGY
The demo board is set for an operating frequency of
approximately 330 kHz. The ROSC pin provides a 2.0 V
reference voltage which is divided down with a resistor
divider and fed into the current limit pin ILIM. Then
calculate the individual RLIM1 and RLIM2 values for the
divider. The series resistors RLIM1 and RLIM2 sink
current from the ILIM pin to ground. This current is
internally mirrored into a capacitor to create an oscillator.
The period is proportional to the resistance and frequency
is inversely proportional to the total resistance. The total
resistance may be estimated by Equation 1. This equation
is valid for the individual phase frequency in both three and
four phase mode.
Decoupling the VCC Pin on the IC
An RC input filter is required as shown in the VCC pin to
minimize supply noise on the IC. The resistor should be
sized such that it does not generate a large voltage drop
between 5 V supply and the IC.
Understanding Soft--Start
The controller supports standard VR11 startup routines.
The Vcore voltage ramps up to the 1.1 V boot voltage, with
a pause to capture the VID code then resume ramping to
target value based on internal slew rate limit. The initial
ramp rate was set to be 0.8 mV/mS.
R osc ≅ 20947 × F SW −1.1262
30.5 kΩ ≅ 20947 × 330
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20
−1.1262
(eq. 1)
NCP5392T
60
The current limit function is based on the total sensed
current of all phases multiplied by a controlled gain
(Acssum*Adrp). DCR sensed inductor current is a function
of the winding temperature. The best approach is to set the
maximum current limit based on expected average
maximum temperature of the inductor windings,
Rosc--kohm
50
40
30
DCR Tmax = DCR 25C(1 + 0.00393 ⋅ (T max − 25)) (eq. 2)
20
10
Calculation
Real
0
100
1000
Freq--kHz
Figure 11. ROSC vs. Frequency
For multiphase controller, the ripple current can be calculated as,
Ipp =
(V in − N ⋅ V out) ⋅ V out
L ⋅ F SW ⋅ V in
(eq. 3)
Therefore calculate the current limit voltage as below,
V LIMIT ≅ A CSSUM ⋅ A DRP ⋅ DCR Tmax ⋅ (I MIN_OCP ⋅ + 0.5 ⋅ Ipp)

V LIMIT ≅ A CSSUM ⋅ A DRP ⋅ DCR Tmax ⋅ I MIN_OCP ⋅ + 0.5 ⋅
(eq. 4)

(V in − N ⋅ V out) ⋅ V out
L ⋅ F SW ⋅ V in
In Equation 4, ACSSUM and ADRP are the gain of current summing amplifier and droop amplifier.
Acssum
I1
I2
I3
I4
Adrp
RNOR
RISO1
+
RSUM
RT2
-+
RISO1 and RISO2 are in series with RT2, the NTC
temperature sense resistor placed near inductor. RSUM is
the resistor connecting between pin VDFB and pin
CSSUM. If PSI = 1, PSI function is off, the current limit
follows the Equation 7; if PSI = 0, the power saving mode
will be enabled, COEpsi is a coefficient for the current
limiting related with power saving function (PSI), the
current limit can be calculated from Equation 8. COEpsi
value is one over the original phase count N. Refer to the
PSI and phase shedding section for more details.
RISO2
+
--
OCP
event
Ilim
Figure 12. ACSSUM and ADRP
As introduced before, VLIMIT comes from a resistor
divider connected to Rosc pin, thus,
V LIMIT = 2 V ⋅
R LIM2
⋅ COEpsi
R LIM1 + R LIM2
(eq. 5)
A CSSUM = −4
A DRP = −
R NOR ⋅ (R ISO1 + R ISO2 + R T2)
(eq. 6)
(R NOR + R ISO1 + R ISO2 + R T2) ⋅ R SUM
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NCP5392T
Final Equations for the Current Limit Threshold
Final equations are described based on two conditions: normal mode and PSI mode.
ILIMIT(normal) ≅
ILIMIT(PSI) ≅
2V⋅R LIM2
R LIM1+R LIM2
4⋅
R
⋅(R
+R
+R T2)
NOR
ISO1
ISO2
(R
+R
+R
+R T2)⋅R
NOR
ISO1
ISO2
SUM
⋅ DCR 25C(1 + 0.00393 ⋅ (T inductor − 25))
2V⋅R LIM2
R LIM1+R LIM2
4⋅
R
⋅(R
+R
+R T2)
NOR
ISO1
ISO2
(R
+R
+R
+R T2)⋅R
NOR
ISO1
ISO2
SUM
⋅ COEpsi
⋅ DCR 25C(1 + 0.00393 ⋅ (T inductor − 25))
− 0.5 ⋅
− 0.5 ⋅
(V in − N ⋅ V out) ⋅ V out
L ⋅ F SW ⋅ V in
(eq. 7)
(V in − V out) ⋅ V out
L ⋅ F SW ⋅ V in
(eq. 8)
Inductor Current Sensing Compensation
N is the number of phases involved in the circuit.
The inductors on the demo board have a DCR at 25C of
0.6 mΩ. Selecting the closest available values of 21.3 kΩ
for RLIM1 and 9.28 kΩ for RLIM2 yields a nominal
operating frequency of 330 kHz. Select RISO1 = 1 k, RISO2
= 1 k, RT2 = 10 K (25C), RNOR/RSUM = 2, (refer to
application diagram). That results to an approximate
current limit of 133 A at 100C for a four phase operation
and 131 A at 25C. The total sensed current can be
observed as a scaled voltage at the VDRP with a positive
no-- load offset of approximately 1.3 V.
The NCP5392T uses the inductor current sensing
method. An RC filter is selected to cancel out the
impedance from inductor and recover the current
information through the inductor’s DCR. This is done by
matching the RC time constant of the sensing filter to the
L/DCR time constant. The first cut approach is to use a 0.1
mF capacitor for C and then solve for R.
(eq. 9)
R sense(T) =
L
0.1 ⋅ mF ⋅ DCR 25C ⋅ (1 + 0.00393(T − 25))
Because the inductor value is a function of load and
inductor temperature final selection of R is best done
experimentally on the bench by monitoring the Vdroop pin
and performing a step load test on the actual solution.
Inductor Selection
When using inductor current sensing it is recommended
that the inductor does not saturate by more than 10% at
maximum load. The inductor also must not go into hard
saturation before current limit trips. The demo board
includes a four phase output filter using the T44-- 8 core
from Micrometals with 3 turns and a DCR target of 0.6 mΩ
@ 25C. Smaller DCR values can be used, however,
current sharing accuracy and droop accuracy decrease as
DCR decreases. Use the NCP5392T design aide for
regulation accuracy calculations for specific value of DCR.
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NCP5392T
Simple Average SPICE Model
A simple state average model shown in Figure 13 can be used to determine a stable solution and provide insight into the
control system.
GAIN = 1
{--2/3*4}
Voff
VRamp_min
1.3V
V3
12V
0
0
0
12
L
LBRD RBRD
2 DCR
1
1
2
{185e--9/4} {0.6E--3/4}
100p 0.75m
CBulk
{560e--6*6}
ESRBulk
{7e--3/6}
2
RSUM
1k
RDFB 22p
Voff
+ E1
+
-- -E
GAIN = {6}
2k
R8
1k
CDFB
1E3
C5
10.6p
Vdrp
RDAC
0
CH
RF
1E3
22p
2.2k
Unity Gain BW=15MHz
R12
5.11k
CF
1.8n
R6
50
CFB1
680P
ESLBulk
{3.5e--9/6}
ESRCer
{1.5e--3/18}
0Aac
2
0Adc
ESLCer
{1.5e--9/18}
1
1
CDAC
12n
RFB1
RFB
CCer
{22e--6*18}
69.8
I1 = 50
I2 = 110 I1
TD = 100u
TR = 50n
TF = 50n
Vout
PW = 100u
PER = 200u
0
VDAC
DC = 1.2V
AC = 0
TRAN = PULSE
(0 0.05 400u 5u 5u 500u 1000u)
0
1k
Voff
R11
Voffset
1.3V
C4 1k
10.6p
0
0
Vdrp
1k
R10
2k
R9
C6
1k 10.6p
Figure 13. NCP5392T Average SPICE Model
1E3
Voff
IMON
0
Compensation and Output Filter Design
If the required output filter and switching frequency are
significantly different, it’s best to use the available PSPICE
models to design the compensation and output filter from
scratch.
The design target for this demo board was 1.0 mΩ up to
2.0 MHz. The phase switching frequency is currently set to
330 kHz. It can easily be seen that the board impedance of
0.75 mΩ between the load and the bulk capacitance has a
large effect on the output filter. In this case the six 560 mF
bulk capacitors have an ESR of 7.0 mΩ. Thus the bulk ESR
plus the board impedance is 1.15 mΩ + 0.75 mΩ or
1.9 mΩ. The actual output filter impedance does not drop
to 1.0 mΩ until the ceramic breaks in at over 375 kHz. The
controller must provide some loop gain slightly less than
one out to a frequency in excess 300 kHz. At frequencies
below where the bulk capacitance ESR breaks with the
bulk capacitance, the DC-- DC converter must have
sufficiently high gain to control the output impedance
completely. Standard Type-- 3 compensation works well
with the NCP5392T.
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NCP5392T
Zout Open Loop
Zout Closed Loop
Open Loop Gain with Current Loop Closed
Voltage Loop Compensation Gain
80
60
40
20
dB
0
--20
--40
--60
1mOhm
--80
--100
100
1000
10000
100000
1000000
10000000
Frequency
Figure 14. NCP5392T Circuit Frequency Response
The goal is to compensate the system such that the
resulting gain generates constant output impedance from
DC up to the frequency where the ceramic takes over
holding the impedance below 1.0 mΩ. See the example of
the locations of the poles and zeros that were set to optimize
the model above.
By matching the following equations a good set of
starting compensation values can be found for a typical
mixed bulk and ceramic capacitor type output filter.
CH
RFB1 CFB1
I Bias
RDRP
RISO2
RT
RFB
Gain = 4
RSUM
-+
CSSUM
Amp
RSx
1.3 V
(eq. 11)
1
1
=
2π ⋅ C Cer ⋅ (RBRD + ESR Bulk)
2π ⋅ CFB1 ⋅ (RFB1 + RFB)
RL
RFB should be set to provide optimal thermal
compensation in conjunction with thermistor RT2, RISO1
and RISO2. With RFB set to 1.0 kΩ, RFB1 is usually set to
100 Ω for maximum phase boost, and the value of RF is
typically set to 3.0 kΩ.
CSx
+
--
Error
Amp
PWM
Comparator
1.3 V
RISO1
1
1
=
(eq. 10)
2π ⋅ (RBRD + ESR Bulk) ⋅ C Bulk
2π ⋅ CF ⋅ RF
-+
1.3 V
Droop
Amp
+
-RNOR
RF CF
+
+
--
+
Gain = 1
Figure 15. Droop Injection and Thermal
Compensation
RDRP determines the target output impedance by the
basic equation:
Droop Injection and Thermal Compensation
The VDRP signal is generated by summing the sensed
output currents for each phase. A droop amplifier is added
to adjust the total gain to approximately eight. VDRP is
externally summed into the feedback network by the
resistor RDRP. This introduces an offset which is
proportional to the output current thereby forcing a
controlled, resistive output impedance.
R ⋅ DCR ⋅ A CSSUM ⋅ A DRP
V out
= Z out = FB
(eq. 12)
Iout
R DRP
R DRP =
R FB ⋅ DCR ⋅ A CSSUM ⋅ A DRP
Z out
(eq. 13)
The value of the inductor’s DCR is a function of
temperature according to the Equation 14:
DCR (T) = DCR 25C ⋅ (1 + 0.00393 ⋅ (T − 25)) (eq. 14)
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NCP5392T
Actual DCR increases by temperature, the system can be
thermally compensated to cancel this effect to a great
degree by adding an NTC in parallel with RNOR to reduce
the droop gain as the temperature increases. The NTC
device is nonlinear. Putting a resistor in series with the NTC
helps make the device appear more linear with
Z out(T) =
temperature. The series resistor is split and inserted on both
sides of the NTC to reduce noise injection into the feedback
loop. The recommended total value for RISO1 plus RISO2 is
approximately 1.0 kΩ.
The output impedance varies with inductor temperature
by the equation:
R FB ⋅ DCR 25C ⋅ (1 + 0.00393 ⋅ (T − 25)) ⋅ A CSSUM ⋅ A DRP
R DRP
(eq. 15)
By including the NTC RT2 and the series isolation resistors the new equation becomes:
Z out(T) =
R FB ⋅ DCR 25C ⋅ (1 + 0.00393 ⋅ (T − 25)) ⋅ A CSSUM ⋅ (R
R
⋅(R
+R
+R T2)
NOR
ISO1
ISO2
+R
+R
+R T2)⋅R
NOR
ISO1
ISO2
SUM
Acssum
The typical equation of an NTC is based on a curve fit
Equation 17
RT2(T) = RT2 25C ⋅ e
β
1   1 
273+T
−
298
Adrp
RNOR
RISO1
I1
I2
I3
I4
(eq. 17)
The demo board use a 10 kΩ NTC with a β value of 3740.
Figure 16 shows the comparison of the compensated output
impedance and uncompensated output impedance varying
with temperature.
+
RSUM
RT2
-+
RISO2
+
--
OCP
event
Ilim
Imon
+
-Gain = 2
0.0013
Zout
Zout(uncomp)
0.0012
(eq. 16)
R DRP
0.001
Figure 17. IMON Circuit
0.0009
0.0008
1.05
0.0007
0.84
0.0006
25
45
65
85
105
Vimon--V
Ohm
0.0011
Celsius
Figure 16. Zout vs. Temperature
Vimon vs. Iout
0.63
0.42
0.21
IMON for Current Monitor
Since VDRP signal reflects the current information of all
phases. It can be fed into the IMON amplifier for current
monitoring as shown in Figure 17. IMON amplifier has a
fixed gain of 2 with an offset when VDRP is equal to 1.3 V,
the internal floating reference voltage. The IMON
amplifier will be saturated at an maximum output of 1.09 V
therefore the total gain of current should be carefully
considered to make the maximum load current indicated by
the IMON output. Figure 18 shows a typical of the relation
between IMON output and the load current.
0
0
10
20
30
40
50 60
Iout--A
70
80
90 100
Figure 18. IMON Output vs. Output Current
Power Saving Indicator (PSI) and Phase Shedding
VR11.1 requires the processor to provide an output
signal to the VR controller to indicate when the processor
is in a low power state. NCP5392T use the status of PSI pin
to decide if there is a need to change its operating state to
maximize efficiency at light loads. When PSI = 0, the PSI
function will be enabled, and VR system will be running at
a single phase power saving mode.
The PSI signal will de-- assert 1 ms prior to moving to a
normal power state.
At power saving mode, NCP5392T works with the
NCP5359 driver to represent diode emulation mode at light
load for further power saving.
When system switches on PSI function, an phase
shedding will be presented. Only one phase is active in the
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NCP5392T
emulation mode while other phases are shed. Figure 19
indicates a PSI-- on transition from a 3-- phase mode to a
single phase mode. While staying stable in PSI mode, the
PWM signal of phase 1 will vary from a mid-- state level
(1.5 V typical) to high level while other phases all go to
mid-- state level. Vice verse, when PSI signal goes high, the
system will go back to the original phase mode such as
shown in Figure 20.
impedance. The following equations can be used to find the
temperature trip points.
RT1(T) = RT1 25C ⋅ e β
1   1 
273+T
−
298
(eq. 18)
With a beta value of 3740, a 68 kΩ NTC resistor is
selected for RT1, RNTC1 is populated with 19.6 kΩ.
VR_HOT threshold is carefully selected to make sure when
board temperature is less than 92C.
VCC
RNTC1
NTC
+
RT1
VRHOT
OUT
-0.268 Vcc
0
0
Figure 21. VRHOT Circuit
OVP Improved Performance
The overvoltage protection threshold is not adjustable.
OVP protection is enabled as soon as soft-- start begins and
is disabled when part is disabled. When OVP is tripped, the
controller commands all four gate drivers to enable their
low side MOSFETs and VR_RDY transitions low. In order
to recover from an OVP condition, VCC must fall below the
UVLO threshold. See the state diagram for further details.
The OVP circuit monitors the output of DIFFOUT. If the
DIFFOUT signal reaches 180 mV (typical) above the
nominal 1.3 V offset the OVP will trip and VRRDY will be
pulled low, after eight consecutive OVP events are
detected, all PWMs will be latched. The DIFFOUT signal
is the difference between the output voltage and the DAC
voltage (minus 19 mV if in VR11.1 modes) plus the 1.3 V
internal offset. This results in the OVP tracking on the DAC
voltage even during a dynamic change in the VID setting
during operation.
Figure 19. PSI turns on, CH1: PWM1, CH2: PWM2,
CH3: PWM3, CH4: PSI
Figure 20. PSI turns off, CH1: PWM1, CH2: PWM2,
CH3: PWM3, CH4: PSI
VRHOT
Thermal monitoring circuit consists of one sensitive
comparator that compares the voltage on the NTC pin with
an internal voltage reference. VR_HOT is an open drain
type of output. In normal temperature, the voltage value on
NTC pin is higher than the internal reference, VR_HOT
will be low impedance. When the temperature is higher
than certain threshold, the VR_HOT will be high
Figure 22. VR11.1, 1.6 V OVP Event
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NCP5392T
Gate Driver and MOSFET Selection
Board Stackup and Board Layout
ON Semiconductor provides the NCP5359 as a
companion gate driver IC. The NCP5359 driver is
optimized to work with a range of MOSFETs commonly
used in CPU applications. The NCP5359 provides special
functionality including power saving mode operation and
is required for high performance dynamic VID operation.
Contact your local ON Semiconductor applications
engineer for MOSFET recommendations.
Close attention should be paid to the routing of the sense
traces and control lines that propagate away from the
controller IC. Routing should follow the demo board
example. For further information or layout review contact
ON Semiconductor.
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NCP5392T
SYSTEM TIMING DIAGRAM
12 V (Gate Driver)
UVLO
5 V (Controller)
UVLO
EN
3.5 ms
VID
Valid VID
DRVON
1 ms min
1.5 ms
500 ms
VSP--VSN
500 ms
VR_RDY
Figure 23. Normal Startup
UVLO
UVLO
EN
12 V (Gate Driver)
5 V (Controller)
POR
3.5 ms
DRVON
VID
Valid VID
1 ms min
1.5 ms
VSP--VSN
1 ms
500 ms
500 ms
VR_RDY
Figure 24. Driver UVLO Limited Startup
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NCP5392T
Diffout ~ 1.3 V
1
2
3
4
5
6
7
8
1
2
3
4
5
6
7
8
185 mV
VR_RDY
DRVON = High
VSP = VID -- 19 mV
185 mV
Figure 25. OVP Shutdown
Ilimit + 1.3
VDRP
VR_RDY
DRVON
Figure 26. Non--PSI Current Limit
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NCP5392T
PACKAGE DIMENSIONS
QFN40 6x6, 0.5P
CASE 488AR--01
ISSUE A
D
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ASME Y14.5M, 1994.
2. CONTROLLING DIMENSIONS: MILLIMETERS.
3. DIMENSION b APPLIES TO PLATED
TERMINAL AND IS MEASURED BETWEEN
0.25 AND 0.30mm FROM TERMINAL
4. COPLANARITY APPLIES TO THE EXPOSED
PAD AS WELL AS THE TERMINALS.
A B
PIN ONE
LOCATION
E
DIM
A
A1
A3
b
D
D2
E
E2
e
L
K
0.15 C
2X
TOP VIEW
0.15 C
2X
(A3)
0.10 C
A
0.08 C
40X
SIDE VIEW A1
C
D2
L
40X
11
6.30
4.20
40X
40X
21
10
SOLDERING FOOTPRINT*
SEATING
PLANE
K
20
MILLIMETERS
MIN
MAX
0.80
1.00
0.00
0.05
0.20 REF
0.18
0.30
6.00 BSC
4.00
4.20
6.00 BSC
4.00
4.20
0.50 BSC
0.30
0.50
0.20
-- -- --
0.65
EXPOSED PAD
1
E2
b
0.10 C A B
40X
0.05 C
4.20 6.30
1
30
40
31
e
36X
BOTTOM VIEW
40X
0.30
36X
0.50 PITCH
DIMENSIONS: MILLIMETERS
*For additional information on our Pb--Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any
liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental
damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over
time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under
its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body,
or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death
may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees,
subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of
personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part.
SCILLC is an Equal Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
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NCP5392T/D