3A High Efficiency Synchronous Regulator

NCP3126
3 A Synchronous PWM
Switching Converter
The NCP3126 is a flexible synchronous PWM Switching Buck
Regulator. The NCP3126 is capable of producing output voltages as
low as 0.8 V. The NCP3126 also incorporates voltage mode control.
To reduce the number of external components, a number of features
are internally set including switching frequency. The NCP3126 is
currently available in an SOIC−8 package.
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Features
•
•
•
•
•
•
•
•
•
•
1
4.5 V to 13.2 V Operating Input Voltage Range
85 mW High−Side, 65 mW Low−Side Switches
Output Voltage Adjustable to 0.8 V
3 A Continuous Output Current
Fixed 350 kHz PWM Operation
1.0% Initial Output Accuracy
75% Max Duty Ratio
Over−Load Protection
Programmable Current Limit
This is a Pb−Free Device
8
SOIC−8 NB
D SUFFIX
CASE 751
3126
A
L
Y
W
G
1
3126
ALYWXG
G
= Specific Device Code
= Assembly Location
= Wafer Lot
= Year
= Work Week
= Pb−Free Package
PIN CONNECTIONS
Typical Application
•
•
•
•
•
MARKING
DIAGRAM
NCP3126
Set Top Boxes
DVD Drives and HDD
LCD Monitors and TVs
Cable Modems
Telecom / Networking / Datacom Equipment
PGND
1
VSW
FB
ISET
COMP
VIN
AGND
BST
(Top View)
ORDERING INFORMATION
See detailed ordering and shipping information in the package
dimensions section on page 22 of this data sheet.
100
BST
VIN
VSW
PGND
NCP3126
ISET
COMP
95
3.3 V
EFFICIENCY (%)
4.5 V − 13.2 V
FB1
AGND
5V
90
85
80
75
70
65
Figure 1. Typical Application Circuit
© Semiconductor Components Industries, LLC, 2011
April, 2011 − Rev. 2
0
0.5
1
1.5
2
OUTPUT CURRENT (A)
2.5
3
Figure 2. Efficiency (VIN = 12 V) vs. Load Current
1
Publication Order Number:
NCP3126/D
NCP3126
CIRCUIT DESCRIPTION
BST
VIN
+
−
−
+
UVLO
POR
10 mA
VREF
Fault
+
OLP
Latch
−
+
0.8 V
FB
VOCTH
+
Fault
PWM
Comp
−
R
PWM
OUT
−
Q
S
COMP
VSW
Counter
DtoA
Clock
Ramp
Count
Latch
&
Logic
−
+
+
+
−
−
2V
OSC
VCC
VREG
OSC
+
0.7 V
−
Fault
AGND
Figure 3. NCP3126 Block Diagram
ISET
PGND
Table 1. PIN DESCRIPTION
Pin
Pin Name
1
PGND
Description
2
FB
Inverting input to the Operational Transconductance Amplifier (OTA). The FB pin in conjunction with the
external compensation, serves to stabilize and achieve the desired output voltage with voltage mode
control.
3
COMP
COMP pin is used to compensate the OTA which stabilizes the operation of the converter stage. Place
compensation components as close to the converter as possible.
4
AGND
The AGND pin serves as small−signal ground. All small−signal ground paths should connect to the AGND
pin at a single point, avoiding any high current ground returns.
5
BST
Supply rail for the floating top gate driver. To form a boost circuit, use an external diode to bring the desired
input voltage to this pin (cathode connected to BST pin). Connect a capacitor (CBST) between this pin and
the VSW pin. Typical values for CBST range from 1 nF to 10 nF. Ensure that CBST is placed near the IC.
6
VIN
The VIN pin powers the internal control circuitry and is monitored by an undervoltage comparator. The VIN
pin is also connected to the internal power NMOSFET switches. The VIN pin has high dI/dt edges and must
be decoupled to PGND pin close to the pin of the device.
7
ISET
Current set pin and bottom gate MOSFET driver. Place a resistor to ground to set the current limit of the
converter.
8
VSW
The VSW pin is the connection of the drain and source of the internal N−MOSFETs. The VSW pin swings
from VIN when the high side switch is on to small negative voltages when the low side switch is on with high
dV/dt transitions.
The PGND pin is the high current ground pin for the low−side MOSFET and the drivers. The pin should be
soldered to a large copper area to reduce thermal resistance.
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NCP3126
Table 2. MAXIMUM RATINGS
Rating
Symbol
Min
Max
Unit
VIN
−0.3
15
V
VBST
−0.3
15
V
Bootstrap Supply Voltage vs Ground (spikes ≤ 50 ns)
VBST spike
−5.0
35
V
Bootstrap Pin Voltage vs VSW
Main Supply Voltage Input
Bootstrap Supply Voltage vs GND
VBST−VSW
−0.3
15
V
High Side Switch Max DC Current
IVSW
0
3.0
A
VSW Pin Voltage
VSW
−0.3
30
V
VSWLIM
−2.0
35
V
Switch Pin voltage (spikes < 50 ns)
VSWtr
−5.0
40
V
FB Pin Voltage
VFB
−0.3
5.5 < VCC
V
VCOMP/DIS
−0.3
5.5 < VCC
V
VISET
−0.3
15 < VCC
V
VISET Spike
−2
15 < VCC
V
Switching Node Voltage Excursion (200 mA)
COMP/DISABLE
Low Side Driver Pin Voltage
Low Side Driver Pin Voltage (spikes v 200 ns)
Rating
Symbol
Rating
Unit
RqJA
110
183
°C/W
Thermal Resistance, Junction−to−Case
RqJC
170
°C/W
Storage Temperature Range
Tstg
−55 to 150
°C
Junction Operating Temperature
TJ
−40 to 125
°C
Lead Temperature Soldering (10 sec):
Reflow (SMD styles only) Pb−Free
RF
260 peak
°C
Thermal Resistance, Junction−to−Ambient
(Note 2)
(Note 3)
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
1. The maximum package power dissipation limit must not be exceeded.
2. The value of qJA is measured with the device mounted on 1 in2 FR−4 board with 1 oz. copper, in a still air environment with TA = 25°C. The
value in any given application depends on the user’s specific board design.
3. The value of qJA is measured with the device mounted on minimum footprint, in a still air environment with TA = 25°C. The value in any given
application depends on the user’s specific board design.
4. 60−180 seconds minimum above 237°C.
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NCP3126
Table 3. ELECTRICAL CHARACTERISTICS (−40°C < TJ < 125°C; VIN = 12 V, BST−VSW = 12 V, BST = 12 V, VSW = 24 V, for
min/max values unless otherwise noted.)
Conditions
Min
Max
Unit
Input Voltage Range
VIN − GND
4.5
13.2
V
Boost Voltage Range
VBST − GND
4.5
26.5
V
Quiescent Supply Current
VFB = 1.0 V, No Switching, VIN = 13.2 V
1.0
−
10.0
mA
Shutdown Supply Current
VFB = 1.0 V, COMP = 0 V, VIN = 13.2 V
−
4.0
−
mA
Boost Quiescent Current
VFB = 1.0 V, No Switching, VIN = 13.2 V
0.1
−
1.0
mA
VIN UVLO Threshold
VIN Rising Edge
3.8
−
4.3
V
VIN UVLO Hysteresis
−
−
430
−
mV
VFB Feedback Voltage,
Control Loop in Regulation
TJ = 0 to 25°C, 4.5 V < VCC < 13.2 V
−40°C v TJ v 125°C, 4.5 v VCC v 13.2 V
792
784
800
800
808
816
mV
Oscillator Frequency
TJ = 0 to 25°C, 4.5 V < VCC < 13.2 V
−40°C v TJ v 125°C, 4.5 v VCC v 13.2 V
300
290
350
350
400
410
kHz
0.8
1.1
1.4
V
Minimum Duty Ratio
−
5.5
−
%
Maximum Duty Ratio
70
75
80
%
3.0
−
5
ms
CO = 1 nF
55
70
−
dB
VFB < 0.8 V
VFB > 0.8 V
60
60
125
125
200
200
mA
−
0.160
1.0
mA
0.3
0.4
0.5
V
3
−
15
ms
Characteristic
Typ
SUPPLY CURRENT
UNDER VOLTAGE LOCKOUT
SWITCHING REGULATOR
Ramp−Amplitude Voltage
PWM COMPENSATION
Transconductance
Open Loop DC Gain
Output Source Current
Output Sink Current
Input Bias Current
ENABLE
Enable Threshold
SOFT−START
Delay to Soft−Start
SS Source Current
VFB < 0.8 V
−
10.5
−
mA
Switch Over Threshold
VFB = 0.8 V
−
100
−
% of Vref
Sourced from ISET pin, before SS
−
10
−
mA
OC Switch−Over Threshold
−
700
−
mV
Fixed OC Threshold
−
375
OVER−CURRENT PROTECTION
OCSET Current Source
mV
PWM OUTPUT STAGE
High−Side Switch On−Resistance
VIN = 12 V (Note 5)
VIN = 5 V (Note 5)
80
105
140
175
mW
Low−Side Switch On−Resistance
VIN = 12 V (Note 5)
VIN = 5 V (Note 5)
45
65
75
100
mW
5. Guaranteed by design.
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NCP3126
TYPICAL CHARACTERISTICS
25
5.0
23
4.0
VCC = 12 V
3.5
3.0
VCC = 5 V
2.5
21
19
17
15
13
0
20
40
60
80
9
100 120 140
VCC = 5 V
0
30
40
50
60
Figure 4. ICC vs. Temperature
Figure 5. Input Current Switching vs.
Temperature
70
808
806
Vref, REFERENCE (mV)
12
11
10
9
804
802
800
798
796
794
0
10
20
30
40
50
60
TJ, JUNCTION TEMPERATURE (°C)
792
70
0
10
20
30
40
50
60
70
TJ, JUNCTION TEMPERATURE (°C)
Figure 6. Soft−Start Sourcing Current vs.
Temperature
Figure 7. Reference Voltage (Vref) vs.
Temperature
6.0
375
5.0
365
DUTY CYCLE (%)
OLP THRESHOLD (mV)
20
TJ, JUNCTION TEMPERATURE (°C)
13
355
345
335
325
10
TJ, JUNCTION TEMPERATURE (°C)
14
8
VCC = 12 V
11
2.0
−60 −40 −20
SOFT−START SOURCING CURRENT (mA)
INPUT CURRENT (mA)
INPUT CURRENT (mA)
4.5
VCC = 12 V
4.0
VCC = 5 V
3.0
2.0
1.0
0
10
20
30
40
50
60
0
70
0
10
20
30
40
50
60
TJ, JUNCTION TEMPERATURE (°C)
TJ, JUNCTION TEMPERATURE (°C)
Figure 8. OLP Threshold vs. Temperature
Figure 9. Minimum Active Duty Cycle vs.
Temperature
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70
NCP3126
TYPICAL CHARACTERISTICS
76
DUTY CYCLE (%)
75.5
VCC = 12 V
75
VCC = 5 V
74.5
74
73.5
73
0
10
20
30
40
50
60
70
TJ, JUNCTION TEMPERATURE (°C)
Figure 10. Duty Cycle Maximum vs. Temperature
100
95
3.3 V
95
85
80
75
70
65
1.5 V
60
1.2 V
1.1 V
1.8 V
0.5
1
1.5
2
LOAD CURRENT (A)
85
80
75
70
50
0
3
1.1 V
0.8 V
1.0 V
0.5
1
1.5
2
LOAD CURRENT (A)
2.5
3
Figure 12. Efficiency (VIN = 5 V) vs. Load
Current
3.5
1.2 V
1.2 V & 1.8 V
3
2.5
2.5 V
OUTPUT CURRENT (A)
3
OUTPUT CURRENT (A)
1.2 V
65
Figure 11. Efficiency (VIN = 12 V) vs. Load
Current
1.8 V
2
1.5
1
2.5
5.0 V
2
3.3 V
1.5
1
0.5
0.5
0
1.5 V
55
2.5
3.5
1.8 V
60
0.8 V
1.0 V
55
50
0
2.5 V Output
90
EFFICIENCY (%)
EFFICIENCY (%)
90
100
5.0 V Output
2.5 V
25
35
45
55
65
75
0
25
85
35
45
55
65
75
TA, AMBIENT TEMPERATURE
TA, AMBIENT TEMPERATURE
Figure 13. Derating Curve 5 V/6 V Input
Figure 14. Derating Curve 12 V Input
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NCP3126
General
COMP
The NCP3126 is a PWM synchronous buck regulator
intended to supply up to a 3 A load for DC−DC conversion
from 5 V and 12 V buses. The NCP3126 is a regulator that
has integrated high−side and low−side NMOSFETs
switches. The output voltage of the converter can be
precisely regulated down to 800 mV $1.0% when the VFB
pin is tied to VOUT. The switching frequency is internally set
to 350 kHz. A high gain operational transconductance
amplifier (OTA) is used for voltage mode control of the
power stage.
Disable
2N7002E
Gate Signal
Enable
COMP
Duty Ratio and Maximum Pulse Width Limits
Disable
In steady state DC operation, the duty ratio will stabilize
at an operating point defined by the ratio of the input to the
output voltage. The device can achieve a 75% duty ratio. The
NCP3126 has a preset off−time of approximately 150 ns,
which ensures that the bootstrap supply is charged every
switching cycle. The preset off time does not interfere with
the conversion of 12 V to 0.8 V.
Base Signal
MMBT3904
Enable
Figure 15. Recommended Disable Circuits
Input Voltage Range (VIN and BST)
The input voltage range for both VIN and BST is 4.5 V to
13.2 V with respect to GND and VSW. Although BST is
rated at 13.2 V with respect to VSW, it can also tolerate
26.5 V with respect to GND.
Power Sequencing
Power sequencing can be achieved with NCP3126 using
two general purpose bipolar junction transistors or
MOSFETs. An example of the power sequencing circuit
using the external components is shown in Figure 16.
External Enable/Disable
Once the input voltage has exceeded the boost and UVLO
threshold at 3 V and VIN threshold at 4 V, the COMP pin
starts to rise. The VSW node is tri−stated until the COMP
voltage exceeds 0.9 V. Once the 0.9 V threshold is exceeded,
the part starts to switch and the part is considered enabled.
When the COMP pin voltage is pulled below the 400 mV
threshold, it disables the PWM logic, the top MOSFET is
driven off, and the bottom MOSFET is driven on. In the
disabled mode, the OTA output source current is reduced to
10 mA.
When disabling the NCP3126 using the COMP / Disable
pin, an open collector or open drain drive should be used as
shown in Figure 15:
VSW
1.0V
VIN
VSW
NCP3126
FB1
NCP3126
FB1
COMP
COMP
3.3 V
Figure 16. Power Sequencing
Input Voltage Shutdown Behavior
Input voltage shutdown occurs when the IC stops
switching because the input supply reaches UVLO
threshold. Undervoltage Lockout (UVLO) is provided to
ensure that unexpected behavior does not occur when VCC
is too low to support the internal rails and power the
converter. For the NCP3126, the UVLO is set to permit
operation when converting from an input voltage of 5 V. If
the UVLO is tripped, switching stops, the internal SS is
discharged, and all MOSFET gates are driven low. The VSW
node enters a high impedance state and the output capacitors
discharge through the load with no ringing on the output
voltage.
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NCP3126
Overcurrent Threshold Setting
NCP3126 overcurrent threshold can be set from 50 mV to
550 mV, by adding a resistor (RSET) between ISET and
GND. During a short period of time following VIN rising
over UVLO threshold, an internal 10 mA current (IOCSET) is
sourced from the ISET pin, creating a voltage drop across
RSET. The voltage drop is compared against a stepped
internal voltage ramp. Once the internal stepped voltage
reaches the RSET voltage, the value is stored internally until
power is cycled. The overall time length for the OC setting
procedure is approximately 9 ms. Connecting an RSET
resistor between ISET and GND, the programmed threshold
will be:
0.9 V
COMP
BG
TG
Figure 17. Enable/Disable Driver State Diagram
I OCth +
I OCSET * R SET
External Soft−Start
The NCP3126 features an external soft−start function,
which reduces inrush current and overshoot of the output
voltage. Soft−start is achieved by using the internal current
source of 10.5 mA (typ), which charges the external
integrator capacitor of the OTA. Figure 18 is a typical
soft−start sequence. The sequence begins once VIN and
VBST surpass their UVLO thresholds and OCP
programming is complete. The current sourced out of the
COMP pin continually increases the voltage until regulation
is reached. Once the voltage reaches 400 mV logic is
enabled. When the voltage exceeds 900 mV, switching
begins. Current is sourced out of the COMP pin, placing the
regulator into open loop operation until 800 mV is sensed at
the FB pin. Once 800 mV is sensed at the FB pin, open loop
operation ends and closed loop operation begins. In closed
loop operation, the OTA is capable of sourcing and sinking
120 mA.
4.2 V
R DS(on)
³ 3.2 A +
10 mA * 24 kW
75 mW
(eq. 1)
IOCSET = Sourced current
IOCth = Current trip threshold
RDS(on) = On resistance of the low side MOSFET
RSET = Current set resistor
The RSET values range from 5 kW to 55 kW. If RSET is not
connected, the device switches the OCP threshold to a fixed
375 mV value, an internal safety clamp on ISET is triggered
as soon as ISET voltage reaches 700 mV, enabling the
375 mV fixed threshold and ending the OCP setting period.
The current trip threshold tolerance is $25 mV. The
accuracy is best at the highest set point (550 mV). The
accuracy will decrease as the set point decreases. MOSFET
tolerances with temperature and input voltage will vary the
over current set threshold operating point. A graph of the
typical current limit set thresholds at 4.5 V and 12 V is
shown in Figure 19.
7
3.85 V
6.5
6
OUTPUT CURRENT (A)
VCC
5.5
0.9 V
COMP
5
4.5
VFB
12 V
4
3.5
3
BG
TG
BG Comparator
DAC Voltage
500 mV
2
1.5
50 mV
BG Comparator Output
1
Vout
UVLO
5.0 V
2.5
POR Current
Soft−Start Normal Operation
Delay Trip Set COMP
Delay
5
UVLO
10
15
20
25 30 35
RSET (kW)
40
45
50
Figure 19. RSET Value for Output Current
Figure 18. Soft−Start Sequence
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NCP3126
Current Limit Protection
In case of an overload, the low−side (LS) FET will
conduct large currents. The regulator will latch off,
protecting the load and MOSFETs from excessive heat and
damage. Low−side RDS(on) sense is implemented at the end
of each LS−FET turn−on duration to sense the current.
While the low side MOSFET is on, the VSW voltage is
compared to the user set internally generated OCP trip
voltage. If the VSW voltage is lower than OCP trip voltage,
an overcurrent condition occurs and a counter counts
consecutive current trips. If the counter reaches 7, the PWM
logic and both HS−FET and LS−FET are turned off. The
regulator has to go through a Power On Reset (POR) cycle
to reset the OCP fault as shown in Figure 20.
Current
−
BG
Flow
+
VOCTH
BG
Drive
Low Side
MOSFET
Current
PHASE
0V
VOCTH
Figure 20. Current Limit Trip
APPLICATION SECTION
Design Procedure
VOUT, VIN, the Low Side Switch Voltage Drop VLSD, and
the High Side Switch Voltage Drop VHSD.
When starting the design of a buck regulator, it is
important to collect as much information as possible about
the behavior of the input and output before starting the
design.
ON Semiconductor has a Microsoft Excel® based design
tool available online under the design tools section of the
NCP3126 product page. The tool allows you to capture your
design point and optimize the performance of your regulator
based on your design criteria.
F SW + 1
T
(VIN)
Output voltage
(VOUT)
Input ripple voltage
(VINRIPPLE)
300 mV
Output ripple voltage
(VOUTRIPPLE)
60 mV
Output current rating
(IOUT)
3A
Operating frequency
(FSW)
350 kHz
T ON
T
and (1 * D) + OFF
T
T
D+
V OUT ) V LSD
V
3.3 V
[ D + OUT ³ 27.5% +
V IN * V HSD ) V LSD
V IN
12 V
D
FSW
T
TOFF
TON
VHSD
VIN
VLSD
VOUT
Example Value
Input voltage
D+
(eq. 3)
(eq. 4)
Table 4. DESIGN PARAMETERS
Design Parameter
(eq. 2)
10.8 V to 13.2 V
3.3 V
= Duty cycle
= Switching frequency
= Switching period
= High side switch off time
= High side switch on time
= High side switch voltage drop
= Input voltage
= Low side switch voltage drop
= Output voltage
Inductor Selection
When selecting an inductor, the designer can employ a
rule of thumb for the design where the percentage of ripple
current in the inductor should be between 10% and 40%.
When using ceramic output capacitors, the ripple current can
be greater because the ESR of the output capacitor is smaller,
thus a user might select a higher ripple current. However,
The buck converter produces input voltage VIN pulses that
are LC filtered to produce a lower DC output voltage VOUT.
The output voltage can be changed by modifying the on time
relative to the switching period T or switching frequency.
The ratio of high side switch on time to the switching period
is called duty ratio D. Duty ratio can also be calculated using
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NCP3126
when using electrolytic capacitors, a lower ripple current
will result in lower output ripple due to the higher ESR of
electrolytic capacitors. The ratio of ripple current to
maximum output current is given in Equation 5.
0.84 A
ra + DI ³ 28% +
Iout
3A
IOUT
= Output current
IPK
= Inductor peak current
ra
= Ripple current ratio
A standard inductor should be found so the inductor will
be rounded to 6.8 mH. The inductor should also support an
RMS current of 3.01 A and a peak current of 3.42 A.
The final selection of an output inductor has both
mechanical and electrical considerations. From a
mechanical perspective, smaller inductor values generally
correspond to smaller physical size. Since the inductor is
often one of the largest components in the regulation system,
a minimum inductor value is particularly important in space
constrained applications. From an electrical perspective, the
maximum current slew rate through the output inductor for
a buck regulator is given by Equation 9.
(eq. 5)
DI
= Ripple current
IOUT
= Output current
ra
= Ripple current ratio
Using the ripple current rule of thumb, the user can
establish acceptable values of inductance for a design using
Equation 6.
L OUT +
V OUT
@ (1 * D) ³
I OUT @ ra @ F SW
(eq. 6)
3.3 V
6.73 mH +
@ (1 * 27.5%)
3 A @ 28% @ 350 kHz
D
FSW
IOUT
LOUT
ra
SlewRate LOUT +
+
= Duty ratio
= Switching frequency
= Output current
= Output inductance
= Ripple current ratio
18
VIN = 12 V
INDUCTANCE (mH)
14
12
10
8
Selected
6
VIN = 8 V
VIN = 4.2 V
4
2
10
13
16 19 22 25 28 31 34
CURRENT RIPPLE RATIO (%)
37
Ipp +
40
2
28%
3.01 A + 3 A * Ǹ1 )
12
IOUT
IRMS
ra
2
(eq. 7)
= Output current
= Inductor RMS current
= Ripple current ratio
ǒ
I PK + I OUT @ 1 )
Ǔ
ǒ
(eq. 9)
6.8 mH
V OUT (1 * D)
³
L OUT @ F SW
3.3 V
(1 * 27.5%)
6.8 mH @ 350 kHz
(eq. 10)
D
= Duty ratio
FSW
= Switching frequency
Ipp
= Peak−to−peak current of the inductor
LOUT
= Output inductance
VOUT
= Output voltage
From Equation 10 it is clear that the ripple current increases
as LOUT decreases, emphasizing the trade−off between
dynamic response and ripple current.
The power dissipation of an inductor falls into two
categories: copper and core losses. The copper losses can be
further categorized into DC losses and AC losses. A good
first order approximation of the inductor losses can be made
using the DC resistance as shown below:
When selecting an inductor, the designer must not exceed
the current rating of the part. To keep within the bounds of
the part’s maximum rating, a calculation of the RMS and
peak inductor current is required.
Ǹ1 ) ra12 ³
A
ms
12 V * 3.3 V
0.84 A +
Figure 21. Inductance vs. Current Ripple Ratio
I RMS + I OUT @
L OUT
³ 1.41
LOUT
= Output inductance
VIN
= Input voltage
VOUT
= Maximum output voltage
Equation 9 implies that larger inductor values limit the
regulator’s ability to slew current through the output
inductor in response to output load transients. Consequently,
output capacitors must supply the load current until the
inductor current reaches the output load current level.
Reduced inductance to increase slew rates results in larger
values of output capacitance to maintain tight output voltage
regulation. In contrast, smaller values of inductance increase
the regulator’s maximum achievable slew rate and decrease
the necessary capacitance, at the expense of higher ripple
current. The peak−to−peak ripple current for NCP3126 is
given by the following equation:
20
16
V IN * V OUT
Ǔ
ra
28%
³ 3.42 A + 3 A @ 1 )
2
2
(eq. 8)
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10
NCP3126
LP _DC + I RMS 2 @ DCR ³
The ESL of capacitors depends on the technology chosen,
but tends to range from 1 nH to 20 nH, where ceramic
capacitors have the lowest inductance and electrolytic
capacitors have the highest. The calculated contributing
voltage ripple from ESL is shown for the switch on and
switch off below:
(eq. 11)
173 mW + 3.01 A 2 @ 19.1 mW
IRMS
= Inductor RMS current
DCR
= Inductor DC resistance
LPCU_DC
= Inductor DC power dissipation
The core losses and AC copper losses will depend on the
geometry of the selected core, core material, and wire used.
Most vendors will provide the appropriate information to
make accurate calculations of the power dissipation at which
point the total inductor losses can be captured by the
equation below:
LP tot + LP CU_DC ) LP CU_AC ) LP Core ³
185 mW + 173 mW ) 0 mW ) 12 mW
LPCU_DC
LPCU_AC
LPCore
V ESLON +
V ESLOFF +
The important factors to consider when selecting an
output capacitor are DC voltage rating, ripple current rating,
output ripple voltage requirements, and transient response
requirements.
The output capacitor must be rated to handle the ripple
current at full load with proper derating. The RMS ratings
given in datasheets are generally for lower switching
frequency than used in switch mode power supplies, but a
multiplier is usually given for higher frequency operation.
The RMS current for the output capacitor can be calculated
below:
ǒ
CoESR
COUT
FSW
IOUT
ra
(eq. 16)
Co ESR ³ 100 mV + 2.0
50 mW
(eq. 17)
= Output capacitor Equivalent Series
Resistance
ITRAN
= Output transient current
DVOUT_ESR = Voltage deviation of VOUT due to the
effects of ESR
A minimum capacitor value is required to sustain the
current during the load transient without discharging it. The
voltage drop due to output capacitor discharge is given by
the following equation:
DV OUT*DIS +
Ǔ
1
³
8 * F SW * C OUT
ǒI TRANǓ
2
4.02 mV +
(eq. 14)
8 * 350 kHz * 470 mF
27.5%
CoESR
CoRMS
= Output capacitor RMS current
IOUT
= Output current
ra
= Ripple current ratio
The maximum allowable output voltage ripple is a
combination of the ripple current selected, the output
capacitance selected, the Equivalent Series Inductance
(ESL), and Equivalent Series Resitance (ESR).
The main component of the ripple voltage is usually due
to the ESR of the output capacitor and the capacitance
selected, which can be calculated as shown in Equation 14:
42.64 mV + 3 * 28% * 50 mW )
(eq. 15)
ESL * Ipp * F SW
DV OUT*ESR + I TRAN
(eq. 13)
1
10 nH * 0.84 A * 350 kHz
D
= Duty ratio
ESL
= Capacitor inductance
FSW
= Switching frequency
Ipp
= Peak−to−peak current
The output capacitor is a basic component for the fast
response of the power supply. For the first few microseconds
of a load transient, the output capacitor supplies current to
the load. Once the regulator recognizes a load transient, it
adjusts the duty ratio, but the current slope is limited by the
inductor value.
During a load step transient, the output voltage initially
drops due to the current variation inside the capacitor and the
ESR (neglecting the effect of the ESL).
Output Capacitor Selection
ǒ
³
³
(1 * D )
10 nH * 0.84 A * 350 kHz
5.79 mV +
ǒ1 * 27.5% Ǔ
= Inductor DC power dissipation
= Inductor AC power dissipation
= Inductor core power dissipation
V ESR_C + I OUT * ra * Co ESR )
D
15.27 mV +
(eq. 12)
ra
28%
Co RMS + I OUT @
³ 0.243 A + 3 A
Ǹ12
Ǹ12
ESL * Ipp * F SW
Ǔ
COUT
DMAX
ITRAN
LOUT
VIN
VOUT
DVOUT_DIS
= Output capacitor ESR
= Output capacitance
= Switching frequency
= Output current
= Ripple current ratio
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11
=
=
=
=
=
=
=
2
D MAX C OUT
L OUT
ǒVIN * V OUTǓ
ǒ2 A Ǔ
2
75%
2
470 mF
6.8 mH
³
(eq. 18)
ǒ12 V * 3.3 VǓ
Output capacitance
Maximum duty ratio
Output transient current
Output inductor value
Input voltage
Output voltage
Voltage deviation of VOUT due to the
effects of capacitor discharge
NCP3126
Starting with the high−side MOSFET, the power
dissipation can be approximated from:
In a typical converter design, the ESR of the output
capacitor bank dominates the transient response. Please note
that DVOUT−DIS and DVOUT−ESR are out of phase with each
other, and the larger of these two voltages will determine the
maximum deviation of the output voltage (neglecting the
effect of the ESL).
P D_HS + P COND ) P SW_TOT
PCOND
= Conduction power losses
PSW_TOT
= Total switching losses
PD_HS
= Power losses in the high side MOSFET
The first term in Equation 21 is the conduction loss of the
high−side MOSFET while it is on.
Input Capacitor Selection
The input capacitor has to sustain the ripple current
produced during the on time of the upper MOSFET, so it
must have a low ESR to minimize the losses. The RMS value
of the input ripple current is:
Iin RMS + I OUT ǸD
(1 * D) ³
1.22 A + 3 A * Ǹ27.58% * (1 * 27.58%)
2
P COND + ǒI RMS_HSǓ @ R DS(on)_HS
I RMS_HS + I OUT @
Ǹ
ǒ
D@ 1)
Ǔ
ra 2
12
(eq. 23)
IRMS_HS
= High side MOSFET RMS current
IOUT
= Output current
D
= Duty ratio
ra
= Ripple current ratio
The second term from Equation 21 is the total switching
loss and can be approximated from the following equations.
2
14.8 mW + 10 mW * ǒ1.22 AǓ
(eq. 22)
IRMS_HS
= RMS current in the high−side MOSFET
RDS(on)_HS = On resistance of the high−side MOSFET
Pcond
= Conduction power losses
Using the ra term from Equation 5, IRMS becomes:
(eq. 19)
D
= Duty ratio
IINRMS
= Input capacitance RMS current
IOUT
= Load current
The equation reaches its maximum value with D = 0.5.
Loss in the input capacitors can be calculated with the
following equation:
P CIN + CIN ESR * ǒIiN RMSǓ ³
(eq. 21)
(eq. 20)
2
CINESR
= Input capacitance Equivalent Series
Resistance
IINRMS
= Input capacitance RMS current
PCIN
= Power loss in the input capacitor
Due to large di/dt through the input capacitors, electrolytic
or ceramics should be used. If a tantalum must be used, it
must be surge protected, otherwise, capacitor failure could
occur.
P SW_TOT + P SW ) P DS ) P RR
(eq. 24)
PDS
= High side MOSFET drain source losses
PRR
= High side MOSFET reverse recovery losses
PSW
= High side MOSFET switching losses
PSW_TOT
= High side MOSFET total switching losses
The first term for total switching losses from Equation 24
are the losses associated with turning the high−side
MOSFET on and off and the corresponding overlap in drain
voltage and current.
Power MOSFET Dissipation
MOSFET power dissipation, package size, and the
thermal environment drive power supply design. Once the
dissipation is known, the thermal impedance can be
calculated to prevent the specified maximum junction
temperatures from being exceeded at the highest ambient
temperature.
Power dissipation has two primary contributors:
conduction losses and switching losses. The high−side
MOSFET will display both switching and conduction
losses. The switching losses of the low side MOSFET will
not be calculated as it switches into nearly zero voltage and
the losses are insignificant. However, the body diode in the
low−side MOSFET will suffer diode losses during the
non−overlap time of the gate drivers.
P SW + P TON ) P TOFF
(eq. 25)
+ 1 @ ǒI OUT @ V IN @ F SWǓ @ ǒt RISE ) t FALLǓ
2
FSW
IOUT
tFALL
tRISE
VIN
PSW
PTON
PTOFF
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12
= Switching frequency
= Load current
= MOSFET fall time
= MOSFET rise time
= Input voltage
= High side MOSFET switching losses
= Turn on power losses
= Turn off power losses
NCP3126
When calculating the rise time and fall time of the high side
MOSFET it is important to know the charge characteristic
shown in Figure 22.
FSW
= Switching frequency
PRR
= High side MOSFET reverse recovery losses
QRR
= Reverse recovery charge
VIN
= Input voltage
The low−side MOSFET turns on into small negative
voltages so switching losses are negligible. The low−side
MOSFET’s power dissipation only consists of conduction
loss due to RDS(on) and body diode loss during the
non−overlap periods.
Vth
P D_LS + P COND ) P BODY
(eq. 30)
PBODY
= Low side MOSFET body diode losses
PCOND
= Low side MOSFET conduction losses
PD_LS
= Low side MOSFET losses
Conduction loss in the low−side MOSFET is described as
follows:
2
P COND + ǒI RMS_LSǓ @ R DS(on)_LS
Figure 22. MOSFET Switching Characteristics
t RISE +
Q GD
I G1
IG1
Q GD
ǒV BST * V THǓńǒR HSPU ) R GǓ
(eq. 26)
Q GD
I G2
+
Q GD
ǒVBST * V THǓńǒR HSPD ) R GǓ
ǒ ǒ ǓǓ
2
(1 * D) @ 1 ) ra
12
(eq. 32)
P BODY + V FD @ I OUT @ F SW @ ǒNOL LH ) NOL HLǓ (eq. 33)
FSW
IOUT
NOLHL
(eq. 27)
NOLLH
PBODY
VFD
= Switching frequency
= Load current
= Dead time between the high−side
MOSFET turning off and the low−side
MOSFET turning on, typically 50 ns
= Dead time between the low−side
MOSFET turning off and the high−side
MOSFET turning on, typically 50 ns
= Low−side MOSFET body diode losses
= Body diode forward voltage drop
Control Dissipation
The control portion of the IC power dissipation is
determined by the formula below:
(eq. 28)
P C + I CC
COSS
= MOSFET output capacitance at 0V
FSW
= Switching frequency
PDS
= MOSFET drain to source charge losses
VIN
= Input voltage
Finally, the loss due to the reverse recovery time of the
body diode in the low−side MOSFET is shown as follows:
P RR + Q RR @ V IN @ F SW
Ǹ
D
= Duty ratio
IOUT
= Load current
IRMS_LS
= RMS current in the low side
ra
= Ripple current ratio
The body diode losses can be approximated as:
IG2
= Output current from the low−side gate drive
QGD
= MOSFET gate to drain gate charge
RG
= MOSFET gate resistance
RHSPD
= Drive pull down resistance
tFALL
= MOSFET fall time
VBST
= Boost voltage
VTH
= MOSFET gate threshold voltage
Next, the MOSFET output capacitance losses are caused
by both the high−side and low−side MOSFETs, but are
dissipated only in the high−side MOSFET.
2
P DS + 1 @ C OSS @ V IN @ F SW
2
= RMS current in the low side
= Low−side MOSFET on resistance
= High side MOSFET conduction losses
I RMS_LS + I OUT @
= Output current from the high−side gate
drive
= MOSFET gate to drain gate charge
= Drive pull up resistance
= MOSFET gate resistance
= MOSFET rise time
= Boost voltage
= MOSFET gate threshold voltage
QGD
RHSPU
RG
tRISE
VBST
VTH
t FALL +
+
IRMS_LS
RDS(on)_LS
PCOND
(eq. 31)
V IN
(eq. 34)
ICC
= Control circuitry current draw
PC
= Control power dissipation
VIN
= Input voltage
Once the IC power dissipations are determined, the
designer can calculate the required thermal impedance to
maintain a specified junction temperature at the worst case
(eq. 29)
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13
NCP3126
ambient temperature. The formula for calculating the
junction temperature with the package in free air is:
T J + T A ) P D @ R qJA
FSW
= Switching frequency
FESR
= Output capacitor ESR zero frequency
If the criteria is not met, the compensation network may
not provide stability and the output power stage must be
modified.
Figure 23 shows a pseudo Type III transconductance error
amplifier.
(eq. 35)
PD
RqJA
= Power dissipation of the IC
= Thermal resistance junction to ambient of
the regulator package
TA
= Ambient temperature
TJ
= Junction temperature
As with any power design, proper laboratory testing
should be performed to ensure the design will dissipate the
required power under worst case operating conditions.
Variables considered during testing should include
maximum ambient temperature, minimum airflow,
maximum input voltage, maximum loading, and component
variations (i.e., worst case MOSFET RDS(on)).
ZIN
CF
IEA
CC
To create a stable power supply, the compensation
network around the transconductance amplifier must be
used in conjunction with the PWM generator and the power
stage. Since the power stage design criteria is set by the
application, the compensation network must correct the over
all system response to ensure stability. The output inductor
and capacitor of the power stage form a double pole at the
frequency as shown in Equation 36:
F LC +
2p ǸL OUT
2.85 kHz +
C OUT
Gm
CP
R2
2p
Ǹ6.8 mH
RC
VREF
Figure 23. Pseudo Type III Transconductance Error
Amplifier
The compensation network consists of the internal OTA
and the impedance networks ZIN (R1, R2, RF, and CF) and
external ZFB (RC, CC, and CP). The compensation network
has to provide a closed loop transfer function with the
highest 0 dB crossing frequency to have fast response and
the highest gain in DC conditions to minimize the load
regulation issues. A stable control loop has a gain crossing
with −20 dB/decade slope and a phase margin greater than
45°. Include worst−case component variations when
determining phase margin. To start the design, a resistor
value should be chosen for R2 from which all other
components can be chosen. A good starting value is 10 kW.
The NCP3126 allows the output of the DC−DC regulator
to be adjusted down to 0.8 V via an external resistor divider
network. The regulator will maintain 0.8 V at the feedback
pin. Thus, if a resistor divider circuit was placed across the
feedback pin to VOUT, the regulator will regulate the output
voltage proportional to the resistor divider network in order
to maintain 0.8 V at the FB pin.
³
(eq. 36)
1
470 mF
COUT
FLC
= Output capacitor
= Double pole inductor and capacitor
frequency
LOUT
= Output inductor value
The ESR of the output capacitor creates a “zero” at the
frequency as shown in Equation 37:
F ESR +
2.773 kHz +
2p
1
CO ESR
C OUT
³
1
2p
0.050 mW
(eq. 37)
470 mF
COESR
= Output capacitor ESR
COUT
= Output capacitor
FLC
= Output capacitor ESR frequency
The two equations above define the bode plot that the
power stage has created or open loop response of the system.
The next step is to close the loop by considering the feedback
values. The closed loop crossover frequency should be
greater than the FLC and less than 1/5 of the switching
frequency, which would place the maximum crossover
frequency at 70 kHz. Further, the calculated FESR frequency
should meet the following:
F ESR t
F SW
5
RF
ZFB
Compensation Network
1
R1
VOUT
R1
FB
R2
Figure 24. Feedback Resistor Divider
The relationship between the resistor divider network
above and the output voltage is shown in Equation 39:
(eq. 38)
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NCP3126
R2 + R1 @
ǒ
V REF
Ǔ
The compensation components for the Pseudo Type III
Transconductance Error Amplifier can be calculated using
the method described below. The method serves to provide
a good starting place for compensation of a power supply.
The values can be adjusted in real time using the
compensation tool comp calc, available for download at
ON Semiconductor’s website.
The value of the feed through resistor should always be at
least 2X the value of R2 to minimize error from feed through
noise. Using the 2X assumption, RF will be set to 20 kW and
the feed through capacitor can be calculated as shown
below:
(eq. 39)
V OUT * V REF
R1
= Top resistor divider
R2
= Bottom resistor divider
VOUT
= Output voltage
VREF
= Regulator reference voltage
The most frequently used output voltages and their
associated standard R1 and R2 values are listed in Table 5.
Table 5. OUTPUT VOLTAGE SETTINGS
VO (V)
R1 (kW)
R2 (kW)
0.8
1.0
Open
1.0
2.55
10
1.1
3.83
10.2
1.2
4.99
10
1.5
10
11.5
1.8
12.7
10.2
2.5
21.5
10
3.3
31.6
10
5.0
52.3
10
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NCP3126
CF +
ǒR 1 ) R 2Ǔ
ǒR 1 ) R F ) R 2
2p
239 pF +
RF ) R2
R 1Ǔ
f cross
(eq. 40)
ǒ31.6 kW ) 10 kWǓ
ǒ31.6 kW
2p
CF
fcross
R1
R2
RF
20 kW ) 10 kW
20 kW ) 10 kW
31.6 kWǓ
30 kHz
= Feed through capacitor
= Crossover frequency
= Top resistor divider
= Bottom resistor divider
= Feed through resistor
The cross over of the overall feedback occurs at FPO:
F PO +
ǒR1 ) RFǓ
2
ǒCFǓ ƪǒR1 ) RFǓ
ǒ2pǓ 2
12.69 kHz +
CF
FLC
FPO
R1
R2
RF
VIN
Vramp
R2 ) R1
V ramp
R Fƫ
ǒRF ) R1Ǔ
FLC
ǒ2pǓ
2
2
ǒ239 pFǓ ƪǒ31.6 kW ) 20 kWǓ
10 kW ) 31.6 kW
The cross over combined compensation network can be
used to calculate the transconductance output compensation
network as follows:
76 nF +
CC
FPO
gm
R1
R2
³
(eq. 41)
ǒ31.6 kW ) 20 kWǓ
= Feed through capacitor
= Frequency of the output inductor and capacitor
= Pole frequency
= Top of resistor divider
= Bottom of resistor divider
= Feed through resistor
= Input voltage
= Peak−to−peak voltage of the ramp
CC +
V IN
R2
1
F PO
R2
R1
gm ³
(eq. 42)
1
10 kW
12.69 kHz
10 kW ) 31.6 kW
4 ms
= Compensation capacitor
= Pole frequency
= Transconductance of amplifier
= Top of resistor divider
= Bottom of resistor divider
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1.1 V
20 kWƫ
ǒ20 kW ) 31.6 kWǓ
2.82 kHz
12 V
NCP3126
RC +
2
F LC
CC
1
Ǹ
ǒ 2 ń2 ) fcross
CO ESR
C OUTǓ
(eq. 43)
1.65 kW +
CC
COESR
COUT
fcross
FLC
RC
1
2
2.82 kHz
0.05 mW
470 mFǓ
= Compensation capacitance
= Output capacitor ESR
= Output capacitance
= Crossover frequency
= Output inductor and capacitor frequency
= Compensation resistor
C P + C OUT
CO ESR
RC
2.27 nF + 470 mF
COESR
COUT
CP
RC
ǒǸ2 ń2 ) 30 kHz
76 nF
2
p
³
(eq. 44)
0.05 mW
2.05 kW
2*p
= Output capacitor ESR
= Output capacitor
= Compensation pole capacitor
= Compensation resistor
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NCP3126
Assuming an output capacitance of 470 mF in parallel with
22 mF with a crossover frequency of 35 kHz, the
compensation values for common output voltages can be
calculated as shown in Table 6:
CP
CC
D
ISS
tSS
Vramp
Table 6. COMPENSATION VALUES
Vin
(V)
Vout
(V)
Lout
(mF)
Cf
(nF)
Cc
(nF)
Rc
(kW)
Cp
(nF)
12
0.8
3.3
NI
180
0.357
2.7
12
1.0
3.3
0.180
120
0.442
2.7
12
1.1
3.3
0.180
120
0.475
2.2
12
1.2
4.7
0.180
120
0.787
2.2
12
1.5
4.7
0.180
120
0.909
1.8
12
1.8
6.8
0.180
100
1.5
1.2
12
2.5
6.8
0.220
100
1.87
1
12
3.3
8.2
0.220
100
2.05
1
12
5.0
10
0.220
100
2.5
1.2
5
0.8
3.3
NI
100
0.887
1.8
5
1.0
3.3
0.180
82
1.1
1.5
5
1.1
3.3
0.180
82
1.65
0.82
5
1.2
4.7
0.180
82
1.82
0.82
5
1.5
4.7
0.180
82
2.21
0.82
5
1.8
4.7
0.180
82
2.61
0.82
5
2.5
4.7
0.180
82
3.4
0.82
V
900 mV
Vcomp
Vout
Figure 25. Soft−Start Ramp
The delay from the charging of the compensation network
to the bottom of the ramp is considered tssdelay. The total
delay time is the addition of the current set delay and tssdelay,
which in this case is 9 ms and 7.45 ms respectively, for a
total of 16.45 ms.
Calculating Input Inrush Current
The input inrush current has two distinct stages: input
charging and output charging. The input charging of a buck
stage is usually not controlled, and is limited only by the
input RC network, and the output impedance of the upstream
power stage. If the upstream power stage is a perfect voltage
source, then the input charge inrush current can be depicted
as shown in Figure 26 and calculated as:
Calculating Soft−Start Time
To calculate the soft−start delay and soft−start time, the
following equations can be used.
t SSdelay +
ǒC P ) C CǓ
7.45 ms +
0.9 V
I SS
³
(eq. 45)
ǒ2.83 nF ) 80 nFǓ
IPK
0.9 V
10 mA
CP
= Compensation pole capacitor
CC
= Compensation capacitor
ISS
= Soft−start current
The time the output voltage takes to increase from 0 V to
a regulated output voltage is tss as shown in Equation 46:
t SS +
ǒC P ) C CǓ
2.51 ms +
= Compensation pole capacitor
= Compensation capacitor
= Duty ratio
= Soft−start current
= Soft−start interval
= Peak−to−peak voltage of the ramp
D
Figure 26. Input Charge Inrush Current
I ICinrush_PK1 +
V ramp
I SS
(eq. 46)
ǒ2.83 nF ) 80 nFǓ
27.5%
120 A +
1.1 V
10 mA
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12
0.1
V IN
CIN ESR
(eq. 47)
NCP3126
I ICinrush_RMS1 +
V IN
CIN ESR
0.316
380 mA +
12 V
0.1 W
0.316
ȡ
ȧ
ȧ1 *
Ȣ eƪ
5
t
NCP3126
OR
ȣ
ȧ
ƫȤ
(eq. 48)
Figure 27. Load Connected to the Output Stage
16.45 ms
0.1W 330 mF
0.1 W
If the load is resistive in nature, the output current will
increase with soft−start linearly which can be quantified in
Equation 50.
330 mF
16.45 ms
I CLR_RMS +
CIN
= Output capacitor
CINESR
= Output capacitor ESR
tDELAY_TOTAL = Total delay interval
= Input voltage
VIN
Once the tDELAY_TOTAL has expired, the buck converter
starts to switch and a second inrush current can be
calculated:
I OCinrush_RMS +
ǒC OUT ) C LOADǓ
t SS
Load
C IN
t DELAY_TOTAL
1
Inrush Current
DELAY_TOTAL
CINESR CIN
CIN ESR
ȡ
ȧ1 *
Ȣ eƪ
5
1
ȣ
ȧ
ƫȧȤ
V OUT D
) I CL
Ǹ3
1
Ǹ3
V OUT
R OUT
I CR_PK +
V OUT
R OUT
(eq. 50)
191 mA +
ROUT
VOUT
ICLR_RMS
ICR_PK
D
(eq. 49)
COUT
= Total converter output capacitance
CLOAD
= Total load capacitance
D
= Duty ratio of the load
= Applied load at the output
ICL
IOCinrush_RMS = RMS inrush current during start−up
tSS
= Soft−start interval
VOUT
= Output voltage
From the above equation, it is clear that the inrush current
is dependant on the type of load that is connected to the
output. Two types of load are considered in Figure 27: a
resistive load and a stepped current load.
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1
Ǹ3
3.3 V
10 W
330 mA +
= Output resistance
= Output voltage
= RMS resistor current
= Peak resistor current
3.3 V
10 W
NCP3126
Layout Considerations
3.3 V
As in any high frequency switching regulator, layout is
very important. Switching current from one power device to
another can generate voltage transients across the
impedances of the interconnecting bond wires and circuit
traces. The interconnecting impedances should be
minimized by using wide short printed circuit traces. The
critical components should be located as close together as
possible using ground plane construction or single point
grounding. For optimal performance, the NCP3126 should
have a layout similar to the one shown in Figure 30. An
important note is that the input voltage to the NCP3126
should have local decoupling to PGND. The recommended
decoupling for input voltage is a 1 mF general purpose
ceramic capacitor and a 0.01 mF COG ceramic capacitor
placed in parallel.
Output
Voltage
Output
Current
tss
Figure 28. Resistive Load Current
Alternatively, if the output has an under voltage lockout,
turns on at a defined voltage level, and draws a consistent
current, then the RMS connected load current is:
Ǹ
V OUT * V OUT_TO
798 mA +
IOUT
VOUT
VOUT_TO
V OUT
Ǹǒ
PGND
I OUT
ǒ3.3 V * 1.2 VǓ
3.3 V
Ǔ
CC
(eq. 51)
BST
AGND
VIN
1A
CP
RC
COMP
RF
ISET
FB
VSW
R1
PGND
COG
0.01 uF
= Output current
= Output voltage
= Output voltage load turn on
CF
R2
1 .0 uF
RISET
I CLI+
Top
AGND
3.3 V
1.0 V
PGND
Output
Voltage
CC
RC
ÎÎ
CP
BST
AGND
ÎÎÎÎÎ
Î
ÎÎÎÎÎ
Î
ÎÎÎÎÎ
ÎÎÎÎÎ
ÎÎÎÎÎ
COMP
RF
R2
R1
Output
Current
t
PGND
VIN
ISET
VSW
COG
0.01 uF
1 .0 uF
RISET
ÎÎ
ÎÎ
AGND
CF
FB
Bottom
Single Point
Grounding
Figure 30. Recommended Layout
tss
Figure 29. Voltage Enable Load Current
If the inrush current is higher than the steady state input
current during max load, then an input fuse should be rated
accordingly using I2t methodology.
The typical applications are shown in Figures 31
and NO TAG for output electrolytic and ceramic bulk
capacitors, respectively.
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20
NCP3126
BST
VIN
C3
470uF
16V
4
3
2
1
Vin
CC
56n
10V
AGND BST
COMP VIN
FB
ISET
PGND VSW
R3
5
6
7
8
RSET
39k
RC
2.1k
GND_IN
D1
U1
NCP3126
C1
10uF
16V
VSW
10R
C11
0.01uF
16V
C10
1uF
16V
CP COMP
1.8nF
50V
MMSD4148T1G
CBST
10nF
25V
C9
1nF
50V
LOUT
VOUT
6.8uH
R9
100R
CF
180pF
10V
R1
21.5k
C4
22uF
6.3V
RF
20k
FB
R2
10k
Figure 31. Standard Application 12 V to 2.5 V 3 A
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21
C7
470uF
6.3V
CHF
820pF
6.3V
Vout
GND
GND
NCP3126
Table 7. NCP3126 BOM
Item
Reference
Qty
Description
Value
Tolerance
FootPrint
Manufacturer
Manufacturer Part Name
1
CP
1
SMT Ceramic
Capacitor
1.8 nF
$10%
603
TDK
06031C182JAT2A
2
CC
1
SMT Ceramic
Capacitor
56 n
$10%
603
AVX
0603YC563KAT2A
3
C11
1
SMT Ceramic
Capacitor
10 nF
5%
603
TDK
C1608C0G1E103J
4
CF
1
SMT Ceramic
Capacitor
180 pF
5%
603
AVX
06035C181KAT2A
5
C10
1
SMT Ceramic
Capacitor
1 mF
10%
603
AVX
06033D105KAT2A
6
CHF
1
SMT Ceramic
Capacitor
820 pF
5%
603
AVX
06035A821JAT2A
7
C8
1
SMT Ceramic
Capacitor
NI
8
CBST
1
SMT Ceramic
Capacitor
10 nF
$20%
805
AVX
08055C103MAT2A
9
C9
1
SMT Ceramic
Capacitor
1 nF
$20%
805
AVX
08055C102KAT2A
10
C1
1
SMT Ceramic
Capacitor
10 mF
$10%
1210
AVX
1210YD106KAT2A
11
C4
1
SMT Ceramic
Capacitor
22 mF
$20%
1210
TDK
C3225X5R0J226M/2.00
12
C5
1
SMT Ceramic
Capacitor
NI
13
C3
1
Surface Mount
E−Cap
470 mF
$20%
8.00mm x 6.20mm
Panasonic
EEE−FP1C471AP
14
C7
1
Surface Mount
E−Cap
470 mF
$20%
(8.30 x 8.30)mm
Panasonic
EEE−FP1C471AP
15
C6
1
Surface Mount
E−Cap
NI
$20%
(10.3 x 10.3)mm
United Chemicon
EMZA160ADA471MHA0G
16
D1
1
Switching Diode
1 A, 30 V
SOD−123
ON Semiconductor
MMSD414851G
17
LOUT
1
INDUCTOR, SM
6.8 mH
20%
(12.3 x 12.3 x
8.1)mm
Coilcraft
MSS1278T−682MLB
18
U1
1
Synchronous PWM
Switching Converter
350 kHz
0.8 V
NA
SOIC−8
ON Semiconductor
NCP3126
19
RCR
1
SMT Resistor
NI
20
RC
1
SMT Resistor
21.1k
$1.0%
603
Vishay / Dale
CRCW060321K7FKEA
21
R2
1
SMT Resistor
10k
$1.0%
603
Vishay / Dale
CRCW060310K0FKEA
22
R3
1
SMT Resistor
1R
$5.0%
603
Vishay / Dale
CRCW06031R00JNEA
23
R4
1
SMT Resistor
20R
$1.0%
603
Vishay / Dale
CRCW060320R0FKEA
24
RF
1
SMT Resistor
20k
$1.0%
603
Vishay / Dale
CRCW060320K0FKEA
25
R1
1
SMT Resistor
21.5k
$1.0%
603
Vishay / Dale
CRCW060321K5FKEA
26
RSET
1
SMT Resistor
39k
$1.0%
603
Vishay / Dale
CRCW060339K0FKEA
27
R9
1
SMT Resistor
100R
$1.0%
1206
Vishay / Dale
CRCW1206100RFKEA
603
1206
ORDERING INFORMATION
Device
NCP3126ADR2G
Package
Shipping†
SOIC−8
(Pb−Free)
2500 / Tape & Reel
†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specifications Brochure, BRD8011/D.
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22
NCP3126
PACKAGE DIMENSIONS
SOIC−8 NB
CASE 751−07
ISSUE AJ
−X−
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
6. 751−01 THRU 751−06 ARE OBSOLETE. NEW
STANDARD IS 751−07.
A
8
5
S
B
0.25 (0.010)
M
Y
M
1
4
−Y−
K
G
C
N
DIM
A
B
C
D
G
H
J
K
M
N
S
X 45 _
SEATING
PLANE
−Z−
0.10 (0.004)
H
D
0.25 (0.010)
M
Z Y
S
X
M
J
S
MILLIMETERS
MIN
MAX
4.80
5.00
3.80
4.00
1.35
1.75
0.33
0.51
1.27 BSC
0.10
0.25
0.19
0.25
0.40
1.27
0_
8_
0.25
0.50
5.80
6.20
INCHES
MIN
MAX
0.189
0.197
0.150
0.157
0.053
0.069
0.013
0.020
0.050 BSC
0.004
0.010
0.007
0.010
0.016
0.050
0 _
8 _
0.010
0.020
0.228
0.244
SOLDERING FOOTPRINT*
1.52
0.060
7.0
0.275
4.0
0.155
0.6
0.024
1.270
0.050
SCALE 6:1
mm Ǔ
ǒinches
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should
Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates,
and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death
associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal
Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
LITERATURE FULFILLMENT:
Literature Distribution Center for ON Semiconductor
P.O. Box 5163, Denver, Colorado 80217 USA
Phone: 303−675−2175 or 800−344−3860 Toll Free USA/Canada
Fax: 303−675−2176 or 800−344−3867 Toll Free USA/Canada
Email: [email protected]
N. American Technical Support: 800−282−9855 Toll Free
USA/Canada
Europe, Middle East and Africa Technical Support:
Phone: 421 33 790 2910
Japan Customer Focus Center
Phone: 81−3−5773−3850
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23
ON Semiconductor Website: www.onsemi.com
Order Literature: http://www.onsemi.com/orderlit
For additional information, please contact your local
Sales Representative
NCP3126/D