LT3509 Dual 36V, 700mA Step-Down Regulator Features Description Two 700mA Switching Regulators with Internal Power Switches n Wide 3.6V to 36V Operating Range n Overvoltage Lockout Protects Circuit Through 60V Supply Transients n Short-Circuit Robust n Low Dropout Voltage: 95% Maximum Duty Cycle n Adjustable 300kHz to 2.2MHz Switching Frequency Synchronizable Over the Full Range n Uses Small Inductors and Ceramic Capacitors n Integrated Boost Diodes n Internal Compensation n Thermally Enhanced 14-Lead (4mm × 3mm) DFN and 16 Lead MSOP Packages The LT®3509 is a dual, current mode, step-down switching regulator, with internal power switches each capable of providing 700mA output current. This regulator provides a compact and robust solution for multi-rail systems in harsh environments. It incorporates several protection features including overvoltage lockout and cycle-by-cycle current limit. Thermal shutdown provides additional protection. The loop compensation components and the boost diodes are integrated on-chip. Switching frequency is set by a single external resistor. External synchronization is also possible. The high maximum switching frequency allows the use of small inductors and ceramic capacitors for low ripple. Constant frequency operation above the AM band avoids interference with radio reception, making the LT3509 well suited for automotive applications. Each regulator has an independent shutdown and soft-start control pin. When both converters are powered down, the common circuitry enters a low current shutdown state. n Applications n n n n n Automotive Electronics Industrial Controls Wall Transformer Regulation Networking Devices CPU, DSP, or FPGA Power L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Typical Application 3.3V and 5V Dual Output Step-Down Converter 2.2µF VIN BD BOOST1 BOOST2 80 0.1µF 10µH SW1 5V 700mA SW2 LT3509 31.6k 53.6k MBRM140 DA1 DA2 FB1 FB2 MBRM140 22µF 10.2k 1nF SYNC 1nF RT GND VOUT = 3.3V 75 70 65 60 RUN/SS1 RUN/SS2 10µF VOUT = 5V 85 0.1µF 6.8µH 3.3V 700mA Efficiency 90 EFFICIENCY (%) 6.5V TO 36V (TRANSIENT TO 60V) 60.4k fSW = 700kHz 55 10.2k 3509 TA01a 50 0.0 VIN = 12V fSW = 700kHz 0.1 0.2 0.3 0.4 0.5 LOAD CURRENT (A) 0.6 0.7 3509 TA01b 3509fd For more information www.linear.com/LT3509 1 LT3509 Absolute Maximum Ratings (Note 1) VIN Pin (Note 2).........................................................60V BD Pin........................................................................20V BOOST Pins...............................................................60V BOOST Pins above SW..............................................30V RUN/SS, FB, RT, SYNC pins.........................................6V Operating Junction Temperature Range (Notes 3, 6) LT3509E............................................. –40°C to 125°C LT3509I.............................................. –40°C to 125°C LT3509H............................................. –40°C to 150°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec.) MSOP Package.................................................. 300°C pin configuration TOP VIEW TOP VIEW DA1 1 14 FB1 BOOST1 2 13 RUN/SS1 SW1 3 VIN 4 SW2 5 10 RT BOOST2 6 9 RUN/SS2 DA2 7 8 FB2 DA1 BOOST1 SW1 VIN VIN SW2 BOOST2 DA2 12 BD 15 11 SYNC 1 2 3 4 5 6 7 8 17 16 15 14 13 12 11 10 9 FB1 RUN/SS1 AGND BD SYNC RT RUN/SS2 FB2 MSE PACKAGE 16-LEAD PLASTIC MSOP DE14 PACKAGE 14-LEAD (4mm × 3mm) PLASTIC DFN θJA = 43°C/W, θJC = 4.3°C/W EXPOSED PAD (PIN 15) IS GND, MUST BE SOLDERED TO PCB θJA = 43°C/W, θJC = 4.3°C/W EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB order information LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3509EDE#PBF LT3509EDE#TRPBF 3509 14-Lead (4mm × 3mm) Plastic DFN –40°C to 125°C LT3509IDE#PBF LT3509IDE#TRPBF 3509 14-Lead (4mm × 3mm) Plastic DFN –40°C to 125°C LT3509EMSE#PBF LT3509EMSE#TRPBF 3509 16-Lead Plastic MSOP with Exposed Pad –40°C to 125°C LT3509IMSE#PBF LT3509IMSE#TRPBF 3509 16-Lead Plastic MSOP with Exposed Pad –40°C to 125°C LT3509HMSE#PBF LT3509HMSE#TRPBF 3509 16-Lead Plastic MSOP with Exposed Pad –40°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. 2 3509fd For more information www.linear.com/LT3509 LT3509 Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C, VIN = 12V. (Note 3) PARAMETER CONDITIONS MIN TYP MAX 3.3 3.6 V 37 38.5 40 V 1.9 2.2 mA 9 15 µA 0.8 0.816 V VIN Undervoltage Lockout VIN Overvoltage Lockout Input Quiescent Current Not Switching VFB > 0.8V Input Shutdown Current V(RUN/SS[1,2]) < 0.3V Feedback Pin Voltage Reference Voltage Line Regulation l 0.784 3.6V < VIN < 36V 0.01 RUN/SS Shutdown Threshold 0.4 0.6 RUN/SS Voltage for Full IOUT RUN/SS Pin Pull-up Current Feedback Pin Bias Current (Note 4) 0.7 VFB = 0.8V l DA Comparator Current Threshold Boost Pin Current %/V 0.8 V 2 V 1 1.3 µA 90 500 nA 1.05 1.4 1.9 A 0.7 0.95 1.2 A 22 36 mA 0.01 1.0 µA l Switch Current Limit UNITS ISW = 0.9A Switch Leakage Current Switch Saturation Voltage ISW = 0.9A (Note 5) 0.32 Minumum Boost Voltage above Switch ISW = 0.9A 1.5 2.2 Boost Diode Forward Voltage IBD= 20mA 0.7 0.9 V Boost Diode Leakage VR = 30V 0.1 5 µA Switching Frequency RT = 40.2kΩ RT = 180kΩ RT = 14.1kΩ 1.0 260 2.15 1.08 290 2.5 l Sync Pin Input Threshold 0.92 237 2.0 V 1.0 Switch Minimum Off-Time l Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2. Absolute Maximum Voltage at the VIN pin is 60V for non-repetitive 1 second transients and 36V for continuous operation. Note 3. The LT3509E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3509I is guaranteed over the full –40°C to 125°C temperature range. The LT3509H is guaranteed 80 V MHz kHz MHz V 150 ns over the full –40°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125°C. Note 4. Current flows out of pin. Note 5. Switch Saturation Voltage is guaranteed by design. Note 6. This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed the maximum operating temperature when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. 3509fd For more information www.linear.com/LT3509 3 LT3509 Typical Performance Characteristics Efficiency vs Load Current VOUT = 5V, fSW = 2.0MHz 95 95 TA = 25ºC 85 EFFICIENCY (%) 80 VIN = 24V 75 70 65 0 0.4 0.2 0.6 75 70 75 65 60 60 55 0 0.2 0.4 0.6 VIN = 12V 70 65 50 0.8 0 0.4 0.2 ILOAD(A) ILOAD(A) 0.6 0.8 ILOAD(A) 3509 G02 3509 G01 Switch VCE(SAT) vs ISW 0.35 80 VIN = 12V 80 55 0.8 TA = 25ºC 85 EFFICIENCY (%) VIN = 12V 85 EFFICIENCY (%) 90 TA = 25ºC 90 90 60 Efficiency vs Load Current VOUT = 1.8V, fSW = 0.7MHz Efficiency vs Load Current VOUT = 3.3V, fSW = 2.0MHz 3509 G03 IBOOST vs ISW 25 TA = 25ºC TA = 25ºC 0.3 20 IBOOST (mA) VCE(SAT) (V) 0.25 0.2 0.15 15 10 0.1 5 0.05 0 0 0.2 0.4 0.6 ISW(A) 0.8 0 1.0 0 0.2 0.4 0.6 ISW(A) 3509 G04 1 3509 G05 Boost Diode Characteristics 1.2 0.8 2.2 TA = 25ºC Frequency vs RT TA = 25ºC 2.0 FREQUENCY (MHz) 1 Vf (V) 0.8 0.6 0.4 1.0 0.5 0.2 0 1.5 0 50 100 BOOST DIODE CURRENT (mA) 150 0 0 20 3509 G06 4 40 60 80 100 120 140 160 180 RT(kΩ) 3509 G07 3509fd For more information www.linear.com/LT3509 LT3509 Typical Performance Characteristics fSW vs Temperature (Measured at 1MHz) FB Pin Voltage vs Temperature 0.810 1.04 30 0.800 0.795 1.02 1.01 1.00 0.99 0 0.97 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 45 160 TA = 25ºC 140 TA = 85ºC 120 MINIMUM ON-TIME (ns) MAXIMUM VIN (V) 30 25 20 15 10 10 0 0.2 0.4 ILOAD (A) 0.6 0.8 3509 G10 Minimum On-Time vs Temperature, ILOAD = 0.3A 100 80 60 40 20 5 0 15 3509 G09 Max VIN for Constant Frequency VOUT = 5V, fSW = 2MHz 35 20 0 25 50 75 100 125 150 TEMPERATURE (˚C) 3509 G08 40 TA = 85ºC 5 0.98 0.790 –50 –25 TA = 25ºC 25 MAXIMUM VIN (V) NORMALIZED (fSW) 0.805 0 0.2 0.4 0.6 0 –50 –25 0.8 ILOAD (A) 0 25 50 75 100 125 150 TEMPERATURE (°C) 3509 G12 3509 G11 ILIM vs Temperature ILIM vs Duty Cycle 1.5 1.8 TA = 25ºC 1.6 1.3 SWITCH 1.4 1.2 1.0 DA ILIM (A) ILIM (A) FEEDBACK PIN VOLTAGE (V) 1.03 Max VIN for Constant Frequency VOUT = 3.3V, fSW = 2MHz 0.9 1 0.8 0.6 0.4 0.7 0.2 0.5 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 0 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 3509 G13 3509 G14 3509fd For more information www.linear.com/LT3509 5 LT3509 Pin Functions (DFN/MSOP) DA1, DA2 (Pins 1, 7/Pins 1, 8): The DA pins are the anode connections for the catch diodes. These are connected internally to the exposed ground pad by current sensing resistors. BOOST1, BOOST2 (Pins 2, 6/Pins 2, 7): The BOOST pins are used to dynamically boost the power transistor base above VIN to minimize the voltage drop and power loss in the switch. These should be tied to the associated switch pins through the boost capacitors. SW1, SW2 (Pins 3, 5/Pins 3, 6): The SW pins are the internal power switch outputs. These should be connected to the associated inductors, catch diode cathodes, and the boost capacitors. VIN (Pin 4/Pins 4, 5): The VIN pins supply power to the internal power switches and control circuitry. In the MSE package the VIN pins must be tied together. The input capacitor should be placed as close as possible to the supply pins. FB1, FB2 (Pins 14, 8/Pins 16, 9): The FB pins are used to set the regulated output voltage relative to the internal reference. These pins should be connected to a resistor divider from the regulated output such that the FB pin is at 0.8V when the output is at the desired voltage. RUN/SS1, RUN/SS2 (Pins 13, 9/Pins 15, 10): The RUN/SS pins enable the associated regulator channel. If both pins are pulled to ground, the device will shut-down to a low power state. In the range 0.8V to 2V, the regulators are enabled but the peak switch current and the DA pin maximum current are limited to provide a soft-start function. Above 2V, the full output current is available. The inputs incorporate a 1µA pull-up so that they will float high or charge an external capacitor to provide a current limited soft-start. The pins are pulled down by approximately 250µA 6 in the case of overvoltage or overtemperature conditions in order to discharge the soft-start capacitors. The pins can also be driven by a logic control signal of up to 5.0V. In this case, it is necessary place a 10k to 50k resistor in series along with a capacitor from the RUN/SS pin to ground to ensure that there will be a soft-start for both initial turn on and in the case of fault conditions. Do not tie these pins to VIN. RT (Pin 10/Pin 11): The RT pin is used to set the internal oscillator frequency. A 40.2k resistor from RT to ground results in a nominal frequency of 1MHz. SYNC (Pin 11/Pin12): The SYNC pin allows the switching frequency to be synchronized to a external clock. Choose RT resistor to set a free-run frequency at least 12% less than the external clock frequency for correct operation. The SYNC pin should not be allowed to float; if not used, it should be tied low through a resistance 10kΩ or less. BD (Pin 12/Pin 13): The BD pin is common anode connection of the internal Schottky boost diodes. This provides the power for charging the BOOST capacitors. It should be locally bypassed for best performance. Exposed Pad (Pin 15/Pin 17): GND. This is the reference and supply ground for the regulator. The exposed pad must be soldered to the PCB and electrically connected to supply ground. Use a large ground plane and thermal vias to optimize thermal performance. The current in the catch diodes also flows through the GND pad to the DA pins. AGND (Pin 14, MSOP Package Only): This is the connected to the ground connection of the chip and may be used as a separate return for the low current control side components. It should not be used as the only ground connection or as a connection return for load side components. 3509fd For more information www.linear.com/LT3509 LT3509 Block Diagram COMMON CIRCUITRY 1 OF 2 REGULATOR CHANNELS SHOWN VIN BD C1 OVERVOLTAGE DETECT BOOST RUN/SS1 MAIN CURRENT COMPARATOR SHUTDOWN AND SOFT-START CONTROL C3 VREF AND CORE VOLTAGE REGULATOR SYNC RT RT SWITCH LOGIC RUN/SS2 C2 NOTE: THE BD PIN IS COMMON TO BOTH CHANNELS BOOST DIODE SWITCH DRIVER DA CURRENT COMPARATOR – POWER SWITCH ERROR AMPLIFIER OSCILLATOR L1 SW VREF 0.8V VOUT D1 –17mV DA VC SLOPE C4 C5 R1 18mΩ FB R2 CLAMP GND 3509 BD Figure 1. Functional Block Diagram 3509fd For more information www.linear.com/LT3509 7 LT3509 Operation Overview The LT3509 is a dual, constant frequency, current mode switching regulator with internal power switches. The two independent channels share a common voltage reference and oscillator and operate in phase. The switching frequency is set by a single resistor and can also be synchronized to an external clock. Operation can be best understood by referring to the Block Diagram (Figure 1). Startup and Shutdown When the RUN/SS[1,2] pins are pulled low (<0.4V) the associated regulator channel is shut down. If both channels are shut down, the common circuitry also enters a low current state. When the RUN/SS pins exceed approximately 0.8V, the common circuitry and the associated regulator are enabled but the output current is limited. From 0.8V up to 2.0V the current limit increases until it reaches the full value. The RUN/SS pins also incorporate a 1µA pullup to approximately 3V, so the regulator will run if they are left open. A capacitor to ground will cause a current limited soft-start to occur at power-up. In the case of undervoltage, overvoltage or overtemperature conditions the internal circuitry will pull the RUN/SS pins down with a current of approximately 250µA. Thus a new soft-start cycle will occur when the fault condition ends. Voltage and Current Regulation The power switches are controlled by a current-mode regulator architecture. The power switch is turned on at the beginning of each clock cycle and turned off by the main current comparator. The inductor current will ramp up while the switch is on until it reaches the peak current threshold. The current at which it turns off is determined by the error amp and the internal compensation network. When the switch turns off, the current in the inductor will cause the SW pin to fall rapidly until the catch diode, D1, conducts. The voltage applied to the inductor will now reverse and the current will linearly fall. The resistor divider, R1 and R2, sets the desired output voltage such 8 that when the voltage at FB reaches 0.8V, the main current comparator threshold will fall and reduce the peak inductor current and hence the average current, until it matches the load current. By making current the controlled variable in the loop, the inductor impedance is effectively removed from the transfer function and the compensation network is simplified. The main current comparator threshold is reduced by the slope compensation signal to eliminate sub-harmonic oscillations at duty cycles >50%. Current Limiting Current mode control provides cycle-by-cycle current limiting by means of a clamp on the maximum current that can be provided by the switch. A comparator monitors the current flowing through the catch diode via the DA pin. This comparator delays switching if the diode current is higher than 0.95A (typical). This current level is indicative of a fault condition such as a shorted output with a high input voltage. Switching will only resume once the diode current has fallen below the 0.95A limit. This way the DA comparator regulates the valley current of the inductor to 0.95A during a short circuit. This will ensure the part will survive a short-circuit event. Over and Undervoltage Shutdown A basic undervoltage lockout prevents switching if VIN is below 3.3V (typical). The overvoltage shutdown stops the part from switching when VIN is greater than 38.5V (typical). This protects the device and its load during momentary overvoltage events. After the input voltage falls below 38.5V, the part initiates a soft start sequence and resumes switching. BOOST Circuit To ensure best efficiency and minimum dropout voltage the output transistor base drive is boosted above VIN by the external boost capacitors (C4). When the SW pin is low the capacitors are charged via the BOOST diodes and the supply on BD. 3509fd For more information www.linear.com/LT3509 LT3509 APPLICATIONS INFORMATION Shutdown and Soft Start Setting The Output Voltage When the RUN/SS pins are pulled to ground, the part will shut down to its lowest current state of approximately 9µA. If driving a large capacitive load it may be desirable to use the current limiting soft-start feature. Connecting capacitors to ground from the RUN/SS pins will control the delay until full current is available. The pull-up current is 1µA and the full current threshold is 2V so the start-up time is given by: The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the resistors according to: T = 2 • C • 106 s For example a 0.005µF capacitor will give a time to full current of 10ms. If both outputs can come up together then the two inputs can be paralleled and tied to one capacitor. In this case use twice the capacitor value to obtain the same start-up time. During the soft-start time both the peak current threshold and the DA current threshold will track so the part will skip pulses as required to limit the maximum inductor current. Starting up into a large capacitor is not much different to starting into a shortcircuit in this respect. VSW 10V/DIV IL 0.2A/DIV VOUT 5V/DIV TIME 1ms/DIV Figure 2. Soft-Start 3509 F02 R1=R2 • VOUT –1 0.8 The designators correspond to Figure 1. R2 should be 20k or less to avoid bias current errors. Frequency Setting The timing resistor, RT , for any desired frequency in the range 264kHz to 2.2MHz can be calculated from the following formula: 1.215 RT = – 0.215 • 40.2 fSW where fSW is in MHz and RT is in kΩ. Table 1. Standard E96 Resistors for Common Frequencies FREQUENCY TIMING RESISTOR RT (kΩ) 264 kHz 178 300 kHz 154 400kHz 113 500kHz 88.7 1MHz 40.2 2MHz 15.8 2.2MHz 13.7 Note: The device is specified for operation down to 300kHz. The 264kHz value is to allow external synchronization at 300kHz 3509fd For more information www.linear.com/LT3509 9 LT3509 APPLICATIONS INFORMATION External Synchronization The external synchronization provides a trigger to the internal oscillator. As such, it can only raise the frequency above the free-run value. To allow for device and component tolerances, the free run frequency should be set to at least 12% lower than the lowest supplied external synchronization reference. The oscillator and hence the switching frequency can then pushed up from 12% above the free-run frequency, set by the selected RT. For example, if the minimum external clock is 300kHz, the RT should be chosen for 264kHz. The SYNC input has a threshold of 1.0V nominal so it is compatible with most logic levels. The duty cycle is not critical provided the high or low pulse width is at least 80ns. If not used, the SYNC input should be tied low with 10kΩ less to avoid noise pickup. Design Procedure Before starting detailed design a number of key design parameters should be established as these may affect design decisions and component choices along the way. One of the main things to determine apart from the desired output voltages is the input voltage range. Both the normal operating range and the extreme conditions of surges and/or dips or brown-outs need to be known. Then the operating frequency should be considered and if there are particular requirements to avoid interference. If there are very specific frequencies that need to be avoided then external synchronization may be needed. This could also be desirable if multiple switchers are used as low frequency beating between similar devices can be undesirable. For efficient operation this converter requires a boost supply so that the base of the output transistor can be pumped above 10 the input voltage during the switch on time. Depending on the input and output voltages the boost supply can be provided by the input voltage, one of the regulated outputs or an independent supply such as an LDO. Input Voltage Range Firstly, the LT3509 imposes some hard limits due to the undervoltage lock-out and the overvoltage protection. A given application will also have a reduced, normal operating range over which maximum efficiency and lowest ripple are obtained. This usually requires that the device is operating at a fixed frequency without skipping pulses. There may also be zones above and below the normal range where regulation is maintained but efficiency and ripple may be compromised. At the low end, insufficient input voltage will cause loss of regulation and increased ripple—this is the dropout range. At the high end if the duty cycle becomes too low this will cause pulse skipping and excessive ripple. This is the pulse‑skip region. Both situations also lead to higher noise at frequencies other than the chosen switching frequency. Occasional excursions into pulse-skip mode, during surges for example, may be tolerable. Pulse skipping will also occur at light loads even within the normal operating range but ripple is usually not degraded because at light load the output capacitor can hold the voltage steady between pulses. For input voltages greater than 30V, there are restrictions on the inductor value. See the Inductor Selection section for details. To ensure the regulator is operating in continuous mode it is necessary to calculate the duty cycle for the required output voltage over the full input voltage range. This must then be compared with minimum and maximum practical duty cycles. 3509fd For more information www.linear.com/LT3509 LT3509 APPLICATIONS INFORMATION In any step-down switcher the duty cycle when operating in continuous, or fixed frequency, mode is dependent on the step-down ratio. This is because for a constant average load current the decay of the inductor current when the switch is off must match the increase in inductor current when the switch is on. The can be estimated by the following formula: DC = where: VOUT + VF VIN − VSW + VF The minimum on time increases with increasing temperature so the value for the maximum operating temperature should be used. See the Minimum On-Time vs Temperature graph in the Typical Performance Characteristics. The maximum input voltage for this duty cycle is given by: DC = Duty Cycle (Fraction of Cycle when Switch is On) VOUT = Output Voltage VIN = Input Voltage VIN(MAX ) = VOUT + VF − VF + VSW DCMIN Above this voltage the only way the LT3509 can maintain regulation is to skip cycles so the effective frequency will reduce. This will cause an increase in ripple and the switching noise will shift to a lower frequency. This calculation will in practice drive the maximum switching frequency for a desired step-down ratio. VF = Catch Diode Forward Voltage VSW = Switch Voltage Drop Note: This formula neglects switching and inductor losses so in practice the duty cycle may be slightly higher. It is clear from this equation that the duty cycle will approach 100% as the input voltage is reduced and become smaller as the input voltage increases. There are practical limits to the minimum and maximum duty cycles for continuous operation due to the switch minimum off and on times. These are independent of operating frequency so it is clear that range of usable duty cycle is inverserly proportional to frequency. Therefore at higher frequency the input voltage range (for constant frequency operation) will narrow. The minimum duty cycle is given by: DCMIN = fSW • t ON(MIN) VOUT 100mV/DIV (AC COUPLED) IL 0.5A/DIV TIME 1µs/DIV 3509 F03 Figure 3. Continuous Mode VOUT 100mV/DIV (AC COUPLED) IL 0.5A/DIV where: fSW = Switching Frequency TIME 1µs/DIV tON(MIN) = Switch Minimum On-Time 3509 F04 Figure 4. Pulse Skipping 3509fd For more information www.linear.com/LT3509 11 LT3509 APPLICATIONS INFORMATION Minimum Input Voltage and Boost Architecture The minimum operating voltage is determined either by the LT3509’s internal undervoltage lockout of ~3.6V or by its maximum duty cycle. The maximum duty cycle for fixed frequency operation is given by: DCMAX = 1− tOFF(MIN) • fSW It follows that: VIN(MIN) = VOUT + VF − VF + VSW DCMAX If a reduction in switching frequency can be tolerated the minimum input voltage can drop to just above output voltage. Not only is the output transistor base pumped above the input voltage by the boost capacitor, the switch can remain on through multiple switching cycles resulting in a high effective duty cycle. Thus, this is a true low dropout regulator. As it is necessary to recharge the boost capacitor from time to time, a minimum width off-cycle will be forced occasionally to maintain the charge. Depending on the operating frequency, the duty cycle can reach 97% to 98%, although at this point the output pulses will be at a sub-multiple of the programmed frequency. One other consideration is that at very light loads or no load the part will go into pulse skipping mode. The part will then have trouble getting enough voltage on to the boost capacitors to fully saturate the switch. This is most problematic when the BD pin is supplied from the regulated output. The net result is that a higher input voltage will be required to start up the boost system. The typical minimum input voltage over a range of loads is shown in Figure 5 for 3.3V and Figure 6 for 5V. When operating at such high duty cycles the peak currents in the boost diodes are greater and this will require a the BD supply to be somewhat higher than would be required at less extreme duty cycles. If operation at low input/output ratios and low BD supply voltages is required it may be desirable to augment the internal boost diodes with external discrete diodes in parallel. Boost Pin Considerations The boost capacitor, in conjunction with the internal boost diode, provides a bootstrapped supply for the power switch that is above the input voltage. For operation at 1MHz and above and at reasonable duty cycles a 0.1µF capacitor will work well. For operation at lower frequencies and/or higher duty cycles something larger may be needed. A good rule of thumb is: C BOOST = where fSW is in MHz and CBOOST is in µF 7 5.5 5 TO START VIN TO START (V) VIN TO START (V) TO RUN 3.5 3 6 TO RUN 5.5 5 4.5 2.5 2 0.001 TO START 6.5 4.5 4 1 10 • fSW 0.1 0.01 LOAD CURRENT (A) 1 4 0.001 3509 F05 Figure 5. Minimum VIN for 3.3V VOUT 12 0.1 0.01 LOAD CURRENT (A) 1 3509 F06 Figure 6. Minimum VIN for 5V VOUT 3509fd For more information www.linear.com/LT3509 LT3509 APPLICATIONS INFORMATION Boost Pin Considerations Figure 7 through Figure 9 show several ways to arrange the boost circuit. The BOOST pin must be more than 2V above the SW pin for full efficiency. For outputs of 3.3V and higher, the standard circuit Figure 7 is best. For lower output voltages, the boost diode can be tied to the input Figure 8. The circuit in Figure 7 is more efficient because the boost pin current comes from a lower voltage source. Finally, as shown in Figure 9, the BD pin can be tied to another source that is at least 3V. For example, if you are generating 3.3V and 1.8V, and the 3.3V is on whenever the 1.8V is on, the 1.8V boost diode can be connected to the 3.3V output. VIN BD VIN BOOST LT3509 CBOOST L1 VOUT SW CIN D1 GND COUT DA 3509 F07 VBOOST – VSW ≅ VOUT MAX VBOOST ≅ VIN + VOUT VOUT ≥ 3V Figure 7. BD Tied to Regulated Output BD BOOST VIN VIN A good first choice for the inductor value is: L = ( VOUT + VF ) • VOUT D1 DA 2.1MHz fSW where VF is the voltage drop of the catch diode (~0.5V) and L is in µH. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For highest efficiency, the series resistance (DCR) should be less than 0.15Ω. Table 2 lists several vendors and types that are suitable. The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to-peak inductor ripple current. The LT3509 limits its switch current in order to protect itself and the system from overcurrent faults. Therefore, the maximum output current that the LT3509 will deliver depends on the switch current limit, the inductor value and the input and output voltages. VBD BD VIN VIN BOOST LT3509 L1 SW GND Inductor Selection and Maximum Output Current CBOOST LT3509 CIN In any case, be sure that the maximum voltage at the BOOST pin is less than 60V and the voltage difference between the BOOST and SW pins is less than 30V. L1 VOUT SW CIN D1 GND COUT CBOOST DA COUT 3509 F09 3509 F08 VBOOST – VSW ≅ VIN MAX VBOOST ≅ 2VIN VBOOST – VSW ≅ VBD MAX VBOOST ≅ VIN + VBD VBD ≥ 3V Figure 9. Separate Boost Supply Figure 8. Supplied from VIN 3509fd For more information www.linear.com/LT3509 13 LT3509 APPLICATIONS INFORMATION When the switch is off, the potential across the inductor is the output voltage plus the catch diode forward voltage. This gives the peak-to-peak ripple current in the inductor: ∆IL =(1– DC) where: VOUT + VF L • fSW DC = Duty Cycle fSW = Switching Frequency L = Inductor Value VF = Diode Forward Voltage The peak inductor and switch current is: ISWPK =ILPK =IOUT + ∆IL 2 To maintain output regulation, this peak current must be less than the LT3509’s switch current limit ILIM. This is dependent on duty cycle due to the slope compensation. For ILIM is at least 1.4A at low duty cycles and decreases linearly to 1.0A at DC = 0.8. The theoretical minimum inductance can now be calculated as: LMIN = 1– DCMIN VOUT + VF • ILIM – IOUT f where DCMIN is the minimum duty cycle called for by the application i.e.: DCMIN = 14 There is a limit to the actual minimum duty cycle imposed by the minimum on-time of the switch. For a robust design it is important that inductor that will not saturate when the switch is at its minimum on-time, the input voltage is at maximum and the output is short circuited. In this case the full input voltage, less the drop in the switch, will appear across the inductor. This doesn’t require an actual short, just starting into a capacitive load will provide the same conditions. The Diode current sensing scheme will ensure that the switch will not turn-on if the inductor current is above the DA current limit threshold, which has a maximum of 1.1A. The peak current under short-circuit conditions can then be calculated from: IPEAK = VIN • tON(MIN) L + 1.1A The inductor should have a saturation current greater than this value. For safe operation with high input voltages this can often mean using a physically larger inductor as higher value inductors often have lower saturation currents for a given core size. As a general rule the saturation current should be at least 1.8A to be short-circuit proof. However, it’s generally better to use an inductor larger than the minimum value. For robust operation at input voltages greater than 30V, use an inductor with a value of 4.2µH or greater, and a saturation current rating of 1.8A or higher. The minimum inductor has large ripple currents which increase core losses and require large output capacitors to keep output voltage ripple low. Select an inductor greater than LMIN that keeps the ripple current below 30% of ILIM. VOUT(MAX ) + VF VIN(MIN) – VSW + VF 3509fd For more information www.linear.com/LT3509 LT3509 APPLICATIONS INFORMATION Table 2. Recommended Inductors MANUFACTURER/ PART NUMBER VALUE (µH) ISAT (A) DCR (W) HEIGHT (mm) LPS4018-222ML 2.2 2.8 0.07 1.7 LPS5030-332ML 3.3 2.5 0.066 2.9 LPS5030-472ML 4.7 2.5 0.083 2.9 LPS6225-682ML 6.8 2.7 0.095 2.4 LPS6225-103ML 10 2.1 0.105 2.4 CDRH4D22/HP-2R2N 2.2 3.2 0.0035 2.4 CDRH4D22/HP-3R5N 3.5 2.5 0.052 2.4 CDRH4D22/HP-4R7N 4.7 2.2 0.066 2.4 CDRH5D28/HP-6R8N 6.8 3.1 0.049 3.0 CDRH5D28/HP-8R2N 8.2 2.7 0.071 3.0 CDRH5D28R/HP-100N 10 2.45 0.074 3.0 SD52-2R2-R 2.2 2.30 0.0385 2.0 SD52-3R5-R 3.5 1.82 0.0503 2.0 SD52-4R7-R 4.7 1.64 0.0568 2.0 SD6030-5R8-R 5.8 1.8 0.045 3.0 SD7030-8R0-R 8.0 1.85 0.058 3.0 SD7030-100-R 10.0 1.7 0.065 3.0 A997AS-2R2N 2.2 1.6 0.06 1.8 A997AS-3R3N 3.3 1.2 0.07 1.8 A997AS-4R7M 4.7 1.07 0.1 1.8 7447745022 2.2 3.5 0.036 2.0 7447745033 3.3 3.0 0.045 2.0 7447745047 4.7 2.4 0.057 2.0 7447745076 7.6 1.8 0.095 2.0 7447445100 10 1.6 0.12 2.0 Coilcraft Sumida Cooper Toko Würth The prior analysis is valid for continuous mode operation (IOUT > ∆ILIM / 2). For details of maximum output current in discontinuous mode operation, see Linear Technology’s Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), a minimum inductance is required to avoid subharmonic oscillations. This minimum inductance is L MIN = ( VOUT + VF )• 1.4 fSW where fSW is in MHz and LMIN is in µH. If using external synchronization, calculate LMIN using the RT frequency and not the SYNC frequency. Frequency Compensation The LT3509 uses current mode control to regulate the output, which simplifies loop compensation and allows the necessary filter components to be integrated. The fixed internal compensation network has been chosen to give stable operation over a wide range of operating conditions but assumes a minimum load capacitance. The LT3509 does not depend on the ESR of the output capacitor for stability so the designer is free to use ceramic capacitors to achieve low output ripple and small PCB footprint. Figure 10 shows an equivalent circuit for the LT3509 control loop. The error amp is a transconductance amplifier with finite output impedance. The power section, consisting of the modulator, power switch and inductor is modeled as a transconductance amplifier generating an output current proportional to the voltage at the COMP-NODE. The gain of the power stage (gmp) is 1.1S. Note that the output capacitor integrates this current and that the internal capacitor integrates the error amplifier output current, resulting in two poles in the loop. In most cases, a zero is required and comes either from the output capacitor ESR 3509fd For more information www.linear.com/LT3509 15 LT3509 APPLICATIONS INFORMATION or from RC. This model works well as long as the inductor current ripple is not too low (∆IRIPPLE > 5% IOUT) and the loop crossover frequency is less than fSW/5. An optional phase lead capacitor (CPL) across the feedback divider may improve the transient response. LT3509 1.1S VOUT VIN CPL RC 260µS R1 + – COMPNODE 75k COUT VREF = 0.8V 95pF 1.73M R2 3509 F10 Figure 10. Small-Signal Equivalent Circuit Output Capacitor Selection The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilize the LT3509’s control loop. Because the LT3509 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option. 16 You can estimate output ripple with the following equations. For ceramic capacitors where low capacitance value is more significant than ESR: VRIPPLE = ∆IL / (8 • fSW • COUT ) For electrolytic capacitors where ESR is high relative to capacitive reactance: VRIPPLE = ∆IL • ESR where ∆IL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula: IC(RMS) = ∆IL / 12 Another constraint on the output capacitor is that it must have greater energy storage than the inductor; if the stored energy in the inductor transfers to the output, the resulting voltage step should be small compared to the regulation voltage. For a 5% overshoot, this requirement indicates: COUT > 10 • L •(ILIM / VOUT )2 The low ESR and small size of ceramic capacitors make them the preferred type for LT3509 applications. Not all ceramic capacitors are the same, however. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V 3509fd For more information www.linear.com/LT3509 LT3509 APPLICATIONS INFORMATION and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT, this loss may be unacceptable. Use X7R and X5R types. the top feedback resistor. The small-signal model shown in Figure 10 can be used to model this in a simulator or to give insight to an empirical design. Figure 11 shows some load step responses with differing output capacitors and CPL combinations. The value of the output capacitor greatly affects the transient response to a load step. It has to supply extra current demand or absorb excess current delivery until the feedback loop can respond. The loop response is dependent on the error amplifier transconductance, the internal compensation capacitor and the feedback network. Higher output voltages necessarily require a larger feedback divider ratio. This will also reduce the loop gain and slow the response time. Fortunately this effect can be mitigated by use of a feed-forward capacitor, CPL, across Input Capacitor The input capacitor needs to supply the pulses of charge demanded during the on time of the switches. Little total capacitance is required as a few hundred millivolts of ripple at the VIN pin will not cause any problems to the device. When operating at 2MHz and 12V, 2µF will work well. At the lowest operating frequency and/or at low input voltages a larger capacitor such as 4.7µF is preferred. ILOAD 700mA 300mA ILOAD 700mA 300mA VOUT (AC) 50mV/DIV VOUT (AC) 50mV/DIV COUT = 10µF CPL = 0 TIME 20µs/DIV COUT = 10µF CPL = 82pF TIME 20µs/DIV 3509 F11 Figure 11. Transient Load Response with Different Combinations of COUT and CPL Load Current Step from 300mA to 700mA R1 = 10k, R2 = 32.4k, VIN = 12V, VOUT = 3.3V, fSW = 2.0MHz 3509fd For more information www.linear.com/LT3509 17 LT3509 APPLICATIONS INFORMATION Diode Selection The catch diode (D1 from Figure 1) conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID( AVG) = IOUT ( VIN – VOUT ) / VIN The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to the typical peak switch current limit. If transient input voltages exceed 40V, use a Schottky diode with a reverse voltage rating of 45V or higher. If the maximum transient input voltage is under 40V, use a Schottky diode with a reverse voltage rating greater than the maximum input voltage. Table 3 lists several Schottky diodes and their manufacturers: Table 3. Schottky Diodes MANUFACTURER/ PART NUMBER VR (V) IAVE (A) VF at 1A (mV) 40 1 550 40 1 450 On Semiconductor MBRM140 MicroSemi UPS140 Diodes Inc. DFLS140L 40 1 550 1N5819HW 40 1 450 providing the boost supply to the BD pin. In this case the voltage drop of the other switch will increase and lower the efficiency. This could eventually cause the part to reach the thermal shutdown limit. One other important feature of the part that needs to be considered is that there is a parasitic diode in parallel with the power switch. In normal operation this is reverse biased but it could conduct if the load can be powered from an alternate source when the LT3509 has no input. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode ORed with one of the LT3509 regulated outputs. If the SW pin is at more than about 4V the VIN pin can attain sufficient voltage for LT3509 control circuitry to power-up to the quiescent bias level and up to 2mA could be drawn from the backup supply. This can be minimized if some discrete FETs or open-drain buffers are used to pull down the RUN/SS pins. Of course the gates need to be driven from the standby or battery backed supply. If there is the possibility of a short circuit at the input or just other parallel circuits connected to VIN it would be best to add a protection diode in series with VIN. This will also protect against a reversed input polarity. These concepts are illustrated in Figure 12. VIN D2 BD VIN CIN BOOST RUN/SS1 Short and Reverse Protection VOUT SW D1 Provided the inductors are chosen to not go deep into their saturation region at the maximum ILIMIT current the LT3509 will tolerate a short circuit on one or both outputs. The excess current in the inductor will be detected by the DA comparator and the frequency will reduced until the valley current is below the limit. This shouldn’t affect the other channel unless the channel that is shorted is also 18 CBOOST L1 LT3509 RUN/SS2 GND COUT DA SLEEP 3509 F12 Figure 12. Reverse Bias Protection 3509fd For more information www.linear.com/LT3509 LT3509 APPLICATIONS INFORMATION Hot Plugging Considerations The small size, reliability and low impedance of ceramic capacitors make them attractive for the input capacitor. Unfortunately they can be hazardous to semiconductor devices if combined with an inductive supply loop and a fast power transition such as through a mechanical switch or connector. The low loss ceramic capacitor combined with the just a small amount of wiring inductance forms an underdamped resonant tank circuit and the voltage at the VIN pin of the LT3509 can ring to twice the nominal input voltage. See Linear Technology Application Note 88 for more details. PCB Layout and Thermal Design The PCB layout is critical to both the electrical and thermal performance of the LT3509. Most important is the connection to the Exposed Pad which provides the main ground connection and also a thermal path for cooling the chip. This must be soldered to a topside copper plane which is also tied to backside and/or internal plane(s) with an array of thermal vias. • The loop from the regulated outputs through the output capacitor back to the ground plane. Excess impedance here will result in excessive ripple at the output. The area of the SW and BOOST nodes should as small as possible. Also the feedback components should be placed as close as possible to the FB pins so that the traces are short and shielded from the SW and BOOST nodes by the ground planes. Figure 13 shows a detail view of a practical board layout showing just the top layer. The complete board is somewhat larger at 7.5cm × 7.5cm. The device has been evaluated on this board in still air running at 700kHz switching frequency. One channel was set to 5V and the other to 3.3V and both channels were fully loaded to 700mA. The device temperature reached approximately 15°C above ambient for input voltages below 12V. At 24V input it was slightly higher at 17°C above ambient. To obtain the best electrical performance particular attention should be paid to keeping the following current paths short: • The loop from the VIN pin through the input capacitor back to the ground pad and plane. This sees high di/dt transitions as the power switches turn on and off. Excess impedance will degrade the minimum usable input voltage and could cause crosstalk between channels. • The loops from the switch pins to the catch diodes and back to the DA pins. The fast changing currents and voltage here combined with long PCB traces will cause ringing on the switch pin and may result in unwelcome EMI. Figure 13. Sample PCB Layout (Top Layer Only) 3509fd For more information www.linear.com/LT3509 19 LT3509 Typical Applications 1.8V and 3.3V Outputs, Synchronized to 300kHz to 600kHz VIN 4.5V TO 36V (TRANSIENT TO 60V) 2.2µF BD BOOST1 BOOST2 0.22µF 15µH 0.22µF 10µH VOUT 1.8V 0.7A VIN SW1 LT3509 UPS140 12.4k CLOCK 22µF 1.6V UPS140 DA1 DA2 FB1 FB2 31.6k RUN/SS1 RUN/SS2 10k SYNC 22nF VOUT 3.3V 0.7A SW2 RT 10k GND 22nF 22µF 178k 0.4V 3509 TA03 NOTE: RT CHOSEN FOR 264kHz Automotive Accessory Application 5V Logic Supply and 8V for LCD Display with Display Power Controlled by Logic VIN 9.4V TO 36V 2.2µF VOUT 5V 0.7A VIN BD BOOST1 BOOST2 0.22µF 10µH 0.22µF 6.8µH SW1 LT3509 DFLS140L 52.3k DFLS140L DA1 DA2 FB1 FB2 RUN/SS1 RUN/SS2 SYNC RT 10k 10µF DISPLAY POWER CONTROL 0V = OFF 3.3V = ON 20 22nF VOUT 8V 0.7A SW2 GND 40.2k 90.9k 10k 0.1µF 10k 10µF 3509 TA04 fSW = 1MHz 3509fd For more information www.linear.com/LT3509 LT3509 package description Please refer to http://www.linear.com/product/LT3509#packaging for the most recent package drawings. DE Package 14-Lead Plastic DFN (4mm × 3mm) (Reference LTC DWG # 05-08-1708 Rev B) 0.70 ±0.05 3.30 ±0.05 3.60 ±0.05 2.20 ±0.05 1.70 ±0.05 PACKAGE OUTLINE 0.25 ±0.05 0.50 BSC 3.00 REF RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 4.00 ±0.10 (2 SIDES) R = 0.05 TYP 3.00 ±0.10 (2 SIDES) R = 0.115 TYP 8 0.40 ±0.10 14 3.30 ±0.10 1.70 ±0.10 PIN 1 NOTCH R = 0.20 OR 0.35 × 45° CHAMFER PIN 1 TOP MARK (SEE NOTE 6) 0.200 REF 0.75 ±0.05 (DE14) DFN 0806 REV B 7 1 0.25 ±0.05 0.50 BSC 3.00 REF 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING PROPOSED TO BE MADE VARIATION OF VERSION (WGED-3) IN JEDEC PACKAGE OUTLINE MO-229 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3509fd For more information www.linear.com/LT3509 21 LT3509 package description Please refer to http://www.linear.com/product/LT3509#packaging for the most recent package drawings. MSE Package 16-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1667 Rev F) BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 ±0.102 (.112 ±.004) 5.10 (.201) MIN 2.845 ±0.102 (.112 ±.004) 0.889 ±0.127 (.035 ±.005) 8 1 1.651 ±0.102 (.065 ±.004) 1.651 ±0.102 3.20 – 3.45 (.065 ±.004) (.126 – .136) 0.305 ±0.038 (.0120 ±.0015) TYP 16 0.50 (.0197) BSC 4.039 ±0.102 (.159 ±.004) (NOTE 3) RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 0.35 REF 0.12 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY 9 NO MEASUREMENT PURPOSE 0.280 ±0.076 (.011 ±.003) REF 16151413121110 9 DETAIL “A” 0° – 6° TYP 3.00 ±0.102 (.118 ±.004) (NOTE 4) 4.90 ±0.152 (.193 ±.006) GAUGE PLANE 0.53 ±0.152 (.021 ±.006) DETAIL “A” 1.10 (.043) MAX 0.18 (.007) SEATING PLANE 1234567 8 0.17 – 0.27 (.007 – .011) TYP 0.50 NOTE: (.0197) 1. DIMENSIONS IN MILLIMETER/(INCH) BSC 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 22 0.86 (.034) REF 0.1016 ±0.0508 (.004 ±.002) MSOP (MSE16) 0213 REV F 3509fd For more information www.linear.com/LT3509 LT3509 Revision History REV DATE DESCRIPTION C 4/10 Changed Pin Name to RT D 01/16 (Revision history begins at Rev C) PAGE NUMBER 1, 2, 6, 7, 20, 21, 24 Revised Absolute Maximum Ratings 2 Updated Notes and Change/Add Values in Electrical Characteristics 3 Revised Values in Typical Performance Characteristics 5 Revised Values in Pin Functions 6 Revised Values in Startup and Shutdown Section 8 Revised Values in Shutdown and Soft-Start, Frequency Setting Sections, and Table 1 9 Clarified Sync Pin Function Description 6 Clarified External Synchronization Applications Information 10 3509fd Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. For more information www.linear.com/LT3509 23 LT3509 Typical Applications 2MHz, 5V and 3.3V Outputs VIN 6.5V TO 16V (TRANSIENT TO 60V) 2.2µF BD BOOST2 0.1µF 0.1µF 6.8µH VOUT 5V 0.7A VIN BOOST1 SW1 MBRM140 52.3k 4.7µH SW2 MBRM140 LT3509 DA1 DA2 FB1 FB2 VOUT 3.3V 0.7A 31.6k RUN/SS1 RUN/SS2 GND SYNC RT 10k 10µF 16.9k 22nF 22nF 10µF 10k fSW = 2MHz 3509 TA02 Related Parts PART NUMBER DESCRIPTION COMMENTS LT1766 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT = 1.20V, IQ = 2.5mA, ISD < 25µA, TSSOP16/E Package LT1936 36V, 1.4A (IOUT) , 500kHz High Efficiency Step-Down DC/DC Converter VIN: 36V to 36V, VOUT = 1.20V, IQ = 1.9mA, ISD < 1µA, MS8E Package LT1939 25V, 2A, 2.5MHz High Efficiency DC/DC Converter and LDO Controller VIN: 3.6V to 25V, VOUT = 0.8V, IQ = 2.5µA, ISD < 10µA, 3mm × 3mm DFN-10 VIN: 3.3V to 60V, VOUT = 1.20V, IQ = 100µA, ISD < 1µA, TSSOP16E LT1976/ 60V, 1.2A (IOUT), 200/500kHz, High Efficiency Step-Down LT1977 DC/DC Converter with Burst Mode® Operation Package LT3434/ 60V, 2.4A (IOUT), 200/500kHz, High Efficiency Step-Down VIN: 3.3V to 60V, VOUT = 1.20V, IQ = 100µA, ISD < 1µA, TSSOP16E LT3435 DC/DC Converter with Burst Mode Operation Package LT3437 60V, 400mA (IOUT), MicroPower Step-Down DC/DC Converter VIN: 3.3V to 60V, VOUT = 1.25V, IQ = 100µA, ISD < 1µA, 3mm × 3mm with Burst Mode Operation DFN-10, TSSOP-16E Package LT3480 36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High VIN: 3.6V to 38V, VOUT = 0.78V, IQ = 70µA, ISD < 1µA, 3mm × 3mm Efficiency Step-Down DC/DC Converter with Burst Mode Operation DFN-10, MSOP-10E Package LT3481 34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, High VIN: 3.6V to 34V, VOUT = 1.26V, IQ = 50µA, ISD < 1µA, 3mm × 3mm Efficiency Step-Down DC/DC Converter with Burst Mode Operation DFN-10, MSOP-10E Package LT3493 36V, 1.4A(IOUT), 750kHz High Efficiency Step-Down DC/DC VIN: 36V to 36V, VOUT = 0.8V, IQ = 1.9mA, ISD < 1µA, 2mm × 3mm Converter DFN-6 Package LT3500 36V, 40Vmax, 2A, 2.5MHz High Efficiency DC/DC Converter VIN: 3.6V to 36V, VOUT = 0.8V, IQ = 2.5mA, ISD < 10µA, 3mm × 3mm and LDO Controller DFN-10 LT3501 25V, Dual 3A (IOUT), 1.5MHz High Efficiency Step-Down VIN: 3.3V to 25V, VOUT = 0.8V, IQ = 3.7mA, ISD = 10µA, TSSOP-20E DC/DC Converter Package LT3505 36V with Transient Protection to 40V, 1.4A (IOUT), 3MHz, VIN: 3.6V to 34V, VOUT = 0.78V, IQ = 2mA, ISD < 2µA, 3mm × 3mm High Efficiency Step-Down DC/DC Converter DFN-8, MSOP-8E Package LT3506/ 25V, Dual 1.6A (IOUT), 575kHz,/1.1MHz High Efficiency VIN: 3.6V to 25V, VOUT = 0.8V, IQ = 3.8mA, ISD = 30µA, 5mm × 4mm LT3506A Step-Down DC/DC Converter DFN-16 TSSOP-16E Package LT3507 36V 2.5MHz, Triple (2.4A + 1.5A + 1.5A (IOUT)) with LDO VIN: 4V to 36V, VOUT = 0.8V, IQ = 7mA, ISD = 1µA, 5mm × 7mm Controller High Efficiency Step-Down DC/DC Converter QFN-38 LT3508 36V with Transient Protection to 40V, Dual 1.4A (IOUT), 3MHz, VIN: 3.7V to 37V, VOUT = 0.8V, IQ = 4.6mA, ISD = 1µA, 4mm × 4mm High Efficiency Step-Down DC/DC Converter QFN-24, TSSOP-16E Package LT3510 25V, Dual 2A (IOUT), 1.5MHz High Efficiency Step-Down VIN: 3.3V to 25V, VOUT = 0.8V, IQ = 3.7mA, ISD = 10µA, TSSOP-20E DC/DC Converter Package LT3684 34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, VIN: 3.6V to 34V, VOUT = 1.26V, IQ = 850µA, ISD < 1µA, 3mm × 3mm High Efficiency Step-Down DC/DC Converter DFN-10, MSOP-10E Package LT3685 36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, VIN: 3.6V to 38V, VOUT = 0.78V, IQ = 70µA, ISD < 1µA, 3mm × 3mm High Efficiency Step-Down DC/DC Converter DFN-10, MSOP-10E Package 24 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 For more information www.linear.com/LT3509 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LT3509 3509fd LT 0116 REV D • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2007