NCP1027ATXGEVB A 5.0 V/2.0 A Standby Power Supply for Intel) Compliant ATX Applications Evaluation Board User's Manual http://onsemi.com EVAL BOARD USER’S MANUAL ATX power supply units (PSU) require a standby section who keeps alive some particular areas of the motherboard. Among the live sections are the USB ports, the Ethernet controller and so on. The Intel ATX Power Supply Design Guide 2.01 (rev. June 04) describes the standby voltage rail needed for this purpose. This is the +5.0 VSB section (3.3.3): • Output voltage: 5.0 V, "5% • Nominal load current: 2.0 A • 500 ms pulses current (USB wake−up event): 2.5 A • Input power less than 1.0 W at 230 Vac for an output power of 500 mW • Power on time: 2.0 s maximum • Short circuit protection with auto−recovery • • On top of these requirements, most of PSU designers use the standby power supply as an auxiliary rail to power the main controller. This is an auxiliary 12−13 V rail or, if an UC384X is implemented, this rail can go up to 20 V. When put in standby, a small switch shuts down the controller by interrupting this rail. To help designers quickly fulfilling the above needs, the NCP1027 has been introduced. This DIP 8 package hosts a high performance controller together with a low RDS(on) 700 V BVdss MOSFET. On top of the standby needs, we have packed other interesting goodies in this circuit. They are summarized below: • Brown−out detection: the controller will not allow operation in low mains conditions. You can adjust the level at which the circuit starts or stops operation. • Ramp compensation: designing in Continuous Conduction Mode helps to reduce conduction losses. However, at low input voltage (85 Vac), the duty−cycle might exceed 50%, and the risk exists to enter a subharmonic mode. A simple resistor to ground injects the right compensation level. • Over Power Protection: a resistive network to the bulk reduces the peak current capability and accordingly harnesses the maximum power at high line. As this is © Semiconductor Components Industries, LLC, 2012 October, 2012 − Rev. 1 done independently from the auxiliary VCC, the design gains in simplicity and execution speed. Latch−off input: some PC manufacturers require a complete latch−off in presence of an external event, e.g. overtemperature. The controller offers this possibility via a dedicated input. Frequency dithering: the switching frequency (here 65 kHz) is modulated during operation. This naturally spreads the harmonic content and reduces the peak value when analyzing the signature. Figure 1. Evaluation Board Photo Design Description A full CCM operation gave us an adequate performance in this particular case, with good full load efficiency results as we will see. The part switches at 65 kHz which represents a good trade−off between switching losses and EMI control. A brown−out circuit was implemented, turning the SMPS on around 80 Vac and turning it off at 60 Vac. Different values can easily be selected by altering the dedicated resistive network. Please note that this network impedance has a direct influence on the standby power. To limit the amount of current the supply can deliver at high line, it is necessary to limit the propagation delay effects. The NCP1027 hosts an exclusive circuitry used to reduce the maximum peak current as the line increases. We will see that, once implemented around the auxiliary diode, it does not affect the standby power and nicely harnesses the maximum power. The electrical schematic of the board appears on Figure 2. 1 Publication Order Number: EVBUM2144/D NCP1027ATXGEVB TR1: Lp = 3.4 mH Np:Ns = 1:0.06 Np:Naux = 1:0.15 Bulk R10 2.8 Meg C7 100 mF 16 V R11 2.2 Meg R7 1k C5 10 nF R6 150 k 0.5 W 13 V + L1 2.2 mH D1 MBRD835L + C1 1500 mF R5 47 + C2 1500 mF R3 100 U3 D4 1N4937 + C10 47 mF 400 V 1 R14 560 kW U1 NCP1027 2 + C13 1 mF 8 7 R4 1k C6 10 nF R9 27 k + 4 + U2 TL431 R2 10 k 5 C9 100 pF R8 78 k R1 10 k C4 0.1 mF 3 C8 10 mF 10 V 0 TR1 D4 1N4937 Aux. C3 220 mF 5 V−2 A R13 47 kW U3b C11 2.2 nF Type = Y1 Figure 2. The Evaluation Board Electrical Schematic without the EMI Filter for Simpler Representation. Let us start the review by the transformer description. where: Lp, the primary inductance, 3.4 mH N, the turn ratio, 0.06 Fsw, the switching frequency, 65 kHz Vin, the input voltage Vout, the output voltage Calculations lead to the following mode transition load values: Transformer The design of the transformer section represents the most difficult part as standby power in no−load and 0.5 W output must be respected. Various iterations have lead us to adopt the following characteristics: Lp = 3.4 mH Np:Ns_power = 1:0.06 Np:Ns_aux = 1:0.152 The auxiliary winding in this case delivers 12.5 V but it can be set to any other value, depending on the main controller VCC. The turn ratio limits the reflected value on the drain to less than 100 V, without risks of biasing the MOSFET body diode at the lowest input voltage. This transformer is available from Coilcraft under a reference detailed in the Bill Of Material (BOM). As we have seen, our design operates in CCM at full load but obviously enters the Discontinuous Conduction Mode (DCM) at light loads. The transition point can be evaluated using the following formula: Rc + 2LpN2Fsw ǒNVinNV)inVoutǓ2 Lowest line, Vin = 120 Vdc: Rc = 4.56 W or Iout = 1.0 A Highest line, Vin = 370 Vdc: Rc = 2.4 W or Iout = 2.0 A Please note that low line is actually 85 Vac but the ATX PSU including a large bulk capacitor, we can neglect the ripple given the low output power of this converter, hence a 120 V value. At the highest line, the design operates at the boundary between CCM and DCM. The duty−cycle at full load and lowest input line can be evaluated via the flyback static transfer function: D+ Vout NVin ) Vout (eq. 2) The highest duty−cycle is found to be 41%. However, this number is likely to increase a little, given the presence of the leakage inductance. (eq. 1) http://onsemi.com 2 NCP1027ATXGEVB Ramp Compensation If the limit is set to 750 mA, then we have: Being in CCM with a duty−cycle close to 50% (or above 50% in transient conditions), we need ramp compensation. Several ways exist to evaluate the amount of ramp compensation. The quickest one evaluates the off slope of the secondary inductance and injects around 50% of this slope through a compensating ramp (Sa ) into the controller. With the NCP1027, the ramp compensation level is set via a simple resistor connected from pin 2 to the ground, as shown in Figure 2. We can calculate the off slope, the one actually needed to evaluate Sa , by reflecting the output voltage over the primary inductance. The slope is projected over a complete switching period. 6 V ) Vf Soff + out Tsw + NLp 0.06 15 m + 441 mAń15 ms 3.4 m 0.375 + 165 mVń15 ms ROPPH + Cbulk (eq. 4) 1 2 U1 NCP1027 8 4 lb1 7 lb3 3 Following the data sheet indications, we evaluate the resistor value to be: 5 lb2 ROPPL (eq. 6) In case no ramp compensation is required, pin 2 must be tied to VCC , the adjacent pin. As experiments were carried at lower input voltages (Cbulk is small on the demo board), it was decided to slightly increase the amount of ramp compensation by reducing Rramp to 78 kW. Figure 3. A Possible Option to Reduce the Peak Excursion consists of Connecting a Resistive Divider to the Bulk Capacitor. By injecting a current Ib3 proportional to Vbulk , the maximum peak current limits reduces. Analytically obtaining the right value for Ib3 might represent a complicated exercise as many parameters play a role. To the opposite, a simple experimental method can be setup to obtain the value of the needed current: 1. Configure your working power supply as suggested by Figure 4 or 5. These figures are purposely simplified for ease of understanding of the added circuitry. Figure 4 represents the safest way to run the measurement as the dc power supply injects current via an optoisolator without sharing the converter’s ground. An amp−meter is inserted to read the current Ib3. Leave the dc bias to zero for now. 2. Power the converter and set the input voltage to the highest of your specification. Let us assume it is 375 Vdc. Over Power Protection (OPP) Power supply controllers sensing the primary current to check whether it goes over a certain limit often face propagation delay problems. That is to say, despite the current limit detection by a dedicated comparator, the information takes a certain amount of time to propagate through the logic circuits and eventually reset the latch. During this time, the primary current keeps increasing by a rate given by the primary inductance and the input voltage. Hence, we can quickly see the effects of a 100 ns delay at high line or low line: V Ifinal + Ipeak_max ) in tproper Lp (eq. 9) (eq. 3) SȀoff + 83 mVń15 ms + 5.53 kVńs (eq. 5) 2 Rramp + 7562 + 91 k 0.083 Ifinal + 750 m ) 374 100 n + 761 mA 3.4 m Bulk If we chose 50% of this down slope, then the final compensation ramp will present a slope of: Sa + (eq. 8) The difference looks small but it often leads to a significant different in output current capabilities, especially with larger propagation delays. Hence, the need to act in high line conditions via a dedicated circuitry. The NCP1027 hosts a special section used to reduce the maximum peak current limit as the mains increases. The system works by connecting a resistive network to the bulk capacitor, as Figure 3 depicts: The NCP1027 features a current−mode architecture using a SENSEFETt device. That is to say, the controller does not directly sense the current via a resistor but through a Kelvin cell. For this particular circuit, the cell ratio can be modeled as an equivalent sense resistor of 350 mW. This current slope will thus become a voltage slope having a value of: SȀoff + 0.441 Ifinal + 750 m ) 100 100 n + 753 mA 3.4 m (eq. 7) http://onsemi.com 3 NCP1027ATXGEVB 3. Increase the output current to the point where the system should shut down per specification. Let’s say 3.2 A for this design. 4. Start to increase the variable power supply dc voltage until the amp−meter deviates. Increase carefully because you deal with hundred of mA only. 5. At a certain time, if you continue to increase the current in pin 7, the converter enters in protection mode. The current flowing in pin 7 and the voltage on it prior to the shutdown, correspond to the variables you look for. In this example, we measured a Vf voltage of 2.45 V and a current of 31 mA. R4 10 k 1 U1 NCP1027 2 R2 10 k + 8 7 S 3 + Aux 5 4 + Variable Power Supply + IB3 R3 10 k Figure 4. An Isolated Way to safely Inject Current into Pin 7 at High Line. 1 2 U1 NCP1027 8 R2 100 k 7 S 3 Aux + + 4 5 + IB3 R3 10 k + Variable Power Supply Figure 5. A Non−isolated Way to Inject Current into Pin 7 at High Line. http://onsemi.com 4 NCP1027ATXGEVB The preliminary curve showing the relationship between the injected current and the maximum peak current setpoint appears in Figure 6. We can see that 31 mA corresponds to a 20% reduction in the maximum peak current capability. The Vf parameter specifies the voltage at which pin 7 starts to pump in current. It is around 2.5 V as we can see it. (IPEAK−IPEAKOPP)/IPEAK (%) 90.0% 80.0% 70.0% 60.0% 50.0% 40.0% 30.0% 20.0% 10.0% 0.0% 0 50 100 150 200 LOPP CURRENT (mA) Figure 6. Maximum Peak Current Setpoint Reduction vs. Pin 7 Injected Current To compute the resistor values, we need to define the range within which the OPP reduction should activate. As we do not want any OPP action at low input voltages, we will select resistors to start reducing at Vin = 200 Vdc, with a clamp at Vin = 375 Vdc. Using the following equations and our collected data leads us to the final values: VbulkH = 375 Vdc VbulkL = 200 Vdc = 31 mA IOPP Vf = 2.45 V VbulkH−VbulkL V + 70 kW IOPP(VbulkL−Vf) f (eq. 10) V −V ROPPH + ROPPL bulkL f + 5.6 MW Vf (eq. 11) ROPPL + Unfortunately, despite the good behavior of the network, its permanent presence on the bulk rail will affect the consumption in standby. When you chase every hidden milli−watt, it can become a nasty problem. To avoid this trouble, a simple solution around the auxiliary winding can be worked out. This is presented in Figure 7. During the on time, where the power switch is turned on, the input voltage appears across the primary transformer. Given the auxiliary diode configuration, N . Vin + Vout also appears on the cathode, N being the primary to secondary turn ratio. As this voltage moves up and down with the bulk level, we can perfectly use it for our OPP purposes. Due to its pulsating low voltage nature, power consumption will be the smallest. VCC C7 100 mF 16 V R7 1k + Aux. D4 1N4937 1 ROPPH 560 kW U1 NCP1027 2 8 7 3 C8 10 mF 10 V + 4 C9 100 pF R8 78 k 5 ROPPL 47 k U3b Figure 7. The Auxiliary Diode lends itself very well to a cheap and energy efficient OPP Implementation. http://onsemi.com 5 NCP1027ATXGEVB Hooking an oscilloscope probe on D4 cathode gives us the bulk image evolution between its minimum and maximum values. We can then run the calculation again to obtain ROPPH and ROPPL : VbulkH = 55 Vdc VbulkL = 37 Vdc = 31 mA IOPP Vf = 2.45 V V −V Rupper + Rlower bulk1 BO VBO Rlower + VBO Rupper Rlower PBO + OUTPUT CURRENT (A) Auxiliary Winding Without OPP The auxiliary winding is set to deliver 13 V nominal when the converter is fully loaded. To avoid any VCC drops during transient loading, e.g. sudden load removal, the main VCC capacitor C7 has been increased to 100 mF, leading to a stable VCC in no−load conditions. The resistor R7 sets the Overvoltage Protection (OVP) level by adjusting the injected current in pin 1 internal shunt in case of problems. With 1.0 kW, the OVP level is set to VOVP + 1.0 k · 7.0 m ) 8.7 + 15.7 V typical on the auxiliary winding. Different values can be obtained by changing the resistor values or the ratio between the power and auxiliary windings. 4 With OPP 2 1 260 280 300 320 3302 + 27 mW. 4.02 Meg On the board, we purposely modified these values to further improve the standby power. 6 0 240 = 4.0 MW = 22 kW Total power dissipation at nominal line is then Figure 8 shows the results before and after OPP implementation. The output current stays below 3.5 A in all cases which is well within specs. 3 (eq. 13) Suppose we want to start at Vin = 110 Vdc (77 Vac) and stop operation at Vin = 70 Vdc (50 Vac), then applying Equations 12 and 13 leads to the following values: ROPPH = 580 kW → 560 kW after tweak ROPPL = 41 kW → 47 kW after tweak 5 Vbulk1−Vbulk2 IBO (Vbulk1−VBO) (eq. 12) 340 360 INPUT VOLTAGE (Vdc) Clamping Section Figure 8. Over Power Protection at work keeps the output current below 3.5 A. The converter uses an RCD clamp to safely limit the voltage excursion on the MOSFET drain. Failure to keep Vds (t) below 700 V will permanently damage the circuit. In application where the input line can be subject to strong high−voltage parasitic pulses, we recommend the usage of a Transient Voltage Suppressor (TVS), that will hard clamp all potentially lethal spikes on the drain. Figure 9 portrays the way to wire the TVS. Finally, the TVS offers superior performance in standby power compared to the RCD clamp. Simply because the RCD clamp always activates since the capacitor discharges via the resistor R as soon the leakage inductance is reset. Unfortunately, even in standby, this mechanism generates light losses. If you want to further save 50 mW, a TVS is of good usage. A 200 V TVS has shown to be a good solution here. You can use the 1.5KE200A from ON Semiconductor for instance. Brown−out Brown−out (BO) detection offers a means to protect the converter in presence of low input voltages by stopping switching operation until the mains comes back to a normal value. The circuit works by observing a fraction of the bulk level via a resistive divider routed to pin 2. When the level on pin 2 lies below 0.6 V, the controller does not allow switching operation, but the high−voltage current source maintains the VCC on pin 1. Then, as soon as pin 2 voltage crosses 0.6 V, the VCC is ready to power the chip and switching can start. As this occurs, pin 2 injects around 12 mA (IBO ) in the resistive bridge to create a hysteresis. The designer must thus select the turn−on and turn−off voltages to further apply the following equations: http://onsemi.com 6 NCP1027ATXGEVB Secondary Side Vbulk D2 1.5KE220A The secondary side uses a Schottky diode featuring an 8.0 A capability and a breakdown voltage of 35 V. This is enough since high line conditions imply a Peak Inverse Voltage (PIV) of: 3 PIV + NVin ) Vout + 0.06 5 D1 1N4937 1 370 ) 5 + 27.2 V The forward voltage Vf goes down to 0.41 V @ Tj = 125° which, neglecting ohmic losses, induces conduction losses of: 4 Pdiode + IRMS2Rd ) IavgVf [ 2 5 DV + D1 MBRD835L DI fcCout L1 2.2 mH +5 V + + C2 1500 mF C3 220 mF 0 R3 100 R4 1k U3 + (eq. 15) (eq. 16) Calculations gave us a value of 2.4 mF, made of two low−impedance 10 V/1200 mF capacitors. A TL431 ensures a stable regulation at 5.0 V via a type−2 amplifier. R3 and the NCP1027 internal pullup resistor set the midband gain, whereas C4 sets the zero position. The small capacitor C9 filters out residual noise and adds a high frequency pole, acting together with the optocoupler one. The step load response is clean, without any ringing at both high and low line conditions. The circuit can be slightly modified to add some more soft−start, in case designers would fear output overshoots. Figure 11 shows how to connect a 1.0 mF/10 V capacitor to soften the start−up sequence: Figure 10. Always Check the Voltage on the Drain at the highest Line Condition (265 Vac). + 0.41 + 0.8 W The output capacitor Cout is selected to a) pass the adequate RMS current b) limit the undershoot DV when the output is banged by a current step DI. The undershoot depth of a closed−loop converter having a bandwidth fc can be evaluated via the following formula: Figure 9. A TVS must be used if the Input Voltage can be subject to High Voltage Spikes. C1 1500 mF (eq. 14) C4 0.1 mF X3 TL431 Css 1 mF R1 10 k R2 10 k Figure 11. A 1.0 mF Capacitor connected on the TL431 strengthens the Startup Sequence. http://onsemi.com 7 NCP1027ATXGEVB The basic description being done, let us have a look at the board performance. Standby measurements were captured with a Yokogawa WT210 on a board after it warmed up for 15 minutes. Please make sure the WT210 volt−mater is placed before the current shunt, otherwise its input impedance will degrade the low−power measurements by 30 mW at high line. Static Measurements Vin = 230 Vac Tambient = 25°C Pout = 0 Pout = 0.5 W Pout = 10 W Pin = 88 mW Pin = 779 mW Pin = 12.3 W Vaux = 10.14 V Vaux = 11.64 V Vaux = 12.14 V h = 64.2% h = 81.3% ³ Please note that replacing R6 by a 200 V TVS as mentioned in the text, reduces the input power at 0.5 W @ 230 Vac down to 715 mW and 73 mW at no load. Pout = 0 Pout = 0 Pin = 98 mW at Vin = 265 Vac with RCD Pin = 85 mW at Vin = 265 Vac with TVS Tcase NCP1027 = 50°C at Pout = 10 W Short−circuit temperature: NCP1027 Tcase = 42°C, Tdiode = 52°C Maximum output current: 3.0 A @ 265 Vac Vin = 85 Vac Tambient = 25°C Pout = 0 Pout = 0.5 W Pout = 10 W Pin = 54 mW Pin = 704 mW Pin = 12.6 W Vaux = 10.15 V Vaux = 11.6 V Vaux = 12.6 V h = 71% h = 79.5% Tcase NCP1027 = 60°C at Pout = 10 W Short−circuit temperature: Tcase NCP1027 = 44°C, Tdiode = 52°C Maximum output current: 3.1 A Dynamic Measurements Some critical waveforms have been captured on the demonstration board and are reproduced below: Figure 12. Short−Circuit Protection, Drain−Source Waveform In Figure 12, we can see the power supply operating in a so−called hiccup mode, trying to re−start as soon as the internal timer has elapsed. The resulting duty−burst stays below 8%, keeping all component temperatures at a moderate level. This is an auto−recovery type of protection, implying a re−start when the fault is removed (Vin = 230 Vac). http://onsemi.com 8 NCP1027ATXGEVB 10 W 0W 0.5 W Figure 13. Startup Sequence at Three Loading Conditions Figure 13 shows a typical startup sequence, captured at different output levels (0, 0.5 W and 10 W) for Vin = 85 Vac. Changing the input voltage does not modify the shape of the waveform. This waveform has been captured with the 1.0 mF wired as suggested by Figure 11. Figure 14. Load step where Vout is banged from 0.1 A to 2.5 A with a 1.0 A/ms slew−rate (Vin = 85 Vac). http://onsemi.com 9 NCP1027ATXGEVB Figure 15. Load step where Vout is banged from 0.1 A to 2.5 A with a 1.0 A/ms slew−rate (Vin = 230 Vac). Figures 14 and 15 represent the step load response from the standby mode to the wake−up mode, featuring a variation from 100 mA up to 2.5 A. The spike you can see on the waveforms comes from the current discontinuity indured by the LC filter inductor L1 . Figure 16. Peak−to−peak ripple, Pout = 0.5 W, Cable length = 30 cm, Vin = 85 Vac. http://onsemi.com 10 NCP1027ATXGEVB Figure 17. Peak−to−peak ripple, Pout = 0.5 W, Cable length = 30 cm, Vin = 230 Vac. Figures 16 and 17 display the ripple amplitude when the board enters skip cycle, mainly at a 0.5 W output power. The measurement has been carried at two line levels and using 30 cm long cables to mimic a real PSU cabling connector. High Line Low Line Figure 18. EMI signature, low line and high line in quasi−peak, Pout = 10 W. Figure 18 shows the advantage of EMI jittering, offering a clean signature at both line levels. As the sweep is successfully made in Quasi−Peak (QP) it implies immediate compliance in average mode. http://onsemi.com 11 NCP1027ATXGEVB High Line Low Line Figure 19. Peak Output Current in Short−Circuit at Both Line Conditions: 6.3 A 54 ms 676 ms Figure 20. Output Current Waveform in Short−Circuit Some PC application specs require the output average and RMS currents in short−circuit to stay within given limits. With this design, we have obtained the following results at high−line: S Ipeak = 6.4 A S Iout,av = 6.4 x 54 / 676 = 511 mA S Iout,rms = 6.4 http://onsemi.com 12 54 + 1.8 A Ǹ676 NCP1027ATXGEVB PCB Views The PCB routing includes large copper areas around the integrated circuit to maximize its power dissipation. The diode is placed on the copper side and due to its low Vf, it leads to good thermal performance. Figure 21. The PCB Copper Area of the Demonstration Board Figure 22. The SMD Positions on the Copper Side http://onsemi.com 13 NCP1027ATXGEVB Figure 23. Component Side View BILL OF MATERIAL Table 1. BILL OF MATERIAL Designator QTY Description U1 1 U2 U3 Manufacturer Part Number Substitution Allowed RoHS Compliant ON Semiconductor NCP1027P065G No Yes ON Semiconductor TL431CLPRPG No Yes Vishay SFH615A No Yes Value Tolerance Footprint Manufacturer High Voltage Switcher NA NA PDIP 8 1 Adjustable Shunt Regulator 2.5 V - 36 V, 1 mA - 100 mA 2% TO-92 1 Optocoupler 70 V, 60 mA NA DIP-4 Semiconductors B1 1 Single-Phase Bridge Rectifier 800 V, 1 A NA DFM Vishay General DF08ME3 Yes Yes C1, C2 2 Electrolytic Capacitor 1200 mF, 16 V 20% Radial Panasonic EEU-FC1C122 Yes Yes C10 1 Electrolytic Capacitor 47 mF, 400 V 20% Radial Panasonic ECA2GHG470 Yes Yes C11 1 Ceramic ac Capacitor, Class Y1 2.2 nF, 500 Vac, 50 Hz 20% Leaded spacing = 12.5 mm Vishay Draloric WKP222MCPR No Yes C12 1 Capacitor, Class X2 220 nF, 275 Vac 20% Leaded spacing = 15 mm Evox Rifa PHE840MX6220M No Yes C13 1 Electrolytic Capacitor 1 mF, 50 V 20% Radial Panasonic EEUFC1H1R0 Yes Yes C3 1 Electrolytic Capacitor 220 mF, 16 V 20% Radial Panasonic EEUFC1C221 Yes Yes C4 1 Multilayer Ceramic Capacitor, X7R 100 nF, 50 V 10% 1206 Epcos B37872K5104K060 Yes Yes C5 1 Metallized Polyester Film Capacitor 10 nF, 630 V 10% Lead spacing = 10 mm Vishay Roederstein MKT1822310635 Yes Yes C6 1 Multilayer Ceramic Capacitor, X7R 10 nF, 50 V 10% 1206 Epcos B37872K5103K060 Yes Yes C7 1 Minature Aluminum Electrolytic Capacitor 10 F, 63 V 20% Radial Panasonic ECA1JM100 Yes Yes http://onsemi.com 14 NCP1027ATXGEVB Table 1. BILL OF MATERIAL Manufacturer Manufacturer Part Number Substitution Allowed RoHS Compliant Panasonic EEUFC1A101 Yes Yes 1206 Epcos B37872K5102K060 Yes Yes NA DPAK-4 ON Semiconductor MBRD835LG No Yes 600 V, 1 A NA Axial ON Semiconductor 1N4937G No Yes NA NA NA Multicomp JR-201S Yes Yes NA NA Weidmuller 1760510000 Yes Yes 20% 7.8 mm x 9.5 mm Wurth Elektonic 744772022 Yes Yes NA 22.5 mm x 21.5 mm Schaffner RN114-0.8/02 Yes Yes 10 kW, 1/4 W 5% Axial Panasonic ERD-S2TJ103V Yes Yes Metal Film Resistor 2.8 MW, 1/4 W 5% Axial Vishay Dale CCF072M80JKE36 Yes Yes 1 Axial Lead Carbon Film Resistor 2.2 MW, 1/4 W 5% Axial Panasonic ERD-S2TJ225V Yes Yes R13 1 Axial Lead Carbon Film Resistor 47 kW, 1/4 W 5% Axial Panasonic ERD-S2TJ473V Yes Yes R14 1 Axial Lead Carbon Film Resistor 560 kW, 1/4 W 5% Axial Panasonic ERD-S2TJ564V Yes Yes R2 1 SMD Resistor 10 kW, 1/4 W 5% 1206 Vishay Dale Yes Yes R3 1 SMD Resistor 100 W, 1/4 W 5% 1206 Vishay Dale Yes Yes R4 1 Axial Lead Carbon Film Resistor 680 W, 1/4 W 5% Axial Panasonic ERD-S2TJ681V Yes Yes R5 1 Axial Lead Carbon Film Resistor 47 W, 1/4 W 5% Axial Panasonic ERD-S2TJ470V Yes Yes R6 1 Power Metal Film Resistor 150 kW, 1 W 5% Axial Vishay BC Components PR01000201503JAC00 Yes Yes R7 1 Axial Lead Carbon Film Resistor 1 kW, 1/4 W 5% Axial Panasonic ERD-S2TJ102V Yes Yes R8 1 SMD Resistor 78 kW, 1/4 W 5% 1206 Vishay Dale CRCW120678KJNEA Yes Yes R9 1 SMD Resistor 27 kW, 1/4 W 5% 1206 Vishay Dale CRCW120627KJNEA Yes Yes T1 1 Transformer 3.4 mH NA 1.1” x 1.1” x 0.95” Coilcraft DA2077-AL No Yes Designator QTY Description C8 1 Electrolytic Capacitor C9 1 D1 Value Tolerance Footprint 100 mF, 10 V 20% Radial Multilayer Ceramic Capacitor, X7R 1nF, 50 V 10% 1 Schottky Barrier Rectifier 35 V, 8 A D2, D4 2 Fast-Recovery Rectifier H1 1 PCB Connector H2 1 PCB Connector NA L1 1 Inductor 2.2 mH, 6 A L2 1 Common Mode Inductor 27 mH, 0.8 A R1 1 Axial Lead Carbon Film Resistor R10 1 R11 http://onsemi.com 15 CRCW120610K0JNEA CRCW1206100RJNEA NCP1027ATXGEVB TEST PROCEDURE Figure 24. Test Setup Required Equipment • • • • • Test Procedure 1. Connect the test setup as shown in Figure 24. 2. Apply an input voltage, 85 Vac < VIN < 265 Vac, 50 Hz or 60 Hz. 3. Connect electronic load to output connector. 4. Test the conditions given in Table 2. ac Power Supply, 85 Vac − 265 Vac, 1 A Electronic Load, 5 V up to 3 A Yokogawa Power Meter WT210 NCP1027ATX Evaluation Board Multimeter Table 2. DESIRED RESULTS Measurements Conditions Results Comments 1 Output Voltage VIN = 100 Vac, no load VOUT = 5 V ± 5% 2 Input consumption at high line and no load VIN = 230 Vac, no load PIN = 0.09 W ± 15%, VOUT = 5 V 3 Input consumption at high line line and low load VIN = 230 Vac, Iload = 0.1 A PIN = 0.83 W ± 10% VOUT = 5 V POUT = 0.5 W 4 Output voltage at high line and full load VIN = 230 Vac, Iload = 2.5 A VOUT = 5 V ± 5% 5 Output voltage at low line and full load VIN = 85 Vac, Iload = 2.5 A VOUT = 5 V ± 5% 6 Brownout detection VIN ± until VOUT = 0 V, IOUT = 2 A VIN = 56 Vac ± 15% Output voltage must collapse under 56 Vac 7 Maximum output at low line VIN = 85 Vac, IOUT ° POUTmax = 14 W ± 15% Output voltage must collapse above 14 W 8 Maximum output at high line VIN = 230 Vac, IOUT ° POUTmax = 18 W ± 15% Output voltage must collapse above 18 W http://onsemi.com 16 No load measurment can fluctuate, run WT210 in average mode NCP1027ATXGEVB SENSEFET is a registered trademark of Semiconductor Components Industries, LLC (SCILLC). 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