LTC3609 32V, 6A Monolithic Synchronous Step-Down DC/DC Converter Description Features n n n n n n n n n n n n n n n n n 6A Output Current Wide VIN Range = 4V to 32V (36V Maximum) Internal N-Channel MOSFETs True Current Mode Control Optimized for High Step-Down Ratios tON(MIN) ≤ 100ns Extremely Fast Transient Response Stable with Ceramic COUT ±1% 0.6V Voltage Reference Power Good Output Voltage Monitor Adjustable On-Time/Switching Frequency Adjustable Current Limit Programmable Soft-Start Output Overvoltage Protection Optional Short-Circuit Shutdown Timer Low Shutdown IQ: 15µA Available in a 7mm × 8mm 52-Pin QFN Package Applications Point of Load Regulation Distributed Power Systems n n L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 6100678, 6580258, 5847554, 6304066. The LTC®3609 is a high efficiency, monolithic synchronous step-down DC/DC converter that can deliver up to 6A output current from a 4V to 32V (36V maximum) input supply. It uses a valley current control architecture to deliver very low duty cycle operation at high frequency with excellent transient response. The operating frequency is selected by an external resistor and is compensated for variations in VIN and VOUT. The LTC3609 can be configured for discontinuous or forced continuous operation at light load. Forced continuous operation reduces noise and RF interference while discontinuous mode provides high efficiency by reducing switching losses at light loads. Fault protection is provided by internal foldback current limiting, an output overvoltage comparator and an optional short-circuit shutdown timer. Soft-start capability for supply sequencing is accomplished using an external timing capacitor. The regulator current limit is user programmable. A power good output voltage monitor indicates when the output is in regulation. The LTC3609 is available in a compact 7mm × 8mm QFN package. Typical Application Efficiency and Power Loss vs Load Current High Efficiency Step-Down Converter VON ION RUN/SS VIN 100pF 187k 100 10µF x3 1.2µH SW 1000pF 0.22µF 15.8k ITH BOOST SGND INTVCC 30.1k FCB VRNG PGOOD EXTVCC 100µF x2 10000 EFFICIENCY 80 1000 70 60 50 40 VFB 9.53k 100 POWER LOSS 30 10 20 10 VOUT = 2.5V EXTVCC = 5V 0 0.01 0.1 1 LOAD CURRENT (A) 4.7µF PGND 3609 TA01a VOUT 2.5V 6A 90 POWER LOSS (mW) LTC3609 VIN 4V TO 32V EFFICIENCY (%) 0.1µF VOUT VIN = 12V VIN = 25V 1 10 3609 TA01b 3609fb LTC3609 Absolute Maximum Ratings Pin Configuration (Note 1) 41 SW 42 SW 43 SW 44 SW 45 SW 46 SW 47 SW 48 PVIN 49 PVIN 50 PVIN 51 PVIN TOP VIEW 52 PVIN Input Supply Voltage (SVIN, PVIN, ION)........ 36V to –0.3V Boosted Topside Driver Supply Voltage (BOOST)................................................. 42V to –0.3V SW Voltage............................................. 36V to –0.3V INTVCC, EXTVCC, (BOOST – SW), RUN/SS, PGOOD Voltages....................................... 7V to –0.3V FCB, VON, VRNG Voltages............. INTVCC + 0.3V to –0.3V ITH, VFB Voltages........................................ 2.7V to –0.3V Operating Junction Temperature Range (Notes 2, 4)......................................... –40°C to 125°C Storage Temperature Range.................... –55°C to 125°C PVIN 1 40 PGND PVIN 2 39 PGND PVIN 3 38 PGND 53 PVIN PVIN 4 PVIN 5 37 PGND 55 SW 36 PGND PVIN 6 35 PGND PVIN 7 34 PGND SW 8 33 SW NC 9 32 INTVCC SGND 10 31 INTVCC BOOST 11 30 SVIN 54 SGND RUN/SS 12 29 EXTVCC VON 13 28 NC SGND 26 NC 25 NC 24 VFB 23 ION 22 NC 21 SGND 20 FCB 19 ITH 18 VRNG 17 SGND 15 27 SGND PGOOD 16 SGND 14 WKG PACKAGE 52-LEAD (7mm s 8mm) QFN MULTIPAD TJMAX = 125°C, θJA = 29°C/W order information LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3609EWKG#PBF LTC3609EWKG#TRPBF LTC3609WKG 52-Lead (7mm × 8mm) Plastic QFN –40°C to 125°C LTC3609IWKG#PBF LTC3609IWKG#TRPBF LTC3609WKG 52-Lead (7mm × 8mm) Plastic QFN –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3609fb LTC3609 Electrical Characteristics The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Main Control Loop SVIN Operating Input Voltage Range IQ Input DC Supply Current Normal Shutdown Supply Current VFB Feedback Reference Voltage 4 ITH = 1.2V (Note 3) –40°C to 85°C –40°C to 125°C l 0.594 0.590 32 V 900 15 2000 30 µA µA 0.600 0.600 0.606 0.610 V V ∆VFB(LINEREG) Feedback Voltage Line Regulation VIN = 4V to 30V, ITH = 1.2V (Note 3) 0.002 ∆VFB(LOADREG) Feedback Voltage Load Regulation ITH = 0.5V to 1.9V (Note 3) –0.05 –0.3 % IFB Feedback Input Current VFB = 0.6V –5 ±50 nA gm(EA) Error Amplifier Transconductance ITH = 1.2V (Note 3) mS VFCB Forced Continuous Threshold IFCB Forced Continuous Pin Current VFCB = 0.6V tON On-Time ION = 60µA, VON = 1.5V ION = 60µA, VON = 0V tON(MIN) Minimum On-Time tOFF(MIN) %/V l 1.4 1.7 2 l 0.54 0.6 0.66 V –1 –2 µA 220 280 110 340 ns ns ION = 180µA, VON = 0V 60 100 ns Minimum Off-Time ION = 30µA, VON = 1.5V 320 500 ns IVALLEY(MAX) Maximum Valley Current VRNG = 0V, VFB = 0.56V, FCB = 0V VRNG = 1.2V, VFB = 0.56V, FCB = 0V IVALLEY(MIN) Maximum Reverse Valley Current VRNG = 0V, VFB = 0.64V, FCB = 0V VRNG = 1.2V, VFB = 0.64V, FCB = 0V ∆VFB(OV) Output Overvoltage Fault Threshold VRUN/SS(ON) RUN Pin Start Threshold VRUN/SS(LE) RUN Pin Latchoff Enable Threshold RUN/SS Pin Rising 4 4.5 V VRUN/SS(LT) RUN Pin Latchoff Threshold RUN/SS Pin Falling 3.5 4.2 V IRUN/SS(C) Soft-Start Charge Current VRUN/SS = 0V –0.5 –1.2 –3 µA IRUN/SS(D) Soft-Start Discharge Current VRUN/SS = 4.5V, VFB = 0V 0.8 1.8 3 µA VIN(UVLO) Undervoltage Lockout VIN Falling l 3.4 3.9 V VIN(UVLOR) Undervoltage Lockout Release VIN Rising l 3.5 4 V RDS(ON) Top Switch On-Resistance Bottom Switch On-Resistance 18 13 27 22 mΩ mΩ l l l 4 6 9 14 A A 4 7 A A 7 10 13 % 0.8 1.5 2 V 3609fb LTC3609 Electrical Characteristics The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS 4.7 5 5.5 V –0.1 ±2 % 300 mV Internal VCC Regulator VINTVCC Internal VCC Voltage 6V < VIN < 30V, VEXTVCC = 4V ∆VLDO(LOADREG) Internal VCC Load Regulation ICC = 0mA to 20mA, VEXTVCC = 4V VEXTVCC EXTVCC Switchover Voltage ICC = 20mA, VEXTVCC Rising ∆VEXTVCC EXTVCC Switch Drop Voltage ICC = 20mA, VEXTVCC = 5V ∆VEXTVCC(HYS) EXTVCC Switchover Hysteresis l l 4.5 4.7 150 V 500 mV PGOOD Output ∆VFBH PGOOD Upper Threshold VFB Rising 7 10 13 % ∆VFBL PGOOD Lower Threshold VFB Falling –7 –10 –13 % ∆VFB(HYS) PGOOD Hysteresis VFB Returning 1 2.5 % VPGL PGOOD Low Voltage IPGOOD = 5mA 0.15 0.4 V Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD as follows: TJ = TA + (PD • 29°C/W) (θJA is simulated per JESD51-7 high effective thermal conductivity test board). θJC = 1°C/W (θJC is simulated when heat sink is applied at the bottom of the package). Note 3: The LTC3609 is tested in a feedback loop that adjusts VFB to achieve a specified error amplifier output voltage (ITH). The specification at 85°C is not tested in production. This specification is assured by design, characterization, and correlation to testing at 125°C. Note 4: The LTC3609 is tested under pulsed load conditions such that TJ ≈ TA. The LTC3609E is guaranteed to meet specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3609I is guaranteed over the full –40°C to 125°C operating junction temperature range. Note that the maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. Typical Performance Characteristics Transient Response Transient Response (Discontinuous Mode) VOUT 200mV/DIV Start-Up VOUT 200mV/DIV RUN/SS 2V/DIV IL 5A/DIV IL 5A/DIV VOUT 1V/DIV ILOAD 5A/DIV ILOAD 5A/DIV 20µs/DIV LOAD STEP 0A TO 5A VIN = 25V VOUT = 2.5V FCB = 0V FIGURE 6 CIRCUIT 3609 G01 IL 5A/DIV 20µs/DIV LOAD STEP 1A TO 6A VIN = 25V VOUT = 2.5V FCB = INTVCC FIGURE 6 CIRCUIT 3609 G02 40ms/DIV VIN = 12V VOUT = 2.5V RLOAD = 0.5Ω FIGURE 6 CIRCUIT 3609 G03 3609fb LTC3609 Typical Performance Characteristics Efficiency vs Load Current Efficiency vs Input Voltage 100 100 40 30 ILOAD = 6A 90 ILOAD = 1A 85 20 10 VIN = 12V FREQUENCY = 550kHz 0 0.01 0.1 1 LOAD CURRENT (A) 80 10 FREQUENCY (kHz) 50 EFFICIENCY (%) EFFICIENCY (%) 60 600 95 VOUT = 5V VOUT = 3.3V VOUT = 2.5V VOUT = 1.8V VOUT = 1.5V VOUT = 1.2V VOUT = 1V 70 650 FCB = 5V FIGURE 6 CIRCUIT 90 80 Frequency vs Load Current 0.80 5 8 11 14 17 20 23 26 INPUT VOLTAGE (V) 32 FCB = 0V FIGURE 6 CIRCUIT 5 300 200 100 2.5 FIGURE 6 CIRCUIT 17 20 23 26 29 32 3609 G06 FIGURE 6 CIRCUIT 2.0 0.20 0 1.5 CONTINUOUS MODE 1.0 0.5 DISCONTINUOUS MODE –0.60 4 6 LOAD CURRENT (A) 14 ITH Voltage vs Load Current –0.40 2 11 3609 G05 –0.20 DISCONTINUOUS MODE 0 8 INPUT VOLTAGE (V) ITH VOLTAGE (V) ∆VOUT (%) FREQUENCY (kHz) 29 400 0.40 400 –0.80 8 0 3609 G07 Load Current vs ITH Voltage at Different VRNG 10000 15 2 4 6 LOAD CURRENT (A) 0 8 0 2 3609 G08 On-Time vs ION Current 4 6 LOAD CURRENT (A) 8 3609 G09 On-Time vs VON Voltage 1000 VVON = 0V VRNG = 1.2V ION = 30µA 800 10 VRNG = 1V 0 1000 ON-TIME (ns) VRNG = 0.7V 5 ON-TIME (ns) LOAD CURRENT (A) ILOAD = 1A 0.60 CONTINUOUS MODE 500 100 600 400 200 –5 –10 500 Load Regulation 700 0 ILOAD = 6A 550 450 3609 G04 600 Frequency vs Input Voltage 0 0.5 1.0 1.5 ITH VOLTAGE (V) 2.0 2.5 3609 G10 10 1 10 ION CURRENT (µA) 100 3609 G11 0 0 1 2 VON VOLTAGE (V) 3 3609 G12 3609fb LTC3609 Typical Performance Characteristics On-Time vs Temperature MAXIMUM VALLEY CURRENT LIMIT (A) 250 ON-TIME (ns) 15 IION = 30µA VVON = 0V 200 150 100 50 0 –50 –25 25 75 0 50 TEMPERATURE (°C) 100 Maximum Valley Current Limit vs RUN/SS Voltage 15 FIGURE 6 CIRCUIT MAXIMUM VALLEY CURRENT LIMIT (A) 300 Maximum Valley Current Limit vs VRNG Voltage 12 9 6 3 0 125 0.5 0.6 0.7 0.8 0.9 1.0 VRNG VOLTAGE (V) 1.1 FIGURE 6 CIRCUIT 12 9 6 3 0 1.65 1.90 2.15 2.40 2.65 2.90 3.15 3.40 RUN/SS VOLTAGE (V) 1.2 3609 G14 3609 G15 3609 G13 Maximum Valley Current Limit vs Temperature 9 6 3 –25 0 25 50 75 TEMPERATURE (°C) 100 6 4 2 4 3609 G16 12 20 28 INPUT VOLTAGE (V) 75 0 25 50 TEMPERATURE (°C) 100 125 1.6 1.4 3609 G19 1.0 –50 0.2 0.3 VFB (V) 0.4 0.5 0.6 3609 G18 40 EXTVCC OPEN 1200 1.2 –25 0.1 0 1400 INPUT CURRENT (µA) gm (mS) 0.59 2 30 1000 25 800 50 0 75 25 TEMPERATURE (°C) 100 125 3609 G20 20 SHUTDOWN 600 15 400 0 10 EXTVCC = 5V 200 –25 35 SHUTDOWN CURRENT (µA) 0.60 4 Input and Shutdown Currents vs Input Voltage 1.8 0.61 6 3609 G17 2.0 0.62 8 0 36 Error Amplifier gm vs Temperature Feedback Reference Voltage vs Temperature FEEDBACK REFERENCE VOLTAGE (V) 8 0 125 MAXIMUM VALLEY CURRENT LIMIT (A) 12 0.58 –50 10 10 MAXIMUM VALLEY CURRENT (A) MAXIMUM VALLEY CURRENT LIMIT (A) 15 0 –50 Maximum Valley Current Limit in Foldback Maximum Valley Current vs Input Voltage 5 0 5 10 20 15 INPUT VOLTAGE (V) 25 30 0 3609 G21 3609fb LTC3609 Typical Performance Characteristics INTVCC Load Regulation EXTVCC Switch Resistance vs Temperature EXTVCC SWITCH RESISTANCE (Ω) 0.20 ∆INTVCC (%) 0.10 0 –0.10 –0.20 –0.30 –0.40 0 40 10 20 30 INTVCC LOAD CURRENT (mA) 10 25 8 20 IEXTVCC (mA) 0.30 6 4 2 VIN = 24V 15 10 5 0 –50 50 IEXTVCC vs Frequency –25 0 50 75 25 TEMPERATURE (°C) 100 0 400 125 500 600 700 800 FREQUENCY (kHz) 3609 G22 900 1000 3609 G28 3609 G23 RUN/SS Pin Current vs Temperature FCB Pin Current vs Temperature 3 RUN/SS PIN CURRENT (µA) –0.25 –0.50 –0.75 –1.00 –1.25 –1.50 –50 –25 25 75 0 50 TEMPERATURE (°C) 100 125 5.0 2 RUN/SS PIN CURRENT (µA) 0 FCB PIN CURRENT (µA) RUN/SS Pin Current vs Temperature PULL-DOWN CURRENT 1 0 –1 –2 –50 PULL-UP CURRENT –25 0 50 75 25 TEMPERATURE (°C) 100 4.5 LATCHOFF ENABLE 4.0 LATCHOFF THRESHOLD 3.5 3.0 –50 125 –25 3609 G25 75 0 25 50 TEMPERATURE (°C) 100 125 3609 G26 3609 G24 Load Step 4.0 Efficiency vs Load Current 100 f = 500kHz 95 IL 5A/DIV 3.5 3.0 90 EFFICIENCY (%) UNDERVOLTAGE LOCKOUT THRESHOLD (V) Undervoltage Lockout Threshold vs Temperature VOUT 200mV/DIV 2.5 2.0 –50 –25 75 0 25 50 TEMPERATURE (°C) 100 125 3609 G27 40µs/DIV LOAD STEP 1A TO 4A VIN = 24V VOUT = 12V FCB = 0V FIGURE 8 CIRCUIT 3609 G29 85 DCM 80 75 70 CCM 65 60 55 50 0.01 VIN = 24V FREQUENCY = 500kHz 0.1 1 LOAD CURRENT (A) FIGURE 8 CIRCUIT 10 3609 G30 3609fb LTC3609 Pin Functions PVIN (Pins 1, 2, 3, 4, 5, 6, 7, 48, 49, 50, 51, 52, 53): Main Input Supply. Decouple this pin to power PGND with the input capacitance, CIN. SW (Pins 8, 33, 41, 42, 43, 44, 45, 46, 47, 55): Switch Node Connection to the Inductor. The (–) terminal of the bootstrap capacitor, CB, also connects here. This pin swings from a diode voltage drop below ground up to VIN. NC (Pins 9, 21, 24, 25, 28): No Connection. SGND (Pins 10, 14, 15, 20, 26, 27, 54): Signal Ground. All small-signal components and compensation components should connect to this ground, which in turn connects to PGND at one point. BOOST (Pin 11): Boosted Floating Driver Supply. The (+) terminal of the bootstrap capacitor, CB, connects here. This pin swings from a diode voltage drop below INTVCC up to VIN + INTVCC. RUN/SS (Pin 12): Run Control and Soft-Start Input. A capacitor to ground at this pin sets the ramp time to full output current (approximately 3s/µF) and the time delay for overcurrent latchoff (see Applications Information). Forcing this pin below 0.8V shuts down the device. VON (Pin 13): On-Time Voltage Input. Voltage trip point for the on-time comparator. Tying this pin to the output voltage or an external resistive divider from the output makes the on-time proportional to VOUT. The comparator input defaults to 0.7V when the pin is grounded and defaults to 2.4V when the pin is tied to INTVCC. Tie this pin to INTVCC in high VOUT applications to use a lower RON value. PGOOD (Pin 16): Power Good Output. Open-drain logic output that is pulled to ground when the output voltage is not within ± 10% of the regulation point. ITH (Pin 18): Current Control Threshold and Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. The voltage ranges from 0V to 2.4V with 0.8V corresponding to zero sense voltage (zero current). FCB (Pin 19): Forced Continuous Input. Tie this pin to ground to force continuous synchronous operation at low load, to INTVCC to enable discontinuous mode operation at low load or to a resistive divider from a secondary output when using a secondary winding. ION (Pin 22): On-Time Current Input. Tie a resistor from VIN to this pin to set the one-shot timer current and thereby set the switching frequency. VFB (Pin 23): Error Amplifier Feedback Input. This pin connects the error amplifier input to an external resistive divider from VOUT. EXTVCC (Pin 29): External VCC Input. When EXTVCC exceeds 4.7V, an internal switch connects this pin to INTVCC and shuts down the internal regulator so that controller and gate drive power is drawn from EXTVCC. Do not exceed 7V at this pin and ensure that EXTVCC < VIN. SVIN (Pin 30): Supply Pin for Internal PWM Controller. INTVCC (Pins 31, 32): Internal 5V Regulator Output. The driver and control circuits are powered from this voltage. Decouple this pin to power ground with a minimum of 4.7µF low ESR tantalum or ceramic capacitor. PGND (Pins 34, 35, 36, 37, 38, 39, 40): Power Ground. Connect this pin closely to the (–) terminal of CVCC and the (–) terminal of CIN. VRNG (Pin 17): Current Limit Range Input. The voltage at this pin adjusts maximum valley current and can be set from 0.7V to 1.2V by a resistive divider from INTVCC. It defaults to 0.7V if the VRNG pin is tied to ground which results in a typical 9A current limit. 3609fb LTC3609 Functional Diagram RON VON ION 13 22 FCB EXTVCC 19 29 SVIN 30 4.7V 0.7V 2.4V + 1µA PVIN – 1, 2, 3, 4, 5, 6, 7, 48, 49, 50, 51, 52, 53 0.6V REF 0.6V CIN 5V REG INTVCC + – 31, 32 F 11 VVON tON = (10pF) IION R S Q FCNT SW + ICMP – L1 DB VOUT 8, 33, 41, 42, 43, 44, 45, 46, 47, 55 SWITCH LOGIC IREV – SHDN 1.4V COUT OV M2 + CVCC 17 PGND s 34, 35, 36, 37, 38, 39, 40 (0.5 TO 2) 0.7V 16 PGOOD 1 240k + 1V Q2 Q4 – Q6 ITHB R2 0.54V UV 23 Q3 Q1 VFB + R1 OV + – – 0.8V – SS + SGND 10, 14, 15, 20, 26, 27, 54 0.66V RUN SHDN 1.2µA EA s3.3 NC 9, 21, 24, 25, 28 + – – + VRNG CB M1 ON 20k + BOOST 6V 0.6V 18 ITH 0.4V 12 3609 FD RUN/SS CSS 3609fb LTC3609 Operation Main Control Loop The LTC3609 is a high efficiency monolithic synchronous, step-down DC/DC converter utilizing a constant on-time, current mode architecture. It operates from an input voltage range of 4V to 32V/36V maximum and provides a regulated output voltage at up to 6A of output current. The internal synchronous power switch increases efficiency and eliminates the need for an external Schottky diode. In normal operation, the top MOSFET is turned on for a fixed interval determined by a one-shot timer OST. When the top MOSFET is turned off, the bottom MOSFET is turned on until the current comparator ICMP trips, restarting the one-shot timer and initiating the next cycle. Inductor current is determined by sensing the voltage between the PGND and SW pins using the bottom MOSFET on-resistance. The voltage on the ITH pin sets the comparator threshold corresponding to inductor valley current. The error amplifier, EA, adjusts this voltage by comparing the feedback signal VFB from the output voltage with an internal 0.6V reference. If the load current increases, it causes a drop in the feedback voltage relative to the reference. The ITH voltage then rises until the average inductor current again matches the load current. At light load, the inductor current can drop to zero and become negative. This is detected by current reversal comparator IREV which then shuts off M2 (see Functional Diagram), resulting in discontinuous operation. Both switches will remain off with the output capacitor supplying the load current until the ITH voltage rises above the zero current level (0.8V) to initiate another cycle. Discontinuous mode operation is disabled by comparator F when the FCB pin is brought below 0.6V, forcing continuous synchronous operation. The operating frequency is determined implicitly by the top MOSFET on-time and the duty cycle required to maintain regulation. The one-shot timer generates an on-time that is proportional to the ideal duty cycle, thus holding frequency approximately constant with changes in VIN. The nominal frequency can be adjusted with an external resistor, RON. Overvoltage and undervoltage comparators OV and UV pull the PGOOD output low if the output feedback voltage exits a ±10% window around the regulation point. Furthermore, in an overvoltage condition, M1 is turned off and M2 is turned on and held on until the overvoltage condition clears. Foldback current limiting is provided if the output is shorted to ground. As VFB drops, the buffered current threshold voltage ITHB is pulled down by clamp Q3 to a 1V level set by Q4 and Q6. This reduces the inductor valley current level to one sixth of its maximum value as VFB approaches 0V. Pulling the RUN/SS pin low forces the controller into its shutdown state, turning off both M1 and M2. Releasing the pin allows an internal 1.2µA current source to charge up an external soft-start capacitor, CSS. When this voltage reaches 1.5V, the controller turns on and begins switching, but with the ITH voltage clamped at approximately 0.6V below the RUN/SS voltage. As CSS continues to charge, the soft-start current limit is removed. INTVCC/EXTVCC Power Power for the top and bottom MOSFET drivers and most of the internal controller circuitry is derived from the INTVCC pin. The top MOSFET driver is powered from a floating bootstrap capacitor, CB. This capacitor is recharged from INTVCC through an external Schottky diode, DB, when the top MOSFET is turned off. When the EXTVCC pin is grounded, an internal 5V low dropout regulator supplies the INTVCC power from VIN. If EXTVCC rises above 4.7V, the internal regulator is turned off, and an internal switch connects EXTVCC to INTVCC. This allows a high efficiency source connected to EXTVCC, such as an external 5V supply or a secondary output from the converter, to provide the INTVCC power. Voltages up to 7V can be applied to EXTVCC for additional gate drive. If the input voltage is low and INTVCC drops below 3.5V, undervoltage lockout circuitry prevents the power switches from turning on. 3609fb 10 LTC3609 Applications Information The basic LTC3609 application circuit is shown on the front page of this data sheet. External component selection is primarily determined by the maximum load current. The LTC3609 uses the on-resistance of the synchronous power MOSFET for determining the inductor current. The desired amount of ripple current and operating frequency also determines the inductor value. Finally, CIN is selected for its ability to handle the large RMS current into the converter and COUT is chosen with low enough ESR to meet the output voltage ripple and transient specification. VON and PGOOD The LTC3609 has an open-drain PGOOD output that indicates when the output voltage is within ±10% of the regulation point. The LTC3609 also has a VON pin that allows the on-time to be adjusted. Tying the VON pin high results in lower values for RON which is useful in high VOUT applications. The VON pin also provides a means to adjust the on-time to maintain constant frequency operation in applications where VOUT changes and to correct minor frequency shifts with changes in load current. VRNG Pin and ILIMIT Adjust The VRNG pin is used to adjust the maximum inductor valley current, which in turn determines the maximum average output current that the LTC3609 can deliver. The maximum output current is given by: IOUT(MAX) = IVALLEY(MAX) + 1/2 ∆IL The IVALLEY(MAX) is shown in the figure “Maximum Valley Current Limit vs VRNG Voltage” in the Typical Performance Characteristics. An external resistor divider from INTVCC can be used to set the voltage on the VRNG pin from 0.7V to 1.2V, or it can be simply tied to ground force a default value equivalent to 0.7V. When setting current limit, ensure that the junction temperature does not exceed the maximum rating of 125°C. Do not float the VRNG pin. Operating Frequency The choice of operating frequency is a tradeoff between efficiency and component size. Low frequency operation improves efficiency by reducing MOSFET switching losses but requires larger inductance and/or capacitance in order to maintain low output ripple voltage. The operating frequency of LTC3609 applications is determined implicitly by the one-shot timer that controls the on-time, tON, of the top MOSFET switch. The on-time is set by the current into the ION pin and the voltage at the VON pin according to: tON = VVON (10pF ) IION Tying a resistor RON from VIN to the ION pin yields an on-time inversely proportional to VIN. The current out of the ION pin is: IION = VIN RON For a step-down converter, this results in approximately constant frequency operation as the input supply varies: f= VOUT [ HZ ] VVON RON(10pF ) To hold frequency constant during output voltage changes, tie the VON pin to VOUT or to a resistive divider from VOUT when VOUT > 2.4V. The VON pin has internal clamps that limit its input to the one-shot timer. If the pin is tied below 0.7V, the input to the one-shot is clamped at 0.7V. Similarly, if the pin is tied above 2.4V, the input is clamped at 2.4V. In high VOUT applications, tying VON to INTVCC so that the comparator input is 2.4V results in a lower value for RON. Figures 1a and 1b show how RON relates to switching frequency for several common output voltages. 3609fb 11 LTC3609 Applications Information Because the voltage at the ION pin is about 0.7V, the current into this pin is not exactly inversely proportional to VIN, especially in applications with lower input voltages. To correct for this error, an additional resistor RON2 connected from the ION pin to the 5V INTVCC supply will further stabilize the frequency. RON2 = 5V RON 0.7 V Changes in the load current magnitude will also cause frequency shift. Parasitic resistance in the MOSFET switches and inductor reduce the effective voltage across the inductance, resulting in increased duty cycle as the load current increases. By lengthening the on-time slightly as current increases, constant frequency operation can be maintained. This is accomplished with a resistive divider from the ITH pin to the VON pin and VOUT. The values required will depend on the parasitic resistances in the specific application. A good starting point is to feed about 25% of the voltage change at the ITH pin to the VON pin as shown in Figure 2a. Place capacitance on the VON pin to filter out the ITH variations at the switching frequency. The resistor load on ITH reduces the DC gain of the error amp and degrades load regulation, which can be avoided by using the PNP emitter follower of Figure 2b. SWITCHING FREQUENCY (kHz) 1000 VOUT = 3.3V VOUT = 1.5V 100 100 VOUT = 2.5V 1000 RON (kΩ) 10000 3609 F01a Figure 1a. Switching Frequency vs RON (VON = 0V) SWITCHING FREQUENCY (kHz) 1000 VOUT = 12V VOUT = 5V VOUT = 3.3V 100 100 1000 RON (kΩ) 10000 3609 F01b Figure 1b. Switching Frequency vs RON (VON = INTVCC) 3609fb 12 LTC3609 Applications Information 2.0 CVON 0.01µF RVON2 100k RC VON LTC3609 ITH CC (2a) VOUT INTVCC RVON1 3k 10k CVON 0.01µF RVON2 10k RC Q1 2N5087 CC VON LTC3609 ITH 1.0 0.5 0 0.25 0.50 0.75 DUTY CYCLE (VOUT/VIN) 1.0 3609 F03 Figure 3. Maximum Switching Frequency vs Duty Cycle Minimum Off-time and Dropout Operation The minimum off-time, tOFF(MIN), is the smallest amount of time that the LTC3609 is capable of turning on the bottom MOSFET, tripping the current comparator and turning the MOSFET back off. This time is generally about 250ns. The minimum off-time limit imposes a maximum duty cycle of tON/(tON + tOFF(MIN)). If the maximum duty cycle is reached, due to a dropping input voltage for example, then the output will drop out of regulation. The minimum input voltage to avoid dropout is: tON + tOFF(MIN) tON A plot of maximum duty cycle vs frequency is shown in Figure 3. Setting the Output Voltage The LTC3609 develops a 0.6V reference voltage between the feedback pin, VFB, and the signal ground as shown in Figure 6. The output voltage is set by a resistive divider according to the following formula: R2 VOUT = 0.6V 1+ R1 DROPOUT REGION 3609 F02 Figure 2. Correcting Frequency Shift with Load Current Changes 1.5 0 (2b) VIN(MIN) = VOUT SWITCHING FREQUENCY (MHz) VOUT RVON1 30k To improve the frequency response, a feed-forward capacitor, C1, may also be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. Inductor Selection Given the desired input and output voltages, the inductor value and operating frequency determine the ripple current: V V ΔIL = OUT 1− OUT f L VIN Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors and output voltage ripple. Highest efficiency operation is obtained at low frequency with small ripple current. However, achieving this requires a large inductor. There is a tradeoff between component size, efficiency and operating frequency. A reasonable starting point is to choose a ripple current that is about 40% of IOUT(MAX). The largest ripple current occurs at the highest VIN. To guarantee that ripple current does not exceed a specified maximum, the inductance should be chosen according to: V VOUT OUT L = 1− V f ΔI IN(MAX) L(MAX) 3609fb 13 LTC3609 Applications Information Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores. A variety of inductors designed for high current, low voltage applications are available from manufacturers such as Sumida, Panasonic, Coiltronics, Coilcraft and Toko. CIN and COUT Selection The input capacitance, CIN, is required to filter the square wave current at the drain of the top MOSFET. Use a low ESR capacitor sized to handle the maximum RMS current. IRMS ≅ IOUT(MAX) VOUT VIN VIN –1 VOUT This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT(MAX)/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to derate the capacitor. The selection of COUT is primarily determined by the ESR required to minimize voltage ripple and load step transients. The output ripple ∆VOUT is approximately bounded by: 1 ΔVOUT ≤ ΔIL ESR+ 8fCOUT Since ∆IL increases with input voltage, the output ripple is highest at maximum input voltage. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the necessary RMS current rating. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic and ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important to only use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have significantly higher ESR, but can be used in cost-sensitive applications providing that consideration is given to ripple current ratings and long-term reliability. Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient and audible piezoelectric effects. The high Q of ceramic capacitors with trace inductance can also lead to significant ringing. When used as input capacitors, care must be taken to ensure that ringing from inrush currents and switching does not pose an overvoltage hazard to the power switches and controller. To dampen input voltage transients, add a small 5µF to 50µF aluminum electrolytic capacitor with an ESR in the range of 0.5Ω to 2Ω. High performance through-hole capacitors may also be used, but an additional ceramic capacitor in parallel is recommended to reduce the effect of their lead inductance. Top MOSFET Driver Supply (CB, DB) An external bootstrap capacitor, CB, connected to the BOOST pin supplies the gate drive voltage for the topside MOSFET. This capacitor is charged through diode DB from INTVCC when the switch node is low. When the top MOSFET turns on, the switch node rises to VIN and the BOOST pin rises to approximately VIN + INTVCC. The boost capacitor needs to store about 100 times the gate charge required by the top MOSFET. In most applications an 0.1µF to 0.47µF, X5R or X7R dielectric capacitor is adequate. Discontinuous Mode Operation and FCB Pin The FCB pin determines whether the bottom MOSFET remains on when current reverses in the inductor. Tying this pin above its 0.6V threshold enables discontinuous operation where the bottom MOSFET turns off when inductor current reverses. The load current at which current reverses and discontinuous operation begins depends on the amplitude of the inductor ripple current and will vary 3609fb 14 LTC3609 Applications Information Fault Conditions: Current Limit and Foldback with changes in VIN. Tying the FCB pin below the 0.6V threshold forces continuous synchronous operation, allowing current to reverse at light loads and maintaining high frequency operation. The LTC3609 has a current mode controller which inherently limits the cycle-by-cycle inductor current not only in steady-state operation but also in transient. To further limit current in the event of a short circuit to ground, the LTC3609 includes foldback current limiting. If the output falls by more than 25%, then the maximum sense voltage is progressively lowered to about one sixth of its full value. In addition to providing a logic input to force continuous operation, the FCB pin provides a means to maintain a flyback winding output when the primary is operating in discontinuous mode. The secondary output VOUT2 is normally set as shown in Figure 4 by the turns ratio N of the transformer. However, if the controller goes into discontinuous mode and halts switching due to a light primary load current, then VOUT2 will droop. An external resistor divider from VOUT2 to the FCB pin sets a minimum voltage VOUT2(MIN) below which continuous operation is forced until VOUT2 has risen above its minimum: An internal P-channel low dropout regulator produces the 5V supply that powers the drivers and internal circuitry within the LTC3609. The INTVCC pin can supply up to 50mA RMS and must be bypassed to ground with a minimum of 4.7µF tantalum or ceramic capacitor. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers. R4 VOUT2(MIN) = 0.6V 1+ R3 SW GND IN4148 NC SGND SVIN EXTVCC INTVCC SW PGND PGND PGND PGND INTVCC NC LTC3609 SW SGND FCB PVIN ITH PVIN VRNG PVIN PGOOD PVIN SGND 26 25 R4 24 23 22 21 20 OPTIONAL EXTVCC CONNECTION 5V < VOUT2 < 7V 19 18 17 R3 16 15 SGND PVIN 1 2 3 4 5 6 7 8 9 10 11 12 13 14 VON 52 SW RUN/SS 51 ION BOOST 50 SW SGND 49 VFB NC 48 NC SW SW CIN + VIN SW PVIN 47 NC PVIN 46 SW PVIN 45 SGND PVIN 44 PGND PGND 43 SW PVIN + 42 PVIN COUT 41 T1 1:N PVIN VOUT1 • CSEC 1µF + PGND 40 39 38 37 36 35 34 33 32 31 30 29 28 27 VOUT2 • INTVCC Regulator and EXTVCC Connection 3609 F04 SW SGND Figure 4. Secondary Output Loop and EXTVCC Connection 3609fb 15 LTC3609 Applications Information The EXTVCC pin can be used to provide MOSFET gate drive and control power from the output or another external source during normal operation. Whenever the EXTVCC pin is above 4.7V the internal 5V regulator is shut off and an internal 50mA P-channel switch connects the EXTVCC pin to INTVCC. INTVCC power is supplied from EXTVCC until this pin drops below 4.5V. Do not apply more than 7V to the EXTVCC pin and ensure that EXTVCC ≤ VIN. The following list summarizes the possible connections for EXTVCC: 1. EXTVCC grounded. INTVCC is always powered from the internal 5V regulator. 2. EXTVCC connected to an external supply. A high efficiency supply compatible with the MOSFET gate drive requirements (typically 5V) can improve overall efficiency. 3. EXTVCC connected to an output derived boost network. The low voltage output can be boosted using a charge pump or flyback winding to greater than 4.7V. The system will start-up using the internal linear regulator until the boosted output supply is available. Soft-Start and Latchoff with the RUN/SS Pin The RUN/SS pin provides a means to shut down the LTC3609 as well as a timer for soft-start and overcurrent latchoff. Pulling the RUN/SS pin below 0.8V puts the LTC3609 into a low quiescent current shutdown (IQ < 30µA). Releasing the pin allows an internal 1.2µA current source to charge up the external timing capacitor, CSS. If RUN/SS has been pulled all the way to ground, there is a delay before starting of about: tDELAY = ( ) 1.5V CSS = 1.3s/µF CSS 1.2µA When the voltage on RUN/SS reaches 1.5V, the LTC3609 begins operating with a clamp on ITH of approximately 0.9V. As the RUN/SS voltage rises to 3V, the clamp on ITH is raised until its full 2.4V range is available. This takes an additional 1.3s/µF, during which the load current is folded back until the output reaches 75% of its final value. After the controller has been started and given adequate time to charge up the output capacitor, CSS is used as a short-circuit timer. After the RUN/SS pin charges above 4V, if the output voltage falls below 75% of its regulated value, then a short-circuit fault is assumed. A 1.8µA current then begins discharging CSS. If the fault condition persists until the RUN/SS pin drops to 3.5V, then the controller turns off both power MOSFETs, shutting down the converter permanently. The RUN/SS pin must be actively pulled down to ground in order to restart operation. The overcurrent protection timer requires that the soft‑start timing capacitor, CSS, be made large enough to guarantee that the output is in regulation by the time CSS has reached the 4V threshold. In general, this will depend upon the size of the output capacitance, output voltage and load current characteristic. A minimum soft-start capacitor can be estimated from: CSS > COUT VOUT RSENSE (10 –4 [F/V s]) Generally 0.1µF is more than sufficient. Overcurrent latchoff operation is not always needed or desired. Load current is already limited during a short circuit by the current foldback circuitry and latchoff operation can prove annoying during troubleshooting. The feature can be overridden by adding a pull-up current greater than 5µA to the RUN/SS pin. The additional current prevents the discharge of CSS during a fault and also shortens the soft-start period. Using a resistor to VIN as shown in Figure 5a is simple, but slightly increases shutdown current. Connecting a resistor to INTVCC as shown in Figure 5b eliminates the additional shutdown current, but requires a diode to isolate CSS. Any pull-up network must be able to pull RUN/SS above the 4.2V maximum threshold of the latchoff circuit and overcome the 4µA maximum discharge current. 3609fb 16 LTC3609 Applications Information INTVCC RSS* VIN 3.3V OR 5V D1 RUN/SS RSS* CSS D2* RUN/SS 2N7002 CSS 3609 F05 *OPTIONAL TO OVERRIDE OVERCURRENT LATCHOFF (5a) (5b) Figure 5. RUN/SS Pin Interfacing with Latchoff Defeated Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Although all dissipative elements in the circuit produce losses, four main sources account for most of the losses in LTC3609 circuits: I2R losses. These arise from the resistance of the 1. DC internal resistance of the MOSFETs, inductor and PC board traces and cause the efficiency to drop at high output currents. In continuous mode the average output current flows through L, but is chopped between the top and bottom MOSFETs. The DC I2R loss for one MOSFET can simply be determined by [RDS(ON) + RL] • IO. 2. Transition loss. This loss arises from the brief amount of time the top MOSFET spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V and can be estimated from: Transition Loss ≅ (1.7A–1) VIN2 IOUT CRSS f 3. INTVCC current. This is the sum of the MOSFET driver and control currents. This loss can be reduced by supplying INTVCC current through the EXTVCC pin from a high efficiency source, such as an output derived boost network or alternate supply if available. 4. CIN loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator. It must have a very low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries. Other losses, including COUT ESR loss, Schottky diode D1 conduction loss during dead time and inductor core loss generally account for less than 2% additional loss. When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency. If you make a change and the input current decreases, then the efficiency has increased. If there is no change in input current, then there is no change in efficiency. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ∆ILOAD (ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The ITH pin external components shown in Figure 6 will provide adequate compensation for most applications. For a detailed explanation of switching control loop theory see Application Note 76. 3609fb 17 LTC3609 Applications Information Design Example Next, set up VRNG voltage and check the ILIMIT. Tying VRNG to GND will set the typical current limit to 9A, and tying VRNG to 1.2V will result in a typical current around 14A. CIN is chosen for an RMS current rating of about 5A at 85°C. The ceramic output capacitors are chosen for an ESR of 0.002Ω to minimize output voltage changes due to inductor ripple current and load steps. The ripple voltage is: As a design example, take a supply with the following specifications: VIN = 5V to 32V (12V nominal), VOUT = 2.5V ± 5%, IOUT(MAX) = 6A, f = 550kHz. First, calculate the timing resistor with VON = VOUT: 2.5V RON = = 187k 2.4V )( 550kHz )(10pF ) ( and choose the inductor for about 40% ripple current at the maximum VIN: 2.5V 2.5V L= =1.8µH 1− (550kHz ) (0.4) (6A ) 32V ∆VOUT(RIPPLE)= ∆IL(MAX) (ESR) = (2.4A) (0.002Ω) = 4.8mV and a 0A to 6A load step will only cause an output change of: ∆VOUT(STEP) = ∆ILOAD (ESR) = (6A) (0.002Ω) = 12mV Selecting a standard value of 1.5µH results in a maximum ripple current of: 2.5V 2.5V ΔIL = 1– = 2.4A 550kHz ) (1.5µH) 12V ( An optional 22µF ceramic output capacitor is included to minimize the effect of ESL in the output ripple. The complete circuit is shown in Figure 6. INTVCC CF 0.1µF 50V CVCC 4.7µF 6.3V VIN RF1 1Ω EXTVCC C4 0.01µF SW GND 50 51 52 NC SGND SVIN EXTVCC INTVCC SW PGND PGND PGND PGND PGND PGND INTVCC NC LTC3609 SW SGND PVIN FCB PVIN ITH PVIN VRNG PVIN PGOOD PVIN SGND 1 2 3 4 5 6 7 8 9 10 11 CIN: MURATA GRM32ER71H475K COUT: MURATA GRM435R60J107M LI: CDEP851R2MC-50 KEEP POWER GROUND AND SIGNAL GROUND SEPARATE. CONNECT AT ONE POINT. INTVCC DB CMDSH-3 CB1 0.22µF SW 26 25 R1 9.53k 1% 24 23 22 VOUT VIN INTVCC JP1 20 CC1 R5 1000pF 15.8k 19 18 17 16 15 CC2 100pF RPG1 100k SGND CVON 0.1µF RON 187k 1% 21 12 13 14 2Ω SW R2 30.1k 1% SGND 49 SW VON CIN 4.7µF 50V x2 48 ION RUN/SS VIN SW BOOST VIN 5V TO 32V VFB SGND 47 SW NC 46 NC SW 45 SW PVIN GND NC PVIN 44 SW PVIN 43 SGND PVIN + SW PVIN COUT1 100µF x2 42 PVIN 41 L1 1.2µH PVIN VOUT 2.5V AT 6A PGND 40 39 38 37 36 35 34 33 32 31 30 29 28 27 VOUT PGOOD INTVCC 3609 F06 CSS 0.1µF Figure 6. Design Example: 5V to 32V Input to 2.5V/6A at 550kHz 3609fb 18 LTC3609 Applications Information How to Reduce SW Ringing As with any switching regulator, there will be voltage ringing on the SW node, especially for high input voltages. The ringing amplitude and duration is dependent on the switching speed (gate drive), layout (parasitic inductance) and MOSFET output capacitance. This ringing contributes to the overall EMI, noise and high frequency ripple. One way to reduce ringing is to optimize layout. A good layout minimizes parasitic inductance. Adding RC snubbers from SW to GND is also an effective way to reduce ringing. Finally, adding a resistor in series with the BOOST pin will slow down the MOSFET turn-on slew rate to dampen ringing, but at the cost of reduced efficiency. Note that since the IC is buffered from the high frequency transients by PCB and bondwire inductances, the ringing by itself is normally not a concern for controller reliability. PC Board Layout Checklist When laying out a PC board follow one of the two suggested approaches. The simple PC board layout requires a dedicated ground plane layer. Also, for higher currents, a multilayer board is recommended to help with heat sinking of power components. • The ground plane layer should not have any traces and it should be as close as possible to the layer with the LTC3609. • Place CIN and COUT all in one compact area, close to the LTC3609. It may help to have some components on the bottom side of the board. • Use a compact plane for the switch node (SW) to improve cooling of the MOSFETs and to keep EMI down. • Use planes for VIN and VOUT to maintain good voltage filtering and to keep power losses low. • Flood all unused areas on all layers with copper. Flooding with copper reduces the temperature rise of power components. Connect these copper areas to any DC net (VIN, VOUT, GND or to any other DC rail in your system). When laying out a printed circuit board without a ground plane, use the following checklist to ensure proper operation of the controller. These items are also illustrated in Figure 7. • Segregate the signal and power grounds. All smallsignal components should return to the SGND pin at one point, which is then tied to the PGND pin. • Connect the input capacitor(s), CIN, close to the IC. This capacitor carries the MOSFET AC current. • Keep the high dV/dT SW, BOOST and TG nodes away from sensitive small-signal nodes. • Connect the INTVCC decoupling capacitor, CVCC, closely to the INTVCC and PGND pins. • Connect the top driver boost capacitor, CB, closely to the BOOST and SW pins. • Connect the VIN pin decoupling capacitor, CF , closely to the VIN and PGND pins. • Keep small-signal components close to the LTC3609. • Ground connections (including LTC3609 SGND and PGND) should be made through immediate vias to the ground plane. Use several larger vias for power components. 3609fb 19 LTC3609 Applications Information CVCC SW NC SGND SVIN EXTVCC INTVCC SW PGND PGND PGND PGND INTVCC NC LTC3609 SW SGND FCB PVIN ITH PVIN VRNG PVIN PGOOD PVIN SGND 2 3 4 5 6 7 8 9 10 11 12 13 14 VON 1 DB CB 26 25 R1 R2 24 23 RON 22 21 20 19 RC 18 CC1 17 16 15 CC2 SGND PVIN RUN/SS 52 SW BOOST 51 ION SGND 50 SW NC 49 VFB SW CIN SW PVIN 48 NC PVIN 47 SW PVIN 46 NC PVIN 45 SW PVIN 44 PGND PGND 43 SGND PVIN 42 SW PVIN 41 VOUT PGND 40 39 38 37 36 35 34 33 32 31 30 29 28 27 COUT CSS RF KEEP POWER GROUND AND SIGNAL GROUND SEPARATE. CONNECT AT ONE POINT. 3609 F07 Figure 7. LTC3609 Layout Diagram 3609fb 20 LTC3609 Typical Applications 3.6V Input to 1.5V/6A at 750kHz INTVCC VBIAS 5V CF 0.1µF 50V CVCC 4.7µF 6.3V EXTVCC C4 0.01µF SW GND R2 60.4k 1% 52 NC SGND SVIN EXTVCC INTVCC SW INTVCC PGND PGND PGND PGND PGND PGND SW SGND PVIN FCB PVIN ITH PVIN VRNG PVIN PGOOD PVIN SGND 1 2 3 4 5 6 7 8 9 10 11 KEEP POWER GROUND AND SIGNAL GROUND SEPARATE. CONNECT AT ONE POINT. INTVCC DB CMDSH-3 24 RON 113k 1% 23 22 INTVCC VIN 21 JP1 20 19 CC1 1500pF R5 8.45k 18 17 16 15 SGND CVON 0.1µF CC2 100pF RPG1 100k 12 13 14 CB1 0.22µF SW 26 25 2Ω SW CIN: MURATA GRM32ER71H475K COUT: MURATA GRM435R60J167M LI: CDEP850R5MC-125 VOUT R1 40.2k 1% SGND 51 NC LTC3609 VON 50 SW RUN/SS 49 ION BOOST 48 SW SGND CIN 4.7µF 50V x2 VFB NC VIN 3.6V VIN SW SW 47 NC PVIN 46 NC SW PVIN 45 SW PVIN 44 SGND PVIN 43 SW PVIN COUT1 100µF x2 42 PVIN GND 41 L1 0.5µH PVIN VOUT 1.5V AT 6A PGND 40 39 38 37 36 35 34 33 32 31 30 29 28 27 PGOOD INTVCC VOUT 3609 TA02 CSS 0.1µF Transient Response Efficiency vs Load Current 100 95 IL 5A/DIV EFFICIENCY (%) 90 VOUT 200mV/DIV 20µs/DIV LOAD STEP 1A TO 5A VIN = 3.6V VOUT = 1.5V FCB = 0V 3609 TA02b 85 80 75 70 65 60 55 50 0.01 VIN = 3.6V FREQUENCY = 750kHz 0.1 1 LOAD CURRENT (A) 10 3609 TA02c 3609fb 21 LTC3609 Typical Applications 5V to 32V Input to 1.2V/6A at 550kHz INTVCC CF 0.1µF 25V CVCC 4.7µF 6.3V VIN RF1 1Ω EXTVCC C4 0.01µF SW GND 50 51 52 NC SGND SVIN EXTVCC INTVCC SW INTVCC PGND PGND PGND PGND PGND PGND SW SGND PVIN FCB PVIN ITH PVIN VRNG PVIN PGOOD PVIN SGND 1 2 3 4 5 6 7 8 9 10 11 C5: TAIYO YUDEN JMK316BJ226ML-T CIN: MURATA GRM32ER71H475K COUT: MURATA GRM435R60J167M LI: CDEP850R8MC-88 INTVCC DB CMDSH-3 KEEP POWER GROUND AND SIGNAL GROUND SEPARATE. CONNECT AT ONE POINT. 25 R1 60.4k 1% 24 23 22 RON 182k 1% VOUT JP1 20 19 R5 8.45k 18 CC1 1500pF 17 16 R3 0Ω 15 CC2 100pF RPG1 100k PGOOD INTVCC VOUT CVON 0.1µF INTVCC VIN 21 SGND CB1 0.22µF SW 26 12 13 14 2Ω SW R2 60.4k 1% SGND 49 NC LTC3609 VON 48 SW RUN/SS CIN 4.7µF 50V x2 ION BOOST VIN VIN 5V TO 32V SW SGND 47 VFB NC 46 SW SW 45 NC PVIN GND SW PVIN 44 NC PVIN 43 SW PVIN 42 SGND PVIN COUT1 100µF x2 SW PVIN 41 L1 0.8µH PVIN VOUT 1.2V AT 6A PGND 40 39 38 37 36 35 34 33 32 31 30 29 28 27 3609 TA03 CSS 0.1µF Transient Response Efficiency vs Load Current 90 85 IL 5A/DIV VIN = 12V FREQUENCY = 550kHz EFFICIENCY (%) 80 VOUT 200mV/DIV 75 70 65 60 20µs/DIV LOAD STEP 1A TO 6A VIN = 12V VOUT = 1.2V FCB = 0V 3609 TA02b 55 50 0.01 0.1 1 LOAD CURRENT (A) 10 3609 TA02c 3609fb 22 LTC3609 Typical Applications 5V to 32V Input to 1.8V/6A All Ceramic 1MHz INTVCC CF 0.1µF 50V CVCC 4.7µF 6.3V VIN RF1 1Ω EXTVCC C4 0.01µF SW GND 51 52 NC SGND SVIN EXTVCC INTVCC SW PGND PGND PGND PGND PGND PGND INTVCC NC LTC3609 SW SGND PVIN FCB PVIN ITH PVIN VRNG PVIN PGOOD PVIN SGND 1 2 3 4 5 6 7 8 9 10 11 INTVCC DB CMDSH-3 KEEP POWER GROUND AND SIGNAL GROUND SEPARATE. CONNECT AT ONE POINT. R1 10k 1% 24 23 22 CVON 0.1µF VOUT RON 102k 1% INTVCC VIN 21 JP1 20 19 R5 5.76k 18 CC1 1500pF 17 16 15 CC2 100pF RPG1 100k SGND CB1 0.22µF SW 26 25 12 13 14 2Ω SW C5: TAIYO YUDEN JMK316BJ226ML-T CIN: MURATA GRM32ER71H475K COUT: MURATA GRM32ER60J107M LI: 1HLP25CZERR80M01 R2 20k 1% SGND 50 SW VON 49 ION RUN/SS 48 SW BOOST CIN 4.7µF x2 VFB SGND VIN 5V TO 32V VIN SW NC 47 NC SW 46 SW PVIN 45 NC PVIN GND SW PVIN 44 SGND PVIN 43 SW PVIN COUT1 100µF x2 42 PVIN 41 L1 0.47µH PVIN VOUT 1.8V AT 6A PGND 40 39 38 37 36 35 34 33 32 31 30 29 28 27 PGOOD INTVCC VOUT 3609 TA04 CSS 0.1µF Transient Response Efficiency vs Load Current 100 90 IL 5A/DIV EFFICIENCY (%) 80 VOUT 200mV/DIV 20µs/DIV LOAD STEP 500mA TO 4A VIN = 12V VOUT = 1.8V FCB = 0V 3609 TA04b 70 60 50 40 30 20 10 0 10 100 1000 LOAD CURRENT (mA) 10000 3609 TA04c 3609fb 23 LTC3609 Package Description WKG Package 52-Lead QFN Multipad (7mm × 8mm) (Reference LTC DWG # 05-08-1768 Rev Ø) SEATING PLANE A 7.00 BSC 0.00 – 0.05 2.625 REF 41 B PAD 1 CORNER 4 2.90 REF 0.50 BSC 1 40 bbb M C A B 7 8.00 BSC PIN 1 ID 52 3.40 REF 3.90 ± 0.10 3.20 ± 0.10 2.025 ± 0.10 2.925 ± 0.10 3.40 REF 33 8 32 9 1.00 REF 10 aaa C 2x NX b TOP VIEW 0.90 ± 0.10 NX 0.08 C // ccc C 8 7.50 ± 0.05 2.90 REF 0.50 BSC 1.35 ± 0.10 1.775 REF 15 0.25 ± 0.05 BOTTOM VIEW (BOTTOM METALLIZATION DETAILS) MLP52 QFN REV Ø 0807 NOTE: 1. DIMENSIONING AND TOLERANCING CONFORM TO ASME Y14.5M-1994 2. ALL DIMENSIONS ARE IN MILLIMETERS, ANGLES ARE IN DEGREES (°) 3. N IS THE TOTAL NUMBER OF TERMINALS 4 2.025 ± 0.10 3.40 REF 2.925 ± 0.10 3.90 ± 0.10 8.50 ± 0.05 4.275 ± 0.10 PACKAGE OUTLINE 0.40 ± 0.10 1.775 REF 1.35 ± 0.10 THE LOCATION OF THE TERMINAL #1 IDENTIFIER AND TERMINAL NUMBERING CONVENTION CONFORMS TO JEDEC PUBLICATION 95 SPP-002 5. ND AND NE REFER TO THE NUMBER OF TERMINALS ON EACH D AND E SIDE RESPECTIVELY 6. NJR REFER TO NON JEDEC REGISTERED 7 DIMENSION b APPLIES TO METALLIZED TERMINAL AND IS MEASURED BETWEEN 0.20mm AND 0.30mm FROM THE TERMINAL TIP. IF THE TERMINAL HAS THE OPTIONAL RADIUS ON THE OTHER END OF THE TERMINAL, THE DIMENSION b SHOULD NOT BE MEASURED IN THAT RADIUS AREA. 8 COPLANARITY APPLIES TO THE TERMINALS AND ALL OTHER SURFACE METALLIZATION 9 DRAWING SHOWN ARE FOR ILLUSTRATION ONLY 1.00 REF 0.25 ± 0.05 0.40 ± 0.10 19 2.625 REF PIN 1 2.25 ± 0.10 14 26 9 3.40 REF 2.25 ± 0.10 27 0.580 ± 0.10 aaa C 2x 3.20 ± 0.10 4.275 ± 0.10 SYMBOL TOLERANCE aaa 0.15 bbb 0.10 ccc 0.10 RECOMMENDED SOLDER PAD LAYOUT TOP VIEW 3609fb 24 LTC3609 Revision History REV DATE B 06 /10 (Revision history begins at Rev B) DESCRIPTION PAGE NUMBER Updated SW voltage range in Absolute Maximum Ratings. 2 Note 4 updated. 4 3609fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 25 LTC3609 Typical Application INTVCC CF 0.1µF 50V CVCC 4.7µF 6.3V VIN RF1 1Ω EXTVCC C4 0.01µF SW GND 50 51 52 NC SGND SVIN EXTVCC INTVCC SW PGND PGND PGND PGND PGND PGND INTVCC NC LTC3609 SW SGND PVIN FCB PVIN ITH PVIN VRNG PVIN PGOOD PVIN SGND 1 2 3 4 5 6 7 8 9 10 11 CIN: MURATA GRM31CR71H475K COUT: SANYO 16SVP180MX LI: CDEP4R3MC-88 INTVCC DB CMDSH-3 KEEP POWER GROUND AND SIGNAL GROUND SEPARATE. CONNECT AT ONE POINT. R1 3.16k RON 1% 1M 1% 24 23 22 VOUT INTVCC VIN 21 JP1 20 19 R5 24.3k 18 CC1 1000pF 17 16 15 RPG1 100k 12 13 14 CB1 0.22µF SW 26 25 PGOOD SGND 2Ω SW R2 60.4k 1% SGND 49 SW VON 48 ION RUN/SS VIN CIN 4.7µF x2 VFB SW BOOST VIN 14V TO 32V SW SGND 47 NC NC 46 SW SW 45 NC PVIN GND SW PVIN 44 SGND PVIN 43 SW PVIN 42 PVIN COUT1 180µF 16V L1 4.3µH PVIN 41 + PVIN VOUT 12V AT 4A PGND 40 39 38 37 36 35 34 33 32 31 30 29 28 27 CVON 0.1µF INTVCC CC2 100pF INTVCC 3609 TA05 CSS 0.1µF Figure 8. 14V to 32V Input to 12V/4A at 500kHz Related Parts PART NUMBER DESCRIPTION COMMENTS LTC3602 2.5A (IOUT), 3MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4.5V to 10V, VOUT(MIN) = 0.6V, IQ = 75µA, ISD <1µA, 4mm × 4mm QFN-20, TSSOP-16E Packages LTC3608 18V, 8A (IOUT), 1MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4V to 18V, VOUT(MIN) = 0.6V, IQ = 900µA, ISD <15µA, 7mm × 8mm QFN-52 Package LTC3610 24V, 12A (IOUT), 1MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4V to 24V, VOUT(MIN) = 0.6V, IQ = 900µA, ISD <15µA, 9mm × 9mm QFN-64 Package LTC3611 32V, 10A (IOUT), 1MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4V to 32V, VOUT(MIN) = 0.6V, IQ = 900µA, ISD <15µA, 9mm × 9mm QFN-64 Package LTC3414/ LTC3416 4A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.25V to 5.5V, VOUT(MIN) = 0.8V, IQ = 64µA, ISD <1µA, TSSOP20E Package LTC3415 7A (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 450µA, ISD <1µA, 5mm × 7mm QFN-38 Package LTC3418 8A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.25V to 5.5V, VOUT(MIN) = 0.8V, IQ = 380µA, ISD <1µA, 5mm × 7mm QFN-38 Package 3609fb 26 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 l FAX: (408) 434-0507 l www.linear.com LT 0610 REV B • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2008