CC1021 CC1021 Single Chip Low Power RF Transceiver for Narrowband Systems Applications • Low power UHF wireless data transmitters and receivers with channel spacings of 50 kHz or higher • 433, 868 and 915 MHz ISM/SRD band systems • AMR ñ Automatic Meter Reading • • • • Wireless alarm and security systems Home automation Low power telemetry Automotive (RKE/TPMS) Product Description is a true single-chip UHF transceiver designed for very low power and very low voltage wireless applications. The circuit is mainly intended for the ISM (Industrial, Scientific and Medical) and SRD (Short Range Device) frequency bands at 433, 868 and 915 MHz, but can easily be programmed for multi-channel operation at other frequencies in the 402 470 and 804 - 940 MHz range. CC1021 In a typical system will be used together with a microcontroller and a few external passive components. CC1021 is based on Chipconís SmartRF02 technology in 0.35 µm CMOS. CC1021 The is especially suited for narrowband systems with channel spacings of 50 kHz and higher complying with EN 300 220 and FCC CFR47 part 15. CC1021 The main operating parameters can be programmed via a serial bus, thus making a very flexible and easy to use transceiver. CC1021 CC1021 Features • True single chip UHF RF transceiver • Frequency range 402 MHz - 470 MHz and 804 MHz - 940 MHz • High sensitivity (up to ñ112 dBm for 38.4 kHz and ñ106 dBm for 102.4 kHz receiver channel filter bandwidths respectively) • Programmable output power • Low current consumption (RX: 19.9 mA) • Low supply voltage (2.3 V to 3.6 V) • Very few external components required • Small size (QFN 32 package) • Pb-free package • Digital RSSI and carrier sense indicator • Data rate up to 153.6 kBaud OOK, FSK and GFSK data modulation Integrated bit synchronizer Image rejection mixer Programmable frequency Automatic frequency control (AFC) Suitable for frequency hopping systems Suited for systems targeting compliance with EN 300 220 and FCC CFR47 part 15 • Development kit available • Easy-to-use software for generating the configuration data • Fully compatible with for receiver channel filter bandwidths of 38.4 kHz and higher • • • • • • • CC1021 CC1020 SWRS045 Page 1 of 91 CC1021 Table of Contents 1. Abbreviations.................................................................................................................4 2. Absolute Maximum Ratings .........................................................................................5 3. Operating Conditions....................................................................................................5 4. Electrical Specifications ...............................................................................................5 4.1. RF Transmit Section.............................................................................................6 4.2. RF Receive Section ..............................................................................................8 4.3. RSSI / Carrier Sense Section .............................................................................11 4.4. IF Section............................................................................................................11 4.5. Crystal Oscillator Section ...................................................................................12 4.6. Frequency Synthesizer Section..........................................................................13 4.7. Digital Inputs / Outputs .......................................................................................14 4.8. Current Consumption .........................................................................................15 5. Pin Assignment ...........................................................................................................15 6. Circuit Description ......................................................................................................17 7. Application Circuit.......................................................................................................18 8. Configuration Overview ..............................................................................................21 8.1. 9. Configuration Software .......................................................................................21 Microcontroller Interface ............................................................................................22 9.1. 4-wire Serial Configuration Interface ..................................................................23 9.2. Signal Interface...................................................................................................25 10. Data Rate Programming .............................................................................................27 11. Frequency Programming ............................................................................................28 11.1. 12. Dithering..........................................................................................................29 Receiver........................................................................................................................30 12.1. IF Frequency ...................................................................................................30 12.2. Receiver Channel Filter Bandwidth ................................................................30 12.3. Demodulator, Bit Synchronizer and Data Decision.........................................31 12.4. Receiver Sensitivity versus Data Rate and Frequency Separation ................32 12.5. RSSI................................................................................................................33 12.6. Image Rejection Calibration............................................................................35 12.7. Blocking and Selectivity ..................................................................................36 12.8. Linear IF Chain and AGC Settings .................................................................38 12.9. AGC Settling ...................................................................................................40 12.10. Preamble Length and Sync Word...................................................................40 12.11. Carrier Sense ..................................................................................................41 12.12. Automatic Power-up Sequencing ...................................................................41 12.13. Automatic Frequency Control .........................................................................42 12.14. Digital FM ........................................................................................................43 SWRS045 Page 2 of 91 CC1021 13. Transmitter...................................................................................................................44 13.1. FSK Modulation Formats ................................................................................44 13.2. Output Power Programming ...........................................................................44 13.3. TX Data Latency .............................................................................................45 13.4. Reducing Spurious Emission and Modulation Bandwidth ..............................46 14. Input / Output Matching and Filtering .......................................................................46 15. Frequency Synthesizer ...............................................................................................50 15.1. VCO, Charge Pump and PLL Loop Filter .......................................................50 15.2. VCO and PLL Self-Calibration ........................................................................51 15.3. PLL Turn-on Time versus Loop Filter Bandwidth ...........................................52 15.4. PLL Lock Time versus Loop Filter Bandwidth ................................................53 16. VCO and LNA Current Control ...................................................................................53 17. Power Management.....................................................................................................54 18. On-Off Keying (OOK)...................................................................................................56 19. Crystal Oscillator.........................................................................................................58 20. Built-in Test Pattern Generator ..................................................................................59 21. Interrupt on Pin DCLK.................................................................................................60 22. 21.1. Interrupt upon PLL Lock..................................................................................60 21.2. Interrupt upon Received Signal Carrier Sense ...............................................60 PA_EN and LNA_EN Digital Output Pins ..................................................................60 22.1. Interfacing an External LNA or PA ..................................................................60 22.2. General Purpose Output Control Pins ............................................................61 22.3. PA_EN and LNA_EN Pin Drive.......................................................................61 23. System Considerations and Guidelines ...................................................................61 24. PCB Layout Recommendations.................................................................................63 25. Antenna Considerations.............................................................................................64 26. Configuration Registers .............................................................................................64 26.1. 27. CC1021 Register Overview ............................................................................65 Package Description (QFN 32) ...................................................................................85 27.1. Package Marking ............................................................................................86 27.2. Recommended PCB Footprint for Package (QFN 32) ...................................87 27.3. Package Thermal Properties ..........................................................................87 27.4. Soldering Information......................................................................................87 27.5. Plastic Tube Specification...............................................................................88 27.6. Carrier Tape and Reel Specification ...............................................................88 28. Ordering Information...................................................................................................88 29. General Information ....................................................................................................89 30. Address Information ...................................................................................................91 SWRS045 Page 3 of 91 CC1021 1. ACP ACR ADC AFC AGC AMR ASK BER BOM bps BT ChBW CW DAC DNM ESR FHSS FM FS FSK GFSK IC IF IP3 ISM kbps LNA LO MCU NRZ OOK PA PD PER PCB PN9 PLL PSEL RF RKE RSSI RX SBW SPI SRD TBD TPMS T/R TX UHF VCO VGA XOSC XTAL Abbreviations Adjacent Channel Power Adjacent Channel Rejection Analog-to-Digital Converter Automatic Frequency Control Automatic Gain Control Automatic Meter Reading Amplitude Shift Keying Bit Error Rate Bill Of Materials bits per second Bandwidth-Time product (for GFSK) Receiver Channel Filter Bandwidth Continuous Wave Digital-to-Analog Converter Do Not Mount Equivalent Series Resistance Frequency Hopping Spread Spectrum Frequency Modulation Frequency Synthesizer Frequency Shift Keying Gaussian Frequency Shift Keying Integrated Circuit Intermediate Frequency Third Order Intercept Point Industrial Scientific Medical kilo bits per second Low Noise Amplifier Local Oscillator (in receive mode) Micro Controller Unit Non Return to Zero On-Off Keying Power Amplifier Phase Detector / Power Down Packet Error Rate Printed Circuit Board Pseudo-random Bit Sequence (9-bit) Phase Locked Loop Program Select Radio Frequency Remote Keyless Entry Received Signal Strength Indicator Receive (mode) Signal Bandwidth Serial Peripheral Interface Short Range Device To Be Decided/Defined Tire Pressure Monitoring Transmit/Receive (switch) Transmit (mode) Ultra High Frequency Voltage Controlled Oscillator Variable Gain Amplifier Crystal oscillator Crystal SWRS045 Page 4 of 91 CC1021 2. Absolute Maximum Ratings The absolute maximum ratings given Table 1 should under no circumstances be violated. Stress exceeding one or more of the limiting values may cause permanent damage to the device. Min Max Unit Supply voltage, VDD Parameter -0.3 5.0 V Voltage on any pin Input RF level Storage temperature range Package body temperature Humidity non-condensing ESD (Human Body Model) -0.3 VDD+0.3, max 5.0 10 150 260 85 ±1 ±0.4 V dBm °C °C % kV kV -50 5 Condition All supply pins must have the same voltage Norm: IPC/JEDEC J-STD-020C 1 All pads except RF RF Pads Table 1. Absolute maximum ratings 1 The reflow peak soldering temperature (body temperature) is specified according to IPC/JEDEC J-STD_020C ìMoisture/Reflow Sensitivity Classification for Nonhermetic Solid State Surface Mount Devicesî. Caution! ESD sensitive device. Precaution should be used when handling the device in order to prevent permanent damage. 3. Operating Conditions The operating conditions for Parameter CC1021 are listed in Table 2. Min Typ Max Unit RF Frequency Range 402 804 470 940 MHz MHz Operating ambient temperature range -40 85 °C Supply voltage 2.3 3.6 V 3.0 Condition / Note Programmable in <300 Hz steps Programmable in <600 Hz steps The same supply voltage should be used for digital (DVDD) and analog (AVDD) power. Table 2. Operating conditions 4. Electrical Specifications Table 3 to Table 10 gives the electrical specifications. All measurements were performed using the 2 layer PCB CC1020EMX reference design. This is the same test circuit as shown in Figure 3. Temperature = 25°C, supply voltage = AVDD = DVDD = 3.0 V if nothing else stated. Crystal frequency = 14.7456 MHz. CC1021 The electrical specifications given for 868 MHz are also applicable for the 902 ñ 928 MHz frequency range. SWRS045 Page 5 of 91 CC1021 4.1. RF Transmit Section Parameter Transmit data rate Min Typ 0.45 Max Unit 153.6 kBaud Condition / Note The data rate is programmable. See section 10 on page 27 for details. NRZ or Manchester encoding can be used. 153.6 kBaud equals 153.6 kbps using NRZ coding and 76.8 kbps using Manchester coding. See section 9.2 on page 25 for details Minimum data rate for OOK is 2.4 kBaud Binary FSK frequency separation 0 0 108 216 kHz kHz in 402 - 470 MHz range in 804 - 940 MHz range 108/216 kHz is the maximum guaranteed separation at 1.84 MHz reference frequency. Larger separations can be achieved at higher reference frequencies. Output power 433 MHz -20 to +10 dBm 868 MHz -20 to +5 dBm -4 +3 dB dB 2nd harmonic, 433 MHz, +10 dBm 3rd harmonic, 433 MHz, +10 dBm -50 -50 dBc dBc 2nd harmonic, 868 MHz, +5 dBm 3rd harmonic, 868 MHz, +5 dBm -50 -50 dBc dBc Output power tolerance Harmonics, radiated CW Adjacent channel power (GFSK) 433 MHz -46 dBc 868 MHz -42 dBc Occupied bandwidth (99.5%,GFSK) 433 MHz 60 kHz 868 MHz 60 kHz Modulation bandwidth, 868 MHz 19.2 kBaud, ±9.9 kHz frequency deviation 48 kHz 38.4 kBaud, ±19.8 kHz frequency deviation 106 kHz SWRS045 Delivered to 50 Ω single-ended load. The output power is programmable and should not be programmed to exceed +10/+5 dBm at 433/868 MHz under any operating conditions. See section 14 on page 46 for details. At maximum output power At 2.3 V, +85oC At 3.6 V, -40oC Harmonics are measured as EIRP values according to EN 300 220. The antenna (SMAFF-433 and SMAFF-868 from R.W. Badland) plays a part in attenuating the harmonics. ACP is measured in a 100 kHz bandwidth at ±100 kHz offset. Modulation: 19.2 kBaud NRZ PN9 sequence, ±19.8 kHz frequency deviation. Bandwidth for 99.5% of total average power. Modulation: 19.2 kBaud NRZ PN9 sequence, ±19.8 kHz frequency deviation. Bandwidth where the power envelope of modulation equals ñ36 dBm. Spectrum analyzer RBW = 1 kHz. Page 6 of 91 CC1021 Parameter Min Typ Max Unit Spurious emission, radiated CW Condition / Note At maximum output power, +10/+5 dBm at 433/868 MHz. 47-74, 87.5-118, 174-230, 470-862 MHz -54 dBm 9 kHz ñ 1 GHz -36 dBm 1 ñ 4 GHz -30 dBm To comply with EN 300 220, FCC CFR47 part 15 and ARIB STD T-67 an external (antenna) filter, as implemented in the application circuit in Figure 25, must be used and tailored to each individual design to reduce out-of-band spurious emission levels. Spurious emissions can be measured as EIRP values according to EN 300 220. The antenna (SMAFF-433 and SMAFF-868 from R.W. Badland) plays a part in attenuating the spurious emissions. If the output power is increased using an external PA, a filter must be used to attenuate spurs below 862 MHz when operating in the 868 MHz frequency band in Europe. Application Note AN036 CC1020/1021 Spurious Emission presents and discusses a solution that reduces the TX mode spurious emission close to 862 MHz by increasing the REF_DIV from 1 to 7. Optimum load impedance 433 MHz 54 + j44 Ω 868 MHz 15 + j24 Ω 915 MHz 20 + j35 Ω Transmit mode. For matching details see section 14 on page 46. Table 3. RF transmit parameters SWRS045 Page 7 of 91 CC1021 4.2. RF Receive Section Parameter Min Typ Max Unit Receiver Sensitivity, 433 MHz, FSK Condition / Note Sensitivity is measured with PN9 sequence at BER = 10 3 38.4 kHz channel filter BW (1) -109 dBm (1) 38.4 kHz receiver channel filter bandwidth: 4.8 kBaud, NRZ coded data, ±4.95 kHz frequency deviation. 102.4 kHz channel filter BW (2) -104 dBm (2) 102.4 kHz receiver channel filter bandwidth: 19.2 kBaud, NRZ coded data, ±19.8 kHz frequency deviation. 102.4 kHz channel filter BW (3) -104 dBm (3) 102.4 kHz receiver channel filter bandwidth: 38.4 kBaud, NRZ coded data, ±19.8 kHz frequency deviation. 307.2 kHz channel filter BW (4) -96 dBm (4) 307.2 kHz receiver channel filter bandwidth: 153.6 kBaud, NRZ coded data, ±72 kHz frequency deviation. 38.4 kHz channel filter BW (1) -108 dBm 102.4 kHz channel filter BW (2) -103 dBm 102.4 kHz channel filter BW (3) -103 dBm 307.2 kHz channel filter BW (4) -94 dBm Receiver Sensitivity, 868 MHz, FSK Receiver sensitivity, 433 MHz, OOK 9.6 kBaud 153.6 kBaud -103 -81 dBm dBm Receiver sensitivity, 868 MHz, OOK 9.6 kBaud 153.6 kBaud Saturation (maximum input level) FSK and OOK System noise bandwidth Noise figure, cascaded 433 and 868 MHz See Table 19 and Table 20 or typical sensitivity figures at other channel filter bandwidths. Sensitivity is measured with PN9 sequence at BER = 10 3 Manchester coded data. See Table 27 for typical sensitivity figures at other data rates. -104 -87 dBm dBm 10 dBm FSK: Manchester/NRZ coded data OOK: Manchester coded data BER = 10 3 38.4 to 307.2 kHz The receiver channel filter 6 dB bandwidth is programmable from 38.4 kHz to 307.2 kHz. See section 12.2 on page 30 for details. 7 dB NRZ coded data SWRS045 Page 8 of 91 CC1021 Parameter Min Typ Max Unit Input IP3 Condition / Note Two tone test (+10 MHz and +20 MHz) 433 MHz 102.4 kHz channel filter BW -23 -18 -16 dBm dBm dBm LNA2 maximum gain LNA2 medium gain LNA2 minimum gain 868 MHz 102.4 kHz channel filter BW -18 -15 -13 dBm dBm dBm LNA2 maximum gain LNA2 medium gain LNA2 minimum gain -11 dB Co-channel rejection, FSK and OOK 433 MHz and 868 MHz, 102.4 kHz channel filter BW, Adjacent channel rejection (ACR) 433 MHz 102.4 kHz channel filter BW 32 dB 868 MHz 102.4 kHz channel filter BW 30 dB Image channel rejection 433/868 MHz No I/Q gain and phase calibration 25/25 dB I/Q gain and phase calibrated 50/50 dB Wanted signal 3 dB above the sensitivity level, CW jammer at operating frequency, BER = 10 3 Wanted signal 3 dB above the sensitivity level, CW jammer at adjacent channel, BER = 10 3. Measured at ±100 kHz offset. See Figure 16 to Figure 19. Wanted signal 3 dB above the sensitivity level, CW jammer at image frequency, BER = 10 3. 102.4 kHz channel filter bandwidth. See Figure 16 to Figure 19. Image rejection after calibration will depend on temperature and supply voltage. Refer to section 12.6 on page 35. Selectivity* 433 MHz 102.4 kHz channel filter BW ±200 kHz offset ±300 kHz offset 45 53 dB dB 868 MHz 102.4 kHz channel filter BW ±200 kHz offset ±300 kHz offset 45 50 dB dB Wanted signal 3 dB above the sensitivity level. CW jammer is swept in 20 kHz steps within ± 1 MHz from wanted channel. BER = 10 3. Adjacent channel and image channel are excluded. See Figure 16 to Figure 19. (*Close-in spurious response rejection) Blocking / Desensitization* 433/868 MHz ± 1 MHz ± 2 MHz ± 5 MHz ± 10 MHz 52/58 56/64 58/64 64/66 dB dB dB dB (*Out-of-band spurious response rejection) Image frequency suppression, 433/868 MHz No I/Q gain and phase calibration 35/35 dB I/Q gain and phase calibrated 60/60 dB SWRS045 Wanted signal 3 dB above the sensitivity level, CW jammer at ± 1, 2, 5 and 10 MHz offset, BER = 10 3. 102.4 kHz channel filter bandwidth. Complying with EN 300 220, class 2 receiver requirements. Ratio between sensitivity for a signal at the image frequency to the sensitivity in the wanted channel. Image frequency is RF 2 IF. BER = 10 3. 102.4 kHz channel filter bandwidth. Page 9 of 91 CC1021 Parameter Min Typ Spurious reception LO leakage, 433/868 MHz Max Unit Condition / Note 37 dB Ratio between sensitivity for an unwanted frequency to the sensitivity in the wanted channel. The signal source is swept over all frequencies 100 MHz ñ 2 GHz. Signal level for BER = 10 3. 102.4 kHz channel filter bandwidth. <-80/-66 dBm -64 dBm 9 kHz ñ 1 GHz <-60 dBm 1 ñ 4 GHz <-60 dBm 433 MHz 58 - j10 Ω 868 MHz 54 - j22 Ω 433 MHz -14 dB 868 MHz -12 dB 433 MHz 39 - j14 Ω 868 MHz 32 - j10 Ω VCO leakage Spurious emission, radiated CW Input impedance Matched input impedance, S11 Matched input impedance Bit synchronization offset 8000 ppm Data latency NRZ mode 4 Baud Manchester mode 8 Baud VCO frequency resides between 1608 ñ 1880 MHz Complying with EN 300 220, FCC CFR47 part 15 and ARIB STD T-67. Spurious emissions can be measured as EIRP values according to EN 300 220. Receive mode. See section 14 on page 46 for details. Using application circuit matching network. See section 14 on page 46 for details. Using application circuit matching network. See section 14 on page 46 for details. The maximum bit rate offset tolerated by the bit synchronization circuit for 6 dB degradation (synchronous modes only) Time from clocking the data on the transmitter DIO pin until data is available on receiver DIO pin Table 4. RF receive parameters SWRS045 Page 10 of 91 CC1021 4.3. RSSI / Carrier Sense Section Parameter Min Typ Max Unit Condition / Note RSSI dynamic range 55 dB See section 12.5 on page 33 for details. RSSI accuracy ±3 dB See section 12.5 on page 33 for details. RSSI linearity ±1 dB RSSI attach time 51.2 kHz channel filter BW 730 µs 102.4 kHz channel filter BW 380 µs 307.2 kHz channel filter BW 140 µs Carrier sense programmable range 40 dB Carrier sense at ±100 kHz and ±200 kHz offset 102.4 kHz channel filter BW, 433 MHz ±100 kHz ±200 kHz -57 -44 dBm dBm 102.4 kHz channel filter BW, 868 MHz ±100 kHz ±200 kHz -60 -44 dBm dBm Shorter RSSI attach times can be traded for lower RSSI accuracy. See section 12.5 on page 33 for details. Shorter RSSI attach times can also be traded for reduced sensitivity and selectivity by increasing the receiver channel filter bandwidth. Accuracy is as for RSSI At carrier sense level 98 dBm, CW jammer at ±100 kHz and ±200 kHz offset. Carrier sense is measured by applying a signal at ±100 kHz and ±200 kHz offset and observe at which level carrier sense is indicated. Table 5. RSSI / Carrier sense parameters 4.4. IF Section Parameter Min Typ Max Unit Condition / Note Intermediate frequency (IF) 307.2 kHz See section 12.1 on page 30 for details. Digital channel filter bandwidth 38.4 to 307.2 kHz The channel filter 6 dB bandwidth is programmable from 9.6 kHz to 307.2 kHz. See section 12.2 on page 30 for details. AFC resolution 1200 Hz At 19.2 kBaud Given as Baud rate/16. See section 12.13 on page 42 for details. Table 6. IF section parameters SWRS045 Page 11 of 91 CC1021 4.5. Crystal Oscillator Section Parameter Crystal Oscillator Frequency Min Typ Max Unit Condition / Note 4.9152 14.7456 19.6608 MHz Recommended frequency is 14.7456 MHz. See section 19 on page 58 for details. Crystal operation Crystal load capacitance Crystal oscillator start-up time External clock signal drive, sine wave External clock signal drive, full-swing digital external clock Parallel 12 12 12 22 16 16 C4 and C5 are loading capacitors. See section 19 on page 58 for details. 30 30 16 pF pF pF 4.9-6 MHz, 22 pF recommended 6-8 MHz, 16 pF recommended 8-19.6 MHz, 16 pF recommended 1.55 1.0 0.90 0.95 0.60 0.63 ms ms ms ms ms ms 4.9152 MHz, 12 pF load 7.3728 MHz, 12 pF load 9.8304 MHz, 12 pF load 14.7456 MHz, 16 pF load 17.2032 MHz, 12 pF load 19.6608 MHz, 12 pF load 300 mVpp 0 - VDD V The external clock signal must be connected to XOSC_Q1 using a DC block (10 nF). Set XOSC_BYPASS = 0 in the INTERFACE register when using an external clock signal with low amplitude or a crystal. The external clock signal must be connected to XOSC_Q1. No DC block shall be used. Set XOSC_BYPASS = 1 in the INTERFACE register when using a full-swing digital external clock. Table 7. Crystal oscillator parameters SWRS045 Page 12 of 91 CC1021 4.6. Frequency Synthesizer Section Parameter Min Phase noise, 402 ñ 470 MHz Typ Max -79 -80 -87 -100 -105 Unit dBc/Hz dBc/Hz dBc/Hz dBc/Hz dBc/Hz Condition / Note Unmodulated carrier At 12.5 kHz offset from carrier At 25 kHz offset from carrier At 50 kHz offset from carrier At 100 kHz offset from carrier At 1 MHz offset from carrier Measured using loop filter components given in Table 13. The phase noise will be higher for larger PLL loop filter bandwidth. Phase noise, 804 ñ 940 MHz -73 -74 -81 -94 -111 dBc/Hz dBc/Hz dBc/Hz dBc/Hz dBc/Hz Unmodulated carrier At 12.5 kHz offset from carrier At 25 kHz offset from carrier At 50 kHz offset from carrier At 100 kHz offset from carrier At 1 MHz offset from carrier Measured using loop filter components given in Table 13. The phase noise will be higher for larger PLL loop filter bandwidth. PLL loop filter bandwidth Loop filter 2, up to 19.2 kBaud 15 kHz Loop filter 3, up to 38.4 kBaud 30.5 kHz Loop filter 2, up to 19.2 kBaud 140 us Loop filter 3, up to 38.4 kBaud 75 us Loop filter 5, up to 153.6 kBaud 14 us PLL lock time (RX / TX turn time) PLL turn-on time. From power down mode with crystal oscillator running. Loop filter 2, up to 19.2 kBaud 1300 us Loop filter 3, up to 38.4 kBaud 1080 us Loop filter 5, up to 153.6 kBaud 700 us After PLL and VCO calibration. The PLL loop bandwidth is programmable. See Table 25 on page 52 for loop filter component values. 307.2 kHz frequency step to RF frequency within ±10 kHz, ±15 kHz, ±50 kHz settling accuracy for loop filter 2, 3 and 5 respectively. Depends on loop filter component values and PLL_BW register setting. See Table 26 on page 53 for more details. Time from writing to registers to RF frequency within ±10 kHz, ±15 kHz, ±50 kHz settling accuracy for loop filter 2, 3 and 5 respectively. Depends on loop filter component values and PLL_BW register setting. See Table 25 on page 53 for more details. Table 8. Frequency synthesizer parameters SWRS045 Page 13 of 91 CC1021 4.7. Digital Inputs / Outputs Parameter Min Typ Max Unit Condition / Note Logic "0" input voltage 0 0.3* VDD V Logic "1" input voltage 0.7* VDD VDD V Logic "0" output voltage 0 0.4 V Output current 2.0 mA, 3.0 V supply voltage Logic "1" output voltage 2.5 VDD V Output current 2.0 mA, 3.0 V supply voltage Logic "0" input current NA 1 µA Input signal equals GND. PSEL has an internal pull-up resistor and during configuration the current will be -350 µA. Logic "1" input current NA DIO setup time DIO hold time 1 µA Input signal equals VDD 20 ns TX mode, minimum time DIO must be ready before the positive edge of DCLK. Data should be set up on the negative edge of DCLK. 10 ns TX mode, minimum time DIO must be held after the positive edge of DCLK. Data should be set up on the negative edge of DCLK. Serial interface (PCLK, PDI, PDO and PSEL) timing specification Pin drive, LNA_EN, PA_EN See Table 14 on page 24 for more details 0.90 0.87 0.81 0.69 mA mA mA mA Source current 0 V on LNA_EN, PA_EN pins 0.5 V on LNA_EN, PA_EN pins 1.0 V on LNA_EN, PA_EN pins 1.5 V on LNA_EN, PA_EN pins 0.93 0.92 0.89 0.79 mA mA mA mA Sink current 3.0 V on LNA_EN, PA_EN pins 2.5 V on LNA_EN, PA_EN pins 2.0 V on LNA_EN, PA_EN pins 1.5 V on LNA_EN, PA_EN pins See Figure 35 on page 61 for more details. Table 9. Digital inputs / outputs parameters SWRS045 Page 14 of 91 CC1021 4.8. Current Consumption Parameter Min Typ Max Unit Condition / Note Power Down mode 0.2 1.8 µA Current Consumption, receive mode 433 and 868 MHz 19.9 mA P = 20 dBm 12.3/14.5 mA P = 5 dBm 14.4/17.0 mA P = 0 dBm 16.2/20.5 mA P = +5 dBm 20.5/25.1 mA P = +10 dBm (433 MHz only) 27.1 mA Current Consumption, crystal oscillator 77 µA 14.7456 MHz, 16 pF load crystal Current Consumption, crystal oscillator and bias 500 µA 14.7456 MHz, 16 pF load crystal Current Consumption, crystal oscillator, bias and synthesizer 7.5 mA 14.7456 MHz, 16 pF load crystal Oscillator core off Current Consumption, transmit mode 433/868 MHz: The output power is delivered to a 50 Ω single-ended load. See section 13.2 on page 44 for more details. Table 10. Current consumption 5. Pin Assignment Table 11 provides an overview of the pinout. CC1021 The comes in a QFN32 type package (see page 85 for details). CC1021 AGND 25 AD_REF 26 AVDD 27 CHP_OUT 28 AVDD 29 DGND 30 DVDD 31 PSEL 32 PCLK PDI PDO DGND DVDD DGND DCLK DIO 1 2 3 4 5 6 7 8 24 VC 23 AVDD 22 AVDD 21 RF_OUT 20 AVDD 19 RF_IN 18 AVDD 17 R_BIAS 16 15 14 13 12 11 10 9 AVDD PA_EN LNA_EN AVDD AVDD XOSC_Q2 XOSC_Q1 LOCK Figure 1. CC1021 AGND Exposed die attached pad package (top view) SWRS045 Page 15 of 91 CC1021 Pin no. - Pin name AGND Pin type Ground (analog) 1 2 3 4 5 6 7 PCLK PDI PDO DGND DVDD DGND DCLK Digital input Digital input Digital output Ground (digital) Power (digital) Ground (digital) Digital output 8 DIO Digital input/output 9 LOCK Digital output 10 11 12 13 14 XOSC_Q1 XOSC_Q2 AVDD AVDD LNA_EN Analog input Analog output Power (analog) Power (analog) Digital output 15 PA_EN Digital output 16 AVDD Power (analog) 17 18 19 20 21 22 R_BIAS AVDD RF_IN AVDD RF_OUT AVDD Analog output Power (analog) RF Input Power (analog) RF output Power (analog) 23 24 25 26 27 28 29 30 31 32 AVDD VC AGND AD_REF AVDD CHP_OUT AVDD DGND DVDD PSEL Power (analog) Analog input Ground (analog) Power (analog) Power (analog) Analog output Power (analog) Ground (digital) Power (digital) Digital input Description Exposed die attached pad. Must be soldered to a solid ground plane as this is the ground connection for all analog modules. See page 63 for more details. Programming clock for SPI configuration interface Programming data input for SPI configuration interface Programming data output for SPI configuration interface Ground connection (0 V) for digital modules and digital I/O Power supply (3 V typical) for digital modules and digital I/O Ground connection (0 V) for digital modules (substrate) Clock for data in both receive and transmit mode. Can be used as receive data output in asynchronous mode Data input in transmit mode; data output in receive mode Can also be used to start power-up sequencing in receive PLL Lock indicator, active low. Output is asserted (low) when PLL is in lock. The pin can also be used as a general digital output, or as receive data output in synchronous NRZ/Manchester mode Crystal oscillator or external clock input Crystal oscillator Power supply (3 V typical) for crystal oscillator Power supply (3 V typical) for the IF VGA General digital output. Can be used for controlling an external LNA if higher sensitivity is needed. General digital output. Can be used for controlling an external PA if higher output power is needed. Power supply (3 V typical) for global bias generator and IF anti-alias filter Connection for external precision bias resistor (82 kΩ, ± 1%) Power supply (3 V typical) for LNA input stage RF signal input from antenna (external AC-coupling) Power supply (3 V typical) for LNA RF signal output to antenna Power supply (3 V typical) for LO buffers, mixers, prescaler, and first PA stage Power supply (3 V typical) for VCO VCO control voltage input from external loop filter Ground connection (0 V) for analog modules (guard) 3 V reference input for ADC Power supply (3 V typical) for charge pump and phase detector PLL charge pump output to external loop filter Power supply (3 V typical) for ADC Ground connection (0 V) for digital modules (guard) Power supply connection (3 V typical) for digital modules Programming chip select, active low, for configuration interface. Internal pull-up resistor. Table 11. Pin assignment overview Note: DCLK, DIO and LOCK are highimpedance (3-state) in power down (BIAS_PD = 1 in the MAIN register). The exposed die attached pad must be soldered to a solid ground plane as this is the main ground connection for the chip. SWRS045 Page 16 of 91 CC1021 Circuit Description ADC RF_IN LNA LNA 2 ADC Multiplexer 0 90 DIGITAL DEMODULATOR - Digital RSSI - Gain Control - Image Suppression - Channel Filtering - Demodulation LOCK :2 DIO 0 90 FREQ SYNTH :2 CONTROL LOGIC 6. DIGITAL INTERFACE TO µC DCLK PDO PDI PCLK Power Control PSEL DIGITAL MODULATOR Multiplexer RF_OUT - Modulation - Data shaping - Power Control PA BIAS PA_EN LNA_EN XOSC XOSC_Q1 XOSC_Q2 R_BIAS Figure 2. CC1021 VC CHP_OUT simplified block diagram is A simplified block diagram of shown in Figure 2. Only signal pins are shown. CC1021 features a low-IF receiver. The received RF signal is amplified by the lownoise amplifier (LNA and LNA2) and down-converted in quadrature (I and Q) to the intermediate frequency (IF). At IF, the I/Q signal is complex filtered and amplified, and then digitized by the ADCs. Automatic gain control, fine channel filtering, demodulation and bit synchronization is performed digitally. outputs the digital demodulated data on the DIO pin. A synchronized data clock is available at the DCLK pin. RSSI is available in digital format and can be read via the serial interface. The RSSI also features a programmable carrier sense indicator. CC1021 CC1021 amplifier (PA). The RF output is frequency shift keyed (FSK) by the digital bit stream that is fed to the DIO pin. Optionally, a Gaussian filter can be used to obtain Gaussian FSK (GFSK). The frequency synthesizer includes a completely on-chip LC VCO and a 90 degrees phase splitter for generating the LO_I and LO_Q signals to the downconversion mixers in receive mode. The VCO operates in the frequency range 1.608-1.880 GHz. The CHP_OUT pin is the charge pump output and VC is the control node of the on-chip VCO. The external loop filter is placed between these pins. A crystal is to be connected between XOSC_Q1 and XOSC_Q2. A lock signal is available from the PLL. The 4-wire SPI serial interface is used for configuration. In transmit mode, the synthesized RF frequency is fed directly to the power SWRS045 Page 17 of 91 CC1021 7. Application Circuit Very few external components are required for the operation of . The recommended application circuit is shown in Figure 3. The external components are described in Table 12 and values are given in Table 13. CC1021 Input / output matching L1 and C1 is the input match for the receiver. L1 is also a DC choke for biasing. L2 and C3 are used to match the transmitter to 50 Ω. Internal circuitry makes it possible to connect the input and output together and match the to 50 Ω in both RX and TX mode. However, it is recommended to use an external T/R switch for optimum performance. See section 14 on page 46 for details. Component values for the matching network are easily found using the SmartRF Studio software. CC1021 Bias resistor The precision bias resistor R1 is used to set an accurate bias current. PLL loop filter The loop filter consists of two resistors (R2 and R3) and three capacitors (C6-C8). C7 and C8 may be omitted in applications Ref C1 C3 C4 C5 C6 C7 C8 C60 L1 L2 R1 R2 R3 R10 XTAL where high loop bandwidth is desired. The values shown in Table 13 are optimized for 38.4 kBaud data rate. Component values for other data rates are easily found using the SmartRF Studio software. Crystal An external crystal with two loading capacitors (C4 and C5) is used for the crystal oscillator. See section 19 on page 58 for details. Additional filtering Additional external components (e.g. RF LC or SAW filter) may be used in order to improve the performance in specific applications. See section 14 on page 46 for further information. Power supply decoupling and filtering Power supply decoupling and filtering must be used (not shown in the application circuit). The placement and size of the decoupling capacitors and the power supply filtering are very important to achieve the optimum performance for narrowband applications. Chipcon provides a reference design that should be followed very closely. Description LNA input match and DC block, see page 46 PA output match and DC block, see page 46 Crystal load capacitor, see page 58 Crystal load capacitor, see page 58 PLL loop filter capacitor PLL loop filter capacitor (may be omitted for highest loop bandwidth) PLL loop filter capacitor (may be omitted for highest loop bandwidth) Decoupling capacitor LNA match and DC bias (ground), see page 46 PA match and DC bias (supply voltage), see page 46 Precision resistor for current reference generator PLL loop filter resistor PLL loop filter resistor PA output match, see page 46 Crystal, see page 58 Table 12. Overview of external components (excluding supply decoupling capacitors) SWRS045 Page 18 of 91 CC1021 AVDD=3V DVDD=3V 25 26 28 27 C7 AGND AD_REF AVDD CHP_OUT DGND 29 PDO 4 AVDD PDI 3 30 2 DGND PCLK DVDD PSEL DVDD=3V 1 31 32 Microcontroller configuration interface and signal interface C6 R2 CC1021 VC 24 AVDD 23 AVDD 22 RF_OUT 21 AVDD 20 5 DVDD 6 DGND RF_IN 19 7 DCLK AVDD 18 8 DIO R_BIAS 17 AVDD=3V R3 R10 C8 L2 Monopole antenna (50 Ohm) C60 C3 LC Filter AVDD=3V T/R Switch C1 AVDD=3V AVDD PA_EN LNA_EN AVDD AVDD XOSC_Q2 LOCK XOSC_Q1 L1 R1 15 16 13 14 12 11 10 9 AVDD=3V XTAL C4 C5 Figure 3. Typical application and test circuit (power supply decoupling not shown) Item C1 C3 C4 C5 C6 C7 C8 C60 L1 L2 R1 R2 R3 R10 XTAL 433 MHz 10 pF, 5%, NP0, 0402 5.6 pF, 5%, NP0, 0402 22 pF, 5%, NP0, 0402 12 pF, 5%, NP0, 0402 3.9 nF, 10%, X7R, 0603 120 pF, 10%, X7R, 0402 33 pF, 10%, X7R, 0402 220 pF, 5%, NP0, 0402 33 nH, 5%, 0402 22 nH, 5%, 0402 82 kΩ, 1%, 0402 12 kΩ, 5%, 0402 39 kΩ, 5%, 0402 82 Ω, 5%, 0402 14.7456 MHz crystal, 16 pF load 868 MHz 47 pF, 5%, NP0, 0402 10 pF, 5%, NP0, 0402 22 pF, 5%, NP0, 0402 12 pF, 5%, NP0, 0402 3.9 nF, 10%, X7R, 0603 120 pF, 10%, X7R, 0402 33 pF, 10%, X7R, 0402 220 pF, 5%, NP0, 0402 82 nH, 5%, 0402 3.6 nH, 5%, 0402 82 kΩ, 1%, 0402 12 kΩ, 5%, 0402 39 kΩ, 5%, 0402 82 Ω, 5%, 0402 14.7456 MHz crystal, 16 pF load 915 MHz 47 pF, 5%, NP0, 0402 10 pF, 5%, NP0, 0402 22 pF, 5%, NP0, 0402 12 pF, 5%, NP0, 0402 3.9 nF, 10%, X7R, 0603 120 pF, 10%, X7R, 0402 33 pF, 10%, X7R, 0402 220 pF, 5%, NP0, 0402 82 nH, 5%, 0402 3.6 nH, 5%, 0402 82 kΩ, 1%, 0402 12 kΩ, 5%, 0402 39 kΩ, 5%, 0402 82 Ω, 5%, 0402 14.7456 MHz crystal, 16 pF load Note: Items shaded vary for different frequencies. Table 13. Bill of materials for the application circuit in Figure 3. The PLL loop filter is optimized for 38.4 kBaud data rate. Note: The PLL loop filter component values in Table 13 (R2, R3, C6-C8) are optimized for 38.4 kBaud data rate. The SmartRF Studio software provides component values for other data rates using the equations on page 50. In the CC1020EMX reference design, which is also applicable for , LQG15HS series inductors from Murata have been used. The switch is SW-456 from M/A-COM. CC1021 SWRS045 Page 19 of 91 CC1021 The filter will reduce the emission of harmonics and the spurious emissions in the TX path as well as increase the receiver selectivity. The sensitivity will be slightly reduced due to the insertion loss of the LC filter. The LC filter in Figure 3 is inserted in the TX path only. The filter will reduce the emission of harmonics and the spurious emissions in the TX path. An alternative is to insert the LC filter between the antenna and the T/R switch as shown in Figure 4. AVDD=3V DVDD=3V 25 26 28 27 29 AGND AD_REF AVDD CHP_OUT AVDD PDI 30 2 DGND PCLK DVDD PSEL DVDD=3V 1 31 32 Microcontroller configuration interface and signal interface C6 R2 C7 VC 24 AVDD 23 AVDD 22 RF_OUT 21 AVDD 20 3 PDO 4 DGND 5 DVDD 6 DGND RF_IN 19 7 DCLK AVDD 18 8 DIO R_BIAS 17 CC1021 AVDD=3V R3 R10 C8 L2 Monopole antenna (50 Ohm) C60 C3 LC Filter AVDD=3V C1 T/R Switch AVDD=3V AVDD PA_EN LNA_EN AVDD AVDD XOSC_Q2 LOCK XOSC_Q1 L1 R1 15 16 13 14 12 11 10 9 AVDD=3V XTAL C4 C5 Figure 4. Alternative application circuit (power supply decoupling not shown) SWRS045 Page 20 of 91 CC1021 8. Configuration Overview can be configured to achieve the optimum performance for different applications. Through the programmable configuration registers the following key parameters can be programmed: CC1021 • Receive / transmit mode • RF output power • Frequency synthesizer key parameters: RF output frequency, FSK frequency 8.1. • • • • • • separation, crystal oscillator reference frequency Power-down / power-up mode Crystal oscillator power-up / powerdown Data rate and data format (NRZ, Manchester coded or UART interface) Synthesizer lock indicator mode Digital RSSI and carrier sense FSK / GFSK / OOK modulation Configuration Software Chipcon provides users of with a software program, SmartRF Studio (Windows interface) that generates all necessary configuration data based on the user's selections of various parameters. These hexadecimal numbers will then be the necessary input to the microcontroller for the configuration of CC1021 CC1021 . In addition, the program will provide the user with the component values needed for the input/output matching circuit, the PLL loop filter and the LC filter. CC1021 Figure 5 shows the user interface of the configuration software. CC1021 Figure 5. SmartRF Studio user interface Note: The CC1020/1070DK Development Kit with a fully assembled CC1020EMX Evaluation Module together with the specific software should be used for evaluation of the transceiver. CC1021 CC1021 SWRS045 Page 21 of 91 CC1021 9. Microcontroller Interface Used in a typical system, interface to a microcontroller. microcontroller must be able to: CC1021 The microcontroller pins connected to PDI, PDO and PCLK can be used for other purposes when the configuration interface is not used. PDI, PDO and PCLK are high impedance inputs as long as PSEL is not activated (active low). will This into different modes via • Program the 4-wire serial configuration interface (PDI, PDO, PCLK and PSEL) • Interface to the bi-directional synchronous data signal interface (DIO and DCLK) • Optionally, the microcontroller can do data encoding / decoding • Optionally, the microcontroller can monitor the LOCK pin for frequency lock status, carrier sense status or other status information. • Optionally, the microcontroller can read back the digital RSSI value and other status information via the 4-wire serial interface CC1021 PSEL has an internal pull-up resistor and should be left open (tri-stated by the microcontroller) or set to a high level during power down mode in order to prevent a trickle current flowing in the pullup. Signal interface A bi-directional pin is usually used for data (DIO) to be transmitted and data received. DCLK providing the data timing should be connected to a microcontroller input. As an option, the data output in receive mode can be made available on a separate pin. See section 9.2 on page for 25 further details. Configuration interface The microcontroller interface is shown in Figure 6. The microcontroller uses 3 or 4 I/O pins for the configuration interface (PDI, PDO, PCLK and PSEL). PDO should be connected to a microcontroller input. PDI, PCLK and PSEL must be microcontroller outputs. One I/O pin can be saved if PDI and PDO are connected together and a bi-directional pin is used at the microcontroller. PLL lock signal Optionally, one microcontroller pin can be used to monitor the LOCK signal. This signal is at low logic level when the PLL is in lock. It can also be used for carrier sense and to monitor other internal test signals. PCLK PDI PDO PSEL (Optional) Microcontroller DIO DCLK LOCK (Optional) Figure 6. Microcontroller interface SWRS045 Page 22 of 91 CC1021 9.1. 4-wire Serial Configuration Interface 14. The clocking of the data on PDI is done on the positive edge of PCLK. Data should be set up on the negative edge of PCLK by the microcontroller. When the last bit, D0, of the 8 data-bits has been loaded, the data word is loaded into the internal configuration register. is configured via a simple 4-wire SPI-compatible interface (PDI, PDO, is the PCLK and PSEL) where slave. There are 8-bit configuration registers, each addressed by a 7-bit address. A Read/Write bit initiates a read or write operation. A full configuration of requires sending 33 data frames of 16 bits each (7 address bits, R/W bit and 8 data bits). The time needed for a full configuration depends on the PCLK frequency. With a PCLK frequency of 10 MHz the full configuration is done in less than 53 µs. Setting the device in power down mode requires sending one frame only and will in this case take less than 2 µs. All registers are also readable. CC1021 CC1021 CC1021 The configuration data will be retained during a programmed power down mode, but not when the power supply is turned off. The registers can be programmed in any order. The configuration registers can also be read by the microcontroller via the same configuration interface. The seven address bits are sent first, then the R/W bit set low then to initiate the data read-back. returns the data from the addressed register. PDO is used as the data output and must be configured as an input by the microcontroller. The PDO is set at the negative edge of PCLK and should be sampled at the positive edge. The read operation is illustrated in Figure 8. During each write-cycle, 16 bits are sent on the PDI-line. The seven most significant bits of each data frame (A6:0) are the address-bits. A6 is the MSB (Most Significant Bit) of the address and is sent as the first bit. The next bit is the R/W bit (high for write, low for read). The 8 databits are then transferred (D7:0). During address and data transfer the PSEL (Program SELect) must be kept low. See Figure 7. CC1021 PSEL must be set high between each read/write operation. The timing for the programming is also shown in Figure 7 with reference to Table TSS THS TCL,min TCH,min THD TSD PCLK Address PDI 6 5 4 Write mode 3 2 1 0 W 7 Data byte 6 5 4 3 2 1 0 PDO PSEL Figure 7. Configuration registers write operation SWRS045 Page 23 of 91 CC1021 TSS THS TCL,min TCH,min PCLK Address PDI 6 5 4 Read mode 3 2 1 0 R Data byte PDO PSEL 7 6 5 4 3 2 1 0 TSH Figure 8. Configuration registers read operation Parameter Symbol Min Max Unit 10 MHz Conditions PCLK, clock frequency FPCLK PCLK low pulse duration TCL,min 50 ns The minimum time PCLK must be low. PCLK high pulse duration TCH,min 50 ns The minimum time PCLK must be high. PSEL setup time TSS 25 ns The minimum time PSEL must be low before positive edge of PCLK. PSEL hold time THS 25 ns The minimum time PSEL must be held low after the negative edge of PCLK. PSEL high time TSH 50 ns The minimum time PSEL must be high. PDI setup time TSD 25 ns The minimum time data on PDI must be ready before the positive edge of PCLK. PDI hold time THD 25 ns The minimum time data must be held at PDI, after the positive edge of PCLK. Rise time Trise 100 ns The maximum rise time for PCLK and PSEL Fall time Tfall 100 ns The maximum fall time for PCLK and PSEL Note: The setup and hold times refer to 50% of VDD. The rise and fall times refer to 10% / 90% of VDD. The maximum load that this table is valid for is 20 pF. Table 14. Serial interface, timing specification SWRS045 Page 24 of 91 CC1021 9.2. Signal Interface The can be used with NRZ (NonReturn-to-Zero) data or Manchester (also known as bi-phase-level) encoded data. can also synchronize the data from the demodulator and provide the data clock at DCLK. The data format is controlled by the DATA_FORMAT[1:0] bits in the MODEM register. CC1021 CC1021 can be configured different data formats: CC1021 for three Synchronous NRZ mode In transmit mode provides the data clock at DCLK and DIO is used as data at the input. Data is clocked into rising edge of DCLK. The data is modulated at RF without encoding. CC1021 CC1021 In receive mode performs the synchronization and provides received data clock at DCLK and data at DIO. The data should be clocked into the interfacing circuit at the rising edge of DCLK. See Figure 9. CC1021 Synchronous Manchester encoded mode In transmit mode provides the data clock at DCLK and DIO is used as data at the input. Data is clocked into rising edge of DCLK and should be in NRZ format. The data is modulated at RF with Manchester code. The encoding is done by . In this mode the effective bit rate is half the baud rate due to the coding. As an example, 19.2 kBaud Manchester encoded data corresponds to 9.6 kbps. CC1021 CC1021 CC1021 In receive mode performs the synchronization and provides received data clock at DCLK and data at DIO. performs the decoding and NRZ data is presented at DIO. The data should be clocked into the interfacing circuit at the rising edge of DCLK. See Figure 10. CC102 CC1021 In synchronous NRZ or Manchester mode the DCLK signal runs continuously both in RX and TX unless the DCLK signal is gated with the carrier sense signal or the PLL lock signal. Refer to section 21 for more details. If SEP_DI_DO = 0 in the INTERFACE register, the DIO pin is the data output in receive mode and data input in transmit mode. As an option, the data output can be made available at a separate pin. This is done by setting SEP_DI_DO = 1 in the INTERFACE register. Then, the LOCK pin will be used as data output in synchronous mode, overriding other use of the LOCK pin. Transparent Asynchronous UART mode In transmit mode DIO is used as data input. The data is modulated at RF without synchronization or encoding. In receive mode the raw data signal from the demodulator is sent to the output (DIO). No synchronization or decoding of and should be the signal is done in done by the interfacing circuit. CC1021 If SEP_DI_DO = 0 in the INTERFACE register, the DIO pin is the data output in receive mode and data input in transmit mode. The DCLK pin is not active and can be set to a high or low level by DATA_FORMAT[0]. If SEP_DI_DO = 1 in the INTERFACE register, the DCLK pin is the data output in receive mode and the DIO pin is the data input in transmit mode. In TX mode the DCLK pin is not active and can be set to a high or low level by DATA_FORMAT[0]. See Figure 11. Manchester encoding and decoding In the Synchronous Manchester encoded mode uses Manchester coding when modulating the data. The also performs the data decoding and synchronization. The Manchester code is based on transitions; a ì0î is encoded as a low-to-high transition, a ì1î is encoded as a high-to-low transition. See Figure 12. CC1021 CC1021 The Manchester code ensures that the signal has a constant DC component, which is necessary in some FSK demodulators. Using this mode also SWRS045 Page 25 of 91 CC1021 ensures compatibility with CC400/CC900 designs. Transmitter side: DCLK Clock provided by CC1021 DIO Data provided by microcontroller ì RFî FSK modulating signal (NRZ), internal in CC1021 Receiver side: ì RFî Demodulated signal (NRZ), internal in CC1021 DCLK Clock provided by CC1021 DIO Data provided by CC1021 Figure 9. Synchronous NRZ mode (SEP_DI_DO = 0) Transmitter side: DCLK Clock provided by CC1021 DIO Data provided by microcontroller ì RFî FSK modulating signal (Manchester encoded), internal in CC1021 Receiver side: ì RFî Demodulated signal (Manchester encoded), internal in CC1021 DCLK Clock provided by CC1021 DIO Data provided by CC1021 Figure 10. Synchronous Manchester encoded mode (SEP_DI_DO = 0) SWRS045 Page 26 of 91 CC1021 Transmitter side: DCLK is not used in transmit mode, and is used as data output in receive mode. It can be set to default high or low in transmit mode. DCLK DIO Data provided by UART (TXD) ì RFî FSK modulating signal, internal in CC1021 Receiver side: ì RFî Demodulated signal (NRZ), internal in CC1021 DCLK DCLK is used as data output provided by CC1021. Connect to UART (RXD) DIO DIO is not used in receive mode. Used only as data input in transmit mode Figure 11. Transparent Asynchronous UART mode (SEP_DI_DO = 1) 1 0 1 1 0 0 0 1 1 0 1 Tx data Time Figure 12. Manchester encoding 10. Data Rate Programming The data rate (baud rate) is programmable and depends on the crystal frequency and the programming of the CLOCK (CLOCK_A and CLOCK_B) registers. The baud rate (B.R) is given by B.R. = f xosc 8 ⋅ ( REF _ DIV + 1) ⋅ DIV 1 ⋅ DIV 2 MCLK_DIV2[1:0] 00 01 10 11 Table 15. DIV2 for different settings of MCLK_DIV2 MCLK_DIV1[2:0] 000 001 010 011 100 101 110 111 where DIV1 and DIV2 are given by the value of MCLK_DIV1 and MCLK_DIV2. Table 17 below shows some possible data rates as a function of crystal frequency in synchronous mode. In asynchronous transparent UART mode any data rate up to 153.6 kBaud can be used. DIV2 1 2 4 8 DIV1 2.5 3 4 7.5 12.5 40 48 64 Table 16. DIV1 for different settings of MCLK_DIV1 SWRS045 Page 27 of 91 CC1021 Data rate [kBaud] 0.45 0.5 0.6 0.9 1 1.2 1.8 2 2.4 3.6 4 4.096 4.8 7.2 8 8.192 9.6 14.4 16 16.384 19.2 28.8 32 32.768 38.4 57.6 64 65.536 76.8 115.2 128 153.6 4.9152 7.3728 X X X X X X X X X X X X X X X X X X X X X X X X X Crystal frequency [MHz] 9.8304 12.288 14.7456 X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X X 17.2032 19.6608 X X X X X X X X X X X X X X X X X X X X X X X X X X Table 17. Some possible data rates versus crystal frequency 11. Frequency Programming Programming the frequency word in the configuration registers sets the operation frequency. There are two frequency words registers, termed FREQ_A and FREQ_B, which can be programmed to two different frequencies. One of the frequency words can be used for RX (local oscillator frequency) and the other for TX (transmitting carrier frequency) in order to be able to switch very fast between RX mode and TX mode. They can also be used for RX (or TX) at two different channels. The F_REG bit in the MAIN register selects frequency word A or B. The frequency word is located in FREQ_2A:FREQ_1A:FREQ_0A and FREQ_2B:FREQ_1B:FREQ_0B for the FREQ_A and FREQ_B word respectively. The LSB of the FREQ_0 registers are used to enable dithering, section 11.1. The PLL output frequency is given by: 3 f c = f ref ⋅ 4 + FREQ + 0.5 ⋅ DITHER 32768 in the frequency band 402 ñ 470 MHz, and 3 f c = f ref ⋅ 2 + FREQ + 0.5 ⋅ DITHER 16384 in the frequency band 804 ñ 940 MHz. The BANDSELECT bit in the ANALOG register controls the frequency band used. BANDSELECT = 0 gives 402 - 470 MHz, and BANDSELECT = 1 gives 804 - 940 MHz. The reference frequency is the crystal oscillator clock frequency divided by REF_DIV (3 bits in the CLOCK_A or SWRS045 Page 28 of 91 CC1021 CLOCK_B register), a number between 1 and 7: f ref = f xosc REF _ DIV + 1 FSK frequency deviation is programmed in the DEVIATION register. The deviation programming is divided into a mantissa (TXDEV_M[3:0]) and an exponent (TXDEV_X[2:0]). Generally REF_DIV should be as low as possible but the following requirements must be met 9.8304 ≥ f ref f > c [MHz ] 256 in the frequency band 402 ñ 470 MHz, and 9.8304 ≥ f ref f > c [MHz ] 512 in the frequency band 804 - 940 MHz. The PLL output frequency equations above give the carrier frequency, fc , in transmit mode (centre frequency). The two FSK modulation frequencies are given by: f0 = fc fdev f1 = fc + fdev where fdev is set by the DEVIATION register: f dev = f ref ⋅ TXDEV _ M ⋅ 2 (TXDEV _ X −16 ) in the frequency band 402 ñ 470 MHz and f dev = f ref ⋅ TXDEV _ M ⋅ 2 (TXDEV _ X −15) in the frequency band 804 - 940 MHz. OOK (On-Off Keying) TXDEV_M[3:0] = 0000. is used if The TX_SHAPING bit in the DEVIATION register controls Gaussian shaping of the modulation signal. In receive mode the frequency must be programmed to be the LO frequency. Low side LO injection is used, hence: fLO = fc fIF where fIF is the IF frequency (ideally 307.2 kHz). 11.1. Dithering Spurious signals will occur at certain frequencies depending on the division ratios in the PLL. To reduce the strength of these spurs, a common technique is to use a dithering signal in the control of the frequency dividers. Dithering is activated by setting the DITHER bit in the FREQ_0 registers. It is recommended to use the dithering in order to achieve the best possible performance. SWRS045 Page 29 of 91 CC1021 12. Receiver 12.1. IF Frequency The IF frequency is derived from the crystal frequency as f IF = f xoscx 8 ⋅ ( ADC _ DIV [2 : 0] + 1) where ADC_DIV[2:0] is set in the MODEM register. The analog filter succeeding the mixer is used for wideband and anti-alias filtering which is important for the blocking performance at 1 MHz and larger offsets. This filter is fixed and centered on the nominal IF frequency of 307.2 kHz. The bandwidth of the analog filter is about 160 kHz. Using crystal frequencies which gives an IF frequency within 300 ñ 320 kHz means that the analog filter can be used (assuming low frequency deviations and low data rates). Large offsets, however, from the nominal IF frequency will give an un-symmetric filtering (variation in group delay and different attenuation) of the signal, resulting in decreased sensitivity and selectivity. See Application Note AN022 Crystal Frequency Selection for more details. For IF frequencies other than 300 ñ 320 kHz and for high frequency deviation and high data rates (typically ≥ 76.8 kBaud) the analog filter must be bypassed by setting FILTER_BYPASS = 1 in the FILTER register. In this case the blocking performance at 1 MHz and larger offsets will be degraded. The IF frequency is always the ADC clock frequency divided by 4. The ADC clock frequency should therefore be as close to 1.2288 MHz as possible. 12.2. Receiver Channel Filter Bandwidth In order to meet different channel spacing requirements, the receiver channel filter bandwidth is programmable. It can be programmed from 38.4 to 307.2 kHz. transmitting a 0-1-0 sequence. In both Manchester and NRZ mode 2·fm is then equal to the programmed baud rate. The equation for SBW can then be rewritten as The minimum receiver channel filter bandwidth depends on data rate, frequency separation and crystal tolerance. SBW = Baud rate + frequency separation The signal bandwidth must be smaller than the available receiver channel filter bandwidth. The signal bandwidth (SBW) can be approximated by (Carsonís rule): SBW = 2 · fm + 2 · frequency deviation where fm is the modulating signal. In Manchester mode the maximum modulating signal occurs when transmitting a continuous sequence of 0ís (or 1ís). In NRZ mode the maximum modulating signal occurs when Furthermore, the frequency offset of the transmitter and receiver must also be considered. Assuming equal frequency error in the transmitter and receiver (same type of crystal) the total frequency error is: f_error = ±2 · XTAL_ppm · f_RF where XTAL_ppm is the total accuracy of the crystal including initial tolerance, temperature drift, loading and ageing. f_RF is the RF operating frequency. The minimum receiver channel filter bandwidth (ChBW) can then be estimated as SWRS045 Page 30 of 91 CC1021 ChBW > SBW + 2 · f_error The DEC_DIV[2:0] bits in the FILTER register control the receiver channel filter bandwidth. The 6 dB bandwidth is given by: ChBW = 307.2 / (DEC_DIV + 1) [kHz] where the IF frequency is set to 307.2 kHz. Table 18 shows the available channel filter bandwidths. There is a tradeoff between selectivity as well as sensitivity and accepted frequency tolerance. In applications where larger frequency drift is expected, the filter bandwidth can be increased, but with reduced adjacent channel rejection (ACR) and sensitivity. Filter bandwidth [kHz] 38.4 43.9 51.2 61.4 76.8 102.4 153.6 307.2 FILTER.DEC_DIV[2:0] [decimal(binary)] 7 (111b) 6 (110b) 5 (101b) 4 (100b) 3 (011b) 2 (010b) 1 (001b) 0 (000b) Table 18. Channel filter bandwidth 12.3. Demodulator, Bit Synchronizer and Data Decision The block diagram for the demodulator, data slicer and bit synchronizer is shown in Figure 13. The built-in bit synchronizer synchronizes the internal clock to the incoming data and performs data decoding. The data decision is done using over-sampling and digital filtering of the incoming signal. This improves the reliability of the data transmission. Using the synchronous modes simplifies the data-decoding task substantially. The recommended preamble is a ë010101Ö í bit pattern. The same bit pattern should also be used in Manchester mode, giving a ë011001100110Ö ëchipí pattern. This is necessary for the bit synchronizer to synchronize to the coding correctly. The data slicer does the bit decision. Ideally the two received FSK frequencies are placed symmetrically around the IF frequency. However, if there is some frequency error between the transmitter and the receiver, the bit decision level should be adjusted accordingly. In CC1021 this is done automatically by measuring the two frequencies and use the average value as the decision level. The digital data slicer in CC1021 uses an average value of the minimum and maximum frequency deviation detected as the comparison level. The RXDEV_X[1:0] and RXDEV_M[3:0] in the AFC_CONTROL register are used to set the expected deviation of the incoming signal. Once a shift in the received frequency larger than the expected deviation is detected, a bit transition is recorded and the average value to be used by the data slicer is calculated. The minimum number of transitions required to calculate a slicing level is 3. That is, a 010 bit pattern (NRZ). The actual number of bits used for the averaging can be increased for better data decision accuracy. This is controlled by the SETTLING[1:0] bits in the AFC_CONTROL register. If RX data is present in the channel when the RX chain is turned on, then the data slicing estimate will usually give correct results after 3 bit transitions. The data slicing accuracy will increase after this, depending on the SETTLING[1:0] bits. If the start of transmission occurs after the RX chain has turned on, the minimum number of bit transitions (or preamble bits) before correct data slicing will depend on the SETTLING[1:0] bits. The automatic data slicer average value function can be disabled by setting SETTLING[1:0] = 00. In this case a symmetrical signal around the IF frequency is assumed. The internally calculated average FSK frequency value gives a measure for the frequency offset of the receiver compared SWRS045 Page 31 of 91 CC1021 to the transmitter. This information can also be used for an automatic frequency control (AFC) as described in section 12.13. Average filter Digital filtering Frequency detector Decimator Data filter Data slicer comparator Bit synchronizer and data decoder Figure 13. Demodulator block diagram 12.4. Receiver Sensitivity versus Data Rate and Frequency Separation The sensitivity is measured using the matching network shown in the application circuit in Figure 3, which includes an external T/R switch. The receiver sensitivity depends on the channel filter bandwidth, data rate, data format, FSK frequency separation and the RF frequency. Typical figures for the receiver sensitivity (BER = 10 3) are shown in Table 19 and Table 20 for FSK. For best performance, the frequency deviation should be at least half the baud rate in FSK mode. Refer to Application Note AN029 CC1020/1021 AFC for plots of sensitivity versus frequency offset. Data rate [kBaud] Deviation [kHz] Filter BW [kHz] 4.8 19.2 19.2 38.4 76.8 153.6 ± 4.95 ± 9.9 ± 19.8 ± 19.8 ± 36.0 ± 72.0 38.4 51.2 102.4 102.4 153.6 307.2 NRZ mode -109 -107 -104 -104 -101 -96 Sensitivity [dBm] Manchester mode -112 -108 -106 -104 -101 -97 UART mode -109 -107 -104 -104 -101 -96 Table 19. Typical receiver sensitivity as a function of data rate at 433 MHz, FSK modulation, BER = 10 3, pseudo-random data (PN9 sequence). Data rate [kBaud] Deviation [kHz] Filter BW [kHz] 4.8 19.2 19.2 38.4 76.8 153.6 ± 4.95 ± 9.9 ± 19.8 ± 19.8 ± 36.0 ± 72.0 38.4 51.2 102.4 102.4 153.6 307.2 NRZ mode -108 -107 -103 -103 -99 -94 Sensitivity [dBm] Manchester mode -111 -107 -106 -103 -100 -94 UART mode -108 -107 -103 -103 -99 -94 Table 20. Typical receiver sensitivity as a function of data rate at 868 MHz, FSK modulation, BER = 10 3, pseudo-random data (PN9 sequence). SWRS045 Page 32 of 91 CC1021 12.5. RSSI CC1021 has a built-in RSSI (Received Signal Strength Indicator) giving a digital value that can be read form the RSSI register. The RSSI reading must be offset and adjusted for VGA gain setting (VGA_SETTING[4:0] in the VGA3 register). The digital RSSI value is ranging from 0 to 106 (7 bits). The RSSI reading is a logarithmic measure of the average voltage amplitude after the digital filter in the digital part of the IF chain: RSSI = 4 log2(signal amplitude) The relative power is then given by RSSI x 1.5 dB in a logarithmic scale. The number of samples used to calculate the average signal amplitude is controlled by AGC_AVG[1:0] in the VGA2 register. The RSSI update rate is given by: f RSSI = f 2 filter _ clock AGC _ AVG [1:0 ]+1 where AGC_AVG[1:0] is set in the VGA2 register and f filter _ clock = 2 ⋅ ChBW . Maximum VGA gain is programmed by the VGA_SETTING[4:0] bits. The VGA gain is programmed in approximately 3 dB/LSB. The RSSI measurement can be referred to the power (absolute value) at the RF_IN pin by using the following equation: P = 1.5·RSSI - 3·VGA_SETTING RSSI_Offset [dBm] The RSSI_Offset depends on the channel filter bandwidth used due to different VGA settings. Figure 14 and Figure 15 show typical plots of RSSI reading as a function of input power for different channel filter bandwidths. Refer to Application Note AN030 CC1020/1021 RSSI for further details. The following method can be used to calculate the power P in dBm from the RSSI readout values in Figure 14 and Figure 15: P = 1.5·[RSSI ñ RSSI_ref] + P_ref where P is the output power in dBm for the current RSSI readout value. RSSI_ref is the RSSI readout value taken from Figure 14 or Figure 15 for an input power level of P_ref. Note that the RSSI readings in decimal value changes for different channel filter bandwidths. The analog filter has a finite dynamic range and is the reason why the RSSI reading is saturated at lower channel filter bandwidths. Higher channel filter bandwidths are typically used for high frequency deviation and data rates. The analog filter bandwidth is about 160 kHz and is bypassed for high frequency deviation and data rates and is the reason why the RSSI reading is not saturated for 153.6 kHz and 307.2 kHz channel filter bandwidths in Figure 14 and Figure 15. SWRS045 Page 33 of 91 CC1021 80 RSSI readout value [decimal] 70 60 50 40 30 20 10 0 -125 -115 -105 -95 -85 -75 -65 -55 -45 -35 -25 Input pow er level [dBm] 38.4 kHz 51.2 kHz 102.4 kHz 153.6 kHz 307.2 kHz Figure 14. Typical RSSI value vs. input power for different channel filter bandwidths, 433 MHz 80 RSSI readout value [decimal] 70 60 50 40 30 20 10 0 -125 -115 -105 -95 -85 -75 -65 -55 -45 -35 -25 Input pow er level [dBm] 38.4 kHz 51.2 kHz 102.4 kHz 153.6 kHz 307.2 kHz Figure 15. Typical RSSI value vs. input power for different channel filter bandwidths, 868 MHz SWRS045 Page 34 of 91 CC1021 12.6. Image Rejection Calibration For perfect image rejection, the phase and gain of the ìIî and ìQî parts of the analog RX chain must be perfectly matched. To improve the image rejection, the ìIî and ìQî phase and gain difference can be finetuned by adjusting the PHASE_COMP and GAIN_COMP registers. This allows compensation for process variations and other nonidealities. The calibration is done by injecting a signal at the image frequency, and adjusting the phase and gain difference for minimum RSSI value. During image rejection calibration, an unmodulated carrier should be applied at the image frequency (614.4 kHz below the desired channel), No signal should be present in the desired channel. The signal level should be 50 - 60 dB above the sensitivity in the desired channel, but the optimum level will vary from application to application. Too large input level gives poor results due to limited linearity in the analog IF chain, while too low input level gives poor results due to the receiver noise floor. For best RSSI accuracy, use AGC_AVG(1:0] = 11 during image rejection calibration (RSSI value is averaged over 16 filter output samples). The RSSI register update rate then equals the receiver channel bandwidth (set in FILTER register) divided by 8, as the filter output rate is twice the receiver channel bandwidth. This gives the minimum waiting time between RSSI register reads (0.5 ms is used below). Chipcon recommends the following image calibration procedure: 1. 2. 3. 4. 5. 6. 7. 8. Define 3 variables: XP = 0, XG = 0 and DX = 64. Go to step 3. Set DX = DX/2. Write XG to GAIN_COMP register. If XP+2∑DX < 127 then write XP+2∑DX to PHASE_COMP register else write 127 to PHASE_COMP register. Wait at least 3 ms. Measure signal strength Y4 as filtered average of 8 reads from RSSI register with 0.5 ms of delay between each RSSI read. Write XP+DX to PHASE_COMP register. Wait at least 3 ms. Measure signal strength Y3 as filtered average of 8 reads from RSSI register with 0.5 ms of delay between each RSSI read. Write XP to PHASE_COMP register. 9. Wait at least 3 ms. Measure signal strength Y2 as filtered average of 8 reads from RSSI register with 0.5 ms of delay between each RSSI read. 10. Write XP-DX to PHASE_COMP register. 11. Wait at least 3 ms. Measure signal strength Y1 as filtered average of 8 reads from RSSI register with 0.5 ms of delay between each RSSI read. 12. Write XP-2∑DX to PHASE_COMP register. 13. Wait at least 3 ms. Measure signal strength Y0 as filtered average of 8 reads from RSSI register with 0.5 ms of delay between each RSSI read. 14. Set AP = 2∑(Y0-Y2+Y4) - (Y1+Y3). 15. If AP > 0 then set DP = ROUND( 7∑DX∑(2∑(Y0-Y4)+Y1Y3) / (10∑AP) ) else if Y0+Y1 > Y3+Y4 then set DP = DX else set DP = -DX. 16. If DP > DX then set DP = DX else if DP < -DX then set DP = -DX. 17. Set XP = XP+DP. 18. Write XP to PHASE_COMP register. 19. If XG+2∑DX < 127 then write XG+2∑DX to GAIN_COMP register else write 127 to GAIN_COMP register. 20. Wait at least 3 ms. Measure signal strength Y4 as filtered average of 8 reads from RSSI register with 0.5 ms of delay between each RSSI read. 21. Write XG+DX to GAIN_COMP register. 22. Wait at least 3 ms. Measure signal strength Y3 as filtered average of 8 reads from RSSI register with 0.5 ms of delay between each RSSI read. 23. Write XG to GAIN_COMP register. 24. Wait at least 3 ms. Measure signal strength Y2 as filtered average of 8 reads from RSSI register with 0.5 ms of delay between each RSSI read. 25. Write XG-DX to GAIN_COMP register. 26. Wait at least 3 ms. Measure signal strength Y1 as filtered average of 8 reads from RSSI register with 0.5 ms of delay between each RSSI read. 27. Write XG-2∑DX to GAIN_COMP register. 28. Wait at least 3 ms. Measure signal strength Y0 as filtered average of 8 reads from RSSI register with 0.5 ms of delay between each RSSI read. 29. Set AG = 2∑(Y0-Y2+Y4) - (Y1+Y3). 30. If AG > 0 then set DG = ROUND( 7∑DX∑(2∑(Y0-Y4)+Y1Y3) / (10∑AG) ) else if Y0+Y1 > Y3+Y4 then set DG = DX else set DG = -DX. 31. If DG > DX then set DG = DX else if DG < -DX then set DG = -DX. 32. Set XG = XG+DG. 33. If DX > 1 then go to step 2. 34. Write XP to PHASE_COMP register and XG to GAIN_COMP register. If repeated calibration gives varying results, try to change the input level or SWRS045 Page 35 of 91 CC1021 increase the number of RSSI reads N. A good starting point is N=8. As accuracy is more important in the last fine-calibration steps, it can be worthwhile to increase N for each loop iteration. bypassed by setting FILTER_BYPASS = 1 in the FILTER register. In this case the image rejection is degraded. The image rejection is reduced for low supply voltages (typically <2.5 V) when operating in the 402 ñ 470 MHz frequency range. For high frequency deviation and high data rates (typically ≥ 76.8 kBaud) the analog filter succeeding the mixer must be 12.7. Blocking and Selectivity and 19.2 kBaud data rate. Figure 19 shows the blocking/selectivity for 102.4 kHz channel filter bandwidth at 868 MHz and 38.4 kBaud data rate. The blocking rejection is the ratio between a blocker (interferer) and a wanted signal 3 dB above the sensitivity limit. Figure 16 shows the blocking/selectivity for 102.4 kHz channel filter bandwidth at 433 MHz and 19.2 kBaud data rate. Figure 17 shows the blocking/selectivity for 102.4 kHz channel filter bandwidth at 433 MHz and 38.4 kBaud data rate. Figure 18 shows the blocking/selectivity for 102.4 kHz channel filter bandwidth at 868 MHz 65 60 55 50 45 40 Blocker rejection [dB] 35 30 25 20 15 10 5 0 -5 -10 -15 -1000 -900 -800 -700 -600 -500 -400 -300 -200 -100 0 100 200 300 400 500 600 700 800 900 1000 Blocke r fre que ncy offset [kHz] Not image calibrated Image calibrated Figure 16. Typical blocker rejection. Carrier frequency set to 434.3072 MHz (102.4 kHz channel filter bandwidth, 19.2 kBaud) SWRS045 Page 36 of 91 CC1021 65 60 55 50 45 40 Blocker rejection [dB] 35 30 25 20 15 10 5 0 -5 -10 -15 -1000 -900 -800 -700 -600 -500 -400 -300 -200 -100 0 100 200 300 400 500 600 700 800 900 1000 Blocke r fre que ncy offset [kHz] Not image calibrated Image calibrated Figure 17. Typical blocker rejection. Carrier frequency set to 434.3072 MHz (102.4 kHz channel filter bandwidth, 38.4 kBaud) 65 60 55 50 45 40 Blocker rejection [dB] 35 30 25 20 15 10 5 0 -5 -10 -15 -1000 -900 -800 -700 -600 -500 -400 -300 -200 -100 0 100 200 300 400 500 600 700 800 900 1000 Blocke r fre quency offse t [kHz] Not image calibrated Image calibrated Figure 18. Typical blocker rejection. Carrier frequency set to 868.3072 MHz (102.4 kHz channel filter bandwidth, 19.2 kBaud) SWRS045 Page 37 of 91 CC1021 65 60 55 50 45 40 Blocker rejection [dB] 35 30 25 20 15 10 5 0 -5 -10 -15 -1000 -900 -800 -700 -600 -500 -400 -300 -200 -100 0 100 200 300 400 500 600 700 800 900 1000 Blocke r fre quency offse t [kHz] Not image calibrated Image calibrated Figure 19. Typical blocker rejection. Carrier frequency set to 868.3072 MHz (102.4 kHz channel filter bandwidth, 38.4 kBaud) 12.8. Linear IF Chain and AGC Settings is based on a linear IF chain where the signal amplification is done in an analog VGA (Variable Gain Amplifier). The gain is controlled by the digital part of the IF chain after the ADC (Analog to Digital Converter). The AGC (Automatic Gain Control) loop ensures that the ADC operates inside its dynamic range by using an analog/digital feedback loop. CC1021 The maximum VGA gain is programmed by the VGA_SETTING[4:0] in the VGA3 register. The VGA gain is programmed in approximately 3 dB/LSB. The VGA gain should be set so that the amplified thermal noise from the front-end balance the quantization noise from the ADC. Therefore the optimum maximum VGA gain setting will depend on the channel filter bandwidth. A digital RSSI is used to measure the signal strength after the ADC. The CS_LEVEL[4:0] in the VGA4 register is used to set the nominal operating point of the gain control (and also the carrier sense level). Further explanation can be found in Figure 20. The VGA gain will be changed according to a threshold set by the VGA_DOWN[2:0] in the VGA3 register and the VGA_UP[2:0] in the VGA4 register. Together, these two values specify the signal strength limits used by the AGC to adjust the VGA gain. To avoid unnecessary tripping of the VGA, an extra hysteresis and filtering of the RSSI samples can be added. The AGC_HYSTERESIS bit in the VGA2 register enables this. The time dynamics of the loop can be altered by the VGA_BLANKING bit in the ANALOG register, and VGA_FREEZE[1:0] and VGA_WAIT[2:0] bits in the VGA1 register. When VGA_BLANKING is activated, the VGA recovery time from DC offset spikes after a gain step is reduced. VGA_FREEZE determines the time to hold bit synchronization, VGA and RSSI levels after one of these events occur: • • RX power-up The PLL has been out of lock SWRS045 Page 38 of 91 CC1021 • Frequency register setting is switched between A and B This feature is useful to avoid AGC operation during start-up transients and to ensure minimum dwell time using frequency hopping. This means that bit synchronization can be maintained from hop to hop. VGA_WAIT determines the time to hold the present bit synchronization and RSSI levels after changing VGA gain. This feature is useful to avoid AGC operation during the settling of transients after a VGA gain change. Some transients are expected due to DC offsets in the VGA. At the sensitivity limit, the VGA gain is set by VGA_SETTING. In order to optimize selectivity, this gain should not be set higher than necessary. The SmartRF Studio software gives the settings for VGA1 ñ VGA4 registers. For reference, the following method can be used to find the AGC settings: 1. Disable AGC and use maximum LNA2 gain by writing BFh to the VGA2 register. Set minimum VGA gain by writing to the VGA3 register with VGA_SETTING = 0. 2. Apply no RF input signal, and measure ADC noise floor by reading the RSSI register. 3. Apply no RF input signal, and write VGA3 register with increasing VGA_SETTING value until the RSSI register value is approximately 4 larger than the value read in step 2. This places the front-end noise floor around 6 dB above the ADC noise floor. 4. Apply an RF signal with strength equal the desired carrier sense threshold. The RF signal should preferably be modulated with correct Baud rate and deviation. Read the RSSI register value, subtract 8, and write to CS_LEVEL in the VGA4 register. Vary the RF signal level slightly and check that carrier sense indication (bit 3 in STATUS register) switches at the desired input level. 5. If desired, adjust the VGA_UP and VGA_DOWN settings according to the explanation in Figure 20. 6. Enable AGC and select LNA2 gain change level. Write 55h to VGA2 register if the resulting VGA_SETTING>10. Otherwise, write 45h to VGA2. Modify AGC_AVG in the above VGA2 value if faster carrier sense and AGC settling is desired. RSSI Level Note that the AGC works with "raw" filter output signal strength, while the RSSI readout value is compensated for VGA gain changes by the AGC. The AGC keeps the signal strength in this range. Minimize VGA_DOWN for best selectivity, but leave some margin to avoid frequent VGA gain changes during reception. The AGC keeps the signal strength above carrier sense level + VGA_UP. Minimize VGA_UP for best selectivity, but increase if first VGA gain reduction occurs too close to the noise floor. (signal strength, 1.5dB/step) AGC decreases gain if above this level (unless at minimum). VGA_DOWN+3 AGC increases gain if below this level (unless at maximum). VGA_UP Carrier sense is turned on here. To set CS_LEVEL, subtract 8 from RSSI readout with RF input signal at desired carrier sense level. Zero level depends on front-end settings and VGA_SETTING value. CS_LEVEL+8 0 Figure 20. Relationship between RSSI, carrier sense level, and AGC settings CS_LEVEL, VGA_UP and VGA_DOWN SWRS045 Page 39 of 91 CC1021 12.9. AGC Settling After turning on the RX chain, the following occurs: at minimum, then the VGA gain is reduced and the AGC continues from B). A) The AGC waits 16-128 ADC_CLK (1.2288 MHz) periods, depending on the VGA_FREEZE setting in the VGA1 register, for settling in the analog parts. E) If the RSSI value is too low according to the CS_LEVEL and VGA_UP settings, and the VGA gain is not already at maximum (given by VGA_SETTING), then the VGA gain is increased and the AGC continues from B). B) The AGC waits 16-48 FILTER_CLK periods, depending on the VGA_WAIT setting in the VGA1 register, for settling in the analog parts and the digital channel filter. 2-3 VGA gain changes should be expected before the AGC has settled. Increasing AGC_AVG increases the settling time, but may be worthwhile if there is the time in the protocol, and for reducing false wake-up events when setting the carrier sense close to the noise floor. C) The AGC calculates the RSSI value as the average magnitude over the next 2-16 FILTER_CLK periods, depending of the AGC_AVG setting in the VGA2 register. The AGC settling time depends on the FILTER_CLK (= 2·ChBW). Thus, there is a trade off between AGC settling time and receiver sensitivity because the AGC settling time can be reduced for data rates lower than 76.8 kBaud by using a wider receiver channel filter bandwidth (i.e. larger ChBW). D) If the RSSI value is higher than CS_LEVEL+8, then the carrier sense indicator is set (if CS_SET = 0). If the RSSI value is too high according to the CS_LEVEL, VGA_UP and VGA_DOWN settings, and the VGA gain is not already 12.10. Preamble Length and Sync Word The rules for choosing a good sync word are as follows: 1. The sync word should be significantly different from the preamble 2. A large number of transitions is good for the bit synchronization or clock recovery. Equal bits reduce the number of transitions. The recommended sync word has at the most 3 equal bits in a row. 3. Autocorrelation. The sync word should not repeat itself, as this will increase the likelihood for errors. 4. In general the first bit of sync should be opposite of last bit in preamble, to achieve one more transition. The recommended sync words for are 2 bytes (D391), 3 bytes (D391DA) or 4 bytes (D391DA26) and are selected as the best compromise of the above criteria. CC1021 Using the register settings provided by the SmartRF Studio software, packet error rates (PER) less than 0.5% can be achieved when using 24 bits of preamble and a 16 bit sync word (D391). Using a preamble longer than 24 bits will improve the PER. When performing the PER measurements described above the packet format consisted of 10 bytes of random data, 2 bytes CRC and 1 dummy byte in addition to the sync word and preamble at the start of each package. For the test 1000 packets were sent 10 times. The transmitter was put in power down between each packet. Any bit error in the packet, either in the sync word, in the data or in the CRC caused the packet to be counted as a failed packet. SWRS045 Page 40 of 91 CC1021 12.11. Carrier Sense CS_LEVEL[4:0] sets the threshold for this specific VGA_SETTING[4:0] value. If the VGA_SETTING[4:0] is changed, the CS_LEVEL[4:0] must be changed accordingly to maintain the same absolute carrier sense threshold. See Figure 20 for an explanation of the relationship between RSSI, AGC and carrier sense settings. The carrier sense signal is based on the RSSI value and a programmable threshold. The carrier sense function can be used to simplify the implementation of a CSMA (Carrier Sense Multiple Access) medium access protocol. Carrier sense threshold level is programmed by CS_LEVEL[4:0] in the VGA4 register and VGA_SETTING[4:0] in the VGA3 register. The carrier sense signal can be read as the CARRIER_SENSE bit in the STATUS register. VGA_SETTING[4:0] sets the maximum gain in the VGA. This value must be set so that the ADC works with optimum dynamic range for a certain channel filter bandwidth. The detected signal strength (after the ADC) will therefore depend on this setting. The carrier sense signal can also be made available at the LOCK pin by setting LOCK_SELECT[3:0] = 0100 in the LOCK register. 12.12. Automatic Power-up Sequencing has a built-in automatic power-up sequencing state machine. By setting the into this mode, the receiver can be powered-up automatically by a wake-up signal and will then check for a carrier signal (carrier sense). If carrier sense is not detected, it returns to power-down mode. A flow chart for automatic power-up sequencing is shown in Figure 21. CC1021 CC1021 The automatic power-up sequencing mode is selected when PD_MODE[1:0] = 11 in the MAIN register. When the automatic power-up sequencing mode is selected, the functionality of the MAIN register is changed and used to control the sequencing. By setting SEQ_PD = 1 in the MAIN is set in power down register, mode. If SEQ_PSEL = 1 in the SEQUENCING register the automatic power-up sequence is initiated by a negative transition on the PSEL pin. CC1021 If SEQ_PSEL = 0 in the SEQUENCING register, then the automatic power-up sequence is initiated by a negative transition on the DIO pin (as long as SEP_DI_DO = 1 in the INTERFACE register). Sequence timing is controlled through RX_WAIT[2:0] and CS_WAIT[3:0] in the SEQUENCING register. VCO and PLL calibration can also be done automatically as a part of the sequence. This is controlled through SEQ_CAL[1:0] in the MAIN register. Calibration can be th done every time, every 16 sequence, th every 256 sequence, or never. See the register description for details. A description of when to do, and how the VCO and PLL self-calibration is done, is given in section 15.2 on page 51. SWRS045 Page 41 of 91 CC1021 Turn on crystal oscillator/bias Frequency synthesizer off Receive chain off Sequencing wake-up event (negative transition on PSEL pin or DIO pin) Power down Crystal oscillator and bias off Frequency synthesizer off Receive chain off Crystal oscillator and bias on Turn on frequency synthesizer Receive chain off Wait for PLL lock or timeout, 127 filter clocks PLL timeout Set SEQ_ERROR flag in STATUS register Optional calibration Programmable: each time, once in 16, or once in 256 Receive chain off PLL in lock Optional waiting time before turning on receive chain Programmable: 32-256 ADC clocks Crystal oscillator and bias on Frequency synthesizer on Turn on receive chain Wait for carrier sense or timeout Programmable: 20-72 filter clocks Carrier sense timeout Carrier sense Receive mode Sequencing power-down event Crystal oscillator and bias on Frequency synthesizer on (Positive transition on SEQ_PD in MAIN register) Receive chain on Figure 21. Automatic power-up sequencing flow chart Notes to Figure 21: Filter clock (FILTER_CLK): ADC clock (ADC_CLK): f ADC = f filter _ clock = 2 ⋅ ChBW where ChBW is defined on page 30. f xoscx 2 ⋅ ( ADC _ DIV [2 : 0] + 1) where ADC_DIV[2:0] is set in the MODEM register. 12.13. Automatic Frequency Control has a built-in feature called AFC (Automatic Frequency Control) that can be used to compensate for frequency drift. CC1021 The average frequency offset of the received signal (from the nominal IF frequency) can be read in the AFC register. The signed (2ís-complement) 8bit value AFC[7:0] can be used to compensate for frequency offset between transmitter and receiver. The frequency offset is given by: ∆F = AFC·Baud rate / 16 The receiver can be calibrated against the transmitter by changing the operating frequency according to the measured SWRS045 Page 42 of 91 CC1021 offset. The new frequency must be calculated and written to the FREQ register by the microcontroller. The AFC can be used for an FSK/GFSK signal, but not for OOK. Application Note AN029 CC1020/1021 AFC provides the procedure and equations necessary to implement AFC. The AFC feature reduces the crystal accuracy requirement. 12.14. Digital FM It is possible to read back the instantaneous IF from the FM demodulator as a frequency offset from the nominal IF frequency. This digital value can be used to perform a pseudo analog FM demodulation. The frequency offset can be read from the GAUSS_FILTER register and is a signed 8-bit value coded as 2-complement. The instantaneous deviation is given by: F = GAUSS_FILTER·Baud rate / 8 The digital value should be read from the register and sent to a DAC and filtered in order to get an analog audio signal. The internal register value is updated at the MODEM_CLK rate. MODEM_CLK is available at the LOCK pin when LOCK_SELECT[3:0] = 1101 in the LOCK register, and can be used to synchronize the reading. For audio (300 ñ 4000 Hz) the sampling rate should be higher than or equal to 8 kHz (Nyquist) and is determined by the MODEM_CLK. The MODEM_CLK, which is the sampling rate, equals 8 times the baud rate. That is, the minimum baud rate, which can be programmed, is 1 kBaud. However, the incoming data will be filtered in the digital domain and the 3-dB cut-off frequency is 0.6 times the programmed Baud rate. Thus, for audio the minimum programmed Baud rate should be approximately 7.2 kBaud. The GAUSS_FILTER resolution decreases with increasing baud rate. A accumulate and dump filter can be implemented in the uC to improve the resolution. Note that each GAUSS_FILTER reading should be synchronized to the MODEM_CLK. As an example, accumulating 4 readings and dividing the total by 4 will improve the resolution by 2 bits. Furthermore, to fully utilize the GAUSS_FILTER dynamic range the frequency deviation must be 16 times the programmed baud rate. SWRS045 Page 43 of 91 CC1021 13. Transmitter 13.1. FSK Modulation Formats The data modulator can modulate FSK, which is a two level FSK (Frequency Shift Keying), or GFSK, which is a Gaussian filtered FSK with BT = 0.5. The purpose of the GFSK is to make a more bandwidth efficient system. The modulation and the Gaussian filtering are done internally in the chip. The TX_SHAPING bit in the DEVIATION register enables the GFSK. Figure 22 shows a typical eye diagram for 153.6 kBaud data rate at 868 MHz operation. Figure 22. GFSK eye diagram. 153.6 kBaud, NRZ, ±79.2 kHz frequency deviation. 13.2. Output Power Programming The RF output power from the device is programmable by the 8-bit PA_POWER register. Figure 23 and Figure 24 shows the output power and total current consumption as a function of the PA_POWER register setting. It is more efficient in terms of current consumption to use either the lower or upper 4-bits in the register to control the power, as shown in the figures. However, the output power can be controlled in finer steps using all the available bits in the PA_POWER register. SWRS045 Page 44 of 91 CC1021 35.0 Current [mA] / Output power [dBm] 30.0 25.0 20.0 15.0 10.0 5.0 0.0 -5.0 -10.0 -15.0 -20.0 -25.0 0 1 2 3 4 5 6 7 8 9 0A 0B 0C 0D 0E 0F 50 60 70 80 90 A0 B0 C0 D0 E0 F0 FF PA_POW ER [he x] Current Consumption Output Power Figure 23. Typical output power and current consumption, 433 MHz 35.0 Current [mA] / Output power [dBm] 30.0 25.0 20.0 15.0 10.0 5.0 0.0 -5.0 -10.0 -15.0 -20.0 -25.0 0 1 2 3 4 5 6 7 8 9 0A 0B 0C 0D 0E 0F 50 60 70 80 90 A0 B0 C0 D0 E0 F0 FF PA_POW ER [hex ] Current Consumption Output Power Figure 24. Typical output power and current consumption, 868 MHz 13.3. TX Data Latency The transmitter will add a delay due to the synchronization of the data with DCLK and further clocking into the modulator. The user should therefore add a delay equivalent to at least 2 bits after the data payload has been transmitted before switching off the PA (i.e. before stopping the transmission). SWRS045 Page 45 of 91 CC1021 13.4. Reducing Spurious Emission and Modulation Bandwidth Modulation bandwidth and spurious emission are normally measured with the PA continuously on and a repeated test sequence. In cases where the modulation bandwidth and spurious emission are measured with the switching from power down mode to TX mode, a PA ramping sequence could be used to minimize modulation bandwidth and spurious emission. CC1021 14. PA ramping should then be used both when switching the PA on and off. A linear PA ramping sequence can be used where register PA_POWER is changed from 00h to 0Fh and then from 50h to the register setting that gives the desired output power (e.g. F0h for +10 dBm output power at 433 MHz operation). The longer the time per PA ramping step the better, but setting the total PA ramping time equal to 2 bit periods is a good compromise between performance and PA ramping time. Input / Output Matching and Filtering When designing the impedance matching the circuit must be network for the matched correctly at the harmonic frequencies as well as at the fundamental tone. A recommended matching network is shown in Figure 25. Component values for various frequencies are given in Table 21. Component values for other frequencies can be found using the SmartRF Studio software. CC1021 As can be seen from Figure 25 and Table 21, the 433 MHz network utilizes a T-type filter, while the 868/915 MHz network has a π-type filter topology. It is important to remember that the physical layout and the components used contribute significantly to the reflection coefficient, especially at the higher harmonics. For this reason, the frequency response of the matching network should Item C1 C3 C60 C71 C72 L1 L2 L70 L71 R10 433 MHz 10 pF, 5%, NP0, 0402 5.6 pF, 5%, NP0, 0402 220 pF, 5%, NP0, 0402 DNM 4.7 pF, 5%, NP0, 0402 33 nH, 5%, 0402 22 nH, 5%, 0402 47 nH, 5%, 0402 39 nH, 5%, 0402 82 Ω, 5%, 0402 be measured and compared to the response of the Chipcon reference design. Refer to Figure 27 and Table 22 as well as Figure 28 and Table 23. The use of an external T/R switch reduces current consumption in TX for high output power levels and improves the sensitivity in RX. A recommended application circuit is available from the Chipcon web site (CC1020EMX). The external T/R switch can be omitted in certain applications, but performance will then be degraded. The match can also be tuned by a shunt capacitor array at the PA output (RF_OUT). The capacitance can be set in 0.4 pF steps and used either in RX mode or TX mode. The RX_MATCH[3:0] and TX_MATCH[3:0] bits in the MATCH register control the capacitor array. 868 MHz 47 pF, 5%, NP0, 0402 10 pF, 5%, NP0, 0402 220 pF, 5%, NP0, 0402 8.2 pF 5%, NP0, 0402 8.2 pF 5%, NP0, 0402 82 nH, 5%, 0402 3.6 nH, 5%, 0402 5.1 nH, 5%, 0402 0 Ω resistor, 0402 82 Ω, 5%, 0402 915 MHz 47 pF, 5%, NP0, 0402 10 pF, 5%, NP0, 0402 220 pF, 5%, NP0, 0402 8.2 pF 5%, NP0, 0402 8.2 pF 5%, NP0, 0402 82 nH, 5%, 0402 3.6 nH, 5%, 0402 5.1 nH, 5%, 0402 0 Ω resistor, 0402 82 Ω, 5%, 0402 Table 21. Component values for the matching network described in Figure 25. (DNM = Do Not Mount) SWRS045 Page 46 of 91 CC1021 AVDD=3V R10 CC1021 ANTENNA L2 C60 L70 C3 RF_OUT RF_IN C71 L71 C72 C1 L1 T/R SWITCH Figure 25. Input/output matching network Figure 26. Typical LNA input impedance, 200 ñ 1000 MHz SWRS045 Page 47 of 91 CC1021 433 M H z Figure 27. Typical optimum PA load impedance, 433 MHz. The frequency is swept from 300 MHz to 2500 MHz. Values are listed in Table 22 Frequency (MHz) Real (Ohms) 433 54 Imaginary (Ohms) 44 866 20 173 1299 288 -563 1732 14 -123 2165 5 -66 Table 22. Impedances at the first 5 harmonics (433 MHz matching network) SWRS045 Page 48 of 91 CC1021 868 M H z Figure 28: Typical optimum PA load impedance, 868/915 MHz. The frequency is swept from 300 MHz to 2800 MHz. Values are listed in Table 23 Frequency (MHz) Real (Ohms) Imaginary (Ohms) 868 15 24 915 20 35 1736 1.5 18 1830 1.7 22 2604 3.2 44 2745 3.6 45 Table 23. Impedances at the first 3 harmonics (868/915 MHz matching network) SWRS045 Page 49 of 91 CC1021 15. Frequency Synthesizer 15.1. VCO, Charge Pump and PLL Loop Filter The VCO is completely integrated and operates in the 1608 ñ 1880 MHz range. A frequency divider is used to get a frequency in the UHF range (402 ñ 470 and 804 ñ 940 MHz). The BANDSELECT bit in the ANALOG register selects the frequency band. The VCO frequency is given by: f VCO = f ref ⋅ 3 + FREQ + 0.5 ⋅ DITHER 8192 The VCO frequency is divided by 2 and by 4 to generate frequencies in the two bands, respectively. The VCO sensitivity (sometimes referred to as VCO gain) varies over frequency and operating conditions. Typically the VCO sensitivity varies between 12 and 36 MHz/V. For calculations the geometrical mean at 21 MHz/V can be used. The PLL calibration (explained below) measures the actual VCO sensitivity and adjusts the charge pump current accordingly to achieve correct PLL loop gain and bandwidth (higher charge pump current when VCO sensitivity is lower). The following equations can be used for calculating PLL loop filter component values, see Figure 3, for a desired PLL loop bandwidth, BW: 2 C7 = 3037 (fref / BW ) ñ7 R2 = 7126 (BW / fref) 2 C6 = 80.75 (fref / BW ) R3 = 21823 (BW / fref) C8 = 839 (fref / BW 2) ñ6 [pF] [kΩ] [nF] [kΩ] [pF] Define a minimum PLL loop bandwidth as BW min = 80.75 ⋅ f ref 220 . If BWmin > Baud rate/3 then set BW = BW min and if BW min < Baud rate/3 then set BW = Baud rate/3 in the above equations. There is one special case when using the recommended 14.7456 MHz crystal: If the data rate is 4.8 kBaud or below the following loop filter components are recommended: C6 = 100 nF C7 = 3900 pF C8 = 1000 pF R2 = 2.2 kΩ R3 = 6.8 kΩ After calibration the PLL bandwidth is set by the PLL_BW register in combination with the external loop filter components calculated above. The PLL_BW can be found from PLL_BW = 174 + 16 log2(fref /7.126) where fref is the reference frequency (in MHz). The PLL loop filter bandwidth increases with increasing PLL_BW setting. After calibration the applied charge pump current (CHP_CURRENT[3:0]) can be read in the STATUS1 register. The charge pump current is approximately given by: I CHP = 16 ⋅ 2 CHP _ CURRENT 4 [uA] The combined charge pump and phase detector gain (in A/rad) is given by the charge pump current divided by 2π. The PLL bandwidth will limit the maximum modulation frequency and hence data rate. SWRS045 Page 50 of 91 CC1021 15.2. VCO and PLL Self-Calibration To compensate for supply voltage, temperature and process variations, the VCO and PLL must be calibrated. The calibration is performed automatically and sets the maximum VCO tuning range and optimum charge pump current for PLL stability. After setting up the device at the operating frequency, the self-calibration can be initiated by setting the CAL_START bit in the CALIBRATE register. The calibration result is stored internally in the chip, and is valid as long as power is not turned off. If large supply voltage drops (typically more than 0.25 V) or temperature variations (typically more than 40oC) occur after calibration, a new calibration should be performed. The nominal VCO control voltage is set by the CAL_ITERATE[2:0] bits in the CALIBRATE register. The CAL_COMPLETE bit in the STATUS register indicates that calibration has finished. The calibration wait time (CAL_WAIT) is programmable and is proportional to the internal PLL reference frequency. The highest possible reference frequency should be used to get the minimum calibration time. It is recommended to use CAL_WAIT[1:0] = 11 in order to get the most accurate loop bandwidth. Calibration time [ms] CAL_WAIT 00 01 10 11 Reference frequency [MHz] 1.8432 49 ms 60 ms 71 ms 109 ms 7.3728 12 ms 15 ms 18 ms 27 ms 9.8304 10 ms 11 ms 13 ms 20 ms Table 24. Typical calibration times The CAL_COMPLETE bit can also be monitored at the LOCK pin, configured by LOCK_SELECT[3:0] = 0101, and used as an interrupt input to the microcontroller. To check that the PLL is in lock the user should monitor the LOCK_CONTINUOUS bit in the STATUS register. The LOCK_CONTINUOUS bit can also be monitored at the LOCK pin, configured by LOCK_SELECT[3:0] = 0010. There are separate calibration values for the two frequency registers. However, dual calibration is possible if all of the below conditions apply: • • • The two frequencies A and B differ by less than 1 MHz Reference frequencies are equal (REF_DIV_A[2:0] = REF_DIV_B[2:0] in the CLOCK_A/CLOCK_B registers) VCO currents are equal (VCO_CURRENT_A[3:0] = VCO_CURRENT_B[3:0] in the VCO register). The CAL_DUAL bit in the CALIBRATE register controls dual or separate calibration. The single calibration algorithm (CAL_DUAL=0) using separate calibration for RX and TX frequency is illustrated in Figure 29. The same algorithm is applicable for dual calibration if CAL_DUAL=1. Application Note AN023 CC1020 MCU Interfacing, available from the Chipcon web site, includes example source code for single calibration. Chipcon recommends that single calibration be used for more robust operation. There is a finite possibility that the PLL self-calibration will fail. The calibration routine in the source code should include a loop so that the PLL is re-calibrated until PLL lock is achieved if the PLL does not lock the first time. Refer to Errata Note 002. CC1021 SWRS045 Page 51 of 91 CC1021 Start single calibration fref is the reference frequency (in MHz) Write FREQ_A, FREQ_B, VCO, CLOCK_A and CLOCK_B registers. PLL_BW = 174 + 16log2(fref/7.126) Calibrate RX frequency register A (to calibrate TX frequency register B write MAIN register = D1h). Register CALIBRATE = 34h Write MAIN register = 11h: RXTX=0, F_REG=0, PD_MODE=1, FS_PD=0, CORE_PD=0, BIAS_PD=0, RESET_N=1 Start calibration Write CALIBRATE register = B4h Wait for T≥100 us Read STATUS register and wait until CAL_COMPLETE=1 Read STATUS register and wait until LOCK_CONTINUOUS=1 Calibration OK? No Yes End of calibration Figure 29. Single calibration algorithm for RX and TX 15.3. PLL Turn-on Time versus Loop Filter Bandwidth If calibration has been performed the PLL turn-on time is the time needed for the PLL to lock to the desired frequency when going from power down mode (with the crystal oscillator running) to TX or RX mode. The PLL turn-on time depends on the PLL loop filter bandwidth. Table 25 gives the PLL turn-on time for different PLL loop filter bandwidths. SWRS045 Page 52 of 91 CC1021 Loop filter no. 1 C6 [nF] 56 C7 [pF] 2200 C8 [pF] 560 R2 [k ] 3.3 Ω R3 [k ] 10 Ω PLL turn-on time [us] 1400 2 15 560 150 5.6 18 1300 3 3.9 120 33 12 39 1080 4 1.0 27 3.3 27 82 950 5 0.2 1.5 - 47 150 700 Comment Up to 9.6 kBaud data rate. ±5 kHz settling accuracy Up to 19.2 kBaud data rate. ±10 kHz settling accuracy Up to 38.4 kBaud data rate. ±15 kHz settling accuracy Up to 76.8 kBaud data rate. ±20 kHz settling accuracy Up to 153.6 kBaud data rate. ±50 kHz settling accuracy Table 25. Typical PLL turn-on time to within specified accuracy for different loop filter bandwidths. 15.4. PLL Lock Time versus Loop Filter Bandwidth PLL lock time depends on the PLL loop filter bandwidth. Table 26 gives the PLL lock time for different PLL loop filter bandwidths. If calibration has been performed the PLL lock time is the time needed for the PLL to lock to the desired frequency when going from RX to TX mode or vice versa. The Loop filter no. 1 C6 [nF] C7 [pF] C8 [pF] R2 [k ] R3 [k ] 56 2200 560 3.3 10 1 400 2 15 560 150 5.6 18 140 3 3.9 120 33 12 39 75 4 1.0 27 3.3 27 82 30 5 0.2 1.5 - 47 150 14 Ω Ω PLL lock time [us] 2 140 (50 kHz) 70 (100 kHz) 50 (150 kHz) 15 (200 kHz) 14 (500 kHz) Comment 3 490 230 180 55 28 Up to 9.6 kBaud data rate. ±5 kHz settling accuracy Up to 19.2 kBaud data rate. ±10 kHz settling accuracy Up to 38.4 kBaud data rate. ±15 kHz settling accuracy Up to 76.8 kBaud data rate. ±20 kHz settling accuracy Up to 153.6 kBaud data rate. ±50 kHz settling accuracy Table 26. Typical PLL lock time to within specified accuracy for different loop filter bandwidths. 1) 307.2 kHz step, 2) step as given in brackets, 3) 1 MHz step. 16. VCO and LNA Current Control The VCO current is programmable and should be set according to operating frequency, RX/TX mode and output power. Recommended settings for the VCO_CURRENT bits in the VCO register are shown in the register overview and also given by SmartRF Studio. The VCO current for frequency FREQ_A and FREQ_B can independently. be programmed The bias currents for the LNA, mixer and the LO and PA buffers are also programmable. The FRONTEND and the BUFF_CURRENT registers control these currents. SWRS045 Page 53 of 91 CC1021 17. Power Management offers great flexibility for power management in order to meet strict power consumption requirements in batteryoperated applications. Power down mode is controlled through the MAIN register. There are separate bits to control the RX part, the TX part, the frequency synthesizer and the crystal oscillator in the MAIN register. This individual control can be used to optimize for lowest possible current consumption in each application. Figure 30 shows a typical power-on and initializing sequence for minimum power consumption. CC1021 then be calibrated in both RX and TX mode. After this is completed, the is ready for use. See the detailed procedure flowcharts in Figure 29 - Figure 31. CC1021 With reference to Application Note AN023 CC1020 MCU Interfacing Chipcon recommends the following sequence. Note that the sub-routines are equally . applicable for the CC1020 CC1021 Figure 31 shows a typical sequence for activating RX and TX mode from power down mode for minimum power consumption. After power up: 1) ResetCC1020 2) Initialize 3) WakeUpCC1020ToRX 4) Calibrate 5) WakeUpCC1020ToTX 6) Calibrate Note that PSEL should be tri-stated or set to a high level during power down mode in order to prevent a trickle current from flowing in the internal pull-up resistor. After calibration is completed, enter TX mode (SetupCC1020TX), RX mode (SetupCC1020RX) or power down mode (SetupCC1020PD) Application Note AN023 CC1020 MCU Interfacing is also applicable for the . This application note includes example source code and is available from the Chipcon web site. From power-down mode to RX: 1) WakeUpCC1020ToRX 2) SetupCC1020RX CC1021 Chipcon recommends resetting the (by clearing the RESET_N bit in the MAIN register) when the chip is powered up initially. All registers that need to be configured should then be programmed (those which differ from their default values). Registers can be programmed should freely in any order. The CC1021 From power-down mode to TX: 1) WakeUpCC1020ToTX 2) SetupCC1020TX Switching from RX to TX mode: 1) SetupCC1020TX Switching from TX to RX mode: 1) SetupCC1020RX CC1021 SWRS045 Page 54 of 91 CC1021 Power Off ResetCC1020 Turn on power Reset CC1021 MAIN: RX_TX=0, F_REG=0, PD_MODE=1, FS_PD=1, XOSC_PD=1, BIAS_PD=1 RESET_N=0 RESET_N=1 WakeupCC1020ToRx/ WakeupCC1020ToTx Program all necessary registers except MAIN and RESET Turn on crystal oscillator, bias generator and synthesizer successively SetupCC1020PD Calibrate VCO and PLL MAIN: PD_MODE=1, FS_PD=1, XOSC_PD=1, BIAS_PD=1 PA_POWER=00h Power Down mode Figure 30. Initializing sequence SWRS045 Page 55 of 91 CC1021 WakeupCC1020ToTx Turn on crystal oscillator core MAIN: PD_MODE=1, FS_PD=1, XOSC_PD=0, BIAS_PD=1 Wait 1.2 ms* Turn on bias generator. MAIN: BIAS_PD=0 Wait 150 us RX or TX? TX Turn on frequency synthesizer MAIN: RXTX=0, F_REG=0, FS_PD=0 Turn on frequency synthesizer MAIN: RXTX=1, F_REG=1, FS_PD=0 Wait until lock detected from LOCK pin or STATUS register Turn on RX: MAIN: PD_MODE = 0 Wait until lock detected from LOCK pin or STATUS register Turn on TX: MAIN: PD_MODE = 0 Set PA_POWER RX mode TX mode SetupCC1020PD Turn off RX/TX: MAIN: PD_MODE = 1, FS_PD=1, XOSC_PD=1, BIAS_PD=1 PA_POWER=00h SetupCC1020Tx RX SetupCC1020Rx *Time to wait depends on the crystal frequency and the load capacitance SetupCC1020PD WakeupCC1020ToRx Power Down mode Power Down mode Figure 31. Sequence for activating RX or TX mode 18. On-Off Keying (OOK) The data modulator can also provide OOK (On-Off Keying) modulation. OOK is an ASK (Amplitude Shift Keying) modulation using 100% modulation depth. OOK modulation is enabled in RX and in TX by setting TXDEV_M[3:0] = 0000 in the DEVIATION register. An OOK diagram is shown in Figure 32. eye The data demodulator can also perform OOK demodulation. The demodulation is done by comparing the signal level with SWRS045 Page 56 of 91 CC1021 the "carrier sense" level (programmed as CS_LEVEL in the VGA4 register). The signal is then decimated and filtered in the data filter. Data decision and bit synchronization are as for FSK reception. In this mode AGC_AVG in the VGA2 register must be set to 3. The channel bandwidth must be 4 times the Baud rate for data rates up to 9.6 kBaud. For the highest data rates the channel bandwidth must be 2 times the Baud rate (see Table 27). Manchester coding must always be used for OOK. The AGC has a certain time-constant determined by FILTER_CLK, which depends on the IF filter bandwidth. There is a lower limit on FILTER_CLK and hence the AGC time constant. For low data rates the minimum time constant is too fast and the AGC will increase the gain when a "0" is received and decrease the gain when a "1" is received. For this reason the minimum data rate in OOK is 9.6 kBaud. Typical figures for the receiver sensitivity (BER = 10 3) are shown in Table 27 for OOK. Note that the automatic frequency control (AFC) cannot be used when receiving OOK, as it requires a frequency shift. Figure 32. OOK eye diagram. 9.6 kBaud. Data rate [kBaud] Filter BW [kHz] 9.6 19.2 38.4 76.8 153.6 38.4 51.2 102.4 153.6 307.2 Sensitivity [dBm] 433 MHz 868 MHz Manchester mode Manchester mode -103 -102 -95 -92 -81 -104 -101 -97 -94 -87 Table 27. Typical receiver sensitivity as a function of data rate at 433 and 868 MHz, OOK 3 modulation, BER = 10 , pseudo-random data (PN9 sequence). SWRS045 Page 57 of 91 CC1021 19. Crystal Oscillator the crystal to oscillate at the specified frequency. The recommended crystal frequency is 14.7456 MHz, but any crystal frequency in the range 4 - 20 MHz can be used. Using a crystal frequency different from 14.7456 MHz might in some applications give degraded performance. Refer to Application Note AN022 Crystal Frequency Selection for more details on the use of other crystal frequencies than 14.7456 MHz. The crystal frequency is used as reference for the data rate (as well as other internal functions) and in the 4 ñ 20 MHz range the frequencies 4.9152, 7.3728, 9.8304, 12.2880, 14.7456, 17.2032, 19.6608 MHz will give accurate data rates as shown in Table 17 and an IF frequency of 307.2 kHz. The crystal frequency will influence the programming of the CLOCK_A, CLOCK_B and MODEM registers. CL = 1 + C parasitic 1 1 + C4 C5 The parasitic capacitance is constituted by pin input capacitance and PCB stray capacitance. Total parasitic capacitance is typically 8 pF. A trimming capacitor may be placed across C5 for initial tuning if necessary. The crystal oscillator circuit is shown in Figure 33. Typical component values for different values of CL are given in Table 28. The crystal oscillator is amplitude regulated. This means that a high current is required to initiate the oscillations. When the amplitude builds up, the current is reduced to what is necessary to maintain approximately 600 mVpp amplitude. This ensures a fast start-up, keeps the drive level to a minimum and makes the oscillator insensitive to ESR variations. As long as the recommended load capacitance values are used, the ESR is not critical. An external clock signal or the internal crystal oscillator can be used as main frequency reference. An external clock signal should be connected to XOSC_Q1, while XOSC_Q2 should be left open. The XOSC_BYPASS bit in the INTERFACE register should be set to ë1í when an external digital rail-to-rail clock signal is used. No DC block should be used then. A sine with smaller amplitude can also be used. A DC blocking capacitor must then be used (10 nF) and the XOSC_BYPASS bit in the INTERFACE register should be set to ë0í. For input signal amplitude, see section 4.5 on page 12. The initial tolerance, temperature drift, aging and load pulling should be carefully specified in order to meet the required frequency accuracy in a certain application. By specifying the total expected frequency accuracy in SmartRF Studio together with data rate and frequency separation, the software will estimate the total bandwidth and compare to the available receiver channel filter bandwidth. The software will report any contradictions and a more accurate crystal will be recommended if required. Using the internal crystal oscillator, the crystal must be connected between the XOSC_Q1 and XOSC_Q2 pins. The oscillator is designed for parallel mode operation of the crystal. In addition, loading capacitors (C4 and C5) for the crystal are required. The loading capacitor values depend on the total load capacitance, CL, specified for the crystal. The total load capacitance seen between the crystal terminals should equal CL for XOSC_Q1 XOSC_Q2 XTAL C4 C5 Figure 33. Crystal oscillator circuit SWRS045 Page 58 of 91 CC1021 Item C4 C5 CL= 12 pF 6.8 pF 6.8 pF CL= 16 pF 15 pF 15 pF CL= 22 pF 27 pF 27 pF Table 28. Crystal oscillator component values 20. Built-in Test Pattern Generator number of received ones. Note that the 9 first received bits should be discarded in this case. Also note that one bit error will generate 3 received ones. The has a built-in test pattern generator that generates a PN9 pseudo random sequence. The PN9_ENABLE bit in the MODEM register enables the PN9 generator. A transition on the DIO pin is required after enabling the PN9 pseudo random sequence. CC1021 Transmitting only ones (DIO = 1), the BER can be tested by counting the number of received zeroes. The PN9 pseudo random sequence is 9 5 defined by the polynomial x + x + 1. The PN9 generator can also be used for transmission of ëreal-lifeí data when measuring narrowband ACP (Adjacent Channel Power), modulation bandwidth or occupied bandwidth. The PN9 sequence is ëXORíed with the DIO signal in both TX and RX mode as shown in Figure 34. Hence, by transmitting only zeros (DIO = 0), the BER (Bit Error Rate) can be tested by counting the Tx pseudo random sequence Tx out (modulating signal) Tx data (DIO pin) XOR 8 7 6 5 4 3 2 1 0 5 4 3 2 1 0 XOR Rx pseudo random sequence Rx in (Demodulated Rx data) 8 7 6 XOR XOR Rx out (DIO pin) Figure 34. PN9 pseudo random sequence generator in TX and RX mode SWRS045 Page 59 of 91 CC1021 21. Interrupt on Pin DCLK 21.1. Interrupt upon PLL Lock In synchronous mode the DCLK pin on can be used to give an interrupt signal to wake the microcontroller when the PLL is locked. CC1021 PD_MODE[1:0] in the MAIN register should be set to 01. If DCLK_LOCK in the INTERFACE register is set to 1 the DCLK signal is always logic high if the PLL is not in lock. When the PLL locks to the desired frequency the DCLK signal changes to logic 0. When this interrupt has been detected write PD_MODE[1:0] = 00. This will enable the DCLK signal. This function can be used to wait for the PLL to be locked before the PA is ramped up in transmit mode. In receive mode, it can be used to wait until the PLL is locked before searching for preamble. 21.2. Interrupt upon Received Signal Carrier Sense In synchronous mode the DCLK pin on can also be used to give an interrupt signal to the microcontroller when the RSSI level exceeds a certain threshold (carrier sense threshold). This function can be used to wake or interrupt the microcontroller when a strong signal is received. CC1021 Gating the DCLK signal with the carrier sense signal makes the interrupt signal. This function should only be used in receive mode and is enabled by setting DCLK_CS = 1 in the INTERFACE register. 22. The DCLK signal is always logic high unless carrier sense is indicated. When carrier sense is indicated the DCLK starts running. When gating the DCLK signal with the carrier sense signal at least 2 dummy bits should be added after the data payload in TX mode. The reason being that the carrier sense signal is generated earlier in the receive chain (i.e. before the demodulator), causing it to be updated 2 bits before the corresponding data is available on the DIO pin. In transmit mode DCLK_CS must be set to Errata Note 001. 0. Refer to CC1021 PA_EN and LNA_EN Digital Output Pins 22.1. Interfacing an External LNA or PA has two digital output pins, PA_EN and LNA_EN, which can be used to control an external LNA or PA. The functionality of these pins are controlled through the INTERFACE register. The outputs can also be used as general digital output control signals. CC1021 EXT_PA_POL and EXT_LNA_POL control the active polarity of the signals. EXT_PA and EXT_LNA control the function of the pins. If EXT_PA = 1, then the PA_EN pin will be activated when the internal PA is turned on. Otherwise, the EXT_PA_POL bit controls the PA_EN pin directly. If EXT_LNA = 1, then the LNA_EN pin will be activated when the internal LNA is turned on. Otherwise, the EXT_LNA_POL bit controls the LNA_EN pin directly. These two pins can therefore also be used as two general control signals, section 21.2. In the Chipcon reference design SWRS045 Page 60 of 91 CC1021 LNA_EN and PA_EN are used to control the external T/R switch. 22.2. General Purpose Output Control Pins The two digital output pins, PA_EN and LNA_EN, can be used as two general control signals by setting EXT_PA = 0 and EXT_LNA = 0. The output value is then set directly by the value written to EXT_PA_POL and EXT_LNA_POL. is controlled by LOCK_SELECT[3:0] in the LOCK register. The LOCK pin is low when LOCK_SELECT[3:0] = 0000, and high when LOCK_SELECT[3:0] = 0001. These features can be used to save I/O pins on the microcontroller when the other functions associated with these pins are not used. The LOCK pin can also be used as a general-purpose output pin. The LOCK pin 22.3. PA_EN and LNA_EN Pin Drive Figure 35 shows the PA_EN and LNA_EN pin drive currents. The sink and source currents have opposite signs but absolute values are used in Figure 35. 1400 1200 Current [uA] 1000 800 600 400 200 3.6 3.4 3.2 3 2.8 2.6 2.4 2.2 2 1.8 1.6 1.4 1.2 1 0.8 0.6 0.4 0.2 0 0 Voltage on PA_EN/LNA_EN pin [V] source current, 3 V sink current, 3V source current, 2.3 V sink current, 2.3 V source current, 3.6 V sink current, 3.6 V Figure 35. PA_EN and LNA_EN pin drive 23. System Considerations and Guidelines SRD regulations International regulations and national laws regulate the use of radio receivers and transmitters. SRDs (Short Range Devices) for license free operation are allowed to operate in the 433 and 868 - 870 MHz bands in most European countries. In the United States, such devices operate in the 260 ñ 470 and 902 - 928 MHz bands. A summary of the most important aspects of these regulations can be found in Application Note AN001 SRD regulations for license free transceiver operation, available from the Chipcon web site. SWRS045 Page 61 of 91 CC1021 Narrowband systems is recommended for narrowband applications with channel spacings of 50 kHz and higher complying with FCC CFR47 part 15 and EN 300 220. CC1021 is recommended in narrowband applications with channel spacings of 12.5 or 25 kHz complying with ARIB STD T-67 and EN 300 220. CC1020 and are fully compatible for channel spacings of 50 kHz and higher (receiver channel filter bandwidths of 38.4 kHz and higher). CC1020 CC1021 Due to on-chip complex filtering, the image frequency is removed. An on-chip calibration circuit is used to get the best possible image rejection. A narrowband preselector filter is not necessary to achieve image rejection. spectrum-shaping feature improves adjacent channel power (ACP) and occupied bandwidth. In ëtrueí FSK systems with abrupt frequency shifting, the spectrum is inherently broad. By making the frequency shift ësofterí, the spectrum can be made significantly narrower. Thus, higher data rates can be transmitted in the same bandwidth using GFSK. Low cost systems As the provide true narrowband multi-channel performance without any external filters, a very low cost high performance system can be achieved. The oscillator crystal can then be a low cost crystal with 50 ppm frequency tolerance using the on-chip frequency tuning possibilities. CC1021 Battery operated systems In low power applications, the power down mode should be used when is not being active. Depending on the start-up time requirement, the oscillator core can be powered during power down. See section 17 page 54 for information on how effective power management can be implemented. CC1021 A unique feature in is the very fine frequency resolution. This can be used for temperature compensation of the crystal if the temperature drift curve is known and a temperature sensor is included in the system. Even initial adjustment can be performed using the frequency programmability. This eliminates the need for an expensive TCXO and trimming in some applications. For more details refer to Application Note AN027 Temperature Compensation available from the Chipcon web site. CC1021 In less demanding applications, a crystal with low temperature drift and low aging could be used without further compensation. A trimmer capacitor in the crystal oscillator circuit (in parallel with C5) could be used to set the initial frequency accurately. The frequency offset between a transmitter and receiver is measured in the and can be read back from the AFC register. The measured frequency offset can be used to calibrate the receiver frequency using the transmitter as the reference. For more details refer to Application Note AN029 CC1020/1021 AFC available from the Chipcon web site. CC1021 also has the possibility to use Gaussian shaped FSK (GFSK). This CC1021 High reliability systems Using a SAW filter as a preselector will improve the communication reliability in harsh environments by reducing the probability of blocking. The receiver sensitivity and the output power will be reduced due to the filter insertion loss. By inserting the filter in the RX path only, together with an external RX/TX switch, only the receiver sensitivity is reduced and output power is remained. The PA_EN and LNA_EN pin can be configured to control an external LNA, RX/TX switch or power amplifier. This is controlled by the INTERFACE register. Frequency hopping spread spectrum systems (FHSS) Due to the very fast locking properties of the PLL, the is also very suitable for frequency hopping systems. Hop rates of 1-100 hops/s are commonly used depending on the bit rate and the amount of data to be sent during each transmission. The two frequency registers (FREQ_A and FREQ_B) are designed such that the ënextí frequency can be programmed while the ëpresentí frequency CC1021 SWRS045 Page 62 of 91 CC1021 TEST3[5:0]: VCO_CO[5:0] TEST3[6]: VCO_CAL_OVERRIDE is used. The switching between the two frequencies is done through the MAIN register. Several features have been included to do the hopping without a need to re-synchronize the receiver. For more details refer to Application Note AN014 Frequency Hopping Systems available from the Chipcon web site. CHP_CO[3:0] is the register setting read from CHP_CURRENT[3:0], VCO_AO[4:0] is the register setting read from VCO_ARRAY[4:0] and VCO_CO[5:0] is the register setting read from VCO_CAL_CURRENT[5:0]. In order to implement a frequency hopping system with do the following: Assume channel 1 defined by register FREQ_A is currently being used and that should operate on channel 2 next (to change channel simply write to register MAIN[6]). The channel 2 frequency can be set by register FREQ_B which can be written to while operating on channel 1. The calibration data must be written to the TEST1-3 registers after switching to the next frequency. That is, when hopping to a new channel write to register MAIN[6] first and the test registers next. The PA should be switched off between each hop and the PLL should be checked for lock before switching the PA back on after a hop has been performed. CC1021 Set the desired frequency, calibrate and store the following register settings in nonvolatile memory: CC1021 STATUS1[3:0]: CHP_CURRENT[3:0] STATUS2[4:0]: VCO_ARRAY[4:0] STATUS3[5:0]:VCO_CAL_CURRENT[5:0] Repeat the calibration for each desired frequency. VCO_CAL_CURRENT[5:0] is not dependent on the RF frequency and the same value can be used for all frequencies. When performing frequency hopping, write the stored values to the corresponding TEST1, TEST2 and TEST3 registers, and enable override: Note that the override bits VCO_OVERRIDE, CHP_OVERRIDE and VCO_CAL_OVERRIDE must be disabled when performing a re-calibration. TEST1[3:0]: CHP_CO[3:0] TEST2[4:0]: VCO_AO[4:0] TEST2[5]: VCO_OVERRIDE TEST2[6]: CHP_OVERRIDE 24. PCB Layout Recommendations The top layer should be used for signal routing, and the open areas should be filled with metallization connected to ground using several vias. to the power line (or power plane) by separate vias. The best routing is from the power line (or power plane) to the decoupling capacitor and then to the supply pin. Supply power filtering is very important, especially for pins 23, 22, 20 and 18. CC1021 The area under the chip is used for grounding and must be connected to the bottom ground plane with several vias. In the Chipcon reference designs we have placed 9 vias inside the exposed die attached pad. These vias should be ìtentedî (covered with solder mask) on the component side of the PCB to avoid migration of solder through the vias during the solder reflow process. Each decoupling capacitor should be placed as close as possible to the supply pin it is supposed to decouple. Each decoupling capacitor should be connected Each decoupling capacitor ground pad should be connected to the ground plane using a separate via. Direct connections between neighboring power pins will increase noise coupling and should be avoided unless absolutely necessary. The external components should ideally be as small as possible and surface mount devices are highly recommended. SWRS045 Page 63 of 91 CC1021 Precaution should be used when placing the microcontroller in order to avoid noise interfering with the RF circuitry. The recommended PCB layout is A the same as for the CC1020/1070DK Development Kit with a CC1021 CC1020. 25. Antenna Considerations can be used together with various types of antennas. The most common antennas for short-range communication are monopole, helical and loop antennas. CC1021 Monopole antennas are resonant antennas with a length corresponding to one quarter of the electrical wavelength (λ/4). They are very easy to design and can be implemented simply as a ìpiece of wireî or even integrated onto the PCB. Non-resonant monopole antennas shorter than λ/4 can also be used, but at the expense of range. In size and cost critical applications such an antenna may very well be integrated onto the PCB. Helical antennas can be thought of as a combination of a monopole and a loop antenna. They are a good compromise in size critical applications. But helical antennas tend to be more difficult to optimize than the simple monopole. Loop antennas are easy to integrate into the PCB, but are less effective due to 26. fully assembled CC1020EMX Evaluation Module is available. It is strongly advised that this reference layout is followed very closely in order to get the best performance. The layout Gerber files are available from the Chipcon web site. difficult impedance matching because of their very low radiation resistance. For low power applications the λ/4monopole antenna is recommended due to its simplicity as well as providing the best range. The length of the λ/4-monopole antenna is given by: L = 7125 / f where f is in MHz, giving the length in cm. An antenna for 868 MHz should be 8.2 cm, and 16.4 cm for 433 MHz. The antenna should be connected as close as possible to the IC. If the antenna is located away from the input pin the antenna should be matched to the feeding transmission line (50 Ω). For a more thorough background on antennas, please refer to Application Note AN003 SRD Antennas available from the Chipcon web site. Configuration Registers The configuration of is done by programming the 8-bit configuration registers. The configuration data based on selected system parameters are most easily found by using the SmartRF Studio software. Complete descriptions of the registers are given in the following tables. After a RESET is programmed, all the registers have default values. The TEST registers also get default values after a CC1021 RESET, and should not be altered by the user. Chipcon recommends using the register settings found using the SmartRF Studio software. These are the register settings that Chipcon can guarantee across temperature, voltage and process. Please check the Chipcon web site for regularly updates to the SmartRF Studio software. SWRS045 Page 64 of 91 CC1021 26.1. CC1021 Register Overview ADDRESS 00h 01h 02h 03h 04h 05h 06h 07h 08h 09h 0Ah 0Bh 0Ch 0Dh 0Eh 0Fh 10h 11h 12h 13h 14h 15h 16h 17h 18h 19h 1Ah 1Bh 1Ch 1Dh 1Eh 1Fh 20h 21h 22h 23h 24h 25h 26h 27h 40h 41h 42h 43h 44h 45h 46h 47h 48h 49h 4Ah 4Bh Byte Name MAIN INTERFACE RESET SEQUENCING FREQ_2A FREQ_1A FREQ_0A CLOCK_A FREQ_2B FREQ_1B FREQ_0B CLOCK_B VCO MODEM DEVIATION AFC_CONTROL FILTER VGA1 VGA2 VGA3 VGA4 LOCK FRONTEND ANALOG BUFF_SWING BUFF_CURRENT PLL_BW CALIBRATE PA_POWER MATCH PHASE_COMP GAIN_COMP POWERDOWN TEST1 TEST2 TEST3 TEST4 TEST5 TEST6 TEST7 STATUS RESET_DONE RSSI AFC GAUSS_FILTER STATUS1 STATUS2 STATUS3 STATUS4 STATUS5 STATUS6 STATUS7 Description Main control register Interface control register Digital module reset register Automatic power-up sequencing control register Frequency register 2A Frequency register 1A Frequency register 0A Clock generation register A Frequency register 2B Frequency register 1B Frequency register 0B Clock generation register B VCO current control register Modem control register TX frequency deviation register RX AFC control register Channel filter / RSSI control register VGA control register 1 VGA control register 2 VGA control register 3 VGA control register 4 Lock control register Front end bias current control register Analog modules control register LO buffer and prescaler swing control register LO buffer and prescaler bias current control register PLL loop bandwidth / charge pump current control register PLL calibration control register Power amplifier output power register Match capacitor array control register, for RX and TX impedance matching Phase error compensation control register for LO I/Q Gain error compensation control register for mixer I/Q Power-down control register Test register for overriding PLL calibration Test register for overriding PLL calibration Test register for overriding PLL calibration Test register for charge pump and IF chain testing Test register for ADC testing Test register for VGA testing Test register for VGA testing Status information register (PLL lock, RSSI, calibration ready, etc.) Status register for digital module reset Received signal strength register Average received frequency deviation from IF (can be used for AFC) Digital FM demodulator register Status of PLL calibration results etc. (test only) Status of PLL calibration results etc. (test only) Status of PLL calibration results etc. (test only) Status of ADC signals (test only) Status of channel filter ìIî signal (test only) Status of channel filter ìQî signal (test only) Status of AGC (test only) SWRS045 Page 65 of 91 CC1021 MAIN Register (00h) REGISTER NAME Active RXTX F_REG Default value - MAIN[7] MAIN[6] MAIN[5:4] PD_MODE[1:0] - - MAIN[3] MAIN[2] MAIN[1] FS_PD XOSC_PD BIAS_PD - H H H MAIN[0] RESET_N - L - Description RX/TX switch, 0: RX , 1: TX Selection of Frequency Register, 0: Register A, 1: Register B Power down mode 0 (00): Receive Chain in power-down in TX, PA in power-down in RX 1 (01): Receive Chain and PA in power-down in both TX and RX 2 (10): Individual modules can be put in power-down by programming the POWERDOWN register 3 (11): Automatic power-up sequencing is activated (see below) Power Down of Frequency Synthesizer Power Down of Crystal Oscillator Core Power Down of BIAS (Global Current Generator) and Crystal Oscillator Buffer Reset, active low. Writing RESET_N low will write default values to all other registers than MAIN. Bits in MAIN do not have a default value and will be written directly through the configuration interface. Must be set high to complete reset. MAIN Register (00h) when using automatic power-up sequencing (RXTX = 0, PD_MODE[1:0] =11) REGISTER NAME Active RXTX F_REG PD_MODE[1:0] SEQ_CAL[1:0] Default value - MAIN[7] MAIN[6] MAIN[5:4] MAIN[3:2] MAIN[1] SEQ_PD - ↑ MAIN[0] RESET_N - L H - Description Automatic power-up sequencing only works in RX (RXTX=0) Selection of Frequency Register, 0: Register A, 1: Register B Set PD_MODE[1:0]=3 (11) to enable sequencing Controls PLL calibration before re-entering power-down 0: Never perform PLL calibration as part of sequence 1: Always perform PLL calibration at end of sequence 2: Perform PLL calibration at end of every 16th sequence 3: Perform PLL calibration at end of every 256th sequence ↑1: Put the chip in power down and wait for start of new power-up sequence Reset, active low. Writing RESET_N low will write default values to all other registers than MAIN. Bits in MAIN do not have a default value and will be written directly through the configuration interface. Must be set high to complete reset. SWRS045 Page 66 of 91 CC1021 INTERFACE Register (01h) REGISTER NAME Active XOSC_BYPASS Default value 0 INTERFACE[7] INTERFACE[6] SEP_DI_DO 0 H H INTERFACE[5] DCLK_LOCK 0 H INTERFACE[4] DCLK_CS 0 H INTERFACE[3] EXT_PA 0 H INTERFACE[2] EXT_LNA 0 H INTERFACE[1] EXT_PA_POL 0 H INTERFACE[0] EXT_LNA_POL 0 H Description Bypass internal crystal oscillator, use external clock 0: Internal crystal oscillator is used, or external sine wave fed through a coupling capacitor 1: Internal crystal oscillator in power down, external clock with rail-to-rail swing is used Use separate pin for RX data output 0: DIO is data output in RX and data input in TX. LOCK pin is available (Normal operation). 1: DIO is always input, and a separate pin is used for RX data output (synchronous mode: LOCK pin, asynchronous mode: DCLK pin). If SEP_DI_DO=1 and SEQ_PSEL=0 in SEQUENCING register then negative transitions on DIO is used to start power-up sequencing when PD_MODE=3 (power-up sequencing is enabled). Gate DCLK signal with PLL lock signal in synchronous mode Only applies when PD_MODE = ì01î 0: DCLK is always 1 1: DCLK is always 1 unless PLL is in lock Gate DCLK signal with carrier sense indicator in synchronous mode Use when receive chain is active (in power-up) Always set to 0 in TX mode. 0: DCLK is independent of carrier sense indicator. 1: DCLK is always 1 unless carrier sense is indicated Use PA_EN pin to control external PA 0: PA_EN pin always equals EXT_PA_POL bit 1: PA_EN pin is asserted when internal PA is turned on Use LNA_EN pin to control external LNA 0: LNA_EN pin always equals EXT_LNA_POL bit 1: LNA_EN pin is asserted when internal LNA is turned on Polarity of external PA control 0: PA_EN pin is ì0î when activating external PA 1: PA_EN pin is ì1î when activating external PA Polarity of external LNA control 0: LNA_EN pin is ì0î when activating external LNA 1: LNA_EN pin is ì1î when activating external LNA Note: If TF_ENABLE=1 or TA_ENABLE=1 in TEST4 register, then INTERFACE[3:0] controls analog test module: INTERFACE[3] = TEST_PD, INTERFACE[2:0] = TEST_MODE[2:0]. Otherwise, TEST_PD=1 and TEST_MODE[2:0]=001. RESET Register (02h) REGISTER NAME Active ADC_RESET_N AGC_RESET_N GAUSS_RESET_N AFC_RESET_N BITSYNC_RESET_N Default value 0 0 0 0 0 RESET[7] RESET[6] RESET[5] RESET[4] RESET[3] RESET[2] RESET[1] RESET[0] SYNTH_RESET_N SEQ_RESET_N CAL_LOCK_RESET_N 0 0 0 L L L L L L L L Description Reset ADC control logic Reset AGC (VGA control) logic Reset Gaussian data filter Reset AFC / FSK decision level logic Reset modulator, bit synchronization logic and PN9 PRBS generator Reset digital part of frequency synthesizer Reset power-up sequencing logic Reset calibration logic and lock detector Note: For reset of CC1021 write RESET_N=0 in the MAIN register. The reset register should not be used during normal operation. Bits in the RESET register are self-clearing (will be set to 1 when the reset operation starts). Relevant digital clocks must be running for the resetting to complete. After writing to the RESET register, the user should verify that all reset operations have been completed, by reading the RESET_DONE status register (41h) until all bits equal 1. SWRS045 Page 67 of 91 CC1021 SEQUENCING Register (03h) REGISTER NAME Active SEQ_PSEL Default value 1 SEQUENCING[7] SEQUENCING[6:4] RX_WAIT[2:0] 0 - SEQUENCING[3:0] CS_WAIT[3:0] 10 - Description H Use PSEL pin to start sequencing 0: PSEL pin does not start sequencing. Negative transitions on DIO starts power-up sequencing if SEP_DI_DO=1. 1: Negative transitions on the PSEL pin will start powerup sequencing Waiting time from PLL enters lock until RX power-up 0: Wait for approx. 32 ADC_CLK periods (26 µs) 1: Wait for approx. 44 ADC_CLK periods (36 µs) 2: Wait for approx. 64 ADC_CLK periods (52 µs) 3: Wait for approx. 88 ADC_CLK periods (72 µs) 4: Wait for approx. 128 ADC_CLK periods (104 µs) 5: Wait for approx. 176 ADC_CLK periods (143 µs) 6: Wait for approx. 256 ADC_CLK periods (208 µs) 7: No additional waiting time before RX power-up Waiting time for carrier sense from RX power-up 0: Wait 20 FILTER_CLK periods before power down 1: Wait 22 FILTER_CLK periods before power down 2: Wait 24 FILTER_CLK periods before power down 3: Wait 26 FILTER_CLK periods before power down 4: Wait 28 FILTER_CLK periods before power down 5: Wait 30 FILTER_CLK periods before power down 6: Wait 32 FILTER_CLK periods before power down 7: Wait 36 FILTER_CLK periods before power down 8: Wait 40 FILTER_CLK periods before power down 9: Wait 44 FILTER_CLK periods before power down 10: Wait 48 FILTER_CLK periods before power down 11: Wait 52 FILTER_CLK periods before power down 12: Wait 56 FILTER_CLK periods before power down 13: Wait 60 FILTER_CLK periods before power down 14: Wait 64 FILTER_CLK periods before power down 15: Wait 72 FILTER_CLK periods before power down FREQ_2A Register (04h) REGISTER NAME FREQ_2A[7:0] FREQ_A[22:15] Default value 131 Active Default value 177 Active Default value 124 1 Active - Description 8 MSB of frequency control word A FREQ_1A Register (05h) REGISTER NAME FREQ_1A[7:0] FREQ_A[14:7] - Description Bit 15 to 8 of frequency control word A FREQ_0A Register (06h) REGISTER NAME FREQ_0A[7:1] FREQ_0A[0] FREQ_A[6:0] DITHER_A H Description 7 LSB of frequency control word A Enable dithering for frequency A SWRS045 Page 68 of 91 CC1021 CLOCK_A Register (07h) REGISTER NAME CLOCK_A[7:5] REF_DIV_A[2:0] Default value 2 Active - CLOCK_A[4:2] MCLK_DIV1_A[2:0] 4 - CLOCK_A[1:0] MCLK_DIV2_A[1:0] 0 - Description Reference frequency divisor (A): 0: Not supported 1: REF_CLK frequency = Crystal frequency / 2 Ö 7: REF_CLK frequency = Crystal frequency / 8 It is recommended to use the highest possible reference clock frequency that allows the desired Baud rate. Modem clock divider 1 (A): 0: Divide by 2.5 1: Divide by 3 2: Divide by 4 3: Divide by 7.5 (2.5∑3) 4: Divide by 12.5 (2.5∑5) 5: Divide by 40 (2.5∑16) 6: Divide by 48 (3∑16) 7: Divide by 64 (4∑16) Modem clock divider 2 (A): 0: Divide by 1 1: Divide by 2 2: Divide by 4 3: Divide by 8 MODEM_CLK frequency is FREF frequency divided by the product of divider 1 and divider 2. Baud rate is MODEM_CLK frequency divided by 8. FREQ_2B Register (08h) REGISTER NAME FREQ_2B[7:0] FREQ_B[22:15] Default value 131 Active Default value 189 Active Default value 124 1 Active - Description 8 MSB of frequency control word B FREQ_1B Register (09h) REGISTER NAME FREQ_1B[7:0] FREQ_B[14:7] - Description Bit 15 to 8 of frequency control word B FREQ_0B Register (0Ah) REGISTER NAME FREQ_0B[7:1] FREQ_0B[0] FREQ_B[6:0] DITHER_B H Description 7 LSB of frequency control word B Enable dithering for frequency B SWRS045 Page 69 of 91 CC1021 CLOCK_B Register (0Bh) REGISTER NAME Active REF_DIV_B[2:0] Default value 2 CLOCK_B[7:5] CLOCK_B[4:2] MCLK_DIV1_B[2:0] 4 - CLOCK_B[1:0] MCLK_DIV2_B[1:0] 0 - - Description Reference frequency divisor (B): 0: Not supported 1: REF_CLK frequency = Crystal frequency / 2 Ö 7: REF_CLK frequency = Crystal frequency / 8 Modem clock divider 1 (B): 0: Divide by 2.5 1: Divide by 3 2: Divide by 4 3: Divide by 7.5 (2.5∑3) 4: Divide by 12.5 (2.5∑5) 5: Divide by 40 (2.5∑16) 6: Divide by 48 (3∑16) 7: Divide by 64 (4∑16) Modem clock divider 2 (B): 0: Divide by 1 1: Divide by 2 2: Divide by 4 3: Divide by 8 MODEM_CLK frequency is FREF frequency divided by the product of divider 1 and divider 2. Baud rate is MODEM_CLK frequency divided by 8. VCO Register (0Ch) REGISTER NAME Active VCO_CURRENT_A[3:0] Default value 8 VCO[7:4] VCO[3:0] VCO_CURRENT_B[3:0] 8 - - Description Control of current in VCO core for frequency A 0: 1.4 mA current in VCO core 1: 1.8 mA current in VCO core 2: 2.1 mA current in VCO core 3: 2.5 mA current in VCO core 4: 2.8 mA current in VCO core 5: 3.2 mA current in VCO core 6: 3.5 mA current in VCO core 7: 3.9 mA current in VCO core 8: 4.2 mA current in VCO core 9: 4.6 mA current in VCO core 10: 4.9 mA current in VCO core 11: 5.3 mA current in VCO core 12: 5.6 mA current in VCO core 13: 6.0 mA current in VCO core 14: 6.4 mA current in VCO core 15: 6.7 mA current in VCO core Recommended setting: VCO_CURRENT_A=4 Control of current in VCO core for frequency B The current steps are the same as for VCO_CURRENT_A Recommended setting: VCO_CURRENT_B=4 SWRS045 Page 70 of 91 CC1021 MODEM Register (0Dh) REGISTER NAME MODEM[7] MODEM[6:4] ADC_DIV[2:0] MODEM[3] MODEM[2] MODEM[1:0] Default value 0 3 PN9_ENABLE Active - 0 0 DATA_FORMAT[1:0] Reserved, write 0 ADC clock divisor 0: Not supported 1: ADC frequency = XOSC frequency / 4 2: ADC frequency = XOSC frequency / 6 3: ADC frequency = XOSC frequency / 8 4: ADC frequency = XOSC frequency / 10 5: ADC frequency = XOSC frequency / 12 6: ADC frequency = XOSC frequency / 14 7: ADC frequency = XOSC frequency / 16 Note that the intermediate frequency should be as close to 307.2 kHz as possible. ADC clock frequency is always 4 times the intermediate frequency and should therefore be as close to 1.2288 MHz as possible. Reserved, write 0 Enable scrambling of TX and RX with PN9 pseudorandom bit sequence 0: PN9 scrambling is disabled 1: PN9 scrambling is enabled (x9+x5+1) H 0 Description The PN9 pseudo-random bit sequence can be used for BER testing by only transmitting zeros, and then counting the number of received ones. Modem data format 0 (00): NRZ operation 1 (01): Manchester operation 2 (10): Transparent asynchronous UART operation, set DCLK=0 3 (11): Transparent asynchronous UART operation, set DCLK=1 - DEVIATION Register (0Eh) REGISTER NAME Active TX_SHAPING Default value 1 DEVIATION[7] DEVIATION[6:4] DEVIATION [3:0] TXDEV_X[2:0] TXDEV_M[3:0] 6 8 - H Description Enable Gaussian shaping of transmitted data Recommended setting: TX_SHAPING=1 Transmit frequency deviation exponent Transmit frequency deviation mantissa Deviation in 402-470 MHz band: FREF ∑TXDEV_M ∑2(TXDEV_X 16) Deviation in 804-940 MHz band: FREF ∑TXDEV_M ∑2(TXDEV_X 15) On-off-keying (OOK) is used in RX/TX if TXDEV_M[3:0]=0 To find TXDEV_M given the deviation and TXDEV_X: TXDEV_M = deviation∑2(16 in 402-470 MHz band, TXDEV_X) TXDEV_M = deviation∑2(15 in 804-940 MHz band. TXDEV_X) /FREF /FREF Decrease TXDEV_X and try again if TXDEV_M < 8. Increase TXDEV_X and try again if TXDEV_M ≥ 16. SWRS045 Page 71 of 91 CC1021 AFC_CONTROL Register (0Fh) REGISTER NAME AFC_CONTROL[7:6] SETTLING[1:0] AFC_CONTROL[5:4] AFC_CONTROL[3:0] Default value 2 RXDEV_X[1:0] RXDEV_M[3:0] Active - 1 12 - Description Controls AFC settling time versus accuracy 0: AFC off; zero average frequency is used in demodulator 1: Fastest settling; frequency averaged over 1 0/1 bit pair 2: Medium settling; frequency averaged over 2 0/1 bit pairs 3: Slowest settling; frequency averaged over 4 0/1 bit pairs Recommended setting: AFC_CONTROL=3 for higher accuracy unless it is essential to have the fastest settling time when transmission starts after RX is activated. RX frequency deviation exponent RX frequency deviation mantissa Expected RX deviation should be: Baud rate ∑RXDEV_M ∑2(RXDEV_X 3) / 3 To find RXDEV_M given the deviation and RXDEV_X: RXDEV_M = 3 ∑deviation ∑2(3 RXDEV_X) / Baud rate Decrease RXDEV_X and try again if RXDEV_M<8. Increase RXDEV_X and try again if RXDEV_M≥16. Note: The RX frequency deviation should be close to half the TX frequency deviation for GFSK at 100 kBaud data rate and below. The RX frequency deviation should be close to the TX frequency deviation for FSK and for GFSK at 100 kBaud data rate and above. FILTER Register (10h) REGISTER NAME FILTER[7] FILTER_BYPASS FILTER[6:5] FILTER[4:3] FILTER[2:0] DEC_SHIFT[1:0] DEC_DIV[2:0] Default value 0 0 0 Active H - - Description Bypass analog image rejection / anti-alias filter. Set to 1 for increased dynamic range at high Baud rates. Recommended setting: FILTER_BYPASS=0 below 76.8 kBaud, FILTER_BYPASS=1 for 76.8 kBaud and up. Number of extra bits to shift decimator input (may improve filter accuracy and lower power consumption). Recommended settings: DEC_SHIFT=0 when DEC_DIV ≤1 (receiver channel bandwidth ≥ 153.6 kHz), DEC_SHIFT=1 when DEC_DIV >1 (receiver channel bandwidth < 153.6 kHz). Reserved Decimation clock divisor 0: Decimation clock divisor = 1, 307.2 kHz channel filter BW. 1: Decimation clock divisor = 2, 153.6 kHz channel filter BW. Ö 6: Decimation clock divisor = 7, 43.9 kHz channel filter BW. 7: Decimation clock divisor = 8, 38.4 kHz channel filter BW. Channel filter bandwidth is 307.2 kHz divided by the decimation clock divisor. SWRS045 Page 72 of 91 CC1021 VGA1 Register (11h) REGISTER NAME VGA1[7:6] CS_SET[1:0] VGA1[5] VGA1[4:2] VGA1[1:0] CS_RESET VGA_WAIT[2:0] VGA_FREEZE[1:0] Default value 1 1 Active - - 1 - 1 - Description Sets the number of consecutive samples at or above carrier sense level before carrier sense is indicated (e.g. on LOCK pin) 0: Set carrier sense after first sample at or above carrier sense level 1: Set carrier sense after second sample at or above carrier sense level 2: Set carrier sense after third sample at or above carrier sense level 3: Set carrier sense after fourth sample at or above carrier sense level Increasing CS_SET reduces the number of ìfalseî carrier sense events due to noise at the expense of increased carrier sense response time. Sets the number of consecutive samples below carrier sense level before carrier sense indication (e.g. on lock pin) is reset 0: Carrier sense is reset after first sample below carrier sense level 1: Carrier sense is reset after second sample below carrier sense level Recommended setting: CS_RESET=1 in order to reduce the chance of losing carrier sense due to noise. Controls how long AGC, bit synchronization, AFC and RSSI levels are frozen after VGA gain is changed when frequency is changed between A and B or PLL has been out of lock or after RX power-up 0: Freeze operation for 16 filter clocks, 8/(filter BW) seconds 1: Freeze operation for 20 filter clocks, 10/(filter BW) seconds 2: Freeze operation for 24 filter clocks, 12/(filter BW) seconds 3: Freeze operation for 28 filter clocks, 14/(filter BW) seconds 4: Freeze operation for 32 filter clocks, 16/(filter BW) seconds 5: Freeze operation for 40 filter clocks, 20/(filter BW) seconds 6: Freeze operation for 48 filter clocks, 24/(filter BW) seconds 7: Freeze present levels unconditionally Controls the additional time AGC, bit synchronization, AFC and RSSI levels are frozen when frequency is changed between A and B or PLL has been out of lock or after RX power-up 0: Freeze levels for approx. 16 ADC_CLK periods (13 µs) 1: Freeze levels for approx. 32 ADC_CLK periods (26 µs) 2: Freeze levels for approx. 64 ADC_CLK periods (52 µs) 3: Freeze levels for approx. 128 ADC_CLK periods (104 µs) SWRS045 Page 73 of 91 CC1021 VGA2 Register (12h) REGISTER NAME Active LNA2_MIN Default value 0 VGA2[7] VGA2[6] LNA2_MAX 1 - VGA2[5:4] LNA2_SETTING[1:0] 3 - - Description Minimum LNA2 setting used in VGA 0: Minimum LNA2 gain 1: Medium LNA2 gain Recommended setting: LNA2_MIN=0 for best selectivity. Maximum LNA2 setting used in VGA 0: Medium LNA2 gain 1: Maximum LNA2 gain Recommended setting: LNA2_MAX=1 for best sensitivity. Selects at what VGA setting the LNA gain should be changed 0: Apply LNA2 change below min. VGA setting. 1: Apply LNA2 change at approx. 1/3 VGA setting (around VGA setting 10). 2: Apply LNA2 change at approx. 2/3 VGA setting (around VGA setting 19). 3: Apply LNA2 change above max. VGA setting. Recommended setting: LNA2_SETTING=0 if VGA_SETTING<10, LNA2_SETTING=1 otherwise. VGA2[3] AGC_DISABLE 0 H VGA2[2] AGC_HYSTERESIS 1 H VGA2[1:0] AGC_AVG[1:0] 1 - If LNA2_MIN=1 and LNA2_MAX=0, then the LNA2 setting is controlled by LNA2_SETTING: 0: Between medium and maximum LNA2 gain 1: Minimum LNA2 gain 2: Medium LNA2 gain 3: Maximum LNA2 gain Disable AGC 0: AGC is enabled 1: AGC is disabled (VGA_SETTING determines VGA gain) Recommended setting: AGC_DISABLE=0 for good dynamic range. Enable AGC hysteresis 0: No hysteresis. Immediate gain change for smallest up/down step 1: Hysteresis enabled. Two samples in a row must indicate gain change for smallest up/down step Recommended setting: AGC_HYSTERESIS=1. Sets how many samples that are used to calculate average output magnitude for AGC/RSSI. 0: Magnitude is averaged over 2 filter output samples 1: Magnitude is averaged over 4 filter output samples 2: Magnitude is averaged over 8 filter output samples 3: Magnitude is averaged over 16 filter output samples Recommended setting: AGC_AVG=1. For best AGC/RSSI accuracy AGC_AVG=3. For automatic power-up sequencing, the AGC_AVG and CS_SET values must be chosen so that carrier sense is available in time to be detected before the chip re-enters power-down. SWRS045 Page 74 of 91 CC1021 VGA3 Register (13h) REGISTER NAME VGA3[7:5] VGA_DOWN[2:0] VGA3[4:0] VGA_SETTING[4:0] Default value 1 24 Active - Description Decides how much the signal strength must be above CS_LEVEL+VGA_UP before VGA gain is decreased. 0: Gain is decreased 4.5 dB above CS_LEVEL+VGA_UP 1: Gain is decreased 6 dB above CS_LEVEL+VGA_UP Ö 6: Gain is decreased 13.5 dB above CS_LEVEL+VGA_UP 7: Gain is decreased 15 dB above CS_LEVEL+VGA_UP See Figure 20 on page 39 for an explanation of the relationship between RSSI, AGC and carrier sense settings. VGA setting to be used when receive chain is turned on This is also the maximum gain that the AGC is allowed to use. H See Figure 20 on page 39 for an explanation of the relationship between RSSI, AGC and carrier sense settings. VGA4 Register (14h) REGISTER VGA4[7:5] VGA4[4:0] NAME VGA_UP[2:0] CS_LEVEL[4:0] Default value 1 24 Active - H Description Decides the level where VGA gain is increased if it is not already at the maximum set by VGA_SETTING. 0: Gain is increased when signal is below CS_LEVEL 1: Gain is increased when signal is below CS_LEVEL+1.5 dB Ö 6: Gain is increased when signal is below CS_LEVEL+9 dB 7: Gain is increased when signal below CS_LEVEL+10.5 dB See Figure 20 on page 39 for an explanation of the relationship between RSSI, AGC and carrier sense settings. Reference level for Received Signal Strength Indication (carrier sense level) and AGC. See Figure 20 on page 39 for an explanation of the relationship between RSSI, AGC and carrier sense settings. SWRS045 Page 75 of 91 CC1021 LOCK Register (15h) REGISTER NAME Active LOCK_SELECT[3:0] Default value 0 LOCK[7:4] LOCK[3] WINDOW_WIDTH 0 - LOCK[2] LOCK_MODE 0 - LOCK[1:0] LOCK_ACCURACY[1:0] 0 - - Description Selection of signals to LOCK pin 0: Set to 0 1: Set to 1 2: LOCK_CONTINUOUS (active low) 3: LOCK_INSTANT (active low) 4: CARRIER_SENSE (RSSI above threshold, active low) 5: CAL_COMPLETE (active low) 6: SEQ_ERROR (active low) 7: FXOSC 8: REF_CLK 9: FILTER_CLK 10: DEC_CLK 11: PRE_CLK 12: DS_CLK 13: MODEM_CLK 14: VCO_CAL_COMP 15: F_COMP Selects lock window width 0: Lock window is 2 prescaler clock cycles wide 1: Lock window is 4 prescaler clock cycles wide Recommended setting: WINDOW_WIDTH=0. Selects lock detector mode 0: Counter restart mode 1: Up/Down counter mode Recommended setting: LOCK_MODE=0. Selects lock accuracy (counter threshold values) 0: Declare lock at counter value 127, out of lock at value 111 1: Declare lock at counter value 255, out of lock at value 239 2: Declare lock at counter value 511, out of lock at value 495 3: Declare lock at counter value 1023, out of lock at value 1007 Note: Set LOCK_SELECT=2 to use the LOCK pin as a lock indicator. SWRS045 Page 76 of 91 CC1021 FRONTEND Register (16h) REGISTER NAME FRONTEND[7:6] LNAMIX_CURRENT[1:0] Default value 2 Active - Description Controls current in LNA, LNA2 and mixer Recommended setting: LNAMIX_CURRENT=1 FRONTEND[5:4] LNA_CURRENT[1:0] 1 - Controls current in the LNA Recommended setting: LNA_CURRENT=3. FRONTEND[3] FRONTEND[2] FRONTEND[1] FRONTEND[0] MIX_CURRENT LNA2_CURRENT SDC_CURRENT LNAMIX_BIAS 0 0 0 1 - Can be lowered to save power, at the expense of reduced sensitivity. Controls current in the mixer - Recommended setting: MIX_CURRENT=1 at 402-470 MHz, MIX_CURRENT=0 at 804-940 MHz.. Controls current in LNA 2 - Recommended settings: LNA2_CURRENT=0 at 402-470 MHz, LNA2_CURRENT=1 at 804-940 MHz. Controls current in the single-to-diff. converter - Recommended settings: SDC_CURRENT=0 at 402-470 MHz, SDC_CURRENT=1 at 804-940 MHz. Controls how front-end bias currents are generated 0: Constant current biasing 1: Constant Gm·R biasing (reduces gain variation) Recommended setting: LNAMIX_BIAS=0. SWRS045 Page 77 of 91 CC1021 ANALOG Register (17h) REGISTER NAME Active BANDSELECT Default value 1 ANALOG[7] ANALOG[6] LO_DC 1 - - Description Frequency band selection 0: 402-470 MHz band 1: 804-940 MHz band Lower LO DC level to mixers 0: High LO DC level to mixers 1: Low LO DC level to mixers ANALOG[5] VGA_BLANKING 1 H ANALOG[4] PD_LONG 0 H ANALOG[3] ANALOG[2] PA_BOOST 0 0 H ANALOG[1:0] DIV_BUFF_CURRENT[1:0] 3 - Recommended settings: LO_DC=1 for 402-470 MHz, LO_DC=0 for 804-940 MHz. Enable analog blanking switches in VGA when changing VGA gain. 0: Blanking switches are disabled 1: Blanking switches are turned on for approx. 0.8µs when gain is changed (always on if AGC_DISABLE=1) Recommended setting: VGA_BLANKING=0. Selects short or long reset delay in phase detector 0: Short reset delay 1: Long reset delay Recommended setting: PD_LONG=0. Reserved, write 0 Boost PA bias current for higher output power Recommended setting: PA_BOOST=1. Overall bias current adjustment for VCO divider and buffers 0: 4/6 of nominal VCO divider and buffer current 1: 4/5 of nominal VCO divider and buffer current 2: Nominal VCO divider and buffer current 3: 4/3 of nominal VCO divider and buffer current Recommended setting: DIV_BUFF_CURRENT=3 BUFF_SWING Register (18h) REGISTER NAME Active PRE_SWING[1:0] Default value 3 BUFF_SWING[7:6] BUFF_SWING[5:3] RX_SWING[2:0] 4 - BUFF_SWING[2:0] TX_SWING[2:0] 1 - - Description Prescaler swing. Fractions for PRE_CURRENT=0: 0: 2/3 of nominal swing 1: 1/2 of nominal swing 2: 4/3 of nominal swing 3: Nominal swing Recommended setting: PRE_SWING=0 LO buffer swing, in RX (to mixers) 0: Smallest load resistance (smallest swing) Ö 7: Largest load resistance (largest swing) Recommended setting: RX_SWING=2. LO buffer swing, in TX (to power amplifier driver) 0: Smallest load resistance (smallest swing) Ö 7: Largest load resistance (largest swing) Recommended settings: TX_SWING=4 for 402-470 MHz, TX_SWING=0 for 804-940 MHz. SWRS045 Page 78 of 91 CC1021 BUFF_CURRENT Register (19h) REGISTER NAME Active PRE_CURRENT[1:0] Default value 1 BUFF_CURRENT[7:6] BUFF_CURRENT[5:3] RX_CURRENT[2:0] 4 - BUFF_CURRENT[2:0] TX_CURRENT[2:0] 5 - - Description Prescaler current scaling 0: Nominal current 1: 2/3 of nominal current 2: 1/2 of nominal current 3: 2/5 of nominal current Recommended setting: PRE_CURRENT=0. LO buffer current, in RX (to mixers) 0: Minimum buffer current Ö 7: Maximum buffer current Recommended setting: RX_CURRENT=4. LO buffer current, in TX (to PA driver) 0: Minimum buffer current Ö 7: Maximum buffer current Recommended settings: TX_CURRENT=2 for 402-470 MHz, TX_CURRENT=5 for 804-940 MHz. PLL_BW Register (1Ah) REGISTER PLL_BW[7:0] NAME Default value 134 PLL_BW[7:0] Active Description - Charge pump current scaling/rounding factor. Used to calibrate charge pump current for the desired PLL loop bandwidth. The value is given by: PLL_BW = 174 + 16 log2(fref/7.126) where fref is the reference frequency in MHz. CALIBRATE Register (1Bh) REGISTER NAME Active CAL_START Default value 0 CALIBRATE[7] CALIBRATE[6] CAL_DUAL 0 H CALIBRATE[5:4] CAL_WAIT[1:0] 0 - CALIBRATE[3] CALIBRATE[2:0] CAL_ITERATE[2:0] 0 5 ↑ - Description ↑ 1: Calibration started 0: Calibration inactive Use calibration results for both frequency A and B 0: Store results in A or B defined by F_REG (MAIN[6]) 1: Store calibration results in both A and B Selects calibration wait time (affects accuracy) 0 (00): Calibration time is approx. 90000 F_REF periods 1 (01): Calibration time is approx. 110000 F_REF periods 2 (10): Calibration time is approx. 130000 F_REF periods 3 (11): Calibration time is approx. 200000 F_REF periods Recommended setting: CAL_WAIT=3 for best accuracy in calibrated PLL loop filter bandwidth. Reserved, write 0 Iteration start value for calibration DAC 0 (000): DAC start value 1, VC<0.49 V after calibration 1 (001): DAC start value 2, VC<0.66 V after calibration 2 (010): DAC start value 3, VC<0.82 V after calibration 3 (011): DAC start value 4, VC<0.99 V after calibration 4 (100): DAC start value 5, VC<1.15 V after calibration 5 (101): DAC start value 6, VC<1.32 V after calibration 6 (110): DAC start value 7, VC<1.48 V after calibration 7 (111): DAC start value 8, VC<1.65 V after calibration Recommended setting: CAL_ITERATE=4. SWRS045 Page 79 of 91 CC1021 PA_POWER Register (1Ch) REGISTER NAME Active PA_HIGH [3:0] Default value 0 PA_POWER[7:4] PA_POWER[3:0] PA_LOW[3:0] 15 - Description - Controls output power in high-power array 0: High-power array is off 1: Minimum high-power array output power Ö 15: Maximum high-power array output power Controls output power in low-power array 0: Low-power array is off 1: Minimum low-power array output power Ö 15: Maximum low-power array output power It is more efficient in terms of current consumption to use either the lower or upper 4-bits in the PA_POWER register to control the power. MATCH Register (1Dh) REGISTER NAME Default value Active MATCH[7:4] RX_MATCH[3:0] 0 - MATCH[3:0] TX_MATCH[3:0] 0 - Description Selects matching capacitor array value for RX. Each step is approximately 0.4 pF. Selects matching capacitor array value for TX. Each step is approximately 0.4 pF. PHASE_COMP Register (1Eh) REGISTER NAME PHASE_COMP[7:0] PHASE_COMP[7:0] Default value 0 Active - Description Signed compensation value for LO I/Q phase error. Used for image rejection calibration. 128: approx. 6.2∞adjustment between I and Q phase 1: approx. 0.02∞adjustment between I and Q phase 0: approx. +0.02∞adjustment between I and Q phase 127: approx. +6.2∞adjustment between I and Q phase GAIN_COMP Register (1Fh) REGISTER NAME GAIN_COMP[7:0] GAIN_COMP[7:0] Default value 0 Active Active H H H H H - Description Signed compensation value for mixer I/Q gain error. Used for image rejection calibration. 128: approx. 1.16 dB adjustment between I and Q gain 1: approx. 0.004 dB adjustment between I and Q gain 0: approx. +0.004 dB adjustment between I and Q gain 127: approx. +1.16 dB adjustment between I and Q gain POWERDOWN Register (20h) REGISTER NAME POWERDOWN[7] POWERDOWN[6] POWERDOWN[5] PA_PD VCO_PD BUFF_PD Default value 0 0 0 POWERDOWN[4] POWERDOWN[3] POWERDOWN[2] POWERDOWN[1] POWERDOWN[0] CHP_PD LNAMIX_PD VGA_PD FILTER_PD ADC_PD 0 0 0 0 0 H H H Description Sets PA in power-down when PD_MODE[1:0]=2 Sets VCO in power-down when PD_MODE[1:0]=2 Sets VCO divider, LO buffers and prescaler in power-down when PD_MODE[1:0]=2 Sets charge pump in power-down when PD_MODE[1:0]=2 Sets LNA/mixer in power-down when PD_MODE[1:0]=2 Sets VGA in power-down when PD_MODE[1:0]=2 Sets image filter in power-down when PD_MODE[1:0]=2 Sets ADC in power-down when PD_MODE[1:0]=2 SWRS045 Page 80 of 91 CC1021 TEST1 Register (21h, for test only) REGISTER NAME Active CAL_DAC_OPEN[3:0] Default value 4 TEST1[7:4] TEST1[3:0] CHP_CO[3:0] 13 - Active - Description Calibration DAC override value, active when BREAK_LOOP=1 Charge pump current override value TEST2 Register (22h, for test only) REGISTER NAME TEST2[7] BREAK_LOOP Default value 0 TEST2[6] CHP_OVERRIDE 0 H TEST2[5] VCO_OVERRIDE 0 H TEST2[4:0] VCO_AO[4:0] 16 - Default value 0 0 Active H Description 0: PLL loop closed 1: PLL loop open 0: use calibrated value 1: use CHP_CO[3:0] value 0: use calibrated value 1: use VCO_AO[4:0] value VCO_ARRAY override value TEST3 Register (23h, for test only) REGISTER NAME TEST3[7] TEST3[6] VCO_CAL_MANUAL VCO_CAL_OVERRIDE TEST3[5:0] VCO_CO[5:0] H H 6 - Active H H H Description Enables ìmanualî VCO calibration (test only) Override VCO current calibration 0: Use calibrated value 1: Use VCO_CO[5:0] value VCO_CAL_OVERRIDE controls VCO_CAL_CLK if VCO_CAL_MANUAL=1. Negative transitions are then used to sample VCO_CAL_COMP. VCO_CAL_CURRENT override value TEST4 Register (24h, for test only) REGISTER NAME TEST4[7] TEST4[6] TEST4[5] TEST4[4:3] CHP_DISABLE CHP_TEST_UP CHP_TEST_DN TM_IQ[1:0] Default value 0 0 0 0 TEST4[2] TEST4[1] TEST4[0] TM_ENABLE TF_ENABLE TA_ENABLE 0 0 0 H H H - Description Disable normal charge pump operation Force charge pump to output ìupî current Force charge pump to output ìdownî current Value of differential I and Q outputs from mixer when TM_ENABLE=1 0: I output negative, Q output negative 1: I output negative, Q output positive 2: I output positive, Q output negative 3: I output positive, Q output positive Enable DC control of mixer output (for testing) Connect analog test module to filter inputs Connect analog test module to ADC inputs If TF_ENABLE=1 or TA_ENABLE=1 in TEST4 register, then INTERFACE[3:0] controls analog test module: INTERFACE[3] = TEST_PD, INTERFACE[2:0] = TEST_MODE[2:0]. Otherwise, TEST_PD=1 and TEST_MODE[2]=1. TEST5 Register (25h, for test only) REGISTER NAME Active F_COMP_ENABLE Default value 0 TEST5[7] TEST5[6] TEST5[5] SET_DITHER_CLOCK ADC_TEST_OUT 1 0 H H TEST5[4] TEST5[3] TEST5[2] TEST5[1:0] CHOP_DISABLE SHAPING_DISABLE VCM_ROT_DISABLE ADC_ROTATE[1:0] 0 0 0 0 H H H - H SWRS045 Description Enable frequency comparator output F_COMP from phase detector Enable dithering of delta-sigma clock Outputs ADC samples on LOCK and DIO, while ADC_CLK is output on DCLK Disable chopping in ADC integrators Disable ADC feedback mismatch shaping Disable rotation for VCM mismatch shaping Control ADC input rotation 0: Rotate in 00 01 10 11 sequence 1: Rotate in 00 10 11 01 sequence 2: Always use 00 position 3: Rotate in 00 10 00 10 sequence Page 81 of 91 CC1021 TEST6 Register (26h, for test only) REGISTER NAME Active VGA_OVERRIDE AC1O Default value 0 0 0 TEST6[7:4] TEST6[3] TEST6[2] TEST6[1:0] AC2O[1:0] 0 - Default value 0 0 0 0 Active - Description Reserved, write 0 Override VGA settings Override value to first AC coupler in VGA 0: Approx. 0 dB gain 1: Approx. 12 dB gain Override value to second AC coupler in VGA 0: Approx. 0 dB gain 1: Approx. 3 dB gain 2: Approx. 12 dB gain 3: Approx. 15 dB gain TEST7 Register (27h, for test only) REGISTER NAME TEST7[7:6] TEST7[5:4] TEST7[3:2] TEST7[1:0] VGA1O[1:0] VGA2O[1:0] VGA3O[1:0] - Description Reserved, write 0 Override value to VGA stage 1 Override value to VGA stage 2 Override value to VGA stage 3 STATUS Register (40h, read only) REGISTER NAME Active CAL_COMPLETE Default value - STATUS[7] STATUS[6] SEQ_ERROR - H STATUS[5] STATUS[4] LOCK_INSTANT LOCK_CONTINUOUS - H H STATUS[3] STATUS[2] STATUS[1] STATUS[0] CARRIER_SENSE LOCK DCLK DIO - H H H H H Description Set to 0 when PLL calibration starts, and set to 1 when calibration has finished Set to 1 when PLL failed to lock during automatic powerup sequencing Instantaneous PLL lock indicator PLL lock indicator, as defined by LOCK_ACCURACY. Set to 1 when PLL is in lock Carrier sense when RSSI is above CS_LEVEL Logical level on LOCK pin Logical level on DCLK pin Logical level on DIO pin RESET_DONE Register (41h, read only) REGISTER NAME Active ADC_RESET_DONE AGC_RESET_DONE GAUSS_RESET_DONE AFC_RESET_DONE BITSYNC_RESET_DONE Default value - RESET_DONE[7] RESET_DONE[6] RESET_DONE[5] RESET_DONE[4] RESET_DONE[3] RESET_DONE[2] SYNTH_RESET_DONE - H RESET_DONE[1] RESET_DONE[0] SEQ_RESET_DONE CAL_LOCK_RESET_DONE - H H H H H H H Description Reset of ADC control logic done Reset of AGC (VGA control) logic done Reset of Gaussian data filter done Reset of AFC / FSK decision level logic done Reset of modulator, bit synchronization logic and PN9 PRBS generator done Reset digital part of frequency synthesizer done Reset of power-up sequencing logic done Reset of calibration logic and lock detector done RSSI Register (42h, read only) REGISTER NAME RSSI[7] RSSI[6:0] RSSI[6:0] Default value - Active - Description Not in use, will read 0 Received signal strength indicator. The relative power is given by RSSI x 1.5 dB in a logarithmic scale. The VGA gain set by VGA_SETTING must be taken into account. See page 33 for more details. SWRS045 Page 82 of 91 CC1021 AFC Register (43h, read only) REGISTER NAME AFC[7:0] AFC[7:0] Default value - Active Description - Average received frequency deviation from IF. This 8-bit 2complement signed value equals the demodulator decision level and can be used for AFC. The average frequency offset from the IF frequency is ∆F = Baud rate ∑AFC / 16 GAUSS_FILTER Register (44h) REGISTER NAME GAUSS_FILTER[7:0] GAUSS_FILTER[7:0] Default value - Active - Description Readout of instantaneous IF frequency offset from nominal IF. Signed 8-bit value. ∆F = Baud rate ∑GAUSS_FILTER/8 STATUS1 Register (45h, for test only) REGISTER NAME STATUS1[7:4] STATUS1[3:0] CAL_DAC[3:0] CHP_CURRENT[3:0] Default value - Active Description - Status vector defining applied Calibration DAC value Status vector defining applied CHP_CURRENT value STATUS2 Register (46h, for test only) REGISTER NAME Active CC1021_VERSION[2:0] Default value - STATUS2[7:5] STATUS2[4:0] VCO_ARRAY[4:0] - - - Description CC1021 version code: 0: Pre-production version 1: First production version 2-7: Reserved for future use Status vector defining applied VCO_ARRAY value STATUS3 Register (47h, for test only) REGISTER NAME STATUS3[7] STATUS3[6] STATUS3[5:0] Active F_COMP Default value - VCO_CAL_COMP - - VCO_CAL_CURRENT[5:0] - - - Description Frequency comparator output from phase detector Readout of VCO current calibration comparator. Equals 1 if current defined by VCO_CURRENT_A/B is larger than the VCO core current Status vector defining applied VCO_CAL_CURRENT value STATUS4 Register (48h, for test only) REGISTER NAME STATUS4[7:6] STATUS4[5:3] STATUS4[2:0] ADC_MIX[1:0] ADC_I[2:0] ADC_Q[2:0] Default value - Active Default value - Active Default value - Active - Description Readout of mixer input to ADC Readout of ADC ìIî output Readout of ADC ìQî output STATUS5 Register (49h, for test only) REGISTER NAME STATUS5[7:0] FILTER_I[7:0] - Description Upper bits of ìIî output from channel filter STATUS6 Register (4Ah, for test only) REGISTER NAME STATUS6[7:0] FILTER_Q[7:0] - SWRS045 Description Upper bits of ìQî output from channel filter Page 83 of 91 CC1021 STATUS7 Register (4Bh, for test only) REGISTER NAME STATUS7[7:5] STATUS7[4:0] VGA_GAIN_OFFSET[4:0] Default value - Active SWRS045 - Description Not in use, will read 0 Readout of offset between VGA_SETTING and actual VGA gain set by AGC Page 84 of 91 CC1021 27. Package Description (QFN 32) SWRS045 Page 85 of 91 CC1021 Quad Flat Pack - No Lead Package (QFN) E A A1 e b L QFN 32 Min 0.8 0.25 0.45 7.0 7.0 0.9 0.203 0.65 0.30 0.55 Max 1.0 0.35 0.65 All dimensions in mm. Angles are in degrees. D D1 4.18 4.28 4.38 E1 4.18 4.28 4.38 P 45° Package is compliant with JEDEC: MO-220. Note: Do not place a via underneath at ìpin #1 cornerî as this pin is internally connected to the exposed die attached pad, which is the main ground connection for the chip. CC1021 27.1. Package Marking When contacting technical support with a chip-related question, please state the entire marking information, not just the date code. Standard leaded 0409123 0409 is the date code (year 04, week 09) 123 is the lot code RoHS compliant Pb-free A409123 409 is the date code (year 4, week 09) 123 is the lot code A means RoHS compliant Pb-free SWRS045 Page 86 of 91 CC1021 27.2. Recommended PCB Footprint for Package (QFN 32) Note: The figure is an illustration only and not to scale. There are nine 14 mil (0.36 mm) diameter via holes distributed symmetrically in the ground plane under the package. See also the CC1020EMX reference design. 27.3. Package Thermal Properties Air velocity [m/s] Rth,j-a [K/W] Thermal resistance 0 1 21.4 18.9 2 17.0 27.4. Soldering Information Recommended soldering profile for both standard leaded packages and Pb-free packages is according to IPC/JEDEC J-STD-020C. SWRS045 Page 87 of 91 CC1021 27.5. Plastic Tube Specification QFN 7x7 mm antistatic tube. Package QFN 32 Tube Specification Tube Height Tube Length 2.2 +0.2/-0.1 mm 315 ± 1.25 mm Tube Width 8.5 ± 0.2 mm Units per Tube 43 27.6. Carrier Tape and Reel Specification Carrier tape and reel is in accordance with EIA Specification 481. 28. Package Tape Width QFN 32 16 mm Tape and Reel Specification Component Hole Reel Pitch Pitch Diameter 12 mm 4 mm 13î Units per Reel 4000 Ordering Information Ordering part number Description 1135 CC1021-RTB1 1136 CC1021-RTR1 1115 CC1020_1070DK-433 CC1021, QFN32 package, standard leaded assembly, tubes with 43 pcs per tube, Single Chip RF Transceiver. CC1021, QFN32 package, RoHS compliant Pb-free assembly, T&R with 4000 pcs per reel, Single Chip RF Transceiver. CC1020_1070 Development Kit, 433 MHz MOQ 1116 CC1020_1070DK-868/915 CC1020/1070 Development Kit, 868/915 MHz 1 1159 CC1021SK RoHS CC1021 Sample Kit, QFN32 package, RoHS compliant Pbfree assembly, 5 pcs 1 43 4000 1 MOQ = Minimum Order Quantity T&R = tape and reel Note: The CC1020/1070DK Development Kit with a fully assembled CC1020EMX Evaluation transceiver. Module should be used for evaluation of the CC1021 SWRS045 Page 88 of 91 CC1021 29. General Information Document Revision History Revision Date 1.0 1.1 February 2004 December 2004 1.2 October 2005 1.3 January 2006 Product Status Definitions Data Sheet Identification Description/Changes First edition The various sections have been reorganized to improve readability Added chapter numbering Reorganized electrical specification section Electrical specifications updated Changes to sensitivity figures Changes to TX spurious emission and harmonics figures Changes to current consumption figures in RX and TX mode and crystal oscillator, bias and synthesizer mode Changes to noise figure Added figures on modulation bandwidth Updates to section on input / output matching Updates to section on VCO and PLL self-calibration Updates to section on VCO, charge pump and PLL loop filter Updates to section on receiver channel filter bandwidth Updates to section on RSSI Updates to section on image rejection calibration Updates to section on preamble length and sync word Description of OOK modulation and demodulation merged into one section New bill of materials for operation at 433 MHz and 868/915 MHz Added recommended PCB footprint for package (QFN 32) Added information that there should be no via at ìpin #1 cornerî (section 27.2) Added list of abbreviations Changes to ordering information RSSI dynamic range changed from 63 dB to 55 dB Recommended CAL_ITERATE changed from 5 to 4 PLL timeout in ìAutomatic power-up sequencing flow chartî changed from 1024 filter clocks to 127 filter clocks Calibration routine flow chart changed in accordance to CC1021 Errata Note 002 Added chapter on TX data latency Updates to Ordering Information and Address Information Product Status Advance Information Planned or Under Development Preliminary Engineering Samples and First Production No Identification Noted Full Production Obsolete Not In Production Definition This data sheet contains the design specifications for product development. Specifications may change in any manner without notice. This data sheet contains preliminary data, and supplementary data will be published at a later date. Chipcon reserves the right to make changes at any time without notice in order to improve design and supply the best possible product. This data sheet contains the final specifications. Chipcon reserves the right to make changes at any time without notice in order to improve design and supply the best possible product. This data sheet contains specifications on a product that has been discontinued by Chipcon. The data sheet is printed for reference information only. SWRS045 Page 89 of 91 CC1021 Disclaimer Chipcon AS believes the information contained herein is correct and accurate at the time of this printing. However, Chipcon AS reserves the right to make changes to this product without notice. Chipcon AS does not assume any responsibility for the use of the described product; neither does it convey any license under its patent rights, or the rights of others. The latest updates are available at the Chipcon website or by contacting Chipcon directly. To the extent possible, major changes of product specifications and functionality will be stated in product specific Errata Notes published at the Chipcon website. Customers are encouraged to sign up for the Developerís Newsletter for the most recent updates on products and support tools. When a product is discontinued this will be done according to Chipconís procedure for obsolete products as described in Chipconís Quality Manual. This includes informing about last-time-buy options. The Quality Manual can be downloaded from Chipconís website. Compliance with regulations is dependent on complete system performance. It is the customerís responsibility to ensure that the system complies with regulations. Trademarks SmartRF is a registered trademark of Chipcon AS. SmartRF is Chipcon's RF technology platform with RF library cells, modules and design expertise. Based on SmartRF technology Chipcon develops standard component RF circuits as well as full custom ASICs based on customer requirements and this technology. All other trademarks, registered trademarks and product names are the sole property of their respective owners. Life Support Policy This Chipcon product is not designed for use in life support appliances, devices, or other systems where malfunction can reasonably be expected to result in significant personal injury to the user, or as a critical component in any life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. Chipcon AS customers using or selling these products for use in such applications do so at their own risk and agree to fully indemnify Chipcon AS for any damages resulting from any improper use or sale. © 2006, Chipcon AS. All rights reserved. SWRS045 Page 90 of 91 SmartRF CC1021 30. 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