AD AD8370 Lf to 750 mhz, digitally controlled vga Datasheet

LF to 750 MHz,
Digitally Controlled VGA
AD8370
FEATURES
FUNCTIONAL BLOCK DIAGRAM
PWUP
4
ICOM
2
INHI
1
GENERAL DESCRIPTION
The AD8370 is a low cost, digitally controlled, variable gain
amplifier (VGA) that provides precision gain control, high IP3,
and low noise figure. The excellent distortion performance and
wide bandwidth make the AD8370 a suitable gain control
device for modern receiver designs.
For wide input, dynamic range applications, the AD8370
provides two input ranges: high gain mode and low gain mode.
A vernier, 7-bit, transconductance (gm) stage provides 28 dB of
gain range at better than 2 dB resolution and 22 dB of gain
range at better than 1 dB resolution. A second gain range, 17 dB
higher than the first, can be selected to provide improved noise
performance.
The AD8370 is powered on by applying the appropriate logic
level to the PWUP pin. When powered down, the AD8370
consumes less than 4 mA and offers excellent input to output
isolation. The gain setting is preserved when operating in a
power-down mode.
VCCO
VCCO
3
11
6
BIAS CELL
PRE
AMP
TRANSCONDUCTANCE
5
VOCM
7
OCOM
8
OPHI
9
OPLO
OUTPUT
AMP
INLO 16
10 OCOM
ICOM 15
SHIFT REGISTER
AND LATCHES
13
03692-001
AD8370
14
12
DATA CLCK LTCH
Figure 1.
40
70
APPLICATIONS
CODE = LAST 7 BITS OF GAIN CODE
(NO MSB)
30
60
20
50
LOW GAIN MODE
40
Δ GAIN ≅ 0.409
Δ CODE
HIGH GAIN MODE
30
10
0
–10
20
10
LOW GAIN MODE
0
0
10
20
30
40
50 60 70 80
GAIN CODE
Δ GAIN ≅ 0.059
Δ CODE
–20
–30
90 100 110 120 130
VOLTAGE GAIN (dB)
HIGH GAIN MODE
VOLTAGE GAIN (V/V)
Differential ADC drivers
IF sampling receivers
RF/IF gain stages
Cable and video applications
SAW filter interfacing
Single-ended-to-differential conversion
VCCI
03692-002
Programmable low and high gain (<2 dB resolution)
Low range: −11 dB to +17 dB
High range: 6 dB to 34 dB
Differential input and output
200 Ω differential input
100 Ω differential output
7 dB noise figure @ maximum gain
Two-tone IP3 of 35 dBm @ 70 MHz
−3 dB bandwidth of 750 MHz
40 dB precision gain range
Serial 8-bit digital interface
Wide input dynamic range
Power-down feature
Single 3 V to 5 V supply
Figure 2. Gain vs. Gain Code at 70 MHz
Gain control of the AD8370 is through a serial 8-bit gain control
word. The MSB selects between the two gain ranges, and the
remaining 7 bits adjust the overall gain in precise linear gain steps.
Fabricated on the ADI high speed XFCB process, the high
bandwidth of the AD8370 provides high frequency and low
distortion. The quiescent current of the AD8370 is 78 mA
typically. The AD8370 amplifier comes in a compact, thermally
enhanced 16-lead TSSOP package and operates over the
temperature range of −40°C to +85°C.
Rev. A
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
© 2005 Analog Devices, Inc. All rights reserved.
AD8370
TABLE OF CONTENTS
Features .............................................................................................. 1
Basic Connections...................................................................... 15
Applications....................................................................................... 1
Gain Codes.................................................................................. 15
General Description ......................................................................... 1
Power-Up Feature....................................................................... 15
Functional Block Diagram .............................................................. 1
Choosing Between Gain Ranges .............................................. 16
Revision History ............................................................................... 2
Layout and Operating Considerations .................................... 16
Specifications..................................................................................... 3
Package Considerations............................................................. 17
Absolute Maximum Ratings............................................................ 5
Single-Ended-to-Differential Conversion............................... 17
ESD Caution.................................................................................. 5
DC-Coupled Operation............................................................. 18
Pin Configuration and Function Descriptions............................. 6
ADC Interfacing ......................................................................... 19
Typical Performance Characteristics ............................................. 7
3 V Operation ............................................................................. 20
Theory of Operation ...................................................................... 13
Evaluation Board and Software .................................................... 22
Block Architecture...................................................................... 13
Appendix ......................................................................................... 25
Preamplifier................................................................................. 13
Characterization Equipment..................................................... 25
Transconductance Stage ............................................................ 13
Composite Waveform Assumption.......................................... 25
Output Amplifier........................................................................ 14
Definitions of Selected Parameters .......................................... 25
Digital Interface and Timing .................................................... 14
Outline Dimensions ....................................................................... 28
Applications..................................................................................... 15
Ordering Guide .......................................................................... 28
REVISION HISTORY
7/05—Rev. 0 to Rev. A
Changes to Features.......................................................................... 1
Changes to Table 1............................................................................ 3
Changes to Figure 11 and Figure 15............................................... 8
Added Figure 12; Renumbered Sequentially ................................ 8
Added Figure 16; Renumbered Sequentially ................................ 9
Changes to Evaluation Board and Software Section.................. 22
Changes to Figure 60...................................................................... 23
Updated Outline Dimensions ....................................................... 28
Changes to Ordering Guide .......................................................... 28
1/04—Revision 0: Initial Version
Rev. A | Page 2 of 28
AD8370
SPECIFICATIONS
VS = 5 V, T = 25°C, ZS = 200 Ω, ZL = 100 Ω at gain code HG127, 70 MHz, 1 V p-p differential output, unless otherwise noted.
Table 1.
Parameter
DYNAMIC PERFORMANCE
−3 dB Bandwidth
Slew Rate
INPUT STAGE
Maximum Input
Input Resistance
Common-Mode Input Range
CMRR
Input Noise Spectral Density
GAIN
Maximum Voltage Gain
High Gain Mode
Conditions
Min
VOUT < 1 V p-p
Gain Code HG127, RL = 1 kΩ, AD8370 in compression
Gain Code LG127, RL = 1 kΩ, VOUT = 2 V p-p
Pins INHI and IHLO
Gain Code LG2, 1 dB compression
Differential
Differential, f = 10 MHz, Gain Code LG127
Gain Code = HG127
Low Gain Mode
Gain Code = LG127
Minimum Voltage Gain
High Gain Mode
Gain Code = HG1
Low Gain Mode
Gain Code = LG1
Gain Step Size
High Gain Mode
Low Gain Mode
Gain Code = HG127
For 6 dB gain step, settled to 10% of final value
Pins OPHI and OPLO
RL ≥ 1 kΩ (1 dB compression)
Differential
VINHI = VINLO, over all gain codes
Gain Temperature Sensitivity
Step Response
OUTPUT INTERFACE
Output Voltage Swing
Output Resistance
Output Differential Offset
NOISE/HARMONIC PERFORMANCE
10 MHz
Gain Flatness
Noise Figure
Second Harmonic 1
Third Harmonic1
Output IP3
Output 1 dB Compression Point
70 MHz
Gain Flatness
Noise Figure
Second Harmonic1
Third Harmonic1
Output IP3
Output 1 dB Compression Point
Within ±10 MHz of 10 MHz
VOUT = 2 V p-p
VOUT = 2 V p-p
Within ±10 MHz of 70 MHz
VOUT = 2 V p-p
VOUT = 2 V p-p
Rev. A | Page 3 of 28
Typ
Max
Unit
750
5750
3500
MHz
V/ns
V/ns
3.2
200
3.2
77
1.9
V p-p
Ω
V p-p
dB
nV/√Hz
34
52
17
7.4
dB
V/V
dB
V/V
−8
0.4
−25
0.06
0.408
0.056
–2
20
dB
V/V
dB
V/V
(V/V)/Code
(V/V)/Code
mdB/°C
ns
8.4
95
±60
V p-p
Ω
mV
±0.01
7.2
−77
−77
35
17
dB
dB
dBc
dBc
dBm
dBm
±0.02
7.2
−65
−62
35
17
dB
dB
dBc
dBc
dBm
dBm
AD8370
Parameter
140 MHz
Gain Flatness
Noise Figure
Second Harmonic1
Third Harmonic1
Output IP3
Output 1 dB Compression Point
190 MHz
Gain Flatness
Noise Figure
Second Harmonic1
Third Harmonic1
Output IP3
Output 1 dB Compression Point
240 MHz
Gain Flatness
Noise Figure
Second Harmonic1
Third Harmonic1
Output IP3
Output 1 dB Compression Point
380 MHz
Gain Flatness
Noise Figure
Output IP3
Output 1 dB Compression Point
POWER-INTERFACE
Supply Voltage
Quiescent Current 3
vs. Temperature 4
Total Supply Current
Power-Down Current
vs. Temperature4
POWER-UP INTERFACE
Power-Up Threshold4
Power-Down Threshold4
PWUP Input Bias Current
GAIN CONTROL INTERFACE
VIH4
VIL4
Input Bias Current
Conditions
Min
Within ±10 MHz of 140 MHz
VOUT = 2 V p-p
VOUT = 2 V p-p
Within ±10 MHz of 240 MHz
VOUT = 2 V p-p
VOUT = 2 V p-p
Within ±10 MHz of 240 MHz
VOUT = 2 V p-p
VOUT = 2 V p-p
Within ±10 MHz of 240 MHz
PWUP High, GC = LG127, RL = ∞, 4 seconds after
power-on, thermal connection made to exposed
paddle under device
−40°C ≤ TA ≤ +85°C
PWUP High, VOUT = 1 V p-p, ZL = 100 Ω reactive,
GC = LG127 (includes load current)
PWUP low
−40°C ≤TA ≤ +85°C
Pin PWUP
Voltage to enable the device
Voltage to disable the device
PWUP = 0 V
Pins CLCK, DATA, and LTCH
Voltage for a logic high
Voltage for a logic low
1
Refer to Figure 22 for performance into a lighter load.
See the 3 V Operation section for more information.
Minimum and maximum specified limits for this parameter are guaranteed by production test.
4
Minimum or maximum specified limit for this parameter is a 6-sigma value and not guaranteed by production test.
2
3
Rev. A | Page 4 of 28
3.0 2
72.5
Typ
Max
Unit
±0.03
7.2
−54
−50
33
17
dB
dB
dBc
dBc
dBm
dBm
±0.03
7.2
−43
−43
33
17
dB
dB
dBc
dBc
dBm
dBm
±0.04
7.4
–28
–33
32
17
dB
dB
dBc
dBc
dBm
dBm
±0.04
8.1
27
14
dB
dB
dBm
dBm
79
5.5
85.5
V
mA
105
mA
mA
82
3.7
5
1.8
mA
mA
0.8
400
V
V
nA
0.8
900
V
V
nA
1.8
AD8370
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
Supply Voltage, VS
PWUP, DATA, CLCK, LTCH
Differential Input Voltage,
VINHI – VINLO
Common-Mode Input Voltage,
VINHI or VINLO, with Respect to
ICOM or OCOM
Internal Power Dissipation
θJA (Exposed Paddle Soldered Down)
θJA (Exposed Paddle Not Soldered Down)
θJC (At Exposed Paddle)
Maximum Junction Temperature
Operating Temperature Range
Storage Temperature Range
Lead Temperature Range
(Soldering 60 sec)
Rating
5.5 V
VS + 500 mV
2V
VS + 500 mV (max),
VICOM – 500 mV,
VOCOM – 500 mV (min)
575 mW
30°C/W
95°C/W
9°C/W
150°C
−40°C to +85°C
−65°C to +150°C
235°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those listed in the operational sections
of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate
on the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. A | Page 5 of 28
AD8370
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
INHI 1
16 INLO
ICOM 2
15 ICOM
VCCI 3
AD8370
14 DATA
PWUP 4
VCCO 6
11 VCCO
OCOM 7
10 OCOM
OPHI 8
9
OPLO
03692-003
13 CLCK
TOP VIEW
VOCM 5 (Not to Scale) 12 LTCH
Figure 3.16-Lead TSSOP
Table 3. Pin Function Descriptions
Pin No.
1
2, 15, PADDLE
Mnemonic
INHI
ICOM
3
4
5
VCCI
PWUP
VOCM
6, 11
7, 10
8
9
12
VCCO
OCOM
OPHI
OPLO
LTCH
13
14
16
CLCK
DATA
INLO
Description
Balanced Differential Input. Internally biased.
Input Common. Connect to a low impedance ground. This node is also connected to the exposed pad
on the bottom of the device.
Input Positive Supply. 3.0 V to 5.5 V. Should be properly bypassed.
Power Enable Pin. Device is operational when PWUP is pulled high.
Common-Mode Output Voltage Pin. The midsupply ((VVCCO − VOCOM)/2) common-mode voltage is delivered
to this pin for external bypassing for additional common-mode supply decoupling. This can be achieved
with a bypass capacitor to ground. This pin is an output only and is not to be driven externally.
Output Positive Supply. 3.0 V to 5.5 V. Should be properly bypassed.
Output Common. Connect to a low impedance ground.
Balanced Differential Output. Biased to midsupply.
Balanced Differential Output. Biased to midsupply.
Serial Data Latch Pin. Serial data is clocked into the shift register via the DATA pin when LTCH is low. Data
in shift register is latched on the next high-going edge.
Serial Clock Input Pin.
Serial Data Input Pin.
Balanced Differential Input. Internally biased.
Rev. A | Page 6 of 28
AD8370
TYPICAL PERFORMANCE CHARACTERISTICS
VS = 5 V, ZS = 200 Ω, ZL = 100 Ω, T = 25°C, unless otherwise noted.
40
40
CODE = LAST 7 BITS OF GAIN CODE
(NO MSB)
HIGH GAIN CODES SHOWN WITH DASHED LINES
HG127
35
30
60
HG77
30
HIGH GAIN MODE
HG102
0
–10
20
0
0
10
20
30
40
50 60 70 80
GAIN CODE
Δ GAIN ≅ 0.059
Δ CODE
LG90
15
LG36
5
HG3
0
–20
–30
90 100 110 120 130
LOW GAIN CODES SHOWN WITH SOLID LINES
–10
10
100
30
50
40
+25°C
25
25
15
20
10
15
5
40
60
80
GAIN CODE
100
120
40
+85°C
35
25
30
20
–40°C
15
SHADING INDICATES ±3σ FROM THE
MEAN. DATA BASED ON 30 PARTS
FROM TWO BATCH LOTS.
5
45
30
0
–5
140
03692-068
OUTPUT IP3 (dBm)
20
UNIT CONVERSION NOTE FOR
100Ω LOAD: dBVrms = dBm–10dB
35
OUTPUT IP3 (dBm) +25°C
LOW GAIN MODE
OUTPUT IP3 (dBV rms)
35
20
1000
Figure 7. Frequency Response vs. Gain Code
HIGH GAIN MODE
0
LG18
LG9
–5
FREQUENCY (MHz)
40
10
HG18
HG9
10
Figure 4. Gain vs. Gain Code at 70 MHz
30
LG127
HG25
20
25
SHADING INDICATES ±3σ FROM THE
MEAN. DATA BASED ON 30 PARTS
FROM TWO BATCH LOTS.
10
0
50
100
150
200
250
300
350
20
400
FREQUENCY (MHz)
Figure 5. Output Third-Order Intercept vs. Gain Code at 70 MHz
Figure 8. Output Third-Order Intercept vs. Frequency at Maximum Gain
45
25
40
20
NOISE FIGURE (dB)
30
380MHz
25
LOW GAIN MODE
20
70MHz
LG127
15
10
HG18
15
HG127
10
HIGH GAIN MODE
70MHz
5
0
20
40
60
80
100
120
03692-009
5
380MHz
03692-006
NOISE FIGURE (dB)
35
0
140
0
GAIN CODE
100
200
300
400
500
FREQUENCY (MHz)
Figure 6. Noise Figure vs. Gain Code at 70 MHz
Figure 9. Noise Figure vs. Frequency at Various Gains
Rev. A | Page 7 of 28
600
OUTPUT IP3 (dBm) –40°C, +85°C
LOW GAIN MODE
25
03692-069
10
VOLTAGE GAIN (dB)
Δ GAIN ≅ 0.409
Δ CODE
HIGH GAIN MODE
30
10
VOLTAGE GAIN (dB)
LOW GAIN MODE
40
03692-004
VOLTAGE GAIN (V/V)
HG51
20
50
03692-007
70
AD8370
20
2.0
LOW GAIN MODE
1.5
16
HIGH GAIN MODE
1.0
GAIN ERROR (dB)
1kΩ LOAD
HIGH GAIN MODE
8
4
UNIT CONVERSION NOTE:
FOR 100Ω LOAD: dBV rms = dBm–10dB
FOR 1kΩ LOAD: dBV rms = dBm
0
–4
0
20
40
60
80
100
120
–40°C
0
+85°C
–0.5
–1.0
ERROR AT –40°C AND +85°C WITH RESPECT TO +25°C.
SHADING INDICATES ±3σ FROM THE MEAN. DATA
BASED ON 30 PARTS FROM ONE BATCH LOT.
–1.5
03692-010
SHADING INDICATES ±3σ FROM THE
MEAN. DATA BASED ON 30 PARTS
FROM TWO BATCH LOTS.
–8
0.5
03692-012
LOW GAIN MODE
12
OUTPUT P1dB (dB)
100Ω LOAD
–2.0
10
140
100
1000
FREQUENCY (MHz)
GAIN CODE
Figure 10. Output P1dB vs. Gain Code at 70 MHz
HIGH GAIN MODE
–30
–80
–40
–90
–50
–100
–60
–110
–70
–120
LOW GAIN MODE
–80
–130
–90
–140
140
0
20
40
60
80
GAIN CODE
100
120
18
+25°C, 100Ω LOAD
+85°C, 100Ω LOAD
18
16
14
16
UNIT CONVERSION NOTE:
RE 100Ω LOAD: dBV rms = dBm – 10dB
RE 1kΩ LOAD: dBV rms = dBm
–40°C, 100Ω LOAD
+25°C, 1kΩ LOAD
14
12
12
10
8
10
+85°C, 1kΩ LOAD
8 SHADING INDICATES ±3σ FROM
THE MEAN. DATA BASED ON 30
PARTS FROM TWO BATCH LOTS.
6
0
50
100
150
200
6
–40°C, 1kΩ LOAD
250
300
350
4
400
FREQUENCY (MHz)
Figure 11. Two-Tone Output IMD3 vs. Gain Code at 70 MHz,
RL = 1 kΩ, VOUT = 2 V p-p Composite Differential
30
20
15
5
LOW GAIN
MODE
10
0
5
–5
0
–10
–5
0
20
40
60
80
GAIN CODE
100
120
–15
140
OUTPUT IMD (dBc)
10
03692-005
20
OUTPUT IP3 (dBV rms)
15
HIGH GAIN
MODE
–50
–52
–54
–56
–58
–60
–62
–64
–66
–68
–70
–72
–74
–76
–78
–80
–82
–84
–40°C
+25°C
+85°C
0
50
100
150
200
250
FREQUENCY (MHz)
300
03692-014
25
25
OUTPUT IP3 (dBm)
Figure 14. Output P1dB vs. Frequency
35
350
400
Figure 15. Two-Tone Output IMD3 vs. Frequency at Maximum Gain,
RL = 1 kΩ, VOUT = 2 V p-p Composite Differential
Figure 12. Output Third-Order Intercept vs. Gain Code at 70 MHz,
RL = 1 kΩ, VOUT = 2 V p-p Composite Differential
Rev. A | Page 8 of 28
OUTPUT P1dB (dBm) +25°C
–70
OUTPUT P1dB (dBm) –40°C, +85°C
–20
HIGH GAIN MODE OUTPUT IMD (dBc)
–60
03692-011
–10
20
03692-013
–50
0
LOW GAIN MODE OUTPUT IMD (dBc)
Figure 13. Gain Error over Temperature vs. Frequency, RL = 100 Ω
AD8370
34
90
24
32
22
120
30
OUTPUT IP3 (dBV rms)
20
18
–40°C
26
16
+25°C
14
22
12
20
10
18
8
16
6
14
0
50
100
150
200
250
FREQUENCY (MHz)
300
S22
5MHz
0
330
210
S11
4
400
350
30
180
03692-008
24
150
240
300
03692-017
28
OUTPUT IP3 (dBm)
60
1GHz
+85°C
270
Figure 16. Output Third-Order Intercept vs. Frequency at Maximum Gain,
RL = 1 kΩ, VOUT = 2 V p-p Composite Differential
Figure 19. Input and Output Reflection Coefficients, S11 and S22,
ZO = 100 Ω Differential
100
250
2.0
16 DIFFERENT GAIN
CODES REPRESENTED
R+jX FORMAT
1.5
200
50
150
0
100
–50
50
–100
+85°C
–0.5
REACTANCE (j Ω)
–40°C
0
ERROR AT –40°C AND +85°C WITH RESPECT TO +25°C.
SHADING INDICATES ±3σ FROM THE MEAN. DATA
BASED ON 30 PARTS FROM ONE BATCH LOT.
–2.0
10
100
0
0
1000
100
200
300
400
500
600
–150
700
FREQUENCY (MHz)
FREQUENCY (MHz)
Figure 17. Gain Error over Temperature vs. Frequency, RL = 1 kΩ
Figure 20. Input Resistance and Reactance vs. Frequency
0
0
–10
–10
–20
LOW GAIN, RL = 1kΩ
–30
–40
HIGH GAIN, RL = 100Ω
LOW GAIN, RL = 100Ω
–50
–60
–80
03692-016
–70
HIGH GAIN, RL = 1kΩ
–90
0
20
40
60
80
100
120
HIGH GAIN RL = 1kΩ
–20
HIGH GAIN RL = 100Ω
–30
LOW GAIN RL = 100Ω
–40
–50
–60
–70
03692-019
HARMONIC DISTORTION (dBc)
LOW GAIN RL = 1kΩ
–80
–90
140
0
20
40
60
80
100
120
140
GAIN CODE
GAIN CODE
Figure 18. Second-Order Harmonic Distortion vs. Gain Code at 70 MHz,
VOUT = 2 V p-p Differential
Figure 21. Third-Order Harmonic Distortion vs. Gain Code at 70 MHz,
VOUT = 2 V p-p Differential
Rev. A | Page 9 of 28
03692-018
–1.0
–1.5
HARMONIC DISTORTION (dBc)
RESISTANCE (Ω)
0.5
03692-015
GAIN ERROR (dB)
1.0
AD8370
0
120
110
HD2 RL = 100Ω
100
–20
HD3 RL = 100Ω
–30
90
–40
–50
80
PSRR (dB)
HD3 RL = 1kΩ
70
60
–60
50
–70
40
03692-020
HD2 RL = 1kΩ
–80
–90
0
50
100
150
200
250
300
350
03692-023
HARMONIC DISTORTION (dBc)
–10
30
20
400
1
10
FREQUENCY (MHz)
100
1000
FREQUENCY (MHz)
Figure 25. Power Supply Rejection Ratio vs. Frequency at Maximum Gain
Figure 22. Harmonic Distortion vs. Frequency at Maximum Gain,
VOUT = 2 V p-p Composite Differential
120
80
100
60
0
FORWARD TRANSMISSION, HG0
–20
60
20
40
0
16 DIFFERENT GAIN
CODES REPRESENTED
R+jX FORMAT
0
0
100
200
300
400
500
600
–40
700
–80
FORWARD TRANSMISSION, PWUP LOW
REVERSE TRANSMISSION, HG127
–120
10
100
1000
FREQUENCY (MHz)
FREQUENCY (MHz)
Figure 23. Output Resistance and Reactance vs. Frequency
Figure 26. Various Forms of Isolation vs. Frequency
860
1400
840
1300
RL = 1kΩ
HIGH GAIN MODE
1200
800
780
760
LOW GAIN MODE
1100
900
800
720
700
03692-022
740
700
0
10
20
30
40
50
60
70
80
RL = 100Ω
1000
03692-025
GROUP DELAY (ps)
820
GROUP DELAY (ps)
–60
–100
–20
03692-021
20
–40
03692-024
40
ISOLATION (dB)
80
REACTANCE (j Ω)
RESISTANCE (Ω)
FORWARD TRANSMISSION, LG0
600
0
90 100 110 120 130
100
200
300
400
500
600
700
800
900
GAIN CODE
FREQUENCY (MHz)
Figure 24. Group Delay vs. Gain Code at 70 MHz
Figure 27. Group Delay vs. Frequency at Maximum Gain
Rev. A | Page 10 of 28
AD8370
80
DIFFERENTIAL OUTPUT (50mV/DIV)
70
LG32, LG127
60
ZERO
CMRR (dB)
HG32, HG127
50
40
PWUP (2V/DIV)
30
GAIN CODE HG127
03692-026
10
0
10
100
GND
INPUT = –30dBm, 70MHz 100 AVERAGES
03692-029
20
1000
FREQUENCY (MHz)
TIME (40ns/DIV)
Figure 28. Common-Mode Rejection Ratio vs. Frequency
Figure 31. PWUP Time Domain Response
12
10
ZERO
8
6dB GAIN STEP (HG36 TO LG127)
LG127
6
LTCH (2V/DIV)
4
HG127
0
10
110
210
310
410
510
GND
INPUT = –30dBm, 70MHz
NO AVERAGING
03692-030
HG18
2
03692-027
NOISE SPECTRAL DENSITY (nV/ Hz)
DIFFERENTIAL OUTPUT (10mV/DIV)
610
FREQUENCY (MHz)
TIME (20ns/DIV)
Figure 29. Input Referred Noise Spectral Density vs.
Frequency at Various Gains
Figure 32. Gain Step Time Domain Response
VOUT DIFFERENTIAL
VOPHI
VOLTAGE (1V/DIV)
DIFFERENTIAL VOUT
DIFFERENTIAL VIN
GND
03692-028
GND
03692-031
VOLTAGE (600mV/DIV)
VOPLO
TIME (2ns/DIV)
TIME (2ns/DIV)
Figure 30. DC-Coupled Large Signal Pulse Response
Figure 33. Overdrive Recovery
Rev. A | Page 11 of 28
85
2.75
80
2.70
75
2.65
70
2.60
HIGH GAIN
2.55
60
2.50
55
2.45
0
16
32
48
64
80
96
112
LOW GAIN MODE
2.40
128
0
32
64
96
HIGH GAIN MODE
0
32
64
96
128
GAIN CODE
GAIN CODE
Figure 34. Supply Current vs. Gain Code
Figure 36. Common-Mode Output Voltage vs. Gain Code at
Various Temperatures
35
MEAN: 51.9
σ: 0.518
30
25
DATA FROM 136 PARTS
FROM ONE BATCH LOT
20
15
10
03692-033
5
0
50
–40°C
03692-034
VCM (V)
LOW GAIN
65
50
COUNT
+85°C
+25°C
03692-032
SUPPLY CURRENT (mA)
AD8370
51
52
53
54
55
GAIN (V/V)
Figure 35. Distribution of Voltage Gain, HG127, 70 MHz, RL = 100 Ω
Rev. A | Page 12 of 28
AD8370
THEORY OF OPERATION
PWUP
4
ICOM
2
INHI
1
VCCO
VCCO
11
6
BIAS CELL
PRE
AMP
TRANSCONDUCTANCE
5
VOCM
7
OCOM
8
OPHI
9
OPLO
OUTPUT
AMP
INLO 16
INHI/INLO
2kΩ
VCC/2
1mA
Figure 38. INHI/INLO Simplified Schematic
TRANSCONDUCTANCE STAGE
Figure 39 shows typical input levels that can be applied to this
amplifier at different gain settings. The maximum input was
determined by finding the 1 dB compression or expansion point
of the VOUT/VSOURCE gain. Note that this is not VOUT/VIN. In this
way, the change in the input impedance of the device is also
taken into account.
3.2
10 OCOM
ICOM 15
SHIFT REGISTER
AND LATCHES
12
DATA CLCK LTCH
Figure 37. Functional Block Diagram
PREAMPLIFIER
There are two selectable input preamplifiers. Selection is made
by the most significant bit (MSB) of the serial gain control dataword. In the high gain mode, the overall device gain is 7.1 V/V
(17 dB) above the low gain setting. The two preamplifiers give
the AD8370 the ability to accommodate a wide range of input
amplitudes. The overlap between the two gain ranges allows the
user some flexibility based on noise and distortion demands.
See the Choosing Between Gain Ranges section for more
information.
Rev. A | Page 13 of 28
34dB
GAIN
2.0
LOW GAIN
<1dB
RES
2.4
VOUT [V peak] (V)
13
03692-035
14
<0.5dB
RES
2.8
AD8370
17dB
GAIN
HIGH GAIN
<0.5dB
RESOLUTION
<2dB
RES
1.6
12dB
GAIN
1.2
0.1dB GAIN
6dB
GAIN
0.8
–5dB GAIN
0.4
–8dB GAIN
<1dB
RES
<2dB
RES
03692-037
The three basic building blocks of the AD8370 are a high/low
gain selectable input preamplifier, a digitally controlled
transconductance (gm) block, and a fixed gain output stage.
3
1mA
The digitally controlled gm section has 42 dB of controllable
gain and makes gain adjustments within each gain range. The
step size resolution ranges from a fine ~ 0.07 dB up to a coarse
6 dB per bit, depending on the gain code. As shown in Figure 39, of
the 42 dB total range, 28 dB has resolution of better than 2 dB,
and 22 dB has resolution of better than 1 dB.
BLOCK ARCHITECTURE
VCCI
The input impedance is approximately 200 Ω differential,
regardless of which preamplifier is selected. Note that the input
impedance is formed by using active circuit elements and is not
set by passive components. See Figure 38 for a simplified
schematic of the input interface.
03692-036
The AD8370 is a low cost, digitally controlled, fine adjustment
variable gain amplifier (VGA) that provides both high IP3 and
low noise figure. The AD8370 is fabricated on an ADI
proprietary high performance 25 GHz silicon bipolar process.
The –3 dB bandwidth is approximately 750 MHz throughout
the variable gain range. The typical quiescent current of the
AD8370 is 78 mA. A power-down feature reduces the current to
less than 4 mA. The input impedance is approximately 200 Ω
differential, and the output impedance is approximately 100 Ω
differential to be compatible with saw filters and matching
networks used in intermediate frequency (IF) radio
applications. Because there is no feedback between the input
and output and stages within the amplifier, the input amplifier
is isolated from variations in output loading and from
subsequent impedance changes, and excellent input to output
isolation is realized. Excellent distortion performance and wide
bandwidth make the AD8370 a suitable gain control device for
modern differential receiver designs. The AD8370 differential
input and output configuration is ideally suited to fully
differential signal chain circuit designs, although it can be
adapted to single-ended system applications, if required.
–11dB GAIN
–25dB GAIN
0
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
VSOURCE [V peak] (V)
Figure 39. Gain Resolution and Nominal Input and
Output Range over the Gain Range
1.8
AD8370
OUTPUT AMPLIFIER
Table 4. Serial Programming Timing Parameters
The output impedance is approximately 100 Ω differential and,
like the input preamplifier, this impedance is formed using
active circuit elements. See Figure 40 for a simplified schematic
of the output interface.
Parameter
Clock Pulse Width (TPW)
Clock Period (TCK)
Setup Time Data vs. Clock (TDS)
Setup Time Latch vs. Clock (TES)
Hold Time Latch vs. Clock (TEH)
Min
25
50
10
20
10
10μA
OPHI/OPLO
740Ω
VCC/2
03692-038
03692-040
CLCK/DATA/LTCH/PWUP
Figure 42. Simplified Circuit for Digital Inputs
Figure 40. OPHI/OPLO Simplified Circuit
The gain of the output amplifier, and thus the AD8370 as a
whole, is load dependent. The following equation can be used to
predict the gain deviation of the AD8370 from that at 100 Ω as
the load is varied.
VOCM
1.98
98
1+
RLOAD
75Ω
VCC/2
For example, if RLOAD is 1 kΩ, the gain is a factor of 1.80 (5.12 dB)
above that at 100 Ω, all other things being equal. If RLOAD is 50
Ω, the gain is a factor of 0.669 (3.49 dB) below that at 100 Ω.
DIGITAL INTERFACE AND TIMING
The digital control port uses a standard TTL interface. The 8-bit
control word is read in a serial fashion when the LTCH pin is
held low. The levels presented to the DATA pin are read on each
rising edge of the CLCK signal. Figure 41 illustrates the timing
diagram for the control interface. Minimum values for timing
parameters are presented in Table 4. Figure 42 is a simplified
schematic of the digital input pins.
TDS
DATA
(PIN 14)
MSB
MSB-1 MSB-2 MSB-3 LSB+3 LSB+2 LSB+1
TCK
LSB
TPW
CLCK
(PIN 13)
TEH
TES
03692-039
LTCH
(PIN 12)
Figure 41. Digital Timing Diagram
Rev. A | Page 14 of 28
03692-041
GainDeviation =
Figure 43. Simplified Circuit for VOCM Output
Unit
ns
ns
ns
ns
ns
AD8370
APPLICATIONS
BASIC CONNECTIONS
GAIN CODES
Figure 44 shows the minimum connections required for basic
operation of the AD8370. Supply voltages between 3.0 V and
5.5 V are allowed. The supply to the VCCO and VCCI pins
should be decoupled with at least one low inductance, surfacemount ceramic capacitor of 0.1 μF placed as close as possible to
the device.
The AD8370’s two gain ranges are referred to as high gain (HG)
and low gain (LG). Within each range, there are 128 possible
gain codes. Therefore, the minimum gain in the low gain range
is given by the nomenclature LG0 whereas the maximum gain
in that range is given by LG127. The same is true for the high
gain range. Both LG0 and HG0 essentially turn off the variable
transconductance stage, and thus no output is available with
these codes (see Figure 26).
SERIAL CONTROL
INTERFACE
1nF
1nF
The theoretical linear voltage gain can be expressed with respect
to the gain code as
11
10
9
OPLO
CLCK
12
OCOM
13
LTCH
14
VCCO
15
DATA
BALANCED
SOURCE
16
INLO
2
ICOM
RS
AV = GainCode Vernier (1 + (PreGain − 1) MSB)
RL
ICOM
VCCI
PWUP
VOCM
VCCO
OCOM
OPHI
RS
INHI
AD8370
1
2
3
4
5
6
7
8
BALANCED
LOAD
AV is the linear voltage gain.
GainCode is the digital gain control word minus the MSB
(the final 7 bits).
2
1nF
100pF
0.1μF
FERRITE
BEAD
0.1μF
1nF
Vernier = 0.055744 V/V
100pF
PreGain = 7.079458 V/V
FERRITE
BEAD
+VS (3.0V TO 5.0V)
03692-042
1nF
where:
Figure 44. Basic Connections
The AD8370 is designed to be used in differential signal chains.
Differential signaling allows improved even-order harmonic
cancellation and better common-mode immunity than can be
achieved using a single-ended design. To fully exploit these
benefits, it is necessary to drive and load the device in a
balanced manner. This requires some care to ensure that the
common-mode impedance values presented to each set of
inputs and outputs are balanced. Driving the device with an
unbalanced source can degrade the common-mode rejection
ratio. Loading the device with an unbalanced load can cause
degradation to even-order harmonic distortion and premature
output compression. In general, optimum designs are fully
balanced, although the AD8370 still provides impressive
performance when used in an unbalanced environment.
The AD8370 is a fine adjustment, VGA. The gain control
transfer function is linear in voltage gain. On a decibel scale,
this results in the logarithmic transfer functions shown in
Figure 4. At the low end of the gain transfer function, the slope
is steep, providing a rather coarse control function. At the high
end of the gain control range, the decibel step size decreases,
allowing precise gain adjustment.
MSB is the most significant bit of the 8-bit gain control word.
The MSB sets the device in either high gain mode (MSB = 1)
or low gain mode (MSB = 0).
For example, a gain control word of HG45 (or 10101101 binary)
results in a theoretical linear voltage gain of 17.76 V/V,
calculated as
45 × 0.055744 × (1 + (7.079458 − 1) × 1)
Increments or decrements in gain within either gain range are
simply a matter of operating on the GainCode. Six –dB gain
steps, which are equivalent to doubling or halving the linear
voltage gain, are accomplished by doubling or halving the
GainCode.
When power is first applied to the AD8370, the device is
programmed to code LG0 to avoid overdriving the circuitry
following it.
POWER-UP FEATURE
The power-up feature does not affect the GainCode, and the
gain setting is preserved when in power-down mode. Powering
down the AD8370 (bringing PWUP low while power is still
applied to the device) does not erase or change the GainCode
from the AD8370, and the same gain code is in place when the
device is powered up, that is, when PWUP is brought high
again. Removing power from the device all together and
reapplying, however, reprograms to LG0.
Rev. A | Page 15 of 28
AD8370
gain is increased beyond this point, which explains the knee in
the OIP3 curve. The IIP3 curve has a knee for the same reason;
however, as the gain is increased beyond the knee, the IIP3
starts to decrease rather than increase. This is because in this
region OIP3 is constant, therefore the higher the gain, the lower
the IIP3. The two gain ranges have equal SFDR at
approximately 13 dB power gain.
CHOOSING BETWEEN GAIN RANGES
There is some overlap between the two gain ranges; users can
choose which one is most appropriate for their needs. When
deciding which preamp to use, consider resolution, noise,
linearity, and spurious-free dynamic range (SFDR). The most
important points to keep in mind are
•
The low gain range has better gain resolution.
•
The high gain range has a better noise figure.
•
The high gain range has better linearity and SFDR at
higher gains.
•
Conversely, the low gain range has higher SFDR at lower
gains.
LAYOUT AND OPERATING CONSIDERATIONS
Each input and output pin of the AD8370 presents either a
100 Ω or 50 Ω impedance relative to their respective ac grounds.
To ensure that signal integrity is not seriously impaired by the
printed circuit board, the relevant connection traces should
provide an appropriate characteristic impedance to the ground
plane. This can be achieved through proper layout.
Figure 45 provides a summary of noise, OIP3, IIP3, and SFDR
as a function of device power gain. SFDR is defined as
SFDR =
2
(IIP3 − NF − N S )
3
When laying out an RF trace with a controlled impedance,
consider the following:
•
Space the ground plane to either side of the signal trace at
least three line-widths away to ensure that a microstrip
(vertical dielectric) line is formed, rather than a coplanar
(lateral dielectric) waveguide.
•
Ensure that the width of the microstrip line is constant and
that there are as few discontinuities as possible, such as
component pads, along the length of the line. Width
variations cause impedance discontinuities in the line and
may result in unwanted reflections.
•
Do not use silkscreen over the signal line because it alters
the line impedance.
where:
IIP3 is the input third-order intercept point, the output
intercept point in dBm minus the gain in dB.
NF is the noise figure in dB.
NS is source resistor noise, –174 dBm for a 1 Hz bandwidth at
300°K (27°C).
In general, NS = 10 log10(kTB), where k = 1.374 ×10−23 , T is the
temperature in degrees Kelvin, and B is the noise bandwidth in
Hertz.
180
40
OIP3 LOW GAIN
OIP3 HIGH GAIN
30
20
IIP3 LOW GAIN
IIP3 HIGH GAIN
10
160
Figure 46 shows the cross section of a PC board, and Table 5
show the dimensions that provide a 100 Ω line impedance for
FR-4 board material with εr = 4.6.
150
Table 5.
170
140
NF HIGH GAIN
0
130
–10
SFDR (dB)
NF LOW GAIN
100 Ω
22 mils
53 mils
2 mils
W
H
T
50 Ω
13 mils
8 mils
2 mils
120
SFDR LOW GAIN
–20
110
SFDR HIGH GAIN
–30
–30
100
–20
–10
0
10
20
30
03692-043
NOISE FIGURE (dB); OIP3 AND IIP3 (dBm)
50
Keep the length of the input and output connection lines as
short as possible.
3W
W
3W
T
40
POWER GAIN (dB)
ER
03692-044
H
Figure 45. OIP3, IIP3, NF, and SFDR Variation with Gain
As the gain increases, the input amplitude required to deliver
the same output amplitude is reduced. This results in less
distortion at the input stage, and therefore the OIP3 increases.
At some point, the distortion of the input stage becomes small
enough such that the nonlinearity of the output stage becomes
dominant. The OIP3 does not improve significantly because the
Figure 46. Cross-Sectional View of a PC Board
It possible to approximate a 100 Ω trace on a board designed
with the 50 Ω dimensions above by removing the ground plane
within 3 line-widths of the area directly below the trace.
Rev. A | Page 16 of 28
AD8370
High transient and noise levels on the power supply, ground,
and digital inputs can, under some circumstances, reprogram the
AD8370 to an unintended gain code. This further reinforces the
need for proper supply bypassing and decoupling. The user
should also be aware that probing the AD8370 and associated
circuitry during circuit debug may also induce the same effect.
PACKAGE CONSIDERATIONS
The package of the AD8370 is a compact, thermally enhanced
TSSOP 16-lead design. A large exposed paddle on the bottom of
the device provides both a thermal benefit and a low inductance
path to ground for the circuit. To make proper use of this packaging feature, the PCB needs to make contact directly under the
device, connected to an ac/dc common ground reference with
as many vias as possible to lower the inductance and thermal
impedance.
0.5
0
9
RL
INHI
ICOM
VCCI
PWUP
VOCM
VCCO
OCOM
OPHI
AD8370
1
2
3
4
5
6
7
8
CAC
400
500
Figure 48 illustrates the differential balance at the output for a
single-ended input drive for multiple gain codes. The differential
balance is better than 0.5 dB for signal frequencies less than
250 MHz. Figure 49 depicts the differential balance over the
entire gain range at 10 MHz. The balance is degraded for lower
gain settings because the finite common gain allows some of the
input signal applied to INHI to pass directly through to the
OPLO pin. At higher gain settings, the differential gain dominates
and balance is restored.
0.6
LOW GAIN MODE
HIGH GAIN MODE
0.4
0.3
0.2
0.1
0
0.1μF
+VS
03692-045
0.1μF
300
Figure 48. Differential Output Balance for a Single-Ended Input Drive at
Maximum Gain (RL = 1 kΩ, CAC = 10 nF)
CAC
1nF
200
Figure 47. Single-Ended-to-Differential Conversion
The AD8370 is primarily designed for differential signal interfacing. The device can be used for single-ended-to-differential
conversion simply by terminating the unused input to ground
using a capacitor as depicted in Figure 47. The ac coupling
capacitors should be selected such that their reactance is
negligible at the frequency of operation. For example, using
1 nF capacitors for CAC presents a capacitive reactance of
03692-047
10
100
FREQUENCY (MHz)
DIFFERENTIAL BALANCE (dB)
CLCK
11
OPLO
DATA
12
OCOM
13
LTCH
14
VCCO
15
INLO
SINGLEENDED
SOURCE
16
ICOM
RS
LOW GAIN MODE
(GAIN CODE LG127)
0.5
CAC
AC
–0.5
–1.0
SINGLE-ENDED-TO-DIFFERENTIAL CONVERSION
SERIAL CONTROL
INTERFACE
C
HIGH GAIN MODE
(GAIN CODE HG255)
0
03692-046
Due to the nature of the AD8370’s circuit design, care must be
taken to minimize parasitic capacitance on the input and output.
The AD8370 could become unstable with more than a few pF of
shunt capacitance on each input. Using resistors in series with
input pins is recommended under conditions of high source
capacitance.
−j1.6 Ω on each input node at 100 MHz. This attenuates the
applied input voltage by 0.003 dB. If 10 pF capacitors had been
selected, the voltage delivered to the input would be reduced by
2.1 dB when operating with a 200 Ω source impedance.
DIFFERENTIAL BALANCE (dB)
The AD8370 contains both digital and analog sections. Care
should be taken to ensure that the digital and analog sections
are adequately isolated on the PC board. The use of separate
ground planes for each section connected at only one point via
a ferrite bead inductor ensures that the digital pulses do not
adversely affect the analog section of the AD8370.
0
32
64
96
0
32
64
96
128
GAIN CODE
Figure 49. Differential Output Balance at 10 MHz for a Single-Ended Drive vs.
Gain Code (RL = 1 kΩ, CAC = 10 nF)
Even though the amplifier is no longer being driven in a balanced
manner, the distortion performance remains adequate for most
applications. Figure 50 illustrates the harmonic distortion
performance of the circuit in Figure 47 over the entire gain range.
Rev. A | Page 17 of 28
AD8370
SERIAL CONTROL
INTERFACE
–50
RT
2
499Ω
10
9
LTCH
OCOM
OPLO
CLCK
11
RL
VCCO
OCOM
OPHI
AD8370
1
2
3
4
5
6
7
8
100Ω
VOCM
1nF
–60
1nF
–70
0.1μF
03692-050
+5V
SINGLE-ENDED GROUND
REFERENCED SOURCE
HD2
HD2
Figure 52. DC Coupling the AD8370. The AD8138 is used as a unity-gain level
shifting amplifier to lift the common-mode level of the source to midsupply.
–80
HD3
HD3
The AD8370 is also a dc accurate VGA. The common-mode dc
voltage present at the output pins is internally set to midsupply
using what is essentially a buffered resistive divider network
connected between the positive supply rail and the common
(ground) pins. The input pins are at a slightly higher dc
potential, typically 250 mV to 550 mV above the output pins,
depending on gain setting. In a typical single-supply
application, it is necessary to raise the common-mode reference
level of the source and load to roughly midsupply to maintain
symmetric swing and to avoid sinking or sourcing strong bias
currents from the input and output pins. It is possible to use
balanced dual supplies to allow ground referenced source and
load, as shown in Figure 51. By connecting the VOCM pin and
unused input to ground, the input and output common-mode
potentials are forced to virtual ground. This allows direct
coupling of ground referenced source and loads. The initial
differential input offset is typically only a few 100 μV. Over
temperature, the input offset could be as high as a few tens of
mVs. If precise dc accuracy is needed over temperature and time,
it may be necessary to periodically measure the input offset and
to apply the necessary opposing offset to the unused differential
input, canceling the resulting output offset.
LOW GAIN MODE
0
32
64
03692-048
–90
–100
12
VCCO
13
VOCM
RS
14
PWUP
499Ω
–40
HARMONIC DISTORTION (dBc)
AD8138
15
VCCI
RT
VOCM
where RTERM is the termination resistor connected to the other
input.
16
DATA
499Ω
INLO
+5V
RinSE = RinDIFF + RTERM
ICOM
100Ω
ICOM
499Ω
VOCM
INHI
If the amplifier is driven in single-ended mode, the input
impedance varies depending on the value of the resistor used to
terminate the other input as
HIGH GAIN MODE
96
0
32
64
96
128
GAIN CODE
Figure 50. Harmonic Distortion of the Circuit in Figure 47
DC-COUPLED OPERATION
–2.5V
SERIAL CONTROL
INTERFACE
0V
1nF
DATA
CLCK
12
11
10
9
OPLO
13
OCOM
14
LTCH
15
VCCO
16
INLO
RS
SINGLEENDED
GROUND
REFERENCED
SOURCE
ICOM
RT
RL
INHI
ICOM
VCCI
PWUP
VOCM
VCCO
OCOM
OPHI
AD8370
1
2
3
4
5
6
7
8
0V
–2.5V
1nF
0.1μF
03692-049
+2.5V
0.1μF
Figure 51. DC Coupling the AD8370. Dual supplies are used to set the input
and output common-mode levels to 0 V.
To address situations where dual supplies are not convenient, a
second option is presented in Figure 52. The AD8138 differential
amplifier is used to translate the common-mode level of the
driving source to midsupply, which allows dc accurate
performance with a ground-referenced source without the need
for dual supplies. The bandwidth of the solution in Figure 52 is
limited by the gain-bandwidth product of the AD8138. The
normalized frequency response of both implementations is shown
in Figure 53.
Rev. A | Page 18 of 28
AD8370
10
Often it is wise to include input and output parasitic suppression
resistors, RIP and ROP. Parasitic suppressing resistors help to
prevent resonant effects that occur as a result of internal bondwire inductance, pad to substrate capacitance, and stray
capacitance of the printed circuit board trace artwork. If
omitted, undesirable settling characteristics may be observed.
Typically, only 10 Ω to 25 Ω of series resistance is all that is
needed to help dampen resonant effects. Considering that most
ADCs present a relatively high input impedance, very little
signal is lost across the RIP and ROP series resistors.
6
AD8370 WITH
AD8138 SINGLE
+5V SUPPLY
4
2
0
–2
AD8370
USING DUAL
±2.5V SUPPLY
–4
–6
03692-051
NORMALIZED RESPONSE (dB)
8
–8
–10
1
10
100
1k
10k
100k
1M
10M
100M
1G
FREQUENCY (Hz)
Figure 53. Normalized Frequency Response of the Two Solutions in
Figure 51 and Figure 52
ADC INTERFACING
Although the AD8370 is designed to provide a 100 Ω output
source impedance, the device is capable of driving a variety of
loads while maintaining reasonable gain and distortion
performance. A common application for the AD8370 is ADC
driving in IF sampling receivers and broadband wide dynamic
range digitizers. The wide gain adjustment range allows the use
of lower resolution ADCs. Figure 54 illustrates a typical ADC
interface network.
ROP
CAC
ZS
RIP
AD8370
VIN
100Ω
ZP
RT
ZIN
ADC
VOCM
ROP
CAC
ZS
RIP
03692-052
VIN
Figure 54. Generic ADC Interface
Many factors need to be considered before defining component
values used in the interface network, such as the desired
frequency range of operation, the input swing, and input
impedance of the ADC. AC coupling capacitors, CAC, should be
used to block any potential dc offsets present at the AD8370
outputs, which would otherwise consume the available low-end
range of the ADC. The CAC capacitors should be large enough
so that they present negligible reactance over the intended
frequency range of operation. The VOCM pin may serve as an
external reference for ADCs that do not include an on-board
reference. In either case, it is suggested that the VOCM pin be
decoupled to ground through a moderately large bypassing
capacitor (1 nF to 10 nF) to help minimize wideband noise
pick-up.
Depending on the input impedance presented by the input
system of the ADC, it may be desirable to terminate the ADC
input down to a lower impedance by using a terminating
resistor, RT. The high frequency response of the AD8370
exhibits greater peaking when driving very light loads. In
addition, the terminating resistor helps to better define the
input impedance at the ADC input. Any part-to-part variability
of ADC input impedance is reduced when shunting down the
ADC inputs by using a moderate tolerance terminating resistor
(typically a 1% value is acceptable).
After defining reasonable values for coupling capacitors,
suppressing resistors, and the terminating resistor, it is time to
design the intermediate filter network. The example in
Figure 54 suggests a second-order, low-pass filter network
comprised of series inductors and a shunt capacitor. The order
and type of filter network used depends on the desired high
frequency rejection required for the ADC interface, as well as
on pass-band ripple and group delay. In some situations, the
signal spectra may already be sufficiently band-limited such
that no additional filter network is necessary, in which case ZS
would simply be a short and ZP would be an open. In other
situations, it may be necessary to have a rather high-order
antialiasing filter to help minimize unwanted high frequency
spectra from being aliased down into the first Nyquist zone of
the ADC.
To properly design the filter network, it is necessary to consider
the overall source and load impedance presented by the AD8370
and ADC input, including the additional resistive contribution
of suppression and terminating resistors. The filter design can
then be handled by using a single-ended equivalent circuit, as
shown in Figure 55. A variety of references that address filter
synthesis are available. Most provide tables for various filter
types and orders, indicating the normalized inductor and
capacitor values for a 1 Hz cutoff frequency and 1 Ω load. After
scaling the normalized prototype element values by the actual
desired cut-off frequency and load impedance, it is simply a
matter of splitting series element reactances in half to realize the
final balanced filter network component values.
Rev. A | Page 19 of 28
AD8370
LOAD
SINGLE-ENDED
EQUIVALENT
VS
RS
2
ZS
2
RL
2
ZP
RL
2
The input of the AD9430 is terminated with a 1.5 kΩ resistor so
that the overall load presented to the filter network is ~1 kΩ.
The variable gain of the AD8370 extends the useable dynamic
range of the ADC. The measured intermodulation distortion of
the combination is presented in Figure 57 at 42 MHz.
03692-053
RS
2
RL
ZP
BALANCED
CONFIGURATION
VS
A complete design example is shown in Figure 58. The AD8370
is configured for single-ended-to-differential conversion with
the input terminated down to present a single-ended 75 Ω input.
A sixth-order Chebyshev differential filter is used to interface
the output of the AD8370 to the input of the AD9430
170 MSPS, 12-bit ADC. The filter minimizes aliasing effects
and improves harmonic distortion performance.
ZS
ZS
2
Figure 55. Single-Ended-to-Differential Network Conversion
As an example, a second-order, Butterworth, low-pass filter
design is presented where the differential load impedance is
1200 Ω, and the padded source impedance of the AD8370 is
assumed to be 120 Ω. The normalized series inductor value for
the 10-to-1, load-to-source impedance ratio is 0.074 H, and the
normalized shunt capacitor is 14.814 F. For a 70 MHz cutoff
frequency, the single-ended equivalent circuit consists of a
200 nH series inductor followed by a 27 pF capacitor. To realize
the balanced equivalent, simply split the 200 nH inductor in
half to realize the network shown in Figure 56.
0
–10
–20
–30
–40
–50
dBFS
SOURCE
RS
–60
–70
–80
–90
–100
RS
= 0.1
RL
–120
NORMALIZED
SINGLE-ENDED
EQUIVALENT
VS
–110
LN = 0.074H
03692-055
RS =
CN
14.814F
–130
RL= 1Ω
0
10
20
30
40
50
60
70
FREQUENCY (MHz)
fC = 1Hz
RS = 120Ω
200nH
DE-NORMALIZED
SINGLE-ENDED
EQUIVALENT
VS
Figure 57. FFT Plot of Two-Tone Intermodulation Distortion at
42 MHz for the Circuit in Figure 58
27pF
RL= 1200Ω
fC = 70MHz
3 V OPERATION
100nH
BALANCED
CONFIGURATION
VS
RS
= 60Ω
2
100nH
27pF
RL
2 = 600Ω
RL
2 = 600Ω
03692-054
RS
= 60Ω
2
In Figure 57, the intermodulation products are comparable to
the noise floor of the ADC. The spurious-free dynamic range of
the combination is better than 66 dB for a 70 MHz measurement
bandwidth.
Figure 56. Second-Order, Butterworth, Low-Pass Filter Design Example
It is possible to operate the AD8370 at voltages as low as 3 V
with only minor performance degradation. Table 6 gives typical
specifications for operation at 3 V.
Table 6.
Parameter
Output IP3
P1dB
−3 dB Bandwidth
IMD3
Rev. A | Page 20 of 28
Typical (70 MHz, RL = 100 Ω)
+23.5 dBm
+12.7 dBm
650 MHz (HG 127)
−82 dBc (RL = 1 kΩ)
AD8370
SERIAL CONTROL INTERFACE
FROM 75Ω
Tx-LINE
CAC
CAC
100nF
68nH
180nH
220nH
25Ω
VINA
INLO
DATA
CLCK
12
11
10
9
OPLO
13
OCOM
14
LTCH
15
VCCO
16
ICOM
120Ω
100nF
AD8370
ICOM
VCCI
PWUP
VOCM
VCCO
OCOM
OPHI
27pF
INHI
1
2
3
4
5
6
7
8
CAC
CAC
68nH
39pF
180nH
27pF
220nH
1.5kΩ
AD9430
25Ω
VINB
100nF
1nF
100nF
0.1μF
0.1μF
03692-056
RS
+VS
Figure 58. ADC Interface Example
Rev. A | Page 21 of 28
AD8370
EVALUATION BOARD AND SOFTWARE
The evaluation board allows quick testing of the AD8370 by
using standard 50 Ω test equipment. The schematic is shown in
Figure 59. Transformers T1 and T2 are used to transform 50 Ω
source and load impedances to the desired input and output
reference levels. The top and bottom layers are shown in
Figure 63 and Figure 64. The ground plane was removed under
the traces between T1 and Pins INHI and INLO to approximate
a 100 Ω characteristic impedance.
1
2
3
14
15
4
16
5
The evaluation board comes with the AD8370 control software
that allows serial gain control from most computers. The
evaluation board is connected via a cable to the parallel port of
the computer. Adjusting the appropriate slider bar in the control
software automatically updates the gain code of the AD8370 in
either a linear or linear-in-dB fashion.
6
17
18
7
19
8
20
9
21
10
22
11
23
12
24
13
25
D-SUB 25 PIN MALE
L2*
C9
R7 R6 R5
1kΩ 1kΩ 1kΩ
C1
13
12
11
10
9
LTCH
VCCO
OCOM
OPLO
R2
0Ω
14
CLCK
1:4
15
DATA
T2
16
INLO
T1
ICOM
R4
0Ω
PWUP
VOCM
VCCO
OCOM
OPHI
1
2
3
4
5
6
7
8
JTX-2-10T
2:1
AD8370
R1
0Ω
VCCI
50Ω Tx LINE
1nF
ICOM
50Ω Tx LINE
OUT+
50Ω Tx LINE
OUT–
R3
0Ω
C2
C4
1nF
1nF
C8
0.1μF
C5
0.1μF
C6
1μF
SW1
PWUP
L1*
VOCM
+VS
R9
OPEN
R8 49.9Ω
C7
0.1μF
GND
C10 OPEN
P2
1
2
3
4
5
VS
GND
*EMI SUPPRESSION FERRITE
HZ1206E601R-00
Figure 59. AD8370 Evaluation Board Schematic
Rev. A | Page 22 of 28
03692-057
IN–
TC4-1W
50Ω Tx LINE
C3
1nF
INHI
IN+
OPEN
03692-058
AD8370
Figure 60. Evaluation Software
Table 7. AD8370 Evaluation Board Configuration Options
Component
VS, GND, VOCM
SW1, R8,
C10, PWUP
P1, R5, R6,
R7, C9
J1, J2, J6, J7
C1, C2, C3, C4
T1, T2
R1, R2, R3, R4
C5, C6, C7,
C8 L1, L2
Function
Power Interface Vector Pins. Apply supply voltage between VS and GND. The VOCM
pin allows external monitoring of the common-mode input and output bias levels.
Device Enable. Set to Position B to power up the device. When in Position A, the PWUP
pin is connected to the PWUP vector pin. The PWUP pin allows external power cycling
of the device. R8 and C10 are provided to allow for proper cable termination.
Serial Control Interfaces. The evaluation board can be controlled using most PCs.
Windows®-based control software is shipped with the evaluation kit. A 25-pin, D-sub
connector cable is required to connect the PC to the evaluation board. It may be
necessary to use a capacitor on the clock line, depending on the quality of the PC port
signals. A 1 nF capacitor for C9 is usually sufficient for reducing clock overshoot.
Input and Output Signal Connectors. These SMA connectors provide a convenient way
to interface the evaluation board with 50 Ω test equipment. Typically, the device is
evaluated using a single-ended source and load. The source should connect to
J1 (IN+), and the load should connect to J6 (OUT+).
AC Coupling Capacitors. Provide ac coupling of the input and output signals.
Impedance Transformers. T1 provides a 50 Ω to 200 Ω impedance transformation.
T2 provides a 100 Ω to 50 Ω impedance transformation.
Single-Ended or Differential. R2 and R4 are used to ground the center tap of the
secondary windings on transformers T1 and T2. R1 and R3 should be used to ground
J2 and J7 when used in single-ended applications.
Power Supply Decoupling. Nominal supply decoupling consists of a ferrite bead
series inductor followed by a 1 μF capacitor to ground followed by a 0.1 μF capacitor
to ground positioned as close to the device as possible. C7 provides additional
decoupling of the input common-mode voltage. L1 provides high frequency
isolation between the input and output power supply. L2 provides high
frequency isolation between the analog and digital ground.
Rev. A | Page 23 of 28
Default Condition
Not applicable
SW1 = installed
R8 = 49.9 Ω (Size 0805)
C10 = open (Size 0805)
P1 = installed
R5, R6, R7 = 1 kΩ (Size 0603)
C9 = open (Size 0603)
Not applicable
C1, C2, C3, C4 = 1 nF (Size 0603)
T1 = TC4 −1W (Mini-Circuits)
T2 = JTX−2−10T (Mini-Circuits)
R1, R2, R3, R4 = 0 Ω (Size 0603)
C6 = 1 μF (Size 0805)
C5, C7, C8 = 0.1 μF (Size 0603)
L1, L2 = HZ1206E601R-00
(Steward, Size 1206)
Figure 61. Evaluation Board Top Silkscreen
03692-061
03692-059
AD8370
Figure 62. Evaluation Board Bottom Silkscreen
03692-062
03692-060
Figure 63. Evaluation Board Top
Figure 64. Evaluation Board Bottom
Rev. A | Page 24 of 28
AD8370
APPENDIX
CHARACTERIZATION EQUIPMENT
DEFINITIONS OF SELECTED PARAMETERS
An Agilent N4441A Balanced-Measurement System was used to
obtain the gain, phase, group delay, reverse isolation, CMRR,
and s-parameter information contained in this data sheet. With
the exception of the s-parameter information, T-attenuator pads
were used to match the 50 Ω impedance of this instrument’s ports
to the AD8370. An Agilent 4795A Spectrum Analyzer was used
to obtain nonlinear measurements IMD, IP3, and P1dB through
matching baluns and/or attenuator networks. Various other
measurements were taken with setups shown in this section.
Common-mode rejection ratio (Figure 28) has been defined for
this characterization effort as
COMPOSITE WAVEFORM ASSUMPTION
The nonlinear two-tone measurements made for this data sheet,
that is, IMD and IP3, are based on the assumption of a fixed
value composite waveform at the output, generally 1 V p-p. The
frequencies of interest dictate the use of RF test equipment, and
because this equipment is generally not designed to work in
units of volts, but rather watts and dBm, an assumption was
made to facilitate equipment setup and operation. Two sinusoidal
tones can be represented as
Differential Mode Gain
Common Mode Gain
where the numerator is the gain into a differential load at the
output due to a differential source at the input, and the
denominator is the gain into a differential-mode load at the
output due to a common-mode source at the input. In terms of
mixed-mode s-parameters, this equates to
SDD21
SDC 21
More information on mixed-mode s-parameters can be
obtained in a reference by Bockelman, D.E. and Eisenstadt,
W.R., Combined Differential and Common-Mode Scattering
Parameters: Theory and Simulation. IEEE Transactions on
Microwave Theory and Techniques, v 43, n 7, 1530 (July 1995).
V1 = V sin (2∏f1t)
Reverse isolation (Figure 26) is defined as SDD12.
V2 = V sin (2∏f2t)
Power supply rejection ratio (PSRR) is defined as
Adm
As
The RMS average voltage of one tone is
2
1T
(V1 ) dt = 1
∫
T0
2
where Adm is the differential mode forward gain (SDD21), and
As is the gain from the power supply pins (VCCI and VCCO,
taken together) to the output (OPLO and OPHI, taken
differentially), corrected for impedance mismatch. The
following reference provides more information: Gray, P.R.,
Hurst, P.J., Lewis, S.H. and Meyer, R.G., Analysis and Design of
Analog Integrated Circuits, 4th Edition, John Wiley & Sons, Inc.,
page 422.
where T is the period of the waveform. The RMS average
voltage of the two-tone composite signal is
2
1T
(V1 + V2 ) dt = 1
T ∫0
It can be shown that the average power of this composite
waveform is twice (3 dB) that of the single tone. This also
means that the composite peak-to-peak voltage is twice (6 dB)
that of a single tone. This principle can be used to set correct
input amplitudes from generators scaled in dBm and is correct
if the two tones are of equal amplitude and are reasonably close
in frequency.
Rev. A | Page 25 of 28
AD8370
–22.5dB
PORT 1
PORT 1
SERIAL DATA
SOURCE
1nF
1nF
1nF
LTCH
VCCO
OCOM
OPLO
VCCO
OCOM
OPHI
2
3
4
5
6
7
8
1nF
1nF
1μF
Figure 65. PSRR Adm Test Setup
HP8133A
3GHz PULSE
GENERATOR
AUX IN
1nF
50Ω
INPUT
Figure 66. PSRR As Test Setup
TEKTRONIX TDS5104
DPO OSCILLOSCOPE
50Ω
INPUT
TRIG
3dB
ATTEN
OUT
6dB
SPLITTER
50Ω
50Ω
INPUT INPUT
SERIAL DATA
SOURCE
VS 5.0V
3dB
ATTEN
475Ω
2dB
ATTEN
52.3Ω
LTCH
VCCO
OCOM
OPLO
OCOM
OPHI
3
4
5
6
7
8
ICOM
2
DATA
VCCO
9
CLCK
10
VOCM
11
PWUP
12
VCCI
6dB
SPLITTER
13
ICOM
OUT
200Ω
14
1
INLO
3dB
ATTEN
15
INHI
16
AD8370
475Ω
3dB
ATTEN
2dB
ATTEN
52.3Ω
VS 5.0V
1μF
1nF
1nF
1nF
1μF
VS 5.0V
Figure 67. DC Pulse Response and Overdrive Recovery Test Setup
Rev. A | Page 26 of 28
03692-065
1nF
1
MINICIRCUITS
TC2-1T
03692-063
1μF
9
AD8370
200Ω
VS 5.0V
VS 5.0V
10
PORT 2
03692-064
1nF
11
AGILENT 8753D
NETWORK ANALYZER
1nF
12
CLCK
8
13
VOCM
7
DATA
6
14
PWUP
5
15
VCCI
4
AD8370
1nF
16
INLO
3
1nF
ICOM
2
PORT 2
ICOM
OPHI
1
MINICIRCUITS
TC2-1T
AGILENT 8753D
NETWORK ANALYZER
OCOM
OCOM
OPLO
LTCH
DATA
VCCO
INLO
VCCO
9
CLCK
10
VOCM
11
PWUP
12
VCCI
13
ICOM
14
INHI
0Ω
15
ICOM
T2
16
INHI
1nF
T1
MINICIRCUITS
TC4-1W
BIAS TEE
CONNECTION
TO PORT 1
SERIAL DATA
SOURCE
VS 5.0V
AD8370
AGILENT 8648D
SIGNAL
GENERATOR
TEKTRONIX
TDS5104 DPO
OSCILLOSCOPE
SERIAL DATA
SOURCE
TEKTRONIX
P6205 ACTIVE
FET PROBE
RF OUT
VS 5.0V
50Ω INPUT
1nF 475Ω
1nF
15
14
13
12
11
10
9
DATA
CLCK
LTCH
VCCO
OCOM
OPLO
0Ω
16
INLO
T2
MINICIRCUITS
TC4-1W
ICOM
T1
MINICIRCUITS
JTX-2-10T
50Ω INPUT
105Ω
INHI
ICOM
VCCI
PWUP
VOCM
VCCO
OCOM
OPHI
AD8370
1
2
3
4
5
6
7
8
1nF 475Ω
1nF
VS 5.0V
1μF
1nF
1nF
03692-066
VS 5.0V
1μF
1nF
Figure 68. Gain Step Time Domain Response Test Setup
AGILENT 8648D
SIGNAL
GENERATOR
10MHz REF OUT
TEKTRONIX
TDS5104 DPO
OSCILLOSCOPE
SERIAL DATA
SOURCE
RF OUT
VS 5.0V
1nF 475Ω
1nF
14
13
12
11
10
9
DATA
CLCK
LTCH
VCCO
OCOM
OPLO
MINICIRCUITS
JTX-2-10T
105Ω
INHI
ICOM
VCCI
PWUP
VOCM
VCCO
OCOM
OPHI
AD8370
1
2
3
4
5
6
7
8
1nF 475Ω
1nF
10MHz IN
OUTPUT
AGILENT 33250A
FUNCTION/ARBITRARY
WAVEFORM
GENERATOR
VS 5.0V
VS 5.0V
1μF
50Ω INPUT
1nF
1nF
52.3Ω
TEKTRONIX
P6205 ACTIVE
FET PROBE
50Ω INPUT
1μF
1nF
Figure 69. PWUP Response Time Domain Test Setup
Rev. A | Page 27 of 28
03692-067
0Ω
15
ICOM
T2
16
INLO
T1
MINICIRCUITS
TC4-1W
AD8370
OUTLINE DIMENSIONS
5.10
5.00
4.90
16
BOTTOM
VIEW
9
4.50
4.40
4.30
TOP
VIEW
1
EXPOSED
PAD
(Pins Up)
6.40
BSC
3.00
SQ
8
1.05
1.00
0.80
1.20 MAX
0.15
0.00 SEATING 0.65
BSC
PLANE
0.30
0.19
0.20
0.09
8°
0°
0.75
0.60
0.45
COMPLIANT TO JEDEC STANDARDS MO-153-ABT
Figure 70. 16-Lead Thin Shrink Small Outline Package with Exposed Pad [TSSOP_EP]
(RE-16-2)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD8370ARE
AD8370ARE-REEL7
AD8370AREZ 1
AD8370AREZ-RL71
AD8370-EVAL
1
Temperature Range
Package Description
16-lead TSSOP, Tube
16-lead TSSOP, 7” Reel
16-lead TSSOP, Tube
16-lead TSSOP, 7” Reel
Evaluation Board
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
Package Option
RE-16-2
RE-16-2
RE-16-2
RE-16-2
Z = Pb-free part.
© 2005 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D03692–0–7/05(A)
T
T
Rev. A | Page 28 of 28