IRF ECJ-1VC1H271J Highly efficient integrated 9a, synchronous buck regulator Datasheet

PD-97514
IR3859MPbF
SupIRBuck
HIGHLY EFFICIENT
TM
INTEGRATED 9A, SYNCHRONOUS BUCK REGULATOR
Features
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Applications
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Description
Greater than 95% Maximum Efficiency
Wide Input Voltage Range 1.5V to 21V
Wide Output Voltage Range 0.7V to 0.9*Vin
Continuous 9A Load Capability
Integrated Bootstrap-diode
High Bandwidth E/A for excellent transient
performance
Programmable Switching Frequency up to 1.5MHz
Programmable Over Current Protection
Over Voltage Protection
Dedicated input for output voltage monitoring
Programmable PGood output
Hiccup Current Limit
Precision Reference Voltage (0.7V, +/-1%)
Programmable Soft-Start
Enable Input with Voltage Monitoring Capability
Enhanced Pre-Bias Start-up
Seq input for Tracking applications
External Synchronization
-40oC to 125oC operating junction temperature
Thermal Protection
4mm x 5mm Power QFN Package
Halogen Free, Lead Free and RoHS compliant
Server Applications
Storage Applications
Embedded Telecom Systems
The IR3859 SupIRBuckTM is an easy-to-use, fully
integrated
and
highly
efficient
DC/DC
synchronous Buck regulator. The MOSFETs copackaged with the on-chip PWM controller make
IR3859 a space-efficient solution, providing
accurate power delivery for low output voltage
applications.
IR3859 is a versatile regulator which offers
programmability of start up time, switching
frequency and current limit while operating in
wide input and output voltage range.
The switching frequency is programmable from
250kHz to 1.5MHz for an optimum solution.
It also features important protection functions,
such as Pre-Bias startup, hiccup current limit and
thermal shutdown to give required system level
security in the event of fault conditions.
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Distributed Point of Load Power Architectures
Netcom Applications
Computing Peripheral Voltage Regulators
General DC-DC Converters
Fig. 1. Typical application diagram
Rev 5.0
1
PD-97514
IR3859MPbF
ABSOLUTE MAXIMUM RATINGS
(Voltages referenced to GND unless otherwise specified)
•
Vin ……………………………………………………. -0.3V to 25V
•
Vcc ……………….….…………….……..……….…… -0.3V to 8V (Note2)
•
Boot
……………………………………..……….…. -0.3V to 33V
•
SW
…………………………………………..……… -0.3V to 25V(DC), -4V to 25V(AC, 100ns)
•
Boot to SW
•
OCSet
•
Input / output Pins
•
PGND to GND ……………...………………………….. -0.3V to +0.3V
•
Storage Temperature Range ................................... -55°C To 150°C
•
Junction Temperature Range ................................... -40°C To 150°C (Note2)
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ESD Classification …………………………… ……… JEDEC Class 1C
•
Moisture sensitivity level………………...………………JEDEC Level [email protected] °C (Note5)
……..…………………………….…..….. -0.3V to Vcc+0.3V (Note1)
………………………………………….……. -0.3V to 30V (Max 30mA)
……………………………….. ... -0.3V to Vcc+0.3V (Note1)
Note1: Must not exceed 8V
Note2: Vcc must not exceed 7.5V for Junction Temperature between -10oC and -40oC
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the
device. These are stress ratings only and functional operation of the device at these or any other
conditions beyond those indicated in the operational sections of the specifications are not implied.
PACKAGE INFORMATION
4mm x 5mm POWER QFN
θ JA( Sync _ FET ) = 45 o C / W *
13
12
VIN
SW
11
θ JA( Ctrl _ FET ) = 45 o C / W *
PGnd
θ J -PCB = 2 o C / W
Exposed pads on underside
are connected to copper
pads of a 4-layer (2 oz.) PCB
*
Boot
14
Enable
15
Seq
16
1
ORDERING INFORMATION
Rev 5.0
10
17
Gnd
2
3
4
5
Fb Vsns COMP Gnd Rt
6
Vcc
9
Sync
8
PGood
7
SS OCSet
PACKAGE
DESIGNATOR
PACKAGE
DESCRIPTION
PIN COUNT
PARTS PER
REEL
M
IR3859MTRPbF
17
4000
M
IR3859MTR1PbF
17
750
2
PD-97514
IR3859MPbF
Block Diagram
Fig. 2. Simplified block diagram of the IR3859
Rev 5.0
3
PD-97514
IR3859MPbF
Pin Description
Pin
Name
Description
1
Fb
2
Vsns
3
Comp
4;17
Gnd
5
Rt
6
SS/SD
7
OCSet
8
PGood
Power Good status pin. Output is open drain. Connect a pull up resistor
from this pin to Vcc.
9
Sync
Sync pin, connect external system clock to synchronize multiple POLs
with the same frequency
10
V CC
11
PGnd
This pin powers the internal IC and the drivers. A minimum of 1uF high
frequency capacitor must be connected from this pin to the power ground
(PGnd).
Power Ground. This pin serves as a separated ground for the MOSFET
drivers and should be connected to the system’s power ground plane.
12
SW
Switch node. This pin is connected to the output inductor.
13
VIN
Input voltage connection pin.
14
Boot
15
Enable
16
Seq
Rev 5.0
Inverting input to the error amplifier. This pin is connected directly to the
output of the regulator via resistor divider to set the output voltage and
provide feedback to the error amplifier.
Sense pin for PGood
Output of error amplifier. An external resistor and capacitor network is
typically connected from this pin to Fb pin to provide loop compensation.
Signal ground for internal reference and control circuitry.
Set the switching frequency. Connect an external resistor from this pin to
Gnd to set the switching frequency. See Table 1 for Fs vs. Rt.
Soft start / shutdown. This pin provides user programmable soft-start
function. Connect an external capacitor from this pin to Gnd to set the
start up time of the output voltage. The converter can be shutdown by
pulling this pin below 0.3V.
Current limit set point. A resistor from this pin to SW pin will set the
current limit threshold.
Supply voltage for high side driver. A 0.1uF capacitor must be connected
from this pin to SW.
Enable pin to turn on and off the device. Use two external resistors to set
the turn on threshold (see Enable section). Connect this pin to Vcc if it is
not used.
Sequence pin. Use two external resistors to set Simultaneous Power up
sequencing. If this pin is not used connect to Vcc.
4
PD-97514
IR3859MPbF
Recommended Operating Conditions
Symbol
Vin
Vcc
Boot to SW
Vo
Io
Fs
Tj
Definition
Input Voltage
Supply Voltage
Supply Voltage
Output Voltage
Output Current
Switching Frequency
Junction Temperature
Min
Max
1.5
4.5
4.5
0.7
0
225
-40
21*
5.5
5.5
0.9*Vin
9
1650
125
Units
V
A
kHz
o
C
* Note: SW node should not exceed 25V
Electrical Specifications
Unless otherwise specified, these specification apply over 4.5V< Vcc<5.5V, Vin=12V, 0oC<Tj< 125oC.
Typical values are specified at Ta = 25oC.
PARAMETER
POWER STAGE
Power Losses
SYMBOL
TEST CONDITION
Ploss
Vcc=5V, Vin=12V, Vo=1.8V,
Io=9A, Fs=600kHz, L=0.68uH,
Note4
Top Switch
Rds(on)_Top
Bottom Switch
Deadband Time
Rds(on)_Bot
Tdb
Bootstrap Diode Forward
Voltage
SW leakage Current
o
VCC Supply Current (Dyn)
UNIT
W
29
11
16
5
10
30
ns
180
260
470
mV
Vcc=5V, ID=9A, Tj=25 C
mΩ
SW=0V, Enable=0V
6
SW=0V, Enable=high, SS=3V,
Vseq=0V, Note4
SUPPLY CURRENT
VCC Supply Current (Standby)
MAX
21
o
Note4
TYP
2.1
VBoot -Vsw =5V, ID=9A, Tj=25 C
I(Boot)= 30mA
Isw
MIN
ICC(Standby)
SS=0V, Vcc=5V, Enable low ,
No Switching
ICC(Dyn)
SS=3V, Vcc=5V, Enable high,
Fs=500kHz
500
5.5
9.97
14
uA
uA
mA
REFERENCE VOLTAGE
Feedback Voltage
VFB
0.7
o
Accuracy
o
0 C<Tj<125 C
o
o
-40 C<Tj<125 C, Note3
V
-1.0
+1.0
-2.0
-2.0
%
SOFT START / SD
Soft Start Current
Soft Start Clamp Voltage
Shutdown Output Threshold
Rev 5.0
ISS
Vss(clamp)
SD
Source
14
20
26
2.7
3.0
3.3
0.3
uA
V
5
PD-97514
IR3859MPbF
Electrical Specifications (continued)
Unless otherwise specified, these specifications apply over 4.5V< Vcc<5.5V, Vin=12V, 0oC<Tj< 125oC.
Typical values are specified at Ta = 25oC.
PARAMETER
ERROR AMPLIFIER
Input Offset Voltage
SYMBOL
MAX
UNIT
-10
+10
mV
Input Bias Current
IFb(E/A)
-1
+1
Input Bias Current
IVseq(E/A)
-1
+1
Sink Current
Isink(E/A)
0.40
0.85
1.2
Isource(E/A)
8
10
13
mA
Source Current
Slew Rate
Gain-Bandwidth Product
DC Gain
Vos
TYP
μA
Note4
7
12
20
V/μs
GBWP
Note4
20
30
40
MHz
Gain
Note4
100
110
120
dB
Vcc=4.5V
3.4
3.5
3.75
V
120
220
mV
1
V
V
Vmax(E/A)
Minimum Voltage
Vmin(E/A)
Note4
Seq Common Mode Voltage
Frequency Range
Vfb-Vseq, Vseq=0.8V
MIN
SR
Maximum Voltage
OSCILLATOR
Rt Voltage
TEST CONDITION
Vrt
FS
0
0.665
0.7
0.735
Rt=59K
225
250
275
Rt=28.7K
450
500
550
Rt=9.31K, Note4
1350
1500
1650
kHz
Vramp
Note4
1.8
Vp-p
Ramp Offset
Ramp(os)
Note4
0.6
V
Min Pulse Width
Dmin(ctrl)
Note4
50
ns
Ramp Amplitude
Max Duty Cycle
Dmax
Fixed Off Time
Toff
Fs=250kHz
92
Note4
130
Sync Frequency Range
Fsync
225
Sync Pulse Duration
Tsync
100
Sync Level Threshold
High
2
Low
Rev 5.0
%
200
ns
1650
kHz
200
ns
V
0.6
6
PD-97514
IR3859MPbF
Electrical Specifications (continued)
Unless otherwise specified, these specification apply over 4.5V< Vcc<5.5V, Vin=12V, 0oC<Tj< 125oC.
Typical values are specified at Ta = 25oC.
P ARAMETER
S YMBOL
TEST CONDITION
MIN
TYP
MAX
Fs =250kHz
20.8
23.6
26.4
Fs =500kHz
43
48.8
54.6
Fs =1500k Hz
136
154
172
Note4
-10
0
+10
UNIT
FAULT PROTECTION
OCSET Current
OC com p Offset Voltage
SS off tim e
OVP Trip Threshold
OVP Fault Prop. Delay
I OCSET
VOFFS ET
SS_Hicc up
OVP(trip)
OVP(delay)
4096
Vsns Rising
110
115
Note4
Therm al Shutdown
Note4
140
Therm al Hysteresis
Note4
20
120
%Vref
150
ns
°C
V CC _UVLO_S tart
Vcc Rising Trip Level
3.95
4.15
4.35
VCC-Stop-Threshold
V CC _UVLO_S top
Vcc Falling Trip Level
3.65
3.85
4.05
INPUT/OUTPUT SIGNAL
Enable-Start-Threshold
E nable_UVLO_Start
Supply ramping up
1.14
1.2
1.36
Enable-Stop-Threshold
E nable_UVLO_Stop
Supply ramping down
0.9
1.0
1.06
Enable leakage current
Ien
Enable= 3.3V
VPG
Vsns Rising
85
15
90
PGood Com parator Delay
PG(Delay)
Vsns Rising
PGood Delay Com parator
Threshold
SS(Delay)
Relative to charge voltage,
SS rising
PGood Delay Com parator
Hysteresis
Delay(SShys)
PGood Leak age Current
PGood Voltage Low
Note4
I(PGDlk)
PG(voltage)
IP good= -5mA
80
mV
Cyc les
VCC-Start-Threshold
Power Good Threshold
uA
256/Fs
V
V
uA
%Vref
s
2
2.1
2.3
V
260
300
340
mV
0
10
uA
0.5
V
Note3: Cold temperature performance is guaranteed via correlation using statistical quality control. Not tested in production.
Note4: Guaranteed by Design but not tested in production.
Note5: Upgrade to industrial/MSL2 level applies from date codes 1227 (marking explained on application note AN1132 page 2).
Products with prior date code of 1227 are qualified with MSL3 for Consumer market.
Rev 5.0
7
PD-97514
IR3859MPbF
Typical Efficiency and Power Loss Curves
Vin=12V, Vcc=5V, Io=0.9A- 9A, Fs=600kHz, Room Temperature, No Air Flow
The table below shows the inductors used for each of the output voltages
in the efficiency measurement.
Vo (V)
1
1.2
1.5
1.8
3.3
5
L (uH)
0.51
0.51
0.68
0.68
1.2
1.2
P/N
59PR9876N
59PR9876N
ETQP4LR68XFC
ETQP4LR68XFC
MPL105-1R2
MPL105-1R2
DCR (mOhm)
0.29
0.29
1.58
1.58
2.9
2.9
97
95
Efficiency (%)
93
91
89
87
85
83
81
79
0.9
1.8
2.7
3.6
4.5
5.4
6.3
7.2
8.1
9.0
Load Current (A)
Power Loss (W)
1.0V
1.2V
1.5V
1.8V
3.3V
5V
3.0
2.8
2.6
2.4
2.2
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0.9
1.8
2.7
3.6
4.5
5.4
6.3
7.2
8.1
9.0
Load Current (A)
1.0V
Rev 5.0
1.2V
1.5V
1.8V
3.3V
5.0V
8
PD-97514
IR3859MPbF
Typical Efficiency and Power Loss Curves
Vin=5V, Vcc=5V, Io=0.9A- 9A, Fs=600kHz, Room Temperature, No Air Flow
The table below shows the inductors used for each of the output voltages
in the efficiency measurement.
Vo (V)
1
1.2
1.5
1.8
3.3
L (uH)
0.4
0.51
0.51
0.51
0.51
P/N
59PR9875N
59PR9876N
59PR9876N
59PR9876N
59PR9876N
DCR (mOhm)
0.29
0.29
0.29
0.29
0.29
97
96
95
94
Efficiency (%)
93
92
91
90
89
88
87
86
85
84
83
0.9
1.8
2.7
1.0V
3.6
4.5
5.4
Load Current (A)
1.2V
6.3
1.5V
7.2
8.1
1.8V
9.0
3.3V
2.7
2.5
2.3
Power Loss (W)
2.1
1.9
1.7
1.5
1.3
1.1
0.9
0.7
0.5
0.3
0.1
0.9
1.8
2.7
3.6
4.5
5.4
6.3
7.2
8.1
9.0
Load Current (A)
1.0Vout
Rev 5.0
1.2Vout
1.5Vout
1.8Vout
3.3Vout
9
PD-97514
IR3859MPbF
Resistance [m Ω ]
Rdson of MOSFETs Over Temperature at Vcc=5V
30
28
26
24
22
20
18
16
14
12
10
8
-40
-20
0
20
40
60
Temperature [ °C]
Sync-FET
80
100
120
Ctrl-FET
Thermal De-rating Curves
Test Conditions: Vin=12V, Vout=1.8V, Vcc=5V, Fs=600kHz, 0- 200LFM
L=0.68uH (ETQP4LR68XFC)
Maximum Load Current (A)
9.0
8.5
8.0
7.5
7.0
6.5
6.0
25
30
35
40
45
50
55
60
65
70
75
80
85
Ambient Temperature (°C)
0 LFM
Rev 5.0
100LFM
200LFM
10
PD-97514
IR3859MPbF
TYPICAL OPERATING CHARACTERISTICS (-40oC - 125oC) Fs=500 kHz
Icc(Dyn)
13.5
270
12.5
250
11.5
230
[mA]
[uA]
Icc(Standby)
290
210
10.5
9.5
8.5
190
7.5
170
6.5
150
-40
-20
0
20
40
60
80
100
5.5
120
-40
o
Temp[ C]
20
60
80
100
120
80
100
120
80
100
120
60
80
100
120
60
80
100
120
IOCSET(500kHz)
54.0
53.0
52.0
520
51.0
510
50.0
[uA]
530
500
490
49.0
48.0
47.0
480
46.0
470
45.0
460
44.0
43.0
450
-40
-20
0
20
40
60
Temp[ oC]
80
100
-40
120
-20
0
20
40
60
Temp[ oC]
Vcc(UVLO) Start
4.46
Vcc(UVLO) Stop
4.16
4.11
4.36
4.06
4.31
4.01
[V]
4.41
4.26
3.96
4.21
3.91
4.16
3.86
4.11
3.81
3.76
4.06
-40
-20
0
20
40
60
o
80
100
-40
120
-20
0
20
Enable(UVLO) Start
1.36
40
60
Temp[ oC]
Temp[ C]
Enable(UVLO) Stop
1.06
1.34
1.04
1.32
1.30
1.02
1.28
1.00
1.26
[V]
[V]
40
Temp[ oC]
540
[kHz]
0
FREQUENCY
550
[V]
-20
1.24
0.98
1.22
0.96
1.20
0.94
1.18
0.92
1.16
1.14
0.90
-40
-20
0
20
40
o
60
80
100
120
-40
-20
0
Temp[ C]
20
40
Temp[ οC]
ISS
26.0
Vfb
711
24.0
706
[uA]
[mV]
22.0
20.0
701
18.0
696
16.0
691
14.0
686
-40
-20
0
20
40
o
60
Temp[ C]
Rev 5.0
80
100
120
-40
-20
0
20
40
o
Temp[ C]
11
PD-97514
IR3859MPbF
Circuit Description
THEORY OF OPERATION
Introduction
The IR3859 uses a PWM voltage mode control
scheme with external compensation to provide
good noise immunity and maximum flexibility in
selecting inductor values and capacitor types.
The switching frequency is programmable from
250kHz to 1.5MHz and provides the capability of
optimizing the design in terms of size and
performance.
IR3859 provides precisely regulated output
voltage programmed via two external resistors
from 0.7V to 0.9*Vin.
If the input to the Enable pin is derived from the
bus voltage by a suitably programmed resistive
divider, it can be ensured that the IR3859 does not
turn on until the bus voltage reaches the desired
level. Only after the bus voltage reaches or
exceeds this level will the voltage at Enable pin
exceed its threshold, thus enabling the IR3859.
Therefore, in addition to being a logic input pin to
enable the IR3859, the Enable feature, with its
precise threshold, also allows the user to
implement an Under-Voltage Lockout for the bus
voltage Vin. This is desirable particularly for high
output voltage applications, where we might want
the IR3859 to be disabled at least until Vin
exceeds the desired output voltage level.
The IR3859 operates with an external bias
supply from 4.5V to 5.5V, allowing an extended
operating input voltage range from 1.5V to 21V.
The device utilizes the on-resistance of the low
side MOSFET as current sense element, this
method enhances the converter’s efficiency and
reduces cost by eliminating the need for external
current sense resistor.
IR3859 includes two low Rds(on) MOSFETs using
IR’s HEXFET technology. These are specifically
designed for high efficiency applications.
Under-Voltage Lockout and POR
The under-voltage lockout circuit monitors the
input supply Vcc and the Enable input. It assures
that the MOSFET driver outputs remain in the off
state whenever either of these two signals drop
below the set thresholds. Normal operation
resumes once Vcc and Enable rise above their
thresholds.
The POR (Power On Ready) signal is generated
when all these signals reach the valid logic level
(see system block diagram). When the POR is
asserted the soft start sequence starts (see soft
start section).
Enable
The Enable features another level of flexibility for
start up. The Enable has precise threshold which
is internally monitored by Under-Voltage Lockout
(UVLO) circuit. Therefore, the IR3859 will turn on
only when the voltage at the Enable pin exceeds
this threshold, typically, 1.2V.
Rev 5.0
Fig. 3a. Normal Start up, Device turns on
when the Bus voltage reaches 10.2V
Figure 3b. shows the recommended start-up
sequence for the non-sequenced operation of
IR3859, when Enable is used as a logic input.
Fig. 3b. Recommended startup sequence,
Non-Sequenced operation
12
PD-97514
IR3859MPbF
Figure 3c. shows the recommended startup
sequence for sequenced operation of IR3859
with Enable used as logic input.
Fig. 5. Pre-Bias startup pulses
Soft-Start
Fig. 3c. Recommended startup sequence,
Sequenced operation
Pre-Bias Startup
The IR3859 has a programmable soft-start to
control the output voltage rise and to limit the
current surge at the start-up. To ensure correct
start-up, the soft-start sequence initiates when
the Enable and Vcc rise above their UVLO
thresholds and generate the Power On Ready
(POR) signal. The internal current source
(typically 20uA) charges the external capacitor
Css linearly from 0V to 3V. Figure 6 shows the
waveforms during the soft start.
The start up time can be estimated by:
(1.4 - 0.7) * CSS
IR3859 is able to start up into pre-charged
output,
which
prevents
oscillation
and
disturbances of the output voltage.
Tstart =
The output starts in asynchronous fashion and
keeps the synchronous MOSFET off until the first
gate signal for control MOSFET is generated.
Figure 4 shows a typical Pre-Bias condition at
start up.
During the soft start the OCP is enabled to
protect the device for any short circuit and over
current condition.
20μA
- - - - - - - - - - - - - - - - - - - - (1)
The synchronous MOSFET always starts with a
narrow pulse width and gradually increases its
duty cycle with a step of 25%, 50%, 75% and
100% until it reaches the steady state value. The
number of these startup pulses for the
synchronous MOSFET is internally programmed.
Figure 5 shows a series of 32, 16, 8 startup
pulses.
Fig. 6. Theoretical operation waveforms
during soft-start
Fig. 4. Pre-Bias startup
Rev 5.0
13
PD-97514
IR3859MPbF
Operating Frequency
The switching frequency can be programmed
between 250kHz – 1500kHz by connecting an
external resistor from Rt pin to Gnd. Table 1
tabulates the oscillator frequency versus Rt.
Table 1. Switching Frequency and IOCSet vs.
External Resistor (Rt)
Rt (kΩ)
47.5
35.7
28.7
23.7
20.5
17.8
15.8
14.3
12.7
11.5
10.7
9.76
9.31
Fs (kHz)
300
400
500
600
700
800
900
1000
1100
1200
1300
1400
1500
Iocset (μA)
29.4
39.2
48.7
59.07
68.2
78.6
88.6
97.9
110.2
121.7
130.8
143.4
150.3
Shutdown
The IR3859 can be shutdown by pulling the
Enable pin below its 1 V threshold. This will tristate both, the high side driver as well as the low
side driver. Alternatively, the output can be
shutdown by pulling the soft-start pin below 0.3V.
Normal operation is resumed by cycling the
voltage at the Soft Start pin.
Over-Current Protection
The over current protection is performed by
sensing current through the RDS(on) of low side
MOSFET. This method enhances the converter’s
efficiency and reduces cost by eliminating a
current sense resistor. As shown in figure 7, an
external resistor (ROCSet) is connected between
OCSet pin and the switch node (SW) which sets
the current limit set point.
An internal current source sources current (IOCSet
) out of the OCSet pin. This current is a function
of Rt and hence, of the free-running switching
frequency.
Rev 5.0
I OCSet (μA ) =
1400
.......... .......... ...............( 2)
R t (kΩ)
Table 1. shows IOCSet at different switching
frequencies. The internal current source
develops a voltage across ROCSet. When the low
side MOSFET is turned on, the inductor current
flows through the Q2 and results in a voltage at
OCSet which is given by:
VOCSet = ( IOCSet ∗ ROCSet ) − ( RDS(on) ∗ I L ) ...........(3)
Fig. 7. Connection of over current sensing resistor
An over current is detected if the OCSet pin goes
below ground. Hence, at the current limit
threshold, VOCset=0. Then, for a current limit
setting ILimit, ROCSet is calculated as follows:
ROCSet =
RDS ( on) * ILimit
IOCSet
........................(4)
An overcurrent detection trips the OCP
comparator, latches OCP signal and cycles the
soft start function in hiccup mode.
The hiccup is performed by shorting the soft-start
capacitor to ground and counting the number of
switching cycles. The Soft Start pin is held low
until 4096 cycles have been completed. The
OCP signal resets and the converter recovers.
After every soft start cycle, the converter stays in
this mode until the overload or short circuit is
removed.
The OCP circuit starts sampling current typically
160 ns after the low gate drive rises to about 3V.
This delay functions to filter out switching noise.
14
PD-97514
IR3859MPbF
Thermal Shutdown
Temperature sensing is provided inside IR3859.
The trip threshold is typically set to 140oC. When
trip threshold is exceeded, thermal shutdown
turns off both MOSFETs and discharges the soft
start capacitor.
1.5V <Vin<16V
4.5V <Vcc<5.5V
Enable
Vin
Boot
Vo(master)
Vcc
SW
PGood
PGood
OCSet
Seq
Automatic restart is initiated when the sensed
temperature drops within the operating range.
There is a 20oC hysteresis in the thermal
shutdown threshold.
RA
Fb
Rt
SS/ SD
RB
Gnd
PGnd
Comp
1.5V <Vin<16V
Output Voltage Sequencing
The
IR3859
can
accommodate
user
programmable sequencing options using Seq,
Enable and Power Good pins.
4.5V <Vcc<5.5V
Enable
Vo(master)
Vin
Boot
Vo(slave)
Vcc
SW
PGood
RE
PGood
OCSet
Seq
RF
Rt
RD
Vo1
SS/ SD
Vo2
RC
Fb
Gnd
PGnd
Comp
Fig. 8b. Application Circuit for Simultaneous
Sequencing
Power-Good and Over-voltage Protection
Simultaneous Powerup
Fig. 8a. Simultaneous Power-up of the slave
with respect to the master.
Through these pins, voltage sequencing such as
simultaneous
and
sequential
can
be
implemented. Figure 8. shows simultaneous
sequencing configurations. In simultaneous
power-up, the voltage at the Seq pin of the slave
reaches 0.7V before the Fb pin of the master. For
RE/RF =RC/RD, therefore, the output voltage of
the slave follows that of the master until the
voltage at the Seq pin of the slave reaches 0.7 V.
After the voltage at the Seq pin of the slave
exceeds 0.85V, the internal 0.7V reference of
the slave dictates its output voltage.
The Vsns pin forms an input to a window
comparator whose upper and lower thresholds
are 0.805V and 0.595V, respectively. Hence, the
Power Good signal is flagged when the Vsns pin
voltage is within the PGood window, i.e.
between 0.595V to 0.805V, as shown in figure 9.
The PGood pin is open drain and it needs to be
externally pulled high. High state indicates that
output is in regulation. Figure 9a shows the
PGood timing diagram for non-tracking
operation. In this case, during startup, PGood
goes high after the SS voltage reaches 2.1V if
the Vsns voltage is within the PGood
comparator window. Figure 9.a and Figure 9.b
also show a 256 cycle delay between the Vsns
voltage entering within the thresholds defined by
the PGood window and PGood going high.
If the output voltage exceeds the over voltage
threshold, an over voltage trip signal asserts, this
will result to turn off the high side driver and turn
on the low side driver until the Vsns voltage
drops below 1.15*Vref threshold. Both drivers are
latched off until a reset performed by cycling
either Vcc or Enable.
The OVP threshold can be externally
programmed to user defined value. Figure 10
shows the response in over-voltage condition.
Rev 5.0
15
PD-97514
IR3859MPbF
TIMING DIAGRAM OF PGOOD FUNCTION
2.1V
1.4V
0.7V
SS
0
1.15*Vref(typical),
+/-5% for Min/Max
PGood window
Vsns
0.85*Vref(typical),
+/-5% for Min/Max
0
At point “A” the power Good
signal goes low, high drive turns
off, low drive turns on till Vsns
is above Over Voltage threshold
and the device latches off. POR
(Vcc/Enable) needs to be
recycled for new start up.
PGood
0
100ns(typical) Delay
100ns(typical) Delay
A
Fig.9a IR3859 Non-Tracking Operation (Seq=Vcc)
256/Fs
Fig.9b IR3859 Tracking Operation
Rev 5.0
16
PD-97514
IR3859MPbF
TIMING DIAGRAM OF Over Voltage Protection
Fig.10 IR3859 Over Voltage Timing Diagram
External Synchronization
The IR3859 incorporates an internal circuit which
enables synchronization of the internal oscillator
(using rising edge) to an external clock. An
external resistor from Rt pin to Gnd is still
required to set the free-running frequency close
to the Sync input frequency. This function is
important to avoid sub-harmonic oscillations due
to beat frequency for embedded systems when
multiple POL (point of load) regulators are used.
The synchronization clock can be applied during
IR3859 normal operation or before IR3859 startup. In any case, IR3859 will perform with the
external after the end of the PreBias cycle.
Applying the external signal to the Sync input
changes the effective value of the ramp signal
(Vramp/Vosc).
Vosc ( eff ) = 1.8 × fFree _ Run fSync ........................ (5)
the frequency of the Sync (fSync) and the freerunning frequency (fFree_Run) results in more
change in the effective amplitude of the ramp
signal.
Therefore, since the ramp amplitude takes part in
calculating the loop-gain and bandwidth of the
regulator, it is recommended not to use a Sync
frequency which is much higher than the freerunning frequency. In addition, the effective value
of the ramp signal, given by equation (5), should
be used when the compensator is designed for
the regulator.
The pulse width of the external clock, which is
applied to the sync, should be greater than 100ns
and its high level should be greater than 2V,
while its lower level is less than 0.6V. If this pin is
left floating, the IC will run with the free running
frequency set by the resistor Rt.
Equation (5) shows that the effective amplitude
of the ramp (Vosc(eff)) is reduced after the external
Sync signal is applied. More difference between
Rev 5.0
17
PD-97514
IR3859MPbF
Minimum on time Considerations
Maximum Duty Ratio Considerations
The minimum ON time is the shortest amount of
time for which the Control FET may be reliably
turned on, and this depends on the internal
timing delays. For the IR3859, the typical
minimum on-time is specified as 50 ns.
Any design or application using the IR3859 must
ensure operation with a pulse width that is higher
than this minimum on-time and preferably higher
than 100 ns. This is necessary for the circuit to
operate without jitter and pulse-skipping, which
can cause high inductor current ripple and high
output voltage ripple.
A fixed off-time of 200 ns maximum is specified
for the IR3859. This provides an upper limit on
the operating duty ratio at any given switching
frequency. It is clear, that higher the switching
frequency, the lower is the maximum duty ratio at
which the IR3859 can operate. To allow a margin
of 50ns, the maximum operating duty ratio in any
application using the IR3859 should still
accommodate about 250 ns off-time. Fig 10.
shows a plot of the maximum duty ratio v/s the
switching frequency, with 250 ns off-time.
=
M a x Duty Cycle
D
Fs
Vout
Vin × Fs
In any application that uses the IR3859, the
following condition must be satisfied:
t on (min) ≤ t on
∴ t on (min) ≤
Vout
Vin × Fs
Vout
t on(min)
The minimum output voltage is limited by the
reference voltage and hence Vout(min) = 0.7 V.
Therefore, for Vout(min) = 0.7 V,
∴ Vin × Fs ≤
∴ Vin × Fs ≤
Max D uty C ycle (%)
t on =
95
90
85
80
75
70
65
60
55
250
450
650
850
1050
1250
1450
1650
S w itchin g Freq uency (kH z )
Fig. 11. Maximum duty cycle v/s switching
frequency.
Vout (min)
t on(min)
0.7 V
∴ Vin × Fs ≤
= 7 × 10 6 V/s
100 ns
Therefore, at the maximum recommended input
voltage 21V and minimum output voltage, the
converter should be designed at a switching
frequency that does not exceed 333 kHz.
Conversely, for operation at the maximum
recommended operating frequency 1.65 MHz
and minimum output voltage, any voltage above
4.2 V may not be stepped down without pulseskipping.
Rev 5.0
18
PD-97514
IR3859MPbF
When an external resistor divider is connected to
the output as shown in figure 12.
Equation (6) can be rewritten as:
Application Information
Design Example:
The following example is a typical application for
IR3859. The application circuit is shown on page
25.
Vin = 12 V ( 13.2V max)
Vo = 1.8 V
Io = 9 A
ΔVo ≤ ± 5% of Vo
Fs = 600 kHz
⎛ V
⎞
R9 = R8 ∗ ⎜⎜ ref ⎟⎟ ..................................(9)
⎝ V o−Vref ⎠
For the calculated values of R8
feedback compensation section.
and R9 see
VOUT
IR3859
IR3624
R8
Fb
R9
Enabling the IR3859
As explained earlier, the precise threshold of the
Enable lends itself well to implementation of a
UVLO for the Bus Voltage.
V in
IR3859
Enable
R1
R2
For a maximum Enable threshold of VEN = 1.36 V
Vin(min) *
R2
= VEN = 1.36V........... (6)
R1 + R2
VEN
R2 = R1
.......... (7)
Vin( min ) − VEN
For a Vin (min)=10.2V, R1=49.9K and R2=7.5K is a
good choice.
Programming the frequency
For Fs = 600 kHz, select Rt = 23.7 kΩ, using
Table. 1.
Output Voltage Programming
Output voltage is programmed by reference
voltage and external voltage divider. The Fb pin
is the inverting input of the error amplifier, which
is internally referenced to 0.7V. The divider is
ratioed to provide 0.7V at the Fb pin when the
output is at its desired value. The output voltage
is defined by using the following equation:
⎛
R ⎞
Vo = Vref ∗ ⎜⎜ 1 + 8 ⎟⎟ .......... .......... .(8)
R9 ⎠
⎝
Rev 5.0
Fig. 12. Typical application of the IR3859 for
programming the output voltage
Soft-Start Programming
The soft-start timing can be programmed by
selecting the soft-start capacitance value. From
(1), for a desired start-up time of the converter,
the soft start capacitor can be calculated by
using:
CSS ( μF ) = Tstart ( ms ) × 0.02857 .......... (10)
Where Tstart is the desired start-up time (ms).
For a start-up time of 3.5ms, the soft-start
capacitor will be 0.099μF. Choose a 0.1μF
ceramic capacitor.
Bootstrap Capacitor Selection
To drive the Control FET, it is necessary to
supply a gate voltage at least 4V greater than
the voltage at the SW pin, which is connected
the source of the Control FET . This is achieved
by using a bootstrap configuration, which
comprises the internal bootstrap diode and an
external bootstrap capacitor (C6), as shown in
Fig. 13. The operation of the circuit is as follows:
When the lower MOSFET is turned on, the
capacitor node connected to SW is pulled down
to ground. The capacitor charges towards Vcc
through the internal bootstrap diode, which has a
forward voltage drop VD. The voltage Vc across
the bootstrap capacitor C6 is approximately
given as
Vc ≅ Vcc − VD .......................... (11)
When the upper MOSFET turns on in the next
cycle, the capacitor node connected to SW rises
to the bus voltage Vin. However, if the value of
C6 is appropriately chosen, the voltage Vc
19
PD-97514
IR3859MPbF
across C6 remains approximately unchanged and
the voltage at the Boot pin becomes:
VBoot ≅ Vin + Vcc − VD ........................................ (12)
Fig. 13. Bootstrap circuit to generate
Vc voltage
A bootstrap capacitor of value 0.1uF is suitable
for most applications.
Inductor Selection
The inductor is selected based on output power,
operating frequency and efficiency requirements.
A low inductor value causes large ripple current,
resulting in the smaller size, faster response to a
load transient but poor efficiency and high output
noise. Generally, the selection of the inductor
value can be reduced to the desired maximum
ripple current in the inductor (Δi ) . The optimum
point is usually found between 20% and 50%
ripple of the output current.
For the buck converter, the inductor value for the
desired operating ripple current can be
determined using the following relation:
1
Δi
; Δt = D ∗
Fs
Δt
....................... (15)
Vo
L = (Vin − Vo ) ∗
Vin ∗ Δi * Fs
Vin − Vo = L ∗
Where:
Vin = Maximum input voltage
Input Capacitor Selection
The ripple current generated during the on time of
the upper MOSFET should be provided by the
input capacitor. The RMS value of this ripple is
expressed by:
Vo = Output Voltage
Δi = Inductor ripple current
F s = Switching frequency
Δt = Turn on time
D = Duty cycle
IRMS = Io ∗ D ∗ ( 1 − D ) ........................(13)
V
D = o ................................ (14)
Vin
If Δi ≈ 42%(Io), then the output inductor is
calculated to be 0.69μH. Select L=0.68 μH.
The ETQP4LR68XFC from Panasonic provides a
compact inductor suitable for this application.
Where:
D is the Duty Cycle
IRMS is the RMS value of the input capacitor
current.
Io is the output current.
For Io=9A and D = 0.15, the IRMS = 3.21A.
Ceramic capacitors are recommended due to
their peak current capabilities. They also feature
low ESR and ESL at higher frequency which
enables better efficiency. For this application, it is
advisable to have 4x10uF 25V ceramic capacitors
C3216X5R1E106M from TDK. In addition to
these, although not mandatory, a 1X330uF, 25V
SMD capacitor EEV-FK1E331P may also be
used as a bulk capacitor and is recommended if
the input power supply is not located close to the
converter.
Rev 5.0
20
PD-97514
IR3859MPbF
Output Capacitor Selection
The voltage ripple and transient requirements
determine the output capacitors type and values.
The criteria is normally based on the value of the
Effective Series Resistance (ESR). However the
actual capacitance value and the Equivalent
Series Inductance (ESL) are other contributing
components. These components can be
described as
ΔVo = ΔVo( ESR ) + ΔVo( ESL ) + ΔVo( C )
The output LC filter introduces a double pole,
–40dB/decade gain slope above its corner
resonant frequency, and a total phase lag of 180o
(see figure 13). The resonant frequency of the LC
filter is expressed as follows:
FLC =
1
2 ∗ π Lo ∗ Co
................................ (17)
Figure 14 shows gain and phase of the LC filter.
Since we already have 180o phase shift from the
output filter alone, the system runs the risk of
being unstable.
ΔVo( ESR ) = ΔIL * ESR
⎛ V − Vo ⎞
ΔVo( ESL ) = ⎜ in
⎟ * ESL
⎝ L ⎠
ΔVo( C ) =
ΔI L
8 * Co * Fs
ΔVo = Output
voltage ripple
ΔIL = Inductor
ripple
............... (16)
current
Since the output capacitor has a major role in the
overall performance of the converter and
determines the result of transient response,
selection of the capacitor is critical. The IR3859
can perform well with all types of capacitors.
As a rule, the capacitor must have low enough
ESR to meet output ripple and load transient
requirements.
The goal for this design is to meet the voltage
ripple requirement in the smallest possible
capacitor size. Therefore it is advisable to select
ceramic capacitors due to their low ESR and ESL
and
small
size.
Six
of
the
TDK
C2102X5R0J226M (22uF, 6.3V, 3mOhm)
capacitors is a good choice.
Feedback Compensation
The IR3859 is a voltage mode controller. The
control loop is a single voltage feedback path
including error amplifier and error comparator. To
achieve fast transient response and accurate
output regulation, a compensation circuit is
necessary. The goal of the compensation
network is to provide a closed-loop transfer
function with the highest 0 dB crossing frequency
and adequate phase margin (greater than 45o).
Rev 5.0
Fig. 14. Gain and Phase of LC filter
The IR3859 uses a voltage-type error amplifier
with high-gain (110dB) and wide-bandwidth. The
output of the amplifier is available for DC gain
control and AC phase compensation.
The error amplifier can be compensated either in
Type-II or Type-III compensation.
Local feedback with Type-II compensation is
shown in figure 14.
This method requires that the output capacitor
should have enough ESR to satisfy stability
requirements.
In
general,
for
Type-II
compensation the output capacitor’s ESR
generates a zero typically at 5kHz to 50kHz
which is essential for an acceptable phase
margin.
The ESR zero of the output capacitor is
expressed as follows:
FESR =
1
........................... (18)
2 ∗ π*ESR*Co
21
PD-97514
IR3859MPbF
Where:
Vin = Maximum Input Voltage
Vosc = Oscillator Ramp Voltage
Fo = Crossover Frequency
FESR = Zero Frequency of the Output Capacitor
FLC = Resonant Frequency of the Output Filter
R8 = Feedback Resistor
To cancel one of the LC filter poles, place the
zero before the LC filter resonant frequency pole:
Fz = 75% FLC
Fz = 0.75 *
Fig. 15. Type II compensation network
and its asymptotic gain plot
The transfer function (Ve/Vo) is given by:
1+ sR3C4
Z
Ve
.....(19)
= H( s ) = − f = −
ZIN
sR8C4
Vo
The (s) indicates that the transfer function varies
as a function of frequency. This configuration
introduces a gain and zero, expressed by:
H (s ) =
Fz =
R3
......... ............................. (20)
R8
1
............................ (21)
2π * R3 * C4
First select the desired zero-crossover frequency
(Fo ):
Fo > FESR and Fo ≤ (1/5 ~ 1/10 ) * Fs
Vosc * Fo * FESR * R8
Vin *
2
FLC
........................... (22)
2π Lo * Co
..................................... (23)
Use equations (21), (22) and (23) to calculate
C4.
One more capacitor is sometimes added in
parallel with C4 and R3. This introduces one
more pole which is mainly used to suppress the
switching noise.
The additional pole is given by:
FP =
1
...............................(24)
C * CPOLE
2π * R3 * 4
C 4 + CPOLE
The pole sets to one half of the switching
frequency which results in the capacitor CPOLE:
CPOLE =
1
1
π*R3*Fs −
C4
≅
1
......................(25)
π*R3*Fs
For a general solution for unconditional stability
for any type of output capacitors, and a wide
range of ESR values, we should implement local
feedback with a Type-III compensation network.
The typically used compensation network for
voltage-mode controller is shown in figure 16.
Again, the transfer function is given by:
Ve
Z
= H(s) = − f
Vo
ZIN
Use the following equation to calculate R3:
R3 =
1
By replacing Zin and Zf according to figure 16,
the transfer function can be expressed as:
− (1 + sR3C4 )[1 + sC7 (R8 + R10 )]
⎡
⎛ C * C3 ⎞ ⎤
⎟⎟⎥ (1 + sR10C7 )
sR8 (C 4 + C3 )⎢1 + sR3 ⎜⎜ 4
⎝ C 4 + C3 ⎠⎦⎥
⎣⎢
....... (26)
H( s ) =
Rev 5.0
22
PD-97514
IR3859MPbF
VOUT
ZIN
C3
C7
R3
R10
C4
R8
Zf
Fb
R9
Gain(dB)
E/A
Comp
Ve
VREF
FZ2
FP2
FP3
Frequency
Fig.16. Type III Compensation network and
its asymptotic gain plot
The compensation network has three poles and
two zeros and they are expressed as follows:
FP 1 = 0 ..............................................................(27)
1
...........................................(28 )
FP 2 =
2π * R10 * C7
1
1
............ (29)
≅
FP 3 =
⎛ C 4 * C3 ⎞ 2π * R3 * C3
⎟⎟
2π * R3 ⎜⎜
⎝ C 4 + C3 ⎠
1
..................................... ........(30)
FZ 1 =
2π * R3 * C 4
1
1
≅
..........(31)
FZ 2 =
2π * C7 * ( R8 + R10 ) 2π * C7 * R8
Cross over frequency is expressed as:
Fo = R3 * C7 *
FESR vs Fo
Output
Capacitor
Type II
FLC<FESR<Fo<Fs/2
Electrolytic
Tantalum
Type III
FLC<Fo<FESR
Tantalum
Ceramic
The higher the crossover frequency, the
potentially faster the load transient response.
However, the crossover frequency should be low
enough to allow attenuation of switching noise.
Typically, the control loop bandwidth or crossover
frequency is selected such that
H(s) dB
FZ1
Compensator
Type
Vin
1
*
....................... (32)
Vosc 2π * Lo * Co
Based on the frequency of the zero generated by
the output capacitor and its ESR, relative to
crossover frequency, the compensation type can
be different. The table below shows the
compensation types for relative locations of the
crossover frequency.
Fo ≤ (1/5 ~ 1/10 ) * Fs
The DC gain should be large enough to provide
high DC-regulation accuracy. The phase margin
should be greater than 45o for overall stability.
For this design we have:
Vin=12V
Vo=1.8V
Vosc=1.8V
Vref=0.7V
Lo=0.68uH
Co=6x22uF, ESR=3mOhm each
It must be noted here that the value of the
capacitance used in the compensator design
must be the small signal value. For instance, the
small signal capacitance of the 22uF capacitor
used in this design (i.e. C3216X5R1E106M from
TDK) is 9.5uF at 1.8 V DC bias and 600 kHz
frequency. It is this value that must be used for all
computations related to the compensation. The
small signal value may be obtained from the
manufacturer’s datasheets, design tools or
SPICE models. Alternatively, they may also be
inferred from measuring the power stage transfer
function of the converter and measuring the
double pole frequency FLC and using equation
(16) to compute the small signal Co.
These result to:
FLC=25.5 kHz
FESR=5.5 MHz
Fs/2=300 kHz
Select crossover frequency Fo=100 kHz
Since FLC<Fo<Fs/2<FESR, Type-III is selected to
place the pole and zeros.
Rev 5.0
23
PD-97514
IR3859MPbF
Detailed calculation of compensation Type-III
Desired Phase Margin Θ = 70o
FZ2 = Fo
1− sin Θ
= 17.63 kHz
1+ sin Θ
FP2 = Fo
1+ sin Θ
= 567.1kHz
1− sin Θ
RDS( on ) = 11 mΩ * 1.25 = 13.75 mΩ
ISET ≅ Io( LIM ) = 9 A * 1.5 = 13.5 A
(50% over nominal output current )
IOCSet = 59.07 μA (at Fs = 600 kHz)
R OCSet = 3.14 kΩ Select
R7 = 3.16 kΩ
Select: FZ1 = 0.5* FZ2 = 8.82 kHz and
FP3 = 0.5* Fs = 300 kHz
Setting the Power Good Threshold
Select: C7 = 2.2nF
Power Good threshold can be programmed by
using two external resistors (R5, R7 on Page 24).
Calculate R3, C3 and C4 :
R3 =
2π * Fo * Lo * Co * Vosc
; R3 = 1.66 kΩ
C7 * Vin
Select: R3 = 1.65 kΩ
1
; C4 = 10.94 nF, Select: C4 = 10 nF
C4 =
2π * FZ1 * R 3
1
C3 =
; C3 = 321 pF, Select: C3 = 270 pF
2π * FP3 * R3
Calculate R10, R8 and R9 :
R10 =
1
; R10 = 128 Ω, Select: R10 = 130 Ω
2π * C7 * FP2
1
- R10; R8 = 3.97 kΩ,
R8 =
2π * C7 * FZ2
V
R9 = ref * R8 ; R9 = 2.56 kΩ Select: R9 = 2.55 kΩ
Vo -Vref
Programming the Current-Limit
The Current-Limit threshold can be set by
connecting a resistor (ROCSET) from the SW pin
to the OCSet pin. The resistor can be calculated
by using equation (4). This resistor ROCSET must
be placed close to the IC.
The RDS(on) has a positive temperature
coefficient and it should be considered for the
worst case operation.
Rev 5.0
ROCSet ∗ IOCSet
R DS ( on )
R6 = (
Vo ( PGood _TH )
0.85 * Vref
− 1) * R7
- - (33)
Where: 0.85*Vref is reference of the internal
comparator, for IR3859.
Vo(PGood_TH) is the selectable output voltage
threshold for power good, for this design it is
1.53V (i.e. 0.85*1.8V).
Select R7=2.55KOhm
Using (24): R5=3.97KOhm
Select R6=4.02KOhm
Select: R8 = 4.02 kΩ
ISET = I L ( critical ) =
The following formula can be used to set the
threshold:
The PGood is an open drain output. Hence, it is
necessary to use a pull up resistor RPG from
PGood pin to Vcc. The value of the pull-up
resistor must be chosen such as to limit the
current flowing into the PGood pin, when the
output voltage is not in regulation, to less than 5
mA. A typical value used is 10kΩ.
.......... . (32)
24
PD-97514
IR3859MPbF
Application Diagram:
Vin=12V
Cin= 4 X 10 uF +
330 uF+1x0.1uF
R1
49.9 K
4.5V <Vcc<5.5V
R2
7.5K
Enable
Seq
Vin
Boot
Vcc
RPG
10 K
R7
4.02 K
OCSet
Vsns
Sync
CSS
0.1 uF
Vo
ROCSet
3.16 K
PGood
Rt
23.7 K
Lo
0.68uH
SW
CVcc
1uF
PGood
C6
0.1 uF
Rt
Fb
SS/ SD
Gnd
PGnd
Comp
R5
2.55 K
C4
10 nF
R3
1.65 K
R8
4.02 K
C7
2.2nF
Co=6X22uF
R10
130
R9
2.55 K
C3
270 pF
Fig. 17. Application circuit diagram for a 12V to 1.8 V, 9A Point Of Load Converter
Suggested Bill of Materials for the application circuit:
Part Reference
Cin
Lo
Co
R1
R2
Rt
RPG
Css C6
R3
C3
C4
R8 R6
R9 R5
Rocset
R10
C7
CVcc
U1
Rev 5.0
Quantity
1
4
1
1
6
1
1
1
1
2
1
1
1
2
2
1
1
1
1
1
Value
330uF
10uF
0.1uF
0.68uH
22uF
49.9k
7.5k
23.7k
10k
0.1uF
1.65k
270pF
10nF
4.02k
2.55k
3.16k
130
2200pF
1.0uF
IR3859
Description
SMD Elecrolytic, Fsize, 25V, 20%
1206, 25V, X5R, 20%
0603, 25V, X7R, 10%
11.7x10x4mm, 20%, 1.58mOhm
0805, 6.3V, X5R, 20%
Thick Film, 0603,1/10 W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
0603, 25V, X7R, 10%
Thick Film, 0603,1/10W,1%
50V, 0603, NPO, 5%
0603, 50V, X7R, 10%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
Thick Film, 0603,1/10W,1%
0603, 50V, X7R, 10%
0603, 16V, X5R, 20%
SupIRBuck, 9A, PQFN 4x5mm
Manufacturer
Panasonic
TDK
Panasonic
Panasonic
TDK
Rohm
Rohm
Rohm
Rohm
Panasonic
Rohm
Panasonic
Panasonic
Rohm
Rohm
Rohm
Panasonic
Panasonic
Panasonic
International Rectifier
Part Number
EEV-FK1E331P
C3216X5R1E106M
ECJ-1VB1E104K
ETQP4LR68XFC
C2102X5R0J226M
MCR03EZPFX4992
MCR03EZPFX7501
MCR03EZPFX2372
MCR03EZPFX1002
ECJ-1VB1E104K
MCR03EZPFX1651
ECJ-1VC1H271J
ECJ-1VB1H103K
MCR03EZPFX4021
MCR03EZPFX2551
MCR03EZPFX3161
ERJ-3EKF1300V
ECJ-1VB1H222K
ECJ-BVB1C105M
IR3859MPbF
25
PD-97514
IR3859MPbF
TYPICAL OPERATING WAVEFORMS
Vin=12.0V, Vcc=5V, Vo=1.8V, Io=0-9A, Room Temperature, No Air Flow
Fig. 18: Start up at 9A Load
Ch1:Vin, Ch2:Vout, Ch3:Vss, Ch4:Enable
Fig. 19: Start up at 9A Load,
Ch1:Vin, Ch2:Vout, Ch3:Vss, Ch4:VPGood
Fig. 20: Start up with 1.62V PreBias, 0A Load, Ch2:Vout, Ch3:VSS
Fig. 21: Output Voltage Ripple, 9A
load Ch2: Vout
Fig. 22: Inductor node at 9A load
Ch2: Switch Node
Fig. 23: Short (Hiccup) Recovery
Ch2:Vout , Ch3:Vss
Rev 5.0
26
PD-97514
IR3859MPbF
TYPICAL OPERATING WAVEFORMS
Vin=12V, Vcc=5V, Vo=1.8V, Io=4.5A- 9A, Room Temperature, No Air Flow
Fig. 24: Transient Response, 4.5A to 9A step 2.5A/μs
Ch2:Vout, Ch4:Iout
Rev 5.0
27
PD-97514
IR3859MPbF
TYPICAL OPERATING WAVEFORMS
Vin=12V, Vcc=5V, Vo=1.8V, Io=9A, Room Temperature, No Air Flow
Fig. 25: Bode Plot at 9A load shows a bandwidth of 92kHz and phase margin of 54
degrees
Fig. 26: Synchronization to 700kHz external clock signal at 9A load
Ch1: SW (Switch Node) Ch2:Sync
Rev 5.0
28
PD-97514
IR3859MPbF
TYPICAL OPERATING WAVEFORMS
Simultaneous Tracking at Power Up and Power Down
Vin=12V, Vo=1.8V, Io=9A, Room Temperature, No Air Flow
VOUT
3.3V
4.02K
R s1
2.55K
Rs2
IR3859
IR3624
Seq
R8 4.02K
Fb
R9 2.55K
Fig. 27: Simultaneous Tracking a 3.3V input at power-up and shut-down
Ch1: SEQ(3.3V) Ch2:SS(1.8V) Ch4: Vout(1.8V)
Rev 5.0
29
PD-97514
IR3859MPbF
Layout Considerations
The layout is very important when designing high
frequency switching converters. Layout will affect
noise pickup and can cause a good design to
perform with less than expected results.
Make all the connections for the power
components in the top layer with wide, copper
filled areas or polygons. In general, it is desirable
to make proper use of power planes and
polygons for power distribution and heat
dissipation.
The inductor, output capacitors and the IR3859
should be as close to each other as possible.
This helps to reduce the EMI radiated by the
power traces due to the high switching currents
through them. Place the input capacitor directly at
the Vin pin of IR3859.
The feedback part of the system should be kept
away from the inductor and other noise sources.
The critical bypass components such as
capacitors for Vcc should be close to their
respective pins. It is important to place the
feedback components including feedback
resistors and compensation components close to
Fb and Comp pins.
The connection between the OCSet resistor and
the SW pin should not share any trace with the
connection between the bootstrap capacitor and
the SW pin. Instead, it is recommended to use a
Kelvin connection of the trace from the OCSet
resistor and the trace from the bootstrap
Vin
PGnd
capacitor at the SW pin.
In a multilayer PCB use one layer as a power
ground plane and have a control circuit ground
(analog ground), to which all signals are
referenced. The goal is to localize the high
current path to a separate loop that does not
Vout analog control
interfere
AGnd with the more sensitive
function. These two grounds must be connected
together on the PC board layout at a single point.
The Power QFN is a thermally enhanced
package. Based on thermal performance it is
recommended to use at least a 4-layers PCB. To
effectively remove heat from the device the
exposed pad should be connected to the ground
plane using vias. Figure 28 illustrates the
implementation of the layout guidelines outlined
above, on the IRDC3859 4 layer demoboard.
Vin
AGnd
PGnd
Vout
Enough copper &
minimum length
ground path between
Input and Output
Compensation parts
should be placed as close
as possible to
the Comp pin.
Vin
PGnd
All bypass caps should
be placed as close as
possible to their
connecting pins.
Resistors Rt, SS cap,
and Rocset should be
placed as close as
possible to their pins.
Vout
AGnd
Fig. 28a. IRDC3859 demoboard layout
considerations – Top Layer
Rev 5.0
30
PD-97514
IR3859MPbF
PGnd
Vin
Single point
connection between
AGND & PGND; It
should be close to the
SupIRBuck, kept
away from noise
sources.
SW
PGnd
Vout
Fig. 28b. IRDC3859 demoboard layout
considerations – Bottom Layer
PGnd
AGnd
Fig. 28c. IRDC3859 demoboard layout
considerations – Mid Layer 1
Use separate trace for
connecting Boost cap and
Rocset to the switch node
and with the minimum
length traces. Avoid big
loops.
Feedback trace should be
kept away form noise
sources
Fig. 28d. IRDC3859 demoboard layout
considerations – Mid Layer 2
Rev 5.0
31
PD-97514
IR3859MPbF
PCB Metal and Components Placement
Evaluations have shown that the best overall performance is achieved using the substrate/PCB layout
as shown in following figures. PQFN devices should be placed to an accuracy of 0.050mm on both X
and Y axes. Self-centering behavior is highly dependent on solders and processes, and experiments
should be run to confirm the limits of self-centering on specific processes. For further information,
please refer to “SupIRBuck™ Multi-Chip Module (MCM) Power Quad Flat No-Lead (PQFN) Board
Mounting Application Note.” (AN-1132)
PCB metal pad sizing (all dimensions in mm)
PCB metal pad spacing (all dimensions in mm)
Rev 5.0
32
PD-97514
IR3859MPbF
Solder Resist
IR recommends that the larger Power or Land Area pads are Solder Mask Defined (SMD.) This allows
the underlying Copper traces to be as large as possible, which helps in terms of current carrying
capability and device cooling capability.
When using SMD pads, the underlying copper traces should be at least 0.05mm larger (on each edge)
than the Solder Mask window, in order to accommodate any layer to layer misalignment. (i.e. 0.1mm in X
& Y.)
However, for the smaller Signal type leads around the edge of the device, IR recommends that these are
Non Solder Mask Defined or Copper Defined.
When using NSMD pads, the Solder Resist Window should be larger than the Copper Pad by at least
0.025mm on each edge, (i.e. 0.05mm in X&Y,) in order to accommodate any layer to layer misalignment.
Ensure that the solder resist in-between the smaller signal lead areas are at least 0.15mm wide, due to
the high x/y aspect ratio of the solder mask strip.
Rev 5.0
33
PD-97514
IR3859MPbF
Stencil Design
Stencils for PQFN can be used with thicknesses of 0.100-0.250mm (0.004-0.010"). Stencils thinner
than 0.100mm are unsuitable because they deposit insufficient solder paste to make good solder
joints with the ground pad; high reductions sometimes create similar problems. Stencils in the range
of 0.125mm-0.200mm (0.005-0.008"), with suitable reductions, give the best results.
Evaluations have shown that the best overall performance is achieved using the stencil design shown
in following figure. This design is for a stencil thickness of 0.127mm (0.005"). The reduction should be
adjusted for stencils of other thicknesses.
Stencil pad sizing (all dimensions in mm)
Stencil pad spacing (all dimensions in mm)
Rev 5.0
34
PD-97514
IR3859MPbF
IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105
TAC Fax: (310) 252-7903
This product has been designed and qualified for the Industrial market (Note5)
Visit us at www.irf.com for sales contact information
Data and specifications subject to change without notice. 08/12
Rev 5.0
35
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