LT8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Description Features n n n n n n n n n n n 6V to 100V Input Voltage Range 260mA, 150V Internal DMOS Power Switch Low Quiescent Current: 70µA in Sleep Mode 330µA in Active Mode Boundary Mode Operation at Heavy Load Low-Ripple Burst Mode® Operation at Light Load Minimum Load <0.5% (Typ) of Full Output VOUT Set with a Single External Resistor No Transformer Third Winding or Opto-Isolator Required for Regulation Accurate EN/UVLO Threshold and Hysteresis Internal Compensation and Soft-Start 5-Lead TSOT-23 Package Applications n n Isolated Telecom, Automotive, Industrial, Medical Power Supplies Isolated Auxiliary/Housekeeping Power Supplies The LT®8300 is a micropower high voltage isolated flyback converter. By sampling the isolated output voltage directly from the primary-side flyback waveform, the part requires no third winding or opto-isolator for regulation. The output voltage is programmed with a single external resistor. Internal compensation and soft-start further reduce external component count. Boundary mode operation provides a small magnetic solution with excellent load regulation. Low ripple Burst Mode operation maintains high efficiency at light load while minimizing the output voltage ripple. A 260mA, 150V DMOS power switch is integrated along with all high voltage circuitry and control logic into a 5-lead ThinSOT™ package. The LT8300 operates from an input voltages range of 6V to 100V and can deliver up to 2W of isolated output power. The high level of integration and the use of boundary and low ripple burst modes result in a simple to use, low component count, and high efficiency application solution for isolated power delivery. L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497, and 7471522. Typical Application 5V Micropower Isolated Flyback Converter Efficiency vs Load Current 100 VOUT+ 5V 1mA TO 300mA 4:1 2.2µF 1M 300µH VIN 19µH • LT8300 EN/UVLO • VOUT– SW 40.2k 210k RFB GND 8300 TA01a 47µF 90 VIN = 36V 80 EFFICIENCY (%) VIN 36V TO 72V 70 VIN = 72V 60 VIN = 48V 50 40 30 20 10 0 0 50 100 150 200 LOAD CURRENT (mA) 250 300 8300 TA01b 8300f 1 LT8300 Absolute Maximum Ratings Pin Configuration (Note 1) TOP VIEW SW (Note 2)............................................................ 150V VIN.......................................................................... 100V EN/UVLO.................................................................... VIN RFB....................................................... VIN – 0.5V to VIN Current into RFB.................................................... 200µA Operating Junction Temperature Range (Notes 3, 4) LT8300E, LT8300I.............................. –40°C to 125°C LT8300H............................................. –40°C to 150°C LT8300MP.......................................... –55°C to 150°C Storage Temperature Range................... –65°C to 150°C EN/UVLO 1 5 VIN GND 2 RFB 3 4 SW S5 PACKAGE 5-LEAD PLASTIC TSOT-23 TJMAX = 150°C, θJA = 150°C/W Order Information LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8300ES5#PBF LT8300ES5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8300IS5#PBF LT8300IS5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8300HS5#PBF LT8300HS5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –40°C to 150°C LT8300MPS5#PBF LT8300MPS5#TRPBF LTGFF 5-Lead Plastic TSOT-23 –55°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 8300f 2 LT8300 Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN unless otherwise noted. SYMBOL PARAMETER VIN Input Voltage Range CONDITIONS MIN TYP 6 MAX UNIT 100 V VIN UVLO Threshold Rising Falling 5.8 3.2 6 V V VIN Quiescent Current VEN/UVLO = 0.3V VEN/UVLO = 1.1V Sleep Mode (Switch Off) Active Mode (Switch On) 1.2 200 70 330 2 µA µA µA µA EN/UVLO Shutdown Threshold For Lowest Off IQ l 0.3 0.75 EN/UVLO Enable Threshold Falling Hysteresis l 1.199 1.223 0.016 1.270 V V IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.3V VEN/UVLO = 1.1V VEN/UVLO = 1.3V –0.1 2.2 –0.1 0 2.5 0 0.1 2.8 0.1 µA µA µA fMAX Maximum Switching Frequency 720 750 780 kHz fMIN Minimum Switching Frequency 6 7.5 9 kHz IQ V tON(MIN) Minimum Switch-On Time 160 ns tOFF(MIN) Minimum Switch-Off Time 350 ns tOFF(MAX) Maximum Switch-Off Time 200 µs ISW(MAX) ISW(MIN) SW Over Current Limit To Initiate Soft-Start 520 mA RDS(ON) Switch On-Resistance ISW = 100mA 10 Ω ILKG Switch Leakage Current VIN = 100V, VSW = 150V 0.1 IRFB RFB Regulation Current Maximum SW Current Limit l 228 260 292 mA Minimum SW Current Limit l 34 52 70 mA RFB Regulation Current Line Regulation tSS Backup Timer l 6V ≤ VIN ≤ 100V Soft-Start Timer Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The SW pin is rated to 150V for transients. Depending on the leakage inductance voltage spike, operating waveforms of the SW pin should be derated to keep the flyback voltage spike below 150V as shown in Figure 5. Note 3: The LT8300E is guaranteed to meet performance specifications from 0°C to 125°C operating junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. 98 0.5 µA 100 102 µA 0.001 0.01 %/V 2.7 ms The LT8300I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT8300H is guaranteed over the full –40°C to 150°C operating junction temperature range. The LT8300MP is guaranteed over the full –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperature greater than 125°C. Note 4: The LT8300 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 150°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. 8300f 3 LT8300 Typical Performance Characteristics Output Load and Line Regulation 5.20 5.5 FRONT PAGE APPLICATION 5.2 5.00 4.95 4.90 5.1 5.0 4.9 4.8 0 50 100 150 200 LOAD CURRENT (mA) 250 4.5 –50 –25 300 VSW 50V/DIV VOUT 50mV/DIV 2µs/DIV FRONT PAGE APPLICATION VIN = 48V, IOUT = 300mA ILPRI 100mA/DIV VSW 50V/DIV VSW 50V/DIV VOUT 50mV/DIV VOUT 50mV/DIV 2µs/DIV FRONT PAGE APPLICATION VIN = 48V, IOUT = 60mA 8300 G06 360 IQ (µA) TJ = 150°C 80 IQ (µA) 300 380 90 8 TJ = 25°C 250 VIN Quiescent Current, Active Mode 100 4 100 150 200 LOAD CURRENT (mA) 20µs/DIV FRONT PAGE APPLICATION VIN = 48V, IOUT = 1mA 8300 G05 VIN Quiescent Current, Sleep Mode 6 50 Burst Mode Waveforms ILPRI 100mA/DIV 8300 G04 VIN Shutdown Current 0 8300 G03 Discontinuous Mode Waveforms ILPRI 100mA/DIV IQ (µA) 0 0 25 50 75 100 125 150 AMBIENT TEMPERATURE (°C) 8300 G02 Boundary Mode Waveforms 2 200 4.6 8300 G01 TJ = 150°C 300 100 4.7 VIN = 36V VIN = 48V VIN = 72V 4.85 FREQUENCY (kHz) 5.05 FRONT PAGE APPLICATION VIN = 48V 400 5.3 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 500 FRONT PAGE APPLICATION VIN = 48V, IOUT = 200mA 5.4 5.10 10 Switching Frequency vs Load Current Output Temperature Variation 5.15 4.80 TA = 25°C, unless otherwise noted. TJ = 25°C 70 60 TJ = 150°C 340 TJ = 25°C 320 TJ = –55°C TJ = –55°C 300 50 TJ = –55°C 0 0 20 40 60 VIN (V) 80 100 8300 G07 40 0 20 40 60 VIN (V) 80 100 8300 G08 280 0 20 40 60 VIN (V) 80 100 8300 G09 8300f 4 LT8300 Typical Performance Characteristics EN/UVLO Enable Threshold TA = 25°C, unless otherwise noted. EN/UVLO Hysteresis Current 1.240 105 5 104 1.235 103 4 1.230 102 1.220 1.215 3 IRFB (µA) 1.225 IHYS (µA) VEN/UVLO (V) RFB Regulation Current 2 101 100 99 98 1.210 97 1 1.205 96 1.200 –50 –25 0 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 95 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 8300 G10 0 25 50 75 100 125 150 TEMPERATURE (°C) 8300 G12 8300 G11 RDS(ON) Switch Current Limit Maximum Switching Frequency 300 25 1000 MAXIMUM CURRENT LIMIT 250 800 10 FREQUENCY (kHz) 200 15 ISW (mA) RESISTANCE (Ω) 20 150 100 5 MINIMUM CURRENT LIMIT 50 0 –50 –25 0 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 Minimum Switching Frequency 0 –50 –25 400 300 300 TIME (ns) FREQUENCY (kHz) TIME (ns) 25 50 75 100 125 150 TEMPERATURE (°C) 200 8300 G16 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) Minimum Switch-Off Time 400 100 0 0 8300 G15 Minimum Switch-On Time 16 4 0 –50 –25 8300 G14 20 8 400 200 25 50 75 100 125 150 TEMPERATURE (°C) 8300 G13 12 600 200 100 0 25 50 75 100 125 150 TEMPERATURE (°C) 8300 G17 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 8300 G18 8300f 5 LT8300 Pin Functions EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The EN/UVLO pin is used to enable the LT8300. Pull the pin below 0.3V to shut down the LT8300. This pin has an accurate 1.223V threshold and can be used to program a VIN undervoltage lockout (UVLO) threshold using a resistor divider from VIN to ground. A 2.5µA current hysteresis allows the programming of VIN UVLO hysteresis. If neither function is used, tie this pin directly to VIN. mary SW pin. The ratio of the RFB resistor to the internal trimmed 12.23k resistor, times the internal bandgap reference, determines the output voltage (plus the effect of any non-unity transformer turns ratio). Minimize trace area at this pin. GND (Pin 2): Ground. Tie this pin directly to local ground plane. VIN (Pin 5): Input Supply. The VIN pin supplies current to internal circuitry and serves as a reference voltage for the feedback circuitry connected to the RFB pin. Locally bypass this pin to ground with a capacitor. RFB (Pin 3): Input Pin for External Feedback Resistor. Connect a resistor from this pin to the transformer pri- SW (Pin 4): Drain of the 150V Internal DMOS Power Switch. Minimize trace area at this pin to reduce EMI and voltage spikes. 8300f 6 LT8300 Block Diagram T1 NPS:1 VIN CIN LPRI • • DOUT VOUT+ LSEC COUT RFB 5 3 VIN VOUT– 4 RFB SW BOUNDARY DETECTOR 1:4 M3 M2 OSCILLATOR – 25µA RREF 12.23kΩ 1.223V + – gm + S A3 R Q DRIVER M1 R1 1 – EN/UVLO 2.5µA R2 1.223V M4 + + A2 A1 VIN REFERENCE REGULATORS – RSENSE 0.3Ω GND 2 8300 BD 8300f 7 LT8300 Operation The LT8300 is a current mode switching regulator IC designed specially for the isolated flyback topology. The key problem in isolated topologies is how to communicate the output voltage information from the isolated secondary side of the transformer to the primary side for regulation. Historically, opto-isolators or extra transformer windings communicate this information across the isolation boundary. Opto-isolator circuits waste output power, and the extra components increase the cost and physical size of the power supply. Opto-isolators can also cause system issues due to limited dynamic response, nonlinearity, unitto-unit variation and aging over lifetime. Circuits employing extra transformer windings also exhibit deficiencies, as using an extra winding adds to the transformer’s physical size and cost, and dynamic response is often mediocre. The LT8300 samples the isolated output voltage through the primary-side flyback pulse waveform. In this manner, neither opto-isolator nor extra transformer winding is required for regulation. Since the LT8300 operates in either boundary conduction mode or discontinuous conduction mode, the output voltage is always sampled on the SW pin when the secondary current is zero. This method improves load regulation without the need of external load compensation components. The LT8300 is a simple to use micropower isolated flyback converter housed in a 5-lead TSOT-23 package. The output voltage is programmed with a single external resistor. By integrating the loop compensation and soft-start inside, the part further reduces the number of external components. As shown in the Block Diagram, many of the blocks are similar to those found in traditional switching regulators including reference, regulators, oscillator, logic, current amplifier, current comparator, driver, and power switch. The novel sections include a flyback pulse sense circuit, a sample-and-hold error amplifier, and a boundary mode detector, as well as the additional logic for boundary conduction mode, discontinuous conduction mode, and low ripple Burst Mode operation. Boundary Conduction Mode Operation The LT8300 features boundary conduction mode operation at heavy load, where the chip turns on the primary power switch when the secondary current is zero. Boundary conduction mode is a variable frequency, variable peakcurrent switching scheme. The power switch turns on and the transformer primary current increases until an internally controlled peak current limit. After the power switch turns off, the voltage on the SW pin rises to the output voltage multiplied by the primary-to-secondary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero, the SW pin voltage collapses and rings around VIN. A boundary mode detector senses this event and turns the power switch back on. Boundary conduction mode returns the secondary current to zero every cycle, so parasitic resistive voltage drops do not cause load regulation errors. Boundary conduction mode also allows the use of smaller transformers compared to continuous conduction mode and does not exhibit sub-harmonic oscillation. Discontinuous Conduction Mode Operation As the load gets lighter, boundary conduction mode increases the switching frequency and decreases the switch peak current at the same ratio. Running at a higher switching frequency up to several MHz increases switching and gate charge losses. To avoid this scenario, the LT8300 has an additional internal oscillator, which clamps the maximum switching frequency to be less than 750kHz. Once the switching frequency hits the internal frequency clamp, the part starts to delay the switch turn-on and operates in discontinuous conduction mode. Low Ripple Burst Mode Operation Unlike traditional flyback converters, the LT8300 has to turn on and off at least for a minimum amount of time and with a minimum frequency to allow accurate sampling of the output voltage. The inherent minimum switch current limit and minimum switch-off time are necessary to guarantee the correct operation of specific applications. As the load gets very light, the LT8300 starts to fold back the switching frequency while keeping the minimum switch current limit. So the load current is able to decrease while still allowing minimum switch-off time for the sampleand-hold error amplifier. Meanwhile, the part switches between sleep mode and active mode, thereby reducing the 8300f 8 LT8300 Operation effective quiescent current to improve light load efficiency. In this condition, the LT8300 operates in low ripple Burst Mode. The typical 7.5kHz minimum switching frequency determines how often the output voltage is sampled and also the minimum load requirement. Applications Information Output Voltage The RFB resistor as depicted in the Block Diagram is the only external resistor used to program the output voltage. The LT8300 operates similar to traditional current mode switchers, except in the use of a unique flyback pulse sense circuit and a sample-and-hold error amplifier, which sample and therefore regulate the isolated output voltage from the flyback pulse. Operation is as follows: when the power switch M1 turns off, the SW pin voltage rises above the VIN supply. The amplitude of the flyback pulse, i.e., the difference between the SW pin voltage and VIN supply, is given as: VFLBK = (VOUT + VF + ISEC • ESR) • NPS VF = Output diode forward voltage ISEC = Transformer secondary current ESR = Total impedance of secondary circuit NPS = Transformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current IRFB by the flyback pulse sense circuit (M2 and M3). This current IRFB also flows through the internal trimmed 12.23k RREF resistor to generate a ground-referred voltage. The resulting voltage feeds to the inverting input of the sampleand-hold error amplifier. Since the sample-and-hold error amplifier samples the voltage when the secondary current is zero, the (ISEC • ESR) term in the VFLBK equation can be assumed to be zero. The bandgap reference voltage VBG, 1.223V, feeds to the non-inverting input of the sample-and-hold error amplifier. The relatively high gain in the overall loop causes the voltage across RREF resistor to be nearly equal to the bandgap reference voltage VBG. The resulting relationship between VFLBK and VBG can be expressed as: VFLBK R • RREF = VBG FB or V VFLBK = BG • RFB = I RFB • RFB RREF VBG = Bandgap reference voltage IRFB = RFB regulation current = 100µA Combination with the previous VFLBK equation yields an equation for VOUT, in terms of the RFB resistor, transformer turns ratio, and diode forward voltage: R VOUT = 100µA • FB − VF NPS Output Temperature Coefficient The first term in the VOUT equation does not have temperature dependence, but the output diode forward voltage VF has a significant negative temperature coefficient (–1mV/°C to –2mV/°C). Such a negative temperature coefficient produces approximately 200mV to 300mV voltage variation on the output voltage across temperature. For higher voltage outputs, such as 12V and 24V, the output diode temperature coefficient has a negligible effect on the output voltage regulation. For lower voltage outputs, such as 3.3V and 5V, however, the output diode temperature coefficient does count for an extra 2% to 5% output voltage regulation. For customers requiring tight output voltage regulation across temperature, please refer to other LTC parts with integrated temperature compensation features. 8300f 9 LT8300 Applications Information Selecting Actual RFB Resistor Value Output Power The LT8300 uses a unique sampling scheme to regulate the isolated output voltage. Due to the sampling nature, the scheme contains repeatable delays and error sources, which will affect the output voltage and force a re-evaluation of the RFB resistor value. Therefore, a simple two-step process is required to choose feedback resistor RFB. A flyback converter has a complicated relationship between the input and output currents compared to a buck or a boost converter. A boost converter has a relatively constant maximum input current regardless of input voltage and a buck converter has a relatively constant maximum output current regardless of input voltage. This is due to the continuous non-switching behavior of the two currents. A flyback converter has both discontinuous input and output currents which make it similar to a non-isolated buck-boost converter. The duty cycle will affect the input and output currents, making it hard to predict output power. In addition, the winding ratio can be changed to multiply the output current at the expense of a higher switch voltage. Rearrangement of the expression for VOUT in the Output Voltage section yields the starting value for RFB: RFB = ( NPS • VOUT + VF 100µA ) VOUT = Output voltage VF = Output diode forward voltage = ~0.3V NPS = Transformer effective primary-to-secondary turns ratio Power up the application with the starting RFB value and other components connected, and measure the regulated output voltage, VOUT(MEAS). The final RFB value can be adjusted to: VOUT RFB(FINAL) = • RFB VOUT(MEAS) Once the final RFB value is selected, the regulation accuracy from board to board for a given application will be very consistent, typically under ±5% when including device variation of all the components in the system (assuming resistor tolerances and transformer windings matching within ±1%). However, if the transformer or the output diode is changed, or the layout is dramatically altered, there may be some change in VOUT. The graphs in Figures 1 to 4 show the typical maximum output power possible for the output voltages 3.3V, 5V, 12V, and 24V. The maximum output power curve is the calculated output power if the switch voltage is 120V during the switch-off time. 30V of margin is left for leakage inductance voltage spike. To achieve this power level at a given input, a winding ratio value must be calculated to stress the switch to 120V, resulting in some odd ratio values. The curves below the maximum output power curve are examples of common winding ratio values and the amount of output power at given input voltages. One design example would be a 5V output converter with a minimum input voltage of 36V and a maximum input voltage of 72V. A six-to-one winding ratio fits this design example perfectly and outputs equal to 2.44W at 72V but lowers to 1.87W at 36V. The following equations calculate output power: POUT = η • VIN • D •ISW(MAX) • 0.5 η = Efficiency = 85% ( VOUT + VF ) • NPS D = DutyCycle = ( VOUT + VF ) • NPS + VIN ISW(MAX) = Maximum switch current limit = 260mA 8300f 10 LT8300 Applications Information 3.5 3.5 MAXIMUM OUTPUT POWER 2.5 N = 8:1 2.0 N = 6:1 1.5 N = 4:1 1.0 0.5 0 MAXIMUM OUTPUT N = 8:1 POWER 3.0 N = 12:1 OUTPUT POWER (W) OUTPUT POWER (W) 3.0 2.5 N = 6:1 N = 4:1 2.0 1.5 N = 2:1 1.0 0.5 0 20 40 60 INPUT VOLTAGE (V) 0 100 80 0 20 40 60 INPUT VOLTAGE (V) 8300 F01 8300 F02 Figure 1. Output Power for 3.3V Output 3.5 3.5 N = 2:1 3.0 3.0 N = 3:1 MAXIMUM OUTPUT POWER OUTPUT POWER (W) OUTPUT POWER (W) Figure 2. Output Power for 5V Output N = 4:1 2.5 N = 2:1 2.0 1.5 N = 1:1 1.0 0.5 0 N = 3:2 MAXIMUM OUTPUT POWER 2.5 2.0 N = 1:1 1.5 N = 1:2 1.0 0.5 0 20 40 60 INPUT VOLTAGE (V) 0 100 80 0 20 40 60 INPUT VOLTAGE (V) 8300 F03 Primary Inductance Requirement The LT8300 obtains output voltage information from the reflected output voltage on the SW pin. The conduction of secondary current reflects the output voltage on the primary SW pin. The sample-and-hold error amplifier needs a minimum 350ns to settle and sample the reflected output voltage. In order to ensure proper sampling, the secondary winding needs to conduct current for a minimum of 350ns. The following equation gives the minimum value for primary-side magnetizing inductance: ( tOFF(MIN) • NPS • VOUT + VF ISW(MIN) ) tOFF(MIN) = Minimum switch-off time = 350ns ISW(MIN) = Minimum switch current limit = 52mA 80 100 8300 F04 Figure 3. Output Power for 12V Output LPRI ≥ 100 80 Figure 4. Output Power for 24V Output In addition to the primary inductance requirement for the minimum switch-off time, the LT8300 has minimum switch-on time that prevents the chip from turning on the power switch shorter than approximately 160ns. This minimum switch-on time is mainly for leading-edge blanking the initial switch turn-on current spike. If the inductor current exceeds the desired current limit during that time, oscillation may occur at the output as the current control loop will lose its ability to regulate. Therefore, the following equation relating to maximum input voltage must also be followed in selecting primary-side magnetizing inductance: LPRI ≥ tON(MIN) • VIN(MAX) ISW(MIN) tON(MIN) = Minimum Switch-On Time = 160ns 8300f 11 LT8300 Applications Information In general, choose a transformer with its primary magnetizing inductance about 20% to 40% larger than the minimum values calculated above. A transformer with much larger inductance will have a bigger physical size and may cause instability at light load. Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed flyback transformers for use with the LT8300. Table 1 shows the details of these transformers. Selecting a Transformer Note that when choosing the RFB resistor to set output voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application. In contrast, the use of simple ratios of small integers, e.g., 4:1, 2:1, 1:1, provides more freedom in settling total turns and mutual inductance. Transformer specification and design is perhaps the most critical part of successfully applying the LT8300. In addition to the usual list of guidelines dealing with high frequency isolated power supply transformer design, the following information should be carefully considered. Turns Ratio Table 1. Predesigned Transformers — Typical Specifications TRANSFORMER PART NUMBER LPRI (µH) LLEAKAGE (µH) NP:NS:NB VENDOR 750312367 400 4.5 8:1 Würth Elektronik 48V to 3.3V/0.51A, 24V to 3.3V/0.37A, 12V to 3.3V/0.24A 750312557 300 2.5 6:1 Würth Elektronik 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A 48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A 750312365 300 1.8 4:1 Würth Elektronik 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A 750312558 300 1.75 2:1:1 Würth Elektronik 48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA 48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA 750312559 300 2 1:1 Würth Elektronik 48V to 24V/67mA, 24V to 24V/50mA, 12V to 24V/33mA 750311019 400 5 6:1:2 Würth Elektronik 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A 48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A 750311558 300 1.5 4:1:1 Würth Elektronik 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A 750311660 350 3 2:1:0.33 Würth Elektronik 48V to 12V/0.134A, 24V to 12V/0.1A, 12V to 12V/0.066A 48V to 15V/0.124A, 24V to 15V/0.088A, 12V to 15V/0.056A 750311838 350 3 2:1:1 Würth Elektronik 48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA 48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA 48V to 24V/67mA, 24V to 24V/50mA, 12V to 24V/33mA TARGET APPLICATIONS 750311659 300 2 1:1:0.2 Würth Elektronik 10396-T026 300 2.5 6:1:2 Sumida 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A 48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A 10396-T024 300 2 4:1:1 Sumida 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A 10396-T022 300 2 2:1:0.33 Sumida 48V to 12V/0.134A, 24V to 12V/0.1A, 12V to 12V/0.066A 48V to 15V/0.124A, 24V to 15V/0.088A, 12V to 15V/0.056A 10396-T028 300 2.5 2:1:1 Sumida 48V to ±12V/67mA, 24V to ±12V/50mA, 12V to ±12V/33mA 48V to ±15V/62mA, 24V to ±15V/44mA, 12V to ±15V/28mA L10-0116 500 7.3 6:1 BH Electronics 48V to 3.3V/0.42A, 24V to 3.3V/0.32A, 12V to 3.3V/0.22A 48V to 5V/0.38A, 24V to 5V/0.27A, 12V to 5V/0.17A L10-0112 230 3.38 4:1 BH Electronics 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A L11-0067 230 2.16 4:1 BH Electronics 48V to 5V/0.29A, 24V to 5V/0.22A, 12V to 5V/0.15A * All the transformers are rated for 1.5kV Isolation. 8300f 12 LT8300 Applications Information Typically, choose the transformer turns ratio to maximize available output power. For low output voltages (3.3V or 5V), a larger N:1 turns ratio can be used with multiple primary windings relative to the secondary to maximize the transformer’s current gain (and output power). However, remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. In addition, leakage inductance will cause a voltage spike (VLEAKAGE) on top of this reflected voltage. This total quantity needs to remain below the 150V absolute maximum rating of the SW pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio, NPS, for a given application. Choose a turns ratio low enough to ensure: NPS < 150V − VIN(MAX) − VLEAKAGE VOUT + VF For lower output power levels, choose a smaller N:1 turns ratio to alleviate the SW pin voltage stress. Although a 1:N turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch, the multiplied parasitic capacitance through turns ratio coupled with the relatively resistive 150V internal power switch may cause the switch turn-on current spike ringing beyond 160ns leading-edge blanking, thereby producing light load instability in certain applications. So any 1:N turns ratio should be fully evaluated before its use with the LT8300. The turns ratio is an important element in the isolated feedback scheme, and directly affects the output voltage accuracy. Make sure the transformer manufacturer specifies turns ratio accuracy within ±1%. Saturation Current The current in the transformer windings should not exceed its rated saturation current. Energy injected once the core is saturated will not be transferred to the secondary and will instead be dissipated in the core. When designing custom transformers to be used with the LT8300, the saturation current should always be specified by the transformer manufacturers. Winding Resistance Resistance in either the primary or secondary windings will reduce overall power efficiency. Good output voltage regulation will be maintained independent of winding resistance due to the boundary/discontinuous conduction mode operation of the LT8300. Leakage Inductance and Snubbers Transformer leakage inductance on either the primary or secondary causes a voltage spike to appear on the primary after the power switch turns off. This spike is increasingly prominent at higher load currents where more stored energy must be dissipated. It is very important to minimize transformer leakage inductance. When designing an application, adequate margin should be kept for the worst-case leakage voltage spikes even under overload conditions. In most cases shown in Figure 5, the reflected output voltage on the primary plus VIN should be kept below 120V. This leaves at least 30V margin for the leakage spike across line and load conditions. A larger voltage margin will be required for poorly wound transformers or for excessive leakage inductance. In addition to the voltage spikes, the leakage inductance also causes the SW pin ringing for a while after the power switch turns off. To prevent the voltage ringing falsely trigger boundary mode detector, the LT8300 internally blanks the boundary mode detector for approximately 250ns. Any remaining voltage ringing after 250ns may turn the power switch back on again before the secondary current falls to zero. So the leakage inductance spike ringing should be limited to less than 250ns. 8300f 13 LT8300 Applications Information VSW VSW <150V VSW <150V <150V VLEAKAGE VLEAKAGE <120V VLEAKAGE <120V <120V tOFF > 350ns tOFF > 350ns tOFF > 350ns tSP < 250ns tSP < 250ns tSP < 250ns TIME TIME No Snubber TIME with DZ Snubber with RC Snubber 8300 F05 Figure 5. Maximum Voltages for SW Pin Flyback Waveform Lℓ Lℓ • Z D • C • R 8300 F06a • 8300 F06b DZ Snubber RC Snubber Figure 6. Snubber Circuits A snubber circuit is recommended for most applications. Two types of snubber circuits shown in Figure 6 that can protect the internal power switch include the DZ (diodeZener) snubber and the RC (resistor-capacitor) snubber. The DZ snubber ensures well defined and consistent clamping voltage and has slightly higher power efficiency, while the RC snubber quickly damps the voltage spike ringing and provides better load regulation and EMI performance. Figure 5 shows the flyback waveforms with the DZ and RC snubbers. For the DZ snubber, proper care must be taken when choosing both the diode and the Zener diode. Schottky diodes are typically the best choice, but some PN diodes can be used if they turn on fast enough to limit the leakage inductance spike. Choose a diode that has a reversevoltage rating higher than the maximum SW pin voltage. The Zener diode breakdown voltage should be chosen to balance power loss and switch voltage protection. The best compromise is to choose the largest voltage breakdown. Use the following equation to make the proper choice: VZENER(MAX) ≤ 150V – VIN(MAX) For an application with a maximum input voltage of 72V, choose a 68V Zener diode, the VZENER(MAX) of which is around 72V and below the 78V maximum. The power loss in the clamp will determine the power rating of the Zener diode. Power loss in the clamp is highest at maximum load and minimum input voltage. The switch current is highest at this point along with the energy stored in the leakage inductance. A 0.5W Zener will satisfy most applications when the highest VZENER is chosen. 8300f 14 LT8300 Applications Information Tables 2 and 3 show some recommended diodes and Zener diodes. Table 2. Recommended Zener Diodes VZENER (V) POWER (W) CASE VENDOR MMSZ5266BT1G 68 0.5 SOD-123 On Semi MMSZ5270BT1G 91 0.5 SOD-123 CMHZ5266B 68 0.5 SOD-123 CMHZ5267B 75 0.5 SOD-123 BZX84J-68 68 0.5 SOD323F NXP BZX100A 100 0.5 SOD323F PART Central Semiconductor Table 3. Recommended Diodes PART I (A) VREVERSE (V) BAV21W 0.625 200 SOD-123 Diodes Inc. BAV20W 0.625 150 SOD-123 CASE VENDOR The recommended approach for designing an RC snubber is to measure the period of the ringing on the SW pin when the power switch turns off without the snubber and then add capacitance (starting with 100pF) until the period of the ringing is 1.5 to 2 times longer. The change in period will determine the value of the parasitic capacitance, from which the parasitic inductance can be determined from the initial period, as well. Once the value of the SW node capacitance and inductance is known, a series resistor can be added to the snubber capacitance to dissipate power and critically dampen the ringing. The equation for deriving the optimal series resistance using the observed periods ( tPERIOD and tPERIOD(SNUBBED)) and snubber capacitance (CSNUBBER) is: CPAR = CSNUBBER 2 Note that energy absorbed by the RC snubber will be converted to heat and will not be delivered to the load. In high voltage or high current applications, the snubber may need to be sized for thermal dissipation. Undervoltage Lockout (UVLO) A resistive divider from VIN to the EN/UVLO pin implements undervoltage lockout (UVLO). The EN/UVLO pin falling threshold is set at 1.223V with 16mV hysteresis. In addition, the EN/UVLO pin sinks 2.5µA when the voltage at the pin is below 1.223V. This current provides user programmable hysteresis based on the value of R1. The programmable UVLO thresholds are: 1.239V • (R1+ R2) + 2.5µA • R1 R2 1.223V • (R1+ R2) VIN(UVLO−) = R2 VIN(UVLO+) = Figure 7 shows the implementation of external shutdown control while still using the UVLO function. The NMOS grounds the EN/UVLO pin when turned on, and puts the LT8300 in shutdown with quiescent current less than 2µA. VIN R1 EN/UVLO LT8300 R2 RUN/STOP CONTROL (OPTIONAL) GND 8300 F07 Figure 7. Undervoltage Lockout (UVLO) tPERIOD(SNUBBED) − 1 t PERIOD L PAR = tPERIOD 2 CPAR • 4π 2 RSNUBBER = LPAR CPAR 8300f 15 LT8300 Applications Information Minimum Load Requirement Design Example The LT8300 samples the isolated output voltage from the primary-side flyback pulse waveform. The flyback pulse occurs once the primary switch turns off and the secondary winding conducts current. In order to sample the output voltage, the LT8300 has to turn on and off at least for a minimum amount of time and with a minimum frequency. The LT8300 delivers a minimum amount of energy even during light load conditions to ensure accurate output voltage information. The minimum energy delivery creates a minimum load requirement, which can be approximately estimated as: Use the following design example as a guide to design applications for the LT8300. The design example involves designing a 12V output with a 120mA load current and an input range from 36V to 72V. I LOAD(MIN) = 2 L PRI • I SW(MIN) • f MIN 2 • VOUT LPRI = Transformer primary inductance ISW(MIN) = Minimum switch current limit = 52mA fMIN = Minimum switching frequency = 7.5kHz The LT8300 typically needs less than 0.5% of its full output power as minimum load. Alternatively, a Zener diode with its breakdown of 20% higher than the output voltage can serve as a minimum load if pre-loading is not acceptable. For a 5V output, use a 6V Zener with cathode connected to the output. Output Short Protection When the output is heavily overloaded or shorted, the reflected SW pin waveform rings longer than the internal blanking time. After the 350ns minimum switch-off time, the excessive ring falsely trigger the boundary mode detector and turn the power switch back on again before the secondary current falls to zero. Under this condition, the LT8300 runs into continuous conduction mode at 750kHz maximum switching frequency. Depending on the VIN supply voltage, the switch current may run away and exceed 260mA maximum current limit. Once the switch current hits 520mA over current limit, a soft-start cycle initiates and throttles back both switch current limit and switch frequency. This output short protection prevents the switch current from running away and limits the average output diode current. VIN(MIN) = 36V, VIN(NOM) = 48V, VIN(MAX) = 72V, VOUT = 12V, IOUT = 120mA Step 1: Select the Transformer Turns Ratio. NPS < 150V − VIN(MAX) − VLEAKAGE VOUT + VF VLEAKAGE = Margin for transformer leakage spike = 30V VF = Output diode forward voltage = ~0.3V Example: NPS < 150V − 72V − 30V = 3.9 12V + 0.3V The choice of transformer turns ratio is critical in determining output current capability of the converter. Table 4 shows the switch voltage stress and output current capability at different transformer turns ratio. Table 4. Switch Voltage Stress and Output Current Capability vs Turns Ratio NPS VSW(MAX) at VIN(MAX) (V) IOUT(MAX) at VIN(MIN) (mA) DUTY CYCLE (%) 1:1 84.3 84 15-25 2:1 96.6 135 25-41 3:1 108.9 168 34-51 Since both NPS = 2 and NPS = 3 can meet the 120mA output current requirement, NPS = 2 is chosen in this example to allow more margin for transformer leakage inductance voltage spike. 8300f 16 LT8300 Applications Information Step 2: Determine the Primary Inductance. Primary inductance for the transformer must be set above a minimum value to satisfy the minimum switch-off and switch-on time requirements: LPRI ≥ LPRI ≥ ( tOFF(MIN) • NPS • VOUT + VF ISW(MIN) ) tON(MIN) • VIN(MAX) ISW(MIN) tOFF(MIN) = 350ns tON(MIN) = 160ns ISW(MIN) = 52mA Example: 350ns • 2 • (12V + 0.3V) = 166µH 52mA 160ns • 72V LPRI ≥ = 222µH 52mA LPRI ≥ Most transformers specify primary inductance with a tolerance of ±20%. With other component tolerance considered, choose a transformer with its primary inductance 20% to 40% larger than the minimum values calculated above. LPRI = 300µH is then chosen in this example. Once the primary inductance has been determined, the maximum load switching frequency can be calculated as: fSW = ISW = 1 1 = LPRI •ISW tON + tOFF LPRI •ISW + VIN NPS • (VOUT + VF ) VOUT •IOUT • 2 η • VIN • D Example: (12V + 0.3V) • 2 = 0.34 (12V + 0.3V) • 2 + 48V 12V • 0.12A • 2 = 0.21A ISW = 0.85 • 48V • 0.34 fSW = 260kHz D= The transformer also needs to be rated for the correct saturation current level across line and load conditions. A saturation current rating larger than 400mA is necessary to work with the LT8300. The 10396-T022 from Sumida is chosen as the flyback transformer. Step 3: Choose the Output Diode. Two main criteria for choosing the output diode include forward current rating and reverse voltage rating. The maximum load requirement is a good first-order guess as the average current requirement for the output diode. A conservative metric is the maximum switch current limit multiplied by the turns ratio, IDIODE(MAX) = ISW(MAX) • NPS Example: IDIODE(MAX) = 0.52A Next calculate reverse voltage requirement using maximum VIN: VREVERSE = VOUT + VIN(MAX) NPS Example: VREVERSE = 12V + 72V = 48V 2 The SBR0560S1 (0.5A, 60V diode) from Diodes Inc. is chosen. 8300f 17 LT8300 Applications Information Step 4: Choose the Output Capacitor. The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor. Use the equation below to calculate the output capacitance: COUT = L PRI • I SW 2 2 • VOUT • ∆VOUT Example: Design for output voltage ripple less than 1% of VOUT, i.e., 120mV. COUT 300µH • (0.21A)2 = = 4.6µF 2 • 12V • 0.12V Remember ceramic capacitors lose capacitance with applied voltage. The capacitance can drop to 40% of quoted capacitance at the maximum voltage rating. So a 10uF, 16V rating ceramic capacitor is chosen. Step 5: Design Snubber Circuit. The snubber circuit protects the power switch from leakage inductance voltage spike. A DZ snubber is recommended for this application because of lower leakage inductance and larger voltage margin. The Zener and the diode need to be selected. The maximum Zener breakdown voltage is set according to the maximum VIN: VZENER(MAX) ≤ 150V – VIN(MAX) A 68V Zener with a maximum of 72V will provide optimal protection and minimize power loss. So a 68V, 0.5W Zener from On Semiconductor (MMSZ5266BT1G) is chosen. Choose a diode that is fast and has sufficient reverse voltage breakdown: VREVERSE > VSW(MAX) VSW(MAX) = VIN(MAX) + VZENER(MAX) Example: VREVERSE > 144V A 150V, 0.6A diode from Diodes Inc. (BAV20W) is chosen. Step 6: Select the RFB Resistor. Use the following equation to calculate the starting value for RFB: RFB = NPS • (VOUT + VF ) 100µA Example: RFB = 2 • (12V + 0.3V) = 246k 100µA Depending on the tolerance of standard resistor values, the precise resistor value may not exist. For 1% standard values, a 243k resistor in series with a 3.01k resistor should be close enough. As discussed in the Application Information section, the final RFB value should be adjusted on the measured output voltage. Example: VZENER(MAX) ≤ 150V – 72V = 78V 8300f 18 LT8300 Applications Information Step 7: Select the EN/UVLO Resistors. Step 8: Ensure minimum load. Determine the amount of hysteresis required and calculate R1 resistor value: The theoretical minimum load can be approximately estimated as: VIN(HYS) = 2.5µA • R1 Example: Choose 2.5V of hysteresis, R1 = 1M Determine the UVLO thresholds and calculate R2 resistor value: VIN(UVLO+) = 1.239V • (R1+ R2) + 2.5µA • R1 R2 ILOAD(MIN) = 300µH • (52mA)2 • 7.5kHz = 0.25mA 2 • 12V Remember to check the minimum load requirement in real application. The minimum load occurs at the point where the output voltage begins to climb up as the converter delivers more energy than what is consumed at the output. The real minimum load for this application is about 0.6mA, 0.5% of 120mA maximum load. In this example, a 20k resistor is selected as the minimum load. Example: Set VIN UVLO rising threshold to 34.5V, R2 = 40.2k VIN(UVLO+) = 34.1V VIN(UVLO–) = 31.6V 8300f 19 LT8300 Typical Applications 5V Micropower Isolated Flyback Converter D1 T1 6:1 VIN 36V TO 72V 2.2µF 1M 8µH 47µF • LT8300 EN/UVLO • 300µH VIN VOUT+ 5V 1mA TO 330mA VOUT– SW 40.2k 316k RFB GND T1: WÜRTH 750312557 D1: DIODES INC. SBR2A30P1 8300 TA02 12V Micropower Isolated Flyback Converter T1 2:1 VIN 36V TO 72V 2.2µF 1M 75µH • LT8300 EN/UVLO • 300µH VIN D1 VOUT+ 12V 0.6mA TO 120mA 10µF VOUT– SW 243k 40.2k RFB GND 8300 TA03 T1: SUMIDA 10396-TO22 D1: DIODES INC. SBR0560S1 8300f 20 LT8300 Typical Applications 24V Micropower Isolated Flyback Converter T1 1:1 VIN 36V TO 72V 2.2µF 1M 300µH • LT8300 EN/UVLO • 300µH VIN D1 VOUT+ 24V 0.3mA TO 60mA 4.7µF VOUT– SW 243k 40.2k RFB GND T1: WÜRTH 750311559 D1: DIODES DFLS 1200-7 8300 TA04 3.3V Micropower Isolated Flyback Converter T1 8:1 VIN 36V TO 72V 2.2µF 1M 6µH • LT8300 EN/UVLO • 400µH VIN D1 VOUT+ 3.3V 2mA TO 440mA 100µF VOUT– SW 40.2k 287k RFB GND 8300 TA05 T1: WÜRTH 750312367 D1: NXP PMEG2020EH 8300f 21 LT8300 Typical Applications VIN to (VIN + 10V) Micropower Converter VOUT+ 10V 50mA 4.7µF VOUT– VIN 15V TO 80V 1µF L1 330µH VIN 1M Z1 LT8300 D1 SW EN/UVLO 102k 118k RFB GND L1: COILTRONICS DR73-331-R D1: DIODES INC. SBR1U150SA Z1: CENTRAL CMDZ12L 8300 TA06 VIN to (VIN – 10V) Micropower Converter VIN 15V TO 80V VOUT+ 10V 100mA 1µF 4.7µF Z1 – VOUT 1M L1 330µH VIN LT8300 EN/UVLO D1 SW 102k 118k RFB GND L1: COILTRONICS DR73-331-R D1: DIODES INC. SBR1U150SA Z1: CENTRAL CMDZ12L 8300 TA07 8300f 22 LT8300 Package Description Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. S5 Package 5-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1635 Rev B) 0.62 MAX 0.95 REF 2.90 BSC (NOTE 4) 1.22 REF 1.4 MIN 3.85 MAX 2.62 REF 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.30 – 0.45 TYP 5 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.20 BSC 0.01 – 0.10 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF 0.09 – 0.20 (NOTE 3) NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 1.90 BSC S5 TSOT-23 0302 REV B 8300f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 23 LT8300 Typical Application 3.3V Isolated Converter (Conforming to DEF-STAN61-5) L1 1:1 VIN 18V TO 32V 1µF 1M VOUT+ 3.3V 0mA TO 20mA OUT LT3009-3.3 150µH Z1 1µF • LT8300 EN/UVLO IN • 150µH VIN D1 SHDN 1µF GND VOUT– SW 42.2k 93.1k RFB GND D1: DIODES INC. SBR0560S1-7 L1: DRQ73-151-R Z1: CENTRAL CMDZ4L7 8300 TA08a Input Current with No Load 400 IVIN (µA) 300 200 100 0 18 20 22 24 26 VIN (V) 28 30 32 8300 TA08b Related Parts PART NUMBER DESCRIPTION COMMENTS LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch, MSOP-16(12) LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No Opto Flyback , MSOP-16 with High Voltage Spacing LT3798 Off-Line Isolated No Opto-Coupler Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components LT3573/LT3574/LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch LT3757/LT3759/LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive LT3957/LT3958 40V/100V Flyback/Boost Converters Monolithic with Integrated 5A/3.3A Switch LTC3803/LTC3803-3/ LTC3803-5 200kHz/300kHz Flyback Controllers in SOT-23 VIN and VOUT Limited by External Components LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited by External Components 8300f 24 Linear Technology Corporation LT 0812 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2012