Richtek C1608JB0J475M 1.5mhz, 600ma, high efficiency pwm step-down dc/dc converter Datasheet

RT8016L
1.5MHz, 600mA, High Efficiency PWM Step-Down DC/DC
Converter
General Description
Features
The RT8016L is a high-efficiency Pulse-Width-Modulated
(PWM) step-down DC/DC converter capable of delivering
up to 600mA output current over a wide input voltage range
from 2.5V to 5.5V. The RT8016L is ideally suited for
portable electronic devices that are powered from 1-cell
Li-ion battery or from other power sources, such as cellular
phones, PDAs and hand-held devices.
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Two operating modes are available including : PWM/low
dropout autoswitch mode and shut-down mode. The
internal synchronous rectifier with low RDS(ON) dramatically
reduces conduction loss at PWM mode. No external
Schottky diode is required in practical application.
The RT8016L enters low dropout mode when normal PWM
can not provide regulated output voltage by continuously
turning on the upper P-MOSFET. The RT8016L enters
Shut-down mode and consumes less than 0.1μA when
the EN pin is pulled low. The RT8016L also offers a fixed
output voltage with a range from 1V to 3.3V with 0.1V per
step or an adjustable output voltage via two external
resistors.
The switching ripple is easily smoothed out by small
package filtering elements due to a fixed operating
frequency of 1.5MHz. Other features include soft-start,
lower internal reference voltage with 2% accuracy, over
temperature protection, and over current protection. The
IC is available in a WQFN-8L 1.6x1.6 (COL) package which
allows small PCB area application.
NC
8
1
7
2
6
5
4
FB/VOUT
GND
LX
NC
WQFN-8L 1.6x1.6 (COL)
DS8016L-02 March 2011
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Applications
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Mobile Phones
Personal Information Appliances
Wireless and DSL Modems
MP3 Players
Portable Instruments
Ordering Information
RT8016LPackage Type
QW : WQFN-8L 1.6x1.6 (COL) (W-Type)
Lead Plating System
G : Green (Halogen Free and Pb Free)
Output Voltage
Default : Adjustable
10 : 1.0V
11 : 1.1V
:
32 : 3.2V
33 : 3.3V
Note :
Richtek products are :
RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020.
(TOP VIEW)
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Pin Configurations
GND
EN
VIN
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Input Range : 2.5V to 5.5V
Adjustable Output Voltage Range : 0.6V to VIN
600mA Output Current
Efficiency up to 95%
No Schottky Diode Required
1.5MHz Fixed-Frequency PWM Operation
RoHS Compliant and Haloger Free
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Suitable for use in SnPb or Pb-free soldering processes.
Marking Information
BSW
BS : Product Code
W : Date Code
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RT8016L
Typical Application Circuit
VIN
2.5V to 5.5V
3
CIN
4.7µF
VIN
LX
5
L
2.2µH
VOUT
RT8016L
2
EN
VOUT
3
VIN
2.5V to 5.5V
COUT
2
10µF
GND
1, 6
LX
5
VOUT
C1
RT8016L
4.7µF
7
VIN
CIN
L
2.2µH
EN
R1
FB
COUT
7
10µF
GND
1, 6
IR2
R2
VOUT = VREF x ⎛⎜ 1 + R1 ⎞⎟
⎝ R2 ⎠
with R2 = 300kΩ to 60kΩ so IR2 = 2μA to 10μA,
Figure 1. Fixed Voltage Regulator
and (R1 x C1) should be in the range between 3x10-6 and
6x10-6 for component selection.
Figure 2. Adjustable Voltage Regulator
Functional Pin Description
Pin No.
1, 6
Pin Name
Pin Function
GND
Ground Pin.
2
EN
Chip Enable (Active High).
3
VIN
Power Input Pin
4, 8
NC
No Internal Connection.
5
LX
Switching Pin for Step-down Converter
7
FB/VOUT
Feedback/Output Voltage Pin.
Function Block Diagram
EN
VIN
RS1
OSC &
Shutdown
Control
Current
Limit
Detector
Slope
Compensation
Current
Sense
FB/VOUT
Error
Amplifier
Control
Logic
PWM
Comparator
UVLO &
Power Good
Detector
LX
Mux
Current
Source
Controller
RC
CCOMP
Driver
Current
Detector
VREF
RS2
GND
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DS8016L-02 March 2011
RT8016L
Absolute Maximum Ratings
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(Note 1)
Supply Input Voltage, VIN ----------------------------------------------------------------------------------------------EN, FB Pin Voltage ------------------------------------------------------------------------------------------------------Power Dissipation, PD @ TA = 25°C
WQFN-8L 1.6x1.6 (COL) -----------------------------------------------------------------------------------------------Package Thermal Resistance (Note 2)
WQFN-8L 1.6x1.6 (COL), θJA ------------------------------------------------------------------------------------------Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------------------Storage Temperature Range -------------------------------------------------------------------------------------------Junction Temperature ----------------------------------------------------------------------------------------------------ESD Susceptibility (Note 3)
HBM (Human Body Mode) ---------------------------------------------------------------------------------------------MM (Machine Mode) ------------------------------------------------------------------------------------------------------
Recommended Operating Conditions
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6.5V
−0.3V to VIN
0.833W
120°C/W
260°C
−65°C to 150°C
150°C
2kV
200V
(Note 4)
Supply Input Voltage, VIN ------------------------------------------------------------------------------------------------ 2.5V to 5.5V
Junction Temperature Range -------------------------------------------------------------------------------------------- −40°C to 125°C
Ambient Temperature Range -------------------------------------------------------------------------------------------- −40°C to 85°C
Electrical Characteristics
(VIN = 3.6V, VOUT = 2.5V, VREF = 0.6V, L = 2.2μH, CIN = 4.7μF, COUT = 10μF, TA = 25°C, IMAX = 600mA unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Quiescent Current
IQ
IOUT = 0mA, VFB = VREF + 5%
--
50
70
μA
Shutdown Current
ISHDN
EN = GND
--
0.1
1
μA
Reference Voltage
VREF
For Adjustable Output Voltage
0.588
0.6
0.612
V
Adjustable Output Range
VOUT
(Note 6)
V REF
--
VIN −
0.2V
V
−3
--
3
%
−3
--
3
%
−50
--
50
nA
VIN = 3.6V
--
0.28
--
VIN = 2.5V
--
0.38
--
VIN = 3.6V
--
0.25
--
VIN = 2.5V
--
0.35
--
0.6
1.5
--
A
Output
Voltage
Accuracy
Fixed
ΔV OUT
Adjustable
ΔV OUT
VIN = (VOUT + ΔV) to 5.5V or
VIN > 2.5V whichever is larger. (Note 5)
VIN = V OUT + ΔV to 5.5V (Note 5)
0A < IOUT < 600mA
FB Input Current
IFB
VFB = VIN
P-MOSFET RON
RDS(ON)_P
IOUT = 200mA
N-MOSFET RON
RDS(ON)_N
IOUT = 200mA
P-Channel Current Limit
ILIM_P
VIN = 2.5V to 5.5 V
Logic High
VEN_H
1.5
--
VIN
V
Logic Low
VEN_L
--
--
0.4
V
UVLO Threshold
VUVLO
--
1.8
--
V
UVLO Hysteresis
ΔV UVLO
--
0.1
--
V
Oscillator Frequency
fOSC
1.2
1.5
1.8
MHz
EN Input
Voltage
VIN = 3.6V, IOUT = 100mA
Ω
Ω
To be continued
DS8016L-02 March 2011
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RT8016L
Parameter
Symbol
Thermal Shutdown Temperature
T SD
Maximum Duty Cycle
DMAX
LX Current Source
Test Conditions
V IN = 3.6V, V LX = 0V or VLX = 3.6V
Min
Typ
Max
Unit
--
160
--
°C
100
--
--
%
1
--
100
μA
Note 1. Stresses listed as the above “Absolute Maximum Ratings” may cause permanent damage to the device. These are for
stress ratings. Functional operation of the device at these or any other conditions beyond those indicated in the
operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended
periods may remain possibility to affect device reliability.
Note 2. θJA is measured in the natural convection at TA = 25°C on a high effective thermal conductivity four-layer test board of
JEDEC 51-7 thermal measurement standard.
Note 3. Devices are ESD sensitive. Handling precaution is recommended.
Note 4. The device is not guaranteed to function outside its operating conditions.
Note 5. ΔV = IOUT x PRDS(ON)
Note 6. Guarantee by design.
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DS8016L-02 March 2011
RT8016L
Typical Operating Characteristics
Output Voltage vs. Input Voltage
Efficiency vs. Output Current
100
1.900
90
1.875
1.850
VIN = 3.3V
VIN =3.6V
VIN =4.2V
70
60
Output Voltage (V)
Efficiency (%)
80
50
40
30
20
1.825
1.800
1.775
1.750
1.725
1.700
1.675
1.650
10
1.625
VOUT = 1.8V
0
1.600
0
0.2
0.4
0.6
0.8
1
2.5
3.0
3.5
Output Current (A)
1.875
1.95
1.850
1.90
1.825
Output Voltage (V)
Output Voltage (V)
2.00
VIN = 3.3V
VIN = 3.6V
VIN = 4.2V
1.775
1.750
1.725
1.700
1.675
5.0
5.5
1.85
1.80
1.75
1.70
1.65
1.60
1.650
1.55
1.625
VIN = 3.6V, VOUT = 1.8V
1.50
1.600
0
0.2
0.4
0.6
0.8
-50
1
-25
0
EN Threshold vs. Input Voltage
1.1
1.1
EN Threshold (V)1
1.2
1.0
Rising
0.8
Falling
0.7
0.6
VIN = 3.6V
0.5
2.5
3.0
3.5
4.0
4.5
Input Voltage (V)
DS8016L-02 March 2011
50
75
100
125
EN Threshold vs. Temperature
1.2
0.9
25
Temperature (°C)
Output Current (A)
EN Threshold (V)1
4.5
Output Voltage vs. Temperature
Output Voltage vs. Output Current
1.900
1.800
4.0
Input Voltage (V)
5.0
5.5
1.0
0.9
Rising
0.8
Falling
0.7
0.6
VIN = 3.6V
0.5
-50
-25
0
25
50
75
100
125
Temperature (°C)
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RT8016L
Frequency vs. Temperature
1.7
1.6
1.6
Frequency(MHz)1
Frequency (MHz)1
Frequency vs. Input Voltage
1.7
1.5
1.4
1.3
1.2
1.5
1.4
1.3
1.2
1.1
1.1
IOUT = 100mA
1.0
2.5
3.0
3.5
4.0
4.5
5.0
VIN = 3.6V, IOUT = 100mA
1.0
-50
5.5
-25
0
Input Voltage threshold vs. Temperature
60
2.0
55
Quiescent Current (μA) 0
Input Voltage (V)
75
100
125
Quiescent Current vs. Temperature
2.1
Rising
1.8
1.7
50
Temperature(°C)
Input Voltage (V)
1.9
25
Falling
1.6
1.5
1.4
50
45
40
35
30
25
1.3
IOUT = 0mA
VIN = 3.6V
20
1.2
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
Temperature (°C)
Temperature (°C)
Current Limit vs. Input Voltage
Power On from EN
100
125
1.8
VIN = 3.6V, VOUT = 1.8V, lOUT = 100mA
1.7
Output Current (A)
1.6
1.5
VEN
(2V/Div)
1.4
1.3
VOUT
(1V/Div)
1.2
1.1
1.0
0.9
VIN = 3.6V
IOUT
(500mA/Div)
0.8
2.5
3.0
3.5
4.0
4.5
5.0
5.5
Time (500μs/Div)
Input Voltage (V)
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DS8016L-02 March 2011
RT8016L
Power Off from EN
Power On from EN
VIN = 3.6V, VOUT = 1.8V, IOUT = 100mA
VEN
(2V/Div)
VEN
(2V/Div)
VOUT
(1V/Div)
VOUT
(1V/Div)
I IN
(500mA/Div)
VIN = 3.6V, VOUT = 1.8V, IOUT = 1A
I IN
(50mA/Div)
Time (500μs/Div)
Time (250μs/Div)
Power Off from EN
Output Ripple Voltage
VIN = 3.6V, VOUT = 1.8V, IOUT = 1A
VEN
(2V/Div)
VLX
(5V/Div)
VOUT
(1V/Div)
VOUT
(10mV/Div)
I IN
(500mA/Div)
VIN = 3.6V, VOUT = 1.8V, IOUT = 1A
Time (50μs/Div)
Time (500ns/Div)
Load Transient Response
Load Transient Response
Output Voltage
(100mV/Div)
Output Voltage
(100mV/Div)
Output Current
(500mA/Div)
Output Current
(500mA/Div)
VIN = 3.6V, IOUT = 50mA to 500mA
Time (100μs/Div)
DS8016L-02 March 2011
VIN = 3.6V, IOUT = 50mA to 1000mA
Time (100μs/Div)
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RT8016L
Application Information
The basic RT8016L application circuit is shown in Typical
Application Circuit. External component selection is
determined by the maximum load current and begins with
the selection of the inductor value and operating frequency
followed by CIN and COUT.
Inductor Selection
For a given input and output voltage, the inductor value
and operating frequency determine the ripple current. The
ripple current ΔIL increases with higher VIN and decreases
with higher inductance:
⎤
⎡V
⎤ ⎡ V
ΔIL = ⎢ OUT ⎥ x ⎢1− OUT ⎥
VIN ⎦
⎣F x L ⎦ ⎣
Having a lower ripple current reduces the ESR losses in
the output capacitors and the output voltage ripple. Highest
efficiency operation is achieved at low frequency with small
ripple current. This, however, requires a large inductor. A
reasonable starting point for selecting the ripple current
is ΔIL = 0.4(IMAX). The largest ripple current occurs at the
highest VIN. To guarantee that the ripple current stays
below a specified maximum, the inductor value should be
chosen according to the following equation :
⎡ VOUT
⎤ ⎡
⎤
V
L=⎢
⎥ x ⎢1− OUT ⎥
⎣⎢ f x ΔIL(MAX) ⎦⎥ ⎢⎣ VIN(MAX) ⎦⎥
Inductor Core Selection
Once the value for L is known, the type of inductor can be
selected. High efficiency converters generally cannot afford
the core loss found in low cost powdered iron cores, forcing
the use of more expensive ferrite or mollypermalloy cores.
Actual core loss is independent of core size for a fixed
inductor value but it is very dependent on the inductance
selected. As the inductance increases, core losses
decrease. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will
increase. Ferrite designs have very low core losses and
are preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard”, which means that
inductance collapses abruptly when the peak design
current is exceeded. This results in an abrupt increase in
inductor ripple current and consequent output voltage ripple.
Do not allow the core to saturate!
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Different core materials and shapes will change the size/
current and price/current relationship of an inductor.
Toroid or shielded pot cores in ferrite or permalloy materials
are small and don't radiate energy but generally cost more
than powdered iron core inductors with similar
characteristics. The choice of which style inductor to use
mainly depends on the price vs. size requirements and
any radiated field/EMI requirements.
CIN and COUT Selection
The input capacitance, C IN, is needed to filter the
trapezoidal current at the source of the top MOSFET. To
prevent large ripple voltage, a low ESR input capacitor
sized for the maximum RMS current should be used. RMS
current is given by :
IRMS = IOUT(MAX)
VOUT
VIN
VIN
−1
VOUT
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT/2. This simple worst case condition is commonly
used for design because even significant deviations do
not offer much relief. Note that ripple current ratings from
capacitor manufacturers are often based on only 2000
hours of life which makes it advisable to further derate the
capacitor, or choose a capacitor rated at a higher
temperature than required. Several capacitors may also
be paralleled to meet size or height requirements in the
design.
The selection of COUT is determined by the effective series
resistance (ESR) that is required to minimize voltage ripple
and load step transients, as well as the amount of bulk
capacitance that is necessary to ensure that the control
loop is stable. Loop stability can be checked by viewing
the load transient response as described in a later section.
The output ripple, ΔVOUT, is determined by :
⎡
1 ⎤
ΔVOUT ≤ ΔIL ⎢ESR +
⎥
8fCOUT ⎦
⎣
The output ripple is highest at maximum input voltage
since ΔIL increases with input voltage. Multiple capacitors
placed in parallel may be needed to meet the ESR and
RMS current handling requirements. Dry tantalum, special
polymer, aluminum electrolytic and ceramic capacitors are
DS8016L-02 March 2011
RT8016L
all available in surface mount packages. Special polymer
capacitors offer very low ESR but have lower capacitance
density than other types. Tantalum capacitors have the
highest capacitance density but it is important to only
use types that have been surge tested for use in switching
power supplies. Aluminum electrolytic capacitors have
significantly higher ESR but can be used in cost sensitive
applications provided that consideration is given to ripple
current ratings and long term reliability. Ceramic capacitors
have excellent low ESR characteristics but can have a
high voltage coefficient and audible piezoelectric effects.
The high Q of ceramic capacitors with trace inductance
can also lead to significant ringing.
Using Ceramic Input and Output Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. However, care must
be taken when these capacitors are used at the input and
output. When a ceramic capacitor is used at the input
and the power is supplied by a wall adapter through long
wires, a load step at the output can induce ringing at the
input, VIN. At best, this ringing can couple to the output
and be mistaken as loop instability. At worst, a sudden
inrush of current through the long wires can potentially
cause a voltage spike at VIN large enough to damage the
part.
Output Voltage Programming
The resistive voltage divider allows the FB pin to sense a
fraction of the output voltage as shown in Figure 3.
VOUT
R1
)
R2
where VREF is the internal reference voltage (0.6V typ.)
VOUT = VREF (1 +
Efficiency Considerations
The efficiency of a switching regulator is equal to the output
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as :
Efficiency = 100% − (L1+ L2+ L3+ ...)
where L1, L2, etc. are the individual losses as a percentage
of input power. Although all dissipative elements in the
circuit produce losses, two main sources usually account
for most of the losses : VIN quiescent current and I2R
losses.
The VIN quiescent current loss dominates the efficiency
loss at very low load currents whereas the I2R loss
dominates the efficiency loss at medium to high load
currents. In a typical efficiency plot, the efficiency curve
at very low load currents can be misleading since the
actual power lost is of no consequence.
1. The VIN quiescent current appears due to two factors
including : the DC bias current as given in the electrical
characteristics and the internal main switch and
synchronous switch gate charge currents. The gate charge
current results from switching the gate capacitance of the
internal power MOSFET switches. Each time the gate is
switched from high to low to high again, a packet of charge
ΔQ moves from VIN to ground. The resulting ΔQ/Δt is the
current out of VIN that is typically larger than the DC bias
current. In continuous mode,
IGATECHG = f (QT+QB)
R1
FB
RT8016L
R2
GND
Figure 3. Setting the Output Voltage
For adjustable voltage mode, the output voltage is set by
an external resistive voltage divider according to the
following equation :
DS8016L-02 March 2011
where QT and QB are the gate charges of the internal top
and bottom switches. Both the DC bias and gate charge
losses are proportional to VIN and thus their effects will
be more pronounced at higher supply voltages.
2. I2R losses are calculated from the resistances of the
internal switches, R SW and external inductor R L. In
continuous mode, the average output current flowing
through inductor L is “chopped” between the main switch
and the synchronous switch. Thus, the series resistance
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RT8016L
looking into the LX pin is a function of both top and bottom
MOSFET RDS(ON) and the duty cycle (DC) as follows :
test board. The maximum power dissipation at TA = 25°C
RSW = RDS(ON)TOP x DC + RDS(ON)BOT x (1−DC)
P D(MAX) = (125°C − 25°C) / (120°C/W) = 0.833W
for WQFN-8L 1.6x1.6 (COL) package
Other losses including CIN and COUT ESR dissipative
losses and inductor core losses generally account for less
than 2% of the total loss.
The maximum power dissipation depends on the operating
ambient temperature for fixed T J (MAX) and thermal
resistance, θJA. For the RT8016L package, the derating
curve in Figure 4 allows the designer to see the effect of
rising ambient temperature on the maximum power
dissipation.
Maximum Power Dissipation (W)1
The RDS(ON) for both the top and bottom MOSFETs can be
obtained from the Typical Operating Characteristics curves.
Thus, to obtain I2R losses, simply add RSW to RL and
multiply the result by the square of the average output
current.
can be calculated by the following formula:
Checking Transient Response
The regulator loop response can be checked by looking
at the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to ΔILOAD (ESR), where ESR is the effective series
resistance of COUT. ΔILOAD also begins to charge or
discharge COUT generating a feedback error signal used
by the regulator to return VOUT to its steady state value.
During this recovery time, VOUT can be monitored for
overshoot or ringing which would indicate a stability
problem.
Thermal Considerations
0.9
Four Layer PCB
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0.0
0
25
50
75
100
125
Ambient Temperature (°C)
(°C)
Figure 4
Layout Considerations
For continuous operation, do not exceed absolute
maximum junction temperature. The maximum power
dissipation depends on the thermal resistance of the IC
package, PCB layout, rate of surrounding airflow, and
difference between junction and ambient temperature. The
maximum power dissipation can be calculated by the
following formula :
Follow the PCB layout guidelines for optimal performance
of the RT8016L.
PD(MAX) = (TJ(MAX) − TA) / θJA
where TJ(MAX) is the maximum junction temperature, TA is
the ambient temperature, and θJA is the junction to ambient
thermal resistance.
For recommended operating condition specifications of
the RT8016L, the maximum junction temperature is 125°C
and TA is the ambient temperature. The junction to ambient
thermal resistance, θJA, is layout dependent. For WQFN8L 1.6x1.6 (COL) packages, the thermal resistance, θJA,
is 120°C/W on a standard JEDEC 51-7 four-layer thermal
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10
`
For the main current paths, keep their traces short and
wide.
`
Put the input capacitor as close as possible to the device
pins (VIN and GND).
`
LX node is with high frequency voltage swing and should
be kept in a small area. Keep analog components away
from LX node to prevent stray capacitive noise pick-up.
`
Connect feedback network behind the output capacitors.
Keep the loop area small. Place the feedback
components near the RT8016L.
` Connect all analog grounds to a common node and then
connect the common node to the power ground behind
the output capacitors.
DS8016L-02 March 2011
RT8016L
The inductor should be placed as close as
possible to the switch pin to minimize the
noise coupling into other circuits.
LX node copper area should be minimized
for reducing EMI.
NC
8
Battery
GND 1
7
FB/VOUT
EN 2
6
GND
VIN 3
5
LX
L
4
COUT
CIN
NC
GND
CIN should be placed as closed as
COUT should be connected
possible to the VIN pin for good filtering.
directly from Pin 7 to ground
Figure 5. Fixed Voltage Regulator layout guide
NC
8
R1
GND
1
7
FB/VOUT
EN
2
6
GND
VIN
3
5
LX
C1
Battery
CIN
R2
VOUT
L
4
COUT
NC
GND
CIN should be placed as closed as
The inductor should be placed as close as
possible to the VIN pin for good filtering. possible to the switch pin to minimize the
noise coupling into other circuits.
LX node copper area should be minimized
COUT should be connected
directly from Pin 7 to ground
for reducing EMI.
Figure 6. Adjustable Voltage Regulator layout guide
DS8016L-02 March 2011
www.richtek.com
11
RT8016L
Supplier
Table 1. Recommended Inductors
Current
DCR
Inductance (µH)
Dimensions(mm)
Rating (mA)
(mΩ)
Series
TAIYO YUDEN
2.2
1480
60
3.00 x 3.00 x 1.50
NR3015
GOTREND
2.2
1500
58
3.85 x 3.85 x 1.80
GTSD32
Sumida
2.2
1500
75
4.50 x 3.20 x 1.55
CDRH2D14
Sumida
4.7
1000
135
4.50 x 3.20 x 1.55
CDRH2D14
TAIYO YUDEN
4.7
1020
120
3.00 x 3.00 x 1.50
NR3015
GOTREND
4.7
1100
146
3.85 x 3.85 x 1.80
GTSD32
Table 2. Recommened Capacitors for CIN and COUT
Supplier
Capacitance (µF)
Package
Part Number
TDK
4.7
0603
C1608JB0J475M
MURATA
4.7
0603
GRM188R60J475KE19
TAIYO YUDEN
4.7
0603
JMK107BJ475RA
TAIYO YUDEN
10
0603
JMK107BJ106MA
TDK
10
0805
C2012JB0J106M
MURATA
10
0805
GRM219R60J106ME19
MURATA
10
0805
GRM219R60J106KE19
TAIYO YUDEN
10
0805
JMK212BJ106RD
www.richtek.com
12
DS8016L-02 March 2011
RT8016L
Outline Dimension
2
1
2
1
DETAIL A
Pin #1 ID and Tie Bar Mark Options
Note : The configuration of the Pin #1 identifier is optional,
but must be located within the zone indicated.
Symbol
Dimensions In Millimeters
Dimensions In Inches
Min
Max
Min
Max
A
0.700
0.800
0.028
0.031
A1
0.000
0.050
0.000
0.002
A3
0.175
0.250
0.007
0.010
b
0.150
0.250
0.006
0.010
D
1.550
1.650
0.061
0.065
E
1.550
1.650
0.061
0.065
e
L
0.400
0.350
0.016
0.450
0.014
0.018
W-Type 8L QFN 1.6x1.6 (COL) Package
Richtek Technology Corporation
Richtek Technology Corporation
Headquarter
Taipei Office (Marketing)
5F, No. 20, Taiyuen Street, Chupei City
5F, No. 95, Minchiuan Road, Hsintien City
Hsinchu, Taiwan, R.O.C.
Taipei County, Taiwan, R.O.C.
Tel: (8863)5526789 Fax: (8863)5526611
Tel: (8862)86672399 Fax: (8862)86672377
Email: [email protected]
Information that is provided by Richtek Technology Corporation is believed to be accurate and reliable. Richtek reserves the right to make any change in circuit
design, specification or other related things if necessary without notice at any time. No third party intellectual property infringement of the applications should be
guaranteed by users when integrating Richtek products into any application. No legal responsibility for any said applications is assumed by Richtek.
DS8016L-02 March 2011
www.richtek.com
13
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