Analogic AAT2514IDE-AA-T1 Dual channel 600ma step-down converter Datasheet

PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
General Description
Features
The AAT2514 SwitchReg™ is a dual channel current mode
PWM DC-DC step-down converter operating at 1.5MHz
constant frequency. The device is ideal for portable equipment requiring two separate power supplies that need
high current up to 600mA. The device operates from
single-cell Lithium-ion batteries while still achieving over
96% efficiency. The AAT2514 also can run at 100% duty
cycle for low dropout operation, extending battery life in
portable systems while light load operation provides very
low output ripple for noise sensitive applications.
•
•
•
•
•
•
•
•
•
•
•
•
•
•
The device has a unique adaptive slope compensation
scheme that makes it possible to operate with a lower
range of inductor values to optimize size and provide
efficient operation. The 1.5MHz switching frequency
minimizes the size of external components while keeping
switching losses low. The AAT2514 can operate from a
2.5V to 5.5V input voltage and can supply up to 600mA
output current for each channel.
VIN Range:2.5V to 5.5V
Up to 600mA Output Current
High Efficiency: Up to 96%
1.5MHz Constant Frequency Operation
100% Duty Cycle Dropout Operation
Low RDS(ON) Internal Switches: 0.35Ω
Current Mode Operation for Excellent Line and Load
Transient Response
Adaptive Slope Compensation
Soft Start
Short-Circuit and Thermal Fault Protection
<1μA Shutdown Current
Power-On Reset Output
Small, Thermally Enhanced TDFN33 -10 Package
-40°C to +85°C Temperature Range
Applications
•
•
•
•
•
The AAT2514 is available in a Pb-free 3 x 3mm 10-lead
TDFN package and operates over the -40°C to +85°C
temperature range.
Cellular Telephones
Digital Still Cameras
PDAs
Portable Media Players
Wireless and DSL Modems
Typical Application
R5
100kΩ
EN1
VIN 2.5V to 5.5V
C1
10μF
EN2
POR
RESET
IN
LX 2
C3
10μF
2514.2008.02.1.1
AAT2514
L2 2.2μH
VOUT2 2.5V
FB2
R4
1MΩ
R3
316kΩ
L1 2.2μH
VOUT1 1.8V
LX1
GND
FB1
R1
316kΩ
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R2
634kΩ
C2
10μF
1
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
Pin Descriptions
Pin #
Symbol
1
FB1
2
EN1
3
4
5
6
8
IN
LX1
GND
N/C
POR
9
EN2
10
FB2
EP
Function
Feedback input for channel 1. Connect FB1 to the center point of an external resistor divider. The feedback threshold voltage is 0.6V.
Channel 1 enable pin. Active high. In shutdown, all functions are disabled drawing <1μA supply current.
Do not leave EN1 floating.
Power supply input pin. Must be closely decoupled to GND with a 2.2μF or greater ceramic capacitor.
Channel 1 switching node pin. Connect the output inductor to this pin.
Ground
No connection
Power-on reset, active low. Open drain. External resistor (100kΩ) is required. Channel 2 enable pin. Active high. In shutdown, all functions are disabled drawing <1μA supply current. Do not leave EN2 floating.
Feedback input for channel 2. Connect FB2 to the center point of an external resistor divider. The feedback threshold voltage is 0.6V.
Exposed paddle. The exposed paddle should be connected to board ground plane and GND. The ground
plane should include a large exposed copper pad under the package for thermal dissipation (see package outline).
Pin Configuration
AAT2514-IDE
TDFN33-10
(Top View)
FB1
1
10
FB2
EN1
2
9
EN2
IN
3
8
POR
LX1
4
7
LX2
GND
5
6
NC
EXPOSED PAD
10-Lead (3mm X 3mm) Plastic Thin DFN
Exposed Pad is PGND
Must be connected to GND.
2
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2514.2008.02.1.1
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
Absolute Maximum Ratings1
Symbol
Description
VIN
VEN1, VEN2
VFB1, VFB2
VLX1, VLX2
VPOR
TA
TJ
TSTORAGE
TLEAD
Input Supply Voltage
EN1, EN2 Voltages
FB1, FB2 Voltages
LX1, LX2 Voltages
POR Voltage
Operating Temperature Range2
Junction Temperature2
Storage Temperature Range
Lead Temperature (Soldering, 10s)
Value
Units
-0.3 to +6.0
-0.3 to VIN + 0.3
-0.3 to VIN + 0.3
-0.3 to VIN + 0.3
-0.3 to +6.0
-40 to +85
+125
-65 to +150
+300
V
V
V
V
V
°C
°C
°C
°C
Value
Units
45
2.2
°C/W
W
Recommended Operating Conditions
Symbol
θJA
PD
Description
Thermal Resistance3
Maximum Power Dissipation at TA = 25°C
1. Absolute Maximum Ratings are those values beyond which the life of a device may be impaired.
2. TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + PD · θJA.
3. Thermal resistance is specified with approximately 1 square inch of 1 oz copper.
2514.2008.02.1.1
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3
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
Electrical Characteristics
VIN = VEN = 3.6V, TA = 25°C, unless otherwise noted.
Symbol
Description
Conditions
Step-Down Converter
VIN
Input Voltage Range
IQ
Input DC Supply Current
VFB
Regulated Feedback
Voltage
IFB
ΔVOUT/
VOUT/ΔVIN
ΔVOUT/
VOUT/ΔIOUT
ILIM
TS
TSD
THYS
FOSC
RDS(ON)
VEN(L)
VEN(H)
IEN
FB Input Bias Current
Output Voltage Line
Regulation
Output Voltage Load
Regulation
Maximum Output Current
Startup Time
Over-Temperature
Shutdown Threshold
Over-Temperature
Shutdown Hysteresis
Oscillator Frequency
P-Channel MOSFET
N-Channel MOSFET
Peak Inductor Current
Enable Threshold Low
Enable Threshold High
EN Input Current
Power-On Reset
Threshold (POR)
Min
Typ
Max
Units
5.5
800
2.0
0.6120
0.6135
0.6150
30
V
500
0.3
0.6000
0.6000
0.6000
2.5
Active Mode, VFB = 0.5V
Shutdown Mode, EN1 = EN2 = 0V, VIN = 4.2V
TA = 25°C, Channel 1 or 2
TA = 0°C ≤ TA ≤ +85°C, Channel 1 or 2
TA = -40°C ≤ TA ≤ +85°C, Channel 1 or 2 (See Note 2)
0.5880
0.5865
0.5850
-30
VIN = 2.5V to 5.5V, IOUT = 10mA
0.11
IOUT = 10mA to 600mA
VIN = 3.0V
From Enable to Output Regulation
VFB = 0.6V
ILX = 300mA
ILX = 300mA
VIN = 3V, VFB = 0.5V; Duty Cycle <35%
0.40
nA
%/V
%/mA
100
mA
μs
140
°C
15
°C
1.5
0.35
0.28
1.20
1.8
0.45
0.45
0.3
1.5
-1.0
VFB Ramping Up
VFB Ramping Down
Power-On Reset Delay
Power-On Reset On-Resistance
V
0.0015
600
1.2
μA
1.0
8.5
-8.5
175
100
MHz
Ω
A
V
V
μA
%
ms
Ω
1. Specifications over the temperature range are guaranteed by design and characterization.
2. The regulated feedback voltage is tested in an internal test mode that connects VFB to the output of the error amplifier.
4
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2514.2008.02.1.1
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
Typical Characteristics
Efficiency vs. Load Current
Efficiency vs. Load Current
(VOUT = 1.8V; TA = 25°C)
100
100
90
90
80
80
70
Efficiency (%)
Efficiency (%)
(VOUT = 2.5V; TA = 25°C)
VIN = 4.2V
60
VIN = 2.7V
50
40
30
VIN = 3.3V
20
VIN = 2.7V
VIN = 3.3V
70
50
40
30
20
10
10
0
0.1
1
10
100
0
1000
0.1
1
Load Current (mA)
Efficiency vs. Load Current
(VOUT = 1.5V; TA = 25°C)
(VOUT = 1.2V; TA = 25°C)
100
90
90
1000
80
Efficiency (%)
Efficiency (%)
100
Efficiency vs. Load Current
100
VIN = 2.7V
70
60
VIN = 4.2V
50
40
VIN = 3.3V
30
70
60
10
10
100
VIN = 3.3V
30
10
1
0
0.1
1000
Load Current (mA)
10
100
ILOAD = 100mA
80
ILOAD = 600mA
60
50
40
30
20
10
3.5
4.0
4.5
5.0
5.5
Output Voltage Error (%)
Load Regulation
(VIN = 3.6V; VOUT = 1.8V; L = 2.2μH)
3.0
1.0
0.8
0.6
0.4
0.2
0
-0.2
-0.4
-0.6
-0.8
-1.0
1
Input Voltage (V)
2514.2008.02.1.1
1000
Load Current (mA)
(VOUT = 1.8V; TA = 25ºC)
90
0
2.5
1
Efficiency vs. Input Voltage
100
70
VIN = 4.2V
40
20
0.1
VIN = 2.7V
50
20
0
10
Load Current (mA)
80
Efficiency (%)
VIN = 4.2V
60
10
100
1000
Load Current (mA)
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PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
Frequency vs. Input Voltage
Frequency vs. Temperature
(VIN = 3.6V; VOUT = 1.8V; ILOAD = 150mA; L = 2.2μH)
(VIN = 3.6V; VOUT = 1.8V; ILOAD = 150mA; L = 2.2μH)
Switching Frequency (MHz)
Switching Frequency (MHz)
Typical Characteristics
1.58
1.56
1.54
1.52
1.50
1.48
1.46
2.5 2.7 2.9 3.1 3.3 3.5 3.7 3.9 4.1 4.3 4.5 4.7 4.9 5.1 5.3 5.5
1.8
1.7
1.6
1.5
1.4
1.3
1.2
-40
-25
-10
5
20
35
50
65
80
Temperature (°C)
Input Voltage (V)
RDS(ON) vs. Input Voltage
VFB vs. Temperature
(TA = 25ºC)
(VIN = 3.6V; VOUT = 1.8V; ILOAD = 0mA)
0.612
0.45
0.609
0.606
0.35
Voltage (V)
RDS(ON) (Ω)
0.40
Main switch
0.30
0.25
0.20
2.0
2.8
3.2
3.6
4.0
4.4
4.8
5.2
0.600
0.597
0.594
Synchronous switch
2.4
0.603
0.591
5.6
6.0
0.588
-45
-30
0
15
30
45
60
75
90
Temperature (°°C)
Input Voltage (V)
Load Transient Response
(Light Load Mode to PWM Mode; L = 2.2µH;
Load Transient Response
(PWM Mode Only; ILOAD = 180mA to 400mA; L = 2.2µH;
CIN = 10µF; COUT = 10µF; VIN = 3.6V; VOUT = 1.8V)
CIN = 10µF; COUT = 10µF; VIN = 3.6V; VOUT = 1.8V)
VSW
(2V/div)
VSW
(2V/div)
VOUT
(200mV/div)
VOUT
(200mV/div)
IOUT
(400mA/div)
IOUT
(500mA/div)
Time (20µs/div)
6
-15
Time (20µs/div)
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2514.2008.02.1.1
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
Functional Block Diagram
Regulator 1
Slope Comp
+
3
IN
4
LX1
8
POR
7
LX2
ISENSE
AMP
-
600mV
FB1
+
+
EA
-
1
650mV
I COMP
R
Q
S _Q
PWM
Logic
+
OVDET
-
NonOverlap
Control
+
I ZERO
COMP
-
EN1
EN2
2
9
Bandgap
Reference
Overtemperature
and Shortcircuit
Protection
FB2 10
OSC
REGULATOR 2 (Same as Regulator 1)
Functional Description
The AAT2514 is a dual high performance 600mA, 1.5MHz
fixed frequency monolithic switch-mode step-down converter which uses current mode architecture with an
adaptive slope compensation scheme. It minimizes external component size and optimizes efficiency over the complete load range. The adaptive slope compensation allows
the device to remain stable over a wider range of inductor
values so that smaller values (1μH to 4.7μH) with associated lower DCR can be used to achieve higher efficiency.
Apart from the small bypass input capacitor, only a small
L-C filter is required at each output. The adjustable outputs can be programmed with external feedback to any
2514.2008.02.1.1
POR
Counter
voltage, ranging from very low output voltages to the
input voltage and by using an internal reference of 0.6V.
The part uses internal MOSFETs for each channel to
achieve high efficiency. At dropout, the converter duty
cycle increases to 100% and the output voltages track
the input voltage minus the low RDS(ON) drop of the
P-channel high-side MOSFETs. The converter efficiency
has been optimized for all load conditions, ranging from
no load to 600mA at VIN = 3V with an input voltage range
from 2.5V to 5.5V. The internal error amplifier and compensation provides excellent transient response, load,
and line regulation. Internal soft start eliminates any
output voltage overshoot when the enable or the input
voltage is applied.
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PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
Current Mode PWM Control
Slope compensated current mode PWM control provides
stable switching and cycle-by-cycle current limit for
excellent load and line response and protection of the
internal main switch (P-channel MOSFET) and synchronous rectifier (N-channel MOSFET). During normal operation, the internal P-channel MOSFET is turned on for a
specified time to ramp the inductor current at each rising
edge of the internal oscillator, and is switched off when
the peak inductor current is above the error voltage. The
current comparator, ICOMP, limits the peak inductor current. When the main switch is off, the synchronous rectifier turns on immediately and stays on until either the
inductor current starts to reverse, as indicated by the
current reversal comparator, IZERO, or the beginning of the
next clock cycle. The OVDET comparator controls output
transient overshoot by turning the main switch off and
keeping it off until the fault is no longer present.
Control Loop
The AAT2514 is a peak current mode step-down converter.
The current through the P-channel MOSFET (high side) is
sensed for current loop control, as well as short circuit and
overload protection. An adaptive slope compensation signal is added to the sensed current to maintain stability for
duty cycles greater than 50%. The peak current mode
loop appears as a voltage-programmed current source in
parallel with the output capacitor. The output of the voltage error amplifier programs the current mode loop for
the necessary peak switch current to force a constant
output voltage for all load and line conditions. Internal
loop compensation terminates the transconductance voltage error amplifier output. For fixed voltage versions, the
error amplifier reference voltage is internally set to program the converter output voltage. For the adjustable
output, the error amplifier reference is fixed at 0.6V.
Enable
The enable pins are active high. When pulled low, the
enable input forces the AAT2514 into a low-power, nonswitching state. The total input current during shutdown
is less than 2μA.
8
Current Limit and
Over-Temperature Protection
For overload conditions, the peak input current is limited. To minimize power dissipation and stresses under
current limit and short-circuit conditions, switching is
terminated after entering current limit for a series of
pulses. Switching is terminated for seven consecutive
clock cycles after a current limit has been sensed for a
series of four consecutive clock cycles. Thermal protection completely disables switching when internal dissipation becomes excessive. The junction over-temperature
threshold is 140°C with 15°C of hysteresis. Once an
over-temperature or over-current fault conditions is
removed, the output voltage automatically recovers.
Dropout Operation
When the input voltage decreases toward the value of
the output voltage, the AAT2514 allows the main switch
to remain on for more than one switching cycle and
increases the duty cycle until it reaches 100%.
The duty cycle D of a step-down converter is defined as:
D = TON · FOSC · 100% ≈
VOUT
· 100%
VIN
Where TON is the main switch on time and FOSC is the
oscillator frequency (1.5MHz).
The output voltage then is the input voltage minus the
voltage drop across the main switch and the inductor. At
low input supply voltage, the RDS(ON) of the P-channel
MOSFET increases and the efficiency of the converter
decreases. Caution must be exercised to ensure the heat
dissipated does not exceed the maximum junction temperature of the IC.
Maximum Load Current
The AAT2514 will operate with an input supply voltage as
low as 2.5V; however, the maximum load current
decreases at lower input due to the large IR drop on the
main switch and synchronous rectifier. The slope compensation signal reduces the peak inductor current as a
function of the duty cycle to prevent sub-harmonic oscillations at duty cycles greater than 50%. Conversely, the
current limit increases as the duty cycle decreases.
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2514.2008.02.1.1
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
Applications Information
Setting the Output Voltage
Figure 1 shows the basic application circuit for the
AAT2514. Resistors R1 and R3 and R2 and R4 program
the output to regulate at a voltage higher than 0.6V. To
limit the bias current required for the external feedback
resistor string while maintaining good noise immunity,
the minimum suggested value for R1 and R3 is 59kΩ.
Although a larger value will further reduce quiescent
current, it will also increase the impedance of the feedback node, making it more sensitive to external noise
and interference. Table 1 summarizes the resistor values
for various output voltages with R1 and R3 set to either
59kΩ for good noise immunity or 316kΩ for reduced no
load input current.
The adjustable feedback resistors, combined with a
external feed forward capacitors (C4 and C5 in Figure 1),
deliver enhanced transient response for extreme pulsed
load applications. The addition of the feed forward
capacitor typically requires a larger output capacitor C2
and C3 for stability. The external resistor sets the output
voltage according to the following equation:
R 2⎞
⎛
VOUT = 0.6 V · ⎜1 +
⎟
R1⎠
⎝
VOUT
(V)
R1, R3 = 59kΩ
R2, R4 (kΩ)
R1, R3 = 316kΩ
R2, R4 (kΩ)
0.8
0.9
1.0
1.1
1.2
1.3
1.4
1.5
1.8
1.85
2.0
2.5
3.3
19.6
29.4
39.2
49.9
59.0
68.1
78.7
88.7
118
124
137
187
267
105
158
210
261
316
365
422
475
634
655
732
1000
1430
Table 1: Resistor Selection for Output
Voltage Setting; Standard 1% Resistor Values
Substituted Closest to the Calculated Values.
Inductor Selection
For most designs, the AAT2514 operates with inductor
values of 1μH to 4.7μH. Low inductance values are
physically smaller, but require faster switching, which
results in some efficiency loss. The inductor value can be
derived from the following equation:
L=
or
⎡
⎣
⎡
⎣
Where ΔIL is inductor ripple current. Large value inductors lower ripple current and small value inductors result
in high ripple currents. Choose inductor ripple current
approximately 35% of the maximum load current 600mA,
or ΔIL = 210mA.
⎛V ⎞
R2 = ⎜ OUT ⎟ - 1 · R1
⎝ VREF ⎠
R5
100kΩ
EN1
VIN 2.5V to 5.5V
C1
10μF
EN2
POR
IN
L2 2.2μH
VOUT2 2.5V
AAT2514
LX2
LX1
FB2
FB1
C5 22pF
C3
10μF
VOUT · (VIN - VOUT)
VIN · ΔIL · fOSC
R4
1MΩ
RESET
L1 2.2μH
VOUT1 1.8V
C4 22pF
R3
316kΩ
GND
R1
316kΩ
R2
634kΩ
C2
10μF
Figure 1: AAT2514 Typical Application Circuit.
2514.2008.02.1.1
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9
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
For output voltages above 2.0V, when light-load efficiency
is important, the minimum recommended inductor size is
2.2μH. For optimum voltage-positioning load transients,
choose an inductor with DC series resistance in the 50mΩ
to 150mΩ range. For higher efficiency at heavy loads
(above 200mA), or minimal load regulation (with some
transient overshoot), the resistance should be kept below
100mΩ. The DC current rating of the inductor should be
at least equal to the maximum load current plus half the
ripple current to prevent core saturation (600mA +
105mA). Table 2 lists some typical surface mount inductors that meet target applications for the AAT2514.
Manufacturer's specifications list both the inductor DC current rating, which is a thermal limitation, and the peak
current rating, which is determined by the saturation characteristics. The inductor should not show any appreciable
saturation under normal load conditions. Some inductors
may meet the peak and average current ratings yet result
in excessive losses due to a high DCR. Always consider the
losses associated with the DCR and its effect on the total
converter efficiency when selecting an inductor. For example, the 2.2μH CR43 series inductor selected from Sumida
has a 71.2mΩ DCR and a 1.75ADC current rating. At full
load, the inductor DC loss is 25mW, which gives a 2.8%
loss in efficiency for a 600mA, 1.5V output.
The AAT2514 step-down converter uses peak current
mode control with a unique adaptive slope compensation
scheme to maintain stability with lower value inductors for
duty cycles greater than 50%. Using lower value inductors
provides better overall efficiency and also makes it easier
to standardize on one inductor for different required out-
CDRH2D11/HP
Sumida CDRH4D18
Toko D312C
VREF
Error Amp
VFB
When below 50% duty cycle, the slope compensation is
0.284A/μs; but when above 50% duty cycle, the slope
compensation is set to 1.136A/μs. The output inductor
value must be selected so the inductor current down slope
meets the internal slope compensation requirements.
Below 50% duty cycle, the slope compensation requirement is:
m=
1.25
= 0.284A/µs
2·L
Therefore:
Slope Compensation
Part
put voltage levels. In order to do this and keep the stepdown converter stable when the duty cycle is greater than
50%, the AAT2514 separates the slope compensation into
2 phases. The required slope compensation is automatically detected by an internal circuit using the feedback
voltage VFB before the error amp comparison to VREF.
L =
0.625
= 2.2µH
m
Above 50% duty cycle,
m=
5
= 1.136A/µs
2·L
L (μH)
Max DCR (mΩ)
Rated DC Current (A)
1.5
2.2
3.3
4.7
1.0
2.2
3.3
4.7
1.5
2.2
3.3
4.7
80
120
173
238
45
75
110
162
120
140
180
240
1.35
1.10
0.9
0.75
1.72
1.32
1.04
0.84
1.29
1.14
0.98
0.79
Size WxLxH (mm)
3.2x3.2x1.2
4.7x4.7x2.0
3.6x3.6x1.2
Table 2: Typical Surface Mount Inductors.
10
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2514.2008.02.1.1
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Therefore:
Dual Channel 600mA Step-Down Converter
2.5
L =
= 2.2µH
m
With these adaptive settings, a 2.2μH inductor can be
used for all output voltages from 0.6V to 5V.
Input Capacitor Selection
The input capacitor reduces the surge current drawn
from the input and switching noise from the device. The
input capacitor impedance at the switching frequency
shall be less than the input source impedance to prevent
high frequency switching current passing to the input.
The calculated value varies with input voltage and is a
maximum when VIN is double the output voltage.
CIN =
V ⎞
VO ⎛
· 1- O
VIN ⎝
VIN ⎠
⎛ VPP
⎞
- ESR · FS
⎝ IO
⎠
VO ⎛
V ⎞
1
· 1 - O = for VIN = 2 · VO
VIN ⎝
VIN ⎠
4
CIN(MIN) =
Output Capacitor Selection
1
⎛ VPP
⎞
- ESR · 4 · FS
⎝ IO
⎠
A low ESR input capacitor sized for maximum RMS current must be used. Ceramic capacitors with X5R or X7R
dielectrics are highly recommended because of their low
ESR and small temperature coefficients. A 22μF ceramic
capacitor for most applications is sufficient. A large value
may be used for improved input voltage filtering.
The maximum input capacitor RMS current is:
IRMS = IO ·
VO ⎛
V ⎞
· 1- O
VIN ⎝
VIN ⎠
The input capacitor RMS ripple current varies with the
input and output voltage and will always be less than or
equal to half of the total DC load current
VO ⎛
V ⎞
· 1- O =
VIN ⎝
VIN ⎠
D · (1 - D) =
IRMS(MAX) =
2514.2008.02.1.1
IO
2
To minimize stray inductance, the capacitor should be
placed as closely as possible to the IC. This keeps the
high frequency content of the input current localized,
minimizing EMI and input voltage ripple. The proper
placement of the input capacitor (C1) can be seen in the
evaluation board layout in Figure 3. A laboratory test setup typically consists of two long wires running from the
bench power supply to the evaluation board input voltage
pins. The inductance of these wires, along with the lowESR ceramic input capacitor, can create a high Q net-work
that may affect converter performance. This problem
often becomes apparent in the form of excessive ringing
in the output voltage during load transients. Errors in the
loop phase and gain measurements can also result. Since
the inductance of a short PCB trace feeding the input voltage is significantly lower than the power leads from the
bench power supply, most applications do not exhibit this
problem. In applications where the input power source
lead inductance cannot be reduced to a level that does
not affect the converter performance, a high ESR tantalum or aluminum electrolytic should be placed in parallel
with the low ESR, ESL bypass ceramic. This dampens the
high Q network and stabilizes the system.
0.52 =
1
2
The function of output capacitance is to store energy to
attempt to maintain a constant voltage. The energy is
stored in the capacitor’s electric field due to the voltage
applied.
The value of output capacitance is generally selected to
limit output voltage ripple to the level required by the
specification. Since the ripple current in the output inductor is usually determined by L, VOUT, and VIN, the series
impedance of the capacitor primarily determines the output voltage ripple. The three elements of the capacitor
that contribute to its impedance (and output voltage
ripple) are equivalent series resistance (ESR), equivalent
series inductance (ESL), and capacitance (C). The output voltage droop due to a load transient is dominated by
the capacitance of the ceramic output capacitor. During a
step increase in load current, the ceramic output capacitor alone supplies the load current until the loop
responds. Within two or three switching cycles, the loop
responds and the inductor current increases to match the
load current demand. The relationship of the output voltage droop during the three switching cycles to the output
capacitance can be estimated by:
COUT =
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3 · ΔILOAD
VDROOP · FS
11
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
In many practical designs, to get the required ESR, a
capacitor with much more capacitance than is needed
must be selected. For both continuous or discontinuous
inductor current mode operation, the ESR of the COUT
needed to limit the ripple to ∆VO, V peak-to-peak is:
ESR ≤
ΔVO
ΔIL
Ripple current flowing through a capacitor’s ESR causes
power dissipation in the capacitor. This power dissipation
causes a temperature increase internal to the capacitor.
Excessive temperature can seriously shorten the expected life of a capacitor. Capacitors have ripple current ratings that are dependent on ambient temperature and
should not be exceeded. The output capacitor ripple current is the inductor current, IL, minus the output current,
IO. The RMS value of the ripple current flowing in the
output capacitance (continuous inductor current mode
operation) is given by:
IRMS = ΔIL ·
There are three types of losses associated with the
AAT2514 step-down converter: switching losses, conduction losses, and quiescent current losses. Conduction
losses are associated with the RDS(ON) characteristics of the
power output switching devices. Switching losses are
dominated by the gate charge of the power output switching devices. At full load, assuming continuous conduction
mode(CCM), a simplified form of the losses is given by:
PTOTAL =
IO2 · (RDSON(HS) · VO + RDSON(LS) · [VIN - VO])
VIN
+ (tsw · F · IO + IQ) · VIN
IQ is the step-down converter quiescent current. The
term tsw is used to estimate the full load step-down converter switching losses.
For the condition where the step-down converter is in
dropout at 100% duty cycle, the total device dissipation
reduces to:
3
ΔIL · 0.289
6
ESL can be a problem by causing ringing in the low
megahertz region but can be controlled by choosing low
ESL capacitors, limiting lead length (PCB and capacitor),
and replacing one large device with several smaller ones
connected in parallel.
In conclusion, in order to meet the requirement of output voltage ripple small and regulation loop stability,
ceramic capacitors with X5R or X7R dielectrics are recommended due to their low ESR and high ripple current
ratings. The output ripple VOUT is determined by:
ΔVOUT ≤
Thermal Calculations
PTOTAL = IO2 · RDSON(HS) + IQ · VIN
Since RDS(ON), quiescent current, and switching losses all
vary with input voltage, the total losses should be investigated over the complete input voltage range. Given the
total losses, the maximum junction temperature can be
derived from the θJA for the MSOP-10 or DFN-10 packages, which is 45°C/W.
TJ(MAX) = PTOTAL · ΘJA + TAMB
⎞
1
VOUT · (VIN - VOUT) ⎛
· ESR +
⎝
8 · fOSC · C3⎠
VIN · fOSC · L
A 10μF ceramic capacitor can satisfy most applications.
Manufacturer
Part Number
Value
Voltage (V)
Temp. Co.
Case
Murata
Murata
Murata
GRM219R60J106KE19
GRM21BR60J226ME39
GRM1551X1E220JZ01B
10μF
22μF
22pF
6.3
6.3
25
X5R
X5R
JIS
0805
0805
0402
Table 3: Typical Surface Mount Capacitors.
12
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2514.2008.02.1.1
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
Layout Guidance
4.
Figure 1 is the schematic for a typical application. When
laying out the PC board, the following layout guidelines
should be followed to ensure proper operation of the
AAT2514:
5.
1.
2.
3.
Exposed pad must be reliably soldered to GND. The
exposed thermal pad should be connected to the
board ground plane and GND. The ground plane
should include a large exposed copper pad under the
package for thermal dissipation.
The power traces, including the GND trace, the LX1/
LX2 traces, and the VIN trace should be kept short,
direct and wide to allow large current flow. The L1/2
connection to the LX1/2 pins should be as short as
possible. Use several VIA pads when routing between
layers.
The input capacitor (C1) should connect as closely as
possible to IN and GND to get good power filtering.
6.
7.
Keep the switching nodes, LX1/LX2, away from the
sensitive FB1/FB2 nodes.
The feedback traces or FB pins should be separate
from any power trace and connected as closely as
possible to the load point. Sensing along a highcurrent load trace will degrade DC load regulation.
The feedback resistors should be placed as close as
possible to the FB pins to minimize the length of the
high impedance feedback trace.
The output capacitors C2/C3 and L1/L2 should be
connected as close as possible and there should not
be any signal lines under the inductor.
The resistance of the trace from the load return to
GND should be kept to a minimum. This will help to
minimize any error in DC regulation due to differences in the potential of the internal signal ground
and the power ground.
Figure 2 shows an example of a layout with 4 layers. The
2nd and 3rd layers are Internal GND Plane.
a: Top Layer
b: Bottom Layer
Figure 2: AAT2514 Typical Application Circuit Layout.
2514.2008.02.1.1
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13
PRODUCT DATASHEET
AAT2514
SwitchRegTM
Dual Channel 600mA Step-Down Converter
Ordering Information
Output Voltage1
Package
Marking2
Part Number (Tape & Reel)3
Adj. 0.6V to VIN
TDFN33-10
ZBXYY
AAT2514IDE-AA-T1
All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” means semiconductor
products that are in compliance with current RoHS standards, including the requirement that lead not exceed
0.1% by weight in homogeneous materials. For more information, please visit our website at
http://www.analogictech.com/about/quality.aspx.
Package Information4
TDFN33-10
Pin 1 dot by marking
0.500 BSC
1.70 ± 0.05
3.00 ± 0.05
0.23 ± 0.05
Pin 1 identification
R0.200
0.40 ± 0.05
3.00 ± 0.05
2.40 ± 0.05
Top View
0.05 ± 0.05
0.203 REF
0.75 ± 0.05
Bottom View
Side View
All dimensions in millimeters.
1.
2.
3.
4.
Please contact Sales for other voltage options.
XYY = assembly and date code.
Sample stock is generally held on part numbers listed in BOLD.
The leadless package family, which includes QFN, TQFN, DFN, TDFN and STDFN, has exposed copper (unplated) at the end of the lead terminals due to the manufacturing
process. A solder fillet at the exposed copper edge cannot be guaranteed and is not required to ensure a proper bottom solder connection.
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Phone (408) 737-4600
Fax (408) 737-4611
© Advanced Analogic Technologies, Inc.
AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech product. No circuit patent licenses, copyrights, mask work rights, or other intellectual
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relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right. In order to minimize risks associated with the customer’s applications, adequate
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14
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2514.2008.02.1.1