AD AD7677 16-bit 1 msps sar unipolar adc with ref Datasheet

PRELIMINARY TECHNICAL DATA
a
16-Bit 1 MSPS SAR Unipolar ADC with Ref
Preliminary Technical Data
AD7667*
FEATURES
Throughput:
1 MSPS (Warp Mode)
800 kSPS (Normal Mode)
INL: ±2.5 LSB Max (±0.0038% of Full Scale)
16 Bits Resolution with No Missing Codes
Analog Input Voltage Range: 0 V to 2.5 V
No Pipeline Delay
Parallel and Serial 5 V/3 V Interface
SPITM/QSPITM/MICROWIRETM/DSP Compatible
Single 5 V Supply Operation
Power Dissipation
112 mW Typ without REF, 122 mW Typ with REF
15 W @ 100 SPS
Power-Down Mode: 7 W Max
Package: 48-Lead Quad Flat Pack (LQFP);
48-Lead Chip Scale Package (LFCSP);
Pin-to-Pin Compatible with PulSAR ADCs
APPLICATIONS
Data Acquisition
Instrumentation
Digital Signal Processing
Spectrum Analysis
Medical Instruments
Battery-Powered Systems
Process Control
FUNCTIONAL
REFBUFIN
BLOCK
DIAGRAM
REF REFGND
AGND
AVDD
DVDD DGND
OVDD
AD7667
2.5 V REF
IN
16
SWITCHED
CAP DAC
INGND
OGND
SERIAL
PORT
DATA[15:0]
BUSY
PARALLEL
INTERFACE
PDREF
PDBUF
CLOCK
CS
PD
CONTROL LOGIC AND
CALIBRATION CIRCUITRY
RESET
RD
SER/PAR
OB/2C
BYTESWAP
WARP
IMPULSE
CNVST
PulSAR Selection
Type / kSPS
Pseudo
Differential
100 - 250
AD7651
AD7660/61
500 - 570
AD7650/52
AD7664/66
1000
AD7653
AD7667
True Bipolar
AD7663
AD7665
AD7671
True
Differential
AD7675
AD7676
AD7677
PRODUCT HIGHLIGHTS
GENERAL DESCRIPTION
The AD7667 is a 16-bit, 1 MSPS, charge redistribution SAR,
analog-to-digital converter that operates from a single 5 V
power supply. The part contains a high-speed 16-bit sampling
ADC, an internal conversion clock, internal reference, error
correction circuits, and both serial and parallel system interface ports.
It features a very high sampling rate mode (Warp) and, for
asynchronous conversion rate applications, a fast mode
(Normal) and, for low power applications, a reduced power
mode (Impulse) where the power is scaled with the throughput.
It is fabricated using Analog Devices’ high-performance, 0.6 micron
CMOS process, with correspondingly low cost and is available in a
48-lead LQFP and a tiny 48-lead LFCSP with operation specified from –40°C to +85°C.
*Patent pending.
SPI and QSPI are trademarks of Motorola Inc.
MICROWIRE ia a trademark of National Semiconductor Corporation
1. Fast Throughput
The AD7667 is a 1 MSPS, charge redistribution, 16-bit
SAR ADC with internal error correction circuitry.
2. Internal Reference
The AD7667 has an internal reference and allows for an
external reference to be used.
3. Superior INL
The AD7667 has a maximum integral nonlinearity of 2.5
LSB with no missing 16-bit code.
4. Single-Supply Operation
The AD7667 operates from a single 5 V supply and dissipates
a typical of 112 mW. In impulse mode, its power dissipation decreases with the throughput. It consumes 7 µW
maximum when in power-down.
5. Serial or Parallel Interface
Versatile parallel or 2-wire serial interface arrangement
compatible with both 3 V or 5 V logic.
REV. PrA
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
Fax: 781/326-8703
www.analog.com
© Analog Devices, Inc., 2002
PRELIMINARY TECHNICAL DATA
AD7667 –SPECIFICATIONS (–40C to +85C, AVDD = DVDD = 5 V, OVDD = 2.7 V to 5.25 V, unless otherwise noted.)
Parameter
RESOLUTION
ANALOG INPUT
Voltage Range
Operating Input Voltage
Analog Input CMRR
Input Current
Input Impedance
THROUGHPUT SPEED
Complete Cycle
Throughput Rate
Time Between Conversions
Complete Cycle
Throughput Rate
Complete Cycle
Throughput Rate
DC ACCURACY
Integral Linearity Error
No Missing Codes
Transition Noise
Full-Scale Error2
Unipolar Zero Error2
Power Supply Sensitivity
AC ACCURACY
Signal-to-Noise
Spurious Free Dynamic Range
Total Harmonic Distortion
Signal-to-(Noise+Distortion)
–3 dB Input Bandwidth
SAMPLING DYNAMICS
Aperture Delay
Aperture Jitter
Transient Response
REFERENCE
Internal Reference Voltage
Internal Reference Source Current
Internal Reference Temp Drift
Internal Reference Temp Drift
Turn-on Settling Time
External Reference Voltage Range
External Reference Current Drain
Temperature Pin
Voltage Output @ 25C
Conditions
Min
Max
16
VIN – VINGND
VIN
VINGND
fIN = 10 kHz
1 MSPS Throughput
VREF
+3
+0.5
V
V
V
dB
µA
1
1000
1
1.25
800
1.5
666
µs
kSPS
ms
µs
kSPS
µs
kSPS
+2.5
LSB 1
Bits
LSB
% of FSR
LSB
LSB
TBD
11
See Analog Input Section
In Warp Mode
In Warp Mode
In Warp Mode
In Normal Mode
In Normal Mode
In Impulse Mode
In Impulse Mode
Unit
Bits
0
–0.1
–0.1
1
0
0
–2.5
16
0.7
REF = 2.5 V
±TBD
±TBD
AVDD = 5 V ± 5%
±TBD
±TBD
fIN = 100 kHz
fIN = 100 kHz
fIN = 45 kHz
fIN = 100 kHz
fIN = 100 kHz
–60 dB Input, fIN = 100 kHz
90
100
-100
-100
90
30
TBD
dB
dB
dB
dB
dB
dB
MHz
2
5
ns
ps rms
ns
Full-Scale Step
250
@ 25C
TBD
–40C to +85C
0C to +70C
2.5
TBD
TBD
TBD
TBD
2.3
1 MSPS Throughput
TBD
2.5
TBD
Output Resistance
DIGITAL INPUTS
Logic Levels
VIL
VIH
IIL
IIH
DIGITAL OUTPUTS
Data Format
Pipeline Delay
AVDD – 1.85
k
+0.8
OVDD + 0.3
+1
+1
Parallel or Serial 16-Bits
Conversion Results Available
Immediately after
Completed Conversion
0.4
ISINK = 1.6 mA
ISOURCE = –500 µA
OVDD – 0.6
POWER SUPPLIES
Specified Performance
AVDD
DVDD
OVDD
4.75
4.75
2.7
–2–
5
5
V
µA
mV
mV/C
4.3
–0.3
2.0
–1
–1
V
µA
ppm/C
ppm/C
313
1
Temperature Sensitivity
VOL
VOH
Typ
5.25
5.25
5.25 9
V
V
µA
µA
V
V
V
V
V
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
Parameter
Operating Current4
AVDD 5
DVDD 5
OVDD 5
Power Dissipation5 without REF
Power Dissipation5 with REF
TEMPERATURE RANGE8
Specified Performance
Conditions
1 MSPS Throughput
Min
Typ
Max
TBD
TBD
TBD
112
15
1 MSPS Throughput
100 SPS Throughput6
In Power-Down Mode7
1 MSPS Throughput
100 SPS Throughput6
In Power-Down Mode7
TBD
mA
mA
µA
mW
µW
µW
mW
mW
µW
+85
°C
TBD
122
10.015
TMIN to TMAX
–40
Unit
NOTES
1
LSB means Least Significant Bit. With the 0 V to 2.5 V input range, one LSB is 38.15 µV.
2
See Definition of Specifications section. These specifications do not include the error contribution from the external reference.
3
All specifications in dB are referred to a full-scale input FS. Tested with an input signal at 0.5 dB below full-scale unless otherwise specified.
4
In warp mode.
5
Tested in parallel reading mode using external reference.
6
In impulse mode with external REF.
7
With all digital inputs forced to DVDD or DGND respectively.
8
Contact factory for extended temperature range.
9
The max should be the minimum of 5.25V and DVDD+0.3 V.
Specifications subject to change without notice.
TIMING SPECIFICATIONS (–40C to +85C, AVDD = DVDD = 5 V, OVDD = 2.7 V to 5.25 V, unless otherwise noted.)
REFER TO FIGURES 11 AND 12
Convert Pulsewidth
Time Between Conversions
(Warp Mode/Normal Mode/Impulse Mode)
CNVST LOW to BUSY HIGH Delay
BUSY HIGH All Modes Except in
Master Serial Read After Convert Mode
(Warp Mode/Normal Mode/Impulse Mode)
Aperture Delay
End of Conversion to BUSY LOW Delay
Conversion Time
(Warp Mode/Normal Mode/Impulse Mode)
Acquisition Time
RESET Pulsewidth
REFER TO FIGURES 13, 14, AND 15 (Parallel Interface Modes)
CNVST LOW to DATA Valid Delay
(Warp Mode/Normal Mode/Impulse Mode)
DATA Valid to BUSY LOW Delay
Bus Access Request to DATA Valid
Bus Relinquish Time
REFER TO FIGURES 16 AND 17 (Master Serial Interface Modes)2
CS LOW to SYNC Valid Delay
CS LOW to Internal SCLK Valid Delay2
CS LOW to SDOUT Delay
CNVST LOW to SYNC Delay
(Warp Mode/Normal Mode/Impulse Mode)
SYNC Asserted to SCLK First Edge Delay
Internal SCLK Period3
Internal SCLK HIGH3
Internal SCLK LOW3
SDOUT Valid Setup Time3
SDOUT Valid Hold Time3
SCLK Last Edge to SYNC Delay3
CS HIGH to SYNC HI-Z
CS HIGH to Internal SCLK HI-Z
CS HIGH to SDOUT HI-Z
BUSY HIGH in Master Serial Read after Convert3
(Warp Mode/Normal Mode/Impulse Mode)
CNVST LOW to SYNC Asserted Delay
(Warp Mode/Normal Mode/Impulse Mode)
SYNC Deasserted to BUSY LOW Delay
REV. PrA
Symbol
Min
t1
t2
5
1/1.25/1.5
Typ
t3
t4
Max
Unit
Note 1
ns
µs
30
0.75/1/1.25
ns
µs
0.75/1/1.25
ns
ns
µs
2
t5
t6
t7
10
t8
t9
250
10
ns
ns
t10
t11
t12
t13
0.75/1/1.25
µs
40
15
ns
ns
ns
45
5
t14
t15
t16
t17
10
10
10
25/275/525
t18
t19
t20
t21
t22
t23
t24
t25
t26
t27
t28
3
25
12
7
4
2
3
40
10
10
10
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
See Table I
ns
ns
ns
µs
t29
0.75/1/1.25
µs
t30
25
ns
–3–
PRELIMINARY TECHNICAL DATA
AD7667
REFER TO FIGURES 18 AND 20 (Slave Serial Interface Modes)2
External SCLK Setup Time
External SCLK Active Edge to SDOUT Delay
SDIN Setup Time
SDIN Hold Time
External SCLK Period
External SCLK HIGH
External SCLK LOW
Symbol
Min
t31
t32
t33
t34
t35
t36
t37
5
3
5
5
25
10
10
Typ
Max
Unit
ns
ns
ns
ns
ns
ns
ns
18
NOTES
1
In warp mode only, the maximum time between conversions is 1 ms; otherwise, there is no required maximum time.
2
In serial interface modes, the SYNC, SCLK, and SDOUT timings are defined with a maximum load CL of 10 pF; otherwise, the load is 60 pF maximum.
3
In serial master read during convert mode. See Table I for serial master read after convert mode.
Specifications subject to change without notice.
Table I. Serial clock timings in Master Read after Convert
DIVSCLK[1]
DIVSCLK[0]
SYNC to SCLK First Edge Delay Minimum
Internal SCLK Period minimum
Internal SCLK Period typical
Internal SCLK HIGH Minimum
Internal SCLK LOW Minimum
SDOUT Valid Setup Time Minimum
SDOUT Valid Hold Time Minimum
SCLK Last Edge to SYNC Delay Minimum
Busy High Width Maximum (Warp)
Busy High Width Maximum (Normal)
Busy High Width Maximum (Impulse)
t18
t19
t19
t20
t21
t22
t23
t24
t24
t24
t24
–4–
0
0
3
25
40
12
7
4
2
3
1.5
1.75
2
0
1
17
50
70
22
21
18
4
60
2
2.25
2.5
1
0
17
100
140
50
49
18
30
140
3
3.25
3.5
1
1
17
200
280
100
99
18
89
300
5.25
5.55
5.75
unit
ns
ns
ns
ns
ns
ns
ns
ns
µs
µs
µs
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
ABSOLUTE MAXIMUM RATINGS*
1.6mA
IN2, TEMP2,REF, REFBUFIN, INGND, REFGND to AGND
. . . . . . . . . . . . . . . . . . . . . . . AVDD + 0.3 V to AGND – 0.3 V
Ground Voltage Differences
AGND, DGND, OGND . . . . . . . . . . . . . . . . . . . . . ±0.3 V
Supply Voltages
AVDD, DVDD, OVDD . . . . . . . . . . . . . . . -0.3V to +7 V
AVDD to DVDD, AVDD to OVDD . . . . . . . . . . . . . ±7 V
DVDD to OVDD . . . . . . . . . . . . . . . . . . . . -0.3V to +7 V
Digital Inputs . . . . . . . . . . . . . . . . –0.3 V to DVDD + 3.0 V
Internal Power Dissipation3 . . . . . . . . . . . . . . . . . . . 700 mW
Internal Power Dissipation4 . . . . . . . . . . . . . . . . . . . . . 2.5 W
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . 150°C
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
Lead Temperature Range
(Soldering 10 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . . 300°C
TO OUTPUT
PIN
IOL
1.4V
CL
60pF*
500A
IOH
*IN SERIAL INTERFACE MODES, THE SYNC, SCLK, AND
SDOUT TIMINGS ARE DEFINED WITH A MAXIMUM LOAD
CL OF 10pF; OTHERWISE, THE LOAD IS 60pF MAXIMUM.
Figure 1. Load Circuit for Digital Interface Timing,
SDOUT, SYNC, SCLK Outputs, CL = 10 pF
2V
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those listed in the operational
sections of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
See Analog Input section.
3
Specification is for the device in free air:
48-Lead LQFP; θJA = 91°C/W, θJC = 30°C/W
4
Specification is for the device in free air:
48-Lead LFCSP; θJA = 26°C/W
0.8V
t DELAY
t DELAY
2V
0.8V
2V
0.8V
Figure 2. Voltage Reference Levels for Timing
ORDERING GUIDE
Model
AD7667AST
AD7667ASTRL
AD7667ACP
AD7667ACPRL
EVAL-AD7667CB1
EVAL-CONTROL BRD22
Temperature
Range
Package Description
Package Option
–40°C
–40°C
–40°C
–40°C
Quad Flatpack (LQFP)
Quad Flatpack (LQFP)
Chip Scale (LFCSP)
Chip Scale (LFCSP)
ST-48
ST-48
CP-48
CP-48
Evaluation Board
Controller Board
to
to
to
to
+85°C
+85°C
+85°C
+85°C
NOTES
1
This board can be used as a standalone evaluation board or in conjunction with the EVAL-CONTROL BRD2 for evaluation/demonstration purposes.
2
This board allows a PC to control and communicate with all Analog Devices evaluation boards ending in the CB designators.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the AD7667 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.
REV. PrA
–5–
WARNING!
ESD SENSITIVE DEVICE
PRELIMINARY TECHNICAL DATA
AD7667
INGND
REFGND
REF
PDBUF
PDREF
REFBUFIN
TEMP
NC
IN
NC
NC
NC
PIN CONFIGURATION
48-Lead LQFP
(ST-48)
48 47 46 45 44 43 42 41 40 39 38 37
AGND 1
AVDD 2
NC
BYTESWAP
OB/2C
WARP
IMPULSE
SER/PAR
3
AGND
CNVST
34 PD
4
33
5
32
D0
D1
D2/SCLK0
D3/SCLK1
9
36
PIN 1
IDENTIFIER
35
6
AD7667
7
TOP VIEW
(Not to Scale)
8
31
30
RD
DGND
29
BUSY
D15
27 D14
26 D13
25 D12
28
10
11
12
D9/SCLK
D10/SYNC
D11/RDERROR
13 14 15 16 17 18 19 20 21 22 23 24
D4/EXT/INT
D5/INVSYNC
D6/INVSCLK
D7/RDC/SDIN
OGND
OVDD
DVDD
DGND
D8/SDOUT
NC = NO CONNECT
RESET
CS
PIN FUNCTION DESCRIPTIONS
Pin No.
Mnemonic
Type
Description
1
2
3, 40–42,
44
4
AGND
AVDD
NC
P
P
Analog Power Ground Pin
Input Analog Power Pins. Nominally 5 V.
No Connect
BYTESWAP
DI
5
OB/2C
DI
6
WARP
DI
7
IMPULSE
DI
8
SER/PAR
DI
9,10
DATA[0:1]
DI
11,12
DATA[2:3]or
DIVSCLK[0:1]
Parallel Mode Selection (8/16 bit). When LOW, the LSB is output on D[7:0] and the
MSB is output on D[15:8]. When HIGH, the LSB is output on D[15:8] and the MSB
is output on D[7:0].
Straight Binary/Binary Two’s Complement. When OB/2C is HIGH, the digital output is
straight binary; when LOW, the MSB is inverted resulting in a two’s complement output
from its internal shift register.
Mode Selection. When HIGH and IMPULSE LOW, this input selects the fastest mode,
the maximum throughput is achievable, and a minimum conversion rate must be applied
in order to guarantee full specified accuracy. When LOW, full accuracy is maintained
independent of the minimum conversion rate.
Mode Selection. When HIGH and WARP LOW, this input selects a reduced power mode.
In this mode, the power dissipation is approximately proportional to the sampling rate.
Serial/Parallel Selection Input. When LOW, the parallel port is selected; when HIGH, the
serial interface mode is selected and some bits of the DATA bus are used as a serial port.
Bit 0 and Bit 1 of the Parallel Port Data Output Bus. When SER/PAR is HIGH, these
outputs are in high impedance.
When SER/PAR is LOW, these outputs are used as Bit 2 and Bit 3 of the Parallel Port
Data Output Bus. When SER/PAR is HIGH, EXT/INT is LOW, and RDC/SDIN is
LOW, which is serial master read after convert, these inputs, part of the serial port, are
used to slow down if desired the internal serial clock which clocks the data output. In
other serial moes, these pins are not used
When SER/PAR is LOW, this output is used as Bit 4 of the Parallel Port Data Output
Bus.
When SER/PAR is HIGH, this input, part of the serial port, is used as a digital select input
for choosing the internal or an external data clock. With EXT/INT tied LOW, the internal
clock is selected on SCLK output. With EXT/INT set to a logic HIGH, output data is syn
chronized to an external clock signal connected to the SCLK input.
When SER/PAR is LOW, this output is used as Bit 5 of the Parallel Port Data Output
Bus.
When SER/PAR is HIGH, this input, part of the serial port, is used to select the active
13
DATA[4]
DI/O
DI/O
or EXT/INT
14
DATA[5]
or INVSYNC
DI/O
–6–
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
Pin No.
15
Mnemonic
DATA[6]
Type
DI/O
or INVSCLK
16
DATA[7]
DI/O
or RDC/SDIN
17
18
OGND
OVDD
P
P
19
20
21
DVDD
DGND
DATA[8]
or SDOUT
P
P
DO
22
DATA[9]
or SCLK
DI/O
23
DATA[10]
DO
or SYNC
24
DATA[11]
DO
or RDERROR
25–28
DATA[12:15] DO
29
BUSY
DO
30
31
DGND
RD
P
DI
32
CS
DI
REV. PrA
Description
state of the SYNC signal. It is active in both master and slave mode. When LOW, SYNC
is active HIGH. When HIGH, SYNC is active LOW.
When SER/PAR is LOW, this output is used as Bit 6 of the Parallel Port Data Output
Bus.
When SER/PAR is HIGH, this input, part of the serial port, is used to invert the SCLK signal.
It is active in both master and slave mode.
When SER/PAR is LOW, this output is used as Bit 7 of the Parallel Port Data Output
Bus.
When SER/PAR is HIGH, this input, part of the serial port, is used as either an external
data input or a read mode selection input depending on the state of EXT/INT.
When EXT/INT is HIGH, RDC/SDIN could be used as a data input to daisy chain the conversion results from two or more ADCs onto a single SDOUT line. The digital data level on
SDIN is output on DATA with a delay of 16 SCLK periods after the initiation of the read
sequence.
When EXT/INT is LOW, RDC/SDIN is used to select the read mode. When RDC/SDIN
is HIGH, the data is output on SDOUT during conversion. When RDC/SDIN is LOW,
the data can be output on SDOUT only when the conversion is complete.
Input/Output Interface Digital Power Ground
Input/Output Interface Digital Power. Nominally at the same supply than the supply of
the host interface (5 V or 3 V).
Digital Power. Nominally at 5 V.
Digital Power Ground
When SER/PAR is LOW, this output is used as Bit 8 of the Parallel Port Data Output Bus.
When SER/PAR is HIGH, this output, part of the serial port, is used as a serial data out
put synchronized to SCLK. Conversion results are stored in an on-chip register. The
AD7667 provides the conversion result, MSB first, from its internal shift register. The
DATA format is determined by the logic level of OB/2C. In serial mode, when EXT/INT
is LOW, SDOUT is valid on both edges of SCLK.
In serial mode, when EXT/INT is HIGH:
If INVSCLK is LOW, SDOUT is updated on SCLK rising edge and valid on the next
falling edge.
If INVSCLK is HIGH, SDOUT is updated on SCLK falling edge and valid on the next rising
edge.
When SER/PAR is LOW, this output is used as the Bit 9 of the Parallel Port Data
Output Bus.
When SER/PAR is HIGH, this pin, part of the serial port, is used as a serial data clock
input or output, dependent upon the logic state of the EXT/INT pin. The active edge
where the data SDOUT is updated depends upon the logic state of the INVSCLK pin.
When SER/PAR is LOW, this output is used as the Bit 10 of the Parallel Port Data Output
Bus.
When SER/PAR is HIGH, this output, part of the serial port, is used as a digital output
frame synchronization for use with the internal data clock (EXT/INT = Logic LOW).
When a read sequence is initiated and INVSYNC is LOW, SYNC is driven HIGH and
remains HIGH while SDOUT output is valid. When a read sequence is initiated and
INVSYNC is HIGH, SYNC is driven LOW and remains LOW while SDOUT output is
valid.
When SER/PAR is LOW, this output is used as the Bit 11 of the Parallel Port Data Output
Bus.
When SER/PAR is HIGH and EXT/INT is HIGH, this output, part of the serial port,
is used as a incomplete read error flag. In slave mode, when a data read is started and
not complete when the following conversion is complete, the current data is lost and
RDERROR is pulsed high.
Bit 12 to Bit 15 of the Parallel Port Data output bus. These pins are always outputs regard
less of the state of SER/PAR.
Busy Output. Transitions HIGH when a conversion is started, and remains HIGH until
the conversion is complete and the data is latched into the on-chip shift register. The fall
ing edge of BUSY could be used as a data ready clock signal.
Must Be Tied to Digital Ground
Read Data. When CS and RD are both LOW, the interface parallel or serial output bus is
enabled.
Chip Select. When CS and RD are both LOW, the interface parallel or serial output bus is
enabled. CS is also used to gate the external clock.
–7–
PRELIMINARY TECHNICAL DATA
AD7667
Pin No.
33
Mnemonic
RESET
Type
DI
34
PD
DI
35
CNVST
DI
36
37
38
39
43
45
46
47
AGND
REF
REFGND
INGND
IN
TEMP
REFBUFIN
PDREF
P
AI
AI
AI
AI
AO
AI/O
DI
48
PDBUF
DI
Description
Reset Input. When set to a logic HIGH, reset the AD7667. Current conversion if any is
aborted. If not used, this pin could be tied to DGND.
Power-Down Input. When set to a logic HIGH, power consumption is reduced and conver
sions are inhibited after the current one is completed.
Start Conversion. A falling edge on CNVST puts the internal sample/hold into the hold state
and initiates a conversion. In impulse mode (IMPULSE HIGH and WARP LOW), if
CNVST is held low when the acquisition phase (t8) is complete, the internal sample/hold
is put into the hold state and a conversion is immediately started.
Must Be Tied to Analog Ground
Reference Input Voltage
Reference Input Analog Ground
Analog Input Ground
Primart Analog Input with a Range of 0 to 2.5 V.
Temperature sensor voltage output.
Reference Input Voltage. The reference output and the reference buffer input.
Allows choice of Internal or External voltage reference. When HIGH, the internal reference is switched off and an external reference must be used. When low,the on-chip refer
ence is turned on.
Allows choice of buffering internal reference. When LOW, the buffer is selected. When
HIGH, the buffer is switched off.
NOTES
AI = Analog Input
AI/O = Bidirectional Analog
AO = Analog Output
DI = Digital Input
DI/O = Bidirectional Digital
DO = Digital Output
P = Power
–8–
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
ENOB = (S/[N+D]dB – 1.76)/6.02
DEFINITION OF SPECIFICATIONS
INTEGRAL NONLINEARITY ERROR (INL)
Linearity error refers to the deviation of each individual code
from a line drawn from “negative full scale” through “positive full
scale.” The point used as “negative full scale” occurs 1/2 LSB
before the first code transition. “Positive full scale” is defined as a
level 1 1/2 LSB beyond the last code transition. The deviation is
measured from the middle of each code to the true straight line.
and is expressed in bits.
TOTAL HARMONIC DISTORTION (THD)
THD is the ratio of the rms sum of the first five harmonic
components to the rms value of a full-scale input signal and is
expressed in decibels.
SIGNAL-TO-NOISE RATIO (SNR)
DIFFERENTIAL NONLINEARITY ERROR (DNL)
In an ideal ADC, code transitions are 1 LSB apart. Differential nonlinearity is the maximum deviation from this ideal
value. It is often specified in terms of resolution for which no
missing codes are guaranteed.
SNR is the ratio of the rms value of the actual input signal to
the rms sum of all other spectral components below the Nyquist
frequency, excluding harmonics and dc. The value for SNR is
expressed in decibels.
FULL-SCALE ERROR
SIGNAL TO (NOISE + DISTORTION) RATIO
(S/[N+D])
The last transition (from 011 . . . 10 to 011 . . . 11 in two’s
complement coding) should occur for an analog voltage 1 1/2 LSB
below the nominal full scale (2.49994278 V for the 0 V–2.5 V
range). The full-scale error is the deviation of the actual level of
the last transition from the ideal level.
S/(N+D) is the ratio of the rms value of the actual input signal
to the rms sum of all other spectral components below the
Nyquist frequency, including harmonics but excluding dc. The
value for S/(N+D) is expressed in decibels.
APERTURE DELAY
UNIPOLAR ZERO ERROR
The first transition should occur at a level 1/2 LSB above analog
ground (19.073 µV for the 0 V–2.5 V range). Unipolar zero error is
the deviation of the actual transition from that point.
Aperture delay is a measure of the acquisition performance and
is measured from the falling edge of the CNVST input to when
the input signal is held for a conversion.
TRANSIENT RESPONSE
SPURIOUS FREE DYNAMIC RANGE (SFDR)
The difference, in decibels (dB), between the rms amplitude of
the input signal and the peak spurious signal.
The time required for the AD7667 to achieve its rated accuracy after a full-scale step function is applied to its input.
OVERVOLTAGE RECOVERY
EFFECTIVE NUMBER OF BITS (ENOB)
ENOB is a measurement of the resolution with a sine wave
input. It is related to S/(N+D) by the following formula:
REV. PrA
The time required for the ADC to recover to full accuracy
after an analog input signal 150% of full-scale is reduced to
50% of the full-scale value.
–9–
PRELIMINARY TECHNICAL DATA
AD7667
0
0
-20
-20
-40
-60
TO BE
-80
SUPPLIED
-40
-60
TO BE
-80
SUPPLIED
-100
-100
-120
-120
-140
-140
-160
-160
1
1
TPC 1. Integral Nonlinearity vs. Code
TPC 4. Differential Nonlinearity vs. Code
0
0
-20
-20
-40
-40
-60
TO BE
-60
TO BE
-80
SUPPLIED
-80
SUPPLIED
-100
-100
-120
-120
-140
-140
-160
-160
1
1
TPC 2. Histogram of 16,384 Conversions of a DC Input at
the Code Transition
TPC 5. Histogram of 16,384 Conversions of a DC Input at
the Code Center
0
0
-20
-20
-40
-40
-60
-80
TO BE
-60
TO BE
-80
SUPPLIED
SUPPLIED
-100
-100
-120
-120
-140
-140
-160
1
-160
1
TPC 6. SNR, THD vs. Temperature
TPC 3. FFT Plot
–10–
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
0
0
-20
-20
-40
-60
TO BE
-80
SUPPLIED
-40
-100
-60
TO BE
-80
SUPPLIED
-120
-100
-140
-120
-140
-160
1
-160
1
TPC 7. SNR, S/(N+D), and ENOB vs. Frequency
TPC 10. THD, Harmonics, and SFDR vs. Frequency
0
-20
0
-40
-20
-60
TO BE
-80
SUPPLIED
-40
-100
-60
TO BE
-80
SUPPLIED
-100
-120
-120
-140
-140
-160
1
-160
1
TPC 8. SNR and S/(N+D) vs. Input Level
(Referred to Full Scale)
TPC 11. Typical Delay vs. Load Capacitance CL
0
0
-20
-20
-40
-60
TO BE
-80
SUPPLIED
-40
-60
TO BE
-100
-80
SUPPLIED
-120
-100
-140
-120
-140
-160
1
-160
1
TPC 9. Operating Currents vs. Sample Rate
TPC 12. Power-Down Operating Currents vs. Temperature
REV. PrA
–11–
PRELIMINARY TECHNICAL DATA
AD7667
CIRCUIT INFORMATION
After the completion of this process, the control logic generates
the ADC output code and brings BUSY output low.
The AD7667 is a very fast, low power, single supply, precise 16-bit analog-to-digital converter (ADC). The
AD7667 features different modes to optimize performances
according to the applications.
Modes of Operation
The AD7667 features three modes of operations, Warp, Normal,
and Impulse. Each of these modes is more suitable for specific
applications.
In warp mode, the AD7667 is capable of converting
1,000,000 samples per second (1MSPS).
The Warp mode allows the fastest conversion rate up to 1
MSPS. However, in this mode, and this mode only, the full
specified accuracy is guaranteed only when the time between
conversion does not exceed 1 ms. If the time between two consecutive conversions is longer than 1 ms, for instance, after
power-up, the first conversion result should be ignored. This
mode makes the AD7667 ideal for applications where both high
accuracy and fast sample rate are required.
The AD7667 provides the user with an on-chip track/hold,
successive approximation ADC that does not exhibit any pipeline or latency, making it ideal for multiple multiplexed channel
applications.
The AD7667 can be operated from a single 5 V supply and
be interfaced to either 5 V or 3 V digital logic. It is housed
in either a 48-lead LQFP package or a 48-lead LFCSP that
saves space and allows flexible configurations as either serial or
parallel interface. The AD7667 is a pin-to-pin compatible
upgrade of the AD7661/64/66.
The normal mode is the fastest mode (800 kSPS) without any
limitation about the time between conversions. This mode
makes the AD7667 ideal for asynchronous applications such as
data acquisition systems, where both high accuracy and fast
sample rate are required.
CONVERTER OPERATION
The AD7667 is a successive-approximation analog-to-digital
converter based on a charge redistribution DAC. Figure 3 shows
the simplified schematic of the ADC. The capacitive DAC consists
of an array of 16 binary weighted capacitors and an additional
“LSB” capacitor. The comparator’s negative input is connected to a
“dummy” capacitor of the same value as the capacitive DAC
array.
The impulse mode, the lowest power dissipation mode, allows
power saving between conversions. When operating at 100 SPS,
for example, it typically consumes only 15 µW. This feature
makes the AD7667 ideal for battery-powered applications.
Transfer Functions
Using the OB/2C digital input, the AD7667 offers two output
codings: straight binary and two’s complement. The LSB size is
VREF/65536, which is about 38.15 µV. The ideal transfer characteristic for the AD7667 is shown in Figure 4 and Table I.
During the acquisition phase, the common terminal of the array
tied to the comparator's positive input is connected to AGND
via SWA. All independent switches are connected to the analog
input IN. Thus, the capacitor array is used as a sampling capacitor and acquires the analog signal on IN input. Similarly, the
“dummy” capacitor acquires the analog signal on INGND input.
1 LSB = VREF/65536
ADC CODE › Straight Binary
When the CNVST input goes low, a conversion phase is initiated. When the conversion phase begins, SWA and SWB are
opened first. The capacitor array and the “dummy” capacitor are
then disconnected from the inputs and connected to the REFGND input. Therefore, the differential voltage between IN and
INGND captured at the end of the acquisition phase is applied
to the comparator inputs, causing the comparator to become
unbalanced. By switching each element of the capacitor array
between REFGND or REF, the comparator input varies by
binary-weighted voltage steps (VREF/2, VREF/4, . . . VREF/65536).
The control logic toggles these switches, starting with the MSB
first, to bring the comparator back into a balanced condition.
111...111
111...110
111...101
000...010
000...001
000...000
0V
1 LSB
0.5 LSB
VREF › 1 LSB
VREF › 1.5 LSB
ANALOG INPUT
Figure 4. ADC Ideal Transfer Function
IN
REF
REFGND
LSB
MSB
32,768C 16,384C
4C
2C
C
SWA
SWITCHES
CONTROL
C
BUSY
COMP
CONTROL
LOGIC
INGND
OUTPUT
CODE
65,536C
SWB
CNVST
Figure 3. ADC Simplified Schematic
–12–
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
Table I. Output Codes and Ideal Input Voltages
Description
Analog
Input
Digital Output Code
Hexa
Straight Two’s
Binary
Comple-
2.499962 V
2.499923 V
1.250038 V
1.25 V
1.249962 V
38 µV
0V
FFFF1
FFFE
8001
8000
7FFF
0001
00002
TYPICAL CONNECTION DIAGRAM
Figure 5 shows a typical connection diagram for the AD7667.
Analog Input
Figure 6 shows an equivalent circuit of the input structure of
the AD7667.
ment
AVDD
FSR –1 LSB
FSR – 2 LSB
Midscale + 1 LSB
Midscale
Midscale – 1 LSB
–FSR + 1 LSB
–FSR
7FFF 1
7FFE
0001
0000
FFFF
8001
80002
D1
IN
OR INGND
R1
D2
Figure 6. Equivalent Analog Input Circuit
The two diodes D1 and D2 provide ESD protection for
the analog inputs IN and INGND. Care must be taken to
ensure that the analog input signal never exceeds the supply
rails by more than 0.3 V. This will cause these diodes to
become forward-biased and start conducting current.
These diodes can handle a forward-biased current of 100
mA maximum. For instance, these conditions could eventually occur when the input buffer’s (U1) supplies are
100
10F
100nF
DIGITAL SUPPLY
(3.3V OR 5V)
10F
AVDD
AGND
C2
AGND
NOTES
1
This is also the code for overrange analog input (VIN – VINGND above
VREF – VREFGND).
2
This is also the code for underrange analog input (VIN below V INGND).
ANALOG
SUPPLY
(5V)
C1
100nF
DGND
DVDD
100nF
OVDD
10F
OGND
SERIAL
PORT
SCLK
REF
47F
100 nF
SDOUT
REFBUFIN1
REFGND
BUSY
C/ P/DSP
AD7667
CNVST
ANALOG INPUT
(0V TO 2.5V)
D3
15
U12
IN
CC
OB/ 2C
SER/ PAR
2.7nF
INGND
PDREF PD
DVDD
WARP
BYTESWAP
IMPULSE
PDBUF
RESET
CS
RD
NOTES:
1
THE CONFIGURATION SHOWN IS USING THE INTERNAL REFERENCE AND INTERNAL BUFFER
THE AD8021 IS RECOMMENDED. SEE DRIVER AMPLIFIER CHOICE SECTION.
3 OPTIONAL LOW JITTER CNVST.
2
Figure 5. Typical Connection Diagram
REV. PrA
–13–
CLOCK
PRELIMINARY TECHNICAL DATA
AD7667
Source Resistance
different from AVDD. In such case, an input buffer with a
short circuit current limitation can be used to protect the
part.
Driver Amplifier Choice
This analog input structure allows the sampling of the
differential signal between IN and INGND. Unlike other
converters, the INGND input is sampled at the same time as
the IN input. By using this differential input, small signals
common to both inputs are rejected, as shown in Figure 7,
which represents the typical CMRR over frequency. For instance, by using INGND to sense a remote signal ground,
difference of ground potentials between the sensor and the
local ADC ground are eliminated.
Although the AD7667 is easy to drive, the driver amplifier
needs to meet at least the following requirements:
− The driver amplifier and the AD7667 analog input circuit
must be able together to settle for a full-scale step the capacitor array at a 16-bit level (0.0015%). In the amplifier’s data
sheet, the settling at 0.1% to 0.01% is more commonly specified. It could significantly differ from the settling time at 16
bit level and it should therefore be verified prior to the
driver selection. The tiny op amp AD8021, which combines ultralow noise and a high-gain bandwidth, meets
this settling time requirement even when used with high
gain up to 13.
0
-20
-40
-60
TO BE
-80
SUPPLIED
− The noise generated by the driver amplifier needs to be
kept as low as possible in order to preserve the SNR and
transition noise performance of the AD7667. The noise
coming from the driver is filtered by the AD7667 analog
input circuit one-pole low-pass filter made by R1 and C2
or the external filter if any is used. The SNR degredation
due to the amplifier is:
-100
-120
-140
-160
1
SNRLOSS = 20 LOG
Figure 7. Analog Input CMR vs. Frequency
During the acquisition phase, the impedance of the analog
input IN can be modeled as a parallel combination of capacitor C1 and the network formed by the series connection
of R1 and C2. Capacitor C1 is primarily the pin capacitance.
The resistor R1 is typically 183 and is a lumped component made up of some serial resistors and the on resistance of
the switches. The capacitor C2 is typically 60 pF and is mainly
the ADC sampling capacitor. During the conversion phase, where
the switches are opened, the input impedance is limited to C1. The
R1, C2 makes a one-pole low-pass filter that reduces undesirable aliasing effect and limits the noise.
When the source impedance of the driving circuit is low,
the AD7667 can be driven directly. Large source impedances will significantly affect the ac performances,
especially the total harmonic distortion. The maximum
source impedance depends on the amount of total harmonic
distortion (THD) that can be tolerated. The THD degrades
in function of the source impedance and the maximum input
frequency as shown in Figure TBD.
(
28
784 + f-3dB ( N eN )2
2
)
where
f–3dB is the –3 dB input bandwidth of the AD7667 in MHz
(14.5) or the cutoff frequency of the input filter if
any used.
N
is the noise gain of the amplifier (1 if in buffer
configuration).
e N is the equivalent input noise voltage of the op amp
in
nV/
(Hz) 1/2 .
For instance, a driver like the AD8021, with an equivalent
input noise of 2 nV/ Hz and configured as a buffer, thus with
a noise gain of 1, the SNR degrades by only 0.13 dB with the
filter used in figure 5.
− The driver needs to have a THD performance suitable to that
of the AD7667.
The AD8021 meets these requirements and is usually appropriate for almost all applications. The AD8021 needs an
external compensation capacitor of 10 pF. This capacitor
should have good linearity as an NPO ceramic or mica type.
0
-20
-40
-60
TO BE
-80
SUPPLIED
The AD8022 could also be used where dual version is needed
and gain of 1 is used.
-100
-120
-140
-160
1
Figure 8. THD vs. Analog Input Frequency and
–14–
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
The AD829 is another alternative where high-frequency
(above 100 kHz) performance is not required. In gain of
1, it requires an 82 pF compensation capacitor.
The AD8610 is another option where low bias current is
needed in low-frequency applications.
Voltage Reference Input
The AD7667 allows the choice of either an internal 2.5 V
voltage reference or an external 2.5 V reference.
To use the internal reference along with the internal buffer,
PDREF and PDBUF should both be LOW. This will produce a voltage on REFBUFIN of 1.25 V and the buffer’s
gain will be 2, resulting in a 2.5 V reference on REF pin.
To use an external reference along with the internal
buffer, PDREF should be HIGH and PDBUF should be
LOW. This powers down the internal reference and allows
for the 2.5 V reference to be applied to REFBUFIN. In
this mode the buffer’s gain is 1.
To use both external reference, PDREF and PDBUF
should both be HIGH. The reference input should be
applied to REF.
It is useful to decouple the REFBUFIN pin with a 100 nF
ceramic capacitor. The output impedance of the REFBUFIN
pin is 4 k. Thus, the 100 nF capacitor provides an RC filter
for noise reduction.
It should be noted that the internal reference and internal
buffer are independent of the power down (PD) pin of the
part. Powering down the part does not power down the internal reference or the internal buffer. Furthermore, powering
down the internal reference and internal buffer, as well as
powering them up, requires time. This is due to the fact that
we have charging and discharging capacitors on the REF
which require some settling time. Therefore, for applications
requiring low power, there will always be a typical of 10 mW
of power dissipated when using the internal reference and
internal buffer even during times with no conversions.
The internal reference is temperature compensated to
2.5V ± TBD mV. The reference is trimmed to provide
a typical drift of TBD ppm/C. This typical drift characteristic is shown in Figure TBD. For improved drift
performance, an external reference such as the AD780
can be used.
0
-20
For the external reference, the voltage reference input REF
of the AD7667 has a dynamic input impedance; it should
therefore be driven by a low-impedance source with an
efficient decoupling between REF and REFGND inputs.
This decoupling depends on the choice of the voltage reference but usually consists of a 1 µF ceramic capacitor and a
low ESR tantalum capacitor connected to the REF and
REFGND inputs with minimum parasitic inductance. 47
µF is an appropriate value for the tantalum capacitor when
using either the internal reference of one of the recommended reference voltages:
− The low noise, low temperature drift ADR421 and
AD780 voltage references
− The low power ADR291 voltage reference
− The low cost AD1582 voltage reference
For applications using multiple AD7667s, it is more effective
to buffer the reference voltage using the internal buffer. To
do so, PDREF should be HIGH, and PDBUF should be low.
Care should also be taken with the reference temperature
coefficient of the voltage reference which directly affects the
full-scale accuracy if this parameter matters. For instance, a
±15 ppm/°C tempco of the reference changes the full scale by
±1 LSB/°C.
VREF , as mentioned in the specification table, could be increased
to AVDD – 1.85 V. The benefit here is the increased SNR
obtained as a result of this increase. Since the input range is
defined in terms of VREF, this would essentially increase the
range to make it a 0 to 3 V input range with an AVDD above
4.85 V. One of the benefits here is the additional SNR obtained as a result of this increase. The theoretical
improvement as a result of this increase in reference is 1.58
dB (20 log [3/2.5]). Due to the theoretical quantization
noise, however, the observed improvement is approximately
1 dB. The AD780 can be selected with a 3 V reference
voltage.
The TEMP pin, which measures the temperature of the
AD7667, can be used as follows. Refer to figure TBD to
see the connectivity. The output of the TEMP pin is applied to one of the inputs of the analog switch (ADG779).
The other input, as shown is the analog signal. The output
of the switch is connected to the AD8021 which is configured as a follower. The output of the op-amp is applied to
the IN pin. Refer to the Specification Table for the appropriate values related to the TEMP pin. This configuration
could be very useful to improve the calibration accuracy
over the temperature range.
-40
-60
TO BE
-80
SUPPLIED
TEMP
ADG779
-100
-140
IN
-160
1
Figure TBD
REV. PrA
IN
ANALOG INPUT
(UNIPOLAR)
-120
–15–
AD8021
CC
temperature
sensor
AD7667
PRELIMINARY TECHNICAL DATA
AD7667
Figure TBD
t2
t1
CNVST
Power Supply
The AD7667 uses three sets of power supply pins: an analog 5 V
supply AVDD, a digital 5 V core supply DVDD, and a digital
input/output interface supply OVDD. The OVDD supply
allows direct interface with any logic working between 2.7 V and
DVDD + 0.3 V. To reduce the number of supplies needed,
the digital core (DVDD) can be supplied through a simple
RC filter from the analog supply as shown in Figure 5. The
AD7667 is independent of power supply sequencing, once
OVDD does not exceed DVDD by more than 0.3V, and thus
free from supply voltage induced latchup.
POWER DISSIPATION Vs. THROUGHPUT
Operating currents are very low during the acquisition phase,
which allows a significant power saving when the conversion rate
is reduced as shown in Figure 10. This power saving depends on
the mode used. In impulse mode, the AD7667 automatically
reduces its power consumption at the end of each conversion
phase. This feature makes the AD7667 ideal for very low power
battery applications. It should be noted that the digital interface
remains active even during the acquisition phase. To reduce the
operating digital supply currents even further, the digital inputs
need to be driven close to the power supply rails (i.e., DVDD or
DGND) and OVDD should not exceed DVDD by more than
0.3V.
BUSY
t4
t3
t6
t5
MODE
ACQUIRE
CONVERT
ACQUIRE
t7
t8
CONVERT
Figure 11. Basic Conversion Timing
In impulse mode, conversions can be automatically initiated. If CNVST is held low when BUSY is low, the
AD7667 controls the acquisition phase and then automatically initiates a new conversion. By keeping CNVST low,
the AD7667 keeps the conversion process running by itself.
It should be noted that the analog input has to be settled
when BUSY goes low. Also, at power-up, CNVST
should be brought low once to initiate the conversion process. In this mode, the AD7667 could sometimes run
slightly faster then the guaranteed limits in the impulse mode
of 666 kSPS. This feature does not exist in warp or normal
modes.
t9
RESET
0
-20
-40
BUSY
-60
TO BE
-80
SUPPLIED
-100
DATA
-120
t8
-140
-160
CNVST
1
Figure 12. RESET Timing
Figure 10. Power Dissipation vs. Sample Rate
CONVERSION CONTROL
Figure 11 shows the detailed timing diagrams of the conversion process. The AD7667 is controlled by the signal CNVST
which initiates conversion. Once initiated, it cannot be
restarted or aborted, even by the power-down input PD, until
the conversion is complete. The CNVST signal operates
independently of CS and RD signals.
Although CNVST is a digital signal, it should be designed with special care with fast, clean edges, and levels
with minimum overshoot and undershoot or ringing.
It is a good thing to shield the CNVST trace with ground and
also to add a low value serial resistor (i.e., 50 ) termination
close to the output of the component that drives this line.
For applications where the SNR is critical, CNVST signal
should have a very low jitter. Some solutions to achieve that is
to use a dedicated oscillator for CNVST generation or, at least,
to clock it with a high-frequency low-jitter clock as shown in
Figure 5.
DIGITAL INTERFACE
The AD7667 has a versatile digital interface; it can be
interfaced with the host system by using either a serial or
–16–
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
low in a single AD7667 design. RD is generally used to
enable the conversion result on the data bus.
parallel interface. The serial interface is multiplexed on
the parallel data bus. The AD7667 digital interface also
accommodates both 3 V or 5 V logic by simply connecting
the OVDD supply pin of the AD7667 to the host system
interface digital supply. Finally, by using the OB/2C input pin, both two’s complement or straight binary coding
can be used.
The two signals CS and RD control the interface. CS and RD
have a similar effect because they are OR’d together internally. When at least one of these signals is high, the interface
outputs are in high impedance. Usually, CS allows the selection of each AD7667 in multicircuits applications and is held
EXT/INT = 0
CS, RD
RDC/SDIN = 0
INVSCLK = INVSYNC = 0
t3
CNVST
t 28
BUSY
t 30
t 29
t 25
SYNC
t 18
t 19
t 14
t 20
1
SCLK
t 24
t 21
2
3
14
15
t 26
16
t 15
t 27
SDOUT
D15
X
t 16
D14
D2
D1
D0
t 23
t 22
Figure 16. Master Serial Data Timing for Reading (Read After Convert)
EXT/INT = 0
RDC/SDIN = 1
INVSCLK = INVSYNC = 0
CS, RD
t1
CNVST
t3
BUSY
t 17
t 25
SYNC
t 14
t 19
t 20
t 15
SCLK
1
t 21
t 24
2
3
14
15
t 26
16
t 18
t 27
SDOUT
X
t 16
t 22
D15
D14
D2
D1
D0
t 23
Figure 17. Master Serial Data Timing for Reading (Read Previous Conversion During Convert)
REV. PrA
–17–
PRELIMINARY TECHNICAL DATA
AD7667
EXT/INT = 1
INVSCLK = 0
RD = 0
CS
BUSY
t 36
SCLK
t 35
t 37
1
2
t 31
3
14
15
16
17
18
t 32
X
SDOUT
t 16
D15
D14
D13
X14
X13
D1
D0
X15
X14
X0
Y15
Y14
t 34
SDIN
X15
X1
t 33
Figure 18. Slave Serial Data Timing for Reading (Read After Convert)
CS = RD = 0
CS
t1
CNVST
RD
t 10
BUSY
t4
t3
DATA
BUS
BUSY
t 11
PREVIOUS CONVERSION DATA
NEW DATA
DATA
BUS
CURRENT
CONVERSION
t 12
Figure 13. Master Parallel Data Timing for Reading
(Continuous Read)
t 13
Figure 14. Slave Parallel Data Timing for Reading
(Read After Convert)
PARALLEL INTERFACE
The AD7667 is configured to use the parallel interface when
the SER/PAR is held low. The data can be read either after
each conversion, which is during the next acquisition phase,
or during the following conversion as shown, respectively, in
Figure 14 and Figure 15. When the data is read during the
conversion, however, it is recommended that it is read only
during the first half of the conversion phase. That avoids
any potential feedthrough between voltage transients on the
digital interface and the most critical analog conversion circuitry.
The BYTESWAP pin allows a glueless interface to a 8 bits
bus. As shown in Figure TBD, the LSB byte is output on
D[7:0] and the MSB is output on D[15:8] when
BYTESWAP is low. When BYTESWAP is high, the LSB and
MSB bytes are swapped and the LSB is output on D[15:8]
and the MSB is output on D[7:0]. By connecting
BYTESWAP to an address line, the 16 bits data can be read
in 2 bytes on either D[15:8] or D[7:0].
–18–
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
EXT/INT = 1
RD = 0
INVSCLK = 0
CS
CNVST
BUSY
t3
t 36
SCLK
t 35
t 37
1
2
t 31
X
SDOUT
3
14
15
16
t 32
D15
D14
D13
D1
D0
t 16
Figure 20. Slave Serial Data Timing for Reading (Read Previous Conversion During Convert)
MASTER SERIAL INTERFACE
Internal Clock
CS
RD
BYTE
Pins D[15:8]
HI-Z
HI-Z
HIGH BYTE
t 12
Pins D[7:0]
HI-Z
LOW BYTE
t 12
LOW BYTE
HIGH BYTE
t 13
HI-Z
Figure TBD, 8-bit Parallel Interface
CS = 0
t1
CNVST
,
RD
The AD7667 is configured to generate and provide the serial
data clock SCLK when the EXT/INT pin is held low. The
AD7667 also generates a SYNC signal to indicate to the host
when the serial data is valid. The serial clock SCLK and the
SYNC signal can be inverted if desired. Depending on RDC/
SDIN input, the data can be read after each conversion or
during the following conversion. Figure 16 and Figure 17
show the detailed timing diagrams of these two modes.
Usually, because the AD7667 is used with a fast throughput,
the mode master, read during conversion is the most recommended serial mode when it can be used.
In read-during-conversion mode, the serial clock and data
toggle at appropriate instants which minimize potential
feedthrough between digital activity and the critical conversion decisions.
In read-after-conversion mode, it should be noted that, unlike in
other modes, the signal BUSY returns low after the 16 data bits
are pulsed out and not at the end of the conversion phase which
results in a longer BUSY width.
BUSY
t4
SLAVE SERIAL INTERFACE
t3
External Clock
DATA
BUS
PREVIOUS
CONVERSION
t 12
t 13
Figure 15. Slave Parallel Data Timing for Reading
(Read During Convert)
SERIAL INTERFACE
The AD7667 is configured to use the serial interface when
the SER/PAR is held high. The AD7667 outputs 16 bits of
data, MSB first, on the SDOUT pin. This data is synchronized with the 16 clock pulses provided on SCLK pin. The
output data is valid on both the rising and falling edge of the
data clock.
REV. PrA
The AD7667 is configured to accept an externally supplied
serial data clock on the SCLK pin when the EXT/INT pin is
held high. In this mode, several methods can be used to read
the data. The external serial clock is gated by CS. When CS and
RD are both low, the data can be read after each conversion
or during the following conversion. The external clock can be
either a continuous or discontinuous clock. A discontinuous
clock can be either normally high or normally low when inactive. Figure 18 and Figure 20 show the detailed timing diagrams
of these methods.
While the AD7667 is performing a bit decision, it is important that voltage transients not occur on digital input/output
pins or degradation of the conversion result could occur.
This is particularly important during the second half of the
conversion phase because the AD7667 provides error correction circuitry that can correct for an improper bit
–19–
PRELIMINARY TECHNICAL DATA
AD7667
decision made during the first half of the conversion
phase. For this reason, it is recommended that when an
external clock is being provided, it is a discontinuous
clock that is toggling only when BUSY is low or, more
importantly, that it does not transition during the latter
half of BUSY high.
External Discontinuous Clock Data Read After Conversion
Though the maximum throughput cannot be achieved using
this mode, it is the most recommended of the serial slave
modes. Figure 18 shows the detailed timing diagrams of this
method. After a conversion is complete, indicated by BUSY
returning low, the result of this conversion can be read while
both CS and RD are low. The data is shifted out, MSB first,
with 16 clock pulses and is valid on both rising and falling
edge of the clock.
Among the advantages of this method, the conversion performance is not degraded because there are no voltage transients
on the digital interface during the conversion process.
Another advantage is to be able to read the data at any speed
up to 40 MHz which accommodates both slow digital host
interface and the fastest serial reading.
Finally, in this mode only, the AD7667 provides a “daisychain” feature using the RDC/SDIN input pin for cascading
multiple converters together. This feature is useful for reducing
component count and wiring connections when desired as, for
instance, in isolated multiconverter applications.
An example of the concatenation of two devices is shown in
Figure 19. Simultaneous sampling is possible by using a common CNVST signal. It should be noted that the RDC/SDIN
input is latched on the edge of SCLK opposite to the one used to
shift out the data on SDOUT. Hence, the MSB of the “upstream”
converter just follows the LSB of the “downstream” converter
on the next SCLK cycle.
BUSY
AD7667
AD7667
#2
(UPSTREAM)
#1
(DOWNSTREAM)
RDC/SDIN
SDOUT
CNVST
RDC/SDIN
SDOUT
DATA
OUT
CNVST
CS
CS
SCLK
SCLK
To reduce performance degradation due to digital activity, a
fast discontinuous clock of, at least 25 MHz, when impulse mode is
used, 32 MHz when normal mode is used or 40 MHz when
warp mode is used, is recommended to ensure that all the bits
are read during the first half of the conversion phase. It is
also possible to begin to read the data after conversion and
continue to read the last bits even after a new conversion has
been initiated. That allows the use of a slower clock speed like
18 MHz in impulse mode, 21 MHz in normal mode and 26
MHz in warp mode.
MICROPROCESSOR INTERFACING
The AD7667 is ideally suited for traditional dc measurement applications supporting a microprocessor, and ac signal
processing applications interfacing to a digital signal processor.
The AD7667 is designed to interface either with a parallel 16bit-wide interface or with a general-purpose serial port or I/O
ports on a microcontroller. A variety of external buffers can be
used with the AD7667 to prevent digital noise from coupling
into the ADC. The following sections illustrate the use of the
AD7667 with an SPI-equipped microcontroller, the ADSP21065L and ADSP-218x signal processors.
SPI Interface (MC68HC11)
Figure 21 shows an interface diagram between the AD7667
and an SPI-equipped microcontroller like the MC68HC11. To
accommodate the slower speed of the microcontroller, the
AD7667 acts as a slave device and data must be read after
conversion. This mode allows also the “daisy chain” feature.
The convert command could be initiated in response to
an internal timer interrupt. The reading of output data,
one byte at a time, if necessary, could be initiated in
response to the end-of-conversion signal (BUSY going
low) using to an interrupt line of the microcontroller. The
Serial Peripheral Interface (SPI) on the MC68HC11 is
configured for master mode (MSTR = 1), Clock Polarity Bit (CPOL) = 0, Clock Phase Bit (CPHA) = 1 and
SPI Interrupt Enable (SPIE = 1) by writing to the SPI Control Register (SPCR). The IRQ is configured for
edge-sensitive-only operation (IRQE = 1 in OPTION
register).
BUSY
OUT
BUSY
valid on both rising and falling edge of the clock. The 16
bits have to be read before the current conversion is complete. If that is not done, RDERROR is pulsed high and
can be used to interrupt the host interface to prevent
incomplete data reading. There is no “daisy chain”
feature in this mode and RDC/SDIN input should always be tied either high or low.
SCLK IN
CS IN
CNVST IN
Figure 19. Two AD7667s in a “Daisy-Chain” Configuration
External Clock Data Read During Conversion
Figure 20 shows the detailed timing diagrams of this
method. During a conversion, while both CS and RD are
both low, the result of the previous conversion can be read.
The data is shifted out, MSB first, with 16 clock pulses and is
–20–
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
DVDD
AD7667*
MC68HC11*
DVDD
SER/PAR
For applications where accurate gain and offset are desired, they can be calibrated by acquiring a ground and a
voltage reference using an analog multiplexer, U2, as
shown for bipolar input ranges in Figure 23.
EXT/INT
BUSY
CS
SDOUT
RD
SCLK
INVSCLK
CNVST
Figure 23 shows a connection diagram which allows
that. Components values required and resulting fullscale ranges are shown in Table II.
IRQ
MISO/SDI
SCK
I/O PORT
CF
*ADDITIONAL PINS OMITTED FOR CLARITY
Figure 21. Interfacing the AD7667 to SPI Interface
R1
ADSP-21065L in Master Serial Interface
The AD7667 is configured for the internal clock mode (EXT/
INT low) and acts, therefore, as the master device. The convert command can be generated by either an external low
jitter oscillator or, as shown, by a FLAG output of the ADSP21065L or by a frame output TFS of one serial port of the
ADSP-21065L which can be used as a timer. The serial port
on the ADSP-21065L is configured for external clock (IRFS
= 0), rising edge active (CKRE = 1), external late framed
sync signals (IRFS = 0, LAFS = 1, RFSR = 1) and active
high (LRFS = 0). The serial port of the ADSP-21065L is
configured by writing to its receive control register
(SRCTL)—see ADSP-2106x SHARC User’s Manual. Because the serial port within the ADSP-21065L will be seeing a
discontinuous clock, an initial word reading has to be done after
the ADSP-21065L has been reset to ensure that the serial
port is properly synchronized to this clock during each following data read operation.
R2
ANALOG
INPUT
As shown in Figure 22, the AD7667 can be interfaced to
the ADSP-21065L using the serial interface in master mode
without any glue logic required. This mode combines the
advantages of reducing the number of wire connections and
being able to read the data during or after conversion at user
convenience.
IN
U1
AD7667
U2
R3
100nF
R4
INGND
REF
CREF
100nF
REFGND
Figure 23. Using the AD7667 in 16-Bit Bipolar and/or
Wider Input Ranges
Table II. Component Values and Input Ranges
Input Range
R1
R2
R3
R4
±10 V
±5 V
0 V to –5 V
250 500 1 k
2 k
2 k
2 k
10 k
10 k
None
8 k
6.67 k
0
Layout
DVDD
AD7667*
ADSP-21065L*
DVDD
SHARC
SER/PAR
RDC/SDIN
RD
EXT/INT
CS
SYNC
SDOUT
INVSYNC
SCLK
INVSCLK
CNVST
RFS
DR
RCLK
FLAG OR TFS
*ADDITIONAL PINS OMITTED FOR CLARITY
Figure 22. Interfacing to the ADSP-21065L Using the
Serial Master Mode
APPLICATION HINTS
Bipolar and Wider Input Ranges
In some applications, it is desired to use a bipolar or wider
analog input range like, for instance, ±10 V, ±5 V or 0 V to 5 V.
Although the AD7667 has only one unipolar range, by simple
modifications of the input driver circuitry, bipolar and wider
input ranges can be used without any performance degradation.
REV. PrA
The AD7667 has very good immunity to noise on the
power supplies as can be seen in Figure 9. However, care
should still be taken with regard to grounding layout.
The printed circuit board that houses the AD7667 should be
designed so the analog and digital sections are separated and
confined to certain areas of the board. This facilitates the use
of ground planes that can be easily separated. Digital and
analog ground planes should be joined in only one place,
preferably underneath the AD7667, or, at least, as close as
possible to the AD7667. If the AD7667 is in a system where
multiple devices require analog-to-digital ground connections,
the connection should still be made at one point only, a star
ground point, which should be established as close as possible
to the AD7667.
It is recommended to avoid running digital lines under
the device as these will couple noise onto the die. The analog ground plane should be allowed to run under the
AD7667 to avoid noise coupling. Fast switching signals
like CNVST or clocks should be shielded with digital
ground to avoid radiating noise to other sections of the
board, and should never run near analog signal paths.
Crossover of digital and analog signals should be avoided.
Traces on different but close layers of the board should run
–21–
PRELIMINARY TECHNICAL DATA
AD7667
at right angles to each other. This will reduce the effect of
feedthrough through the board.
The power supplies lines to the AD7667 should use as
large trace as possible to provide low impedance paths and
reduce the effect of glitches on the power supplies lines.
Good decoupling is also important to lower the supplies
impedance presented to the AD7667 and reduce the magnitude of the supply spikes. Decoupling ceramic capacitors,
typically 100 nF, should be placed on each power supplies
pins AVDD, DVDD, and OVDD close to, and ideally right
up against, these pins and their corresponding ground pins.
Additionally, low ESR 10 µF capacitors should be located in
the vicinity of the ADC to further reduce low frequency
ripple.
The DVDD supply of the AD7667 can be either a separate
supply or come from the analog supply AVDD or the digital interface supply OVDD. When the system digital supply
is noisy, or fast switching digital signals are present, it is
recommended that if no separate supply available, connect
the DVDD digital supply to the analog supply, AVDD,
through an RC filter as shown in Figure 5, and connect
the system supply to the interface digital supply, OVDD,
and the remaining digital circuitry. When DVDD is powered
from the system supply, it is useful to insert a bead to further reduce high-frequency spikes.
The AD7667 has five different ground pins: INGND, REFGND, AGND, DGND, and OGND. INGND is used to
sense the analog input signal. REFGND senses the reference voltage and should be a low impedance return to the
reference because it carries pulsed currents. AGND is the
ground to which most internal ADC analog signals are
referenced. This ground must be connected with the least
resistance to the analog ground plane. DGND must be
tied to the analog or digital ground plane depending on the
configuration. OGND is connected to the digital system
ground.
Evaluating the AD7667 Performance
A recommended layout for the AD7667 is outlined in the
evaluation board for the AD7667. The evaluation board
package includes a fully assembled and tested evaluation
board, documentation, and software for controlling the board
from a PC via the Eval-Control Board.
–22–
REV. PrA
PRELIMINARY TECHNICAL DATA
AD7667
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
48-Lead Quad Flatpack (LQFP)
(ST-48)
0.063 (1.60)
MAX
0.030 (0.75)
0.018 (0.45)
0.354 (9.00) BSC SQ
37
48
36
1
0.276
(7.00)
BSC
SQ
TOP VIEW
(PINS DOWN)
COPLANARITY
0.003 (0.08)
0
MIN
25
12
13
24
0.019 (0.5) 0.011 (0.27)
BSC
0.006 (0.17)
0.008 (0.2)
0.004 (0.09)
0.057 (1.45)
0.053 (1.35)
7
0
0.006 (0.15) SEATING
0.002 (0.05) PLANE
48-Lead Frame Chip Scale Package (LQFP)
(CP-48)
0.024 (0.60)
0.017 (0.42)
0.009 (0.24)
0.024 (0.60)
0.017 (0.42)
37
0.009 (0.24)
36
0.276 (7.0)
BSC SQ
PIN 1
INDICATOR
0.266 (6.75)
BSC SQ
TOP
VIEW
0.039 (1.00) MAX
0.033 (0.85) NOM
0.031 (0.80) MAX
0.026 (0.65) NOM
0.020 (0.50)
BSC
0.008 (0.20)
REF
1
12
25
24
13
0.012 (0.30)
0.009 (0.23)
0.007 (0.18)
0.002 (0.05)
0.0004 (0.01)
0.0 (0.0)
CONTROLLING DIMENSIONS ARE IN MILLIMETERS
REV. PrA
0.215 (5.45)
0.209 (5.30) SQ
0.203 (5.15)
BOTTOM
VIEW
0.020 (0.50)
0.016 (0.40)
0.012 (0.30)
12 MAX
48
–23–
Pad Connected to AGND