AD ADP1073AR-12 Micropower dc.dc converter adjustable and fixed 3.3 v, 5 v, 12 v Datasheet

a
Micropower DC–DC Converter
Adjustable and Fixed 3.3 V, 5 V, 12 V
ADP1073
FEATURES
Operates at Supply Voltages from 1.0 V to 30 V
Ground Current 100 mA
Works in Step-Up or Step-Down Mode
Very Few External Components Required
Low Battery Detector On-Chip
User-Adjustable Current Limit
Internal 1 A Power Switch
Fixed and Adjustable Output Voltage Versions
8-Lead DIP or SO-8 Package
APPLICATIONS
Single-Cell to 5 V Converters
Laptop and Palmtop Computers
Pagers
Cameras
Battery Backup Supplies
Cellular Telephones
Portable Instruments
4 mA–20 mA Loop Powered Instruments
Hand-Held Inventory Computers
FUNCTIONAL BLOCK DIAGRAMS
SET
ADP1073
A2
AO
GAIN BLOCK/
ERROR AMP
ILIM
VIN
SW1
212mV
REFERENCE
A1
OSCILLATOR
COMPARATOR
GND
DRIVER
FB
ADP1073
SET
ADP1073-3.3
ADP1073-5
ADP1073-12
A2
VIN
AO
GAIN BLOCK/
ERROR AMP
212mV
REFERENCE
GENERAL DESCRIPTION
The ADP1073 is part of a family of step-up/step-down switching regulators that operates from an input supply voltage of as
little as 1.0 V. This extremely low input voltage allows the
ADP1073 to be used in applications requiring use of a single
cell battery as the primary power source.
SW2
R1
SW1
A1
OSCILLATOR
COMPARATOR
DRIVER
SW2
ADP1073-3.3: R1 = 62.1kV
ADP1073-5: R1 = 40kV
ADP1073-12: R1 = 16.3kV
R2
904kV
GND
ILIM
SENSE
ADP1073-3.3, 5, 12
The ADP1073 can be configured to operate in either step-up or
step-down mode but for input voltages greater than 3 V, the
ADP1173 is recommended.
An auxiliary gain amplifier can serve as a low battery detector or
linear regulator. Quiescent current on the ADP1073-5 is only
100 µA unloaded, making it ideal for systems where long battery
life is required.
The ADP1073 can deliver 40 mA at 5 V from an input voltage
range as low as 1.25 V, or 10 mA at 5 V from a 1.0 V input.
Current limiting is available by adding an external resistor.
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1997
ADP1073–SPECIFICATIONS (@ T = 08C to +708C, V
A
IN
= 1.5 V unless otherwise noted)
Parameter
Conditions
Symbol
QUIESCENT CURRENT
Switch Off
QUIESCENT CURRENT, STEP-UP
MODE CONFIGURATION
Typ
Max
Units
IQ
100
165
µA
No Load, ADP1073-3.3
ADP1073-5
ADP1073-12, TA = +25°C
IQ
100
100
100
INPUT VOLTAGE
Step-Up Mode
Step-Up Mode, TA = +25°C
Step-Down Mode
VIN
COMPARATOR TRIP POINT VOLTAGE
ADP10731
2
VOUT
Min
1.15
1.0
µA
µA
µA
12.6
12.6
30
V
V
V
200
212
222
mV
3.14
4.75
11.4
3.30
5.00
12.00
3.47
5.25
12.6
V
V
V
OUTPUT SENSE VOLTAGE
ADP1073-3.3
ADP1073-52
ADP1073-122
COMPARATOR HYSTERESIS
ADP1073
5
10
mV
OUTPUT HYSTERESIS
ADP1073-3.3
ADP1073-5
ADP1073-12
90
125
300
130
250
600
mV
mV
mV
OSCILLATOR FREQUENCY
MAXIMUM DUTY CYCLE
Full Load (VFB < VREF)
SWITCH ON TIME
fOSC
14
19
24
kHz
DC
57
72
80
%
tON
28
38
50
µs
FEEDBACK PIN BIAS CURRENT
ADP1073 VFB = 0 V
IFB
60
300
nA
SET PIN BIAS CURRENT
VSET = VREF
ISET
100
220
nA
AO OUTPUT LOW
IAO = 100 µA
VAO
0.15
0.4
V
REFERENCE LINE REGULATION
1.0 V ≤ VIN ≤ 1.5 V
1.5 V ≤ VIN ≤ 12 V
0.35
0.05
0.15
%/V
%/V
450
600
550
750
1000
1500
mV
mV
mV
mV
mV
mV
SWITCH SATURATION VOLTAGE
STEP-UP MODE
VIN = 1.5 V, ISW = 400 mA, +25°C
TMIN to TMAX
VIN = 1.5 V, ISW = 500 mA, +25°C
TMIN to TMAX
VIN = 5 V, ISW = 1 A, +25°C
TMIN to TMAX
VCESAT
RL = 100 kΩ3
AV
REVERSE BATTERY CURRENT
TA = +25°C
CURRENT LIMIT
220 Ω Between ILIM and VIN
TA = +25°C
A2 ERROR AMP GAIN
4
400
700
1000
V/V
IREV
750
mA
ILIM
400
mA
–0.3
%/°C
CURRENT LIMIT TEMPERATURE
COEFFICIENT
SWITCH-OFF LEAKAGE CURRENT
MAXIMUM EXCURSION BELOW GND
300
400
Measured at SW1 Pin
TA = +25°C
ILEAK
1
15
µA
ISW1 ≤ 10 µA, Switch Off
TA = +25°C
VSW2
–400
–350
mV
NOTES
1
This specification guarantees that both the high and low trip point of the comparator fall within the 200 mV to 222 mV range.
2
This specification guarantees that the output voltage of the fixed versions will always fall within the specified range. The waveform at the sense pin will exhibit a
sawtooth shape due to the comparator hysteresis.
3
100 kΩ resistor connected between a 5 V source and the AO pin.
4
The ADP1073 is guaranteed to withstand continuous application of +1.6 V applied to the GND and SW2 pins while VIN, ILIM and SW1 pins are grounded.
All limits at temperature extremes are guaranteed via correlation using standard Quality Control methods.
Specifications subject to change without notice.
–2–
REV. 0
ADP1073
ABSOLUTE MAXIMUM RATINGS
PIN FUNCTION DESCRIPTIONS
Input Supply Voltage, Step-Up Mode . . . . . . . . . . . . . . . 15 V
Input Supply Voltage, Step-Down Mode . . . . . . . . . . . . . 36 V
SW1 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50 V
SW2 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . .–0.4 V to VIN
Feedback Pin Voltage (ADP1073) . . . . . . . . . . . . . . . . . . . 5 V
Switch Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .1.5 A
Maximum Power Dissipation . . . . . . . . . . . . . . . . . . 500 mW
Operating Temperature Range (A) . . . . . . . . . . 0°C to +70°C
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . +300°C
Pin
Mnemonic
Function
1
ILIM
For normal conditions this pin is connected to VIN. When a lower current
limit is required, a resistor should be
connected between ILIM and V IN. Limiting the switch current to 400 mA is
achieved by connecting a 220 Ω resistor.
2
VIN
Input Voltage.
3
SW1
Collector Node of Power Transistor.
For step-down configuration, connect to
VIN; for step-up configuration, connect
to an inductor/diode.
4
SW2
Emitter Node of Power Transistor. For
step- down configuration, connect to
inductor/diode; for step-up configuration, connect to ground. Do not allow
this pin to drop more than a diode drop
below ground.
5
GND
Ground.
6
AO
Auxiliary Gain (GB) Output. The open
collector can sink 100 µA.
7
SET
Gain Amplifier Input. The amplifier’s
positive input is connected to the SET
pin and its negative input is connected
to the 212 mV reference.
8
FB/SENSE
On the ADP1073 (adjustable) version
this pin is connected to the comparator
input. On the ADP1073-3.3, ADP10735 and ADP1073-12, the pin goes directly to the internal application resistor
that sets output voltage.
CADDELL-BURNS
7200-12
82mH
1N5818
ILIM
+5V
40mA
VIN
SW1
1.5V
AA CELL*
ADP1073-5
SENSE
GND
SW2
100mF
SANYO
OS-CON
OPERATES WITH CELL VOLTAGE$1.0V
*ADD 10mF DECOUPLING CAPACITOR IF BATTERY
IS MORE THAN 2 INCHES AWAY FROM ADP1073
Figure 1. Typical Application
ORDERING GUIDE
Model*
Output
Voltage
Package
Options**
ADP1073AN
ADP1073AR
ADP1073AN-3.3
ADP1073AR-3.3
ADP1073AN-5
ADP1073AR-5
ADP1073AN-12
ADP1073AR-12
ADJ
ADJ
3.3 V
3.3 V
5V
5V
12 V
12 V
N-8
SO-8
N-8
SO-8
N-8
SO-8
N-8
SO-8
PIN CONFIGURATIONS
8-Lead Plastic DIP
(N-8)
NOTES
**Temperature Range: 0°C to +70°C.
**N = Plastic DIP; SO = Small Outline Package.
8 FB (SENSE)*
ILIM 1
VIN 2
ADP1073
7 SET
TOP VIEW
SW1 3 (Not to Scale) 6 AO
SW2 4
5 GND
* FIXED VERSIONS
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the ADP1073 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
REV. 0
–3–
8-Lead Small Outline Package
(SO-8)
8 FB (SENSE)*
ILIM 1
ADP1073
7 SET
TOP VIEW
SW1 3 (Not to Scale) 6 AO
VIN 2
5 GND
SW2 4
* FIXED VERSIONS
WARNING!
ESD SENSITIVE DEVICE
ADP1073 –Typical Performance Characteristics
VIN = 1.5V
0.8
VIN = 3.0V
VIN = 2.0V
0.6
0.4
VIN = 1.25V
0.2
1.4
1.2
1
0.8
0.6
0.4
0
0.05 0.1
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 1.1 1.2
SWITCH CURRENT – Amps
SET PIN BIAS CURRENT – nA
OUTPUT CURRENT – mA
800
600
VIN = 1.5V
WITH L = 82mH
400
100
FOR VIN > 1.6V,
RLIM = 68V
110
140
100
90
80
70
60
10 30
50 70
90 200 400 600 800 1000
RLIM – V
Figure 4. Maximum Switch Current
vs. RLIM
160
VIN = 1.5V
120
100
80
60
50
40
240
3.5
Figure 5. Guaranteed Minimum
Output Current at VOUT = 5 V vs.
Input Voltage
0.7
120
0
1.5
2
2.5
3
INPUT VOLTAGE – Volts
0
0.2
0.3
0.4
0.5
0.6
SWITCH CURRENT – Amps
Figure 3. Switch ON Voltage vs.
Switch Current in Step-Down Mode
1000
0
25
70
TEMPERATURE – 8C
40
240
85
70
21
68
0
85
25
70
TEMPERATURE – 8C
Figure 7. Supply Current vs.
Temperature
Figure 6. Set Pin Bias Current vs.
Temperature
22
34.5
34
33.5
20
DUTY CYCLE – %
66
19
18
17
16
64
62
60
58
15
14
240
VIN = 12V
WITH L = 150mH
200
SUPPLY CURRENT – mA
Figure 2. Saturation Voltage vs.
Switch Current in Step-Up Mode
1
VIN = 3V
WITH L = 82mH
1000
0.2
VIN = 1.0V
10
1200
SATURATION VOLTAGE
SWITCH-ON TIME – ms
VCE (SAT) – Volts
VIN = 5.0V
1.6
SWITCH CURRENT – mA
SWITCH ON VOLTAGE – Volts
1.8
1
OSCILLATOR FREQUENCY – kHz
1400
2
1.2
0
25
70
TEMPERATURE – 8C
85
Figure 8. Oscillator Frequency vs.
Temperature
56
240
33
32.5
32
31.5
31
30.5
0
25
70
TEMPERATURE – 8C
85
Figure 9. Duty Cycle vs. Temperature
–4–
30
240
0
25
70
TEMPERATURE – 8C
85
Figure 10. Switch ON Time vs.
Temperature
REV. 0
ADP1073
2300
GAIN BLOCK GAIN – V/V
2100
VIN = 1.5V
RL = 100kV
1900
1700
1500
1300
1100
240
0
25
70
TEMPERATURE – 8C
85
Figure 11. “Gain Block” Gain vs. Temperature
THEORY OF OPERATION
The ADP1073 is a flexible, low power switch mode power
supply (SMPS) controller. The regulated output voltage can be
greater than the input voltage (boost or step-up mode) or less
than the input (buck or step-down mode). This device uses a
gated-oscillator technique to provide very high performance
with low quiescent current.
A functional block diagram of the ADP1073 is shown on the
front page. The internal 212 mV reference is connected to one
input of the comparator, while the other input is externally
connected (via the FB pin) to a feedback network connected to
the regulated output. When the voltage at the FB pin falls below
212 mV, the 19 kHz oscillator turns on. A driver amplifier provides base drive to the internal power switch and the switching
action raises the output voltage. When the voltage at the FB pin
exceeds 212 mV, the oscillator is shut off. While the oscillator is
off, the ADP1073 quiescent current is only 100 µA. The comparator includes a small amount of hysteresis, which ensures
loop stability without requiring external components for frequency compensation.
The maximum current in the internal power switch can be set
by connecting a resistor between VIN and the ILIM pin. When
the maximum current is exceeded, the switch is turned OFF.
The current limit circuitry has a time delay of about 2 µs. If an
external resistor is not used, connect ILIM to VIN. Further information on ILIM is included in the Limiting the Switch Current
section of this data sheet.
The ADP1073 internal oscillator provides 38 µs ON and 15 µs
OFF times, which is ideal for applications where the ratio between VIN and VOUT is roughly a factor of three (such as generating +5 V from a single 1.5 V cell). Wider range conversions,
as well as step-down converters, can also be accomplished with
a slight loss in the maximum output power that can be obtained.
REV. 0
An uncommitted gain block on the ADP1073 can be connected
as a low-battery detector, linear post-regulator or undervoltage
lockout detector. The inverting input of the gain block is internally connected to the 212 mV reference. The noninverting
input is available at the SET pin. A resistor divider, connected
between VIN and GND with the junction connected to the SET
pin, causes the AO output to go LOW when the input voltage
goes below the low battery set point. The AO output is an open
collector NPN transistor that can sink 100 µA.
The ADP1073 provides external connections for both the collector and emitter of its internal power switch, which permits
both step-up and step-down modes of operation. For the stepup mode, the emitter (Pin SW2) is connected to GND and the
collector (Pin SW1) drives the inductor. For step-down mode,
the emitter drives the inductor while the collector is connected
to VIN.
The output voltage of the ADP1073 is set with two external
resistors. Three fixed-voltage models are also available:
ADP1073-3.3 (+3.3 V), ADP1073-5 (+5 V) and ADP1073-12
(+12 V). The fixed-voltage models are identical to the ADP1073,
except that laser-trimmed voltage-setting resistors are included
on the chip. Only three external components are required to
form a +3.3 V, +5 V or +12 V converter. On the fixed-voltage
models of the ADP1073, simply connect the feedback pin (Pin
8) directly to the output voltage.
The ADP1073 oscillator only turns on when the output voltage
is below the programmed voltage. When the output voltage is
above the programmed voltage, the ADP1073 remains in its
quiescent state to conserve power. Output ripple, which is inherent in gated oscillator converters, is typically 125 mV for a
5 V output and 300 mV for a 12 V output. This ripple voltage
can be greatly reduced by inserting the gain-block between the
output and the FB pin. Further information and a typical circuit
are shown in the Programming the Gain Block section.
–5–
ADP1073
where L is in henrys and R' is the sum of the switch equivalent
resistance (typically 0.8 Ω at +25°C) and the dc resistance of
the inductor. If the voltage drop across the switch is small compared to VIN, a simpler equation can be used:
COMPONENT SELECTION
General Notes on Inductor Selection
When the ADP1073 internal power switch turns on, current
begins to flow in the inductor. Energy is stored in the inductor
core while the switch is on, and this stored energy is then transferred to the load when the switch turns off. Both the collector
and the emitter of the switch transistor are accessible on the
ADP1073, so the output voltage can be higher, lower or of
opposite polarity than the input voltage.
I L (t) =
EL =
P L = (5V + 0.5V – 2V )×(25 mA) = 87.5 mW
On each switching cycle, the inductor must supply:
P L 87.5 mW
=
= 4.6 µJ
f OSC 19 kHz
Since the inductor power is low, the peak current can also be
low. Assuming a peak current of 100 mA as a starting point,
Equation 4 can be rearranged to recommend an inductor value:
Selecting the proper inductor value is a simple three-step process:
1. Define the operating parameters: minimum input voltage,
maximum input voltage, output voltage and output current.
L=
2. Select the appropriate conversion topology (step-up, stepdown or inverting).
Inductor Selection—Step-Up Converter
I PEAK =
In a step-up, or boost, converter (Figure 15), the inductor must
store enough power to make up the difference between the
input voltage and the output voltage. The power that must be
stored is calculated from the equation:
(1)
The inductor energy of 5.2 µJ is greater than the PL/f OSC requirement of 4.6 µJ, so the 470 µH inductor will work in this
application. The optimum inductor value can be determined
by substituting other inductor values into the same equations.
When selecting an inductor, the peak current must not exceed
the maximum switch current of 1.5 A.
(2)
The peak current must be evaluated for both minimum and
maximum values of input voltage. If the switch current is high
when VIN is at its minimum, then the 1.5 A limit may be exceeded
at the maximum value of VIN. In this case, the ADP1073’s current
When the internal power switch turns ON, current flow in the
inductor increases at the rate of:
–R′t 
V IN 
1– e L 

R′ 

–2.0 Ω × 38 µs 

1 – e 470 µH  = 149 mA


1
E L = (470 µH )× (149 mA)2 = 5.2 µJ
2
in order for the ADP1073 to regulate the output voltage.
I L (t) =
2V
2.0 Ω
Once the peak current is known, the inductor energy can be
calculated from Equation 5:
where VD is the diode forward voltage (≈ 0.5 V for a 1N5818
Schottky). Energy is only stored in the inductor while the
ADP1073 switch is ON, so the energy stored in the inductor on
each switching cycle must be must be equal to or greater than:
PL
f OSC
V IN
2V
t=
38 µs = 760 µH
I L(MAX ) 100 mA
Substituting a standard inductor value of 470 µH, with 1.2 Ω dc
resistance, will produce a peak switch current of:
3. Calculate the inductor value, using the equations in the
following sections.
)
(5)
In practice, the inductor value is easily selected using the equations above. For example, consider a supply that will generate
5 V at 25 mA from two alkaline batteries with a 2 V end-of-life
voltage. The inductor power required is, from Equation 1:
Calculating the Inductor Value
) (
1
L × I 2 PEAK
2
As previously mentioned, EL must be greater than PL/fOSC so the
ADP1073 can deliver the necessary power to the load. For best
efficiency, peak current should be limited to 1 A or less. Higher
switch currents will reduce efficiency because of increased saturation voltage in the switch. High peak current also increases
output ripple. As a general rule, keep peak current as low as
possible to minimize losses in the switch, inductor and diode.
To minimize Electro-Magnetic Interference (EMI), a toroid or
pot core type inductor is recommended. Rod core inductors are
a lower cost alternative if EMI is not a problem.
(
(4)
Replacing t in the above equation with the ON time of the ADP1073
(38 µs, typical) will define the peak current for a given inductor
value and input voltage. At this point, the inductor energy can
be calculated as follows:
To specify an inductor for the ADP1073, the proper values of
inductance, saturation current and dc resistance must be determined. This process is not difficult, and specific equations for
each circuit configuration are provided in this data sheet.
In general terms, however, the inductance value must be low
enough to store the required amount of energy (when both
input voltage and switch ON time are at a minimum) but high
enough that the inductor will not saturate when both VIN and
switch ON time are at their maximum values. The inductor
must also store enough energy to supply the load without saturating. Finally, the dc resistance of the inductor should be low
so that excessive power will not be wasted by heating the
windings. For most ADP1073 applications, an 82 µH to
1000 µH inductor with a saturation current rating of 300 mA to
1 A is suitable. Ferrite core inductors that meet these specifications are available in small, surface-mount packages.
P L = V OUT +V D –V IN(MIN ) × IOUT
V IN
t
L
(3)
–6–
REV. 0
ADP1073
limit feature can be used to limit switch current. Simply select a
resistor (using Figure 4) that will limit the maximum switch
current to the IPEAK value calculated for the minimum value of
VIN. This will improve efficiency by producing a constant IPEAK
as VIN increases. See the Limiting the Switch Current section of
this data sheet for more information.
Note that the switch current limit feature does not protect the
circuit if the output is shorted to ground. In this case, current is
limited only by the dc resistance of the inductor and the forward
voltage of the diode.
Inductor Selection—Step-Down Converter
The step-down mode of operation is shown in Figure 16. Unlike
the step-up mode, the ADP1073’s power switch does not saturate when operating in the step-down mode. Switch current
should therefore be limited to 600 mA for best performance in
this mode. If the input voltage will vary over a wide range, the
ILIM pin can be used to limit the maximum switch current.
The first step in selecting the step-down inductor is to calculate
the peak switch current as follows:
I PEAK =
2 × IOUT  V OUT +V D 
DC V IN –V SW +V D 
(6)
where DC = duty cycle (0.72 for the ADP1073)
To avoid exceeding the maximum switch current when the
input voltage is at +9 V, an RLIM resistor should be specified.
Inductor Selection—Positive-to-Negative Converter
The configuration for a positive-to-negative converter using the
ADP1073 is shown in Figure 17. As with the step-up converter,
all of the output power for the inverting circuit must be supplied
by the inductor. The required inductor power is derived from
the formula:
P L = (|V OUT|+V D ) × ( IOUT )
The ADP1073 power switch does not saturate in positive-tonegative mode. The voltage drop across the switch can be
modeled as a 0.75 V base-emitter diode in series with a 0.65 Ω
resistor. When the switch turns on, inductor current will rise at
a rate determined by:
I L (t) =
where
PL
413 mW
=
= 21.7 µJ
f OSC 19 kHz
Using a standard inductor value of 330 µH, with 1 Ω dc resistance, will produce a peak switch current of:
I PEAK =
(7)
where tON = switch ON time (38 µs)
If the input voltage will vary (such as an application which must
operate from a battery), an RLIM resistor should be selected
from Figure 4. The RLIM resistor will keep switch current constant as the input voltage rises. Note that there are separate
RLIM values for step-up and step-down modes of operation.
For example, assume that +3.3 V at 150 mA is required from a
9 V battery with a 6 V end-of-life voltage. Deriving the peak
current from Equation 6 yields:
2 × 150 mA  3.3 + 0.5 
 6 – 1.5 + 0.5  = 317 mA
0.72


1
E L = (330 µH ) × (393 mA)2 = 25.5 µJ
2
The inductor energy of 25.5 µJ is greater than the PL/f OSC
requirement of 21.7 µJ, so the 330 µH inductor will work in this
application.
The input voltage varies between only 4.5 V and 5.5 V in this
example. Therefore, the peak current will not change enough to
require an RLIM resistor and the ILIM pin can be connected directly to VIN. Care should be taken, of course, to ensure that the
peak current does not exceed 800 mA.
6 –1.5 – 3.3
× 38 µs =144 µH
317 mA
Since 144 µH is not a standard value, the next lower standard
value of 100 µH would be specified.
REV. 0
–1.65 Ω × 38 µs 
4.5V – 0.75V 
1– e 330 µH  = 393 mA

0.65 Ω +1Ω 

Once the peak current is known, the inductor energy can be
calculated from Equation 9:
The peak current can than be inserted into Equation 7 to calculate the inductor value:
L=
R' = 0.65 Ω + RL(DC)
VL = VIN – 0.75 V
For example, assume that a –5 V output at 75 mA is to be generated from a +4.5 V to +5.5 V source. The power in the inductor is calculated from Equation 8:
The inductor value can now be calculated:
I PEAK =
(9)
During each switching cycle, the inductor must supply the following energy:
As previously mentioned, the switch voltage is higher in stepdown mode than in step-up mode. VSW is a function of switch
current and is therefore a function of VIN, L, time and VOUT.
For most applications, a VSW value of 1.5 V is recommended.
V IN(MIN ) –V SW –V OUT
× tON
I PEAK
–R't 
VL 
1– e L 

R' 

P L = (|−5V|+ 0.5V ) × (75 mA) = 413 mW
VSW = voltage drop across the switch
VD = diode drop (0.5 V for a 1N5818)
IOUT = output current
VOUT = the output voltage
VIN = the minimum input voltage
L=
(8)
–7–
ADP1073
Capacitor Selection
For optimum performance, the ADP1073’s output capacitor
must be carefully selected. Choosing an inappropriate capacitor
can result in low efficiency and/or high output ripple.
Ordinary aluminum electrolytic capacitors are inexpensive, but
often have poor Equivalent Series Resistance (ESR) and
Equivalent Series Inductance (ESL). Low ESR aluminum
capacitors, specifically designed for switch mode converter
applications, are also available, and these are a better choice
than general purpose devices. Even better performance can be
achieved with tantalum capacitors, although their cost is higher.
Very low values of ESR can be achieved by using OS-CON
capacitors (Sanyo Corporation, San Diego, CA). These devices
are fairly small, available with tape-and-reel packaging and have
very low ESR.
Figure 14. OS-CON Capacitor
If low output ripple is important, the user should consider using
the ADP3000. This device switches at 400 kHz, and the higher
switching frequency simplifies the design of the output filter.
Consult the ADP3000 data sheet for additional details.
The effects of capacitor selection on output ripple are demonstrated in Figures 12, 13 and 14. These figures show the output
of the same ADP1073 converter, which was evaluated with
three different output capacitors. In each case, the peak switch
current is 500 mA and the capacitor value is 100 µF. Figure 12
shows a Panasonic HF-series radial aluminum electrolytic. When
the switch turns off, the output voltage jumps by about 90 mV
and then decays as the inductor discharges into the capacitor.
The rise in voltage indicates an ESR of about 0.18 Ω. In
Figure 13, the aluminum electrolytic has been replaced by a
Sprague 593D-series device. In this case the output jumps
about 35 mV, which indicates an ESR of 0.07 Ω. Figure 14
shows an OS-CON SA series capacitor in the same circuit, and
ESR is only 0.02 Ω.
All potential current paths must be considered when analyzing
very low power applications, and this includes capacitor leakage
current. OS-CON capacitors have leakage in the 5 µA to 10 µA
range, which will reduce efficiency when the load is also in the
microampere range. Tantalum capacitors, with typical leakage
in the 1 µA to 5 µA range, are recommended for very low power
applications.
Diode Selection
In specifying a diode, consideration must be given to speed,
forward voltage drop and reverse leakage current. When the
ADP1073 switch turns off, the diode must turn on rapidly if
high efficiency is to be maintained. Schottky rectifiers, as well as
fast signal diodes such as the 1N4148, are appropriate. The
forward voltage of the diode represents power that is not delivered to the load, so VF must also be minimized. Again, Schottky
diodes are recommended. Leakage current is especially important in low current applications, where the leakage can be a
significant percentage of the total quiescent current.
For most circuits, the 1N5818 is a suitable companion to the
ADP1073. This diode has a VF of 0.5 V at 1 A, 4 µA to 10 µA
leakage and fast turn-on and turn-off times. A surface mount
version, the MBRS130T3, is also available. For applications
where the ADP1073 is “off” most of the time, such as when the
load is intermittent, a silicon diode may provide higher overall
efficiency due to lower leakage. For example, the 1N4933 has a
1 A capability, but with a leakage current of less than 1 µA. The
higher forward voltage of the 1N4933 reduces efficiency when
the ADP1073 delivers power, but the lower leakage may outweigh
the reduction in efficiency.
Figure 12. Aluminum Electrolytic
For switch currents of 100 mA or less, a Schottky diode such as
the BAT85 provides a VF of 0.8 V at 100 mA and leakage less
than 1 µA. A similar device, the BAT54, is available in an SOT-23
package. Even lower leakage, in the 1 nA to 5 nA range, can be
obtained with a 1N4148 signal diode.
General purpose rectifiers, such as the 1N4001, are not suitable
for ADP1073 circuits. These devices, which have turn-on times
of 10 µs or more, are too slow for switching power supply applications. Using such a diode “just to get started” will result in
wasted time and effort. Even if an ADP1073 circuit appears to
function with a 1N4001, the resulting performance will not be
indicative of the circuit performance when the correct diode is
used.
Figure 13. Tantalum Electrolytic
–8–
REV. 0
ADP1073
Circuit Operation, Step-Up (Boost) Mode
VIN
In boost mode, the ADP1073 produces an output voltage that is
higher than the input voltage. For example, +5 V can be
derived from one alkaline cell (+1.5 V), or +12 V can be
generated from a +5 V logic power supply.
Figure 15 shows an ADP1073 configured for step-up operation.
The collector of the internal power switch is connected to the
output side of the inductor, while the emitter is connected to
GND. When the switch turns on, Pin SW1 is pulled near ground.
This action forces a voltage across L1 equal to VIN – VCE(SAT) and
current begins to flow through L1. This current reaches a final
value (ignoring second-order effects) of:
I PEAK ≅
V IN –V CE(SAT )
× 38 µs
L
where 38 µs is the ADP1073 switch’s “on” time.
L1
D1
VIN
VOUT
R3*
1
2
ILIM
VIN
ADP1073
GND
SW2
5
4
R1
SW1 3
C1
FB 8
R2
C2
R3
220V
2
1
ILIM
 R1
V OUT = 212 mV × 1+ 
 R2
The circuit of Figure 15 shows a direct current path from VIN to
VOUT, via the inductor and D1. Therefore, the boost converter
is not protected if the output is short circuited to ground.
Circuit Operation, Step-Down (Buck) Mode)
The ADP1073’s step-down mode is used to produce an output
voltage that is lower than the input voltage. For example, the
output of four NiCd cells (+4.8 V) can be converted to a +3.3 V
logic supply.
A typical configuration for step-down operation of the ADP1073
is shown in Figure 16. In this case, the collector of the internal
power switch is connected to VIN and the emitter drives the
inductor. When the switch turns on, SW2 is pulled up toward
VIN. This forces a voltage across L1 equal to (VIN – VCE ) – VOUT,
and causes current to flow in L1. This current reaches a final
value of:
I PEAK
V –V CE –V OUT
≅ IN
× 38 µs
L
where 38 µs is the ADP1073 switch’s “on” time.
REV. 0
L1
VOUT
SW2 4
GND
R1
5
D1
1N5818
C1
R2
Figure 16. Step-Down Mode Operation
When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes and the switch side of the
inductor is driven below ground. Schottky diode D1 then turns
on and current flows into the load. Notice that the Absolute
Maximum Rating for the ADP1073’s SW2 pin is 0.5 V below
ground. To avoid exceeding this limit, D1 must be a Schottky
diode. Using a silicon diode in this application will generate
forward voltages above 0.5 V, which will cause potentially damaging power dissipation within the ADP1073.
The output voltage of the buck regulator is fed back to the
ADP1073’s FB pin by resistors R1 and R2. When the voltage at
pin FB falls below 212 mV, the internal power switch turns
“on” again and the cycle repeats. The output voltage is set by
the formula:
 R1
V OUT = 212 mV × 1+ 
 R2
Figure 15. Step-Up Mode Operation
The output voltage is fed back to the ADP1073 via resistors R1
and R2. When the voltage at pin FB falls below 212 mV, SW1
turns “on” again and the cycle repeats. The output voltage is
therefore set by the formula:
FB 8
ADP1073
*OPTIONAL
When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, current begins to flow
through D1 into the load and the output voltage is driven above
the input voltage.
3
VIN SW1
The output voltage should be limited to 6.2 V or less when
using the ADP1073 in step-down mode.
If the input voltage to the ADP1073 varies over a wide range, a
current limiting resistor at Pin 1 may be required. If a particular
circuit requires high peak inductor current with minimum input
supply voltage the peak current may exceed the switch maximum
rating and/or saturate the inductor when the supply voltage is at the
maximum value. See the Limiting the Switch Current section of
this data sheet for specific recommendations.
Positive-to-Negative Conversion
The ADP1073 can convert a positive input voltage to a negative
output voltage, as shown in Figure 17. This circuit is essentially
identical to the step-down application of Figure 16, except that
the “output” side of the inductor is connected to power ground.
When the ADP1073’s internal power switch turns off, current
flowing in the inductor forces the output (–VOUT) to a negative
1VIN
R3
ILIM
C2
VIN
SW1
ADP1073
FB
L1
SW2
GND
R2
D1
1N5818
C1
R1
2VOUT
Figure 17. A Positive-to-Negative Converter
potential. The ADP1073 will continue to turn the switch on
until its FB pin is 212 mV above its GND pin, so the output
voltage is determined by the formula:
–9–
ADP1073
 R1
V OUT = 212 mV × 1+ 
 R2
The design criteria for the step-down application also apply to
the positive-to-negative converter. The output voltage should be
limited to |6.2 V| and D1 must be a Schottky diode to prevent
excessive power dissipation in the ADP1073.
Negative-to-Positive Conversion
The circuit of Figure 18 converts a negative input voltage to a
positive output voltage. Operation of this circuit configuration is
similar to the step-up topology of Figure 16, except that the current through feedback resistor R1 is level-shifted below ground
by a PNP transistor. The voltage across R1 is (VOUT – VBE(Q1)).
However, diode D2 level-shifts the base of Q1 about 0.6 V below
ground, thereby cancelling the VBE of Q1. The addition of D2
also reduces the circuit’s output voltage sensitivity to temperature, which would otherwise be dominated by the –2 mV/°C VBE
contribution of Q1. The output voltage for this circuit is determined by the formula:
the switch turns on for the next cycle, the inductor current
begins to ramp up from the residual level. If the switch ON time
remains constant, the inductor current will increase to a high
level (see Figure 19). This increases output ripple and can
require a larger inductor and capacitor. By controlling switch
current with the ILIM resistor, output ripple current can be maintained at the design values. Figure 20 illustrates the action of the
ILIM circuit.
 R1
V OUT = 212 mV × 1+ 
 R2
Figure 19. (I LIM Operation, RLIM = 0 Ω)
Unlike the positive step-up converter, the negative-to-positive
converter’s output voltage can be either higher or lower than the
input voltage.
D1
1N5818
L1
POSITIVE
OUTPUT
RLIM
R1
ILIM
C2
VIN
SW1
ADP1073
FB
Q1
2N3906
CL
D2
1N4148
AO SET GND SW2
R2
NEGATIVE
INPUT
NC
10kV
NC
Figure 18. A Negative-to-Positive Converter
Figure 20. (I LIM Operation, RLIM = 240 Ω)
Limiting the Switch Current
The ADP1073’s RLIM pin permits the switch current to be limited with a single resistor. This current limiting action occurs on
a pulse by pulse basis. This feature allows the input voltage to
vary over a wide range without saturating the inductor or exceeding the maximum switch rating. For example, a particular
design may require peak switch current of 800 mA with a 2.0 V
input. If VIN rises to 4 V, however, the switch current will exceed
1.6 A. The ADP1073 limits switch current to 1.5 A and thereby
protects the switch, but the output ripple will increase. Selecting
the proper resistor will limit the switch current to 800 mA, even
if VIN increases. The relationship between RLIM and maximum
switch current is shown in Figure 4.
The ILIM feature is also valuable for controlling inductor current
when the ADP1073 goes into continuous conduction mode. This
occurs in the step-up mode when the following condition is met:
V OUT +V DIODE
1
<
V IN –V SW
1– DC
The internal structure of the ILIM circuit is shown in Figure 21.
Q1 is the ADP1073’s internal power switch, which is paralleled
by sense transistor Q2. The relative sizes of Q1 and Q2 are
scaled so that IQ2 is 0.5% of IQ1. Current flows to Q2 through an
internal 80 Ω resistor and through the RLIM resistor. These two
resistors parallel the base-emitter junction of the oscillatordisable transistor, Q3. When the voltage across R1 and RLIM
exceeds 0.6 V, Q3 turns on and terminates the output pulse. If
only the 80 Ω internal resistor is used (i.e., the ILIM pin is connected directly to VIN), the maximum switch current will be
1.5 A. Figure 4 gives RLIM values for lower current-limit values.
The delay through the current limiting circuit is approximately
2 µs. If the switch ON time is reduced to less than 5 µs, accuracy of the current trip point is reduced. Attempting to program
a switch ON time of 2 µs or less will produce spurious responses
in the switch ON time. However, the ADP1073 will still provide
a properly regulated output voltage.
where DC is the ADP1073’s duty cycle.
When this relationship exists, the inductor current does not go
all the way to zero during the time that the switch is OFF. When
–10–
REV. 0
ADP1073
RLIM
(EXTERNAL)
VIN
ADP1073
+5V
ILIM
Q3
VBAT
SW1
Q1
Programming the Gain Block
The gain block of the ADP1073 can be used as a low battery
detector, error amplifier or linear post regulator. The gain block
consists of an op amp with PNP inputs and an open-collector
NPN output. The inverting input is internally connected to the
ADP1073’s 212 mV reference, while the noninverting input is
available at the SET pin. The NPN output transistor will sink
about 100 µA.
Figure 22a shows the gain block configured as a low-battery
monitor. Resistors R1 and R2 should be set to high values to
reduce quiescent current, but not so high that bias current in
the SET input causes large errors. A value of 100 kΩ for R2 is a
good compromise. The value for R1 is then calculated from the
formula:
V LOBATT − 212 mV
212 mV
R2
where VLOBATT is the desired low battery trip point. Since the
gain block output is an open-collector NPN, a pull-up resistor
should be connected to the positive logic power supply.
+5V
ADP1073
100kV
VIN
212mV
REF
SET
VBAT
TO
PROCESSOR
R2
33kV
R3
1.6MV
Figure 21. Current Limit Operation
R1
47kV
AO
GND
Q2
SW2
R1 =
VIN
212mV
REF
SET
DRIVER
OSCILLATOR
ADP1073
R1
R1
80V
(INTERNAL)
AO
TO
PROCESSOR
GND
R2
R1 = R2
VLB
–1)
(212mV
VLB = BATTERY TRIP POINT
Figure 22b. Adding Hysteresis to the Low Battery Detector
The circuit of Figure 22a may produce multiple pulses when
approaching the trip point, due to noise coupled into the SET
input. To prevent multiple interrupts to the digital logic, hysteresis can be added to the circuit (Figure 22b). Resistor RHYS, with
a value of 1 MΩ to 10 MΩ, provides the hysteresis. The addition of RHYS will change the trip point slightly, so the new value
for R1 will be:
R1=
V LOBATT – 212 mV
 212 mV  V L – 212 mV 
 R2  –  R + R


  L
HYS 
where VL is the logic power supply voltage, RL is the pull-up
resistor and R HYS creates the hysteresis.
The gain block can also be used as a control element to reduce
output ripple. The ADP3000 is normally recommended for lowripple applications, but its minimum input voltage is 2 V. The
gain-block technique using the ADP1073 can be useful for stepup converters operating down to 1 V.
A step-up converter using this technique is shown in Figure 23.
This configuration uses the gain block to sense the output voltage and control the comparator. The result is that the comparator hysteresis is reduced by the open loop gain of the gain block.
Output ripple can be reduced to only a few millivolts with this
technique, versus a typical value of 150 mV for a +5 V converter
using just the comparator. For best results, a large output
capacitor (1000 µF or more) should be specified. This technique can also be used for step-down or inverting applications,
but the ADP3000 is usually a more appropriate choice. See the
ADP3000 data sheet for further details.
Figure 22a. Setting the Low Battery Detector Trip Point
D1
L1
VOUT
R3
680kV
VBAT
AO
ILIM
SW1
ADP1073
FB
VOUT =
VIN
GND
SW2
R1
C1
SET
R2
+1) (212mV)
( R1
R2
Figure 23. Using the Gain Block to Reduce Output Ripple
REV. 0
–11–
ADP1073 –Typical Application Circuits
L1*
120mH
1N5818
1MV
12V OUTPUT
5mA AT VBATTERY = 1.0V
12mA AT VBATTERY = 1.5V
+12V
1mF*
ILIM
100V
1.5 VOLT
CELL
ADP1073
CIRCUIT
V2
V1
47mF
Figure 28. 1.5 V to 12 V Step-Up Converter
1N5818
L1*
68mH
1.00MV**
VIN
SW1
ADP1073
FB
GND
SW2
*L1 = GOWANDA GA10-123k
OR CADDELL-BURNS 7300-14
220V
ILIM
ADP1073-12 SENSE
GND
Figure 24. Test Circuit Measures No Load Quiescent
Current of ADP1073 Converter
1.5 VOLT
CELL
SW1
100mF
V2 – V1
VSET
IIN =
100V
*NON-POLARIZED
L1*
120mH
VIN
9V OUTPUT
5mA AT VBATTERY = 1.00V
12mA AT VBATTERY = 1.5V
SW2
23.3kV**
12V OUTPUT
25mA AT VBATTERY = 2.0V
51V
ILIM
TWO
1.5 VOLT
CELLS
47mF
1N5818
VIN
SW1
ADP1073-12 SENSE
GND
47mF
SW2
*L1
= GOWANDA GA10-123k
OR CADDELL-BURNS 7300-14
**1% METAL FILM
*L1 = GOWANDA GA10-682k
OR CADDELL-BURNS 7300-11
Figure 25. 1.5 V to 9 V Step-Up Converter
L1*
68mH
L1*
68mH
ILIM
51V
5V OUTPUT
100mA
AT VBATTERY = 2.0V
VIN
1.00MV**
100mF
ADP1073-5 SENSE
ILIM
TWO
1.5 VOLT
CELLS
SW1
GND
1N5818
1N5818
56V
TWO
1.5 VOLT
CELLS
Figure 29. 3 V to 12 V Step-Up Converter
VIN
SW1
ADP1073
GND
15V OUTPUT
20mA AT VBATTERY = 2.0V
47mF
FB
SW2
14.3kV**
SW2
*L1
= GOWANDA GA10-682k
OR CADDELL-BURNS 7300-11
**1% METAL FILM
*L1 = GOWANDA GA10-682k
OR CADDELL-BURNS 7300-11
Figure 30. 3 V to 15 V Step-Up Converter
Figure 26. 3 V to 5 V Step-Up Converter
L1*
120mH
1N5818
220V
536kV**
ILIM
1.5 VOLT
CELL
VIN
GND
3V OUTPUT
15mA AT VBATTERY = 1.00V
SW1
ADP1073
ILIM
100mF
VIN
GND
40.2kV**
*L1
1N5818
SW1
ADP1073
100mF
FB
SW2
L1*
150mH
5 VIN
1MV**
100mF
FB
SW2
15V OUTPUT
100mA AT 4.5 VIN
14.3kV**
*L1
= GOWANDA GA10-123k
OR CADDELL-BURNS 7300-14
**1% METAL FILM
= GOWANDA GA10-153k
OR CADDELL-BURNS 7200-15
**1% METAL FILM
Figure 27. 1.5 V to 3 V Step-Up Converter
Figure 31. 5 V to 15 V Step-Up Converter
–12–
REV. 0
ADP1073
L1*
150mH
5 VIN
3V OUTPUT
1N5818
220V
536kV
ILIM
100mF
VIN
12V OUTPUT
100mA AT 4.5 VIN
SW1
ADP1073-12 SENSE
GND
ILIM
9 VOLT
BATTERY
VIN
ADP1073 FB
GND
100mF
SW1
SW2
SW2
40.2kV
L1*
100mH
*L1
= GOWANDA GA20-153k
OR CADDELL-BURNS 7200-15
1N5818
Figure 32. 5 V to 12 V Step-Up Converter
L1*
82mH
100mF
*L1 = GOWANDA GA10-103k
OR CADDELL-BURNS 7300-13
Figure 35. 9 V to 3 V Step-Down Converter
1N5818
5V OUTPUT
ILIM
1.5 VOLT
CELL
VIN
ADP1073
GND
220V
909kV**
SW1
FB
100mF
ILIM
SW2
9 VOLT
BATTERY
40.2kV**
1N4148
SHUTDOWN
VIN
SW1
ADP1073-5
GND
SENSE
SW2
L1*
100mH
OPERATE
74C04
5V OUTPUT
*L1
= GOWANDA GA10-822k
OR CADDELL-BURNS 7200-12
**1% METAL FILM
1N5818
*L1 = GOWANDA GA10-103k
OR CADDELL-BURNS 7300-13
Figure 33. 1.5 V to 5 V Step-Up Converter with Logic
Shutdown
Figure 36. 9 V to 5 V Step-Down Converter
L1*
82mH
100mF
L1*
47mH
1N5818
1N5818
5V OUTPUT
25mA
5V OUTPUT
442kV**
1.5 VOLT
CELL
ILIM VIN
7
SET
SW1
SENSE
ADP1073-5
100kV**
GND
SW2
2N3906
100kV
AO
100mF
LO BATT
GOES LOW
AT VBATTERY = 1.15V
2.2V
1.5 VOLT
CELL
56V
100mF
ILIM
VIN
SW1
ADP1073-5 SENSE
*L1
GND
= GOWANDA GA10-822k
OR CADDELL-BURNS 7300-12
**1% METAL FILM
SW2
*L1 = GOWANDA GA10-472k
OR CADDELL-BURNS 7300-14
MINIMUM START-UP VOLTAGE = 1.1V
Figure 34. 1.5 V to 5 V Step-Up Converter with Low
Battery Detector
Figure 37. 1.5 V to 5 V Bootstrapped Step-Up Converter
REV. 0
–13–
ADP1073
L1*
82mH
L1*
470mH
5V TO MEMORY
4.5V WHEN MAIN
SUPPLY OPEN
5V
MAIN SUPPLY
680kV
1N5818
ILIM
1.5 VOLT
CELL
806kV**
ILIM VIN
1.5 VOLT
CELL
SW1
ADP1073
GND
FB
SW2
40.2kV**
*L1
= GOWANDA GA10-473k
OR CADDELL-BURNS 7300-21
**1% METAL FILM
EFFICIENCY = 83% AT 5mA LOAD
51V
100kV
1MV**
ILIM
VIN
AO
2.2MV
100mF
OS-CON
SW2
40.2kV**
Figure 41. 1.5 V to 5 V Very Low Noise Step-Up Converter
L1*
68mH 1N5818
100kV
ADP1073 SET
GND
Figure 38. Memory Backup Supply
2N3906
SW1
AO
= GOWANDA GA10-822k
OR CADDELL-BURNS 7300-12
**1% METAL FILM
***OPTIONAL
3 VOLT
CELL
909kV**
VIN
5V OUTPUT
5mA AT
VBATTERY = 1.00V
10mV p-p RIPPLE
FB
100mF***
*L1
909kV**
1N5818
5V OUTPUT
100mA
LOCKOUT
AT 1.6V
6.5V
TO 12V
680kV
ILIM
VIN
SW1
L1*
47mH
FB
5VOUT
90mA
AT 6.5VIN
ADP1073 SET
SW1
AO
ADP1073
100mF
GND
1N5818
SW2
100mF
OS-CON
900kV**
SET
FB
GND SW2
40.3kV**
100kV
*L1
= GOWANDA GA10-472k
OR CADDELL-BURNS 7300-09
**1% METAL FILM
EFFICIENCY = 80%
IQ = 130µA
OUTPUT RIPPLE = 100mV p-p
*L1
= GOWANDA GA10-682k
OR CADDELL-BURNS 7300-11
**1% METAL FILM
Figure 42. 9 V to 5 V Reduced Noise Step-Down Converter
Figure 39. 3 V to 5 V Step-Up Converter with Undervoltage
Lockout
L1*
82mH
ILIM
1N5818
1000mF
10V
OUTPUT
+6V, 1A
AT VIN = 3V
5V OUTPUT
20mV p-p RIPPLE
909kV**
VIN
560kV
SW1
FB
ADP1073 SET
AO
L1*
25mH
INPUT
3V TO 6V
(2 LITHIUM CELLS)
680kV
1.5 VOLT
CELL
40.2kV**
GND
SW2
ILIM VIN
FB
100mF
OS-CON
1N5820
SW1
ADP1073
549kV**
40.2kV**
2200mF
10V
AO
SET
GND SW2
*L1
= GOWANDA GA10-822k
OR CADDELL-BURNS 7300-12
**1% METAL FILM
1N5818
20kV**
51V
MTP3055EL
Figure 40. 1.5 V to 5 V Low Noise Step-Up Converter
2N3906
5.1kV
*L1
= COILTRONICS
CTX25-5-52
**1% METAL FILM
Figure 43. 3 V to 6 V @ 1 A Step-Up Converter
–14–
REV. 0
ADP1073
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
8-Lead Plastic DIP
(N-8)
0.430 (10.92)
0.348 (8.84)
8
5
0.280 (7.11)
0.240 (6.10)
1
4
0.060 (1.52)
0.015 (0.38)
PIN 1
0.210 (5.33)
MAX
0.325 (8.25)
0.300 (7.62)
0.195 (4.95)
0.115 (2.93)
0.130
(3.30)
MIN
0.160 (4.06)
0.115 (2.93)
0.022 (0.558) 0.100 0.070 (1.77)
0.014 (0.356) (2.54) 0.045 (1.15)
BSC
0.015 (0.381)
0.008 (0.204)
SEATING
PLANE
8-Lead Small Outline Package
(SO-8)
0.1968 (5.00)
0.1890 (4.80)
0.1574 (4.00)
0.1497 (3.80)
PIN 1
0.0098 (0.25)
0.0040 (0.10)
SEATING
PLANE
REV. 0
8
5
1
4
0.2440 (6.20)
0.2284 (5.80)
0.0688 (1.75)
0.0532 (1.35)
0.0500 0.0192 (0.49)
(1.27) 0.0138 (0.35)
BSC
0.0196 (0.50)
x 45°
0.0099 (0.25)
0.0098 (0.25)
0.0075 (0.19)
–15–
8°
0°
0.0500 (1.27)
0.0160 (0.41)
–16–
PRINTED IN U.S.A.
C2965–8–10/97