AD AD8302-EVAL Lf-2.7 ghz rf/if gain and phase detector Datasheet

a
FEATURES
Measures Gain/Loss and Phase up to 2.7 GHz
Dual Demodulating Log Amps and Phase Detector
Input Range –60 dBm to 0 dBm in a 50 ⍀ System
Accurate Gain Measurement Scaling (30 mV/dB)
Typical Nonlinearity < 0.5 dB
Accurate Phase Measurement Scaling (10 mV/Degree)
Typical Nonlinearity < 1 Degree
Measurement/Controller/Level Comparator Modes
Operates from Supply Voltages of 2.7 V–5.5 V
Stable 1.8 V Reference Voltage Output
Small Signal Envelope Bandwidth from DC to 30 MHz
APPLICATIONS
RF/IF PA Linearization
Precise RF Power Control
Remote System Monitoring and Diagnostics
Return Loss/VSWR Measurements
Log Ratio Function for AC Signals
PRODUCT DESCRIPTION
The AD8302 is a fully integrated system for measuring gain/loss
and phase in numerous receive, transmit, and instrumentation
applications. It requires few external components and a single
supply of 2.7 V–5.5 V. The ac-coupled input signals can range
from –60 dBm to 0 dBm in a 50 Ω system, from low frequencies
up to 2.7 GHz. The outputs provide an accurate measurement
of either gain or loss over a ± 30 dB range scaled to 30 mV/dB,
and of phase over a 0°–180° range scaled to 10 mV/degree.
Both subsystems have an output bandwidth of 30 MHz, which
may optionally be reduced by the addition of external filter
capacitors. The AD8302 can be used in controller mode to
force the gain and phase of a signal chain toward predetermined
setpoints.
The AD8302 comprises a closely matched pair of demodulating
logarithmic amplifiers, each having a 60 dB measurement range.
By taking the difference of their outputs, a measurement of
the magnitude ratio or gain between the two input signals is
available. These signals may even be at different frequencies,
allowing the measurement of conversion gain or loss. The AD8302
may be used to determine absolute signal level by applying the
unknown signal to one input and a calibrated ac reference signal
to the other. With the output stage feedback connection disabled, a comparator may be realized, using the setpoint pins
MSET and PSET to program the thresholds.
LF–2.7 GHz
RF/IF Gain and Phase Detector
AD8302
FUNCTIONAL BLOCK DIAGRAM
AD8302
VIDEO OUTPUT – A
INPA
OFSA
MFLT
+
+
–
VMAG
–
60dB LOG AMPS
(7 DETECTORS)
MSET
PHASE
DETECTOR
COMM
PSET
OFSB
INPB
–
60dB LOG AMPS
(7 DETECTORS)
VPHS
+
PFLT
VIDEO OUTPUT – B
1.8V
VPOS
BIAS
x3
VREF
The signal inputs are single-ended, allowing them to be matched
and connected directly to a directional coupler. Their input
impedance is nominally 3 kΩ at low frequencies.
The AD8302 includes a phase detector of the multiplier type,
but with precise phase balance driven by the fully limited signals
appearing at the outputs of the two logarithmic amplifiers.
Thus, the phase accuracy measurement is independent of signal
level over a wide range.
The phase and gain output voltages are simultaneously available
at loadable ground referenced outputs over the standard output
range of 0 V to 1.8 V. The output drivers can source or sink up
to 8 mA. A loadable, stable reference voltage of 1.8 V is available for precise repositioning of the output range by the user.
In controller applications, the connection between the gain
output pin VMAG and the setpoint control pin MSET is broken.
The desired setpoint is presented to MSET and the VMAG
control signal drives an appropriate external variable gain device.
Likewise, the feedback path between the phase output pin VPHS
and its setpoint control pin PSET may be broken to allow
operation as a phase controller.
The AD8302 is fabricated on Analog Devices’ proprietary, high
performance 25 GHz SOI complementary bipolar IC process. It is
available in a 14-lead TSSOP package and operates over a –40°C
to +85°C temperature range. An evaluation board is available.
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329–4700
www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 2002
AD8302–SPECIFICATIONS
(TA = 25ⴗC, VS = 5 V, VMAG shorted to MSET, VPHS shorted to PSET, 52.3 ⍀ shunt
resistors connected to INPA and INPB, for Phase measurement PINPA = PINPB, unless otherwise noted.)
Parameter
Conditions
OVERALL FUNCTION
Input Frequency Range
Gain Measurement Range
Phase Measurement Range
Reference Voltage Output
PIN at INPA, PIN at INPB = –30 dBm
φIN at INPA > φIN at INPB
Pin VREF, –40°C ≤ TA ≤ +85°C
1.72
Pins INPA and INPB
To AC Ground, f ≤ 500 MHz
AC-Coupled (0 dBV = 1 V rms)
re: 50 Ω
–73
–60
INPUT INTERFACE
Input Simplified Equivalent Circuit
Input Voltage Range
Min
>0
1.88
mV
V
mV
mA
MHz
V/µs
Any 20 dB Change, 10%–90%
Any 20 dB Change, 90%–10%
Full-Scale 60 dB Change, to 1% Settling
50
60
300
ns
ns
ns
30
1.8
900
8
25
30
40
500
mV
V
mV
mA
V/µs
MHz
ns
ns
58
55
42
29
dB
dB
dB
mV/dB
0.25
dB
0.25
0.2
dB
dB
145
143
10
Degree
Degree
mV/Degree
0.7
Degree
0.7
Degree
MAGNITUDE OUTPUT
± 1 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
± 0.5 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
± 0.2 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
From Linear Regression
Deviation from Output at 25°C
–40°C ≤ TA ≤ +85°C, PINPA = PINPB = –30 dBm
Deviation from Best Fit Curve at 25°C
–40°C ≤ TA ≤ +85°C, PINPA = ± 25 dB, PINPB = –30 dBm
PINPA = PINPB = –5 dBm to –50 dBm
Slope (Absolute Value)
Deviation vs. Temperature
MHz
dB
Degree
V
30
1.8
900
8
30
25
100 MHz
Dynamic Range
Dynamic Range
2700
Pin VMAG
20 × Log (VINPA/VINPB) = –30 dB
20 × Log (VINPA/VINPB) = +30 dB
VINPA = VINPB
Source/Sink
Pin MFLT Open
40 dB Change, Load 20 pF储10 kΩ
Pin VPHS
Phase Difference 180 Degrees
Phase Difference 0 Degrees
When φINPA = φINPB ± 90°
Source/Sink
Gain Measurement Balance
Unit
–43
–30
PHASE OUTPUT
Output Voltage Minimum
Output Voltage Maximum
Phase Center Point
Output Current Drive
Slew Rate
Small Signal Envelope Bandwidth
Response Time
Slope
Deviation vs. Temperature
± 30
± 90
1.8
Max
kΩ储pF
dBV
dBm
dBV
dBm
Center of Input Dynamic Range
MAGNITUDE OUTPUT
Output Voltage Minimum
Output Voltage Maximum
Center Point of Output (MCP)
Output Current
Small Signal Envelope Bandwidth
Slew Rate
Response Time
Rise Time
Fall Time
Settling Time
Typ
Any 15 Degree Change, 10%–90%
120 Degree Change CFILT = 1 pF, to 1% Settling
PHASE OUTPUT
Less than ± 1 Degree Deviation from Best Fit Line
Less than 10% Deviation in Instantaneous Slope
From Linear Regression about –90° or +90°
Deviation from Output at 25°C
–40°C ≤ TA ≤ +85°C, Delta Phase = 90 Degrees
Deviation from Best Fit Curve at 25°C
–40°C ≤ TA ≤ +85°C, Delta Phase = ± 30 Degrees
–2–
3储2
–13
0
REV. A
AD8302
Parameter
Conditions
900 MHz
Dynamic Range
MAGNITUDE OUTPUT
± 1 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
± 0.5 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
± 0.2 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
From Linear Regression
Deviation from Output at 25°C
–40°C ≤ TA ≤ +85°C, PINPA = PINPB = –30 dBm
Deviation from Best Fit Curve at 25°C
–40°C ≤ TA ≤ +85°C, PINPA = ± 25 dB, PINPB = –30 dBm
PINPA = PINPB = –5 dBm to –50 dBm
Slope
Deviation vs. Temperature
Gain Measurement Balance
Dynamic Range
Slope (Absolute Value)
Deviation
Phase Measurement Balance
1900 MHz
Dynamic Range
Slope
Deviation vs. Temperature
Gain Measurement Balance
Dynamic Range
Slope (Absolute Value)
Deviation
Phase Measurement Balance
2200 MHz
Dynamic Range
Slope
Deviation vs. Temperature
Gain Measurement Balance
Dynamic Range
Slope (Absolute Value)
Deviation
Min
PHASE OUTPUT
Less than ± 1 Degree Deviation from Best Fit Line
Less than 10% Deviation in Instantaneous Slope
From Linear Regression about –90° or +90°
Linear Deviation from Best Fit Curve at 25°C
–40°C ≤ TA ≤ +85°C, Delta Phase = 90 Degrees
–40°C ≤ TA ≤ +85°C, Delta Phase = ± 30 Degrees
Phase @ INPA = Phase @ INPB, PIN = –5 dBm to –50 dBm
MAGNITUDE OUTPUT
± 1 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
± 0.5 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
± 0.2 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
From Linear Regression
Deviation from Output at 25°C
–40°C ≤ TA ≤ +85°C, PINPA = PINPB = –30 dBm
Deviation from Best Fit Curve at 25°C
–40°C ≤ TA ≤ +85°C, PINPA = ±25 dB, PINPB = –30 dBm
PINPA = PINPB = –5 dBm to –50 dBm
PHASE OUTPUT
Less than ± 1 Degree Deviation from Best Fit Line
Less than 10% Deviation in Instantaneous Slope
From Linear Regression about –90° or +90°
Linear Deviation from Best Fit Curve at 25°C
–40°C ≤ TA ≤ +85°C, Delta Phase = 90 Degrees
–40°C ≤ TA ≤ +85°C, Delta Phase = ± 30 Degrees
Phase @ INPA = Phase @ INPB, PIN = –5 dBm to –50 dBm
MAGNITUDE OUTPUT
± 1 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
± 0.5 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
± 0.2 dB Linearity PREF = –30 dBm (VREF = –43 dBV)
From Linear Regression
Deviation from Output at 25°C
–40°C ≤ TA ≤ +85°C, PINPA = PINPB = –30 dBm
Deviation from Best Fit Curve at 25°C
–40°C ≤ TA ≤ +85°C, PINPA = ± 25 dB, PINPB = –30 dBm
PINPA = PINPB = –5 dBm to –50 dBm
PHASE OUTPUT
Less than ± 1 Degree Deviation from Best Fit Line
Less than 10% Deviation in Instantaneous Slope
From Linear Regression about –90° or +90°
Linear Deviation from Best Fit Curve at 25°C
–40°C ≤ TA ≤ +85°C, Delta Phase = 90 Degrees
–40°C ≤ TA ≤ +85°C, Delta Phase = ± 30 Degrees
REFERENCE VOLTAGE
Output Voltage
PSRR
Output Current
Pin VREF
Load = 2 kΩ
VS = 2.7 V to 5.5 V
Source/Sink (Less than 1% Change)
POWER SUPPLY
Supply
Operating Current (Quiescent)
Pin VPOS
VS = 5 V
–40°C ≤ TA ≤ +85°C
Specifications subject to change without notice.
REV. A
–3–
Typ
Max
Unit
58
54
42
28.7
dB
dB
dB
mV/dB
0.25
dB
0.25
0.2
dB
dB
143
143
10.1
Degree
Degree
mV/Degree
0.75
0.75
0.8
Degree
Degree
Degree
57
54
42
27.5
dB
dB
dB
mV/dB
0.27
dB
0.33
0.2
dB
dB
128
120
10.2
Degree
Degree
mV/Degree
0.8
0.8
1
Degree
Degree
Degree
53
51
38
27.5
dB
dB
dB
mV/dB
0.28
dB
0.4
0.2
dB
dB
115
110
10
Degree
Degree
mV/Degree
0.85
0.9
Degree
Degree
1.7
1.8
0.25
5
1.9
V
mV/V
mA
2.7
5.0
19
21
5.5
25
27
V
mA
mA
AD8302
ABSOLUTE MAXIMUM RATINGS 1
PIN CONFIGURATION
Supply Voltage VS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5 V
PSET, MSET Voltage . . . . . . . . . . . . . . . . . . . . . . VS + 0.3 V
INPA, INPB Maximum Input . . . . . . . . . . . . . . . . . . –3 dBV
Equivalent Power Re. 50 Ω . . . . . . . . . . . . . . . . . . 10 dBm
θJA2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C/W
Maximum Junction Temperature . . . . . . . . . . . . . . . . 125°C
Operating Temperature Range . . . . . . . . . . . –40°C to +85°C
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . . 300°C
COMM 1
14
MFLT
INPA 2
13
VMAG
OFSA 3
12
MSET
AD8302
TOP VIEW 11 VREF
(Not to Scale)
10 PSET
OFSB 5
VPOS 4
INPB 6
9
VPHS
COMM 7
8
PFLT
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
JEDEC 1S Standard (2-layer) board data.
PIN FUNCTION DESCRIPTIONS
Equivalent
Circuit
Pin No.
Mnemonic
Function
1, 7
2
3
COMM
INPA
OFSA
4
5
VPOS
OFSB
6
8
9
INPB
PFLT
VPHS
10
PSET
11
12
VREF
MSET
13
VMAG
Device Common. Connect to low impedance ground.
High Input Impedance to Channel A. Must be ac-coupled.
A capacitor to ground at this pin sets the offset compensation filter corner
and provides input decoupling.
Voltage Supply (VS), 2.7 V to 5.5 V
A capacitor to ground at this pin sets the offset compensation filter corner
and provides input decoupling.
Input to Channel B. Same structure as INPA.
Low Pass Filter Terminal for the Phase Output
Single-Ended Output Proportional to the Phase Difference between INPA
and INPB.
Feedback Pin for Scaling of VPHS Output Voltage in Measurement Mode.
Apply a setpoint voltage for controller mode.
Internally Generated Reference Voltage (1.8 V Nominal)
Feedback Pin for Scaling of VMAG Output Voltage Measurement Mode.
Accepts a set point voltage in controller mode.
Single-Ended Output. Output voltage proportional to the decibel ratio
of signals applied to INPA and INPB.
Low Pass Filter Terminal for the Magnitude Output
14
MFLT
Circuit A
Circuit A
Circuit A
Circuit A
Circuit E
Circuit B
Circuit D
Circuit C
Circuit D
Circuit B
Circuit E
ORDERING GUIDE
Model
Temperature Range
Package Description
AD8302ARU
AD8302ARU-REEL
AD8302ARU-REEL7
AD8302-EVAL
–40°C to +85°C
Tube, 14-Lead TSSOP
13" Tape and Reel
7" Tape and Reel
Evaluation Board
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the AD8302 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.
–4–
Package
Option
RU-14
WARNING!
ESD SENSITIVE DEVICE
REV. A
AD8302
VPOS
VPOS
100mV
4k⍀
INPA(INPB)
4k⍀
OFSA(OFSB)
+
ON TO
LOG-AMP
–
25⍀
750⍀
2k⍀
10pF
VMAG
(VPHS)
CLASS A-B
CONTROL
COMM
COMM
Circuit A
Circuit B
VPOS
VPOS
VPOS
VREF
10k⍀
MSET
(PSET)
MFLT
(PFLT)
10k⍀
10k⍀
5k⍀
COMM
Circuit C
1.5pF
ACTIVE LOADS
COMM
COMM
Circuit D
Figure 1. Equivalent Circuits
REV. A
–5–
Circuit E
AD8302–Typical Performance Characteristics
1.80
3.0
1.65
2.5
1.50
2.0
1.35
1.5
900
1.8
100
1.6
1.4
1.20
1.2
1900
0.8
0.6
0.4
0.2
0
–30 –25 –20 –15 –10 –5
0
5
10
MAGNITUDE RATIO – dB
15
20
25
TPC 1. Magnitude Output (VMAG) vs. Input Level Ratio
(Gain) VINPA/VINPB, Frequencies 100 MHz, 900 MHz,
1900 MHz, 2200 MHz, 2700 MHz, 25 ⴗC, PINPB = –30 dBm,
(Re: 50 Ω)
2.0
–0.5
0.60
–1.0
0.45
–1.5
0.30
–2.0
0.15
–2.5
–20
–10
0
10
MAGNITUDE RATIO – dB
20
1.4
3.0
1.65
2.5
1.50
2.0
1.35
1.5
1.20
2700
VMAG – V
1.2
1.0
0.8
0.6
2200
0.4
0.2
900
0
–30 –25 –20 –15 –10 –5
0
5
10
MAGNITUDE RATIO – dB
100
15
20
25
–3.0
30
1.80
1900
1.6
VMAG – V
0.0
+85ⴗC
0.75
TPC 4. VMAG and Log Conformance vs. Input Level Ratio
(Gain), Frequency 900 MHz, –40 ⴗC, +25 ⴗC, and +85ⴗ C,
Reference Level = –30 dBm
1.8
30
1.0
1.05
0.5
–40ⴗC
0.90
0.0
+25ⴗC
0.75
–0.5
+85ⴗC
0.60
–1.0
0.45
–1.5
0.30
–2.0
0.15
–2.5
0
–30
TPC 2. VMAG vs. Input Level Ratio (Gain) VINPA/VINPB,
Frequencies 100 MHz, 900 MHz, 1900 MHz, 2200 MHz,
2700 MHz, PINPA = –30 dBm
–20
–10
0
10
MAGNITUDE RATIO – dB
20
–3.0
30
TPC 5. VMAG and Log Conformance vs. Input Level Ratio
(Gain), Frequency 1900 MHz, –40 ⴗC, +25 ⴗC, and +85 ⴗC,
Reference Level = –30 dBm
1.80
3.0
1.80
3.0
1.65
2.5
1.65
2.5
1.50
2.0
1.50
2.0
1.35
1.5
1.35
1.5
1.20
1.0
0.5
+25ⴗC
0.90
0.0
+85ⴗC
0.75
–0.5
1.20
VMAG – V
–40ⴗC
1.05
ERROR IN VMAG – dB
VMAG – V
0.5
+25ⴗC
0.90
0
–30
30
1.0
–40ⴗC
1.05
1.0
1.05
0.5
–40ⴗC
0.90
0.0
+25ⴗC
0.75
–0.5
+85ⴗC
0.60
–1.0
–1.5
0.45
–1.5
0.30
–2.0
0.30
–2.0
0.15
–2.5
0.15
–2.5
0.60
–1.0
0.45
0
–30
–20
–10
0
10
MAGNITUDE RATIO – dB
20
ERROR IN VMAG – dB
2700
1.0
VMAG – V
VMAG – V
2200
–3.0
30
0
–30
TPC 3. VMAG Output and Log Conformance vs. Input
Level Ratio (Gain), Frequency 100 MHz, –40 ⴗC, +25 ⴗC,
and +85 ⴗC, Reference Level = –30 dBm
–20
–10
0
10
MAGNITUDE RATIO – dB
20
ERROR IN VMAG – dB
2.0
ERROR IN VMAG – dB
(VS = 5 V, VINPB is the reference input and VINPA is swept, unless otherwise noted. All references to dBm are referred to 50 ⍀. For the phase output
curves, the input signal levels are equal, unless otherwise noted.)
–3.0
30
TPC 6. VMAG Output and Log Conformance vs. Input
Level Ratio (Gain), Frequency 2200 MHz, –40 ⴗC, +25 ⴗC,
and +85 ⴗC, Reference Level = –30 dBm
–6–
REV. A
AD8302
3.0
2.0
2.5
1.8
2.0
1.6
–40 C
+85 C
1.4
1.0
0.5
VMAG – V
0.0
–0.5
–1.0
0.8
+25 C
0.4
–40 C
–2.0
1.0
0.6
+85 C
–1.5
1.2
0.2
–2.5
–3.0
0
5
10
–30 –25 –20 –15 –10 –5
MAGNITUDE RATIO – dB
15
20
25
0.0
–30 –25 –20 –15 –10 –5
0
5
10
MAGNITUDE RATIO – dB
30
TPC 7. Distribution of Magnitude Error vs. Input Level
Ratio (Gain), Three Sigma to Either Side of Mean,
Frequency 900 MHz, –40 ⴗC, +25 ⴗC, and +85 ⴗC, Reference Level = –30 dBm
15
20
25
30
TPC 10. Distribution of VMAG vs. Input Level Ratio (Gain),
Three Sigma to Either Side of Mean, Frequency 1900 MHz,
Temperatures Between –40 ⴗC and +85 ⴗC, Reference Level
= –30 dBm
1.8
3.0
2.5
3.0
–45dBm
2.5
1.6
2.0
2.0
1.4
–40 C
1.0
+85 C
0.5
0.0
–0.5
–2.0
–30dBm
1.5
1.0
–15dBm
1.0
0.5
0.0
0.8
–30dBm
–0.5
–15dBm
0.6
–1.0
–1.5
–45dBm
1.2
VMAG – V
ERROR IN VMAG – dB
1.5
+25 C
–40 C
–1.0
–1.5
0.4
+85 C
–2.0
0.2
–2.5
–3.0
0
5
10
–30 –25 –20 –15 –10 –5
MAGNITUDE RATIO – dB
ERROR IN VMAG – dB
ERROR IN VMAG – dB
1.5
15
20
25
0.0
–30
30
TPC 8. Distribution of Error vs. Input Level Ratio (Gain),
Three Sigma to Either Side of Mean, Frequency 1900 MHz,
–40 ⴗC, +25 ⴗC, and +85 ⴗC, Reference Level = –30 dBm
–2.5
–20
–10
0
10
MAGNITUDE RATIO – dB
20
–3.0
30
TPC 11. VMAG Output and Log Conformance vs. Input
Level Ratio (Gain), Reference Level = –15 dBm, –30 dBm,
and –45 dBm, Frequency 1900 MHz
3.0
1.10
2.5
1.05
2.0
PINPA = PINPB + 5dB
–40 C
+85 C
1.00
1.0
0.5
VMAG – V
ERROR IN VMAG – dB
1.5
0.0
–0.5
0.95
PINPA = PINPB
0.90
–1.0
–1.5
–2.0
0.85
+25 C +85 C
–40 C
0.80
–2.5
–3.0
0
5
10
–30 –25 –20 –15 –10 –5
MAGNITUDE RATIO – dB
15
20
25
0.75
–65 –60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
INPUT LEVEL – dBm
30
TPC 9. Distribution of Magnitude Error vs. Input Level
Ratio (Gain), Three Sigma to Either Side of Mean,
Frequency 2200 MHz, Temperatures –40 ⴗC, +25 ⴗC, and
+85 ⴗC, Reference Level = –30 dBm
REV. A
PINPA = PINPB – 5dB
0
TPC 12. VMAG Output vs. Input Level for P INPA = PINPB,
PINPA = PINPB + 5 dB, PINPA = PINPB – 5 dB, Frequency 1900 MHz
–7–
AD8302
1.06
18
1.04
1.02
PINPA = PINPB + 5dB
1.00
15
0.98
0.96
12
PERCENT
VMAG – V
0.94
0.92
0.90
PINPA = PINPB
0.88
9
0.86
6
0.84
0.82
0.80
3
PINPA = PINPB – 5dB
0.78
0.76
0.74
0
200
400
600
0
0.80
800 1000 1200 1400 1600 1800 2000 2200
FREQUENCY – MHz
0.85
0.90
0.95
1.00
MCP – V
TPC 13. VMAG Output vs. Frequency, for PINPA = PINPB, PINPA
= PINPB + 5 dB, and PINPA = PINPB – 5 dB, PINPB = –30 dBm
TPC 16. Center Point of Magnitude Output (MCP)
Distribution Frequencies 900 MHz, 17,000 Units
0.4
18
0.2
15
–0.2
12
–0.4
PERCENT
CHANGE IN SLOPE – mV
0
–0.6
–0.8
–1.0
9
6
–1.2
–1.4
3
–1.6
–1.8
–40
–20
0
20
40
60
TEMPERATURE – ⴗC
80
0
27.0
85
27.5
28.0
28.5
29.0
29.5
30.0
VMAG SLOPE – mV/dB
TPC 17. VMAG Slope, Frequency 900 MHz, 17,000 Units
TPC 14. Change in VMAG Slope vs. Temperature, Three
Sigma to Either Side of Mean, Frequencies 1900 MHz
25
0.032
20
15
0.030
SLOPE OF VMAG – V
VMAG – mV
10
5
0
–5
–10
0.028
0.026
–15
2800
2600
2400
2200
2000
1800
1600
1400
90
1200
80
800
70
1000
60
600
10 20 30 40 50
TEMPERATURE – ⴗC
400
0.024
0
0
–25
–40 –30 –20 –10
200
–20
FREQUENCY – MHz
TPC 15. Change in Center Point of Magnitude Output
(MCP) vs. Temperature, Three Sigma to Either Side of
Mean, Frequencies 1900 MHz
TPC 18. VMAG Slope vs. Frequency
–8–
REV. A
AD8302
10000
VMAG – nV/ Hz
INPUT –50dBm
20mV PER
VERTICAL
DIVISION
25ns
HORIZONTAL
1000
INPUT –30dBm
INPUT –10dBm
100
10
1k
10k
100k
1M
FREQUENCY – Hz
10M
100M
TPC 22. Magnitude Output Noise Spectral
Density, PINPA = PINPB = –10 dBm, –30 dBm,
–50 dBm, No Filter Capacitor
TPC 19. Magnitude Output Response to 4 dB Step, for
PINPB = –30 dBm, PINPA = –32 dBm to –28 dBm, Frequency
1900 MHz, No Filter Capacitor
10000
VMAG – nV/ Hz
INPUT –50dBm
20mV PER
VERTICAL
DIVISION
1000
INPUT –30dBm
100
INPUT –10dBm
1.00␮s
HORIZONTAL
10
1k
TPC 20. Magnitude Output Response to 4 dB Step, for
PINPB = –30 dBm, PINPA = –32 dBm to –28 dBm, Frequency
1900 MHz, 1 nF Filter Capacitor
10k
100k
1M
FREQUENCY – Hz
10M
100M
TPC 23. Magnitude Output Noise Spectral Density, PINPA = PINPB
= –10 dBm, –30 dBm, –50 dBm, with Filter Capacitor, C = 1 nF
0.18
VMAG (PEAK-TO-PEAK) – V
0.16
200mV PER
VERTICAL
DIVISION
100ns
HORIZONTAL
0.14
0.12
0.10
0.08
2700
1900
0.06
0.02
100
0.00
–25
TPC 21. Magnitude Output Response to 40 dB Step, for
PINPB = –30 dBm, PINPA = –50 dBm to –10 dBm, Supply 5 V,
Frequency 1900 MHz, No Filter Capacitor
REV. A
2200
900
0.04
–20
–15
–10
–5
0
5
10
MAGNITUDE RATIO – dB
15
20
25
TPC 24. VMAG Peak-to-Peak Output Induced by Sweeping
Phase Difference through 360 Degrees vs. Magnitude Ratio,
Frequencies 100 MHz, 900 MHz, 1900 MHz, 2200 MHz, and
2700 MHz
–9–
1.80
10
1.62
8
1.44
6
1.26
4
1.08
2
0.90
0
0.72
–2
0.54
–4
0.4
0.36
–6
0.2
0.18
–8
1.8
1.6
900MHz
1.4
1.2
PHASE OUT – V
PHASE OUT – V
1900MHz
2200MHz
2700MHz
1.0
0.8
0.6
0.0
–180
–140
–100 –60
–20
20
60
100
PHASE DIFFERENCE – Degrees
140
0.00
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
180
TPC 28. VPHS Output and Nonlinearity vs. Input Phase
Difference, Input Levels –30 dBm, Frequency 1900 MHz
1.80
10
1.80
10
1.62
8
1.62
8
1.44
6
1.44
6
1.26
4
1.26
4
1.08
2
1.08
2
0.90
0
0.72
–2
PHASE OUT – V
ERROR – Degrees
PHASE OUT – V
TPC 25. Phase Output (VPHS) vs. Input Phase Difference,
Input Levels –30 dBm, Frequencies 100 MHz, 900 MHz,
1900 MHz, 2200 MHz, Supply 5 V, 2700 MHz
0.90
0
0.72
–2
0.54
–4
0.54
–4
0.36
–6
0.36
–6
0.18
–8
0.18
–8
0.00
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
0.00
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
–10
120 150 180
10
10
1.62
8
8
1.44
6
6
1.26
4
4
1.08
2
0.90
0
0.72
–2
0.54
–4
0.36
–6
0.18
–8
ERROR – Degrees
ERROR – Degrees
1.80
0.00
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
–10
120 150 180
TPC 29. VPHS Output and Nonlinearity vs. Input Phase
Difference, Input Levels –30 dBm, Frequency 2200 MHz
TPC 26. VPHS Output and Nonlinearity vs. Input Phase
Difference, Input Levels –30 dBm, Frequency 100 MHz
PHASE OUT – V
–10
120 150 180
ERROR – Degrees
100MHz
ERROR – Degrees
AD8302
0
–2
–4
+85ⴗC
–40ⴗC
–6
–8
–10
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
–10
120 150 180
TPC 27. VPHS Output and Nonlinearity vs. Input Phase
Difference, Input Levels –30 dBm, Frequency 900 MHz
+25ⴗC
2
120 150 180
TPC 30. Distribution of VPHS Error vs. Input Phase Difference, Three Sigma to Either Side of Mean, Frequency
900 MHz, –40 ⴗC, +25 ⴗC, and +85 ⴗC, Input Levels –30 dBm
–10–
REV. A
AD8302
0.15
8
0.10
6
0.05
CHANGE IN VPHS SLOPE – mV
10
ERROR – Degrees
4
+25ⴗC
–40ⴗC
2
0
–2
–4
+85ⴗC
–6
MEAN +3 SIGMA
0.00
–0.05
–0.10
–0.15
MEAN –3 SIGMA
–0.20
–0.25
–0.30
–8
–10
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
–0.35
–40 –30 –20 –10
120 150 180
TPC 31. Distribution of VPHS Error vs. Input Phase
Difference, Three Sigma to Either Side of Mean, Frequency
1900 MHz, –40 ⴗC, +25 ⴗC, and +85 ⴗC, Supply 5 V, Input
Levels PINPA = PINPB = –30 dBm
0
10 20 30 40 50
TEMPERATURE – ⴗC
60
70
80
90
TPC 34. Change in VPHS Slope vs. Temperature, Three
Sigma to Either Side of Mean, Frequency 1900 MHz
10
10
8
5
6
0
+3 SIGMA
–5
+85ⴗC +25ⴗC
2
PERCENT
ERROR – Degrees
4
0
–2
–4
–10
–20
–25
–40ⴗC
–6
–30
–8
–35
–10
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
–3 SIGMA
–15
–40
–40 –30 –20 –10
120 150 180
TPC 32. Distribution of VPHS Error vs. Input Phase Difference, Three Sigma to Either Side of Mean, Frequency
2200 MHz, –40 ⴗC, +25 ⴗC, and +85 ⴗC, Input Levels –30 dBm
0
10 20 30 40 50
VPHS – mV/Degree
60
70
80
90
TPC 35. Change in Phase Center Point (PCP) vs.
Temperature, Three Sigma to Either Side of Mean,
Frequency 1900 MHz
1.8
18
1.6
15
1.4
12
PERCENT
VPHS – V
1.2
1.0
0.8
0.6
9
6
0.4
3
0.2
0.0
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
TPC 33. Distribution of VPHS vs. Input Phase Difference, Three Sigma to Either Side of Mean, Frequency
900 MHz, Temperature between –40 ⴗC and +85 ⴗC, Input
Levels –30 dBm
REV. A
0
0.75
120 150 180
0.80
0.85
0.90
PCP – V
0.95
1.00
1.05
TPC 36. Phase Center Point (PCP) Distribution, Frequency
900 MHz, 17,000 Units
–11–
AD8302
16
14
12
PERCENT
10
100mV PER
VERTICAL
DIVISION
8
6
4
2
50ns HORIZONTAL
0
9.5
9.7
9.9
10.1
10.3
10.5
VPHS – mV/Degree
10.7
10.9
11.1
TPC 37. VPHS Slope Distribution, Frequency
900 MHz
TPC 40. VPHS Output Response to 40 ⴗ Step with Nominal
Phase Shift of 90 ⴗ, Input Levels PINPA = PINPB = –30 dBm,
Frequency 1900 MHz,1 pF Filter Capacitor
10000
VPHS – nV/ Hz
INPUT –50dBm
10mV PER
VERTICAL
DIVISION
1000
INPUT –30dBm
INPUT –10dBm
100
50ns HORIZONTAL
10
1k
TPC 38. VPHS Output Response to 4 ⴗ Step with Nominal
Phase Shift of 90ⴗ, Input Levels –30 dBm, Frequency
1900 MHz, 25ⴗC, 1 pF Filter Capacitor
10k
100k
1M
FREQUENCY – Hz
10M
100M
TPC 41. VPHS Output Noise Spectral Density vs. Frequency,
PINPA = –30 dBm, PINPB = –10 dBm, –30 dBm, –50 dBm, and
90ⴗ Input Phase Difference
1.80
PINPA = –30dBm
1.62
1.44
PINPA = –15dBm
PHASE OUT – V
1.26
10mV PER
VERTICAL
DIVISION
1.08
0.90
PINPA = –45dBm
0.72
0.54
0.36
0.18
2␮s HORIZONTAL
0.00
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
TPC 39. VPHS Output Response to 4 ⴗ Step with Nominal
Phase Shift of 90ⴗ, Input Levels PINPA = PINPB = –30 dBm,
Supply 5 V, Frequency 1900 MHz, 25ⴗC, with 100 pF Filter
Capacitor
120 150 180
TPC 42. Phase Output vs. Input Phase Difference, PINPA =
PINPB, PINPA = PINPB + 15 dB, PINPA = PINPB – 15 dB, Frequency
900 MHz
–12–
REV. A
AD8302
12
PINPA = –15dBm
1.80
PINPA = –30dBm
PINPA = –20dBm
1.62
8
1.44
PINPA = –45dBm
1.26
PINPA = –40dBm
PHASE OUT – V
ABSOLUTE VALUE OF VPHS
INSTANTANEOUS SLOPE – mV
10
6
4
1.08
0.90
0.72
0.54
0.36
2
0.18
0
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
PINPA = –30dBm
0.00
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
120 150 180
TPC 43. Phase Output Instantaneous Slope,
PINPA = PINPB, PINPA = PINPB + 15 dB, PINPA = PINPB – 15 dB,
Frequency 900 MHz
120 150 180
TPC 46. Phase Output vs. Input Phase Difference,
PINPA = PINPB, PINPA = PINPB + 10 dB, PINPA = PINPB – 10 dB,
Frequency 2200 MHz
12
1.80
PINPA = –20dBm
PINPA = –20dBm
1.62
10
ABSOLUTE VALUE OF VPHS
INSTANTANEOUS SLOPE – mV
1.44
PHASE OUT – V
1.26
PINPA = –40dBm
1.08
0.90
0.72
PINPA = –30dBm
0.54
0.36
PINPA = –40dBm
PINPA = –30dBm
8
6
4
2
0.18
0.00
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
0
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
120 150 180
TPC 44. Phase Output vs. Input Phase Difference,
PINPA = PINPB, PINPA = PINPB + 10 dB, PINPA = PINPB – 10 dB,
Frequency 1900 MHz, Supply 5 V
120 150 180
TPC 47. Phase Output Instantaneous Slope, PINPA = PINPB,
PINPA = PINPB + 10 dB, PINPA = PINPB – 10 dB, Frequency
2200 MHz
12
4000
4.0
3500
3.5
REAL SHUNT Z (⍀)
3.0
PINPA = –40dBm
6
4
2.5
2500
2.0
2000
SHUNT R
1.5
1500
SHUNT C
CAPACITANCE – pF
3000
PINPA = –30dBm
8
RESISTANCE – ⍀
ABSOLUTE VALUE OF VPHS
INSTANTANEOUS SLOPE – mV
10
1.0
1000
CAPACITANCE SHUNT Z (pF)
2
0.5
500
PINPA = –20dBm
0
–180 –150 –120 –90 –60 –30 0
30 60 90
PHASE DIFFERENCE – Degrees
TPC 45. Phase Output Instantaneous Slope, PINPA =
PINPB, PINPA = PINPB + 10 dB, PINPA = PINPB – 10 dB,
Frequency 1900 MHz, Supply 5 V
REV. A
0.0
0
0
120 150 180
500
1000
1500
FREQUENCY – MHz
2000
2500
TPC 48. Input Impedance, Modeled as Shunt R in Parallel
with Shunt C
–13–
AD8302
18
8
6
15
4
PERCENT
VREF – mV
12
2
0
9
6
–2
3
–4
–6
–40 –30 –20 –10
0
10 20 30 40 50
TEMPERATURE – ⴗC
60
70
80
0
1.74
90
TPC 49. Change in VREF vs. Temperature, Three Sigma to
Either Side of Mean
1.76
1.78
1.80
1.82
VREF – V
1.84
1.86
1.88
TPC 51. VREF Distribution, 17,000 Units
120
NOISE – nV/ Hz
100
80
60
40
20
0
1k
10k
100k
1M
FREQUENCY – Hz
10M
100M
TPC 50. VREF Output Noise Spectral Density vs.
Frequency
–14–
REV. A
AD8302
[
The AD8302 measures the magnitude ratio, defined here as
gain, and phase difference between two signals. A pair of
matched logarithmic amplifiers provide the measurement, and
their hard-limited outputs drive the phase detector.
Basic Theory
Logarithmic amplifiers (log amps) provide a logarithmic compression function that converts a large range of input signal
levels to a compact decibel-scaled output. The general mathematical form is:
VOUT = VSLP log (VIN / VZ )
(1)
where VIN is the input voltage, VZ is called the intercept (voltage),
and VSLP is called the slope (voltage). It is assumed throughout
that log(x) represents the log10(x) function. VSLP is thus the
volts/decade, and since a decade of voltage corresponds to
20 dB, V SLP /20 is the volts/dB. V Z is the value of input
signal that results in an output of zero and need not correspond
to a physically realizable part of the log amp signal range.
While the slope is fundamentally a characteristic of the log amp,
the intercept is a function of the input waveform as well.1
Furthermore, the intercept is typically more sensitive to temperature and frequency than the slope. When single log amps
are used for power measurement, this variability introduces
errors into the absolute accuracy of the measurement since the
intercept represents a reference level.
(2)
where VINA and VINB are the input voltages, VMAG is the output
corresponding to the magnitude of the signal level difference,
and VSLP is the slope. Note that the intercept, VZ, has dropped
out. Unlike the measurement of power, when measuring a dimensionless quantity such as relative signal level, no independent
reference or intercept need be invoked. In essence, one signal
serves as the intercept for the other. Variations in intercept due
to frequency, process, temperature, and supply voltage affect both
channels identically and hence do not affect the difference. This
technique depends on the two log amps being well matched
in slope and intercept to ensure cancellation. This is the case
for an integrated pair of log amps. Note that if the two signals
have different waveforms (e.g., different peak-to-average ratios)
or different frequencies, an intercept difference may appear, introducing a systematic offset.
The log amp structure consists of a cascade of linear/limiting
gain stages with demodulating detectors. Further details about
the structure and function of log amps can be found in data
sheets for other log amps produced by Analog Devices.2 The
output of the final stage of a log amp is a fully limited signal
over most of the input dynamic range. The limited outputs from
both log amps drive an exclusive-OR style digital phase detector.
Operating strictly on the relative zero-crossings of the limited signals, the extracted phase difference is independent of the original
input signal levels. The phase output has the general form:
NOTES
1
See the data sheet for the AD640 for a description of the effect of waveform on
the intercept of log amps.
2
For example, see the data sheet for the AD8307.
REV. A
(3)
where VΦ is the phase slope in mV/degree and Φ is each signal’s
relative phase in degrees.
Structure
The general form of the AD8302 is shown in Figure 2. The
major blocks consist of two demodulating log amps, a phase
detector, output amplifiers, a biasing cell, and an output reference voltage buffer. The log amps and phase detector process
the high frequency signals and deliver the gain and phase information in current form to the output amplifiers. The output
amplifiers determine the final gain and phase scaling. External
filter capacitors set the averaging time constants for the respective outputs. The reference buffer provides a 1.80 V reference
voltage that tracks the internal scaling constants.
VIDEO OUTPUT – A
INPA
OFSA
60dB LOG AMPS
(7 DETECTORS)
+
MFLT
+
–
VMAG
–
MSET
PHASE
DETECTOR
COMM
PSET
OFSB
INPB
The AD8302 takes the difference in the output of two identical
log amps, each driven by signals of similar waveforms but at
different levels. Since subtraction in the logarithmic domain
corresponds to a ratio in the linear domain, the resulting
output becomes:
VMAG = VSLP log (VINA / VINB )
]
VPHS = VΦ Φ (VINA) − Φ (VINB )
GENERAL DESCRIPTION AND THEORY
60dB LOG AMPS
(7 DETECTORS)
–
VPHS
+
PFLT
VIDEO OUTPUT – B
VPOS
BIAS
x3
1.8V
VREF
Figure 2. General Structure
Each log amp consists of a cascade of six 10 dB gain stages with
seven associated detectors. The individual gain stages have 3 dB
bandwidths in excess of 5 GHz. The signal path is fully differential to minimize the effect of common-mode signals and noise.
Since there is a total of 60 dB of cascaded gain, slight dc offsets
can cause limiting of the latter stages, which may cause measurement errors for small signals. This is corrected by a feedback
loop. The nominal high-pass corner frequency, fHP, of this loop
is set internally at 200 MHz but can be lowered by adding external
capacitance to the OFSA and OFSB pins. Signals at frequencies
well below the high-pass corner are indistinguishable from dc
offsets and are also nulled. The difference in the log amp outputs is performed in the current domain, yielding by analogy to
Equation 2:
ILA = ISLP log(VINA / VINB )
(4)
where ILA and ISLP are the output current difference and the
characteristic slope (current) of the log amps, respectively. The
slope is derived from an accurate reference designed to be insensitive to temperature and supply voltage.
The phase detector uses a fully symmetric structure with respect
to its two inputs to maintain balanced delays along both signal
paths. Fully differential signaling again minimizes the sensitivity
to common-mode perturbations. The current-mode equivalent
to Equation 3 is:
[
]
IPD = IΦ Φ (VINA) − Φ (VINB ) − 90°
(5)
where IPD and IΦ are the output current and characteristic slope
associated with the phase detector, respectively. The slope is
derived from the same reference as the log amp slope.
–15–
AD8302
Note that by convention, the phase difference is taken in the range
from –180° to +180°. Since this style of phase detector does not
distinguish between ±90°, it is considered to have an unambiguous
180° phase difference range that can be either 0° to +180° centered
at +90° or 0° to –180° centered at –90°.
VP
C7
R4
(6)
where IFB is the feedback current equal to (VSET – VCP)/RF, VSET
is the setpoint input, and T is the integration time constant equal
to RFCAVE/K, where CAVE is the parallel combination of the internal 1.5 pF and the external capacitor CFLT.
1.5pF
MFLT/PFLT
+
K
IIN = ILA OR IPD
VMAG/VPHS
CFLT
–
VMAG
2 INPA
VMAG 13
3 OFSA
MSET 12
4 VPOS
VREF 11
5 OFSB
PSET 10
6 INPB
VPHS 9
7 COMM
PFLT 8
C2
R1
C4
C6
VINB
R2
VPHS
C5
C3
C8
Figure 4. Basic Connections in Measurement Mode with
30 mV/dB and 10 mV/Degree Scaling
In the low frequency limit, the gain and phase transfer functions
given in Equations 4 and 5 become:
VMAG = RF ISLP log(VINA / VINB ) + VCP or
(8a)
VMAG = (RF ISLP / 20) (PINA − PINB ) + VCP
(8b)
(
)
VPHS = –RF IΦ |Φ (VINA) − Φ (VINB ) |–90° + VCP
(9)
which are illustrated in Figure 5. In Equation 8b, PINA and PINB are
the power in dBm equivalent to VINA and VINB at a specified reference impedance. For the gain function, the slope represented by
RF ISLP is 600 mV/decade or, dividing by 20 dB/decade, 30 mV/dB.
With a center point of 900 mV for 0 dB gain, a range of –30 dB to
+30 dB covers the full-scale swing from 0 V to 1.8 V. For the phase
function, the slope represented by RFIΦ is 10 mV/degree. With a
center point of 900 mV for 90°, a range of 0° to 180° covers the
full-scale swing from 1.8 V to 0 V. The range of 0° to –180° covers
the same full-scale swing but with the opposite slope.
VCP = 900mV
IFB
MFLT 14
C1
VINA
The basic structure of both output interfaces is shown in Figure 3. It
accepts a setpoint input and includes an internal integrating/averaging capacitor and a buffer amplifier with gain K. External access to
these setpoints provides for several modes of operation and enables
flexible tailoring of the gain and phase transfer characteristics. The
setpoint interface block, characterized by a transresistance RF, generates a current proportional to the voltage presented to its input pin,
MSET or PSET. A precise offset voltage of 900 mV is introduced
internally to establish the center-point (VCP) for the gain and phase
functions, i.e., the setpoint voltage that corresponds to a gain of 0 dB
and a phase difference of 90°. This setpoint current is subtracted
from the signal current, IIN, coming from the log amps in the gain
channel or from the phase detector in the phase channel. The resulting difference is integrated on the averaging capacitors at either pin
MFLT or PFLT and then buffered by the output amplifier to the
respective output pins, VMAG and VPHS. With this open-loop
arrangement, the output voltage is a simple integration of the difference between the measured gain/phase and the desired setpoint:
VOUT = RF (IIN − IFB ) / (sT )
AD8302
1 COMM
1.8V
+
+
MSET/PSET
20k⍀
30mV/dB
VMAG
RF
Figure 3. Simplified Block Diagram of the Output Interface
900mV
VCP
BASIC CONNECTIONS
Measurement Mode
The basic function of the AD8302 is the direct measurement of gain
and phase. When the output pins, VMAG and VPHS, are connected
directly to the feedback setpoint input pins, MSET and PSET, the
default slopes and center points are invoked. This basic connection
shown in Figure 4 is termed the measurement mode. The current
from the setpoint interface is forced by the integrator to be equal to
the signal currents coming from the log amps and phase detector.
The closed loop transfer function is thus given by:
(7)
The time constant T represents the single-pole response to the envelope of the dB-scaled gain and the degree-scaled phase functions. A
small internal capacitor sets the maximum envelope bandwidth to
approximately 30 MHz. If no external CFLT is used, the AD8302
can follow the gain and phase envelopes within this bandwidth. If
longer averaging is desired, CFLT can be added as necessary according to T (ns) = 3.3 × CAVE (pF). For best transient response with
minimal overshoot, it is recommended that 1 pF minimum value
external capacitors be added to the MFLT and PFLT pins.
–16–
–30
0
+30
MAGNITUDE RATIO – dB
1.8V
+10mV/DEG
VPHS
VOUT = ( I IN RF + VCP ) / (1 + sT )
0V
–10mV/DEG
VCP
900mV
0V
–180
–90
0
90
180
PHASE DIFFERENCE – Degrees
Figure 5. Idealized Transfer Characteristics for the Gain
and Phase Measurement Mode
REV. A
AD8302
Interfacing to the Input Channels
Dynamic Range
The single-ended input interfaces for both channels are identical.
Each consists of a driving pin, INPA and INPB, and an acgrounding pin, OFSA and OFSB. All four pins are internally
dc-biased at about 100 mV from the positive supply and should
be externally ac-coupled to the input signals and to ground. For
the signal pins, the coupling capacitor should offer negligible
impedance at the signal frequency. For the grounding pins, the
coupling capacitor has two functions: It provides ac grounding
and sets the high-pass corner frequency for the internal offset
compensation loop. There is an internal 10 pF capacitor to ground
that sets the maximum corner to approximately 200 MHz.
The corner can be lowered according the formula f HP (MHz) =
2/CC(nF), where CC is the total capacitance from OFSA or OFSB
to ground, including the internal 10 pF.
The maximum measurement range for the gain subsystem is limited to a total of 60 dB distributed from –30 dB to +30 dB. This
means that both gain and attenuation can be measured. The limits
are determined by the minimum and maximum levels that each
individual log amp can detect. In the AD8302, each log amp can
detect inputs ranging from –73 dBV [(223 µV, –60 dBm re: 50 Ω
to –13 dBV (223 mV, 0 dBm re: 50 Ω)]. Note that log
amps respond to voltages and not power. An equivalent power
can be inferred given an impedance level, e.g., to convert from
dBV to dBm in a 50 Ω system, simply add 13 dB. To cover
the entire range, it is necessary to apply a reference level to one log
amp that corresponds precisely to its midrange. In the AD8302,
this level is at –43 dBV, which corresponds to –30 dBm in a 50 Ω
environment. The other channel can now sweep from its low end,
30 dB below midrange, to its high end, 30 dB above midrange. If
the reference is displaced from midrange, some measurement
range will be lost at the extremes. This can occur either if the log
amps run out of range or if the rails at ground or 1.8 V are reached.
Figure 7 illustrates the effect of the reference channel level placement.
If the reference is chosen lower than midrange by 10 dB, then the
lower limit will be at –20 dB rather than –30 dB. If the reference chosen
is higher by 10 dB, the upper limit will be 20 dB rather than 30 dB.
The input impedance to INPA and INPB is a function of
frequency, the offset compensation capacitor, and package
parasitics. At moderate frequencies above fHP, the input network
can be approximated by a shunt 3 kΩ resistor in parallel with a
2 pF capacitor. At higher frequencies, the shunt resistance
decreases to approximately 500 Ω. The Smith Chart in Figure 6
shows the input impedance over the frequency range 100 MHz
to 3 GHz.
MAX RANGE FOR VREF = VREFOPT
VMAG – V
1.80
100MHz
900MHz
0.90
VREF > VREFOPT
VREF < VREFOPT
1.8GHz
2.7GHz
2.2GHz
3.0GHz
–30
0
+30
GAIN MEASUREMENT RANGE – dB
Figure 6. Smith Chart Showing the Input Impedance of a
Single Channel from 100 MHz to 3 GHz
A broadband resistive termination on the signal side of the coupling
capacitors can be used to match to a given source impedance.
The value of the termination resistor, RT, is determined by:
RT = RIN RS / (RIN − RS )
(10)
where RIN is the input resistance and RS the source impedance.
At higher frequencies, a reactive, narrow-band match might be
desirable to tune out the reactive portion of the input impedance.
An important attribute of the two-log-amp architecture is that if
both channels are at the same frequency and have the same input
network, then impedance mismatches and reflection losses become
essentially common-mode and hence do not impact the relative
gain and phase measurement. However, mismatches in these
external components can result in measurement errors.
REV. A
Figure 7. The Effect of Offsetting the Reference Level Is to
Reduce the Maximum Dynamic Range
The phase measurement range is of 0° to 180°. For phase differences of 0° to –180°, the transfer characteristics are mirrored as
shown in Figure 5, with a slope of the opposite sign. The phase
detector responds to the relative position of the zero crossings
between the two input channels. At higher frequencies, the finite
rise and fall times of the amplitude limited inputs create an
ambiguous situation that leads to inaccessible dead zones at the
0° and 180° limits. For maximum phase difference coverage, the
reference phase difference should be set to 90°.
–17–
AD8302
Cross Modulation of Magnitude and Phase
At high frequencies, unintentional cross coupling between signals
in Channels A and B inevitably occurs due to on-chip and boardlevel parasitics. When the two signals presented to the AD8302
inputs are at very different levels, the cross coupling introduces
cross modulation of the phase and magnitude responses. If the two
signals are held at the same relative levels and the phase between
them is modulated then only the phase output should respond.
Due to phase-to-amplitude cross modulation, the magnitude output shows a residual response. A similar effect occurs when the
relative phase is held constant while the magnitude difference is
modulated, i.e., an expected magnitude response and a residual
phase response are observed due to amplitude-to-phase cross
modulation. The point where these effects are noticeable depends
on the signal frequency and the magnitude of the difference. Typically, for differences <20 dB, the effects of cross modulation are
negligible at 900 MHz.
Modifying the Slope and Center Point
The default slope and center point values can be modified with
the addition of external resistors. Since the output interface
blocks are generalized for both magnitude and phase functions,
the scaling modification techniques are equally valid for both
outputs. Figure 8 demonstrates how a simple voltage divider
from the VMAG and VPHS pins to the MSET and PSET pins
can be used to modify the slope. The increase in slope is given by
1 + R1/(R2储20 kΩ). Note that it may be necessary to account for
the MSET and PSET input impedance of 20 kΩ which has a ±20%
manufacturing tolerance. As is generally true in such feedback
systems, envelope bandwidth is decreased and the output noise
transferred from the input is increased by the same factor. For
example, by selecting R1 and R2 to be 10 kΩ and 20 kΩ,
respectively, gain slope increases from the nominal 30 mV/dB
by a factor of 2 to 60 mV/dB. The range is reduced by a factor
of 2 and the new center point is at –15 dB, i.e., the range now
extends from –30 dB, corresponding to VMAG = 0 V, to 0 dB,
corresponding to VMAG = 1.8 V.
20k⍀
NEW SLOPE = 30mV/dB ⴛ 1ⴙ
VMAG
R1
10k⍀
R1
MSET
20k⍀
20k⍀
VREF
Figure 9. The Center Point Is Repositioned with the Help
of the Internal Reference Voltage of 1.80 V
Comparator and Controller Modes
The AD8302 can also operate in a comparator mode if used in
the arrangement shown in Figure 10 where the DUT is the element
to be evaluated. The VMAG and VPHS pins are no longer
connected to MSET and PSET. The trip-point thresholds for the
gain and phase difference comparison are determined by the
voltages applied to pins MSET and PSET according to:
VMSET (V ) = 30 mV / dB × GainSP (dB ) + 900 mV
(11)
(
(12)
)
VPSET (V ) = −10 mV/ ° × | Phase SP ( °)|–90° + 900 mV
where GainSP (dB) and PhaseSP (°) are the desired gain and
phase thresholds. If the actual gain and phase between the two
input channels differ from these thresholds, the VMAG and VPHS
outputs toggle like comparators, i.e.,
VMAG =
VPHS =
R1
NEW SLOPE = 30mV/dB ⴛ 1ⴙ
R2||R20k⍀
VMAG
MSET
reference that determines the nominal center point, their
tracking with temperature, supply, and part-to-part variations
should be better in comparison to a fixed external voltage. If the
center point is shifted to 0 dB in the previous example where
the slope was doubled, then the range spans from –15 dB at
VMAG = 0 V to 15 dB at VMAG = 1.8 V.
R1
1.8 V if Gain > Gain SP
(13)
0 V if Gain < Gain SP
1.8 V if Phase > PhaseSP
(14)
0 V if Phase < PhaseSP
R2
VP
Figure 8. Increasing the Slope Requires the Inclusion of a
Voltage Divider
C7
R4
Repositioning the center point back to its original value of 0 dB
simply requires that an appropriate voltage be applied to the
grounded side of the lower resistor in the voltage divider. This
voltage may be provided externally or derived from the internal
reference voltage on pin VREF. For the specific choice of R2 =
20 kΩ, the center point is easily readjusted to 0 dB by connecting
the VREF pin directly to the lower pin of R2 as shown in Figure 9.
The increase in slope is now simplified to 1 + R1/10 kΩ. Since this
1.80 V reference voltage is derived from the same band gap
AD8302
1 COMM
MFLT 14
C1
VINA
C2
2 INPA
VMAG 13
VMAG
3 OFSA
MSET 12
VMSET
4 VPOS
VREF 11
5 OFSB
PSET 10
VPSET
6 INPB
VPHS 9
VPHS
7 COMM
PFLT 8
R1
C4
C6
VINB
R2
C5
C3
C8
Figure 10. Disconnecting the Feedback to the Setpoint
Controls, the AD8302 Operates in Comparator Mode
–18–
REV. A
AD8302
The comparator mode can be turned into a controller mode by
closing the loop around the VMAG and VPHS outputs.
Figure 11 illustrates a closed loop controller that stabilizes the gain
and phase of a DUT with gain and phase adjustment elements.
If VMAG and VPHS are properly conditioned to drive gain and
phase adjustment blocks preceding the DUT, the actual gain and
phase of the DUT will be forced toward the prescribed setpoint
gain and phase given in Equations 11 and 12. These are essentially
AGC and APC loops. Note that as with all control loops of this kind,
loop dynamics and appropriate interfaces all must be considered
in more detail.
When the insertion phase is nominal, the VPHS output is 900 mV.
Deviations from the nominal are reported with a 10 mV/degree
scaling. Table I gives suggested component values for the
measurement of an amplifier with a nominal gain of 10 dB and
an input power of –10 dBm.
ATTENA
DCA
VP
OUTPUT
C7
AD8302
R4
1 COMM
MFLT 14
C1
2 INPA
⌬MAG
INPA
MSET
AD8302
PSET
INPB
MAG
SETPOINT
3 OFSA
MSET 12
4 VPOS
VREF 11
5 OFSB
PSET 10
6 INPB
VPHS 9
R5
C4
“BLACK BOX”
PHASE
SETPOINT
C6
R2
VPHS
⌬⌽
H
R6
C5
7 COMM
PFLT 8
C3
INPUT
Figure 11. By Applying Overall Feedback to a DUT Via
External Gain and Phase Adjusters, the AD8302 Acts
as a Controller
DCB
C8
ATTENB
Figure 12. Using the AD8302 to Measure the Gain and
Insertion Phase of an Amplifier or Mixer
APPLICATIONS
Measuring Amplifier Gain and Compression
The most fundamental application of AD8302 is the monitoring
of the gain and phase response of a functional circuit block such as
an amplifier or a mixer. As illustrated in Figure 12, directional
couplers, DCB and DCA, sample the input and output signals of
the “Black Box” DUT. The attenuators ensure that the signal
levels presented to the AD8302 fall within its dynamic range.
From the discussion in the Dynamic Range section, the optimal
choice places both channels at POPT = –30 dBm referenced to 50 Ω,
which corresponds to –43 dBV. To achieve this, the combination
of coupling factor and attenuation are given by:
Table I. Component Values for Measuring a 10 dB Amplifier
with an Input Power of –10 dBm
CB + LB = PIN − POPT
(15)
C A + L A = PIN + GAIN NOM − POPT
(16)
where CB and CA are the coupling coefficients, LB and LA are the
attenuation factors, and GAINNOM is the nominal DUT gain. If
identical couplers are used for both ports, then the difference in the
two attenuators compensates for the nominal DUT gain. When the
actual gain is nominal, the VMAG output is 900 mV, corresponding
to 0 dB. Variations from nominal gain appear as a deviation from
900 mV or 0 dB with a 30 mV/dB scaling. Depending on the nominal
insertion phase associated with DUT, the phase measurement may
require a fixed phase shift in series with one of the channels to bring
the nominal phase difference presented to the AD8302 near the
optimal 90° point.
REV. A
H
VMAG 13
R1
VMAG
C2
Component
Value
Quantity
R1, R2
R5, R6
C1, C4, C5, C6
C2, C8
C3
C7
AttenA
AttenB
DCA, DCB
52.3 Ω
100 Ω
0.001 µF
Open
100 pF
0.1 µF
10 dB (See Text)
1 dB (See Text)
20 dB
2
2
4
1
1
1
1
2
The gain measurement application can also monitor gain and
phase distortion in the form of AM-AM (gain compression) and
AM-PM conversion. In this case, the nominal gain and phase
corresponds to those at low input signal levels. As the input level
is increased, output compression and excess phase shifts are
measured as deviations from the low level case. Note that the signal
levels over which the input is swept must remain within the dynamic
range of the AD8302 for proper operation.
–19–
AD8302
Reflectometer
The AD8302 can be configured to measure the magnitude ratio
and phase difference of signals that are incident on and reflected
from a load. The vector reflection coefficient, ⌫, is defined as,
Γ = Reflected Voltage / Incident Voltage = ( Z L − ZO ) / ( Z L + ZO ) (17)
where ZL is the complex load impedance and ZO is the characteristic system impedance.
The measurement accuracy can be compromised if board
level details are not addressed. Minimize the physical distance
between the series connected couplers since the extra path
length adds phase error to ⌫. Keep the paths from the couplers
to the AD8302 as well matched as possible since any differences
introduce measurement errors. The finite directivity, D, of the
couplers sets the minimum detectable reflection coefficient, i.e.,
| ΓMIN(dB)|<|D(dB)|.
The measured reflection coefficient can be used to calculate the
level of impedance mismatch or standing wave ratio (SWR) of a
particular load condition. This proves particularly useful in diagnosing varying load impedances such as antennas that can degrade
performance and even cause physical damage. The vector
reflectometer arrangement given in Figure 13 consists of a pair
of directional couplers that sample the incident and reflected signals. The attenuators reposition the two signal levels within the
dynamic range of the AD8302. In analogy to Equations 15 and
16, the attenuation factors and coupling coefficients are given by:
CB + LB = PIN − POPT
(18)
SOURCE
INCIDENT
WAVE
20dB
R2
C5
C6
ZLOAD
REFLECTED
WAVE
1dB
R1
C4
C1
C3
R4
VP
C7
C A + LA = PIN + ΓNOM − POPT
(19)
where ⌫NOM is the nominal reflection coefficient in dB and is
negative for passive loads. Consider the case where the incident
signal is 10 dBm and the nominal reflection coefficient is –19 dB.
As shown in Figure 13, using 20 dB couplers on both sides and
–30 dBm for POPT, the attenuators for Channel A and B paths
are 1 dB and 20 dB, respectively. The magnitude and phase of
the reflection coefficient are available at the VMAG and VPHS
pins scaled to 30 mV/dB and 10 mV/degree. When ⌫ is –19 dB,
the VMAG output is 900 mV.
AD8302
1 COMM
2 INPA
C2
MFLT 14
⌫
VMAG 13
R5
3 OFSA
MSET 12
4 VPOS
VREF 11
5 OFSB
PSET 10
6 INPB
VPHS 9
⌫
R6
7 COMM
PFLT 8
C8
Figure 13. Using the AD8302 to Measure the Vector
Reflection Coefficient Off an Arbitrary Load
–20–
REV. A
AD8302
VP
VP
C7
AD8302
R4
1 COMM
MFLT 14
C2
C1
INPA
2 INPA
VMAG 13
3 OFSA
MSET 12
GAIN
Table II. P1 Pin Allocations
R5
R1
SW1
GSET
R7
C4
4 VPOS
GND
VREF 11
C6
VREF
SW2
5 OFSB
PSET 10
6 INPB
VPHS 9
7 COMM
PFLT 8
R3
R9
1
2
3
Common
VPOS
Common
R8
R2
INPB
PSET
C5
C3
PHASE
C8
R6
Figure 14. Evaluation Board Schematic
Figure 15a. Component Side Metal of Evaluation Board
Figure 15b. Component Side Silkscreen of Evaluation Board
Table III. Evaluation Board Configuration Options
Component
Function
Default Condition
P1
R1, R2
R3
Not Applicable
R1 = R2 = 52.3 Ω (Size 0402)
R3 = 1 kΩ (Size 0603)
R5, R6, R9
Power Supply and Ground Connector: Pin 2 VPOS and Pins 1 and 3 Ground.
Input Termination. Provide termination for input sources.
VREF Output Load. This load is optional and is meant to allow the user to simulate
their circuit loading of the device.
Snubbing Resistor
C3, C7, R4
Supply Decoupling
C1, C5
C2, C8
Input AC-Coupling Capacitors
Video Filtering. C2 and C8 limit the video bandwidth of the gain and phase
output respectively.
Offset Feedback. These set the high-pass corner of the offset cancellation loop
and thus with the input ac-coupling capacitors the minimum operating frequency.
GSET Signal Source. When SW1 is in the position shown, the device is in gain
measure mode; when switched, it operates in comparator mode and a signal
must be applied to GSET.
PSET Signal Source. When SW2 is in the position shown, the device is in phase
measure mode; when switched, it operates in comparator mode and a signal
must be applied to PSET.
C4, C6
SW1
SW2
REV. A
–21–
R5 = R6 = 0 Ω (Size 0603)
R9 = 0 Ω (Size 0603)
C3 = 100 pF (Size 0603)
C7 = 0.1 µF (Size 0603)
R4 = 0 Ω (Size 0603)
C1 = C5 = 1 nF (Size 0603)
C2 = C8 = Open (Size 0603)
C4 = C6 = 1 nF (Size 0603)
SW1 = Installed
SW1 = Installed
AD8302
CHARACTERIZATION SETUPS AND METHODS
Phase
The general hardware configuration used for most of the AD8302
characterization is shown in Figure 16. The characterization board
is similar to the Customer Evaluation Board. Two reference-locked
R and S SMT03 signal generators are used as the inputs to
INPA and INPB, while the gain and phase outputs are monitored
using both a TDS 744A oscilloscope with 10× high impedance
probes and Agilent 34401A multimeters.
The majority of the VPHS output data was collected by generating
phase change, again by operating the two input sources with a
small frequency offset (normally 100 kHz) using the same
configuration shown in Figure 16. Although this method gives
excellent linear phase change, good for measurement of slope
and linearity, it lacks an absolute phase reference point. In the
curves showing swept phase, the phase at which the VPHS is the
same as VPHS with no input signal is taken to be –90° and all
other angles are references to there. Typical Performance Curves
show two figures of merit; instantaneous slope and error. Instantaneous slope, as shown in TPCs 43, 44, 45, and 47, was calculated
simply by taking the delta in VPHS over angular change for adjacent
measurement points.
Gain
The basic technique used to evaluate the static gain (VMAG)
performance was to set one source to a fixed level and sweep the
amplitude of the other source, while measuring the VMAG output
with the DMM. In practice, the two sources were run at 100 kHz
frequency offset and average output measured with the DMM to
alleviate errors that might be induced by gain/phase modulation
due to phase jitter between the two sources.
TEKTRONIX
TDS 744A
OSCILLOSCOPE
TEKTRONIX
VX1410A
The errors stated are the difference between a best fit line calculated by a linear regression and the actual measured data divided
by the slope of the line to give an error in V/dB. The referred to
25°C error uses this same method while always using the slope
and intercept calculated for that device at 25°C.
R&S
SIGNAL GENERATOR
SMTO3
R&S
SIGNAL GENERATOR
SMTO3
Response measurement made of the VMAG output used the
configuration shown in Figure 17. The variable attenuator,
Alpha AD260, is driven with a HP8112A pulse generator producing a change in RF level within 10 ns.
VMAG
3dB INPA
EVB
VREF
MULTIMETER/
OSCILLOSCOPE
HP 34401A
MULTIMETER
VPHS
3dB INPB
SAME SETUP AS
VMAG
Figure 16. Primary Characterization Setup
Noise spectral density measurements were made using a
HP3589A with the inputs delivered through a Narda 4032C
90° phase splitter.
TEKTRONIX
VX1410A
To measure the modulation of VMAG due to phase variation
again the sources were run at a frequency offset, fOS, effectively
creating a continuous linear change in phase going through 360°
once every 1/fOS seconds. The VMAG output is then measured
with a DSO. When perceivable, only at high frequencies and
large input magnitude differences, the linearly ramping phase
creates a near sinusoid output riding on the expected VMAG dc
output level. The curves in TPC 24 show the peak-to-peak output level measured with averaging.
R&S
SIGNAL
GENERATOR
SMTO3
FIXED
ATTEN
3dB INPA
EVB
VARIABLE
ATTEN
3dB INPB
VMAG P
TEKTRONIX
TDS 744A
OSCILLOSCOPE
VREF
VPHS
SPLITTER
PULSE
GENERATOR
Figure 17. VMAG Dynamic Performance Measurement Setup
–22–
REV. A
AD8302
OUTLINE DIMENSIONS
14-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-14)
Dimensions shown in millimeters
5.10
5.00
4.90
14
8
4.50
4.40
4.30
6.40
BSC
1
7
PIN 1
1.05
1.00
0.80
COPLANARITY
0.65
BSC
1.20
MAX
0.15
0.05
0.30
0.19
SEATING
PLANE
0.20
0.09
8ⴗ
0ⴗ
COMPLIANT TO JEDEC STANDARDS MO-153AB-1
REV. A
–23–
0.75
0.60
0.45
Revision History
Location
Page
7/02—Data Sheet changed from REV. 0 to REV. A.
PRINTED IN U.S.A.
C02492–0–7/02(A)
TPCs 3 through 6 replaced . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
–24–
REV. A