AD AD8045ACPZ-REEL7 3 nv/hz ultralow distortion, high speed op amp Datasheet

3 nV/√Hz Ultralow Distortion,
High Speed Op Amp
AD8045
FEATURES
APPLICATIONS
Ultralow distortion
SFDR
−101 dBc @ 5 MHz
−90 dBc @ 20 MHz
−63 dBc @ 70 MHz
Third-order intercept
43 dBm @ 10 MHz
Low noise
3 nV/√Hz
3 pA/√Hz
High speed
1 GHz, −3 dB bandwidth (G = +1)
1350 V/µs slew rate
7.5 ns settling time to 0.1%
Standard and low distortion pinout
Supply current: 15 mA
Offset voltage: 1.0 mV max
Wide supply voltage range: 3.3 V to 12 V
Instrumentation
IF and baseband amplifiers
Active filters
ADC drivers
DAC buffers
CONNECTION DIAGRAMS
1
8
+VS
2
7
OUTPUT
–IN
3
6
NC
+IN
4
5
–VS
04814-0-001
NC
FEEDBACK
FEEDBACK 1
8
NC
–IN 2
7
+ VS
+IN 3
6
OUTPUT
–VS 4
5
NC
04814-0-001
Figure 1. 8-Lead AD8045 LFCSP (CP-8)
Figure 2. 8-Lead AD8045 SOIC/EP (RD-8)
GENERAL DESCRIPTION
The AD8045 features a low distortion pinout for the LFCSP,
which improves second harmonic distortion and simplifies the
layout of the circuit board.
The AD8045 has 1 GHz bandwidth, 1350 V/µs slew rate, and
settles to 0.1% in 7.5 ns. With a wide supply voltage range (3.3 V
to 12 V) and low offset voltage (200 µV), the AD8045 is an ideal
candidate for systems that require high dynamic range, precision, and high speed.
The AD8045 amplifier is available in a 3 mm × 3 mm LFCSP
and the standard 8-lead SOIC. Both packages feature an
exposed paddle that provides a low thermal resistance path to
the PCB. This enables more efficient heat transfer, and increases
reliability. The AD8045 works over the extended industrial
temperature range (−40°C to +125°C).
–20
HARMONIC DISTORTION (dBc)
The AD8045 is a unity gain stable voltage feedback amplifier
with ultralow distortion, low noise, and high slew rate. With a
spurious-free dynamic range of −90 dBc @ 20 MHz, the
AD8045 is an ideal solution in a variety of applications,
including ultrasound, ATE, active filters, and ADC drivers.
ADI’s proprietary next generation XFCB process and innovative
architecture enables such high performance amplifiers.
G = +1
–30 VS = ±5V
VOUT = 2V p-p
–40 RL = 1kΩ
RS = 100Ω
–50
–60
–70
–80
HD3 LFCSP
–90
HD2 LFCSP
–100
–120
0.1
1
10
FREQUENCY (MHz)
100
04814-0-079
–110
Figure 3. Harmonic Distortion vs. Frequency for Various Packages
Rev. A
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.326.8703
© 2004 Analog Devices, Inc. All rights reserved.
AD8045
TABLE OF CONTENTS
Specifications with ±5 V Supply ..................................................... 3
Applications..................................................................................... 19
Specifications with +5 V Supply ..................................................... 4
Low Distortion Pinout............................................................... 19
Absolute Maximum Ratings............................................................ 5
High Speed ADC Driver ........................................................... 19
Thermal Resistance ...................................................................... 5
90 MHz Active Low-Pass Filter (LPF) ..................................... 20
ESD Caution.................................................................................. 5
Printed Circuit Board Layout ....................................................... 22
Pin Configurations and Function Descriptions ........................... 6
Signal Routing............................................................................. 22
Typical Performance Characteristics ............................................. 7
Power Supply Bypassing ............................................................ 22
Circuit Configurations................................................................... 16
Grounding ................................................................................... 22
Wideband Operation ................................................................. 16
Exposed Paddle........................................................................... 23
Theory of Operation ...................................................................... 17
Driving Capacitive Loads.......................................................... 23
Frequency Response................................................................... 17
Outline Dimensions ....................................................................... 24
DC Errors .................................................................................... 17
Ordering Guide .......................................................................... 24
Output Noise............................................................................... 18
REVISION HISTORY
9/04—Data Sheet Changed from Rev. 0 to Rev. A
Changes to Features......................................................................... 1
Changes to Specifications ............................................................... 4
Changes to Figure 58..................................................................... 15
Changes to Figure 63..................................................................... 17
Changes to Frequency Response Section ................................... 17
Changes to Figure 64..................................................................... 17
Changes to DC Errors Section..................................................... 17
Changes to Figure 65..................................................................... 17
Changes to Figure 66..................................................................... 18
Changes to Output Noise Section ............................................... 18
Changes to Ordering Guide ......................................................... 24
7/04—Revision 0: Initial Version
Rev. A | Page 2 of 24
AD8045
SPECIFICATIONS WITH ±5 V SUPPLY
TA = 25°C, G = +1, RS = 100 Ω, RL = 1 kΩ to ground, unless noted otherwise. Exposed paddle must be floating or connected to −VS.
Table 1.
Parameter
DYNAMIC PERFORMANCE
–3 dB Bandwidth
Bandwidth for 0.1 dB Flatness
Slew Rate
Settling Time to 0.1%
NOISE/HARMONIC PERFORMANCE
Harmonic Distortion (dBc) HD2/HD3
Input Voltage Noise
Input Current Noise
Differential Gain Error
Differential Phase Error
DC PERFORMANCE
Input Offset Voltage
Input Offset Voltage Drift
Input Bias Current
Input Bias Current Drift
Input Bias Offset Current
Open-Loop Gain
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
Input Common-Mode Voltage Range
Common-Mode Rejection
OUTPUT CHARACTERISTICS
Output Overdrive Recovery Time
Output Voltage Swing
Output Current
Short-Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Positive Power Supply Rejection
Negative Power Supply Rejection
Conditions
Min
G = +1, VOUT = 0.2 V p-p
G = +1, VOUT = 2 V p-p
G = +2, VOUT = 0.2 V p-p
G = +2, VOUT = 2 V p-p, RL = 150 Ω
G = +1, VOUT = 4 V step
G = +2, VOUT = 2 V step
300
320
1000
fC = 5 MHz, VOUT = 2 V p-p
LFCSP
SOIC
fC = 20 MHz, VOUT = 2 V p-p
LFCSP
SOIC
fC = 70 MHz, VOUT = 2 V p-p
LFCSP
SOIC
f = 100 kHz
f = 100 kHz
NTSC, G = +2, RL = 150 Ω
NTSC, G = +2, RL = 150 Ω
Typ
Max
Unit
1000
350
400
55
1350
7.5
MHz
MHz
MHz
V/µs
ns
−102/−101
−106/−101
dBc
dBc
−98/−90
−97/−90
dBc
dBc
−71/−71
−60/−71
3
3
0.01
0.01
dBc
dBc
nV/√Hz
pA/√Hz
%
Degrees
62
0.2
8
2
8
0.2
64
VCM = ±1 V
−83
3.6/1.0
1.3
±3.8
−91
MΩ
pF
V
dB
VIN = ±3 V, G = +2
RL = 1 kΩ
RL = 100 Ω
−3.8 to +3.8
−3.4 to +3.5
8
−3.9 to +3.9
−3.6 to +3.6
70
90/170
18
ns
V
V
mA
mA
pF
See Figure 54
VOUT = −3 V to +3 V
Common-mode/differential
Common-mode
Sinking/sourcing
30% overshoot, G = +2
±1.65
+VS = +5 V to +6 V, −VS = −5 V
+VS = +5 V, −VS = −5 V to −6 V
Rev. A | Page 3 of 24
−61
−66
±5
16
−68
−73
1.0
6.3
1.3
±6
19
mV
µV/°C
µA
nA/°C
µA
dB
V
mA
dB
dB
AD8045
SPECIFICATIONS WITH +5 V SUPPLY
TA = 25°C, G = +1, RS = 100 Ω, RL = 1 kΩ to midsupply, unless otherwise noted. Exposed paddle must be floating or connected to −VS.
Table 2.
Parameter
DYNAMIC PERFORMANCE
–3 dB Bandwidth
Bandwidth for 0.1 dB Flatness
Slew Rate
Settling Time to 0.1%
NOISE/HARMONIC PERFORMANCE
Harmonic Distortion (dBc) HD2/HD3
Input Voltage Noise
Input Current Noise
Differential Gain Error
Differential Phase Error
DC PERFORMANCE
Input Offset Voltage
Input Offset Voltage Drift
Input Bias Current
Input Bias Current Drift
Input Bias Offset Current
Open-Loop Gain
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
Input Common-Mode Voltage Range
Common-Mode Rejection
OUTPUT CHARACTERISTICS
Output Overdrive Recovery Time
Output Voltage Swing
Output Current
Short-Circuit Current
Capacitive Load Drive
POWER SUPPLY
Operating Range
Quiescent Current
Positive Power Supply Rejection
Negative Power Supply Rejection
Conditions
G = +1, VOUT = 0.2 V p-p
G = +1, VOUT = 2 V p-p
G = +2, VOUT = 0.2 V p-p
G = +2, VOUT = 2 V p-p, RL = 150 Ω
G = +1, VOUT = 2 V step
G = +2, VOUT = 2 V step
Min
160
320
480
fC = 5 MHz, VOUT = 2 V p-p
LFCSP
SOIC
fC = 20 MHz, VOUT = 2 V p-p
LFCSP
SOIC
fC = 70 MHz, VOUT = 2 V p-p
LFCSP
SOIC
f = 100 kHz
f = 100 kHz
NTSC, G = +2, RL = 150 Ω
NTSC, G = +2, RL = 150 Ω
Typ
Max
Unit
900
200
395
60
1060
10
MHz
MHz
MHz
MHz
V/µs
ns
−89/−83
−92/−83
dBc
dBc
−81/−70
−83/−70
dBc
dBc
−57/−46
−57/−46
3
3
0.01
0.01
dBc
dBc
nV/√Hz
pA/√Hz
%
Degrees
61
0.5
7
2
7
0.2
63
VCM = 2 V to 3 V
−78
3/0.9
1.3
1.2 to 3.8
−94
MΩ
pF
V
dB
VIN = −0.5 V to +3 V, G = +2
RL = 1 kΩ
RL = 100 Ω
2.2 to 3.7
2.5 to 3.5
10
1.1 to 4.0
1.2 to 3.8
55
70/140
15
ns
V
V
mA
mA
pF
See Figure 54
VOUT = 2 V to 3 V
Common-mode/differential
Common-mode
Sinking/sourcing
30% overshoot, G = +2
3.3
+VS = +5 V to +6 V, −VS = 0 V
+VS = +5 V, −VS = 0 V to −1 V
Rev. A | Page 4 of 24
−65
−70
5
15
−67
−73
1.4
6.6
1.3
12
18
mV
µV/°C
µA
nA/°C
µA
dB
V
mA
dB
dB
AD8045
ABSOLUTE MAXIMUM RATINGS
Table 3.
Rating
12.6 V
See Figure 4
−VS − 0.7 V to +VS + 0.7 V
±VS
−VS
−65°C to +125°C
−40°C to +125°C
300°C
150°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
THERMAL RESISTANCE
θJA is specified for the worst-case conditions, i.e., θJA is specified
for device soldered in circuit board for surface-mount packages.
Table 4. Thermal Resistance
θJA
80
93
θJC
30
35
Unit
°C/W
°C/W
PD = Quiescent Power + (Total Drive Power – Load Power)
⎛V V
PD = (VS × I S ) + ⎜⎜ S × OUT
RL
⎝ 2
⎞ VOUT 2
⎟–
⎟
RL
⎠
RMS output voltages should be considered. If RL is referenced to
−VS, as in single-supply operation, the total drive power is VS ×
IOUT. If the rms signal levels are indeterminate, consider the
worst case, when VOUT = VS/4 for RL to midsupply.
PD = (VS × I S ) +
(VS / 4 )2
RL
In single-supply operation with RL referenced to −VS, worst case
is VOUT = VS/2.
Airflow increases heat dissipation, effectively reducing θJA.
Also, more metal directly in contact with the package leads and
exposed paddle from metal traces, through holes, ground, and
power planes reduce θJA.
Figure 4 shows the maximum safe power dissipation in the
package versus the ambient temperature for the exposed paddle
SOIC (80°C/W) and LFCSP (93°C/W) package on a JEDEC
standard 4-layer board. θJA values are approximations.
Maximum Power Dissipation
The maximum safe power dissipation for the AD8045 is limited
by the associated rise in junction temperature (TJ) on the die. At
approximately 150°C, which is the glass transition temperature,
the properties of the plastic change. Even temporarily exceeding
this temperature limit may change the stresses that the package
exerts on the die, permanently shifting the parametric performance of the AD8045. Exceeding a junction temperature of
175°C for an extended period of time can result in changes in
silicon devices, potentially causing degradation or loss of
functionality.
4.0
3.5
3.0
2.5
2.0
1.5
SOIC
1.0
LFCSP
0.5
0.0
–40
–20
0
20
40
60
80
AMBIENT TEMPERATURE (°C)
100
120
04814-0-080
Package Type
SOIC
LFCSP
The power dissipated in the package (PD) is the sum of the quiescent power dissipation and the power dissipated in the die
due to the AD8045 drive at the output. The quiescent power is
the voltage between the supply pins (VS) times the quiescent
current (IS).
MAXIMUM POWER DISSIPATION (Watts)
Parameter
Supply Voltage
Power Dissipation
Common-Mode Input Voltage
Differential Input Voltage
Exposed Paddle Voltage
Storage Temperature
Operating Temperature Range
Lead Temperature Range
(Soldering 10 sec)
Junction Temperature
Figure 4. Maximum Power Dissipation vs. Temperature for a 4-Layer Board
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate
on the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation
and loss of functionality.
Rev. A | Page 5 of 24
AD8045
AD8045
+VS 7
OUTPUT 6
NC 5
1
FEEDBACK
2
–IN
3
BOTTOM VIEW
(Not to Scale)
4
+IN
–VS
NC = NO CONNECT
04814-0-003
NC 8
+VS
8
OUTPUT
7
NC
6
–VS
5
BOTTOM
VIEW
(Not to Scale)
1
NC
2
FEEDBACK
3
–IN
4
+IN
NC = NO CONNECT
Figure 5. SOIC Pin Configuration
04814-0-004
PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS
Figure 6 . 8-Lead LFCSP Pin Configuration
Note: The exposed paddle must be connected to −VS or it must be electrically isolated (floating).
Table 5. 8-Lead SOIC Pin Function Descriptions
Table 6. 8-Lead LFCSP Pin Function Descriptions
Pin No.
1
2
3
4
5
6
7
8
9
Pin No.
1
2
3
4
5
6
7
8
9
Mnemonic
FEEDBACK
−IN
+IN
−VS
NC
OUTPUT
+VS
NC
Exposed Paddle
Description
Feedback Pin
Inverting Input
Noninverting Input
Negative Supply
NC
Output
Positive Supply
NC
Must Be Connected to −VS or
Electrically Isolated
Rev. A | Page 6 of 24
Mnemonic
NC
FEEDBACK
−IN
+IN
−VS
NC
OUTPUT
+VS
Exposed Paddle
Description
No Connect
Feedback Pin
Inverting Input
Noninverting Input
Negative Supply
No Connect
Output
Positive Supply
Must Be Connected to −VS or
Electrically Isolated
AD8045
TYPICAL PERFORMANCE CHARACTERISTICS
12
VS = ±5V
RL = 1kΩ
0
G = +2
11 VS = ±5V
RL = 1kΩ
10 R = 499Ω
F
G = +2
CLOSED-LOOP GAIN (dB)
–1
–2
G = –1
G = +10
–3
–4
–5
8
7
6
5
5pF
4
0pF
3
10
100
FREQUENCY (MHz)
1000
Figure 7. Small Signal Frequency Response for Various Gains
0
10
4
G = +1
3 VS = ±5V
RL = 1kΩ
RL = 1kΩ
2
CLOSED-LOOP GAIN (dB)
RL = 500Ω
1
0
RL = 100Ω
–2
–3
–4
1000
–4
100
FREQUENCY (MHz)
1000
Figure 11. Small Signal Frequency Response for Various Temperatures
6.3
G = +2
VS = ±5V
RF = 499Ω
6.2 R = 150Ω
L
VS = ±2.5V
CLOSED-LOOP GAIN (dB)
2
+125°C
–3
–6
10
5
3
–40°C
–2
+25°C
04814-0-050
100
FREQUENCY (MHz)
Figure 8. Small Signal Frequency Response for Various Loads
G = +1
4 RL = 1kΩ
RS = 100Ω
0
–1
–5
–5
–6
10
1
04814-0-052
–1
1000
Figure 10. Small Signal Frequency Response for Various Capacitive Loads
4
G = +1
3 VS = ±5V
RS = 100Ω
2
100
FREQUENCY (MHz)
04814-0-048
1
1
VS = ±5V
1
0
–1
–2
–3
6.1
VOUT = 2V p-p
6.0
VOUT = 200mV p-p
5.9
5.8
–5
10
100
FREQUENCY (MHz)
1000
04814-0-051
–4
Figure 9. Small Signal Frequency Response for Various Supplies
5.7
1
10
FREQUENCY (MHz)
100
Figure 12. 0.1 dB Flatness vs. Frequency for Various Output Voltages
Rev. A | Page 7 of 24
04814-0-039
CLOSED-LOOP GAIN (dB)
9
2
–7
CLOSED-LOOP GAIN (dB)
18pF
10pF
–6
04814-0-049
NORMALIZED CLOSED-LOOP GAIN (dB)
1
AD8045
0
–1
VS = ±5V
–2
–3
–4
VS = ±2.5V
–5
VS = ±5V
RL = 1kΩ
60
OPEN-LOOP GAIN (dB)
CLOSED-LOOP GAIN (dB)
70
G = +1
RL = 1kΩ
RS = 100Ω
VOUT = 2V p-p
1
–6
–7
0
–45
50
–90
40
–135
30
–180
20
–225
10
–270
0
–315
–8
OPEN-LOOP PHASE (Degrees)
2
–10
0.01
Figure 13. Large Signal Frequency Response for Various Supplies
2
–1
RL = 1kΩ
–2
–3
–4
RL = 100Ω
–5
–6
–7
–8
1000
04814-0-042
100
FREQUENCY (MHz)
–60
–70
–80
HD3 SOIC AND LFCSP
–90
HD2 LFCSP
–100
HD2 SOIC
1
10
FREQUENCY (MHz)
100
Figure 17. Harmonic Distortion vs. Frequency for Various Packages
2
–30
G = +2
1
HARMONIC DISTORTION (dBc)
0
–1
–2
G = +10
G = –1
–3
–4
–5
G = +1
V = ±5V
–40 VS = 4V p-p
OUT
RL = 1kΩ
–50
HD2 SOIC
HD2 LFCSP
–60
–70
HD3 LFCSP AND SOIC
–80
–90
–100
–110
10
100
FREQUENCY (MHz)
1000
04814-0-041
NORMALIZED CLOSED-LOOP GAIN (dB)
1000
–50
–120
0.1
Figure 14. Large Signal Frequency Response for Various Loads
–6 V = ±5V
S
RF = 499Ω
–7 R = 1kΩ
L
VOUT = 2V p-p
–8
1
100
G = +1
–30 VS = ±5V
VOUT = 2V p-p
–40 RL = 1kΩ
RS = 100Ω
–110
–9
–10
10
1
10
FREQUENCY (MHz)
–20
HARMONIC DISTORTION (dBc)
CLOSED-LOOP GAIN (dB)
0
–360
0.1
Figure 16. Open-Loop Gain and Phase vs. Frequency
G = +1
VS = ±5V
RS = 100Ω
VOUT = 2V p-p
1
04814-0-064
1000
04814-0-030
100
FREQUENCY (MHz)
Figure 15. Large Signal Frequency Response for Various Gains
–120
0.1
1
10
FREQUENCY (MHz)
100
Figure 18. Harmonic Distortion vs. Frequency for Various Packages
Rev. A | Page 8 of 24
04814-0-028
–10
10
04814-0-043
–9
AD8045
–30
HARMONIC DISTORTION (dBc)
G = +1
V = ±5V
–30 S
VOUT = 2V p-p
RL = 100Ω
–40 RS = 100Ω
–50
–60
–70
HD2 SOIC
–80
–90
HD2 LFCSP
10
FREQUENCY (MHz)
100
1
10
FREQUENCY (MHz)
100
–40
G = +10
VS = ±5V
V
–50 OUT = 2V p-p
RL = 1kΩ
HARMONIC DISTORTION (dBc)
HARMONIC DISTORTION (dBc)
HD3 SOIC AND LFCSP
–110
0.1
–50
–60
–70
–80
HD2
–90
HD3
10
FREQUENCY (MHz)
100
04814-0-036
1
HD2 SOIC
–60
HD2 LFCSP
–70
–80
–90
HD3 SOIC AND LFCSP
–100
–100
–110
0.1
1
10
FREQUENCY (MHz)
100
Figure 23. Harmonic Distortion vs. Frequency for Various Packages
Figure 20. Harmonic Distortion vs. Frequency for Various Packages
–30
–50
HARMONIC DISTORTION (dBc)
G = –1
VS = ±5V
–40 RL = 150Ω
VOUT = 2V p-p
–50
HD2 LFCSP
–60
HD2 SOIC
–80
–90
G = +1
VS = ±5V
–60 RL = 1kΩ
RS = 100Ω
f = 10MHz
–70
HD3 SOIC AND LFCSP
–80
–90
–100
HD2 SOIC
–110
–100
HD2 LFCSP
HD3 SOIC AND LFCSP
1
10
FREQUENCY (MHz)
100
Figure 21. Harmonic Distortion vs. Frequency for Various Packages
04814-0-037
HARMONIC DISTORTION (dBc)
HD2 LFCSP
–90
Figure 22. Harmonic Distortion vs. Frequency for Various Packages
G = –1
V = ±5V
–30 S
VOUT = 2V p-p
RL = 1kΩ
–40 SOIC AND LFCSP
–110
0.1
–80
04814-0-033
1
–20
–70
HD2 SOIC
–70
–100
HD3 SOIC AND LFCSP
Figure 19. Harmonic Distortion vs. Frequency for Various Packages
–110
0.1
–60
04814-0-034
–110
0.1
G = +2
VS = ±5V
–40 VOUT = 2V p-p
RL = 150Ω
R = 499Ω
–50 F
–120
0
1
2
3
4
5
6
OUTPUT AMPLITUDE (V p-p)
7
8
04814-0-025
–100
04814-0-032
HARMONIC DISTORTION (dBc)
–20
Figure 24. Harmonic Distortion vs. Output Voltage for Various Packages
Rev. A | Page 9 of 24
–40
–30
G = +1
VS = ±5V
R
–50
L = 150Ω
RS = 100Ω
f = 10MHz
–60
G = +1
VS = ±2.5
–40 VOUT = 2V p-p
RL = 1kΩ
RS = 100Ω
–50
HD2 LFCSP
–70
HD2 SOIC
–80
–90
–100
–60
HD3 SOIC AND LFCSP
–70
–80
HD2 LFCSP
–90
HD2 SOIC
–110
0
1
2
3
4
5
6
OUTPUT AMPLITUDE (V p-p)
7
8
04814-0-024
HD3 SOIC AND LFCSP
–100
1
Figure 25. Harmonic Distortion vs. Output Voltage for Various Packages
–20
G = –1
VS = ±5V
–50 RL = 1kΩ
f = 10MHz
SOIC AND LFCSP
–60
HARMONIC DISTORTION (dBc)
–70
–80
HD2
–90
HD3
–100
G = +1
VS = ±2.5V
–30 VOUT = 2V p-p
RL = 100Ω
–40 RS = 100Ω
–50
–60
HD3 SOIC AND LFCSP
–70
–80
HD2 LFCSP
0
1
2
3
4
5
6
OUTPUT VOLTAGE (V p-p)
7
8
HD2 SOIC
04814-0-026
–120
–100
1
10
FREQUENCY (MHz)
100
04814-0-031
–90
–110
Figure 29. Harmonic Distortion vs. Frequency for Various Packages
Figure 26. Harmonic Distortion vs. Output Voltage
–20
–40
HARMONIC DISTORTION (dBc)
G = –1
VS = ±5V
–50 R = 150Ω
L
f = 10MHz
–60
–70
HD2 SOIC
HD2 LFCSP
–80
–90
HD3 SOIC AND LFCSP
–100
G = –1
VS = ±2.5V
–30 VOUT = 2V p-p
RL = 1kΩ
SOIC AND LFCSP
–40
–50
–60
HD3
–70
–80
HD2
–90
–120
0
1
2
3
4
5
6
OUTPUT VOLTAGE (V p-p)
7
8
04814-0-027
–110
–100
0.1
1
10
FREQUENCY (MHz)
100
Figure 30. Harmonic Distortion vs. Frequency for Various Packages
Figure 27. Harmonic Distortion vs. Output Voltage
Rev. A | Page 10 of 24
04814-0-035
HARMONIC DISTORTION (dBc)
100
Figure 28. Harmonic Distortion vs. Frequency for Various Packages
–40
HARMONIC DISTORTION (dBc)
10
FREQUENCY (MHz)
04814-0-029
HARMONIC DISTORTION (dBc)
HARMONIC DISTORTION (dBc)
AD8045
AD8045
–40
0.15
–50
–60
OUTPUT VOLTAGE (V)
HARMONIC DISTORTION (dBc)
RS = 100Ω
RL = 150Ω
G = +1
0.10 V = ±2.5
S
OR VS = ±5V
G = +1
VS = +5V
RL = 1kΩ
RS = 100Ω
f = 10MHz
HD3 SOIC AND LFCSP
–70
–80
–90
HD2 SOIC
0.05
0
–0.05
–0.10
–100
1.0
1.5
2.0
OUTPUT VOLTAGE (V p-p)
2.5
3.0
Figure 31. Harmonic Distortion vs. Output Voltage for Various Packages
–0.15
0
10
15
20
25
TIME (ns)
Figure 34. Small Signal Transient Response for Various Supplies and Loads
–40
0.15
RL = 1kΩ
CL = 10pF
RSNUB = 30Ω
0.10 V = ±5V
S
G = +1
–70
HD3 SOIC AND LFCSP
–80
–90
0.05
0
–0.05
RSNUB
30Ω
–0.10
–100
–110
0.5
0.7
0.9
1.1
1.3
1.5
1.7
1.9
OUTPUT VOLTAGE (V p-p)
2.1
2.3
2.5
Figure 32. Harmonic Distortion vs. Output Voltage for Various Packages
1600
1400
RL
1kΩ
–0.15
0
5
10
15
20
25
TIME (ns)
Figure 35. Small Signal Transient Response for Various Supplies and Loads
0.15
POSITIVE SLEW RATE
RL = 1kΩ
VS = ±5V
CL
10pF
HD2 LFCSP
04814-0-023
HD2 SOIC
04814-0-013
G = +1
VS = +5V
–50 RL = 150Ω
RS = 100Ω
f = 10MHz
–60
OUTPUT VOLTAGE (V)
HARMONIC DISTORTION (dBc)
5
04814-0-012
–110
0.5
04814-0-022
HD2 LFCSP
VS = ±2.5V
G = +2
RC = 1kΩ
OR RC = 150kΩ
0.10
1200
OUTPUT VOLTAGE (V)
800
600
400
0.05
0
–0.05
–0.10
0
0
1
2
3
OUTPUT VOLTAGE STEP (V)
4
5
Figure 33. Slew Rate vs. Output Voltage
–0.15
0
5
10
15
20
25
TIME (ns)
Figure 36. Small Signal Transient Response for Various Loads
Rev. A | Page 11 of 24
04814-0-014
200
04814-0-076
SLEW RATE (V/µs)
NEGATIVE SLEW RATE
1000
AD8045
3
VS = ±5V
RL = 1kΩ
G = +2
18pF
0.15
2
OUTPUT VOLTAGE (V)
0pF
0.05
0
–0.05
–0.10
–0.15 G = +2
VS = ±5V
RL = 1kΩ
–0.20
0
5
10
15
20
25
TIME (ns)
0pF
–2
10pF
18pF
25
1
0
–1
5
10
15
20
25
TIME (ns)
G = –1
VS = ±5V
RL = 1kΩ
–3
0
5
10
20
25
200
TIME (ns)
Figure 38. Large Signal Transient Response for Various Loads
Figure 41. Large Signal Transient Response, Inverting
6
RL = 1kΩ
RS = 100Ω
G = +1
15
04814-0-019
–3
04814-0-016
–2
04814-0-061
–1
–2
INPUT AND OUTPUT VOLTAGE (V)
VS = ±5V
1
VS = ±2.5V
0
–1
–2
G = +1
5 VS = ±5V
f = 5MHz
4
INPUT
3
OUTPUT
2
1
0
–1
–2
–3
–4
–5
–3
0
5
10
15
20
25
TIME (ns)
04814-0-017
OUTPUT VOLTAGE (V)
20
2
0
2
15
3
LOAD = 1kΩ OR 150Ω
3
10
Figure 40. Large Signal Transient Response with Capacitive Load
1
0
5
TIME (ns)
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
–1
0
VS = ±5V
RS = 100Ω
G = +2
2
0
–4
Figure 37. Small Signal Transient Response with Capacitive Load
3
1
–3
04814-0-015
OUTPUT VOLTAGE (V)
0.10
04814-0-018
0.20
Figure 39. Large Signal Transient Response for Various Supplies
–6
0
20
40
60
80
100 120
TIME (ns)
140
160
Figure 42. Input Overdrive Recovery
Rev. A | Page 12 of 24
180
AD8045
0
G = +2
5 VS = ±5V
f = 5MHz
4
–10
3
2
VS = ±5V
2 × INPUT
POWER SUPPLY REJECTION (dB)
INPUT AND OUTPUT VOLTAGE (V)
6
OUTPUT
1
0
–1
–2
–3
–4
–20
–30
–PSR
–40
+PSR
–50
–60
–70
40
60
80
100 120
TIME (ns)
140
160
180
200
–80
0.01
Figure 43. Output Overdrive Recovery
COMMON-MODE REJECTION (dB)
10
1k
10k
100k
1M
FREQUENCY (Hz)
10M
100M
1G
04814-0-053
VOLTAGE NOISE (nV/ Hz)
–30
100
–50
–60
–70
–80
CLOSED-LOOP INPUT IMPEDANCE (Ω)
100M
04814-0-078
CURRENT NOISE (pA/ Hz)
10
10M
1
10
FREQUENCY (MHz)
100
1000
1000
Figure 47. Common-Mode Rejection vs. Frequency
100k
10k
100k
1M
FREQUENCY (Hz)
1000
–40
–90
0.1
100
1k
100
VS = ±5V
RF = 499Ω
Figure 44. Voltage Noise vs. Frequency
1
100
1
10
FREQUENCY (MHz)
Figure 46. Power Supply Rejection vs. Frequency
100
1
10
0.1
04814-0-020
20
04814-0-054
0
04814-0-045
–6
04814-0-062
–5
VS = ±5V
G = +1
10k
1000
100
10
1
10
100
FREQUENCY (MHz)
Figure 48. Input Impedance vs. Frequency
Figure 45. Current Noise vs. Frequency
Rev. A | Page 13 of 24
AD8045
G = +1
VS = ±5V
100
100
VS = ±5V
N = 450
X = 50µV
σ = 180µV
80
60
1
40
0.1
20
0.01
1
10
100
FREQUENCY (MHz)
1000
0
–900
–600
Figure 49. Output Impedance vs. Frequency
G = +10
VS = ±5V
RL = 1kΩ
80
46
THIRD-ORDER INTERCEPT (dBm)
0
VOFFSET (µV)
300
600
900
Figure 52. VOS Distribution for VS = ±5 V
50
48
–300
04814-0-063
COUNT
10
04814-0-055
CLOSED-LOOP OUTPUT IMPEDANCE (Ω)
1000
44
VS = +5V
N = 450
X = 540µV
σ = 195µV
60
COUNT
42
40
40
38
36
20
34
10
20
FREQUENCY (MHz)
30
40
0
–300
0
Figure 50. Third-Order Intercept vs. Frequency
0
0.10
–0.14
–0.16
PHASE
0.05
OFFSET VOLTAGE (µV)
–0.10
DIFFERENTIAL PHASE (Degrees)
0.15
100
0
10
–0.20
1
1500
NUMBER OF 150Ω LOADS
125
VS = +5V
–100
–300
–500
VS = ±5V
–700
–900
–0.18
04814-0-021
DIFFERENTIAL GAIN (%)
–0.06
–0.12
1200
300
0.20
–0.08
900
500
GAIN
G = +2
VS = ±5V
–0.04
600
VOFFSET (µV)
Figure 53. VOS Distribution for VS = +5 V
0.25
–0.02
300
04814-0-077
5
04814-0-040
30
04814-0-058
32
–1100
–40
–25
–10
5
20
35
50
65
TEMPERATURE (°C)
80
95
110
Figure 54. Offset Voltage vs. Temperature for Various Supplies
Figure 51. Differential Gain and Phase vs. Number of 150 Ω Loads
Rev. A | Page 14 of 24
AD8045
1.5
–1.0
–1.2
+VS – VOUT
OUTPUT SATURATION VOLTAGE (V)
–1.8
IB+, VS = ±5V
–2.0
–2.2
IB–, VS = 5V
–2.4
IB+, VS = 5V
–2.6
–3.0
–40
–25
–10
5
20
35
50
65
TEMPERATURE (°C)
80
95
110
125
VS = ±5V
–0.5
–1.0
–VS – VOUT
–1.5
0.1
Figure 55. Input Bias Current vs. Temperature for Various Supplies
4
RL = 1kΩ
3
2
–VS + VOUT
1.10
VOS (mV)
1
1.05
1.00
+VS – VOUT
RL = 1kΩ
–1
–2
–VS + VOUT
+VS – VOUT
0.95
0
RL = 150Ω
–3
–25
–10
5
20
35
50
65
TEMPERATURE (°C)
80
95
110
125
–4
–4
Figure 56. Output Saturation Voltage vs. Temperature for Various Supplies
–2
–1
0
VOUT (V)
1
2
3
4
Figure 59. Input Offset Voltage vs. Output Voltage for Various Loads
0.30
17.0
G = +2
VS = ±5V
VOUT = 2V p-p
RL = 150Ω
RF = 499Ω
0.20
16.5
VS = ±5V
SETTLING (%)
0.10
16.0
VS = 5V
15.5
15.0
0
–0.10
–0.20
–25
–10
5
20
35
50
65
TEMPERATURE (°C)
80
95
110
125
04814-0-056
14.5
–40
–3
04814-0-047
VS = 5V
04814-0-057
OUTPUT SATURATION VOLTAGE (V)
10
1.15
0.90
–40
SUPPLY CURRENT (mA)
1
LOAD (kΩ)
Figure 58. Output Saturation Voltage vs. Load for Various Supplies
1.20
VS = ±5V
VS = +5V
0
04814-0-059
–2.8
0.5
04814-0-044
IB–, VS = ±5V
–0.30
0
2.5
5.0
7.5
10.0 12.5
TIME (ns)
15.0
17.5
Figure 60. Short Term 0.1% Settling Time
Figure 57. Supply Current vs. Temperature for Various Supplies
Rev. A | Page 15 of 24
20.0
22.5
04814-0-046
INPUT BIAS CURRENT (µA)
–1.4
–1.6
1.0
AD8045
CIRCUIT CONFIGURATIONS
WIDEBAND OPERATION
The resistor at the output of the amplifier, labeled RSNUB, is used
only when driving large capacitive loads. Using RSNUB improves
stability and minimizes ringing at the output. For more information, see the Driving Capacitive Loads section.
RF
+VS
10µF
+
RG
RSNUB
AD8045
VOUT
0.1µF
10µF
+
–VS
04814-0-074
VIN
+VS
10µF
+
0.1µF
VIN
RG
RSNUB
AD8045
Figure 61. Noninverting Configuration
Rev. A | Page 16 of 24
VOUT
0.1µF
R = RG||RF
10µF
+
–VS
Figure 62. Inverting Configuration
0.1µF
RS
RF
04814-0-075
Figure 61 and Figure 62 show the recommended circuit
configurations for noninverting and inverting amplifiers. In
unity gain (G = +1) applications, RS helps to reduce high
frequency peaking. It is not needed for any other configurations.
For more information on layout, see the Printed Circuit Board
Layout section.
AD8045
THEORY OF OPERATION
The AD8045 is a high speed voltage feedback amplifier fabricated on ADI’s second generation eXtra Fast Complementary
Bipolar (XFCB) process. An H-bridge input stage is used to
attain a 1400 V/µs slew rate and low distortion in addition to a
low 3 nV/√Hz input voltage noise. Supply current and offset
voltage are laser trimmed for optimum performance.
RS
+ VOUT –
VIN
04814-0-009
RF
RG
FREQUENCY RESPONSE
The AD8045’s open-loop response over frequency can be
approximated by the integrator response shown in Figure 63.
Figure 64. Noninverting Configuration
DC ERRORS
Figure 65 shows the dc error contributions. The total output
error voltage is
VOUT
VOUT/VIN (dB)
VIN
⎛R +R ⎞
⎛R +R ⎞
VOUT (ERROR)= −I B+ RS ⎜ G F ⎟ + I B− RF + VOS ⎜ G F ⎟
R
G
⎠
⎝ RG ⎠
⎝
VOS
RS
fCROSSOVER
IB+
fCROSSOVER = 400MHz
f
10
100
FREQUENCY (MHz)
1000
RF
Figure 63. Open-Loop Response
RG
The closed-loop transfer function for the noninverting configuration is shown in Figure 64 and is written as
Figure 65. Amplifier DC Errors
The voltage error due to IB+ and IB− is minimized if RS = RF||RG.
To include the effects of common-mode and power supply
rejection, model VOS as
2 π × f CROSSOVER × (RG + RF )
VOUT
=
(RF + RG )s + 2 π × f CROSSOVER × RG
VIN
where:
VOS = VOS nom +
s is (2 πj)f.
fCROSSOVER is the frequency where the amplifier’s open-loop gain
equals 1 (0 dB).
DC gain is therefore
VOUT (RG + RF )
=
VIN
RG
Closed-loop −3 dB bandwidth equals
VOUT
RG
= f CROSSOVER ×
(RG + RF )
VIN
04814-0-010
0
1
+ VOUT –
IB–
04814-0-008
VOUT/VIN =
∆VS ∆VCM
+
PSR CMR
where:
Vos nom
is the offset voltage at nominal conditions.
ΔVS is the change in the power supply voltage from nominal
conditions.
PSR is the power supply rejection.
CMR is the common-mode rejection.
ΔVCM is the change in common-mode voltage from nominal
conditions.
The closed-loop bandwidth is inversely proportional to the
noise gain of the op amp circuit, (RF + RG)/RG. This simple
model can be used to predict the −3 dB bandwidth for noise
gains above +2. The actual bandwidth of circuits with noise
gains at or below +2 is higher due to the influence of other
poles present in the real op amp.
Rev. A | Page 17 of 24
AD8045
OUTPUT NOISE
Ven , IN+, and IN− are due to the amplifier. VR F , VRG , and
Figure 66 shows the contributors to the noise at the output of a
noninverting configuration.
VR S are due to the feedback network resistors. RG and RF, and
VEN
RS
source resistor, RS. Total output voltage noise, VOUT _ EN , is the
rms sum of all the contributions.
IEN+
+ VOUT –
VOUT _ EN =
IEN–
(Gn × Ven)2 + (IN + × RS × Gn )2 + (IN − × RF||RG × Gn )2 + 4kTR f + 4kTRG (Gn )2 + 4kTRS (Gn )2
VRF
where:
RF
RG
VRG
Figure 66. Amplifier DC Errors
04814-0-011
VRS
⎛ RF + RG ⎞
⎟.
⎝ RG ⎠
Gn is the noise gain ⎜
Ven is the op amp input voltage noise.
IN is the op amp input current noise.
Table 7 lists the expected output voltage noise spectral density
for several gain configurations.
Table 7. Noise and Bandwidth for Various Gains
Gain
+1
+2
+5
+10
−1
1
RL = 1 kΩ.
Rev. A | Page 18 of 24
RF
0
499
499
499
499
RG
−
499
124
56
499
RS
100
0
0
0
N/A
−3 dB
Bandwidth1
1 GHz
400 MHz
90 MHz
40 MHz
300 MHz
Output
Noise
(nV/√Hz)
3.3
7.4
16.4
31
7.4
AD8045
APPLICATIONS
This dc-coupled differential driver is best suited for ±5 V
operation in which optimum distortion performance is required
and the input signal is ground referenced.
511Ω
511Ω
33Ω
VIN
The traditional SOIC pinout has been slightly modified as well
to incorporate a dedicated feedback pin. Pin 1, previously a no
connect pin on the amplifier, is now a dedicated feedback pin. The
new pinout reduces parasitics and simplifies the board layout.
HIGH SPEED ADC DRIVER
When used as an ADC driver, the AD8045 offers results comparable to transformers in distortion performance. Many ADC
applications require that the analog input signal be dc-coupled
and operate over a wide frequency range. Under these requirements, operational amplifiers are very effective interfaces to
ADCs. An op amp interface provides the ability to amplify and
level shift the input signal to be compatible with the input range
of the ADC. Unlike transformers, operational amplifiers can be
operated over a wide frequency range down to and including dc.
Figure 67 shows the AD8045 as a dc-coupled differential driver
for the AD9244, a 14-bit 65 MSPS ADC. The two amplifiers are
configured in noninverting and inverting modes. Both amplifiers are set with a noise gain of +2 to provide better bandwidth
matching. The inverting amplifier is set for a gain of –1, while
the noninverting is set for a gain of +2. The noninverting input
is divided by 2 in order to normalize its output and make it
equal to the inverting output.
511Ω
511Ω
511Ω
511Ω
VINA
20pF
33Ω
511Ω
AD9244
VCML + VIN
AD8045
511Ω
VINB
2.5kΩ
0.1µF
100Ω
CML
0.1µF
1µF
OP27
Figure 67. High Speed ADC Driver
The outputs of the AD8045s are centered about the AD9244’s
common-mode range of 2.5 V. The common-mode reference
voltage from the AD9244 is buffered and filtered via the OP27
and fed to the noninverting resistor network used in the level
shifting circuit.
The spurious-free dynamic range (SFDR) performance is
shown in Figure 68. Figure 69 shows a 50 MHz single-tone FFT
performance.
120
100
AD8045
80
SFDR (dBc)
Existing applications that use the traditional SOIC pinout can
take full advantage of the outstanding performance offered by
the AD8045. An electrical insulator may be required if the SOIC
rests on the ground plane or other metal trace. This is covered
in more detail in the Exposed Paddle section of this data sheet.
In existing designs, which have Pin 1 tied to ground or to
another potential, simply lift Pin 1 of the AD8045 or remove the
potential on the Pin 1 solder pad. The designer does not need to
use the dedicated feedback pin to provide feedback for the
AD8045. The output pin of the AD8045 can still be used to provide feedback to the inverting input of the AD8045.
VCML – VIN
AD8045
04814-0-066
The AD8045 LFCSP package features Analog Devices new low
distortion pinout. The new pinout provides two advantages
over the traditional pinout. First, improved second harmonic
distortion performance, which is accomplished by the physical
separation of the noninverting input pin and the negative power
supply pin. Second, the simplification of the layout due to the
dedicated feedback pin and easy routing of the gain set resistor
back to the inverting input pin. This allows a compact layout,
which helps to minimize parasitics and increase stability.
Rev. A | Page 19 of 24
60
40
20
0
1
10
INPUT FREQUENCY (MHz)
Figure 68. SFDR vs. Frequency
100
04814-0-067
LOW DISTORTION PINOUT
AD8045
0
–20
–40
fc =
–60
1
2πRC
The quality factor, or Q, is shown in the equation
–80
Q=
–120
0
5
10
15
20
FREQUENCY (MHz)
25
30
04814-0-068
–100
1
3−K
The gain, or K, of the circuits are
First Stage K =
Figure 69. Single-Tone FFT, FIN = 50 MHz, Sample Rate = 65 MSPS
Shown in the First Nyquist Zone
Active filters are used in many applications such as antialiasing
filters and high frequency communication IF strips.
With a 400 MHz gain bandwidth product and high slew rate,
the AD8045 is an ideal candidate for active filters. Figure 70
shows the frequency response of the 90 MHz LPF. In addition to
the bandwidth requirements, the slew rate must be capable of
supporting the full power bandwidth of the filter. In this case, a
90 MHz bandwidth with a 2 V p-p output swing requires at least
1200 V/µs. This performance is achievable only at 90 MHz
because of the AD8045’s wide bandwidth and high slew rate.
The circuit shown in Figure 73 is a 90 MHz, 4-pole, Sallen-Key,
LPF. The filter comprises two identical cascaded Sallen-Key LPF
sections, each with a fixed gain of G = +2. The net gain of the
filter is equal to G = +4 or 12 dB. The actual gain shown in
Figure 70 is only 6 dB. This is due to the output voltage being
divided in half by the series matching termination resistor, RT,
and the load resistor.
Resistor values are kept low for minimal noise contribution,
offset voltage, and optimal frequency response. Due to the low
capacitance values used in the filter circuit, the PCB layout and
minimization of parasitics is critical. A few picofarads can detune
the filters corner frequency, fc. The capacitor values shown in
Figure 73 actually incorporate some stray PCB capacitance.
Capacitor selection is critical for optimal filter performance.
Capacitors with low temperature coefficients, such as NPO
ceramic capacitors and silver mica, are good choices for filter
elements.
20
10
0
–10
–20
GAIN (dB)
90 MHZ ACTIVE LOW-PASS FILTER (LPF)
R8
R3
+ 1, Second Stage K =
+1
R7
R4
–30
–40
–50
–60
–70
–80
–90
0.1
1
10
FREQUENCY (MHz)
100
Figure 70. 90 MHz Low-Pass Filter Response
Rev. A | Page 20 of 24
1000
04814-0-006
DISTORTION (dBc)
Setting the resistors and capacitors equal to each other greatly
simplifies the design equations for the Sallen-Key filter. The
corner frequency, or −3 dB frequency, can be described by the
equation
AIN = –1dBFS
SNR = 69.9dBc
SFDR = 65.3dBc
AD8045
M4.00ns
A CH1
0.00V
CH1 500mV
Figure 71. Small Signal Transient Response of 90 MHz LPF
C3
7.1pF
10µF
+5V
INPUT
RT
49.9Ω
R1
249Ω
A CH1
10µF
0.1µF
R2
249Ω
U1
C2
7.1pF
0.1µF
R6
249Ω
10µF
R5
249Ω
U1
C4
7.1pF
RT
49.9Ω
10µF
0.1µF
0.1µF
–5V
R4
499Ω
R3
499Ω
0.00V
Figure 72. Large Signal Transient Response of 90 MHz LPF
C1
7.1pF
+5V
M4.00ns
OUTPUT
R9
24.9Ω
C5
5pF
–5V
R7
499Ω
R8
499Ω
Figure 73. 4-Pole, 90 MHz, Sallen-Key Low-Pass Filter
Rev. A | Page 21 of 24
04814-0-005
CH1 50.0mV
04814-0-070
1
04814-0-069
1
AD8045
PRINTED CIRCUIT BOARD LAYOUT
Laying out the printed circuit board (PCB) is usually the last
step in the design process and often proves to be one of the
most critical. A brilliant design can be rendered useless because
of a poor or sloppy layout. Since the AD8045 can operate into
the RF frequency spectrum, high frequency board layout considerations must be taken into account. The PCB layout, signal
routing, power supply bypassing, and grounding all must be
addressed to ensure optimal performance.
SIGNAL ROUTING
The AD8045 LFCSP features the new low distortion pinout with
a dedicated feedback pin and allows a compact layout. The
dedicated feedback pin reduces the distance from the output to
the inverting input, which greatly simplifies the routing of the
feedback network.
When laying out the AD8045 as a unity gain amplifier, it is recommended that a short, but wide, trace between the dedicated
feedback pin and the inverting input to the amplifier be used to
minimize stray parasitic inductance.
To minimize parasitic inductances, ground planes should be
used under high frequency signal traces. However, the ground
plane should be removed from under the input and output pins
to minimize the formation of parasitic capacitors, which
degrades phase margin. Signals that are susceptible to noise
pickup should be run on the internal layers of the PCB, which
can provide maximum shielding.
POWER SUPPLY BYPASSING
Power supply bypassing is a critical aspect of the PCB design
process. For best performance, the AD8045 power supply pins
need to be properly bypassed.
A parallel connection of capacitors from each of the power
supply pins to ground works best. Paralleling different values
and sizes of capacitors helps to ensure that the power supply
pins “see” a low ac impedance across a wide band of frequencies.
This is important for minimizing the coupling of noise into the
amplifier. Starting directly at the power supply pins, the smallest
value and sized component should be placed on the same side
of the board as the amplifier, and as close as possible to the
amplifier, and connected to the ground plane. This process
should be repeated for the next larger value capacitor. It is
recommended for the AD8045 that a 0.1 µF ceramic 0508 case
be used. The 0508 offers low series inductance and excellent
high frequency performance. The 0.1 µF case provides low
impedance at high frequencies. A 10 µF electrolytic capacitor
should be placed in parallel with the 0.1 µF. The 10 µf capacitor
provides low ac impedance at low frequencies. Smaller values
of electrolytic capacitors may be used depending on the circuit
requirements. Additional smaller value capacitors help to
provide a low impedance path for unwanted noise out to higher
frequencies but are not always necessary.
Placement of the capacitor returns (grounds), where the capacitors enter into the ground plane, is also important. Returning
the capacitors grounds close to the amplifier load is critical for
distortion performance. Keeping the capacitors distance short,
but equal from the load, is optimal for performance.
In some cases, bypassing between the two supplies can help to
improve PSRR and to maintain distortion performance in
crowded or difficult layouts. It is brought to the designer’s
attention here as another option to improve performance.
Minimizing the trace length and widening the trace from the
capacitors to the amplifier reduce the trace inductance. A series
inductance with the parallel capacitance can form a tank circuit,
which can introduce high frequency ringing at the output. This
additional inductance can also contribute to increased distortion due to high frequency compression at the output. The use
of vias should be minimized in the direct path to the amplifier
power supply pins since vias can introduce parasitic inductance,
which can lead to instability. When required, use multiple large
diameter vias because this lowers the equivalent parasitic
inductance.
GROUNDING
The use of ground and power planes is encouraged as a method
of proving low impedance returns for power supply and signal
currents. Ground and power planes can also help to reduce stray
trace inductance and to provide a low thermal path for the
amplifier. Ground and power planes should not be used under
any of the pins of the AD8045. The mounting pads and the
ground or power planes can form a parasitic capacitance at the
amplifiers input. Stray capacitance on the inverting input and
the feedback resistor form a pole, which degrades the phase
margin, leading to instability. Excessive stray capacitance on the
output also forms a pole, which degrades phase margin.
Rev. A | Page 22 of 24
AD8045
The AD8045 features an exposed paddle, which lowers the
thermal resistance by 25% compared to a standard SOIC plastic
package. The exposed paddle of the AD8045 is internally connected to the negative power supply pin. Therefore, when laying
out the board, the exposed paddle must either be connected to
the negative power supply or left floating (electrically isolated).
Soldering the exposed paddle to the negative power supply metal
ensures maximum thermal transfer. Figure 74 and Figure 75 show
the proper layout for connecting the SOIC and LFCSP exposed
paddle to the negative supply.
THERMAL CONDUCTIVE INSULATOR
04814-0-072
EXPOSED PADDLE
Figure 76. SOIC with Thermal Conductive Pad Material
The thermal pad provides high thermal conductivity but
isolates the exposed paddle from ground or other potential. It is
recommended, when possible, to solder the paddle to the negative power supply plane or trace for maximum thermal transfer.
Note that soldering the paddle to ground shorts the negative
power supply to ground and can cause irreparable damage to
the AD8045.
04814-0-071
DRIVING CAPACITIVE LOADS
Figure 74. SOIC Exposed Paddle Layout
04814-0-073
The use of thermal vias or “heat pipes” can also be incorporated
into the design of the mounting pad for the exposed paddle.
These additional vias help to lower the overall theta junction to
ambient (θJA). Using a heavier weight copper on the surface to
which the amplifier’s exposed paddle is soldered can greatly
reduce the overall thermal resistance “seen” by the AD8045.
Figure 75. LFCSP Exposed Paddle Layout
For existing designs that want to incorporate the AD8045,
electrically isolating the exposed paddle is another option. If the
exposed paddle is electrically isolated, the thermal dissipation is
primarily through the leads, and the thermal resistance of the
package now approaches 125°C/W, the standard SOIC θJA.
However, a thermally conductive and electrically isolated pad
material may be used. A thermally conductive spacer, such as
the Bergquist Company’s Sil-Pad, is an excellent solution to this
problem. Figure 76 shows a typical implementation using
thermal pad material.
In general, high speed amplifiers have a difficult time driving
capacitive loads. This is particularly true in low closed-loop
gains, where the phase margin is the lowest. The difficulty arises
because the load capacitance, CL, forms a pole with the output
resistance, RO, of the amplifier. The pole can be described by the
equation
fP =
1
2πRO C L
If this pole occurs too close to the unity gain crossover point,
the phase margin degrades. This is due to the additional phase
loss associated with the pole.
The AD8045 output can drive 18 pF of load capacitance directly,
in a gain of +2 with 30% overshoot, as shown in Figure 37.
Larger capacitance values can be driven but must use a snubbing resistor (RSNUB) at the output of the amplifier, as shown in
Figure 61 and Figure 62. Adding a small series resistor, RSNUB,
creates a zero that cancels the pole introduced by the load
capacitance. Typical values for RSNUB can range from 25 Ω to
50 Ω. The value is typically arrived at empirically and based on
the circuit requirements.
Rev. A | Page 23 of 24
AD8045
OUTLINE DIMENSIONS
5.00 (0.197)
4.90 (0.193)
4.80 (0.189)
4.00 (0.157)
3.90 (0.154)
3.80 (0.150)
8
BOTTOM VIEW
(PINS UP)
2.29 (0.092)
5
1
2.29 (0.092)
6.20 (0.244)
6.00 (0.236)
5.80 (0.228)
TOP VIEW
4
1.27 (0.05)
BSC
0.50 (0.020)
× 45°
0.25 (0.010)
1.75 (0.069)
1.35 (0.053)
0.25 (0.0098)
0.10 (0.0039)
COPLANARITY
SEATING
0.10
PLANE
0.51 (0.020)
0.31 (0.012)
8°
0.25 (0.0098) 0° 1.27 (0.050)
0.40 (0.016)
0.17 (0.0068)
COMPLIANT TO JEDEC STANDARDS MS-012
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
Figure 77. 8-Lead Standard Small Outline Package with Exposed Pad [SOIC_N_EP], Narrow Body (RD-8-1)—Dimensions shown in millimeters and (inches)
3.00
BSC SQ
0.50
0.40
0.30
0.60 MAX
0.45
1
8
PIN 1
INDICATOR
0.90
0.85
0.80
SEATING
PLANE
2.75
BSC SQ
TOP
VIEW
0.50
BSC
0.25
MIN
0.80 MAX
0.65 TYP
12° MAX
1.50
REF
EXPOSED
PAD
(BOTTOM VIEW)
5
PIN 1
INDICATOR
1.90
1.75
1.60
4
1.60
1.45
1.30
0.05 MAX
0.02 NOM
0.30
0.23
0.18
0.20 REF
Figure 78. 8-Lead Lead Frame Chip Scale Package [LFCSP], 3 mm × 3 mm Body (CP-8-2)—Dimensions shown in millimeters
ORDERING GUIDE
Model
AD8045ARD
AD8045ARD-REEL
AD8045ARD-REEL7
AD8045ARDZ1
AD8045ARDZ-REEL1
AD8045ARDZ-REEL71
AD8045ACP-R2
AD8045ACP-REEL
AD8045ACP-REEL7
AD8045ACPZ-R21
AD8045ACPZ-REEL1
AD8045ACPZ-REEL71
1
Minimum
Ordering Quantity
1
2,500
1,000
1
2,500
1,000
250
5,000
1,500
250
5,000
1,500
Temperature Range
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
–40°C to +125°C
Z = Pb-free part.
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D04814–0–9/04(A)
Rev. A | Page 24 of 24
Package Description
8-Lead SOIC_N_EP
8-Lead SOIC_N_EP
8-Lead SOIC_N_EP
8-Lead SOIC_N_EP
8-Lead SOIC_N_EP
8-Lead SOIC_N_EP
8-Lead LFCSP
8-Lead LFCSP
8-Lead LFCSP
8-Lead LFCSP
8-Lead LFCSP
8-Lead LFCSP
Package
Option
RD-8-1
RD-8-1
RD-8-1
RD-8-1
RD-8-1
RD-8-1
CP-8-2
CP-8-2
CP-8-2
CP-8-2
CP-8-2
CP-8-2
Branding
H8B
H8B
H8B
H8B
H8B
H8B