Allegro A8580 Pwm/pfmn mode control input pin Datasheet

A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
Not for New Design
This part is in production but has been determined to be NOT FOR
NEW DESIGN. Sale of this part is currently restricted to existing
customer programs already using the part. The part should not be
purchased for new programs or designed into new applications.
Samples are no longer available.
Date of status change: September 1, 2016
Recommended Substitutions: A8589, A8590, or A8591
For existing customer transition, and for new customers or new applications, contact Allegro Sales.
NOTE: For detailed information on purchasing options, contact your
local Allegro field applications engineer or sales representative.
Allegro MicroSystems, LLC reserves the right to make, from time to time, revisions to the anticipated product life cycle plan
for a product to accommodate changes in production capabilities, alternative product availabilities, or market demand. The
information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its use; nor for any infringements of patents or other rights of third parties which may result from its use.
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
FEATURES AND BENEFITS
•
•
•
•
•
•
•
•
•
•
•
DESCRIPTION
Designed to provide the power supply requirements of next
generation car audio and infotainment systems, the A8580
provides all the control and protection circuitry to produce a
high current regulator with ±1.0% output voltage accuracy. The
A8580 employs pulse frequency modulation (PFM) to draw
less than 50 µA from 12VIN while supplying 3.3 V/40 µA.
After startup, the A8580 operates down to at least 3.6 VIN
(VIN falling).
Automotive AEC-Q100 qualified
Withstands surge voltages up to 40 V
Operates as low as 3.4 VIN (typ) with VIN decreasing
Utilizes pulse frequency modulation (PFM) to draw only
tens of microamperes from VIN while maintaining
keep-alive VOUT
PWM/PFMn mode control input pin
Delivers up to 2.5 A of output current with integrated
110 mΩ high voltage MOSFET
SLEEPn input pin commands ultra-low current
shutdown mode
Adjustable output voltage with ±1.0% accuracy from 0°C
to 85°C, ±1.5% from –40°C to 150°C
Programmable switching frequency: 250 kHz to 2.4 MHz
Synchronization capability: applying a clock input to the
PWM/PFMn input pin will increase the PWM frequency
Active low, power-on reset (NPOR) open-drain output
Features of the A8580 include a PWM/PFMn mode control
input to enable PWM (logic high) or PFM (logic low). If
the PWM/PFMn input is driven by an external clock signal
higher than the base frequency ( fOSC ) the PWM frequency
synchronizes to the incoming clock frequency. The SLEEPn
input pin commands an ultra-low current shutdown mode
requiring less than 5 µA for internal circuitry and 10 µA (max)
for MOSFET leakage at 16 VIN , 85ºC. The A8580 has external
compensation to accommodate a wide range of frequencies
and external components, and provides a power-on reset
(NPOR) signal validated by the output voltage.
Continued on the next page…
Continued on the next page…
Package: 16-pin TSSOP with exposed
thermal pad (suffix LP)
APPLICATIONS:
• Automotive:
□□ Instrument
clusters
□□ Audio Systems
• Home Audio
Not to scale
V IN
CIN1
4.7 µF
50 V
1210
CP
15 pF
CIN2
4.7 µF
50 V
1210
RZ
26.1 kΩ
CZ
560 pF
CIN3
0.47 µF
100 V
0805
CSS
22 nF
CIN4
47 nF
50 V
0603
1
2
5
13
9
3
8
12
RFSET
59 kΩ
VIN
VIN
A8580
GND
GND
COMP
SS
FSET
VREG
SW
SW
BOOT
BIAS
FB
C1
1 µF
EN
4
Mode
6
15
14
CBOOT
47 nF
16
LO 8.2 µH
D1
2 A/40 V
SMP
11
10
RFB2
47 kΩ
RFB1 147 kΩ
CFB
10 pF
VOUT
CO1
22 µF
16 V X7R
1210
CO2
22 µF
16 V X7R
1210
CO3
0.47µF
100 V
0805
CO4
47 nF
50 V
0603
COUT = 38.2 µF to 42.4 µF total
at 3.3 V BIAS and 10% tolerance
3.3 V
R PU
10 kΩ
SLEEPn
PWM/PFMn
□□ Navigation
□□ HVAC
NPOR
7
NPOR
Typical Application Diagram
Typical application schematic, configured for VIN = 12 V , VOUT = 3.3 V, IOUT = 2.5 A at 425 kHz (see Applications Information section
for configuration producing VOUT = 5.0 V)
A8580-DS, Rev. 8
December 5, 2016
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
Features and Benefits (continued)
•
•
•
•
Pre-bias startup capable, VOUT will not cause a reset
External compensation for maximum flexibility
Stable with ceramic or electrolytic output capacitors
Excellent set of protection features to satisfy the most
demanding applications
• Overvoltage, pulse-by-pulse current limit, hiccup mode short
circuit, and thermal protection
• Robust FMEA, with pin open/short and component faults
• Thermally enhanced, surface mount package
Description (continued)
Extensive protection features of the A8580 include pulse-by-pulse
current limit, hiccup mode short circuit protection, open/short
asynchronous diode protection, BOOT open/short voltage protection,
VIN undervoltage lockout, VOUT overvoltage protection and thermal
shutdown.
The A8580 is supplied in a low profile 16-pin TSSOP package
with exposed power pad (suffix LP). It is lead (Pb) free, with 100%
matte-tin leadframe plating.
Selection Guide
Part Number
Operating Ambient Temperature Range
TA, (°C)
Packing
A8580KLPTR-T
–40 to 125
4000 pieces per 13-in. reel
Contact Allegro for additional packing options.
Table of Contents
Specifications
Absolute Maximum Ratings
Thermal Characteristics
Functional Block Diagram
Pin-out Diagram and Terminal List
Electrical Characteristics
Characteristic Performance
Functional Description
3
3
3
4
5
6
10
12
Overview
12
Reference Voltage
12
PWM Switching Frequency
12
SLEEPn Input
12
PWM/PFMn Input and PWM Synchronization
13
BIAS Input Functionality, Ratings, and Connections13
Transconductance Error Amplifier
13
Slope Compensation
14
Current Sense Amplifier
14
Power MOSFETs
14
BOOT Regulator
14
Pulse Width Modulation (PWM) Mode
14
Low-IQ Pulse Frequency Modulation (PFM) Mode 15
Reduced Current (Low-IP) PWM Mode
17
Soft Start (Startup) and Inrush Current Control
17
Pre-Biased Startup
18
Not Power-On Reset (NPOR) Output
18
Protection Features
19
Undervoltage Lockout (UVLO)
19
Pulse-by-Pulse Overcurrent Protection (OCP) 19
Overcurrent Protection (OCP) and Hiccup Mode 19
20
BOOT Capacitor Protection
Asynchronous Diode Protection
20
Output Overvoltage Protection (OVP)
20
Pin-to-Ground and Pin-to-Pin Short Protections 21
Thermal Shutdown (TSD)
21
Application Information
Design and Component Selection
Setting the Output Voltage (VOUT)
PWM Base Switching Frequency (fOSC, RFSET)
Output Inductor (LO)
Output Capacitors
Low‑IQ PFM Output Voltage Ripple Calculation
Input Capacitors
Asynchronous Diode (D1)
Bootstrap Capacitor
Soft Start and Hiccup Mode Timing (CSS)
Compensation Components (RZ, CZ, and CP)
A Generalized Tuning Procedure
Power Dissipation and Thermal Calculations
PCB Component Placement and Routing
Typical Applications Schematics
Package Outline Drawing
25
25
25
25
26
27
28
28
29
29
29
30
32
35
36
38
39
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
2
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
SPECIFICATIONS
Absolute Maximum Ratings*
Characteristic
Symbol
Notes
Rating
Unit
–0.3 to 40
V
–0.3 to VIN + 0.3
V
–1.0 to VIN + 3
V
Continuous
VSW – 0.3 to VSW + 5.5
V
BOOT OV Fault Condition
VSW – 0.3 to VSW + 7.0
V
VIN, SLEEPn, SS Pin Voltage
SW Pin Voltage
BOOT Pin Voltage
BIAS Pin Voltage
VSW
VBOOT
VBIAS
Continuous (minimum limit is a function of
temperature)
t < 50 ns
Continuous
–0.3 to 5.5
BIAS OV Fault Condition
All Other Pins Voltage
–0.3 to 6
V
–0.3 to 5.5
V
Operating Ambient Temperature
TA
–40 to 125
ºC
Maximum Junction Temperature
TJ(max)
150
ºC
Tstg
–55 to 150
ºC
Storage Temperature
K temperature range
*Operation at levels beyond the ratings listed in this table may cause permanent damage to the device. The Absolute Maximum ratings are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the Electrical Characteristics table is not implied. Exposure to Absolute Maximumrated conditions for extended periods may affect device reliability.
Thermal Characteristics may require derating at maximum conditions, see application information
Characteristic
Symbol
Test Conditions*
Value
Unit
Package Thermal Resistance
RθJA
On 4-layer PCB based on JEDEC standard
34
ºC/W
*Additional thermal information available on the Allegro website.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
3
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
VIN
BIAS
VREG
–
–
3.8V
3.4V
LDO OFF
V BIAS >
LDOOUT + 50 mV
+
BOOT REG
5.0V
UVLO
B OOT
OFF
1.205 V
BG
–
DELAY
103µs↓
2.90V
Digital
POR
+
2V, 4.1V
VPWMOFFS
400 mV
BOOT
OFF
P WM
P WM
+
SE
s leep
P WM
fSYNC
DELAY
PWM/PFMn
minOff
750mA
P FM
–
GCSA
S
Q
R
Q
110 mΩ
Q
fSW
Current
Comp
VREG
PFM
Controller
2048 ↓
IFB
Error
Amplifier
s leep
P WM
SW
10 Ω
BOOT > 4.1V
DIODEOK
FB < 0.8 V
FB
BOOT
+
Low IP P WM
CLK
FB < 0.4V
f OSC
FB < 0.2V
FSET
BOOT
FAULT
2x
ISENSE
fSYNC>1.2 x fOSC
F
F/2
F/4
OC
250 mA
EN
B OOT
RE G
Q
VREG
SLEEPn
REGOV
+
5.75V
V BIAS rising
LDO
CLAMP
OCL
V REF 800 mV
COMP
s leep
P WM
+
s leep
P WM
–
Soft Start
Offset 400 mV
20µA
SS
5µA
FB < 700 mV (PFM)
FB > 880 mV (PWM)
OCL
DIODEOK
BOOT FAULT
REGOV
UVLO
POR
TSD
FAULT
LOGIC
(See Fault
Table)
s leep
P WM
HIC SET
HICCUP
LOGIC
HICCUP
1 kΩ
2 kΩ
HIC RST
PULL DOWN
BOOT OFF
NPOR
DELAY
FB <700 mV (PFM )
FB >880 mV (PWM)
FB <740 mV (PWM )
7.5ms↓
Functional Block Diagram
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
4
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
Pin-out Diagram and Terminal List Table
VIN 1
16 BOOT
VIN 2
15 SW
SS 3
14 SW
SLEEPn 4
GND 5
PWM/PFMn 6
NPOR 7
FSET 8
PAD
13 GND
12 VREG
11 BIAS
10 FB
9 COMP
Package LP, 16-pin TSSOP with Exposed Thermal Pad
Pin-out Diagram
Terminal List Table
Name
Number
11
BIAS
Bias input, supplies internal circuitry.
16
BOOT
High-side gate drive boost input. This pin supplies the drive for the high-side N-channel MOSFET. Connect a 47 nF
ceramic capacitor from BOOT to SW.
9
COMP
Output of the error amplifier and compensation node for the current mode control loop. Connect a series RC network from
this pin to GND for loop compensation. See the Design and Component Selection section of this datasheet for further
details.
10
FB
Feedback (negative) input to the error amplifier. Connect a resistor divider from the regulator output, VOUT, to this pin to
program the output voltage.
8
FSET
Frequency setting pin. A resistor, RFSET, from this pin to GND sets the base PWM switching frequency (fOSC). See the
Design and Component Selection section for information on determining the value of RFSET.
5, 13
GND
Ground pins.
7
NPOR
Active low, power-on reset output signal. This pin is an open drain output that transitions from low to high impedance after
the output has maintained regulation for tD(NPOR).
–
PAD
Exposed pad of the package providing enhanced thermal dissipation. This pad must be connected to the ground plane(s)
of the PCB with at least 6 vias, directly in the pad land.
6
Function
Sets operating output mode (fSW). Setting this pin low forces Low-IQ PFM mode (fSW set by load). Setting this pin high
PWM/PFMn forces PWM mode switching at the the base frequency (fOSC), set by RFSET. Applying an external clock input to this pin
forces synchronization of PWM to the clock input rate (fSYNC), at a rate higher than fOSC. SLEEPn low overrides this pin.
Setting this pin low forces sleep mode (very low current shutdown mode: VOUT = 0 V). This pin must be set high to enable
the A8580. If the application does not require a sleep mode, then this pin can be tied directly to VIN. Do not float this pin.
4
SLEEPn
3
SS
Soft start and hiccup pin. Connect a capacitor, CSS , from this pin to GND to set soft start mode duration. The capacitor
also determines the hiccup period during overcurrent.
14, 15
SW
The source of the high-side N-channel MOSFET. The external free-wheeling diode (D1) and output inductor (LO) should
be connected to this pin. Both D1 and LO should be placed close to this pin and connected with relatively wide traces.
1, 2
VIN
Power input for the control circuits and the drain of the high-side N-channel MOSFET. Connect this pin to a power supply
providing from 4.0 to 35 V. A high quality ceramic capacitor should be placed and grounded very close to this pin.
12
VREG
Internal voltage regulator bypass capacitor pin. Connect a 1 µF ceramic capacitor from this pin to ground and place it very
close to the A8580.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
5
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
ELECTRICAL CHARACTERISTICS: valid at 4.0 V ≤ VIN ≤ 35 V, –40°C ≤ TA = TJ ≤ 150°C; unless otherwise specified
Characteristic
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
Input Voltage
Input Voltage Range1
4.0
–
35
V
VIN UVLO Start
VINUV(ON)
VIN
VIN rising
3.6
3.8
4.0
V
VIN UVLO Stop
VINUV(OFF) VIN falling
3.2
3.4
3.6
V
VIN UVLO Hysteresis
VINUV(HYS)
–
400
–
mV
Input Supply Current
Sleep Mode Input Supply Current2,6
PWM Mode Input Supply
Current2
Low-IQ PFM Input Supply Current2,3
VSLEEPn ≤ 0.5 V, TJ = 85°C, VIN =16 V
–
5
15
µA
VSLEEPn ≤ 0.5 V, TJ = 85°C, VIN = 35 V
–
7
25
µA
IIN(PWM)
VBIAS > 3.2 V, IOUT = 0 mA
–
2.5
5.0
mA
ILO_IQ(0)
VIN = 12 V, VOUT = 3.3 V, VPWMPFMn ≤ 0.8 V,
IOUT = 40 µA, TA =25ºC, components selected
per table 3
–
–
50
µA
ILO_IQ(1)
VIN = 12 V, VOUT = 5.0 V, VPWMPFMn ≤ 0.8 V,
IOUT = 200 µA, TA = 25ºC, components selected
per table 3
–
–
250
µA
ILO_IQ(2)
VIN = 12 V, VOUT = 6.5 V, VPWMPFMn ≤ 0.8 V,
IOUT = 1 mA, TA = 25ºC, components selected
per table 3
–
–
750
µA
0ºC < TJ < 85ºC, VIN ≥ 4.1 V, VFB = VCOMP
792
800
808
mV
–40ºC < TJ < 150ºC, VIN ≥ 4.1 V, VFB = VCOMP
788
800
812
mV
3.0 V < VBIAS < 5.5 V and ILO_IQ specifications
satisfied
3.3
–
6.5
V
VBIAS = GND, PWM only, no PFM mode
0.8
–
10
V
TA = 85°C, DCRLO ≤ 75 mΩ, VIN = 3.7 V,
IOUT = 1 A, fSW = 425 kHz
3.21
3.3
–
V
TA = 85°C, DCRLO ≤ 75 mΩ, VIN = 5.6 V,
IOUT = 1 A, fSW = 425 kHz
4.96
5.0
–
V
TA = 85°C, DCRLO ≤ 50 mΩ, VIN = 4.8 V,
IOUT = 1 A, fSW = 2 MHz
3.28
3.3
–
V
TA = 85°C, DCRLO ≤ 50 mΩ, VIN = 7.1 V,
IOUT = 1 A, fSW = 2 MHz
4.94
5.0
–
V
8 V < VIN < 12 V, components selected per
table 3
–
30
65
mVPP
fSW < 750 kHz
–
750
–
mAPEAK
fSW > 750 kHz
–
850
–
mAPEAK
400
550
700
mA
IIN(SLEEP)
Voltage Regulation
Feedback Voltage Accuracy4
Low-IQ PFM Mode Output Voltage
Setting Range1,3
PWM Output Voltage Setting Range3
Output Dropout Voltage3
VFB
VOUT(LO_IQ)
VOUT
VOUT(SAT)
Low-IQ PFM Mode Ripple Voltage3
ΔVOUT(LO_IQ)
Low-IQ PFM Mode Peak Current
Threshold
IPEAK(LO_IQ)
Low-IQ PFM Mode DC Load Current3
IOUT(LO_IQ)
Maximum load to maintain ΔVOUT(LO_IQ) ,
components selected per table 3
Continued on the next page…
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
6
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
ELECTRICAL CHARACTERISTICS (continued): valid at 4.0 V ≤ VIN ≤ 35 V, –40°C ≤ TA = TJ ≤ 150°C; unless otherwise specified
Characteristic
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
–38
–
–16
nA
–
65
–
dB
Error Amplifier
Feedback Input Bias Current2
Open Loop Voltage Gain
Transconductance
IFB
AVOL
400 mV < VFB
550
750
950
μA/V
0 V < VFB < 400 mV
275
375
475
μA/V
VCOMP = 1.2 V
–
±75
–
μA
FAULT = 1 or HICCUP = 1
–
1
–
KΩ
PWMOFFS VCOMP level required for 0% duty cycle
–
400
–
mV
–
95
135
ns
–
95
130
ns
gm
Output Current
IEA
COMP Pull-Down Resistance
VCOMP = 1.2 V
RCOMP
Pulse Width Modulation (PWM)
PWM Ramp Offset
Minimum Controllable PWM On-Time
tON(MIN)PWM
Minimum Switch Off-Time
tOFF(MIN)PWM
COMP to SW Current Gain
gmPOWER
Slope Compensation3
SE
12 V < VIN < 16 V, IOUT = 1 A,
VBOOT – VSW = 4.5 V
–
2.85
–
A/V
fOSC = 2.44 MHz
2.1
3.0
3.9
A/μs
fOSC = 1.00 MHz
0.60
0.91
1.2
A/μs
fOSC = 252 kHz
0.14
0.20
0.26
A/μs
MOSFET Parameters1
High-Side MOSFET On-Resistance5
High-Side MOSFET
Leakage2,6
SW Node Slew Rate3
Low-Side MOSFET On-Resistance5
RDS(on)HS
Ilkg(HS)
SRSW
RDS(on)LS
TJ =25ºC, VBOOT – VSW = 4.5 V, IDS = 0.4 A
–
110
125
mΩ
TJ =150ºC, VBOOT – VSW = 4.5 V, IDS = 0.4 A
–
190
215
mΩ
TJ < 85°C, VSLEEPn ≤ 0.5 V, VSW = 0 V,
VIN = 16 V
–
–
10
µA
TJ ≤ 150°C, VSLEEPn ≤ 0.5 V, VSW = 0 V,
VIN = 16 V
–
60
150
µA
12 V < VIN < 16 V
–
0.72
–
V/ns
TJ = 25ºC, VIN ≥ 6 V, IDS = 0.1 A
–
–
10
Ω
RFSET = 8.06 kΩ, VPWM/PFMn= high
2.20
2.44
2.70
MHz
RFSET = 23.7 kΩ, VPWM/PFMn= high
0.90
1.00
1.10
MHz
RFSET = 102 kΩ, VPWM/PFMn= high
–
252
–
kHz
PWM Switching Frequency
Base PWM Switching Frequency
fOSC
PWM Synchronization Timing
Synchronization Frequency Range
fSYNC(MULT)
1.2 ×
fOSC(typ)
–
1.5 ×
fOSC(typ)
−
Synchronized PWM Frequency
fSYNC(PWM)
–
–
2.9
MHz
Synchronization Input Duty Cycle
DSYNC
–
–
80
%
Synchronization Input Pulse Width
twSYNC
200
–
–
ns
Synchronization Input Rise Time3
trSYNC
–
10
15
ns
Synchronization Input Fall Time3
tfSYNC
–
10
15
ns
Continued on the next page…
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
7
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
ELECTRICAL CHARACTERISTICS (continued): valid at 4.0 V ≤ VIN ≤ 35 V, –40°C ≤ TA = TJ ≤ 150°C; unless otherwise specified
Characteristic
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
–
–
2.0
V
PWM/PFMn Pin Input Thresholds
PWM/PFMn High Threshold
PWM/PFMn Low Threshold
PWM/PFMn Hysteresis
VPWMPFMn(H)
VPWMPFMn(L)
VPWMPFMnhys
PWM/PFMn Input Resistance
RPWMPFMn
Low-IQ PFM Transition Delay
tD(LO_IQ)
3.0 V < VBIAS < 3.6 V, VPWMPFMn rising
4.5 V < VBIAS < 5.5 V, VPWMPFMn rising
–
–
2.6
V
3.0 V < VBIAS < 3.6 V, VPWMPFMn falling
0.8
–
–
V
4.5 V < VBIAS < 5.5 V, VPWMPFMn falling
1.2
–
–
V
–
200
–
mV
3.0 V < VBIAS < 3.6 V, VPWMPFM(H) – VPWMPFM(L)
4.5 V < VBIAS < 5.5 V, VPWMPFM(H) – VPWMPFM(L)
–
400
–
mV
120
200
280
kΩ
PWM/PFMn = low, VSS > HIC/PFMEN ,
NPOR = high
–
2048
–
counts
fOSC < 1.5 MHz
–
435
–
ns
fOSC > 1.5 MHz
–
275
–
ns
–
4.1
–
µs
PFM Mode Timing
Constant PFM Off-Time
tOFF(PFM)
Maximum PFM On-Time
tON(PFM)MAX
SLEEPn Pin Input Thresholds
SLEEPn High Threshold
VSLEEP(H)
VSLEEPn rising
–
1.3
2.1
V
SLEEPn Low Threshold
VSLEEP(L)
VSLEEPn falling
0.5
1.2
–
V
tD(SLEEP)
VSLEEPn transitioning low
55
103
150
µs
–
500
–
nA
–
3.05
–
V
3.2
–
5.5
V
SLEEPn Delay
SLEEPn Input Bias Current
ISLEEPBIAS VSLEEPn = 5 V
VREG Pin Output
VREG Output Voltage
VVREG
VBIAS = 0 V
BIAS Pin Input
BIAS Input Voltage Range
VBIAS
BOOT Regulator
BOOT Voltage Enable Threshold
VBOOT(EN)
1.7
2.0
2.2
V
BOOT Voltage Enable Hysteresis
VBOOT(HYS)
–
200
–
mV
BOOT Voltage Low-Side Switch
Disable Threshold
VBOOTLS(DIS) VBOOT rising
–
4.1
–
V
–
200
275
mV
–
2.3
–
V
–
VVREG
–
−
VBOOT rising
Soft Start Pin
FAULT, HICCUP Reset Voltage
Hiccup OCP (and Low IQ PFM
Counter Enable) Threshold
Maximum Charge Voltage
VSSRST
VSS falling due to RSS(FLT)
HIC/PFMEN VSS rising
VSS(MAX)
Startup (Source) Current
ISSSU
HICCUP = FAULT = 0
–30
–20
–10
µA
Hiccup (Sink) Current
ISSHIC
HICCUP = 1
2.4
5
10
µA
–
2
–
kΩ
Pull-Down Resistance
RSS(FLT)
FAULT = 1 or VSLEEPn = low
Continued on the next page…
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8
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
ELECTRICAL CHARACTERISTICS (continued): valid at 4.0 V ≤ VIN ≤ 35 V, –40°C ≤ TA = TJ ≤ 150°C; unless otherwise specified
Characteristic
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
0 V < VFB < 200 mV
–
fOSC / 4
–
−
200 mV < VFB < 400 mV
–
fOSC / 2
–
−
400 mV < VFB
–
fOSC
–
−
tD(SS)
CSS = 22 nF
–
440
–
µs
tSS
CSS = 22 nF
–
880
–
µs
VSS > 2.3 V and OCL = 1
–
120
–
counts
Soft Start Pin (continued)
Soft Start Frequency Foldback
Soft Start Delay
Time3
Soft Start Output Ramp Time3
fSW(SS)
Hiccup Modes
Hiccup, OCP Count
OCPLIM
Hiccup, BOOT Undervoltage
(Shorted) Count
BOOTUV
–
120
–
counts
Hiccup, BOOT Overvoltage (Open)
Count
BOOTOV
–
7
–
counts
3.6
4.1
4.6
A
2.3
3.1
3.9
A
860
880
902
mV
–
–10
–
mV
715
740
765
mV
Overcurrent Protection (OCP)
PWM Pulse-by-Pulse Limit
ILIM(TONMIN) tON = tON(MIN)PWM
ILIM(TONMAX)
tON = (1 / fSW) – tOFF(MIN)PWM, no PWM
synchronization
Output Voltage Protection (OVP)
VOUT Overvoltage PWM Threshold
VOUT(OV)PWM VFB rising, PWM mode
VOUT Overvoltage Hysteresis
VOUT(OV)HYS VFB falling, relative to VOUT(OV)PWM
VOUT Undervoltage PWM Threshold
VOUT(UV)PWM VFB falling, PWM mode
VOUT Undervoltage Hysteresis
VOUT(UV)HYS VFB rising, relative to VOUT(UV)PWM
VOUT Undervoltage PFM Threshold
VOUT(UV)PFM VFB falling, Low-IQ PFM mode
–
10
–
mV
665
700
735
mV
5
7.5
10
ms
Power-On Reset (NPOR) Output
NPOR Rising Delay
NPOR Low Output Voltage
NPOR Leakage Current2
tD(NPOR)
VFB rising only
VNPOR(L)
INPOR = 5 mA
INPOR(LKG) VNPOR = 5.5 V
–
185
400
mV
–1
–
1
µA
155
170
185
ºC
–
20
–
ºC
Thermal Protection
Thermal Shutdown Rising Threshold3
Thermal Shutdown Hysteresis3
TSD
PWM stops immediately and COMP and SS are
pulled low
TSDHYS
1Thermally
limited depending on input voltage, output voltage, duty cycle, regulator load currents, PCB layout, and airflow.
2Negative current is defined as coming out of the node or pin, positive current is defined as going into the node or pin.
3Ensured by design and characterization, not production tested.
4Performance at the 0°C and 85°C ranges ensured by design and characterization, not production tested.
5Performance at 25°C ensured by design and characterization, not production tested.
6Performance at 85°C ensured by design and characterization, not production tested.
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9
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
CHARACTERISTIC PERFORMANCE
Reference Voltage versus Temperature
Switching Frequency versus Temperature
808
3.50
806
fOSC = 2.44 MHz
3.00
fOSC = 1.00 MHz
2.50
802
fOSC (MHz)
V VREF (mV)
804
800
798
2.00
1.50
796
1.00
794
792
0.50
-50
-25
0
25
50
75
100
125
150
-50
-25
0
Temperature (°C)
3.9
950
3.8
900
3.7
START, VINUV(ON)
STOP, VINUV(OFF)
3.6
3.5
3.4
3.3
125
150
850
VOUT(OV)PWM
VOUT(UV)PWM
800
VOUT(UV)PFM
750
700
650
-50
-25
0
25
50
75
Temperature (°C)
100
125
150
-50
Pulse-by-Pulse Current Limit at t ON(MIN)PWM
(ILIM(TONMIN)) versus Temperature
4.50
-25
0
25
50
75
Temperature (°C)
100
125
150
Error Amplifier Transconductance
versus Temperature
900
4.40
800
4.30
700
Transconductane (µA/V)
ILIM(TONMIN) (A)
100
VOUT Overvoltage and Undervoltage Thresholds
versus Temperature
VOUT OV and UV Thresholds (V)
VIN UVLO Thresholds (V)
VIN UVLO Start and Stop Thresholds
versus Temperature
25
50
75
Temperature (°C)
4.20
4.10
4.00
3.90
3.80
V
VFB>400
mV
FB>400 mV
V
VFB<400mV
FB<400 mV
600
500
400
300
200
3.70
-50
-25
0
25
50
75
Temperature (°C)
100
125
150
100
-50
-25
0
25
50
75
Temperature (°C)
100
125
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150
10
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
PWM/PFMn High and Low Voltage Thresholds
versus Temperature, VBIAS = 5.0 V
PWM/PFMn High and Low Voltage Thresholds
versus Temperature, VBIAS = 3.3 V
2.3
1.60
VPWMPFMn(H)
1.50
1.45
1.40
1.35
1.30
1.25
2.0
1.9
1.8
1.7
1.15
1.6
-50
-25
0
25
50
75
Temperature (°C)
100
125
-50
150
-25
0
25
50
75
100
125
150
Temperature (°C)
SLEEPn High and Low Voltage Thresholds
versus Temperature
SS Start and Hiccup Currents versus Temperature
1.60
25.0
VSLEEP(H)
VSLEEP(L)
1.40
Startup, ISSSU
Hiccup, ISSHIC
20.0
1.20
Current (µA)
SLEEPn Thresholds (V)
VPWMPFMn(L)
2.1
1.20
1.00
0.80
15.0
10.0
5.0
0.60
-50
-25
0
25
50
75
100
125
0
150
-50
Temperature (°C)
NPOR Low Output Voltage at 5 mA
versus Temperature
400
-25
0
25
50
75
Temperature (°C)
100
125
150
NPOR Time Delay versus Temperature
8.00
7.90
350
7.80
300
7.70
250
tD(NPOR) (ms)
V NPOR (mV)
VPWMPFMn(H)
2.2
VPWMPFMn(L)
PWM/PFMn Thresholds (V)
PWM/PFMn Thresholds (V)
1.55
200
150
7.60
7.50
7.40
7.30
100
7.20
50
7.10
7.00
0
-50
-25
0
25
50
75
Temperature (°C)
100
125
150
-50
-25
0
25
50
75
Temperature (°C)
100
125
150
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11
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
FUNCTIONAL DESCRIPTION
Overview
The A8580 is an asynchronous, current mode, buck regulator that
incorporates all the control and protection circuitry necessary to
provide the power supply requirements of car audio and infotainment systems.
The A8580 has three modes of operation. First, the A8580 can
deliver up to 2.5 A in pulse width modulation (PWM) mode.
Second, in Low-IQ pulse frequency modulation (PFM) mode,
the A8580 will draw only tens of microamperes from VIN while
maintaining VOUT (at no load). Under most conditions, Low-IQ
PFM mode is typically capable of supporting up to 550 mA.
Third, with the SLEEPn pin low, the A8580 will enter an ultralow current shutdown (sleep) mode where VOUT = 0 V and the
total current drawn from VIN will typically be less than 10 µA.
The PWM/PFMn input pin is used to select either PWM or
Low-IQ PFM mode. In PFM mode the A8580 is able to supply a
relatively high amount of current (typically 550 mA). This allows
enough current for a microcontroller or DSP to fully power-up.
After power-up, to obtain the full current capability of the A8580,
the microcontroller or DSP must change the PWM/PFMn input
from a logic low to a logic high to force PWM mode. This will
provide full current to the remainder of the system.
The A8580 was designed to support up to 2.5 A. However, the
exact amount of current it will supply, before possible thermal
shutdown, depends heavily on: duty cycle, ambient temperature,
airflow, PCB layout, and PCB construction. Figure 1 shows
calculated current ratings versus ambient temperature for VIN =
12 V, and VOUT = 3.3 V and 5.0 V, at both fSW = 425 kHz and
fSW = 2 MHz. This analysis assumed a 4-layer PCB constructed
according to the JEDEC standard (vielding a thermal resistance
of 34°C/W), with no nearby heat sources, and no airflow.
Reference Voltage
The A8580 incorporates an internal reference that allows output
voltages ( VOUT ) as low as 0.8 V. The accuracy of the internal
reference is ±1.0% from 0°C to 85°C and ±1.5% from −40°C to
150°C. The output voltage is programmed by connecting a resistor divider from VOUT to the FB pin of the A8580, as shown in
the Typical Applications schematics.
PWM Switching Frequency
The PWM switching frequency of the A8580 is adjustable from
250 kHz to 2.4 MHz and has an accuracy of about ±10% across
the operating temperature range.
During startup, the PWM switching frequency changes from 25%
to 50% and finally to 100% of fOSC , as VOUT rises from 0 V to
the regulation voltage. The startup switching frequency is discussed in more detail in the section describing soft start, below.
If the regulator output is shorted to ground, VFB ≈ 0 V, the PWM
frequency will be 25% of fOSC . In this case, the extra low switching frequency allows extra off-time between SW pulses. The
extra off-time allows the output inductor current to decay back to
0 A before the next SW pulse occurs. This prevents the inductor
current from climbing to a value that could damage the A8580 or
the output inductor.
SLEEPn Input
VIN = 12 V, VOUT = 5 V, fSW = 2 MHz
VIN = 12 V, VOUT = 5 V, fSW = 425 kHz
VIN = 12 V, VOUT = 3.3 V, fSW = 2 MHz
VIN = 12 V, VOUT = 3.3 V, fSW = 425 kHz
The A8580 has a SLEEPn logic level input pin. To get the A8580
to operate, the SLEEPn pin must be a logic high (>2.1 V). The
SLEEPn pin is rated to 40 V, allowing the SLEEPn pin to be connected directly to VIN if there is no suitable logic signal available
to wake up the A8580.
When SLEEPn transitions low, the A8580 waits approximately
103 µs before shutting down. This delay provides plenty of
filtering to prevent the A8580 from prematurely entering sleep
mode because of any small glitch coupling onto the PCB trace or
SLEEPn pin.
Figure 1: A8580 Typical Current Derating
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12
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
PWM/PFMn Input and PWM Synchronization
The PWM/PFMn pin provides two major functions. It is a control
input that sets the operating mode, and also an optional clock
input for setting PWM frequency.
If PWM/PFMn is a logic high, the A8580 operates in PWM
mode. If PWM/PFMn is a logic low, the A8580 operates in
Low‑IQ PFM (keep alive) mode. When PWM/PFMn transitions
from logic high to logic low, the A8580 checks for VSS >2.3 V
and NPOR at logic high. If these two conditions are satisfied,
then the A8580 will wait 2048 internal clock cycles and then
enter Low‑IQ PFM mode. This delay provides plenty of filtering to prevent the regulator from prematurely entering PFM
mode because of any small glitch coupling onto the PCB trace or
PWM/PFMn pin.
Also, note that the SLEEPn pin must be a logic high or the
PWM/PFMn input has no effect. The interaction between the
SLEEPn pin and PWM/PFMn pin is summarized in Table 1.
If an external clock is applied to the PWM/PFMn pin, the A8580
synchronizes its PWM frequency to the external clock. The
external clock may be used to increase the A8580 base PWM
frequency (fOSC) set by RFSET. Synchronization operates from
1.2 × fOSC(typ) to 1.5 × fOSC(typ) . The external clock pulses must
satisfy the pulse width, duty cycle, and rise/fall time requirements
shown in the Electrical Characteristics table in this datasheet.
BIAS Input Functionality, Ratings, and Connections
When the A8580 is powering up, it operates from an internal
LDO regulator, directly from VIN. However, VIN can be a relaTable 1: A8580 Modes of Operation
Pin Inputs
Operating Mode
SLEEPn
PWM/PFMn
Name
Low
Don’t care
Sleep
High
High
The BIAS pin is designed to operate in the range from
3.2 to 5.5 V. If the output of the regulator is in this range then
VOUT should be routed directly to the BIAS pin. However, if
the output of the regulator is above 5.6 V then a very small
LDO, capable of at least 5 mA, must be used to reduce the voltage to either 3.3 V or 5.0 V before routing it to the BIAS pin.
Operating with an external LDO will reduce the efficiency in
Low‑IQ PFM mode.
The BIAS pin may be driven by an external power supply. For
startup, there are no sequencing requirements between VIN and
BIAS. However, for shutdown, VIN should be removed before
BIAS. If BIAS is removed before VIN it will cause the A8580 to
reset. The reset will cause the A8580 to terminate PWM switching and VOUT will decay. Also, NPOR, VSS , and VCOMP will be
pulled low. Ideally, the SLEEPn pin should be used to set the
mode of the A8580 before VIN and/or BIAS are turned on or off.
If the BIAS pin is grounded, the A8580 will simply operate
continuously from VIN. However, during PFM mode, the input
current will increase and the PFM efficiency will be significantly
reduced.
Transconductance Error Amplifier
The transconductance error amplifier primary function is to control the regulator output voltage. The error amplifier is shown in
Figure 2. Here, it is shown as a three-terminal input device with
two positive and one negative input. The negative input is simply
VOUT = 0 V
fSW = fOSC
PWM
High
Description
tively high voltage and an LDO is very inefficient and generates
extra heat. To improve efficiency, especially in Low‑IQ PFM
mode, a BIAS pin is utilized. For most applications, the BIAS pin
should be connected directly to the output of the regulator, VOUT .
When VOUT rises to an adequate level (approximately 3.1 V), the
A8580 will shut down the inefficient LDO and begin running its
control circuitry directly from the output of the regulator. This
makes the A8580 much more efficient and cooler.
fSW =
PWM/PFMn
clock in
VOUT = OK
and
IOUT ≤ 2.5 A
400 mV
+
Enter Low‑IQ PFM after 2048 cycles,
if VSS > 2.3 V (typ) and NPOR = high
High
High
Low
Low‑IQ PFM
High
Low
Low‑IP PWM
fSW is VOUT
dependent
VOUT = OK
and
IOUT ≤
550 mA (typ)
Fault, ILIM at 50%
Error Amplifier
SS Pin
+
VREF
800 mV
-
COMP
Pin
FB Pin
Figure 2: The A8580 Error Amplifier
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13
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
connected to the FB pin and is used to sense the feedback voltage
for regulation. The two positive inputs are used for soft start and
steady-state regulation. The error amplifier performs an analog
OR selection between its two positive inputs. The error amplifier
regulates to either the soft start pin voltage (minus 400 mV) or
the A8580 internal reference, VREF, whichever is lower.
To stabilize the regulator, a series RC compensation network
(RZ and CZ) must be connected from the error amplifier output
(the COMP pin) to GND, as shown in the Typical Applications schematics. In most instances an additional, relatively low
value, capacitor (CP) should be connected in parallel with the
RZ-CZ components to reduce the loop gain at very high frequencies. However, if the CP capacitor is too large, the phase margin of the regulator may be reduced. Calculating RZ, CZ, and
CP is covered in detail in the Component Selection section of
this datasheet.
If a fault occurs or the regulator is disabled (SLEEPn = low), the
COMP pin is pulled to GND via approximately 1 kΩ and PWM
switching is inhibited.
Slope Compensation
The A8580 incorporates internal slope compensation (SE ) to
allow PWM duty cycles above 50% for a wide range of input/output voltages and inductor values. The slope compensation signal
is added to the sum of the current sense amplifier output and the
PWM ramp offset. As shown in the Electrical Characteristics
table, the amount of slope compensation scales with the base
switching frequency set by RFSET (fOSC ). The amount of slope
compensation does not change when the regulator is synchronized to an external clock.
The value of the output inductor should be chosen such that SE is
from 0.5× to 1× the falling slope of the inductor current (SF).
BOOT Regulator
The A8580 contains a regulator to charge the boot capacitor.
The voltage across the BOOT capacitor is typically 5.0 V. If
the BOOT capacitor is missing, the A8580 detects a boot overvoltage. Similarly, if the BOOT capacitor is shorted the A8580
detects a boot undervoltage. Also, the BOOT regulator has a
current limit to protect itself during a short circuit condition. The
details of how each type of boot fault is handled by the A8580 are
shown in Figures 13 and 14 and summarized in table 2.
Pulse Width Modulation (PWM) Mode
The A8580 utilizes fixed-frequency, peak current mode control to
provide excellent load and line regulation, fast transient response,
and ease of compensation. A high-speed comparator and control
logic, capable of typical pulse widths of 95 ns, are included in the
A8580. The inverting input of the PWM comparator is connected
to the output of the error amplifier. The non-inverting input is
connected to the sum of the current sense signal, the slope compensation, and a DC offset voltage (VPWMOFFS , 400 mV (typ) ).
At the beginning of each PWM cycle, the CLK signal sets the
PWM flip flop and the high-side MOSFET is turned on. When
the summation of the DC offset, slope compensation, and current
sense signal rises above the error amplifier voltage, the PWM flip
flop is reset and the high-side MOSFET is turned off.
The PWM flip flop is reset-dominant, so the error amplifier
may override the CLK signal in certain situations. For example,
at very light loads or extremely high input voltages the error
amplifier reduces (temporarily) output voltage below the 400 mV
DC offset and the PWM flip flop will ignore one or more of the
incoming CLK pulses. The high-side MOSFET will not turn
on, and the regulator will skip pulses to maintain output voltage
regulation.
The A8580 incorporates a high-bandwidth current sense amplifier to monitor the current in the high-side MOSFET. This current
signal is used by both the PWM and PFM control circuitry to
regulate the peak current. The current signal is also used by the
protection circuitry to prevent damage to the A8580.
Power MOSFETs
The A8580 includes a 40 V, 110 mΩ high-side N-channel
MOSFET, capable of delivering at least 2.5 A. The A8580 also
includes a 10 Ω, low-side MOSFET to help ensure the BOOT
capacitor is always charged. The typical RDS(on) increase versus
temperature is shown in Figure 3.
Normalized RDS(on)
Current Sense Amplifier
Temperature (°C)
Figure 3: Typical MOSFET RDS(on) versus Temperature
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14
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
In PWM mode all of the A8580 fault detection circuits are active.
See Figure 13 for a timing diagram showing how faults are handled when in PWM mode. Also, the Protection Features section
of this datasheet provides a detailed description of each fault and
Table 2 presents a summary.
Low-IQ Pulse Frequency Modulation (PFM)
Mode
The A8580 enters Low‑IQ PFM mode after 2048 internal clock
cycles, if SLEEPn is high, VSS > HIC/PFMEN (2.3 V (typ)), and
NPOR is high. In Low-IQ PFM mode, the regulator operates with
a switching frequency, fSW , that depends on the load condition.
In Low-IQ PFM mode, a comparator monitors the voltage at
the FB pin. If VFB is above about 800 mV, the A8580 remains
in coast mode and draws extremely low current from the input
supply.
If the voltage at the FB pin drops below about 800 mV, the A8580
will fully power-up, delay approximately 2.5 µs while it wakes
up, and then turn on the high-side MOSFET . VOUT will rise at
a rate dependent on the input voltage, inductor value, output
capacitance, and load. The high-side MOSFET will be turned off
when either:
• current in the high-side MOSFET reaches IPEAK(LO_IQ) , or
• the high-side MOSFET has been on for tON(PFM)MAX .
VOUT
3.3 V
After the high-side MOSFET is turned off, the A8580 will again
delay approximately tOFF(PFM) and either:
• turn on the MOSFET again, if VFB < 800 mV, or
• return to the Low-IQ PFM mode
Figures 4 and 5 demonstrate Low-IQ PFM mode operation for a
light load (66 mA) and a heavy load (330 mA), respectively.
In Low-IQ PFM mode the average current drawn from the input
supply depends primarily on both the load, and how often the
A8580 must fully power-up to maintain regulation. In Low-IQ
PFM mode the following faults are detected: a missing asynchronous diode, an open or shorted boot capacitor, VOUT shorted
to ground, and SW shorted to ground. As described in the next
section, if any of these faults occur the A8580 will transition
from Low-IQ PFM mode to Low-IP PWM mode, with operation
at 50% of the current limit of the PWM switching mode. See
Figure 14 for a timing diagram showing operation of the A8580
in Low‑IQ PFM mode.
In Low‑IQ PFM mode the A8580 dissipates very little power, so
the thermal monitoring circuit (TSD) is not needed and is disabled to minimize the quiescent current and improve efficiency.
Figure 6 shows PWM to Low‑IQ PFM transitions for a typical
microcontroller or DSP system. The system starts in PWM mode
at IOUT = 250 mA and then transitions to Low‑IQ PFM mode,
also at IOUT = 250 mA (time A). While in Low‑IQ PFM mode the
VOUT
3.3 V
18.5 µs
tOFF(PFM) = 435 ns
VSW
VSW
IPEAK(LO_IQ)
IPEAK(LO_IQ)
IL
Figure 4: Low-IQ PFM Mode Operation at VIN = 12 V,
VOUT = 3.3 V, and IOUT = 66 mA. SW turns on only once
every 18.5 µs to regulate VOUT
IL
Figure 5: Low-IQ PFM Mode Operation at VIN = 12 V,
VOUT = 3.3 V, and IOUT = 330 mA. SW turns on only
twice every 5 µs to regulate VOUT
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15
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
VOUT
B
D
Overall
Waveforms
C
A
E
IOUT
VPWM/PFMn
VOUT
VOUT
B
C
A
IOUT
IOUT
VPWM/PFMn
VPWM/PFMn
Time A: Transition from PWM to PFM at 250 mA
Time B: Load steps from 250 mA to 0 A in Low-IQ PFM
mode
VOUT
Time C: Load steps from 0 A to 100 mA and back to
0 A in Low-IQ PFM mode
VOUT
D
IOUT
E
IOUT
VPWM/PFMn
Time D: Load steps from 0 A to 250 mA in Low-IQ PFM
mode
VPWM/PFMn
Time E: Transition from Low-IQ PFM to PWM mode at 250
mA
Figure 6: Transitions between PWM Mode and Low-IQ PFM Mode, and Load Transient Responses;
using circuit in typical application schematic B (VIN = 12 V, VOUT = 5 V, fSW = 425 kHz)
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16
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
current drops from 250 mA to 0 A (time B) and also cycles from
no load to 100 mA (time C). In Low-IQ PFM mode the load steps
from IOUT = 0 A to 250 mA (time D) and then the A8580 transitions back to PWM mode (time E). For this example, the output
ripple voltage is always less than 30 mVPP and the transient
deflection between modes is always less than 50 mVPEAK.
Reduced Current (Low-IP) PWM Mode
The A8580 supports two different levels of current limiting in
PWM modes:
• 100% current, which is during normal PWM, and
• Low-IP, in which the current is limited to about 50% of the
typical current limit
The Low-IP PWM mode is invoked when the A8580 is supposed
to be in PFM mode but a fault occurs. The purpose of the Low-IP
PWM mode is to give priority to maintaining reliable regulation of VOUT while enabling all the protection circuits inside the
A8580 that are normally debiased during Low-IQ PFM mode
(high precision comparators, timers, and counters).
There are several faults that cause a transition from Low-IQ
PFM to Low-IP PWM mode: a missing asynchronous diode, an
open or shorted boot capacitor, VOUT shorted to ground, or SW
shorted to ground. See Figure 14 for a timing diagram showing
operation when the A8580 transitions from Low-IQ PFM mode
to Low-IP PWM mode.
Soft Start (Startup) and Inrush Current Control
Inrush current is controlled by a soft start function. When the
A8580 is enabled and all faults are cleared, the soft start pin will
source ISSSU and the voltage on the soft start capacitor, CSS , will
ramp upward from 0 V. When the voltage at the soft start pin
exceeds approximately 400 mV, the error amplifier will slew its
output voltage above the PWM Ramp Offset ( VPWMOFFS ). At that
instant, the high-side and low-side MOSFETs will begin switching. As shown in Figure 7, there is a small delay (tD(SS) ) between
when the enable pin transitions high, and when both the soft start
voltage exceeds 400 mV and the error amplifier slews its output
high enough to initiate PWM switching.
After the A8580 begins switching, the error amplifier will regulate the voltage at the FB pin to the soft start pin voltage minus
approximately 400 mV. During the active portion of soft start,
the voltage at the soft start pin rises from 400 mV to 1.2 V (a
difference of 800 mV), the voltage at the FB pin rises from 0 V
to 800 mV, and the regulator output voltage rises from 0 V to the
targeted setpoint, which is determined by the feedback resistor
divider on the FB pin.
During startup, the PWM switching frequency is reduced to 25%
of fOSC while VFB is below 200 mV. If VFB is above 200 mV but
below 400 mV, the switching frequency is reduced to 50% of
fOSC . Also, if VFB is below 400 mV, the gm of the error amplifier
is reduced to gm / 2. When VFB is above 400 mV the switching
frequency will be fOSC and the error amplifier gain will be gm .
The reduced switching frequencies and error amplifier gain are
necessary to help improve output regulation and stability when
VOUT is at a very low voltage. When VOUT is very low, the
PWM control loop requires on-times near the minimum controllable on-time, as well as extra-low duty cycles that are not possible at the base operating switching frequencies.
When the voltage at the soft start pin reaches approximately
1.2 V, the error amplifier will change mode and begin regulating
the voltage at the FB pin to the A8580 internal reference, 800 mV.
The voltage at the soft start pin will continue to rise to approximately VREG . Complete soft start operation from VOUT = 0 V is
shown in Figure 7.
VSLEEPn
tD(SS)
VOUT
3.3 V
tSS
VSS = 1.2 V
VSS = 400 mV
VCOMP
fSW
VSS
IL
fSW/4
fSW/2
Figure 7: Normal Startup to VOUT = 3.3 V and IOUT =
1.6 A; PWM/PFMn pin = high, SLEEPn pin transitions
from low to high
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17
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
If the A8580 is disabled or a fault occurs, the internal fault latch
will be set and the capacitor on the soft start pin will be discharged to ground very quickly by an internal 2 kΩ pull-down
resistor. The A8580 will clear the internal fault latch when the
voltage at the soft start pin decays to approximately 200 mV
(VSSRST). Conversely, if the A8580 enters hiccup mode, the
capacitor on the soft start pin is slowly discharged by a current
sink, ISSHIC . Therefore, the soft start capacitor (CSS) not only
controls the startup time but also the time between soft start
attempts in hiccup mode. Hiccup mode operation is discussed in
more detail in the Protection Features section of this datasheet.
Pre-Biased Startup
If the output of the regulator (VOUT) is pre-biased to some voltage, the A8580 will modify the normal startup routine to prevent
discharging the output capacitors. As described previously, the
error amplifier usually becomes active when the voltage at the
soft start pin exceeds 400 mV. If the output is pre-biased, the
FB pin will be at some non-zero voltage. The A8580 will not
start switching until the voltage at the soft start pin increases
to approximately VFB + 400 mV. When the soft start pin voltage exceeds this value: the error amplifier becomes active, the
voltage at the COMP pin rises, PWM switching starts, and VOUT
ramps upward from the pre-bias level. Figure 8 shows startup
when the output voltage is pre-biased to 1.6 V.
VSLEEPn
VOUT
VOUT rises from 1.6 V, it
is not pulled to 0 V
Not Power-On Reset (NPOR) Output
The A8580 has an inverted power-on reset output (NPOR) with a
fixed delay of its rising edge ( tD(NPOR) ). The NPOR output is an
open drain output so an external pull-up resistor must be used, as
shown in the Typical Applications schematics. NPOR transitions
high when the output voltage ( VOUT ), sensed at the FB pin, is
within regulation. In PWM mode, NPOR is high when the output
voltage is typically within 92.5% to 110% of the target value. In
PFM mode, NPOR is high when the output voltage is typically
above 87.5% of the target value. The NPOR overvoltage and
undervoltage comparators incorporate a small amount of hysteresis (10 mV typically) and filtering (5 µs typically) to help reduce
chattering due to voltage ripple at the FB pin.
The NPOR output is immediately pulled low either: if an output
overvoltage or an undervoltage condition occurs, or if the A8580
junction temperature exceeds the thermal shutdown threshold
(TSD). For other faults, NPOR behavior depends on the output
voltage. Table 2 summarizes all the A8580 fault modes and their
effect on NPOR.
At power-up, NPOR must be initialized (set to a logic low)
when VIN is relatively low. Figure 9 shows VIN ramping up, and
also NPOR being set to a logic low when VIN is only 2.2 V. For
this test, NPOR was pulled up to an external 3.3 V supply via a
2 kΩ resistor.
3.3 V
VSS = 1.2 V
1.6 V
Switching delayed until VSS
= VFB +400 mV
VSS = 400 mV
VIN = 2.2 V
VIN
VNPOR
VCOMP
VSS
IL
fSW
fSW/2
Figure 8: Pre-biased Startup from VOUT = 1.6 V to VOUT
= 3.3 V, at IOUT = 1.6 A
Figure 9: Initialization of NPOR as VIN Ramps Up
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18
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
At power-down, NPOR must be held in the logic low state as
long as possible. Figure 10 shows VIN ramping down and also
NPOR being held low until VIN is only 1.3 V. For this test, NPOR
was pulled up to an external 3.3 V supply via a 2 kΩ resistor.
Protection Features
The A8580 was designed to satisfy the most demanding automotive and non-automotive applications. In this section, a description of each protection feature is described and Table 2 summarizes the protection features and operation.
UNDERVOLTAGE LOCKOUT (UVLO)
An undervoltage lockout (UVLO) comparator monitors the voltage at the VIN pin and keeps the regulator disabled if the voltage
is below the stop threshold ( VINUV(OFF) ). The UVLO comparator
incorporates some hysteresis ( VINUV(HYS) ) to help reduce on-off
cycling of the regulator due to resistive or inductive drops in the
VIN path during heavy loading or during startup.
PULSE-BY-PULSE OVERCURRENT PROTECTION (OCP)
The A8580 monitors the current in the high-side MOSFET and
if the current exceeds the pulse-by-pulse overcurrent threshold
( ILIM ) then the high-side MOSFET is turned off. Normal PWM
operation resumes on the next clock pulse from the internal
oscillator. The A8580 includes leading edge blanking to prevent
falsely triggering the pulse-by-pulse current limit when the highside MOSFET is turned on.
Because of the addition of the slope compensation ramp to the
inductor current, the A8580 delivers more current at lower duty
cycles and less current at higher duty cycles. Also, the slope
compensation is not a perfectly linear function of switching
frequency. For a given duty cycle, this results in a little more current being available at lower switching frequencies than higher
frequencies. Figure 11 shows the typical and worst case min/max
pulse-by-pulse current limits versus duty cycle at fSW = 250 kHz
and 2.45 MHz.
If the synchronization input (PWM/PFMn) is used to increase
the switching frequency, the on-time and the current ripple will
decrease. This will allow slightly more current than at the base
switching frequency ( fOSC ).
The exact current the buck regulators can support is heavily
dependent on: duty cycle ( VIN, VOUT , Vf ), ambient temperature,
thermal resistance of the PCB, airflow, component selection, and
nearby heat sources.
OVERCURRENT PROTECTION (OCP) AND HICCUP
MODE
An OCP counter and hiccup mode circuit protect the buck regulator when the output of the regulator is shorted to ground or when
the load is too high. When the voltage at the soft start pin is
below the hiccup OCP threshold ( HIC/PFMEN ) the hiccup mode
counter is disabled. Two conditions must be met for the OCP
counter to be enabled and begin counting:
• VSS > HIC/PFMEN (2.3 V (typ)) and
• VCOMP is clamped at its maximum voltage (OCL =1)
4.6
4.4
4.2
4.0
VIN
3.8
ILIM (A)
VIN = 1.3 V
VNPOR
3.6
3.4
3.2
3.0
Min. at
at 2.45
2.45 MHz
MHz
Min.
2.8
Typ
Typ.atat2.45
2.45MHz
MHz
2.6
MAX_2.45
MHz
Max. at 2.45
MHz
2.4
TYP_250
kHz
Typ. at 250
kHz
2.2
MIN_250
kHz
Min. at 250
kHz
MAX_250
kHz
Max. at 250
kHz
0
5 10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 90 95 100
Duty Cycle (%)
Figure 10: NPOR being Held Low as VIN Ramps Down
Figure 11: Pulse-by-Pulse Current Limiting versus
Duty Cycle; at fSW = 250 kHz (dashed curves) and fSW =
2.45 MHz (solid curves)
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19
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
As long as these two conditions are met, the OCP counter
remains enabled and will count pulses from the overcurrent comparator. If the COMP pin voltage decreases ( OCL = 0 ) the OCP
counter is cleared.
If the OCP counter reaches OCPLIM counts (120), a hiccup latch
is set and the COMP pin is quickly pulled down by a relatively
low resistance (1 kΩ). The hiccup latch also enables a small current sink connected to the soft start pin ( ISSHIC ). This causes the
voltage at the soft start pin to slowly ramp downward. When the
voltage at the soft start pin decays to a low enough level (VSSRST ,
200 mV (typ)) the hiccup latch is cleared and the small current
sink turned off. At that instant, the soft start pin will begin to
source current ( ISSSU ) and the voltage at the soft start pin will
ramp upward. This marks the beginning of a new, normal soft
start cycle as described earlier. (Note: OCP is the only fault that
results in hiccup mode that is ignored when VSS < 2.3 V.)
When the voltage at the soft start pin exceeds the soft start
offset (typically 400 mV) the error amplifier forces the voltage
at the COMP pin to quickly slew upward and PWM switching
will resume. If the short circuit at the regulator output remains,
another hiccup cycle will occur. Hiccups will repeat until the
short circuit is removed or the regulator is disabled. If the short
circuit is removed, the A8580 will soft start normally and the
output voltage will automatically recover to the target level, as
shown in Figure 12.
Short removed
VOUT
VCOMP
120 OCP counts
VSS
IL
The A8580 monitors the voltage across the BOOT capacitor to
detect if the capacitor is missing or short circuited. If the BOOT
capacitor is missing, the regulator will enter hiccup mode after
7 PWM cycles. If the BOOT capacitor is short circuited, the
regulator will enter hiccup mode after 120 PWM cycles, provided
there is no VOUT overvoltage detection. At no load or very light
loads, the boot charging circuit will increase the output voltage
(via the output inductor) and cause an overvoltage condition to be
detected if VIN > VOUT + 5.7 V.
For a boot fault, hiccup mode will operate virtually the same
as described previously for an output short circuit fault (OCP)
with the soft start pin ramping up and down as a timer to initiate
repeated soft start attempts. Boot faults are a non-latched condition, so the A8580 will automatically recover when the fault is
corrected.
ASYNCHRONOUS DIODE PROTECTION
If the asynchronous diode (D1 in the Typical Applications schematics) is missing or damaged (open) the SW pin will be subject
to unusually high negative voltages. These negative voltages may
cause the A8580 to malfunction and could lead to damage.
The A8580 includes protection circuitry to detect when the
asynchronous diode is missing. If the SW pin is below typically
−1.25 V for more than about 50 ns , the A8580 will enter hiccup mode after detecting one missing diode fault. Also, if the
asynchronous diode is short circuited, the A8580 will experience
extremely high currents in the high-side MOSFET. If this occurs
the A8580 will enter hiccup mode after detecting one short circuited diode fault.
OUTPUT OVERVOLTAGE PROTECTION (OVP)
The A8580 provides a basic level of overvoltage protection by
monitoring the voltage level at the FB pin. Two overvoltage conditions can be detected:
2.3 V
ILIM(TONMIN)
BOOT CAPACITOR PROTECTION
200 mV
Figure 12. Hiccup Mode Operation and Recovery to
VOUT = 3.3 V, IOUT = 1.6 A
• The FB pin is disconnected from its feedback resistor divider.
In this case, a tiny internal current source forces the voltage at
the FB pin to rise. When the voltage at the FB pin exceeds the
overvoltage threshold ( VOUT(OV)PWM , 880 mV (typ)) PWM
switching will stop and NPOR will be pulled low.
• A higher, external voltage supply is accidently shorted to the
A8580s output. VFB will probably rise above the overvoltage
threshold and be detected as an overvoltage condition. In
this case, the low-side MOSFET will continue to operate
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20
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
and can correct the OVP condition, provided that only a few
milliamperes of pull-down current are required. In either
case, if the condition causing the overvoltage is corrected the
regulator will automatically recover.
PIN-TO-GROUND AND PIN-TO-PIN SHORT PROTECTIONS
The A8580 is designed to satisfy the most demanding automotive
applications. For example, the A8580 has been carefully designed
from the very beginning to withstand a short circuit to ground at
each pin without suffering damage.
In addition, care was taken when defining the A8580 pin-out to
optimize protection against pin-to-pin adjacent short circuits. For
example, logic pins and high voltage pins are separated as much
as possible. Inevitably, some low voltage pins are located adjacent to high voltage pins, but in these instances the low voltage
pins are designed to withstand increased voltages, with clamps
and/or series input resistance, to prevent damage to the A8580.
THERMAL SHUTDOWN (TSD)
The A8580 monitors junction temperature and will stop PWM
switching and pull NPOR low if it becomes too hot. Also, to prepare for a restart, the soft start and COMP pins will be pulled low
until VSS < VSS(RST) . TSD is a non-latched fault, so the A8580
will automatically recover if the junction temperature decreases
by approximately 20°C.
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21
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
TABLE 2: Summary of A8580 Fault Modes and Operation
Fault
Mode
VSS
Output
overcurrent,
VFB< 200 mV
During fault counting,
before Hiccup mode
BOOT
Charging
NPOR
State
Latched?
Reset
Condition
VCOMP
High-Side
MOSFET
Low-Side
MOSFET
Hiccup, after
120 OCP
faults
Clamped for
ILIM, then
pulled low for
hiccup
fOSC / 4 due to
VFB < 200 mV,
responds to
VCOMP
Can be
activated if
VBOOT is too
low
Not affected
Depends on
VOUT
No
Automatic,
after remove
the short
Output
overcurrent,
VFB > 400 mV
Hiccup, after
120 OCP
faults
Clamped for
ILIM, then
pulled low for
hiccup
fOSC due to
VFB > 400 mV,
responds to
VCOMP
Can be
activated if
VBOOT is too
low
Not affected
Depends on
VOUT
No
Automatic,
after decrease
load current
Boot capacitor
open/missing
(BOOTOV)
Hiccup, after
7 BOOTOV
faults
Pulled low for
hiccup
Forced
off when
BOOTOV fault
occurs
Forced off
when BOOT
fault occurs
Off after
BOOT fault
occurs
Depends on
VOUT
No
Automatic,
after replace
capacitor
Boot capacitor
shorted
(BOOTUV)
Hiccup, after
120 BOOTUV
faults
Not affected,
pulled low for
hiccup
Forced
off when
BOOTUV fault
occurs
Forced off
only during
hiccup
Off only during
hiccup
Depends on
VOUT
No
Automatic,
after unshort
capacitor
Asynchronous
diode missing
Hiccup after
1 fault
Pulled low for
hiccup
Forced off
after 1 fault
Can be
activated if
VBOOT is too
low
Not affected
Depends on
VOUT
No
Automatic,
after install
diode
Asynchronous
diode (or SW)
hard short to
ground
Hiccup after
1 fault
Pulled low for
hiccup
Forced off
after 1 fault
Can be
activated if
VBOOT is too
low
Not affected
Depends on
VOUT
No
Automatic,
after remove
short
Asynchronous
diode (or SW)
soft short to
ground
Hiccup, after
120 OCP
faults
Clamped for
ILIM, then
pulled low for
hiccup
Active,
responds to
VCOMP
Can be
activated if
VBOOT is too
low
Not affected
Depends on
VOUT
No
Automatic,
after remove
short
FB pin open
(FB floats
high)
Begins to
ramp up for
soft start
Transitions
low via loop
response
Forced off by
low VCOMP
Active during
tOFF(MIN)PWM
Off when VFB
is too high
Pulled low
when VFB is
too high
No
Automatic,
after connect
FB pin
Not affected
Transitions
low via loop
response
Forced off by
low VCOMP
Active during
tOFF(MIN)PWM
Off when VFB
is too high
Pulled low
when VFB is
too high
No
Automatic,
after VFB
returns to
normal range
Output
undervoltage
Not affected
Transitions
high via loop
response
Active,
responds to
VCOMP
Can be
activated if
VBOOT is too
low
Not affected
Pulled low
when VFB is
too low
No
Automatic,
after VFB
returns to
normal range
Thermal
shutdown
Pulled low and
latched until
VSS < VSSRST
Pulled low and
latched until
VSS < VSSRST
Forced off by
low VCOMP
Disabled
Off
Pulled low
No
Automatic,
after part
cools down
Not affected
Transitions
low via loop
response
Forced off by
low VCOMP
Active during
tOFF(MIN)PWM
No
Automatic,
VREG or
BIAS to
normal range
Output
overvoltage
(VFB >
880 mV)
VREG
or BIAS
overvoltage
(REGOV)
Off
Pulled low
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22
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
Figure 13: Operation with SLEEPn = high and PWM/PFMn = high (PWM mode)
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23
COMP
SS
VOUT
SW
VIN
DIODE
FAULT
BOOT
FAULT
OC
FAULT
HICCUP
OC
HIC_EN
NPOR
TSD
PWM/
PFMn
SLEEPn
MODE OFF
HICCUP
Vout shorted to GND
OC
SS
~500 mV
FSW /4
x120
Note: NPOR=1 already, so VSS>HIC/PFMEN
starts the 2048 PFM delay counter
2048
LO_IQ
SS>2.3V •
FB>0.74V
FSW
7.5ms
PWM
FSW/4
then
FSW/2
SS
FSW/4
OC
x120
LOW-IP PWM
HICCUP
FSW/4
then
FSW/2
SS
SS>2.3V •
FB>0.74V
FSW
7.5ms
2048
~500 mV
LO_IQ
x7 OV
x120 UV
BOOT
FAULTS
HICCUP
TO
2.3V
FSW/4
FROM
2.3V
HICC
UP
TO
2.3 V
FROM
2 .3V
FSW/4
x7 OV
x120 UV
HICC S
UP S
LOW-IP PWM
x7 OV
x120 UV
S
S
FSW/4
then
FSW/2
SS
SS>2.3V •
FB>0.74V
FSW
7.5ms
2048
HICCUP
~500 mV
x1
DIODE or SW
FAULTS
LO_IQ
x1
FSW/4
SS
HI
C
SS
7.5ms
2048
LO_IQ
PWM
SS>2.3V •
FB>0.74V
FSW/4
then FSW
FSW/2
~500 mV
FSW
Note: Faster SS shown here, so NPOR↑ starts
the 2048 PFM delay counter, instead of VSS
TO FROM TO FROM
2.3V 2.3V 2.3V 2 .3V
FSW/4
x1
HI
C
LOW-IP PWM
SS
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
Figure 14: Operation with SLEEPn = high and PWM/PFMn = low (Low-IQ PFM mode and transition to Low-IP PWM
mode)
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24
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
APPLICATION INFORMATION
Design and Component Selection
SETTING THE OUTPUT VOLTAGE (VOUT)
The output voltage of the regulator is determined by connecting
a resistor divider from the output node (VOUT) to the FB pin
as shown in Figure 15. There are trade-offs when choosing the
value of the feedback resistors. If the series combination (RFB1
+ RFB2) is too low, then the light load efficiency of the regulator will be reduced. So to maximize the efficiency, it is best to
choose higher values of resistors. On the other hand, if the parallel combination (RFB1 // RFB2) is too high, then the regulator
may be susceptible to noise coupling onto the FB pin.
The feedback resistors must satisfy the ratio shown in the following equation to produce the target output voltage, VOUT :
VOUT
–1
0.8 (V)
For Low‑IQ PFM mode, a feedforward capacitor (CFB) should
be connected in parallel with RFB1, as shown in Figure 16.
The purpose of this capacitor is to offset any stray capacitance
(CSTRAY) from the FB pin to ground. Without CFB, the stray
capacitance and the relatively high resistor values used for the
feedback network form a low pass filter and introduce lag to the
Low‑IQ PFM feedback path. The feedforward capacitor helps to
maintain sensitivity during Low‑IQ PFM mode and to assure the
output voltage ripple is minimized.
In general, CFB should be calculated as:
(2)
where CSTRAY is typically 15 to 25 pF.
FB PIN
RFB2
Figure 15: Connecting a Feedback Resistor Divider to
Set the Output Voltage
CFB
(1)
Compared to typical buck regulators, a PFM capable buck
regulator presents some unique challenges when determining its
feedback divider. This resistor divider must draw minimal current
from VOUT or it will reduce the efficiency during Low‑IQ PFM
operation. With this in mind, Allegro recommends the resistor
values show in Table 3 on page 34.
CFB > (1.5 × CSTRAY) × ( RFB2 / RFB1 )
V OUT
RFB1
V OUT
FB PIN
RFB2
CSTRAY
15 to 25 pF
Figure 16: Addition of CFB to Cancel Stray Capacitance
at the FB Pin in PFM Mode
2.50
2.25
2.00
1.75
Frequency (MHz)
RFB1
=
RFB2
RFB1
1.50
1.25
1.00
0.75
0.50
PWM BASE SWITCHING FREQUENCY (FOSC, RFSET)
0.25
The PWM base switching frequency, fOSC, is set by connecting a
resistor from the FSET pin to ground. Figure 17 is a graph showing the relationship between the typical switching frequency and
the FSET resistor. The base frequency is the output frequency,
0.00
5.0
15.0
25.0
35.0
45.0
55.0
65.0
75.0
85.0
95.0
RFSET (kΩ)
Figure 17: PWM Switching Frequency versus RFSET
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Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
fSW , when PWMPFMn is high (no external clocking signal). For
a given base switching frequency ( fOSC ), the FSET resistor can
be calculated as follows:
RFSET =
26385
– 2.75
fOSC
(3)
where fOSC is in kHz and RFSET is in kΩ.
When the PWM base switching frequency is chosen the
designer should be aware of the minimum controllable on-time,
tON(MIN)PWM of the A8580. If the system required on-time is less
than the A8580 minimum controllable on-time, switch node jitter
occurs and the output voltage will have increased ripple or oscillations.
The PWM base switching frequency required should be calculated as follows:
fOSC <
VOUT
tON(MIN)PWM × VIN(MAX)REQ
(4)
where
it is important to calculate an inductor value such that the falling
slope of the inductor current (SF) will work well with the A8580
slope compensation. The following equation can be used to calculate a range of values for the output inductor based on the well
known approach of providing slope compensation that matches
50% to 100% of the falling slope of the inductor current:
VOUT + Vf
2 × SE
≤ LO ≤
VOUT + Vf
SE
where Vf is the forward voltage of the asynchronous diode, and
LO is in µH.
In equation 6, the slope compensation (SE ) is a function of
switching frequency according the following:
SE = 0.23 × fOSC2 + 0.63 × fOSC + 0.038
More recently, Dr. Raymond Ridley presented a formula to calculate the amount of slope compensation required to critically damp
the double poles at half the PWM switching frequency:
tON(MIN)PWM is the minimum controllable on-time of the A8580
(95 ns (typ), 135 ns (max)), and
LO ≥
VOUT+Vf
SE
1 – 0.18
D
VIN(MAX)REQ is the maximum required operational input voltage
(not the peak surge voltage).
=
VOUT+Vf
SE
1 – 0.18 ×
fOSC < 0.66 ×
VOUT
tON(MIN)PWM × VIN(MAX)REQ
(5)
OUTPUT INDUCTOR (LO)
For a peak current mode regulator it is common knowledge that,
without adequate slope compensation, the system will become
unstable when the duty cycle is near or above 50%. However, the
slope compensation in the A8580 is a fixed value (SE). Therefore,
(7)
where SE is in A/µs and fOSC is in MHz.
VOUT is the output voltage,
If the A8580 PWM synchronization function is employed, then
the base switching frequency should be chosen such that jitter
will not result at the maximum synchronized switching frequency, determined from equation 4:
(6)
(VIN(min)+Vf )
VOUT+Vf
(8)
This formula allows the inclusion of the duty cycle (D), which
should be calculated at the minimum input voltage to insure
optimal stability. Also, to avoid dropout (that is, saturation of
the buck regulator), VIN(min) must be approximately 1 to 1.5 V
above VOUT when calculating the inductor value with equation 8.
If equations 7 or 8 yield an inductor value that is not a standard
value, then the next highest available value should be used. The
final inductor value should allow for 10% to 20% of initial tolerance and 20% to 30% of inductor saturation.
The saturation current of the inductor should be higher than the
peak current capability of the A8580. Ideally, for output short circuit conditions, the inductor should not saturate even at the high-
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Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
est pulse-by-pulse current limit at minimum duty cycle, 4.6 A.
This may be too costly. At the very least, the inductor should not
saturate at the peak operating current according to the following:
IPEAK = 4.1 –
SE × (VOUT +Vf )
(9)
1.15 × fOSC × (VIN (max)+Vf )
where VIN(max) is the maximum continuous input voltage, such
as 18 V (not a surge voltage, such as 40 V).
Starting with equation 9, and subtracting half of the inductor
ripple current, provides us with an interesting equation to predict
the typical DC load capability of the regulator at a given duty
cycle (D):
IOUT(DC) = 4.1 –
SE× D
fOSC
VOUT × (1– D)
2 × fOSC × LO
(10)
After an inductor is chosen, it should be tested during output
short circuit conditions. The inductor current should be monitored
using a current probe. A good design would ensure neither the
inductor nor the regulator are damaged when the output is shorted
to ground at maximum input voltage and the highest expected
ambient temperature.
OUTPUT CAPACITORS
The output capacitors filter the output voltage to provide an
acceptable level of ripple voltage and they store energy to help
maintain voltage regulation during a load transient. The voltage
rating of the output capacitors must support the output voltage
with sufficient design margin.
The output voltage ripple (ΔVOUT ) is a function of the output
capacitor parameters: COUT , ESRCOUT , and ESLCOUT :
∆VOUT = ∆IL × ESRCOUT
+
VIN –VOUT
LO
∆IL
+
8 fSW C OUT
× ESLCOUT
(11)
The type of output capacitors will determine which terms of
equation 11 are dominant. For ceramic output capacitors the ESRCOUT and ESLCOUT are virtually zero, so the output voltage ripple
will be dominated by the third term of equation 11:
∆VOUT =
∆IL
8 fSW C OUT
(12)
To reduce the voltage ripple of a design using ceramic output
capacitors, simply: increase the total capacitance, reduce the
inductor current ripple (that is, increase the inductor value), or
increase the switching frequency.
For electrolytic output capacitors the value of capacitance will
be relatively high, so the third term in equation 11 will be very
small. The output voltage ripple will be determined primarily by
the first two terms of equation 11:
∆VOUT = ∆IL × ESRCOUT
+
VIN –VOUT
LO
× ESLCOUT
(13)
To reduce the voltage ripple of a design using electrolytic output
capacitors, simply: decrease the equivalent ESRCO and ESLCO
by using a high(er) quality capacitor, or add more capacitors in
parallel, or reduce the inductor current ripple (that is, increase the
inductor value).
The ESR of some electrolytic capacitors can be quite high so
Allegro recommends choosing a quality capacitor for which the
ESR or the total impedance is clearly documented in the datasheet. Also, the ESR of electrolytic capacitors usually increases
significantly at cold ambients, as much as 10×, which increases
the output voltage ripple and, in most cases, reduces the stability
of the system.
The transient response of the regulator depends on the quantity
and type of output capacitors. In general, minimizing the ESR of
the output capacitance will result in a better transient response.
The ESR can be minimized by simply adding more capacitors in
parallel or by using higher quality capacitors. At the instant of a
fast load transient (di/dt), the output voltage will change by the
amount:
∆VOUT = ∆ILOAD × ESRCOUT +
di
ESLCOUT
dt
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(14)
27
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
After the load transient occurs, the output voltage will deviate
from its nominal value for a short time. This time will depend
on the system bandwidth, the output inductor value, and output
capacitance. Eventually, the error amplifier will bring the output
voltage back to its nominal value.
The speed at which the error amplifier brings the output voltage
back to its setpoint depends mainly on the closed-loop bandwidth
of the system. A higher bandwidth usually results in a shorter
time to return to the nominal voltage. However, with a higher
bandwidth system, it may be more difficult to obtain acceptable
gain and phase margins. Selection of the compensation components (RZ, CZ, and CP) are discussed in more detail in the
Compensation Components section of this datasheet.
LOW‑IQ PFM OUTPUT VOLTAGE RIPPLE CALCULATION
After choosing an output inductor and output capacitor(s), its
important to calculate the output voltage ripple (ΔVOUT (PFM))
that will occur during Low‑IQ PFM mode. With ceramic output
capacitors the output voltage ripple in PWM mode is usually
negligible, but that is not the case during Low‑IQ PFM mode.
First, calculate the high-side MOSFET on-time and off-time. The
on-time is defined as the time it takes for the inductor current to
reach the peak current threshold, IPEAK(LO_IQ) :
tON =
IPEAK(LO_IQ) × LO
VIN – VOUT – IPEAK(LO_IQ) × ( RDS(on)HS + DCRLO ) (15)
If the Low‑IQ PFM output voltage ripple appears to be too high,
then the output capacitance should be increased and/or the output
inductance should be decreased. Decreasing the inductor value
has the drawback of increasing the ripple current, so a higher load
current will be required to transition from discontinuous conduction mode (DCM) to continuous conduction mode (CCM). This
might not be acceptable.
In general, the Low‑IQ PFM output voltage ripple increases as
the input voltage decreases. Also, from equation 15, note that
tON increases as the VOUT / VIN ratio increases (that is, as VIN
decreases). If the VOUT / VIN ratio is too high, the system is not
able to achieve IPEAK(LO_IQ) in only one PFM pulse. In this case
the on-time is limited to approximately 4.1 µs and a second PFM
pulse is required, about tOFF(PFM) later, as shown in Figure 5.
INPUT CAPACITORS
Three factors should be considered when choosing the input
capacitors. First, they must be chosen to support the maximum
expected input surge voltage with adequate design margin.
Second, the capacitor rms current rating must be higher than the
expected rms input current to the regulator. Third, they must have
enough capacitance and a low enough ESR to limit the input
voltage dV/dt to something much less than the hysteresis of the
VIN pin UVLO circuitry ( VINUV(HYS) , nominally 400 mV for the
A8580), at maximum loading and minimum input voltage.
The input capacitors must deliver the rms current according to:
0.55
0.50
The off-time is defined as the time it takes for the inductor current to decay from IPEAK(LO_IQ) to 0 A:
(16)
Finally, the Low‑IQ PFM output voltage ripple can be calculated:
∆VOUT(LO_IQ) =
IPEAK(LO_IQ) × (tON + tOFF)
2 × COUT
(17)
0.45
0.40
Ir m s / IOUT
tOFF
IPEAK(LO_IQ) × LO
=
VOUT+Vf
(18)
Irms = IOUT D × (1– D)
Where RDS(on) is the on-resistance (110 mΩ (typ)) of the highside MOSFET and DCRLO is the DC resistance of the output
inductor, LO. For relatively low input voltages, the on‑time during Low-IQ PFM mode is internally limited to about 4.1 µs.
0.35
0.30
0.25
0.20
0.15
0.10
0.05
0
0
10
20
30
40
50
60
70
80
90
100
Duty Cycle (%)
Figure 18: Input Capacitor Ripple versus Duty Cycle
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Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
where the duty cycle is:
D ≈ (VOUT+ Vf ) / ( VIN + Vf )
(19)
and Vf is the forward voltage of the asynchronous diode, D1 .
Figure 18 shows the normalized input capacitor rms current
versus duty cycle. To use this graph, simply find the operational
duty cycle (D) on the x-axis and determine the input/output current multiplier on the y-axis. For example, at a 20% duty cycle,
the input/output current multiplier is 0.40. Therefore, if the
regulator is delivering 2.5 A of steady-state load current, the input
capacitor(s) must support 0.40 × 2.5 A, or 1.0 Arms.
The input capacitor(s) must limit the voltage deviations at the
VIN pin to something significantly less than the A8580 VIN pin
UVLO hysteresis during maximum load and minimum input
voltage. The minimum input capacitance can be calculated as
follows:
CIN ≥
IOUT × D × (1– D)
0.85 × fOSC × ∆VIN(MIN)
(20)
where ΔVIN(MIN) is chosen to be much less than the hysteresis of
the VIN pin UVLO comparator (ΔVIN(MIN) ≤ 150 mV is recommended).
The D × (1-D) term in equation 20 has an absolute maximum
value of 0.25 at 50% duty cycle. So, for example, a very conservative design, based on: IOUT = 2.5 A, fOSC = 85% of 425 kHz,
D × (1-D) = 0.25, and ΔVIN = 150 mV, yields:
CIN ≥
2.5 (A) × 0.25
= 11.5 µF
361 (kHz) × 150 (mV)
A good design should consider the DC bias effect on a ceramic
capacitor: as the applied voltage approaches the rated value, the
capacitance value decreases. This effect is very pronounced with
the Y5V and Z5U temperature characteristic devices (as much as
90% reduction) so these types should be avoided. The X5R and
X7R type capacitors should be the primary choices due to their
stability versus both DC bias and temperature.
For all ceramic capacitors, the DC bias effect is even more pronounced on smaller sizes of device case, so a good design uses
the largest affordable case size (such as 1206 or 1210). Also, it is
advisable to select input capacitors with plenty of design margin
in the voltage rating to accommodate the worst case transient
input voltage (such as a load dump as high as 40 V for automotive applications).
ASYNCHRONOUS DIODE (D1)
There are three requirements for the asynchronous diode. First,
the asynchronous diode must be able to withstand the regulator
input voltage when the high-side MOSFET is on. Therefore, one
should choose a diode with a reverse voltage rating ( VR ) higher
than the maximum expected input voltage (that is, the surge voltage).
Second, the forward voltage of the diode ( Vf ) should be minimized or the regulator efficiency suffers. Also if Vf is too high,
the A8580 missing diode protection function could be falsely
activated. A Schottky type diode that can maintain a very low
Vf when the regulator output is shorted to ground, at the coldest
ambient temperature, is highly recommended.
Third, the asynchronous diode must conduct the output current
when the high-side MOSFET is turned off. Therefore, the average
forward current rating of this diode (If (AVG)) must be high enough
to deliver the load current according to
If (AVG) ≥ IOUT(MAX) ( 1 – DMIN )
(21)
where DMIN is the minimum duty cycle defined in equation 19,
and IOUT(MAX) is the maximum continuous output current of the
regulator.
BOOTSTRAP CAPACITOR
A bootstrap capacitor must be connected between the BOOT and
SW pins to provide the floating gate drive to the high-side MOSFET. Usually, 47 nF is an adequate value. This capacitor should
be a high-quality ceramic capacitor, such as an X5R or X7R, with
a voltage rating of at least 16 V.
The A8580 incorporates a 10 Ω low-side MOSFET to ensure that
the bootstrap capacitor is always charged, even when the regulator is lightly loaded or pre-biased.
SOFT START AND HICCUP MODE TIMING (CSS)
The soft start time of the A8580 is determined by the value of the
capacitance at the soft start pin, CSS . When the A8580 is enabled,
the voltage at the soft start pin starts from 0 V and is charged by
the soft start current, ISSSU . However, PWM switching does not
begin instantly because the voltage at the soft start pin must rise
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Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
above 400 mV. The soft start delay (tD(SS) ) can be calculated as:
tD(SS) = CSS ×
400 (mV)
ISSSU
(22)
If the A8580 is starting with a very heavy load, a very fast soft
start time may cause the regulator to exceed the pulse-by-pulse
overcurrent threshold. This occurs because the total of the full
load current, the inductor ripple current, and the additional current required to charge the output capacitors:
ICO = COUT × VOUT / tSS
(23)
is higher than the pulse-by-pulse current threshold, as shown in
Figure 19. This phenomena is more pronounced when using high
value electrolytic type output capacitors. To avoid prematurely
triggering hiccup mode the soft start capacitor, CSS , should be
calculated according to:
CSS ≥
ISSSU × VOUT × COUT
0.8 (V) × ICO
}
tSS
tSS = VOUT ×
or
COUT
ICO
tSS = 0.8 (V) × CSS
ISSSU
(25)
(26)
When the A8580 is in hiccup mode, the soft start capacitor is
used as a timing capacitor and sets the hiccup period. The soft
start pin charges the soft start capacitor with ISSSU during a
startup attempt and discharges the same capacitor with ISSHIC
between startup attempts. Because the ratio ISSSU / ISSHIC is
approximately 4:1, the time between hiccups will be about four
times as long as the startup time. Therefore, the effective dutycycle of the A8580 will be very low and the junction temperature
will be kept low.
COMPENSATION COMPONENTS (RZ, CZ, AND CP)
(24)
where VOUT is the output voltage, COUT is the output capacitance, ICO is the amount of current allowed to charge the output
capacitance during soft start (recommended: 0.1 A < ICO < 0.3 A).
Higher values of ICO result in faster soft start times. However,
lower values of ICO ensure that hiccup mode is not falsely triggered. Allegro recommends starting the design with an ICO of
0.1 A and increasing it only if the soft start time is too slow. If a
non-standard capacitor value for CSS is calculated, the next larger
value should be used.
ILIM
ILOAD
The output voltage ramp time, tSS , can be calculated by using
either of the following methods:
To properly compensate the system, it is important to understand
where the buck power stage, load resistance, and output capacitance form their poles and zeros in frequency. Also, it is important to understand that the (Type II) compensated error amplifier
introduces a zero and two more poles, and where these should be
placed to maximize system stability, provide a high bandwidth,
and optimize the transient response.
First, consider the power stage of the A8580, the output capacitors, and the load resistance. This circuitry is commonly referred
as the control-to-output transfer function. The low frequency
gain of this circuitry depends on the COMP to SW current gain
(gmPOWER ), and the value of the load resistor (RL). The DC gain
(GCO(0Hz)) of the control-to-output is:
GCO(0Hz) = gmPOWER × RL (27)
Output
capacitor
current, ICO
Figure 19: Output Current (ICO) During Startup
The control-to-output transfer function has a pole (fP1), formed
by the output capacitance (COUT) and load resistance (RL),
located at:
fP1 =
1
2� × RL × COUT
(28)
The control-to-output transfer function also has a zero (fZ1)
formed by the output capacitance (COUT) and its associated ESR:
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Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
fZ1 =
following equations are very accurate.
1
2� × ESR × COUT
(29)
For a design with very low-ESR type output capacitors (such as
ceramic or OS-CON™ output capacitors), the ESR zero (fZ1 )
is usually at a very high frequency, so it can be ignored. On the
other hand, if the ESR zero falls below or near the 0 dB crossover
frequency of the system (as is the case with electrolytic output
capacitors), then it should be cancelled by the pole formed by the
CP capacitor and the RZ resistor (discussed and identified later
as fP3).
A Bode plot of the control-to-output transfer function for the
configuration shown in Typical Application Schematic B (VOUT
= 5.0 V, IOUT = 2.5 A, RL = 2 Ω) is shown in Figure 20. The pole
at fP1 can easily be seen at 1.9 kHz while the ESR zero (fZ1)
occurs at a very high frequency, 636 kHz (this is typical for a
design using ceramic output capacitors). Note: There is more
than 90° of total phase shift because of the double-pole at half the
switching frequency.
Next, consider the feedback resistor divider (RFB1 and
RFB2), and the error amplifier (gm) and compensation network
RZ-CZ-CP. It greatly simplifies the transfer function derivation if RO >> RZ , and CZ >> CP . In most cases, RO > 2 MΩ,
1 kΩ < RZ < 100 kΩ, 220 pF < CZ < 47 nF, and CP < 50 pF, so the
The low frequency gain of the control section (GC(0Hz)) is formed
by the feedback resistor divider and the error amplifier. It can be
calculated using:
GC(0Hz) =
=
RFB2
RFB1 +RFB2
VFB
=
Gain (dB)
G CO(0Hz) = 15.1 dB
fZ1 = 636 kHz
-20
-40
Phase (°)
-60
180
gm is the error amplifier transconductance (750 µA/V ), and
RO is the error amplifier output impedance (AVOL/gm ).
The transfer function of the Type-II compensated error amplifier
has a (very) low frequency pole (fP2 ) dominated by the output
error amplifier output impedance (RO) and the CZ compensation
capacitor:
90
-180
101
102
103
104
105
Frequency (Hz)
Figure 20: Control-to-Output Bode Plot
1
2� × RO × CZ
(31)
1
2� × RZ × CZ
(32)
Lastly, the transfer function of the Type-II compensated error
amplifier has a (very) high frequency pole (fP3) dominated by the
RZ resistor and the CP capacitor:
Double Pole at
212.5 kHz
-90
(30)
VFB is the reference voltage (0.8 V),
fZ2 =
0
× AVOL
The transfer function of the Type-II compensated error amplifier
also has frequency zero (fZ2) dominated by the RZ resistor and
the CZ capacitor:
fP1 = 1.9 kHz
0
VOUT
VOUT is the output voltage,
60
20
VFB
× gm × RO
where
fP2 =
40
VOUT
× gm × RO
106
fP3 =
1
2� × RZ × CP
(33)
A Bode plot of the error amplifier and its compensation network
is shown in Figure 21, fP2 , fP3 , and fZ2 are indicated on the mag-
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Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
nitude plot. Notice that the zero (fZ2 at 3.8 kHz) has been placed
so that it is just above the pole at fP1 previously shown in the
control-to-output Bode plot (Figure 20) at 1.9 kHz. Placing fZ2
just above fP1 will result in excellent phase margin, but relatively
slow transient recovery time, as will be shown later.
Finally, consider the combined Bode plot of both the controlto-output and the compensated error amplifier (figure 22).
60
Gain (dB)
40
20
fZ2 ≈ 3.8 kHz
This section presents a methodology to systematically apply the
design considerations provided above.
fP3 ≈ 90 kHz
0
-20
1. Choose the system bandwidth (fC ). This is the frequency at
which the magnitude of the gain crosses 0 dB. Recommended
values for fC , based on the PWM switching frequency, are in the
range fSW / 20 < fC < fSW / 7.5. A higher value of fC generally
provides a better transient response, while a lower value of fC
generally makes it easier to obtain higher gain and phase margins.
-40
Phase (°)
-60
180
90
0
-90
-180
101
Complete designs for several common output voltages, at fSW of
425 kHz, 1 MHz, and 2 MHz are provided in Table 3 on page 34.
A GENERALIZED TUNING PROCEDURE
fP2 ≈ 80 Hz
G C(0Hz) = 49 dB
Careful examination of this plot shows that the magnitude and
phase of the entire system (red curve) are simply the sum of the
error amplifier response (blue curve) and the control-to-output
response (green curve). The bandwidth of this system (fC) is
60 kHz, the phase margin is 69 degrees, and the gain margin is
14 dB.
102
103
104
105
2. Calculate the RZ resistor value. This sets the system bandwidth
(fC):
2� × COUT
RZ = fC × VOUT ×
(34)
VFB
gmPOWERx × gm
106
Frequency (Hz)
Figure 21: Type-II Compensated Error Amplifier
3. Determine the frequency of the pole (fP1). This pole is formed
by COUT and RL. Use equation 28 (repeated here):
60
Gain (dB)
40
fP1 =
fC = 60 kHz
20
GM = 14 dB
0
-20
-60
180
Phase (°)
2� × RL × COUT
4. Calculate a range of values for the CZ capacitor. Use the following:
-40
4
1
< CZ <
2� × RZ × fC
2� × RZ × 1.5 × fP1
90
PM = 69°
0
-90
-180
101
1
102
103
104
105
106
Frequency (Hz)
Figure 22. Bode Plot of the Complete System (Red
Curves)
(35)
To maximize system stability (that is, to have the greatest gain
margin), use a higher value of CZ . To optimize transient recovery
time, although at the expense of some phase margin, use a lower
value of CZ .
Figure 23 compares the output voltage recovery time due to a 1 A
load transient for the system shown in Figure 22 (fZ2 = 3.8 kHz,
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32
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
69° phase margin) and a system with fZ2 at 1/4 the crossover frequency (15 kHz). The system with fZ2 at 15 kHz has 60° of phase
margin, but recovers much faster (≈3×) than the other system.
5. Calculate the frequency of the ESR zero (fZ1) formed by the
output capacitor(s). Use equation 29 (repeated here):
1
2� × ESR × COUT
Alternatively, if fZ1 is near or below the target crossover frequency (fC), then use equation 33 to calculate the value of CP by
setting fP3 equal to fZ1. This is usually the case for a design using
high ESR electrolytic output capacitors.
5.00
fZ2 = 15 kHz
4.99
4.98
fZ2 = 3.8 kHz
Voltage (V)
fZ1 =
If fZ1 is at least one decade higher than the target crossover
frequency (fC) then fZ1 can be ignored. This is usually the case
for a design using ceramic output capacitors. Use equation 33 to
calculate the value of CP by setting fP3 to either 5 × fC or fSW / 2,
whichever is higher.
4.97
4.96
4.95
4.94
4.93
0
40
80
120
160
200
240
Time (µs)
Figure 23: Transient Recovery Comparison for fZ2 at
3.8 kHz / 69° and 15 kHz / 60°
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33
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
Table 3: Recommended Component Values
LO
(µH)
CO2
(µF)
RFB1
(kΩ)
RFB2
(kΩ)
CFB
(pF)
1.51
3.3
80
63.4 +3.83
76.8
N/A
3.3
8.2
40
147
47.0
10
10
50
221 + 0.499
42.2
8
6.5
15
60
287 +6.5
41.2
6
3.3
3.3
20
147
47.0
10
4.7
30
221 + 0.499
42.2
8
6.5
6.8
40
287 +6.5
41.2
6
3.3
1.5
10
147
47.0
10
2.2
15
221 + 0.499
42.2
8
3.3
20
287 +6.5
41.2
6
VOUT
(V)
5.0
5.0
5.0
6.5
1If
fSW
(MHz)
0.425
1
2
RFSET
(kΩ)
59.0
23.7
10.5
RZ + CZ // CP
RZ
(kΩ)
CZ
(pF)
CP
(pF)
BIAS Pin
Modes
24.3
560
15
3.3 V EXT
PWM and PFM
26.1
560
15
Connected to
VOUT
PWM and PFM
49.9
270
8
Connected to
VOUT
PWM and PFM
78.7
180
4.7
3.3 V or 5.0 V
LDO
PWM and PFM
18.2
560
15
Connected to
VOUT
PWM and PFM
41.2
270
8
Connected to
VOUT
PWM and PFM
71.5
180
4.7
3.3 V or 5.0 V
LDO
PWM and PFM
11.5
680
15
Connected to
VOUT
PWM and PFM
26.1
330
8
Connected to
VOUT
PWM and PFM
45.3
180
4.7
3.3 V or 5.0 V
LDO
PWM and PFM
BIAS is not connected to VOUT, then the minimum external load must be ≥75 µA at all temperatures. No load operation is okay at approximately 25ºC to 75ºC only.
tolerance and DC-bias effect must be considered when choosing components to obtain CO.
2Negative
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34
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
Power Dissipation and Thermal Calculations
The power dissipated in the A8580 is the sum of the power dissipated from the VIN supply current (PIN), the power dissipated
due to the switching of the high-side power MOSFET (PSW ), the
power dissipated due to the rms current being conducted by the
high-side power MOSFET (PCOND ), and the power dissipated by
the gate driver (PDRIVER).
The power dissipated from the VIN supply current can be calculated using the following equation:
PIN = VIN × IQ + (VIN – VGS) × QG × fSW
(36)
where
IQ is the input quiescent current drawn by the A8580 (nominally 2.5 mA),
VGS is the MOSFET gate drive voltage (typically 5 V),
QG is the MOSFET gate charge (approximately 2.5 nC), and
fSW is the PWM switching frequency.
The power dissipated by the high-side MOSFET during PWM
switching can be calculated using the following equation:
where
VIN × IOUT × (tr + tf ) × fSW
2
IOUT is the regulator output current,
ΔIL is the peak-to-peak inductor ripple current,
RDS(on)HS is the on-resistance of the high-side MOSFET, and
Vf is the forward voltage of the asynchronous diode.
The RDS(on)of the high-side MOSFET has some initial tolerance
plus an increase from self-heating and elevated ambient temperatures. A conservative design should accommodate an RDS(on) with
at least a 15% initial tolerance plus 0.39%/°C increase due to
temperature.
The sum of the power dissipated by the internal gate driver can
be calculated using the following equation:
VIN is the input voltage,
PSW =
where
(37)
PDRIVER = QG × VGS × fSW
(39)
where
VGS is the gate drive voltage (typically 5 V),
QG is the gate charge to drive MOSFET to VGS = 5 V (about
2.5 nC), and
fSW is the PWM switching frequency.
Finally, the total power dissipated (PTOTAL) is the sum of the
previous equations:
PTOTAL = PIN + PSW + PCOND + PDRIVER
(40)
The average junction temperature can be calculated with the following equation:
VIN is the input voltage,
IOUT is the regulator output current,
fSW is the PWM switching frequency, and
where PTOTAL is the total power dissipated as described in
equation 40, RθJA is the junction-to-ambient thermal resistance
(34°C/W on a 4-layer PCB), and TA is the ambient temperature.
tr and tf are the rise and fall times measured at the SW node.
The exact rise and fall times at the SW node depend on the
external components and PCB layout so each design should be
measured at full load. Approximate values for both tr and tf range
from 10 to 15 ns.
The power dissipated by the internal high-side MOSFET while it
is conducting can be calculated using the following equation:
2
PCOND = Irms(FET)
× RDS(on)HS
VOUT +Vf
∆IL2
2
I
=
× OUT + 12 × RDS(on)HS
VIN +Vf
(38)
TJ = PTOTAL + RθJA + TA
(41)
The maximum junction temperature will be dependent on how
efficiently heat can be transferred from the PCB to ambient air. It
is critical that the thermal pad on the bottom of the IC should be
connected to a at least one ground plane using multiple vias.
As with any regulator, there are limits to the amount of heat that
can be dissipated before risking thermal shutdown. There are
tradeoffs between: ambient operating temperature, input voltage,
output voltage, output current, switching frequency, PCB
thermal resistance, airflow, and other nearby heat sources. Even
a small amount of airflow will reduce the junction temperature
considerably.
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115 Northeast Cutoff
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35
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
PCB Component Placement and Routing
A good PCB layout is critical if the A8580 is to provide clean,
stable output voltages. Follow these guidelines to insure a good
PCB layout. Figure 24 shows a typical buck converter schematic
with the critical power paths/loops. Figure 25 shows an example
PCB component placement and routing with the same critical
power paths/loops as shown in the schematic.
1.
2.
3.
4.
By far, the highest di/dt in the asynchronous buck regulator occurs at the instant the high-side MOSFET turns on
and the capacitance of the asynchronous Schottky diode
(200 to 1000 pF) is quickly charged to VIN . The ceramic
input capacitors must deliver this fast, short pulse of current. Therefore the loop, from the ceramic input capacitors
through the high-side MOSFET and into the asynchronous
diode to ground, must be minimized. Ideally these components are all connected using only the top metal layer (that is,
do not use vias to other power/signal layers).
When the high-side MOSFET is on, current flows from the
input supply and capacitors, through the high-side MOSFET,
into the load via the output inductor, and back to ground.
This loop should be minimized and have relatively wide
traces.
When the high-side MOSFET is off, free-wheeling current
flows from ground, through the asynchronous diode, into the
load via the output inductor, and back to ground. This loop
should be minimized and have relatively wide traces.
The voltage on the SW node transitions from 0 V to VIN very
quickly and is the root cause of many noise issues. It is best
to place the asynchronous diode and output inductor close
to the A8580 to minimize the size of the SW polygon. Also,
keep low level analog signals (like FB and COMP) away
from the SW polygon.
5.
6.
7.
8.
9.
10.
11.
12.
13.
14.
Place the feedback resistor divider (RFB1 and RFB2) very
close to the FB pin. Ground this resistor divider as close as
possible to the A8580.
To have the highest output voltage accuracy, the output voltage sense trace (from VOUT to RFB1) should be connected
as close as possible to the load.
Place the compensation components (RZ, CZ, and CP ) as
close as possible to the COMP pin. Place vias to the GND
plane as close as possible to these components.
Place the soft start capacitor (CSS) as close as possible to the
SS pin. Place a via to the GND plane as close as possible to
this component.
Place the boot strap capacitor (CBOOT) near the BOOT pin
and keep the routing from this capacitor to the SW polygon
as short as possible.
When connecting the input and output ceramic capacitors,
use multiple vias to GND and place the vias as close as possible to the pads of the components.
To minimize PCB losses and improve system efficiency, the
input and output traces should be as wide as possible and be
duplicated on multiple layers, if possible.
To improve thermal performance, place multiple vias to the
GND plane around the anode of the asynchronous diode.
The thermal pad under the A8580 must connect to the GND
plane using multiple vias. More vias will ensure the lowest
junction temperature and highest efficiency.
EMI/EMC issues are always a concern. Allegro recommends
having component locations for an RC snubber from SW to
ground. The resistor should be 1206 size.
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115 Northeast Cutoff
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36
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
SW
VIN
VOUT
LO
CIN3
CIN2
CIN1
Q1
D1
Rsnub
CO1
CO2
CO3
CO4
LOAD
Csnub
2
1
3
Figure 24: Typical Buck Converter with Critical Paths/Loops Shown
Loop 1 (red). At the instant Q1 turns on, Schottky diode D1, which
is very capacitive, must be very quickly shut off (only 5 to 15 ns of
charging time). This spike of charging current must come from the
local input ceramic capacitor, CIN1. This spike of current is quite
large and will be an EMI/EMC issue if loop 1 (red) is not minimized.
Therefore, the input capacitor CIN1 and Schottky diode D1 must be
placed be on the same (top) layer, be located near each other, and be
grounded at virtually the same point on the PCB.
Loop 2 (magenta). When Q1 is off, free-wheeling inductor current
must flow from ground through diode D1 (SW will be at –Vf ), into
the output inductor, out to the load and return via ground. While Q1
is off, the voltage on the output capacitors decreases. The output
capacitors and Schottky diode D1 should be placed on the same
(top) layer, be located near each other, and be sharing a good, low
inductance ground connection.
Loop 3 (blue). When Q1 is on, current will flow from the input supply
and input capacitors through the output inductor and into the load
and the output capacitors. At this time the voltage on the output
capacitors increases.
2
3
1
Figure 25: Example PCB Component Placement and Routing
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37
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
Typical Applications Schematics
Vin
CIN1
4.7uF
50V
1210
CP
15pF
CIN2
4.7uF
50V
1210
RZ
26.1K
CIN3
0.47uF
100V
0805
CSS
22nF
CZ
560pF
CIN4
47nF
50V
0603
1
2
5
13
9
3
8
12
RFSET
59K
VIN
VIN
A8580
GND
GND
SW
SW
BOOT
COMP
SS
FSET
VREG
BIAS
FB
C1
1uF
15
14
CBOOT
47 nF
16
4
Mode
6
D1
2A/40V
SMP
10
VOUT
CO1
22uF
16V X7R
1210
CO2
22uF
16V X7R
1210
CO3
0.47uF
100V
0805
CO4
47nF
50V
0603
COUT = 38.2uF to 42.4uF total
at 3.3V bias and 10% tolerance
11
RFB2
47K
EN
LO 8.2 µH
RFB1 147K
3.3V
CFB
10pF
RPU
10K
SLEEPn
PWM/PFMn
NPOR
NPOR
7
Typical Application Schematic A. Configuration for VIN = 12 V , VOUT = 3.3 V, IOUT = 2.5 A at 425 kHz
Vin
CIN1
4.7uF
50V
1210
CP
8.0pF
CIN2
4.7uF
50V
1210
RZ
49.9K
CZ
270pF
CIN3
0.47uF
100V
0805
CSS
22nF
CIN4
47nF
50V
0603
1
2
5
13
9
3
8
12
RFSET
59K
VIN
VIN
A8580
GND
GND
COMP
SS
FSET
VREG
SW
SW
BOOT
BIAS
FB
C1
1uF
15
14
CBOOT
47 nF
16
4
Mode
6
10
RFB1
221K
VOUT
CO1
22uF
16V X7R
1210
RFB
499
CFB
8.0 pF
CO2
22uF
16V X7R
1210
CO3
22uF
16V X7R
1210
CO4
0.47uF
100V
0805
CO5
47nF
50V
0603
COUT = 53.9uF to 59.9uF total
at 5.0V bias and 10% tolerance
3.3 V
RPU
10 kΩ
SLEEPn
PWM/PFMn
D2
2A/40V
SMP
11
RFB2
42.2K
EN
LO 10 µH
NPOR
7
NPOR
Typical Application Schematic B. Configured for VIN = 12 V , VOUT = 5.0 V, IOUT = 2.5 A at 425 kHz
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115 Northeast Cutoff
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38
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
A8580
Package Outline Drawing
For Reference Only – Not for Tooling Use
(Reference MO-153 ABT)
Dimensions in millimeters. NOT TO SCALE
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
0.65
0.45
8º
0º
5.00 ±0.10
16
16
0.20
0.09
1.70
B
3 NOM 4.40 ±0.10
3.00
6.40 ±0.20
A
6.10
0.60 ±0.15
1.00 REF
1
2
3 NOM
1
0.25 BSC
2
Branded Face
3.00
SEATING PLANE
C
16X
0.10
SEATING
PLANE
C
0.30
0.19
GAUGE PLANE
C
PCB Layout Reference View
1.20 MAX
0.65 BSC
NNNNNNN
YYWW
LLLL
0.15
0.00
A
Terminal #1 mark area
B
Exposed thermal pad (bottom surface); dimensions may vary with device
C
Reference land pattern layout (reference IPC7351 SOP65P640X110-17M);
All pads a minimum of 0.20 mm from all adjacent pads; adjust as necessary
to meet application process requirements and PCB layout tolerances; when
mounting on a multilayer PCB, thermal vias at the exposed thermal pad land
can improve thermal dissipation (reference EIA/JEDEC Standard JESD51-5)
D
1
D Standard Branding Reference View
N = Device part number
= Supplier emblem
Y = Last two digits of year of manufacture
W = Week of manufacture
L = Characters 5-8 of lot number
Branding scale and appearance at supplier discretion
Figure 26: Package LP, 16-Pin TSSOP with Exposed Thermal Pad
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115 Northeast Cutoff
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1.508.853.5000; www.allegromicro.com
39
A8580
Wide Input Voltage, 2.4 MHz, 2.5 A Asynchronous Buck Regulator
With Low-IQ Standby, Sleep Mode, External Synchronization, and NPOR Output
Revision History
Number
Date
4
January 20, 2014
Description
Update Table 3
5
July 15, 2014
6
September 9, 2014
Updated Table 3 and reformatted document
Revised equation 8 on p. 26
7
February 11, 2015
Revised Table 2 and PWM Base Frequency section
8
December 5, 2016
Updated product status to Not for New Design
OS-CON is a trademark of SANYO North America Corporation.
Copyright ©2013-2016, Allegro MicroSystems, LLC
Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to
permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that
the information being relied upon is current.
Allegro’s products are not to be used in any devices or systems, including but not limited to life support devices or systems, in which a failure of
Allegro’s product can reasonably be expected to cause bodily harm.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its
use; nor for any infringement of patents or other rights of third parties which may result from its use.
For the latest version of this document, visit our website:
www.allegromicro.com
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115 Northeast Cutoff
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1.508.853.5000; www.allegromicro.com
40