AD AD8312 50 mhz to 3.5 ghz, 45 db rf detector Datasheet

50 MHz to 3.5 GHz, 45 dB
RF Detector
AD8312
FEATURES
Its high sensitivity allows measurement at low power levels, thus
reducing the amount of power that needs to be coupled to the
detector. It is essentially a voltage-responding device, with a
typical signal range of 1.25 mV to 224 mV rms or −45 dBm to
0 dBm, re 50 Ω.
Complete RF detector function
Typical range: −45 dBm to 0 dBm, re 50 Ω
Frequency response from 50 MHz to 3.5 GHz
Temperature-stable linear-in-dB response
Accurate to 3.5 GHz
Rapid response: 85/120 ns (rise/fall)
Low power: 12 mW at 2.7 V
For convenience, the signal is internally ac-coupled, using a
5 pF capacitor to a load of 3 kΩ in shunt with 1.3 pF. This highpass coupling, with a corner at approximately 16 MHz,
determines the lowest operating frequency. Therefore, the
source may be dc grounded.
APPLICATIONS
Cellular handsets (GSM, CDMA, WCDMA)
RSSI and TSSI for wireless terminal devices
Transmitter power measurement
The AD8312 output, VOUT, increases from close to ground to
about 1.2 V because the input signal level increases from
1.25 mV to 224 mV. A capacitor may be connected between the
VOUT and CFLT pins when it is desirable to increase the time
interval over which averaging of the input waveform occurs.
GENERAL DESCRIPTION
The AD8312 is a complete, low cost subsystem for the
measurement of RF signals in the frequency range of 50 MHz to
3.5 GHz. It has a typical dynamic range of 45 dB and is intended
for use in a wide variety of cellular handsets and other wireless
devices. It provides a wider dynamic range and better accuracy
than possible using discrete diode detectors. In particular, its
temperature stability is excellent over the full operating range of
−40°C to +85°C.
The AD8312 is available in a 6-ball, 1.0 mm × 1.5 mm, waferlevel chip scale package and consumes 4.2 mA from a 2.7 V to
5.5 V supply.
FUNCTIONAL BLOCK DIAGRAM
CFLT
V-I
VSET
I-V
VOUT
BAND-GAP
REFERENCE
VPOS
–
+
DET
DET
DET
DET
RFIN
10dB
10dB
10dB
OFFSET
COMPENSATION
10dB
AD8312
COMM
05260-001
DET
Figure 1.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
© 2005 Analog Devices, Inc. All rights reserved.
AD8312
TABLE OF CONTENTS
Specifications..................................................................................... 3
Input Coupling Options ........................................................ 14
Absolute Maximum Ratings............................................................ 6
Increasing the Logarithmic Slope ........................................ 15
ESD Caution.................................................................................. 6
Effect of Waveform Type on Intercept ................................ 15
Pin Configuration and Function Descriptions............................. 7
Temperature Drift .................................................................. 16
Typical Performance Characteristics ............................................. 8
Operation Above 2.5 GHz .................................................... 16
General Description ....................................................................... 12
Device Handling..................................................................... 16
Applications..................................................................................... 13
Evaluation Board.................................................................... 16
Basic Connections ...................................................................... 13
Outline Dimensions ....................................................................... 19
Transfer Function in Terms of Slope and Intercept ............... 13
Ordering Guide .......................................................................... 19
Filter Capacitor ....................................................................... 14
REVISION HISTORY
4/05—Revision 0: Initial Version
Rev. 0| Page 2 of 20
AD8312
SPECIFICATIONS
VS = 3 V, CFLT = open, TA = 25°C, light condition = 600 LUX, 52.3 Ω termination resistor at RFIN, unless otherwise noted.
Table 1.
Parameter
SIGNAL INPUT INTERFACE
Specified Frequency Range
Input Voltage Range
Equivalent Power Range
DC Resistance to COMM
MEASUREMENT MODE
f = 50 MHz
Input Impedance
±1 dB Dynamic Range
Maximum Input Level
Minimum Input Level
Slope
Intercept
Output Voltage—High Power In
Output Voltage—Low Power In
Temperature Sensitivity
f =100 MHz
Input Impedance
±1 dB Dynamic Range
Maximum Input Level
Minimum Input Level
Slope
Intercept
Output Voltage—High Power In
Output Voltage—Low Power In
Temperature Sensitivity
f = 900 MHz
Input Impedance
±1 dB Dynamic Range
Maximum Input Level
Minimum Input Level
Slope
Intercept
Output Voltage—High Power In
Output Voltage—Low Power In
Temperature Sensitivity
Conditions
RFIN (Pin 6)
Min
Internally ac-coupled
52.3 Ω external termination
Typ
Max
Unit
3.5
224
0
100
GHz
mV rms
dBm
kΩ
3050 || 1.4
50
42
3
−47
20.25
−51.5
0.841
0.232
Ω || pF
dB
dB
dBm
dBm
mV/dB
dBm
V
V
0.0010
0.0073
dB/°C
dB/°C
2900 || 1.3
48
40
2
−46
21.0
−50.5
0.850
0.222
Ω || pF
dB
dB
dBm
dBm
mV/dB
dBm
V
V
0.05
1.25
−45
VOUT (Pin 2) shorted to VSET (Pin 3), sinusoidal input signal
TA = 25°C
−40°C < TA < +85°C
±1 dB error
±1 dB error
PIN = −10 dBm
PIN = −40 dBm
PIN = −10 dBm
25°C ≤ TA ≤ +85°C
−40°C ≤ TA ≤ +25°C
TA = 25°C
−40°C < TA < +85°C
±1 dB error
±1 dB error
19.0
−56.0
PIN = −10 dBm
PIN = −40 dBm
PIN = −10 dBm
25°C ≤ TA ≤ +85°C
−40°C ≤ TA ≤ +25°C
TA = 25°C
−40°C < TA < +85°C
±1 dB error
±1 dB error
PIN = −10 dBm
PIN = −40 dBm
PIN = −10 dBm
25°C ≤ TA ≤ 85°C
−40°C ≤ TA ≤ +25°C
Rev. 0| Page 3 of 20
23.0
−47.0
0.0002
0.0060
dB/°C
dB/°C
890 || 1.15
49
42
1
−48.0
20.25
−51.9
0.847
0.237
Ω || pF
dB
dB
dBm
dBm
mV/dB
dBm
V
V
0.0019
0.0010
dB/°C
dB/°C
AD8312
Parameter
f = 1.9 GHz
Input Impedance
±1 dB Dynamic Range
Maximum Input Level
Minimum Input Level
Slope
Intercept
Output Voltage − High Power In
Output Voltage − Low Power In
Temperature Sensitivity
f = 2.2 GHz
Input Impedance
±1 dB Dynamic Range
Maximum Input Level
Minimum Input Level
Slope
Intercept
Output Voltage—High Power In
Output Voltage—Low Power In
Temperature Sensitivity
f = 2.5 GHz
Input Impedance
±1 dB Dynamic Range
Maximum Input Level
Minimum Input Level
Slope
Intercept
Output Voltage—High Power In
Output Voltage—Low Power In
Temperature Sensitivity
OUTPUT INTERFACE
Minimum Output Voltage
Maximum Output Voltage1
General Limit
Available Output Current
Residual RF (at 2f)
Output Noise
Fall Time
Rise Time
VSET INTERFACE
Input Resistance
Bias Current Source
Conditions
Min
TA = 25°C
−40°C < TA < +85°C
±1 dB error
±1 dB error
PIN = −10 dBm
PIN = −40 dBm
PIN = −10 dBm
25°C ≤ TA ≤ 85°C
−40°C ≤ TA ≤ +25°C
TA = 25°C
−40°C < TA < +85°C
±1 dB error
±1 dB error
PIN = −10 dBm
PIN = −40 dBm
PIN = −10 dBm
25°C ≤ TA ≤ 85°C
−40°C ≤ TA ≤ +25°C
TA = 25°C
−40°C < TA < +85°C
±1 dB error
±1 dB error
PIN = −10 dBm
PIN = −40 dBm
PIN = −10 dBm
25°C ≤ TA ≤ 85°C
−40°C ≤ TA ≤ +25°C
VOUT (Pin 2)
No signal at RFIN, RL ≥ 10 kΩ
RL ≥ 10 kΩ
2.7 V ≤ VS ≤ 5.5 V
Sourcing/sinking
f = 0.1 GHz (worst condition)
RF input = 2.2 GHz, −10 dBm, fNOISE = 100 kHz, CFLT open
Input level = off to 0 dBm, 90% to 10%
Input level = 0 dBm to off, 10% to 90%
VSET (Pin 3)
RFIN = −10 dBm; VSET = 1.2 V
Rev. 0| Page 4 of 20
1.8
VS − 1.2
Typ
Max
Unit
450 || 1.13
48
40
1
−47
19.47
−52.4
0.826
0.240
Ω || pF
dB
dB
dBm
dBm
mV/dB
dBm
V
V
0.004
0.005
dB/°C
dB/°C
430 || 1.09
48
40
1
−47
19.1
−52.1
0.803
0.230
Ω || pF
dB
dB
dBm
dBm
mV/dB
dBm
V
V
−0.0023
0.0055
dB/°C
dB/°C
400 || 1.03
49
40
1
−48
18.6
−51.2
0.762
0.204
Ω || pF
dB
dB
dBm
dBm
mV/dB
dBm
V
V
0.005
−0.0126
dB/°C
dB/°C
0.02
2.0
VS − 1
2/0.1
100
1.4
120
85
13
75
0.2
V
V
V
mA
µV
µV/√Hz
ns
ns
kΩ
µA
AD8312
Parameter
POWER INTERFACE
Supply Voltage
Quiescent Current
vs. Temperature
1
Conditions
VPOS (Pin 1)
−40°C ≤ TA ≤ +85°C
Increased output is possible when using an attenuator between VOUT and VSET to raise the slope.
Rev. 0| Page 5 of 20
Min
Typ
Max
Unit
2.7
2.8
3.0
4.2
4.3
5.5
5.7
V
mA
mA
AD8312
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
Supply Voltage VPOS
VOUT, VSET
Input Voltage
Equivalent Power
Internal Power Dissipation
θJA (WLCSP)
Maximum Junction Temperature
Operating Temperature Range
Storage Temperature Range
Value
5.5 V
0 V, VPOS
1.6 V rms
17 dBm
200 mW
200°C/W
125°C
−40°C to +85°C
−65°C to +150°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. 0| Page 6 of 20
AD8312
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
AD8312
1
6
RFIN
VOUT
2
5
COMM
VSET
3
4
CFLT
TOP VIEW
(Not to Scale)
05260-002
VPOS
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Ball No.
1
2
3
Mnemonic
VPOS
VOUT
VSET
4
CFLT
5
6
COMM
RFIN
Description
Positive Supply Voltage (VS), 2.7 V to 5.5 V.
Logarithmic Output. Output voltage increases with increasing input amplitude.
Setpoint Input. Connect VSET to VOUT for measurement-mode operation. The nominal logarithmic slope of
20 mV/dB can be increased to an arbitrarily high value by attenuating the signal between VOUT and VSET
(see the Increasing the Logarithmic Slope section).
Connection for an External Capacitor to Slow the Response of the Output. Capacitor is connected between
CFLT and VOUT.
Device Common (Ground).
RF Input.
Rev. 0| Page 7 of 20
AD8312
TYPICAL PERFORMANCE CHARACTERISTICS
VS = 3 V; TA = 25°C; CFLT = open; light condition = 600 LUX, 52.3 Ω termination; unless otherwise noted. Colors: +25°C = Black,
−40°C = Blue, +85°C = Red.
1.25
2.5
2.5
+85°C
+25°C
–40°C
2.0
1.5
1.00
1.5
0.5
0.75
0
0.50
–0.5
–1.0
–1.0
0.25
–2.0
0
–60
–50
–40
–30
–20
PIN (dBm)
–10
0
–2.5
10
–2.0
0
–60
Figure 3. VOUT and Log Conformance vs. Input Amplitude at 50 MHz;
Typical Device at −40°C, +25°C, and +85°C
1.25
–50
–40
–10
0
1.25
1.5
2.0
1.5
1.00
0
0.50
–0.5
VOUT (V)
0.5
1.0
ERROR (dB)
VOUT (V)
1.0
0.75
0.5
0.75
0
0.50
–0.5
–1.0
0.25
–1.0
0.25
–50
–40
–30
–20
PIN (dBm)
–10
0
–2.5
10
05260-004
–1.5
–2.0
0
–60
–50
–40
–10
0
1.25
2.0
1.5
1.00
1.5
1.0
0
0.50
–0.5
VOUT (V)
0.5
ERROR (dB)
VOUT (V)
1.0
0.75
0.5
0.75
0
0.50
–0.5
–1.0
–1.0
0.25
0.25
–50
–40
–30
–20
PIN (dBm)
–10
0
–2.5
10
05260-005
–1.5
–2.0
0
–60
–2.5
10
2.5
+85°C
+25°C
–40°C
2.0
1.00
–30
–20
PIN (dBm)
Figure 7. VOUT and Log Conformance vs. Input Amplitude at 2.2 GHz;
Typical Device at −40°C, +25°C, and +85°C
2.5
+85°C
+25°C
–40°C
–1.5
–2.0
0
–60
Figure 4. VOUT and Log Conformance vs. Input Amplitude at 100 MHz;
Typical Device at −40°C, +25°C, and +85°C
1.25
–2.5
10
2.5
+85°C
+25°C
–40°C
2.0
1.00
–30
–20
PIN (dBm)
Figure 6. VOUT and Log Conformance vs. Input Amplitude at 1.9 GHz;
Typical Device at −40°C, +25°C, and +85°C
2.5
+85°C
+25°C
–40°C
–1.5
05260-006
05260-003
–1.5
0.25
ERROR (dB)
–0.5
VOUT (V)
0
0.50
ERROR (dB)
VOUT (V)
0.5
0.75
ERROR (dB)
1.0
1.0
05260-007
1.00
2.0
ERROR (dB)
+85°C
+25°C
–40°C
Figure 5. VOUT and Log Conformance vs. Input Amplitude at 900 MHz;
Typical Device at −40°C, +25°C, and +85°C
Rev. 0| Page 8 of 20
–1.5
–2.0
0
–60
–50
–40
–30
–20
PIN (dBm)
–10
0
–2.5
10
Figure 8. VOUT and Log Conformance vs. Input Amplitude at 2.5 GHz;
Typical Device at −40°C, +25°C, and +85°C
05260-008
1.25
AD8312
2.5
2.5
+85°C
+25°C
–40°C
1.5
1.0
1.0
0.5
0
–0.5
–0.5
–1.0
–1.5
–1.5
–50
–40
–30
–20
PIN (dBm)
–10
0
–2.0
–2.5
–60
10
Figure 9. Distribution of Error at −40°C, +25°C, and +85°C After Ambient
Normalization vs. Input Amplitude at 50 MHz for 80 Devices
–30
–20
PIN (dBm)
–10
0
10
1.0
1.0
0.5
0
–0.5
0.5
0
–0.5
–1.0
–1.5
–1.5
05260-010
–1.0
–2.0
–40
–30
–20
PIN (dBm)
–10
0
–2.0
–2.5
–60
10
Figure 10. Distribution of Error at −40°C, +25°C, and +85°C After Ambient
Normalization vs. Input Amplitude at 100 MHz for 80 Devices
05260-013
ERROR (dB)
1.5
–50
+85°C
+25°C
–40°C
2.0
1.5
–50
–40
–30
–20
PIN (dBm)
–10
0
10
Figure 13. Distribution of Error at −40°C, +25°C, and +85°C After Ambient
Normalization vs. Input Amplitude at 2.2 GHz for 80 Devices
2.5
2.5
+85°C
+25°C
–40°C
1.5
1.0
1.0
ERROR (dB)
1.5
0.5
0
–0.5
0.5
0
–0.5
–1.0
–1.5
–1.5
05260-011
–1.0
–2.0
–50
–40
–30
–20
PIN (dBm)
+85°C
+25°C
–40°C
2.0
–10
0
–2.0
–2.5
–60
10
Figure 11. Distribution of Error at −40°C, +25°C, and +85°C After Ambient
Normalization vs. Input Amplitude at 900 MHz for 80 Devices
05260-014
2.0
–2.5
–60
–40
2.5
+85°C
+25°C
–40°C
2.0
–2.5
–60
–50
Figure 12. Distribution of Error at −40°C, +25°C, and +85°C After Ambient
Normalization vs. Input Amplitude at 1.9 GHz for 80 Devices
2.5
ERROR (dB)
0
–1.0
–2.0
ERROR (dB)
0.5
05260-012
ERROR (dB)
1.5
–2.5
–60
+85°C
+25°C
–40°C
2.0
05260-009
ERROR (dB)
2.0
–50
–40
–30
–20
PIN (dBm)
–10
0
10
Figure 14. Distribution of Error at −40°C, +25°C, and +85°C After Ambient
Normalization vs. Input Amplitude at 2.5 GHz for 80 Devices
Rev. 0| Page 9 of 20
AD8312
100ns/HORIZ DIV
RISE TIME 85ns
FALL TIME 120ns
500mV/
VERT DIV
VOUT
500mV/VERT DIV
VPOS
2V/VERT DIV
05260-015
PULSED RF
0.1GHz, 0dBm
200mV/VERT DIV
1µs/HORIZ DIV
Figure 15. VOUT Response Time, RF Off to 0 dBm
ROHDE & SCHWARZ
SMT06 GENERATOR
PULSE MODULATION
10MHz
HP8648B
REF OUTPUT
SIGNAL
GENERATOR
TRIG OUT
1
VPOS
RFIN
6
2
VOUT
COMM
5
AD8312
52.3Ω
CH3*
CFLT 4 NC
NC = NO CONNECT
TEKTRONIX
TDS51504
SCOPE
CH1
*50Ω TERMINATION
VPOS
2
VOUT
3
VSET
RFIN
6
52.3Ω
COMM 5
CFLT 4 NC
TRIG
NC = NO CONNECT
Figure 16. Test Setup for Pulse Response
TEKTRONIX
TDS784C
SCOPE
TEKTRONIX
P6204
FET PROBE
Figure 19. Test Setup for Power-On and Power-Off Response
+j1
10k
+j0.5
0.5
1
2
0.1GHz
0.9GHz
–j0.2
3.5GHz
2.2GHz
1k
100
0dBm
–10dBm
–20dBm
–40dBm
–60dBm
RF OFF
10
1.9GHz
2.5GHz
–j2
05260-017
1
–j0.5
1
3
10
30
100
300
FREQUENCY (kHz)
1k
3k
–j1
Figure 20. Noise Spectral Density of Output; CFLT = Open
Figure 17. Input Impedance vs. Frequency; No Termination Resistor on RFIN
Rev. 0| Page 10 of 20
10k
05260-020
NOISE SPECTRAL DENSITY (nV/ Hz)
+j2
+j0.2
0.2
1
TRIG
05260-016
VSET
49.9Ω
732Ω
–3dB
0.1µF
0
PULSE
OUT
AD811
–3dB
RF
SPLITTER
AD8312
3
TRIG
HP8116A
OUT
PULSE
GENERATOR
EXT
TRIG
20dBm RF OUT
RF OUT
+3dB
3V
Figure 18. Power-On and Power-Off Response
05260-019
RF INPUT
05260-018
VOUT
AD8312
Table 4. Typical Specifications at Selected Frequencies at 25°C (Mean and Σ)
±1 dB Dynamic Range1 (dBm)
µ
Slope (mV/dB)
σ
µ
Intercept (dBm)
σ
µ
High Point
σ
µ
Low Point
σ
Frequency (GHz)
0.05
20.25
0.3
−51.5
0.4
+3.0
0.12
−48.0
0.13
0.1
0.9
1.9
2.2
2.5
3.0
3.5
21.0
20.25
19.47
19.1
18.6
17.5
17.1
0.2
0.3
0.3
0.4
0.6
0.7
0.7
−50.5
−51.9
−52.4
−52.1
−51.2
−46.9
−42.6
0.4
0.4
0.6
0.85
1.2
2.5
2.5
+2.0
+0.2
+1.5
+1.5
+2.0
−4
−1
0.1
0.1
0.12
0.2
0.3
0.3
0.3
−46.0
−49.0
−48.8
−48.5
−47.7
−46
−39
0.1
0.2
0.3
0.4
0.5
0.4
0.3
1
Refer to Figure 23.
Rev. 0| Page 11 of 20
AD8312
GENERAL DESCRIPTION
measure of the RF input voltage with a slope and intercept
controlled by the design. For a fixed termination resistance at
the input of the AD8312, a given voltage corresponds to a
certain power level.
The AD8312 is a logarithmic amplifier (log amp) similar in
design to the AD8313; further details about the structure and
function may be found in the AD8313 data sheet and the data
sheets of other log amplifiers produced by ADI. Figure 21 shows
the main features of the AD8312 in block schematic form.
The external termination added before the AD8312 determines
the effective power scaling. This often takes the form of a
simple resistor (52.3 Ω provides a net 50 Ω input), but more
elaborate matching networks may be used. This impedance
determines the logarithmic intercept, the input power for which
the output would cross the baseline (VOUT = 0) if the function
were continuous for all values of input. Since this is never the
case for a practical log amp, the intercept refers to the value
obtained by the minimum-error, straight-line fit to the actual
graph of VOUT vs. input power. The quoted values assume a
sinusoidal (CW) signal. Where there is complex modulation, as
in CDMA, the calibration of the power response needs to be
adjusted accordingly. Where a true power (waveformindependent) response is needed, the use of an rms-responding
detector, such as the AD8361, should be considered.
The AD8312 combines two key functions needed for the
measurement of signal level over a moderately wide dynamic
range. First, it provides the amplification needed to respond to
small signals in a chain of four amplifier/limiter cells, each
having a small-signal gain of 10 dB and a bandwidth of
approximately 3.5 GHz. At the output of each amplifier stage is
a full-wave rectifier, essentially a square-law detector cell, which
converts the RF signal voltages to a fluctuating current with an
average value that increases with signal level. A further passive
detector stage is added ahead of the first stage. Therefore, there
are five detectors, each separated by 10 dB, spanning some 50
dB of dynamic range. The overall accuracy at the extremes of
this total range, viewed as the deviation from an ideal
logarithmic response, that is, the law-conformance error, can be
judged by reference to Figure 3 through Figure 8, which show
that errors across the central 40 dB are moderate. These figures
show how the conformance to an ideal logarithmic function
varies with temperature and frequency.
However, in terms of the logarithmic slope, the amount by
which the output VOUT changes for each decibel of input
change (voltage or power), is, in principle, independent of
waveform or termination impedance. In practice, it usually falls
off at higher frequencies because of the declining gain of the
amplifier stages and other effects in the detector cells. For the
AD8312, the slope at low frequencies is nominally 21.0 mV/dB,
falling almost linearly with frequency to about 18.6 mV/dB at
2.5 GHz. These values are sensibly independent of temperature
and almost totally unaffected by supply voltages of 2.7 V to 5.5 V.
The output of these detector cells is in the form of a differential
current, making their summation a simple matter. It can easily
be shown that such summation closely approximates a
logarithmic function. This result is then converted to a voltage
at the VOUT pin through a high gain stage. In measurement
modes, this output is connected back to a voltage-to-current
(V-to-I) stage, in such a manner that VOUT is a logarithmic
CFLT
V-I
VSET
I-V
VOUT
BAND-GAP
REFERENCE
VPOS
–
+
DET
DET
DET
DET
RFIN
10dB
10dB
OFFSET
COMPENSATION
10dB
10dB
AD8312
COMM
Figure 21. Block Schematic
Rev. 0| Page 12 of 20
05260-021
DET
AD8312
APPLICATIONS
1.2
BASIC CONNECTIONS
52.3Ω
VOUT
1
VPOS
2
VOUT
3
VSET
RFIN
6
1
0.6
0
0.4
–1
0
–60
–30
–20
PIN (dBm)
0
–3
10
TRANSFER FUNCTION IN TERMS OF SLOPE AND
INTERCEPT
4
05260-022
CF
OPTIONAL
(SEE TEXT)
–10
Figure 23. VOUT and Log Conformance Error vs.
Input Level vs. Input Level at 900 MHz
COMM 5
CFLT
INTERCEPT
–40
INPUT
OPTIONAL
(SEE TEXT)
–2
±3dB DYNAMIC RANGE
–50
ERROR (dB)
0.8
0.2
AD8312
VS
2
±1dB DYNAMIC RANGE
05260-023
0.1µF
1.0
VOUT (V)
Figure 22 shows the basic connections for measurement mode.
A supply voltage of 2.7 V to 5.5 V is required. The supply to the
VPOS pin should be decoupled with a low inductance 0.1 µF
surface-mount ceramic capacitor. A series resistor of about 10 Ω
may be added; this resistor slightly reduces the supply voltage to
the AD8312 (maximum current into the VPOS pin is
approximately 5.7 mA). Its use should be avoided in
applications where the power supply voltage is very low (that is,
2.7 V). A series inductor provides similar power supply filtering
with minimal drop in supply voltage.
3
VS= 2.7V
RT = 52.3Ω
Figure 22. Basic Connections for Operation in Measurement Mode
The AD8312 has an internal input coupling capacitor. This
eliminates the need for external ac coupling. In this example, a
broadband input match is achieved by connecting a
52.3 Ω resistor between RFIN and ground. This resistance
combines with the internal input impedance of approximately
3 kΩ to give an overall broadband input resistance of 50 Ω.
Several other coupling methods are possible; these are
described in the Input Coupling Options section.
The measurement mode is selected by connecting VSET to
VOUT, which establishes a feedback path and sets the
logarithmic slope to its nominal value. The peak voltage range
of the measurement extends from −49 dBm to 0 dBm at
0.9 GHz and is only slightly less at higher frequencies up to
2.5 GHz. At a slope of 21.0 mV/dB, this would amount to an
output span of 1.029 V. Figure 23 shows the transfer function
for VOUT at a supply voltage of 2.7 V and an input frequency of
900 MHz.
The load resistance on VOUT should not be lower than 4 kΩ so
that the full-scale output can be generated with the limited
available current of 1 mA maximum. Figure 23 shows the
logarithmic conformance under the same conditions.
The transfer function of the AD8312 is characterized in terms
of its slope and intercept. The logarithmic slope is defined as the
change in the RSSI output voltage for a 1 dB change at the input.
For the AD8312, the slope is nominally 20 mV/dB. Therefore, a
10 dB change at the input results in a change at the output of
approximately 200 mV. Figure 23 shows the range over which
the device maintains its constant slope. The dynamic range can
be defined as the range over which the error remains within a
certain band, usually ±1 dB or ±3 dB. In Figure 23, for example,
the ±1 dB dynamic range is approximately 51 dB (from
−49 dBm to +2 dBm).
The intercept is the point at which the extrapolated linear
response would intersect the horizontal axis (see Figure 23).
Using the slope and intercept, the output voltage can be
calculated for any input level within the specified input range by
VOUT = VSLOPE × (PIN − PO )
where:
VOUT is the demodulated and filtered RSSI output.
VSLOPE is the logarithmic slope, expressed in V/dB.
PIN is the input signal, expressed in decibels relative to some
reference level (dBm in this case).
PO is the logarithmic intercept, expressed in decibels relative to
the same reference level.
For example, at an input level of −27 dBm, the output voltage is
[
]
VOUT = 0.020 V/dB × − 27 dBm − (− 50 dBm ) = 0.46 V
Rev. 0| Page 13 of 20
AD8312
Filter Capacitor
The video bandwidth of VOUT is approximately 3.5 MHz. In
CW applications where the input frequency is much higher
than this, no further filtering of the demodulated signal is
required. Where there is a low frequency modulation of the
carrier amplitude, however, the low-pass corner must be
reduced by the addition of an external filter capacitor, CF (see
Figure 22). The video bandwidth is related to CF by
1
Video Bandwidth =
2 π × 13 kΩ × (3.5 pF + C F )
The impedance matching characteristics of a reactive matching
network provide voltage gain ahead of the AD8312, which
increases device sensitivity (see Table 5). The voltage gain is
calculated by
R2
R1
Voltage GaindB = 20 log 10
where:
R2 is the input impedance of the AD8312.
R1 is the source impedance to which the AD8312 is being
matched.
At frequencies above 2 GHz, the input impedance drops below
450 Ω; therefore, it is appropriate to use a larger shunt resistor
value. This value is calculated by plotting the input impedance
(resistance and capacitance) on a Smith Chart and by choosing
the best shunt resistor value to bring the input impedance
closest to the center of the chart (see Figure 17). At 2.5 GHz, a
shunt resistor of 57.6 Ω is recommended.
Note that this gain is only achieved for a perfect match.
Component tolerances and the use of standard values tend to
reduce gain.
50Ω
RFIN
CC
CIN
RIN
05260-024
RSHUNT
52.3Ω
VBIAS
Figure 24. Broadband Resistive Method for Input Coupling
AD8312
50Ω SOURCE
50Ω
X1
RFIN
CC
X2
CIN
RIN
VBIAS
Figure 25. Narrow-Band Reactive Method for Input Coupling
A reactive match can also be implemented as shown in Figure 25.
This is not recommended at low frequencies because device
tolerances dramatically vary the quality of the match due to the
large input resistance. For low frequencies, Figure 24 or Figure 26
is recommended.
In Figure 25, the matching components are drawn as general
reactances. Depending on the frequency, the input impedance
at that frequency and the availability of standard value
components, either a capacitor or an inductor, is used. As in the
previous case, the input impedance at a particular frequency is
plotted on a Smith Chart and matching components are chosen
(Shunt or Series L, or Shunt or Series C) to move the impedance
to the center of the chart. Matching components for specific
frequencies can be calculated using the Smith Chart (see
Figure 17). Table 5 outlines the input impedances for some
commonly used frequencies.
AD8312
50Ω SOURCE
05260-025
The internal 5 pF coupling capacitor of the AD8312, along with
the low frequency input impedance of 3 kΩ, gives a high-pass
input corner frequency of approximately 16 MHz. This sets the
minimum operating frequency. Figure 24 to Figure 26 show
three options for input coupling. A broadband resistive match
can be implemented by connecting a shunt resistor to ground at
RFIN (see Figure 24). This 52.3 Ω resistor (other values can also
be used to select different overall input impedances) combines
with the input impedance of the AD8312 (2.9 kΩ || 1.3 pF) to
give a broadband input impedance of 50 Ω. While the input
resistance and capacitance (RIN and CIN) varies by approximately
±20% from device to device, the dominance of the external
shunt resistor means that the variation in the overall input
impedance is close to the tolerance of the external resistor.
AD8312
RFIN
STRIPLINE
RATTN
CC
CIN
RIN
VBIAS
05260-026
Input Coupling Options
Figure 26. Series Attenuation Method for Input Coupling
Figure 26 shows a third method for coupling the input signal
into the AD8312, which is applicable in applications where the
input signal is larger than the input range of the log amp. A
series resistor, connected to the RF source, combines with the
input impedance of the AD8312 to resistively divide the input
signal being applied to the input. This has the advantage of very
little power being tapped off in RF power transmission
applications.
Rev. 0| Page 14 of 20
AD8312
4.0
Table 5. Input Impedance for Select Frequency
S11
Imaginary
−0.043
−0.081
−0.535
−0.891
−0.832
−0.845
−0.849
−0.826
Impedance Ω
(Series)
1090 − j 1461
422.6 − j 1015
25.6 − j 148.5
11.5 − j 72.69
9.91 − j 64.74
9.16 − j 59.91
8.83 − j 57.21
10.5 − j 58.54
3.5
3.0
2.5
2.0
3.0×
1.5
2.7V
SUPPLY
2.0×
1.0
1.0×
0.5
Increasing the Logarithmic Slope
The nominal logarithmic slope of 20 mV/dB can be increased to
an arbitrarily high value by attenuating the signal between VOUT
and VSET, as shown in Figure 27. The ratio R1/R2 is set by
0
–60
–50
–40
–30
–20
PIN (dBm)
–10
0
10
Figure 28. VOUT vs. Input Level at Various Logarithmic Slopes
⎛ New Slope ⎞
⎟ −1
R1/R2 = ⎜
⎜ Original Slope ⎟
⎠
⎝
Effect of Waveform Type on Intercept
In the example shown, two 2 kΩ resistors combine to change
the slope at 1900 MHz from approximately 20 mV/dB to
40 mV/dB. Note that R2 is in parallel with the input resistance of
VSET, typically 13 kΩ. Therefore, the exact R1/R2 ration may vary.
AD8312
R1
2kΩ
~40mV/dB
@ 1900MHz
R2
2kΩ
05260-027
VOUT
VSET
5.0V
SUPPLY
05260-028
Real
0.967
0.962
0.728
0.322
0.230
0.165
0.126
0.146
VOUT (V)
Frequency
(GHz)
0.05
0.1
0.9
1.9
2.2
2.5
3.0
3.5
Figure 27. Increasing the Output Slope
The slope can be increased to higher levels, as shown in
Figure 28. This, however, reduces the usable dynamic range of
the device, depending on the supply voltage.
Output loading should be considered when choosing resistor
values for slope adjustment to ensure proper output swing.
Note that the load resistance on VOUT should not be lower
than 4 kΩ in order that the full-scale output can be generated
with the limited available current of 1 mA.
Although specified for input levels in dBm (dB relative to
1 mW), the AD8312 fundamentally responds to voltage and not
to power. A direct consequence of this characteristic is that
input signals of equal rms power but differing crest factors,
produce different results at the log amplifier’s output.
The effect of differing signal waveforms is to shift the effective
value of the intercept upwards or downwards. Graphically, this
looks like a vertical shift in the log amplifier’s transfer function.
The logarithmic slope, however, is not affected. For example,
consider the case of the AD8312 being alternately fed by an
unmodulated sine wave and by a 64 QAM signal of the same
rms power. The AD8312’s output voltage differs by the
equivalent of 1.6 dB (31 mV) over the complete dynamic range
of the device (with the output for a 64 QAM input being lower).
Figure 29 shows the transfer function of the AD8312 when
driven by both an unmodulated sine wave and several different
signal waveforms. For precision operation, the AD8312 should
be calibrated for each signal type that is driving it. To measure
the rms power of a 64 QAM input, for example, the mV
equivalent of the dB value (19.47 mV/dB × 1.6 dB) should be
subtracted from the output voltage of the AD8312.
Rev. 0| Page 15 of 20
AD8312
1.0
4
0.9
3
0.8
3
0.7
2
0.6
1
0.5
0
0.4
–1
0.3
–2
0.2
–3
0.1
–4
1
CW
VOUT (V)
2
VOUT (V)
5
+85°C
+25°C
–40°C
IS-95 REV
0
4
ERROR (dB)
5
–1
–3
64 QAM
05260-029
–50
–40
–30
–20
INPUT (dBm)
–10
0
–5
–45
–35
10
–25
PIN (dBm)
–15
–5
5
Figure 31.Output Voltage and Error at −40°C, +25°C, and +85°C After
Ambient Normalization vs. Input Amplitude at 3.0 GHz for 60 Devices
Figure 29. Shift in Transfer Function due to
Several Different Signal Waveforms
1.0
5
+85°C
+25°C
–40°C
0.9
Figure 30 shows the log slope and error over temperature for a
0.9 GHz input signal. Error due to drift over temperature
consistently remains within ±0.5 dB and only begins to exceed
this limit when the ambient temperature goes above 70°C. For
all frequencies using a reduced temperature range, higher
measurement accuracy is achievable.
0.8
3
0.7
2
0.6
1
0.5
0
0.4
–1
2.5
0.3
–2
2.0
0.2
–3
1.5
0.1
–4
1.3
1.0
VOUT (V)
Temperature Drift
0
–55
0.5
0
0.5
–0.5
+85°C
+70°C
+25°C
0°C
–10°C
–20°C
–40°C
0.3
0
–60
–50
–40
–30
–20
PIN (dBm)
–10
ERROR (dB)
0.8
–1.5
–2.0
0
–2.5
10
Figure 30. Typical Drift at 900 GHz for Various Temperatures
Operation Above 2.5 GHz
The AD8312 works at high frequencies, but exhibits slightly
higher output voltage temperature drift. Figure 31 and Figure 32
show the transfer functions and error distributions of a large
population of devices at 3.0 GHz and 3.5 GHz over
temperature. Due to the repeatability of the drift from part-topart, compensation can be applied to reduce the effects of
temperature drift. In the case of the 3.5 GHz distribution, an
intercept correction of 2.0 dB at 85°C would improve the
accuracy of the distribution to ±2 dB over a +40 dB range.
–5
–45
–35
–25
PIN (dBm)
–15
–5
5
Figure 32. Output Voltage and Error at −40°C, +25°C, and +85°C After
Ambient Normalization vs. Input Amplitude at 3.5 GHz for 30 Devices
Device Handling
–1.0
05260-030
VOUT (V)
1.0
4
ERROR (dB)
–5
–60
0
–55
05260-032
WCDMA 64-CH
–4
05260-031
–2
The wafer-level chip scale package consists of solder bumps
connected to the active side of the die. The part is lead-free with
95.5% tin, 4.0% silver, and 0.5% copper solder bump composition. The WLCSP package can be mounted on printed circuit
boards using standard surface-mount assembly techniques;
however, caution should be taken to avoid damaging the die.
See the AN-617 application note for additional information.
WLCSP devices are bumped die, and exposed die can be
sensitive to light condition, which can influence specified limits.
Evaluation Board
Figure 33 shows the schematic of the AD8312 evaluation board.
The layout and silkscreen of the component and circuit sides
are shown in Figure 34 to Figure 37. The board is powered by a
single supply in the 2.7 V to 5.5 V range. The power supply is
decoupled by a single 0.1 µF capacitor.
Table 6 details the various configuration options of the
evaluation board.
Rev. 0| Page 16 of 20
AD8312
C2
0.1µF
R1
52.3Ω
AD8312
VPOS
1
VPOS
RFIN 6
2
VOUT
COMM 5
3
VSET
CFLT 4
INPUT
VOUT
C4
(OPEN)
R7
0Ω
VSET
R5
(OPEN)
R8
(OPEN)
R6
(OPEN)
TO EDGE
CONNECTOR
R4
0Ω
R2
(OPEN)
C3
(OPEN)
TO EDGE
CONNECTOR
05260-033
R3
0Ω
05260-037
05260-036
Figure 33. Evaluation Board Schematic
05260-034
05260-035
Figure 36. Silkscreen of Circuit Side (WLCSP)
Figure 34. Silkscreen of Component Side (WLCSP)
Figure 37. Layout of Circuit Side (WLCSP)
Figure 35. Layout of Component Side (WLCSP)
Rev. 0| Page 17 of 20
AD8312
Table 6. Evaluation Board Configuration Options
Component
VPOS, GND
C2
Function
Supply and Ground Vector Pins.
Power Supply Decoupling. The nominal supply decoupling consists of a 0.1 µF capacitor (C1).
R1
Input Interface. The 52.3 Ω resistor in Position R1 combines with the AD8312’s internal input impedance
to give a broadband input impedance of around 50 Ω.
Slope Adjust. By installing resistors in R2 and R4, the nominal slope of 20 mV/dB can be changed. See the
Increasing the Logarithmic Slope section for more details.
R2, R4
C3
R3, R8, C4
Filter Capacitor. The response time of VOUT can be modified by placing a capacitor between CFLT (Pin 4)
and VOUT (Pin 2).
Output Interface. R3, R8, and C4 can be used to check the response of VOUT to capacitive and resistive
loading. R3/R8 can be used to attenuate VOUT.
R7
VSET Interface. R7 can be used to reduce capacitive loading from transmission lines.
R5, R6
Alternate Interface. R5 and R6 allow for VOUT and VSET to be accessible from the edge connector, which
is only used for characterization.
Rev. 0| Page 18 of 20
Default
Condition
Not Applicable
C2 = 0.1 µF
(Size 0603)
R1 = 52.3 Ω
(Size 0603)
R2 = Open
(Size 0402)
R4 = 0 Ω
(Size 0402)
C3 = Open
(Size 0603)
R3 = 0 Ω
(Size 0603)
R8 = C4 = Open
(Size 0402)
R7 = 0 Ω
(Size 0603)
R5 = R6 = Open
(Size 0402)
AD8312
OUTLINE DIMENSIONS
0.675
0.595
0.515
0.380
0.355
0.330
1.00
0.95
0.90
SEATING
PLANE
A1 BALL
CORNER
B
A
1
0.345
0.295
0.245
1.50
1.45
1.40
1.00
BSC
2
0.50
BSC
3
TOP VIEW
(BUMP SIDE DOWN)
0.270
0.240
0.210
0.075
COPLANARITY
0.50 BSC
BOTTOM VIEW
(BUMP SIDE UP)
Figure 38. 6-Ball Wafer-Level Chip Scale Package [WLCSP]
(CB-6)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD8312ACBZ-P71
AD8312ACBZ-P21
AD8312-EVAL
1
Temperature Range
–40°C to +85°C
–40°C to +85°C
Package Description
6-Ball WLCSP, 7” Pocket Tape and Reel
6-Ball WLCSP, 7” Pocket Tape and Reel
Evaluation Board
Z = Pb-free part.
Rev. 0| Page 19 of 20
Package
Outline
CB-6
CB-6
Branding
Information
Q00
Q00
Ordering
Quantity
3000
250
AD8312
NOTES
©2005 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D05260–0–4/05(0)
Rev. 0| Page 20 of 20