AD AD812 Dual, current feedback low power op amp Datasheet

a
Dual, Current Feedback
Low Power Op Amp
AD812
PIN CONFIGURATION
8-Lead Plastic
Mini-DIP and SOIC
FEATURES
Two Video Amplifiers in One 8-Lead SOIC Package
Optimized for Driving Cables in Video Systems
Excellent Video Specifications (RL = 150 ⍀):
Gain Flatness 0.1 dB to 40 MHz
0.02% Differential Gain Error
0.02ⴗ Differential Phase Error
Low Power
Operates on Single +3 V Supply
5.5 mA/Amplifier Max Power Supply Current
High Speed
145 MHz Unity Gain Bandwidth (3 dB)
1600 V/␮s Slew Rate
Easy to Use
50 mA Output Current
Output Swing to 1 V of Rails (150 ⍀ Load)
PRODUCT DESCRIPTION
The AD812 is a low power, single supply, dual video amplifier.
Each of the amplifiers have 50 mA of output current and are
optimized for driving one back-terminated video load (150 Ω)
each. Each amplifier is a current feedback amplifier and features gain flatness of 0.1 dB to 40 MHz while offering differential gain and phase error of 0.02% and 0.02°. This makes the
AD812 ideal for professional video electronics such as cameras
and video switchers.
–IN1 2
8 V+
7 OUT2
+
6 –IN2
+IN1 3
+
V–
4
AD812
5 +IN2
The AD812 offers low power of 4.0 mA per amplifier max (VS =
+5 V) and can run on a single +3 V power supply. The outputs
of each amplifier swing to within one volt of either supply rail to
easily accommodate video signals of 1 V p-p. Also, at gains of
+2 the AD812 can swing 3 V p-p on a single +5 V power supply. All this is offered in a small 8-lead plastic DIP or 8-lead
SOIC package. These features make this dual amplifier ideal
for portable and battery powered applications where size and
power is critical.
The outstanding bandwidth of 145 MHz along with 1600 V/µs
of slew rate make the AD812 useful in many general purpose
high speed applications where a single +5 V or dual power supplies up to ± 15 V are available. The AD812 is available in the
industrial temperature range of –40°C to +85°C.
0.4
0.06
G = +2
RL = 150V
0.3
0.04
0.2
0.1
DIFFERENTIAL PHASE – Degrees
NORMALIZED GAIN – dB
DIFFERENTIAL GAIN
0
–0.1
–0.2
VS = 615V
–0.3
65V
–0.4
5V
–0.5
3V
–0.6
100k
1M
10M
FREQUENCY – Hz
100M
Figure 1. Fine-Scale Gain Flatness vs. Frequency, Gain
= +2, RL = 150 Ω
0.08
0.02
0.06
DIFFERENTIAL GAIN – %
APPLICATIONS
Video Line Driver
Professional Cameras
Video Switchers
Special Effects
OUT1 1
DIFFERENTIAL PHASE
0.04
0.02
0
5
6
7
8
9
10
11
12
SUPPLY VOLTAGE – 6Volts
13
14
15
Figure 2. Differential Gain and Phase vs. Supply Voltage,
Gain = +2, RL = 150 Ω
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1998
AD812–SPECIFICATIONS
Dual Supply (@ T = +25ⴗC, R = 150 ⍀, unless otherwise noted)
A
L
Model
DYNAMIC PERFORMANCE
–3 dB Bandwidth
Bandwidth for 0.1 dB Flatness
Conditions
VS
Min
G = +2, No Peaking
±5 V
± 15 V
± 15 V
±5 V
± 15 V
±5 V
± 15 V
±5 V
± 15 V
50
75
100
20
25
275
1400
Gain = +1
G = +2
Slew Rate1
G = +2, RL = 1 kΩ
20 V Step
G = –1, RL = 1 kΩ
Settling Time to 0.1%
G = –1, RL = 1 kΩ
VO = 3 V Step
VO = 10 V Step
NOISE/HARMONIC PERFORMANCE
Total Harmonic Distortion
Input Voltage Noise
Input Current Noise
Differential Gain Error
fC = 1 MHz, RL = 1 kΩ
f = 10 kHz
f = 10 kHz, +In
f = 10 kHz, –In
NTSC, G = +2, RL = 150 Ω
Differential Phase Error
DC PERFORMANCE
Input Offset Voltage
TMIN –TMAX
Offset Drift
–Input Bias Current
TMIN –T MAX
+Input Bias Current
Open-Loop Voltage Gain
Open-Loop Transresistance
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
Input Common Mode
Voltage Range
Common-Mode Rejection Ratio
Input Offset Voltage
–Input Current
+Input Current
Input Offset Voltage
–Input Current
+Input Current
TMIN –T MAX
VO = ± 2.5 V, RL = 150 Ω
TMIN –T MAX
VO = ± 10 V, RL = 1 kΩ
TMIN –T MAX
VO = ± 2.5 V, RL = 150 Ω
TMIN –T MAX
VO = ± 10 V, RL = 1 kΩ
TMIN –T MAX
±5 V
± 15 V
50
40
ns
ns
± 15 V
± 5 V, ± 15 V
± 5 V, ± 15 V
± 5 V, ± 15 V
±5 V
± 15 V
±5 V
± 15 V
–90
3.5
1.5
18
0.05
0.02
0.07
0.02
dBc
nV/√Hz
pA/√Hz
pA/√Hz
%
%
Degrees
Degrees
± 5 V, ± 15 V
2
± 5 V, ± 15 V
± 5 V, ± 15 V
15
7
± 5 V, ± 15 V
0.3
±5 V
± 15 V
±5 V
± 15 V
±5 V
VCM = ± 12 V
± 15 V
–2–
Units
MHz
MHz
MHz
MHz
MHz
V/µs
V/µs
V/µs
V/µs
68
69
76
75
350
270
450
370
0.1
0.06
0.15
0.06
5
12
25
38
1.5
2.0
76
82
550
800
15
65
1.7
4.0
13.5
±5 V
± 15 V
VCM = ± 2.5 V
Max
65
100
145
30
40
425
1600
250
600
± 15 V
+Input
–Input
+Input
AD812A
Typ
51
55
58
2
0.07
60
1.5
0.05
mV
mV
µV/°C
µA
µA
µA
µA
dB
dB
dB
dB
kΩ
kΩ
kΩ
kΩ
MΩ
Ω
pF
±V
±V
3.0
0.15
3.3
0.15
dB
µA/V
µA/V
dB
µA/V
µA/V
REV. B
AD812
Model
OUTPUT CHARACTERISTICS
Output Voltage Swing
Conditions
VS
Min
RL = 150 Ω, TMIN –TMAX
RL = 1 kΩ, TMIN –TMAX
±5 V
± 15 V
±5 V
± 15 V
± 15 V
3.5
13.6
30
40
Output Resistance
MATCHING CHARACTERISTICS
Dynamic
Crosstalk
Gain Flatness Match
DC
Input offset Voltage
–Input Bias Current
POWER SUPPLY
Operating Range
Quiescent Current
Units
±V
±V
mA
mA
mA
± 15 V
15
Ω
G = +2, f = 5 MHz
G = +2, f = 40 MHz
± 5 V, ± 15 V
± 15 V
–75
0.1
dB
dB
TMIN –TMAX
TMIN –TMAX
± 5 V, ± 15 V
± 5 V, ± 15 V
0.5
2
3.6
25
mV
µA
Per Amplifier
±5 V
± 15 V
± 15 V
3.5
4.5
± 18
4.0
5.5
6.0
V
mA
mA
mA
0.6
0.05
dB
µA/V
µA/V
G = +2, RF = 715 Ω
VIN = 2 V
Open-Loop
TMIN –TMAX
Power Supply Rejection Ratio
Input Offset Voltage
–Input Current
+Input Current
Max
3.8
14.0
40
50
100
Output Current
Short Circuit Current
AD812A
Typ
VS = ± 1.5 V to ± 15 V
± 1.2
70
80
0.3
0.005
NOTES
1
Slew rate measurement is based on 10% to 90% rise time in the specified closed-loop gain.
Specifications subject to change without notice.
Single Supply
(@ TA = +25ⴗC, RL = 150 ⍀, unless otherwise noted)
Model
DYNAMIC PERFORMANCE
–3 dB Bandwidth
Bandwidth for 0.1 dB
Flatness
Slew Rate1
NOISE/HARMONIC PERFORMANCE
Input Voltage Noise
Input Current Noise
Differential Gain Error2
Differential Phase Error 2
REV. B
AD812A
Typ
Conditions
VS
Min
G = +2, No Peaking
+5 V
+3 V
35
30
50
40
MHz
MHz
G = +2
+5 V
+3 V
+5 V
+3 V
13
10
20
18
125
60
MHz
MHz
V/µs
V/µs
3.5
1.5
18
0.07
0.15
0.06
0.15
nV/√Hz
pA/√Hz
pA/√Hz
%
%
Degrees
Degrees
G = +2, RL = 1 kΩ
f = 10 kHz
f = 10 kHz, +In
f = 10 kHz, –In
NTSC, G = +2, RL = 150 Ω
G = +1
G = +2
G = +1
–3–
+5 V, +3 V
+5 V, +3 V
+5 V, +3 V
+5 V
+3 V
+5 V
+3 V
Max
Units
AD812–SPECIFICATIONS
Single Supply (Continued)
Model
Conditions
VS
DC PERFORMANCE
Input Offset Voltage
Min
AD812A
Typ
+5 V, +3 V
1.5
+5 V, +3 V
+5 V, +3 V
7
2
+5 V, +3 V
0.2
TMIN –TMAX
Offset Drift
–Input Bias Current
TMIN –TMAX
+Input Bias Current
Open-Loop Voltage Gain
Open-Loop Transresistance
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
Input Common Mode
Voltage Range
Common-Mode Rejection Ratio
Input Offset Voltage
–Input Current
+Input Current
Input Offset Voltage
–Input Current
+Input Current
OUTPUT CHARACTERISTICS
Output Voltage Swing p-p
TMIN –TMAX
VO = +2.5 V p-p
VO = +0.7 V p-p
VO = +2.5 V p-p
VO = +0.7 V p-p
+5 V
+3 V
+5 V
+3 V
+Input
–Input
+Input
MATCHING CHARACTERISTICS
Dynamic
Crosstalk
Gain Flatness Match
DC
Input offset Voltage
–Input Bias Current
POWER SUPPLY
Operating Range
Quiescent Current
VCM = 1.25 V to 3.75 V
1.0
1.0
+5 V
52
+3 V
RL = 1 kΩ, TMIN –TMAX
RL = 150 Ω, TMIN –TMAX
+5 V
+5 V
+3 V
+5 V
+3 V
+5 V
G = +2, RF = 715 Ω
VIN = 1 V
Units
4.5
7.0
mV
mV
µV/°C
µA
µA
µA
µA
dB
dB
kΩ
kΩ
20
30
1.5
2.0
73
70
400
300
15
90
2
+5 V
+3 V
VCM = 1 V to 2 V
3.0
2.8
1.0
20
15
4.0
2.0
55
3
0.1
52
3.5
0.1
5.5
0.2
MΩ
Ω
pF
V
V
dB
µA/V
µA/V
dB
µA/V
µA/V
3.2
3.1
1.3
30
25
40
V p-p
V p-p
V p-p
mA
mA
mA
dB
dB
G = +2, f = 5 MHz
G = +2, f = 20 MHz
+5 V, +3 V
+5 V, +3 V
–72
0.1
TMIN –TMAX
TMIN –TMAX
+5 V, +3 V
+5 V, +3 V
0.5
2
3.5
25
mV
µA
Per Amplifier
+5 V
+3 V
+5 V
3.2
3.0
36
4.0
3.5
4.5
V
mA
mA
mA
0.6
0.05
dB
µA/V
µA/V
2.4
TMIN –TMAX
Power Supply Rejection Ratio
Input Offset Voltage
–Input Current
+Input Current
250
+5 V
+5 V
Output Current
Short Circuit Current
67
Max
VS = +3 V to +30 V
70
TRANSISTOR COUNT
80
0.3
0.005
56
NOTES
1
Slew rate measurement is based on 10% to 90% rise time in the specified closed-loop gain.
2
Single supply differential gain and phase are measured with the ac coupled circuit of Figure 53.
Specifications subject to change without notice.
–4–
REV. B
AD812
ABSOLUTE MAXIMUM RATINGS 1
MAXIMUM POWER DISSIPATION
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V
Internal Power Dissipation2
Plastic (N) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Watts
Small Outline (R) . . . . . . . . . . . . . . . . . . . . . . . . . . 0.9 Watts
Input Voltage (Common Mode) . . . . . . . . . . . . . . . . . . . . ± VS
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . ± 1.2 V
Output Short Circuit Duration
. . . . . . . . . . . . . . . . . . . . . . Observe Power Derating Curves
Storage Temperature Range N, R . . . . . . . . . –65°C to +125°C
Operating Temperature Range . . . . . . . . . . . . –40°C to +85°C
Lead Temperature Range (Soldering, 10 sec) . . . . . . . +300°C
The maximum power that can be safely dissipated by the
AD812 is limited by the associated rise in junction temperature.
The maximum safe junction temperature for the plastic encapsulated parts is determined by the glass transition temperature
of the plastic, about 150°C. Exceeding this limit temporarily
may cause a shift in parametric performance due to a change in
the stresses exerted on the die by the package. Exceeding a
junction temperature of 175°C for an extended period can result
in device failure.
While the AD812 is internally short circuit protected, this may
not be sufficient to guarantee that the maximum junction temperature (150 degrees) is not exceeded under all conditions. To
ensure proper operation, it is important to observe the derating
curves.
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
Specification is for device in free air: 8-lead plastic package: θJA = 90°C/Watt;
8-lead SOIC package: θJA = 150°C/Watt.
It must also be noted that in high (noninverting) gain configurations (with low values of gain resistor), a high level of input
overdrive can result in a large input error current, which may
result in a significant power dissipation in the input stage. This
power must be included when computing the junction temperature rise due to total internal power.
ORDERING GUIDE
AD812AN
–40°C to +85°C
AD812AR
–40°C to +85°C
AD812AR-REEL
AD812AR-REEL7
Package
Description
Package
Option
2.0
MAXIMUM POWER DISSIPATION – Watts
Temperature
Range
Model
8-Lead Plastic DIP N-8
8-Lead Plastic SOIC SO-8
13" Reel
7" Reel
METALIZATION PHOTO
Dimensions shown in inches and (mm).
0.0783
(1.99)
V+
8
OUT2
7
–IN2
6
TJ = +1508C
8-LEAD MINI-DIP PACKAGE
1.5
1.0
8-LEAD SOIC PACKAGE
0.5
0
–50 –40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90
AMBIENT TEMPERATURE – 8C
5 +IN2
Figure 3. Plot of Maximum Power Dissipation vs.
Temperature
0.0539
(1.37)
4 V–
1
OUT1
2
–IN1
3
+IN1
4
V–
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD812 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
REV. B
–5–
WARNING!
ESD SENSITIVE DEVICE
AD812–Typical Performance Characteristics
16
14
TOTAL SUPPLY CURRENT – mA
COMMON-MODE VOLTAGE RANGE – 6Volts
20
15
10
5
0
0
5
10
15
SUPPLY VOLTAGE – 6Volts
VS = 615V
12
10
VS = 65V
8
6
4
–60
20
–40
–20
0
20
40
60
80
100
120
140
JUNCTION TEMPERATURE – 8C
Figure 4. Input Common-Mode Voltage Range vs. Supply
Voltage
Figure 7. Total Supply Current vs. Junction Temperature
10
20
TA = +25 C
TOTAL SUPPLY CURRENT – mA
OUTPUT VOLTAGE – V p-p
NO LOAD
15
10
RL = 150V
5
9
8
7
6
5
0
0
5
10
15
SUPPLY VOLTAGE – 6Volts
0
2
4
20
Figure 5. Output Voltage Swing vs. Supply Voltage
10
12
6
8
SUPPLY VOLTAGE – 6Volts
14
16
Figure 8. Total Supply Current vs. Supply Voltage
30
25
615V SUPPLY
20
INPUT BIAS CURRENT – mA
OUTPUT VOLTAGE – Volts p-p
25
20
15
10
65V SUPPLY
15
–IB, VS = 65V
10
5
0
+IB, VS = 65V, 615V
–5
–10
–IB, VS = 615V
–15
5
–20
0
10
100
1k
LOAD RESISTANCE – V
–25
–60
10k
Figure 6. Output Voltage Swing vs. Load Resistance
–40
–20
0
20
40
60
80 100
JUNCTION TEMPERATURE – 8C
120
140
Figure 9. Input Bias Current vs. Junction Temperature
–6–
REV. B
AD812
70
4
2
60
OUTPUT CURRENT – mA
INPUT OFFSET VOLTAGE – mV
VS = 65V
0
–2
–4
VS = 615V
–6
–8
–10
50
40
30
–12
–14
–16
–60
20
–40
–20
0
20
40
60
80 100
JUNCTION TEMPERATURE – 8C
120
140
Figure 10. Input Offset Voltage vs. Junction Temperature
10
15
SUPPLY VOLTAGE – 6Volts
20
1k
CLOSED-LOOP OUTPUT RESISTANCE – V
SHORT CIRCUIT CURRENT – mA
5
Figure 13. Linear Output Current vs. Supply Voltage
160
140
120
100
SOURCE
80
60
–40
–20
0
20
40
60
80 100
JUNCTION TEMPERATURE – 8C
G = +2
100
VS = 615V
SINK
40
–60
0
120
10
1
65VS
0.1
615VS
0.01
10k
140
Figure 11. Short Circuit Current vs. Junction Temperature
100k
1M
FREQUENCY – Hz
10M
100M
Figure 14. Closed-Loop Output Resistance vs. Frequency
80
30
70
25
OUTPUT VOLTAGE – V p-p
OUTPUT CURRENT – mA
VS = 615V
60
50
VS = 65V
40
VS = 615V
–40
–20
0
20
40
60
80
100
120
10
0
100k
140
JUNCTION TEMPERATURE – 8C
Figure 12. Linear Output Current vs. Junction Temperature
REV. B
RL = 1kV
15
VS = 65V
5
30
20
–60
20
1M
10M
FREQUENCY – Hz
100M
Figure 15. Large Signal Frequency Response
–7–
AD812
100
0
120
VOLTAGE NOISE
NONINVERTING INPUT
CURRENT NOISE
100
1k
FREQUENCY – Hz
TRANSIMPEDANCE – dB
10
10
CURRENT NOISE – pA/ Hz
VOLTAGE NOISE – nV/ Hz
–90
INVERTING INPUT
CURRENT NOISE
1
10
1
100k
10k
–135
100
GAIN
80
100k
1M
FREQUENCY – Hz
10M
100M
Figure 19. Open-Loop Transimpedance vs. Frequency
(Relative to 1 Ω)
–30
681V
VIN
VOUT
681V
70
G = +2
VS = 2V p-p
VS = 615V ; RL = 1kV
681V
HARMONIC DISTORTION – dBc
80
COMMON-MODE REJECTION – dB
VS = 615V
VS = 3V
40
10k
90
681V
60
50
VS = 615V
40
VS = 3V
30
–50
VS = 65V ; RL = 150V
–70
VS = 65V
VS = 615V
2ND HARMONIC
–90
3RD HARMONIC
–110
20
2ND
100k
1M
FREQUENCY – Hz
10M
3RD
–130
100M
1k
Figure 17. Common-Mode Rejection vs. Frequency
10k
100k
1M
FREQUENCY – Hz
10M
100M
Figure 20. Harmonic Distortion vs. Frequency
80
10
70
8
615V
OUTPUT SWING FROM 6V TO 0
POWER SUPPLY REJECTION – dB
–180
VS = 3V
60
Figure 16. Input Current and Voltage Noise vs. Frequency
10
10k
–45
VS = 615V
PHASE
PHASE – Degrees
100
60
50
61.5V
40
30
20
GAIN = –1
VS = 615V
6
4
2
0
1%
0.1%
0.025%
–2
–4
–6
10
–8
0
10k
100k
1M
FREQUENCY – Hz
10M
–10
20
100M
Figure 18. Power Supply Rejection vs. Frequency
30
40
SETTLING TIME – ns
50
60
Figure 21. Output Swing and Error vs. Settling Time
–8–
REV. B
AD812
1400
1400
G = +1
VS = 615V
RL = 500V
G = +1
1200
1000
1000
SLEW RATE – V/ms
SLEW RATE – V/ms
1200
G = +2
800
G = +10
600
400
G = +2
800
G = +10
600
400
G = –1
G = –1
200
200
0
0
0
1
2
3
4
5
6
7
8
10
9
0
3.0
1.5
Figure 22. Slew Rate vs. Output Step Size
9.0
10.5
12.0
13.5 15.0
20ns
100
90
VIN
90
VIN
10
VOUT
10
VOUT
0%
0%
500mV
2V
Figure 26. Small Signal Pulse Response, Gain = +1,
(RF = 750 Ω, RL = 150 Ω, VS = ± 5 V)
Figure 23. Large Signal Pulse Response, Gain = +1,
(RF = 750 Ω, RL = 150 Ω, VS = ± 5 V)
65V
1
GAIN
–90
–180
5V
–270
0
3V
–1
160
VS = 615V
–2
65V
–3
G = +1
RL = 150V
180
–3dB BANDWIDTH – MHz
VS = 615V
0
200
PHASE SHIFT – Degrees
G = +1
RL = 150V
PHASE
CLOSED-LOOP GAIN – dB
7.5
500mV
100
5V
RF = 750V
140
RF = 866V
120
PEAKING
1dB
100
PEAKING
80
0.2dB
60
40
–4
3V
20
–5
0
–6
1
10
100
FREQUENCY – MHz
0
1000
2
4
6
8
10
12
14
16
18
20
SUPPLY VOLTAGE – 6Volts
Figure 24. Closed-Loop Gain and Phase vs. Frequency,
G = +1
REV. B
6.0
Figure 25. Maximum Slew Rate vs. Supply Voltage
50ns
2V
4.5
SUPPLY VOLTAGE – 6Volts
OUTPUT STEP SIZE – Vp-p
Figure 27. –3 dB Bandwidth vs. Supply Voltage, G = +1
–9–
AD812
100
100
90
VIN
90
VIN
10
VOUT
10
VOUT
0%
0%
500mV
5V
Figure 28. Large Signal Pulse Response, Gain = +10,
(RF = 357 Ω, RL = 500 Ω, VS = ± 15 V)
–90
65V
5V
–180
1
3V
GAIN
–270
0
VS = 615V
–1
PHASE
5V
–2
3V
–3
–4
65V
–5
–6
1
10
100
FREQUENCY – MHz
1000
Figure 29. Closed-Loop Gain and Phase vs. Frequency,
Gain = +10, RL = 150 Ω
100
G = +10
RL = 1kV
VS = 615V
0
65V
3V
–90
5V
1
–180
GAIN
0
–270
–1
–360
5V
–2
PHASE SHIFT – Degrees
0
CLOSED-LOOP GAIN (NORMALIZED) – dB
G = +10
RL = 150V
VS = 615V
Figure 31. Small Signal Pulse Response, Gain = +10,
(RF = 357 Ω, RL = 150 Ω, VS = ± 5 V)
PHASE SHIFT – Degrees
CLOSED-LOOP GAIN (NORMALIZED) – dB
PHASE
VS = 615V
3V
–3
65V
–4
–5
–6
1
10
100
FREQUENCY – MHz
1000
Figure 32. Closed-Loop Gain and Phase vs. Frequency,
Gain = +10, RL = 1 k Ω
110
G = +10
RL= 150V
90
G = +10
RL = 1kV
100
80
90
–3dB BANDWIDTH – MHz
–3dB BANDWIDTH – MHz
20ns
50mV
50ns
500mV
70
PEAKING
1dB
RF = 357V
60
RF = 154V
50
RF = 649V
40
30
20
RF = 357V
80
70
RF = 154V
60
RF = 649V
50
40
30
10
20
0
0
2
4
6
8
10
12
14
SUPPLY VOLTAGE – 6Volts
16
18
10
20
0
Figure 30. –3 dB Bandwidth vs. Supply Voltage,
Gain = +10, RL = 150 Ω
2
4
6
8
10
12
14
SUPPLY VOLTAGE – 6Volts
16
18
20
Figure 33. –3 dB Bandwidth vs. Supply Voltage,
Gain = +10, RL = 1 k Ω
–10–
REV. B
AD812
50ns
2V
500mV
100
90
VIN
90
VIN
10
VOUT
10
VOUT
0%
0%
500mV
2V
0
–90
3V
1
–180
GAIN
0
–270
–1
VS = 615V
65V
–4
5V
–5
3V
–6
1
10
100
FREQUENCY – MHz
1000
Figure 35. Closed-Loop Gain and Phase vs. Frequency,
Gain = –1, RL = 150 Ω
–90
–180
5V
1
GAIN
3V
–270
0
–1
VS = 615V
–2
65V
–3
5V
–4
3V
–5
–6
1
10
100
FREQUENCY – MHz
1000
100
G = –1
RL = 150V
120
G = –10
RL = 1kV
90
RF = 681V
110
80
PEAKING # 1.0dB
100
–3dB BANDWIDTH – MHz
–3dB BANDWIDTH – MHz
0
Figure 38. Closed-Loop Gain and Phase vs. Frequency,
Gain = –10, RL = 1 kΩ
130
RF = 715V
90
80
PEAKING # 0.2dB
70
60
50
RF = 357V
70
60
RF = 154V
RF = 649V
50
40
30
20
40
10
0
2
4
6
8
10
12
14
16
18
0
20
0
SUPPLY VOLTAGE – 6Volts
Figure 36. –3 dB Bandwidth vs. Supply Voltage,
Gain = –1, RL = 150 Ω
REV. B
G = –10
RL = 1kV
65V
–2
–3
VS = 615V
PHASE
PHASE SHIFT – Degrees
CLOSED-LOOP GAIN (NORMALIZED) – dB
G = –1
RL = 150V
65V
5V
CLOSED-LOOP GAIN (NORMALIZED) – dB
VS = 615V
PHASE
Figure 37. Small Signal Pulse Response, Gain = –1,
(RF = 750 Ω, RL = 150 Ω, VS = ± 5 V)
PHASE SHIFT – Degrees
Figure 34. Large Signal Pulse Response, Gain = –1,
(RF = 750 Ω, RL = 150 Ω, VS = ± 5 V)
30
20ns
100
2
4
6
8
10
12
14
SUPPLY VOLTAGE – 6Volts
16
18
20
Figure 39. –3 dB Bandwidth vs. Supply Voltage,
Gain = –10, RL = 1 kΩ
–11–
AD812
General Considerations
To estimate the –3 dB bandwidth for closed-loop gains or feedback resistors not listed in the above table, the following two
pole model for the AD812 many be used:
The AD812 is a wide bandwidth, dual video amplifier which
offers a high level of performance on less than 5.5 mA per amplifier of quiescent supply current. It is designed to offer outstanding performance at closed-loop inverting or noninverting
gains of one or greater.
ACL =
Built on a low cost, complementary bipolar process, and achieving bandwidth in excess of 100 MHz, differential gain and phase
errors of better than 0.1% and 0.1° (into 150 Ω), and output
current greater than 40 mA, the AD812 is an exceptionally
efficient video amplifier. Using a conventional current feedback
architecture, its high performance is achieved through careful
attention to design details.
where:
Choice of Feedback and Gain Resistors
Because it is a current feedback amplifier, the closed-loop bandwidth of the AD812 depends on the value of the feedback resistor. The bandwidth also depends on the supply voltage. In
addition, attenuation of the open-loop response when driving
load resistors less than about 250 Ω will affect the bandwidth.
Table I contains data showing typical bandwidths at different
supply voltages for some useful closed-loop gains when driving a
load of 150 Ω. (Bandwidths will be about 20% greater for load
resistances above a few hundred ohms.)
(
)
Table II. Two-Pole Model Parameters at Various
Supply Voltages
VS
rIN (⍀)
CT (pF)
f2 (MHz)
± 15
±5
+5
+3
85
90
105
115
2.5
3.8
4.8
5.5
150
125
105
95
VS (V)
Gain
RF (⍀)
BW (MHz)
± 15
+1
+2
+10
–1
–10
866
715
357
715
357
145
100
65
100
60
where:
+1
+2
+10
–1
–10
750
681
154
715
154
90
65
45
70
45
and:
+1
+2
+10
–1
–10
750
681
154
715
154
60
50
35
50
35
+1
+2
+10
–1
–10
750
681
154
715
154
50
40
30
40
25
+3
)
As discussed in many amplifier and electronics textbooks (such
as Roberge’s Operational Amplifiers: Theory and Practice), the
–3 dB bandwidth for the 2-pole model can be obtained as:
Table I. –3 dB Bandwidth vs. Closed-Loop Gain and
Feedback Resistor (RL = 150 Ω)
+5
(
 RF + GrIN CT 
 + S RF + GrIN CT + 1
S2 
2πf 2


ACL = closed-loop gain
G = 1 + RF /RG
rIN = input resistance of the inverting input
CT = “transcapacitance,” which forms the open-loop
dominant pole with the tranresistance
RF = feedback resistor
RG = gain resistor
f2 = frequency of second (nondominant) pole
S = 2 πj f
Appropriate values for the model parameters at different supply
voltages are listed in Table II. Reasonable approximations for
these values at supply voltages not found in the table can be
obtained by a simple linear interpolation between those tabulated values which “bracket” the desired condition.
The choice of feedback resistor is not critical unless it is important to maintain the widest, flattest frequency response. The
resistors recommended in the table are those (metal film values)
that will result in the widest 0.1 dB bandwidth. In those applications where the best control of the bandwidth is desired, 1%
metal film resistors are adequate. Wider bandwidths can be
attained by reducing the magnitude of the feedback resistor (at
the expense of increased peaking), while peaking can be reduced
by increasing the magnitude of the feedback resistor.
±5
G
f3 = fN [1 – 2d2 + (2 – 4d2 + 4d4 )1/2]1/2
1/ 2


f2

fN = 
 R + Gr

C
IN
T 
 F
(
)
d = (1/2) [f2 (RF + GrIN ) CT]1/2
This model will predict –3 dB bandwidth within about 10 to
15% of the correct value when the load is 150 Ω. However, it is
not an accurate enough to predict either the phase behavior or
the frequency response peaking of the AD812.
Printed Circuit Board Layout Guidelines
As with all wideband amplifiers, printed circuit board parasitics
can affect the overall closed-loop performance. Most important
for controlling the 0.1 dB bandwidth are stray capacitances at
the output and inverting input nodes. Increasing the space between
signal lines and ground plane will minimize the coupling. Also,
signal lines connecting the feedback and gain resistors should be
kept short enough that their associated inductance does not
cause high frequency gain errors.
–12–
REV. B
AD812
The input and output signal return paths must also be kept from
overlapping. Since ground connections are not of perfectly zero
impedance, current in one ground return path can produce a
voltage drop in another ground return path if they are allowed
to overlap.
Power Supply Bypassing
Adequate power supply bypassing can be very important when
optimizing the performance of high speed circuits. Inductance
in the supply leads can (for example) contribute to resonant
circuits that produce peaking in the amplifier’s response. In
addition, if large current transients must be delivered to a load,
then large (greater than 1 µF) bypass capacitors are required to
produce the best settling time and lowest distortion. Although
0.1 µF capacitors may be adequate in some applications, more
elaborate bypassing is required in other cases.
Electric field coupling external to (and across) the package can
be reduced by arranging for a narrow strip of ground plane to be
run between the pins (parallel to the pin rows). Doing this on
both sides of the board can reduce the high frequency crosstalk
by about 5 dB or 6 dB.
When multiple bypass capacitors are connected in parallel, it is
important to be sure that the capacitors themselves do not form
resonant circuits. A small (say 5 Ω) resistor may be required in
series with one of the capacitors to minimize this possibility.
Driving Capacitive Loads
As discussed below, power supply bypassing can have a significant impact on crosstalk performance.
Achieving Low Crosstalk
Measured crosstalk from the output of amplifier 2 to the input
of amplifier 1 of the AD812 is shown in Figure 40. The crosstalk
from the output of amplifier 1 to the input of amplifier 2 is a few
dB better than this due to the additional distance between critical signal nodes.
When used with the appropriate output series resistor, any load
capacitance can be driven without peaking or oscillation. In
most cases, less than 50 Ω is all that is needed to achieve an
extremely flat frequency response. As illustrated in Figure 44,
the AD812 can be very attractive for driving largely capacitive
loads. In this case, the AD812’s high output short circuit
current allows for a 150 V/µs slew rate when driving a 510 pF
capacitor.
RF
+VS
A carefully laid-out PC board should be able to achieve the level
of crosstalk shown in the figure. The most significant contributors to difficulty in achieving low crosstalk are inadequate power
supply bypassing, overlapped input and/or output signal paths,
and capacitive coupling between critical nodes.
0.1mF
1.0mF
RG
8
RS
AD812
The bypass capacitors must be connected to the ground plane at
a point close to and between the ground reference points for the
two loads. (The bypass of the negative power supply is particularly important in this regard.) There are two amplifiers in the
package, and low impedance signal return paths must be provided for each load. (Using a parallel combination of 1 µF,
0.1 µF, and 0.01 µF bypass capacitors will help to achieve optimal crosstalk.)
VIN
4
VO
1.0mF
CL
RL
RT
0.1mF
–VS
Figure 41. Circuit for Driving a Capacitive Load
–10
VS = 65V
G = +2
RF = 750V
RL = 1kV
CL = 10pF
–20
RL = 150V
–30
12
CLOSED-LOOP GAIN – dB
CROSSTALK – dB
–40
–50
–60
–70
–80
–90
1M
10M
RS = 30V
3
RS = 50V
0
–3
1
100M
FREQUENCY – Hz
10
100
FREQUENCY – MHz
1000
Figure 42. Response to a Small Load Capacitor at ± 5 V
Figure 40. Crosstalk vs. Frequency
REV. B
RS = 0
6
–6
–100
–110
100k
9
–13–
AD812
VS = 615V
G = +2
RF = 750V
RL = 1kV
100
12
CLOSED-LOOP GAIN – dB
50ns
1V
90
VIN
10
VOUT
9
6
CL = 150pF, RS = 30V
3
0
0%
–3
CL = 510pF, RS = 15V
–6
2V
–9
1
10
100
FREQUENCY – MHz
1000
Figure 45. 6 dB Overload Recovery; G = 10, RL = 500 Ω,
VS = ± 5 V
Figure 43. Response to Large Load Capacitor, VS = ± 15 V
5V
In the case of high gains with very high levels of input overdrive,
a longer recovery time may occur. For example, if the input
common-mode voltage range is exceeded in a gain of +10, the
recovery time will be on the order of 100 ns. This is primarily
due to current overloading of the input stage.
100ns
VIN
100
90
As noted in the warning under “Maximum Power Dissipation,”
a high level of input overdrive in a high noninverting gain circuit
can result in a large current flow in the input stage. For differential input voltages of less than about 1.25 V, this will be internally limited to less than 20 mA (decreasing with supply voltage).
For input overdrives which result in higher differential input
voltages, power dissipation in the input stage must be considered. It is recommended that external diode clamps be used in
cases where the differential input voltage is expected to exceed
1.25 V.
VOUT
10
0%
5V
Figure 44. Pulse Response of Circuit of Figure 41 with
CL = 510 pF, RL = 1 kΩ, RF = RG = 715 Ω, RS = 15 Ω
High Performance Video Line Driver
Overload Recovery
There are three important overload conditions to consider.
They are due to input common mode voltage overdrive, input
current overdrive, and output voltage overdrive. When the
amplifier is configured for low closed-loop gains, and its input
common-mode voltage range is exceeded, the recovery time will
be very fast, typically under 10 ns. When configured for a higher
gain, and overloaded at the output, the recovery time will also
be short. For example, in a gain of +10, with 6 dB of input
overdrive, the recovery time of the AD812 is about 10 ns.
At a gain of +2, the AD812 makes an excellent driver for a backterminated 75 Ω video line. Low differential gain and phase
errors and wide 0.1 dB bandwidth can be realized over a wide
range of power supply voltage. Outstanding gain and group
delay matching are also attainable over the full operating supply
voltage range.
RG
RF
+VS
0.1mF
75V
75V CABLE
8
75V
CABLE
VOUT
AD812
VIN
75V
4
75V
0.1mF
–VS
Figure 46. Gain of +2 Video Line Driver (RF = RG from
Table I)
–14–
REV. B
G = +2
RL = 150V
0
–90
3V
VS = 615V
5V
1
65V
GAIN
CLOSED-LOOP GAIN – dB
–180
–270
0
5V
–1
3V
–2
0.4
0.2
VS = 615V
–3
65V
–4
G = +2
RL = 150V
0.3
NORMALIZED GAIN – dB
90
PHASE
PHASE SHIFT – Degrees
AD812
0.1
0
–0.1
–0.2
VS = 615V
–0.3
65V
–0.4
5V
–5
–0.5
3V
–0.6
100k
–6
1
10
100
FREQUENCY –MHz
1000
Figure 47. Closed-Loop Gain and Phase vs. Frequency for
the Line Driver
1.0
RF = 590V
G = +2
RL = 150V
110
RL = 150V
VS = 3V
0.6
RF = 750V
PEAKING # 1dB
G = +2
0.8
RF = 715V
100
RF = 681V
90
0.4
GAIN MATCH – dB
–3dB BANDWIDTH – MHz
100M
Figure 50. Fine-Scale Gain Flatness vs. Frequency,
Gain = +2, RL = 150 Ω
120
80
NO PEAKING
70
60
50
0.2
0
RF = 715V
–0.4
40
–0.6
30
–0.8
0
2
4
6
8
10
12
14
SUPPLY VOLTAGE – 6Volts
16
18
–1.0
20
1
Figure 48. –3 dB Bandwidth vs. Supply Voltage,
Gain = +2, RL = 150 Ω
0.02
0.06
1000
DELAY
8
3V
GROUP DELAY – ns
0.04
DIFFERENTIAL GAIN
10
100
FREQUENCY – MHz
Figure 51. Closed-Loop Gain Matching vs. Frequency,
Gain = +2, RL = 150 Ω
DIFFERENTIAL GAIN – %
0.06
0.08
VS = 615V
–0.2
20
DIFFERENTIAL PHASE – Degrees
1M
10M
FREQUENCY – Hz
DIFFERENTIAL PHASE
0.04
6
5V
4
65V
615V
2
0
DELAY MATCHING
0.4
0.2
VS = 3V TO 615V
0
0.02
–0.2
–0.4
100k
0
5
6
7
8
9
10
11
12
13
14
15
SUPPLY VOLTAGE – 6Volts
10M
FREQUENCY – Hz
100M
Figure 52. Group Delay and Group Delay Matching vs.
Frequency, G = +2, RL = 150 Ω
Figure 49. Differential Gain and Phase vs. Supply Voltage,
Gain = +2, RL = 150 Ω
REV. B
1M
–15–
90
The AD812 will operate with total supply voltages from 36 V
down to 2.4 V. With proper biasing (see Figure 53), it can be an
outstanding single supply video amplifier. Since the input and
output voltage ranges extend to within 1 volt of the supply rails,
it will handle a 1.3 V p-p signal on a single 3.3 V supply, or a
3 V p-p signal on a single 5 V supply. The small signal, 0.1 dB
bandwidths will exceed 10 MHz in either case, and the large
signal bandwidths will exceed 6 MHz.
PHASE
CLOSED-LOOP GAIN – dB
0
–180
–0.5
–270
–1.0
–1.5
–2.0
–2.5
–3.0
–3.5
1
10
100
FREQUENCY – MHz
1000
Figure 54. Closed-Loop Gain and Phase vs. Frequency,
Circuit of Figure 53
649V
C3
30mF
–90
GAIN
The capacitively coupled cable driver in Figure 53 will achieve
outstanding differential gain and phase errors of 0.07% and 0.06
degrees respectively on a single 5 V supply. Resistor R2, in this
circuit, is selected to optimize the differential gain and phase by
operating the amplifier in its most linear region. To optimize the
circuit for a 3 V supply, a value of 8 kΩ is recommended for R2.
649V
0
VS = 5V
0.5
C1859b–0–9/98
Operation Using a Single Supply
PHASE SHIFT – Degrees
AD812
R3
1kV
+VS
C2
1mF
1V
R1
9kV
COUT
8
C1
2mF
47mF
75V
75V
CABLE
100
VIN
90
VOUT
AD812
VIN
50ns
75V
4
R2
11.8kV
VOUT
Figure 53. Biasing for Single Supply Operation
10
0%
500mV
Figure 55. Pulse Response of the Circuit of Figure 53 with
VS = 5 V
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
8-Lead Plastic SOIC
(SO-8)
0.39 (9.91)
8
0.1968 (5.00)
0.1890 (4.80)
5
0.25
(6.35)
1
4
PIN 1
0.165 60.01
(4.19 60.25)
0.125 (3.18)
MIN
0.060 (1.52)
0.015 (0.38)
SEATING
0.018 60.003 0.10 0.033 (0.84)
PLANE
(0.46 +0.08) (2.54)
NOM
BSC
0.1574 (4.00)
0.1497 (3.80)
0.325 (8.25)
0.300 (7.62)
0.195 (4.95)
0.115 (2.93)
PIN 1
0.0098 (0.25)
0.0040 (0.10)
0.015 (0.381)
0.008 (0.204)
8
5
1
4
0.2440 (6.20)
0.2284 (5.80)
0.0688 (1.75)
0.0532 (1.35)
0.0500 0.0192 (0.49)
SEATING (1.27)
0.0098 (0.25)
PLANE BSC 0.0138 (0.35) 0.0075 (0.19)
–16–
0.0196 (0.50)
3 458
0.0099 (0.25)
88
08 0.0500 (1.27)
0.0160 (0.41)
REV. B
PRINTED IN U.S.A.
8-Lead Plastic DIP
(N-8)