AD EVAL-AD7707EB 3 v/5 v, -10 v input range, 1 mw 3-channel 16-bit, sigma-delta adc Datasheet

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3 V/5 V, ⴞ10 V Input Range, 1 mW
3-Channel 16-Bit, Sigma-Delta ADC
AD7707
FEATURES
Charge Balancing ADC
16 Bits No Missing Codes
0.003% Nonlinearity
High Level (ⴞ10 V) and Low Level (ⴞ10 mV) Input
Channels
True Bipolar ⴞ100 mV Capability on Low Level Input
Channels Without Requiring Charge Pumps
Programmable Gain Front End
Gains from 1 to 128
Three-Wire Serial Interface
SPI™, QSPI™, MICROWIRE™ and DSP Compatible
Schmitt Trigger Input on SCLK
Ability to Buffer the Analog Input
2.7 V to 3.3 V or 4.75 V to 5.25 V Operation
Power Dissipation 1 mW max @ 3␣ V
Standby Current 8 ␮A max
20-Lead SOIC and TSSOP Packages
GENERAL DESCRIPTION
The AD7707 is a complete analog front end for low frequency
measurement applications. This three-channel device can accept
either low level input signals directly from a transducer or high
level (± 10 V) signals and produce a serial digital output. It
employs a sigma-delta conversion technique to realize up to
16 bits of no missing codes performance. The selected input
signal is applied to a proprietary programmable gain front end
based around an analog modulator. The modulator output is
processed by an on-chip digital filter. The first notch of this
digital filter can be programmed via an on-chip control register
allowing adjustment of the filter cutoff and output update rate.
The AD7707 operates from a single 2.7 V to 3.3 V or 4.75 V to
5.25 V supply. The AD7707 features two low level pseudodifferential analog input channels, one high level input channel
and a differential reference input. Input signal ranges of 0 mV to
+20 mV through 0 V to +2.5 V can be accommodated on both
low level input channels when operating with a VDD of 5 V and a
reference of 2.5 V. They can also handle bipolar input signal
ranges of ± 20 mV through ± 2.5 V, which are referenced to the
LCOM input. The AD7707, with a 3 V supply and a 1.225 V
reference, can handle unipolar input signal ranges of 0 mV to
+10 mV through 0 V to +1.225 V. Its bipolar input signal ranges
are ± 10 mV through ± 1.225 V.
The high level input channel can accept input signal ranges of
± 10 V, ± 5 V, 0 V to +10 V and 0 V to +5 V. The AD7707 thus
performs all signal conditioning and conversion for a threechannel system.
SPI and QSPI are trademarks of Motorola, Inc.
MICROWIRE is a trademark of National Semiconductor Corporation.
FUNCTIONAL BLOCK DIAGRAM
DVDD
AVDD
REF IN(–)
REF IN(+)
AD7707
CHARGE
BALANCING
A/D CONVERTER
AIN1
AIN2
MUX
LOCOM
BUF
30kV
PGA
AIN3
VBIAS
5kV
A = 1<128
S–D
MODULATOR
DIGITAL FILTER
5kV
SERIAL INTERFACE
15kV
HICOM
REGISTER BANK
SCLK
CS
30kV
DIN
MCLK IN
MCLK OUT
CLOCK
GENERATION
AGND
DGND
DOUT
DRDY
RESET
The AD7707 is ideal for use in smart, microcontroller or DSPbased systems. It features a serial interface that can be configured for three-wire operation. Gain settings, signal polarity and
update rate selection can be configured in software using the
input serial port. The part contains self-calibration and system
calibration options to eliminate gain and offset errors on the
part itself or in the system.
CMOS construction ensures very low power dissipation, and the
power-down mode reduces the standby power consumption to
20␣ µW typ. These parts are available in a 20-lead wide body
(0.3 inch) small outline (SOIC) package and a low profile 20-lead
TSSOP.
PRODUCT HIGHLIGHTS
1. The AD7707 consumes less than 1 mW at 3 V supplies and
1␣ MHz master clock, making it ideal for use in low power
systems. Standby current is less than 8␣ µA.
2. On-chip thin-film resistors allow ± 10 V, ± 5 V, 0 V to +10 V
and 0 V to +5 V high level input signals to be directly accommodated on the analog inputs without requiring split supplies
or charge-pumps.
3. The low level input channels allow the AD7707 to accept
input signals directly from a strain gage or transducer removing a considerable amount of signal conditioning.
4. The part features excellent static performance specifications
with 16 bits, no missing codes, ± 0.003% accuracy and low
rms noise. Endpoint errors and the effects of temperature
drift are eliminated by on-chip calibration options, which
remove zero-scale and full-scale errors.
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 2000
(AV = DV = +3 V or 5 V, REF IN(+) = +1.225␣ V with AV = 3 V and +2.5 V with AV
= 5 V; REF␣ IN(–) = GND; VBIAS = REFIN(+); MCLK IN = 2.4576␣ MHz unless otherwise
AD7707–SPECIFICATIONS
noted. All specifications T to T unless otherwise noted.)
DD
MIN
DD
DD
DD
MAX
Parameter
STATIC PERFORMANCE
Low Level Input Channels (AIN1 and AIN2)
No Missing Codes
Output Noise
Integral Nonlinearity2
Unipolar Offset Error
Unipolar Offset Drift4
Bipolar Zero Error
Bipolar Zero Drift4
Positive Full-Scale Error5
Full-Scale Drift4, 6
Gain Error7
Gain Drift4, 8
Bipolar Negative Full-Scale Error2
Bipolar Negative Full-Scale Drift4
HIGH LEVEL INPUT CHANNEL (AIN3)
No Missing Codes
Output Noise
Integral Nonlinearity2
Unipolar Offset Error9
Unipolar Offset Drift
Bipolar Zero Error9
Bipolar Zero Drift
Gain Error
Gain Drift
Negative Full-Scale Error2
B Version1
Units
Conditions/Comments
16
See Tables I and III
± 0.003
See Note 3
0.5
See Note 3
0.5
0.1
See Note 3
0.5
See Note 3
0.5
± 0.003
1
0.6
Bits min
Guaranteed by Design. Filter Notch < 60 Hz
Depends on Filter Cutoffs and Selected Gain
Filter Notch < 60␣ Hz. Typically ± 0.0003%
16
See Tables IV and VI
± 0.003
± 10
4
± 10
4
1
± 0.2
0.5
± 0.0012
Bits min
LOW LEVEL ANALOG INPUTS/REFERENCE INPUTS
Input Common-Mode Rejection (CMR)2
AVDD = 5 V
Gain = 1
100
Gain = 2
105
Gain = 4
110
Gain = 8 to 128
130
AVDD = 3 V
Gain = 1
105
Gain = 2
110
Gain = 4
120
Gain = 8 to 128
130
98
Normal-Mode 50 Hz Rejection2
Normal-Mode 60 Hz Rejection2
98
Common-Mode 50 Hz Rejection2
150
Common-Mode 60 Hz Rejection2
150
Absolute/Common-Mode REF IN Voltage2
AGND to AVDD
AGND – 100 mV
Absolute/Common-Mode AIN Voltage2, 10
AVDD + 30␣ mV
AGND + 50␣ mV
AVDD – 1.5␣ V
AIN DC Input Current2
1
AIN Sampling Capacitance2
10
AIN Differential Voltage Range11
0 to +VREF/GAIN12
± VREF/GAIN
AIN Input Sampling Rate, fS
GAIN × fCLKIN/64
fCLKIN/8
Reference Input Range
REF IN(+) – REF IN(–) Voltage
1/1.75
REF IN(+) – REF IN(–) Voltage
1/3.5
REF IN Input Sampling Rate, fS
fCLKIN/64
% of FSR max
µV/°C typ
µV/°C typ
µV/°C typ
µV/°C typ
ppm of FSR/°C typ
% of FSR max
µV/°C typ
µV/°C typ
% of FSR max
mV max
µV/°C typ
mV max
µV/°C typ
µV/°C typ
% typ
ppm of FSR/°C typ
% of FSR typ
± 0.003
80
90
Typically ± 0.0007%
For Gains of 1 to 4
For Gains of 8 to 128
Guaranteed by Design. Filter Notch < 60␣ Hz
Depends on Filter Cutoffs and Selected Gain
Filter Notch < 60␣ Hz. Typically ± 0.0003%
Typically Within ± 1.5 mV
Typically Within ± 1.5 mV
For Gains 1, 2 and 4
For Gains 8, 16, 32, 64 and 128
Typically Within ± 0.05%
Specifications for AIN and REF IN Unless Noted
Low Level Input Channels, AIN1 and AIN2
dB typ
dB typ
dB typ
dB typ
dB typ
dB typ
dB typ
dB typ
dB typ
dB typ
dB typ
dB typ
V min to V max
V min
V max
V min
V max
nA max
pF max
nom
nom
V min/max
V min/max
± 100 mV INPUT RANGE
INL2
Input Common-Mode Rejection (CMR)2
Power Supply Rejection (PSR)2
For Gains 1, 2 and 4
For Gains 8, 16, 32, 64 and 128
% of FSR max
dB typ
dB typ
–2–
For Filter Notches of 10 Hz, 25 Hz, 50 Hz, ± 0.02 × fNOTCH
For Filter Notches of 10 Hz, 20 Hz, 60 Hz, ± 0.02 × fNOTCH
For Filter Notches of 10 Hz, 25 Hz, 50 Hz, ± 0.02 × fNOTCH
For Filter Notches of 10 Hz, 20 Hz, 60 Hz, ± 0.02 × fNOTCH
BUF Bit of Setup Register = 0
BUF Bit of Setup Register = 1
BUF = 0
Unipolar Input Range (B/U Bit of Setup Register = 1)
Bipolar Input Range (B/U Bit of Setup Register = 0)
For Gains of 1 to 4
For Gains of 8 to 128
AVDD = 2.7 V to 3.3 V. VREF = 1.225 V ± 1% for
Specified Performance
AVDD = 4.75 V to 5.25 V. VREF = 2.5 V ± 1% for
Specified Performance
Low Level Input Channels, (AIN1 and AIN2)
Gain = 16, Unbuffered Mode
Filter Notch < 60 Hz
REV. A
AD7707
Parameter
1
B Version
HIGH LEVEL ANALOG INPUT CHANNEL (AIN3)
AIN3 Voltage Range
+10
–10
Normal Mode 50 Hz Rejection
78
Normal Mode 60 Hz Rejection
78
AIN3 Input Sampling Rate, fS
GAIN × fCLKIN/64
fCLKIN/8
AIN3 Input Impedance2
27
10
AIN3 Sampling Capacitance2
VBIAS Input Range
0 V/AVDD
LOGIC INPUTS
Input Current
All Inputs Except MCLK IN
MCLK
All Inputs Except SCLK and MCLK IN
VINL, Input Low Voltage
VINH, Input High Voltage
SCLK Only (Schmitt Triggered Input)
VT+
VT–
VT+ – VT–
SCLK Only (Schmitt Triggered Input)
VT+
VT–
VT+ – VT–
MCLK IN Only
VINL, Input Low Voltage
VINH, Input High Voltage
MCLK IN Only
VINL, Input Low Voltage
VINH, Input High Voltage
LOGIC OUTPUTS (Including MCLK OUT)
VOL, Output Low Voltage
VOH, Output High Voltage
Floating State Leakage Current
Floating State Output Capacitance14
Data Output Coding
SYSTEM CALIBRATION
Low Level Input Channels (AIN1 and AIN2)
Positive Full-Scale Calibration Limit15
Negative Full-Scale Calibration Limit15
Offset Calibration Limit16
Input Span16
High Level Input Channels (AIN3)
Positive Full-Scale Calibration Limit15
Negative Full-Scale Calibration Limit15
Offset Calibration Limit16
Input Span16
POWER REQUIREMENTS
Power Supply Voltages
AVDD Voltage
DVDD Voltage
Power Supply Currents
AVDD Current
REV. A
Units
Conditions/Comments
AIN3 is with respect to HICOM.
V max
V min
dB typ
dB typ
kΩ min
pF max
V min/max
For Filter Notches of 10 Hz, 25 Hz, 50 Hz, ± 0.02 × fNOTCH
For Filter Notches of 10 Hz, 20 Hz, 60 Hz, ± 0.02 × fNOTCH
For Gains of 1 to 4
For Gains of 8 to 128
Typically 30 kΩ ± 10%; Typical Resistor Tempco is –30 ppm/°C
Typically = REFIN(+) = 2.5 V
±1
± 10
µA max
µA max
Typically ± 20 nA
Typically ± 2 µA
0.8
0.4
2.0
V max
V max
V min
DVDD = 5 V
DVDD = 3 V
DVDD = 3 V and 5 V
DVDD = 5 V Nominal
1.4/3
0.8/1.4
0.4/0.8
V min/V max
V min/V max
V min/V max
1/2.5
0.4/1.1
0.375/0.8
V min/V max
V min/V max
V min/V max
0.8
3.5
V max
V min
0.4
2.5
V max
V min
0.4
0.4
4
DVDD – 0.6
± 10
9
Binary
Offset Binary
V max
V max
V min
V min
µA max
pF typ
(1.05 × VREF)/GAIN
– (1.05 × VREF)/GAIN
– (1.05 × VREF)/GAIN
(0.8 × VREF)/GAIN
(2.1 × VREF)/GAIN
V max
V max
V max
V min
V max
GAIN Is The Selected PGA Gain (1 to 128)
GAIN Is The Selected PGA Gain (1 to 128)
GAIN Is The Selected PGA Gain (1 to 128)
GAIN Is The Selected PGA Gain (1 to 128)
GAIN Is The Selected PGA Gain (1 to 128)
(8.4 × VREF)/GAIN
– (8.4 × VREF)/GAIN
– (8.4 × VREF)/GAIN
(6.4 × VREF)/GAIN
(16.8 × VREF)/GAIN
V max
V max
V max
V min
V max
GAIN Is The Selected PGA Gain (1 to 128)
GAIN Is The Selected PGA Gain (1 to 128)
GAIN Is The Selected PGA Gain (1 to 128)
GAIN Is The Selected PGA Gain (1 to 128)
GAIN Is The Selected PGA Gain (1 to 128)
+2.7 to +3.3 or
+4.75 to +5.25
+2.7 to +5.25
V
V
For Specified Performance
For Specified Performance
0.27
mA max
0.6
mA max
0.5
1.1
mA max
mA max
DVDD = 3 V Nominal
DVDD = 5 V Nominal
DVDD = 3 V Nominal
ISINK = 800␣ µA Except for MCLK OUT.13 DVDD = 5 V
ISINK = 100␣ µA Except for MCLK OUT.13 DVDD = 3 V
ISOURCE = 200 µA Except for MCLK OUT.13 DVDD = 5 V
ISOURCE = 100␣ µA Except for MCLK OUT.13 DVDD = 3 V
Unipolar Mode
Bipolar Mode
–3–
AVDD = 3␣ V or 5␣ V. Gain = 1 to 4
Typically 0.22 mA. BUF = 0. fCLK IN = 1 MHz
or 2.4576␣ MHz
Typically 0.45 mA. BUF = 1. fCLK IN = 1 MHz
or 2.4576 MHz
AVDD = 3 V or 5␣ V. Gain = 8 to 128
Typically 0.38␣ mA. BUF = 0. fCLK IN = 2.4576␣ MHz
Typically 0.81␣ mA. BUF = 1. fCLK IN = 2.4576␣ MHz
AD7707–SPECIFICATIONS
Parameter
B Version1
Units
0.080
0.15
0.18
0.35
See Note 20
mA max
mA max
mA max
mA max
dB typ
Conditions/Comments
POWER REQUIREMENTS (Continued)
DVDD Current17
Power Supply Rejection 19
Normal Mode Power Dissipation17
1.05
2.04
1.35
mW max
mW max
mW max
2.34
mW max
2.1
3.75
3.1
4.75
18
mW max
mW max
mW max
mW max
µA max
8
µA max
Digital I/Ps = 0␣ V or DVDD. External MCLK IN
Typically 0.06␣ mA. DVDD = 3␣ V. fCLK IN = 1␣ MHz
Typically 0.13 mA. DVDD = 5␣ V. fCLK IN = 1␣ MHz
Typically 0.15␣ mA. DVDD = 3␣ V. fCLK IN = 2.4576␣ MHz
Typically 0.3␣ mA. DVDD = 5␣ V. fCLK IN = 2.4576␣ MHz
AVDD = DVDD = +3 V. Digital I/Ps = 0 V or DVDD.
External MCLK IN Excluding Dissipation in the AIN3
Attenuator
Typically 0.84 mW. BUF = 0. fCLK IN = 1␣ MHz, All Gains.
Typically 1.53 mW. BUF = 1. fCLK IN = 1␣ MHz, All Gains.
Typically 1.11 mW. BUF = 0. fCLK IN = 2.4576 MHz,
Gain = 1 to 4.
Typically 1.9 mW. BUF = 1. fCLK IN = 2.4576 MHz,
Gain = 1 to 4.
AVDD = DVDD = +5 V. Digital I/Ps = 0␣ V or DVDD.
External MCLKIN
Typically 1.75 mW. BUF = 0. fCLK IN = 1␣ MHz, All Gains.
Typically 2.9 mW. BUF = 1. fCLK IN = 1␣ MHz, All Gains.
Typically 2.6 mW. BUF = 0. fCLK IN = 2.4576 MHz.
Typically 3.75 mW. BUF = 1. fCLK IN = 2.4576 MHz.
External MCLK IN = 0 V or DVDD. Typically 9␣ µA.
AVDD = +5 V
External MCLK IN = 0 V or DVDD. Typically 4␣ µA.
AVDD = +3 V␣
Normal Mode Power Dissipation17
Standby (Power-Down) Current18
NOTES
1
Temperature range as follows: B Version, –40°C to +85°C.
These numbers are established from characterization or design at initial product release.
3
A calibration is effectively a conversion so these errors will be of the order of the conversion noise shown in Tables I and III for the low level input channels AIN1
and AIN2. This applies after calibration at the temperature of interest.
4
Recalibration at any temperature will remove these drift errors.
5
Positive full-scale error includes zero-scale errors (unipolar offset error or bipolar zero error) and applies to both unipolar and bipolar input ranges.
6
Full-scale drift includes zero-scale drift (unipolar offset drift or bipolar zero drift) and applies to both unipolar and bipolar input ranges.
7
Gain error does not include zero-scale errors. It is calculated as full-scale error–unipolar offset error for unipolar ranges and full-scale error–bipolar zero error for
bipolar ranges.
8
Gain error drift does not include unipolar offset drift/bipolar zero drift. It is effectively the drift of the part if zero scale calibrations only were performed.
9
Error is removed following a system calibration.
10
This common-mode voltage range is allowed provided that the input voltage on analog inputs does not go more positive than AVDD + 30 mV or go more negative
than AGND – 100␣ mV. Parts are functional with voltages down to AGND – 200 mV, but with increased leakage at high temperature.
11
The analog input voltage range on AIN(+) is given here with respect to the voltage on LCOM on the low level input channels (AIN1 and AIN2) and is given with
respect to the HCOM input on the high level input channel AIN3. The absolute voltage on the low level analog inputs should not go more positive than AVDD +
100␣ mV, or go more negative than GND␣ – 100␣ mV for specified performance. Input voltages of AGND – 200 mV can be accommodated, but with increased leakage
at high temperature.
12
VREF = REF IN(+) – REF IN(–).
13
These logic output levels apply to the MCLK OUT only when it is loaded with one CMOS load.
14
Sample tested at +25°C to ensure compliance.
15
After calibration, if the analog input exceeds positive full scale, the converter will output all 1s. If the analog input is less than negative full scale, the device will
output all 0s.
16
These calibration and span limits apply provided the absolute voltage on the analog inputs does not exceed AVDD + 30␣ mV or go more negative than AGND –
30␣ mV. The offset calibration limit applies to both the unipolar zero point and the bipolar zero point.
17
When using a crystal or ceramic resonator across the MCLK pins as the clock source for the device, the DVDD current and power dissipation will vary depending on
the crystal or resonator type (see Clocking and Oscillator Circuit section).
18
If the external master clock continues to run in standby mode, the standby current increases to 150␣ µA typical at 5 V and 75 µA at 3 V. When using a crystal or
ceramic resonator across the MCLK pins as the clock source for the device, the internal oscillator continues to run in standby mode and the power dissipation
depends on the crystal or resonator type (see Standby Mode section).
19
Measured at dc and applies in the selected passband. PSRR at 50␣ Hz will exceed 120␣ dB with filter notches of 25 Hz or 50␣ Hz. PSRR at 60␣ Hz will exceed 120␣ dB
with filter notches of 20 Hz or 60␣ Hz.
20
PSRR depends on both gain and AVDD.
2
Low Level Input Channels, AIN1 and AIN2
High Level Input Channel, AIN3
Gain
AVDD = 3 V
AVDD = 5 V
Gain
AVDD = 3 V
AVDD = 5 V
1
86
90
2
78
78
4
85
84
8–128
93
91
1
68
72
2
60
60
4
67
66
8–128
75
73
Specifications subject to change without notice.
–4–
REV. A
AD7707
(AVDD = DVDD = +2.7 V TO +5.25 V, AGND = DGND = 0 V; fCLKIN = 2.4576 MHz; Input
DD unless otherwise noted.)
TIMING CHARACTERISTICS1, 2 Logic = 0, Logic 1 = DV
Parameter
fCLKIN
3, 4
tCLKIN LO
tCLKIN HI
t1
t2
Read Operation
t3
t4
t5 5
t6
t7
t8
t9 6
t10
Write Operation
t11
t12
t13
t14
t15
t16
Limit at
TMIN, TMAX
(B Version)
Units
Conditions/Comments
400
5
0.4 × tCLKIN
0.4 × tCLKIN
500 × tCLKIN
100
kHz min
MHz max
ns min
ns min
ns nom
ns min
Master Clock Frequency: Crystal Oscillator or Externally Supplied for
Specified Performance
Master Clock Input Low Time. tCLKIN = 1/fCLKIN
Master Clock Input High Time
DRDY High Time
RESET Pulsewidth
0
120
0
80
100
100
100
0
10
60
100
100
ns min
ns min
ns min
ns max
ns max
ns min
ns min
ns min
ns min
ns max
ns max
ns max
DRDY to CS Setup Time
CS Falling Edge to SCLK Rising Edge Setup Time
SCLK Falling Edge to Data Valid Delay
DVDD = +5␣ V
DVDD = +3.0␣ V
SCLK High Pulsewidth
SCLK Low Pulsewidth
CS Rising Edge to SCLK Rising Edge Hold Time
Bus Relinquish Time after SCLK Rising Edge
DVDD = +5␣ V
DVDD = +3.0␣ V
SCLK Falling Edge to DRDY High7
120
30
20
100
100
0
ns min
ns min
ns min
ns min
ns min
ns min
CS Falling Edge to SCLK Rising Edge Setup Time
Data Valid to SCLK Rising Edge Setup Time
Data Valid to SCLK Rising Edge Hold Time
SCLK High Pulsewidth
SCLK Low Pulsewidth
CS Rising Edge to SCLK Rising Edge Hold Time
NOTES
1
Sample tested at +25°C to ensure compliance. All input signals are specified with tr = tf = 5 ns (10% to 90% of DVDD) and timed from a voltage level of 1.6 V.
2
See Figures 16 and 17.
3
fCLKIN Duty Cycle range is 45% to 55%. f CLKIN must be supplied whenever the AD7707 is not in Standby mode. If no clock is present in this case, the device can
draw higher current than specified and possibly become uncalibrated.
4
The AD7707 is production tested with f CLKIN at 2.4576␣ MHz (1␣ MHz for some I DD tests). It is guaranteed by characterization to operate at 400␣ kHz.
5
These numbers are measured with the load circuit of Figure 1 and defined as the time required for the output to cross the VOL or VOH limits.
6
These numbers are derived from the measured time taken by the data output to change 0.5␣ V when loaded with the circuit of Figure 1. The measured number is then
extrapolated back to remove effects of charging or discharging the 50 pF capacitor. This means that the times quoted in the timing characteristics are the true bus
relinquish times of the part and as such are independent of external bus loading capacitances.
7
DRDY returns high after the first read from the device after an output update. The same data can be read again, if required, while DRDY is high, although care
should be taken that subsequent reads do not occur close to the next output update.
ISINK (800mA AT VDD = +5V
100mA AT VDD = +3V)
TO OUTPUT
PIN
+1.6V
50pF
ISOURCE (200mA AT VDD = +5V
100mA AT VDD = +3V)
Figure 1. Load Circuit for Access Time and Bus Relinquish Time
REV. A
–5–
AD7707
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
SOIC Package, Power Dissipation . . . . . . . . . . . . . . . 450 mW
θJA Thermal Impedance . . . . . . . . . . . . . . . . . . . . . . 75°C/W
Lead Temperature, Soldering
Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . +215°C
Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . +220°C
TSSOP Package, Power Dissipation . . . . . . . . . . . . . 450 mW
θJA Thermal Impedance . . . . . . . . . . . . . . . . . . . . . 139°C/W
Lead Temperature, Soldering
Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . +215°C
␣ ␣ ␣ ␣ Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . +220°C
ESD Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 kV
ABSOLUTE MAXIMUM RATINGS*
(TA = +25°C unless otherwise noted)
AVDD to AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +7␣ V
AVDD to DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +7␣ V
DVDD to AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +7␣ V
DVDD to DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +7␣ V
AVDD to DVDD . . . . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +7 V
DGND to AGND . . . . . . . . . . . . . . . . . . . . . –0.3␣ V to +0.3␣ V
AIN1, AIN2 Input Voltage to
LOCOM . . . . . . . . . . . . . . . . . . . . –0.3 V to AVDD + 0.3␣ V
AIN3 Input Voltage to HICOM . . . . . . . . . . . –11 V to +30␣ V
VBIAS to AGND . . . . . . . . . . . . . . . . –0.3 V to AVDD + 0.3␣ V
HICOM, LOCOM to AGND . . . . . . –0.3 V to AVDD + 0.3␣ V
REF(+), REF(–) to AGND . . . . . . . . –0.3 V to AVDD + 0.3␣ V
Digital Input Voltage to DGND . . . . –0.3 V to DVDD + 0.3 V
Digital Output Voltage to DGND . . . –0.3 V to DVDD + 0.3 V
Operating Temperature Range
Industrial (B Version) . . . . . . . . . . . . . . . . –40°C to +85°C
*Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
ORDERING GUIDE
Model
AD7707BR
AD7707BRU
EVAL-AD7707EB
VDD
Supply
Temperature
Range
Package
Description
Package
Options
2.7 V to 5.25 V
2.7 V to 5.25 V
–40°C to +85°C
–40°C to +85°C
Evaluation Board
SOIC
TSSOP
R-20
RU-20
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD7707 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
–6–
WARNING!
ESD SENSITIVE DEVICE
REV. A
AD7707
PIN CONFIGURATION
20 DGND
SCLK 1
19 DVDD
MCLK IN 2
MCLK OUT 3
18 DIN
CS 4
17 DOUT
16 DRDY
TOP VIEW 15 AGND
(Not to Scale)
14 REF IN(–)
AIN1 7
RESET 5
AVDD
AD7707
6
LOCOM 8
13 REF IN(+)
AIN2 9
12 VBIAS
AIN3 10
11 HICOM
PIN FUNCTION DESCRIPTIONS
Pin No.
Mnemonic
Function
1
SCLK
Serial Clock. Schmitt-Triggered Logic Input. An external serial clock is applied to this input to
access serial data from the AD7707. This serial clock can be a continuous clock with all data
transmitted in a continuous train of pulses. Alternatively, it can be a noncontinuous clock with
the information being transmitted to the AD7707 in smaller batches of data.
2
MCLK IN
Master Clock signal for the device. This can be provided in the form of a crystal/resonator or
external clock. A crystal/resonator can be tied across the MCLK IN and MCLK OUT pins.
Alternatively, the MCLK IN pin can be driven with a CMOS-compatible clock and MCLK
OUT left unconnected. The part can be operated with clock frequencies in the range 500 kHz to
5 MHz.
3
MCLK OUT
When the master clock for the device is a crystal/resonator, the crystal/resonator is connected
between MCLK IN and MCLK␣ OUT. If an external clock is applied to MCLK IN, MCLK
OUT provides an inverted clock signal. This clock can be used to provide a clock source for
external circuitry and is capable of driving one CMOS load. If the user does not require it, this
MCLK OUT can be turned off via the CLK DIS bit of the Clock Register. This ensures that the
part is not wasting unnecessary power driving capacitive loads on MCLK OUT.
4
CS
Chip Select. Active low Logic Input used to select the AD7707. With this input hard-wired low,
the AD7707 can operate in its three-wire interface mode with SCLK, DIN and DOUT used to
interface to the device. CS can be used to select the device in systems with more than one device
on the serial bus or as a frame synchronization signal in communicating with the AD7707.
5
RESET
Logic Input. Active low input that resets the control logic, interface logic, calibration coefficients, digital filter and analog modulator of the part to power-on status.
6
AVDD
Analog Supply Voltage, +2.7 V to +5.25 V operation.
7
AIN1
Low Level Analog Input Channel 1. This is used as a pseudo-differential input with respect to
LOCOM.
8
LOCOM
COMMON Input for low level input channels. Analog inputs on AIN1 and AIN2 must be referenced to this input.
9
AIN2
Low Level Analog Input Channel 2. This is used as a pseudo-differential input with respect to
LOCOM.
10
AIN3
Single-Ended High Level Analog Input Channel with respect to HICOM.
11
HICOM
COMMON Input for high level input channel. Analog input on AIN3 must be referenced to
this input.
12
VBIAS
VBIAS is used to level shift the high level input channel signal. This signal is used to ensure that
the AIN(+) and AIN(–) signals seen by the internal modulator are within its common-mode
range. VBIAS is normally connected to 2.5 V when AVDD = 5 V and 1.225 V when AVDD = 3 V.
13
REF IN(+)
Reference Input. Positive input of the differential reference input to the AD7707. The reference
input is differential with the provision that REF IN(+) must be greater than REF IN(–).
REF␣ IN(+) can lie anywhere between AVDD and AGND.
REV. A
–7–
AD7707
Pin No.
Mnemonic
Function
14
REF IN(–)
Reference Input. Negative input of the differential reference input to the AD7707. The
REF␣ IN(–) can lie anywhere between AVDD and AGND provided REF␣ IN(+) is greater than
REF␣ IN(–).
15
AGND
Analog Ground. Ground reference point for the AD7707’s internal analog circuitry.
16
DRDY
Logic Output. A logic low on this output indicates that a new output word is available from the
AD7707 data register. The DRDY pin will return high upon completion of a read operation of a
full output word. If no data read has taken place between output updates, the DRDY line will
return high for 500 × tCLK␣ IN cycles prior to the next output update. While DRDY is high, a read
operation should neither be attempted nor in progress to avoid reading from the data register as
it is being updated. The DRDY line will return low again when the update has taken place.
DRDY is also used to indicate when the AD7707 has completed its on-chip calibration
sequence.
17
DOUT
Serial Data Output with serial data being read from the output shift register on the part. This
output shift register can contain information from the setup register, communications register,
clock register or data register, depending on the register selection bits of the Communications
Register.
18
DIN
Serial Data Input with serial data being written to the input shift register on the part. Data from
this input shift register is transferred to the setup register, clock register or communications
register, depending, on the register selection bits of the Communications Register.
19
DVDD
Digital Supply Voltage, +2.7 V to +5.25 V operation.
20
DGND
Ground reference point for the AD7707’s internal digital circuitry.
OUTPUT NOISE FOR LOW LEVEL INPUT CHANNELS (5 V OPERATION)
Table I shows the AD7707 output rms noise and peak-to-peak resolution in unbuffered mode for the selectable notch and –3␣ dB
frequencies for the part, as selected by FS0, FS1 and FS2 of the Clock Register. The numbers given are for the bipolar input ranges
with a VREF of +2.5␣ V and AVDD = 5 V. These numbers are typical and are generated at an analog input voltage of 0 V. Table II
shows the rms noise and peak-to-peak resolution when operating in unbuffered mode. It is important to note that the peak-to-peak numbers represent the resolution for which there will be no code flicker. They are not calculated based on rms noise but on peak-to-peak noise. The
numbers given are for bipolar input ranges with a VREF of +2.5 V. These numbers are typical and are rounded to the nearest LSB.
The numbers apply for the CLK DIV bit of the Clock Register set to 0. The output noise comes from two sources. The first is the
electrical noise in the semiconductor devices (device noise) used in the implementation of the modulator. Secondly, when the analog
input is converted into the digital domain, quantization noise is added. The device noise is at a low level and is independent of frequency. The quantization noise starts at an even lower level but rises rapidly with increasing frequency to become the dominant noise
source. The numbers in the tables are given for the bipolar input ranges. For the unipolar ranges the rms noise numbers will be the
same as the bipolar range but the peak-to-peak resolution is now based on half the signal range which effectively means losing 1 bit of
resolution.
Table I. Output RMS Noise/Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +5 V
AIN1 and AIN2 Unbuffered Mode Only
Filter First
Notch and O/P –3␣ dB
Gain of
Data Rate
Frequency 1
Typical Output RMS Noise in ␮V (Peak-to-Peak Resolution in Bits)
Gain of
Gain of
Gain of
Gain of
Gain of
Gain of
Gain of
2
4
8
16
32
64
128
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
1.2 (16)
3.6 (16)
4.7 (16)
95 (13)
600 (10.5)
0.7 (16)
2.1 (16)
2.6 (16)
65 (13)
316 (10.5)
0.7 (16)
1.25 (16)
1.5 (16)
23.4 (13)
138 (10.5)
0.54 (16)
0.89 (16)
0.94 (16)
11.6 (13)
71 (10.5)
0.28 (16)
0.62 (16)
0.73 (16)
6.5 (13)
38 (10.5)
0.28 (16)
0.60 (15.5)
0.68 (15.5)
3.4 (13)
18 (10.5)
0.28 (15.5)
0.56 (14.5)
0.66 (14.5)
2.1 (12.5)
10 (10)
0.27 (14.5)
0.56 (13.5)
0.63 (13.5)
1.5 (12)
5.7 (10)
MCLK IN = 1 MHz
4.05␣ Hz
1.06␣ Hz
20␣ Hz
5.24␣ Hz
25␣ Hz
6.55␣ Hz
100␣ Hz
26.2␣ Hz
200␣ Hz
52.5␣ Hz
1.19 (16)
3.68 (16)
4.78 (16)
100 (13)
543 (10.5)
0.69 (16)
2.18 (16)
2.66 (16)
50.1 (13)
318 (10.5)
0.71 (16)
1.19 (16)
1.51 (16)
23.5 (13)
132 (10.5)
0.63 (16)
0.94 (16)
1.07 (16)
11.9 (13)
68.1 (10.5)
0.27 (16)
0.6 (16)
0.7 (16)
5.83 (13)
33.1 (10.5)
0.27 (16)
0.6 (15.5)
0.67 (15.5)
3.64 (13)
17.6 (10.5)
0.26 (15.5)
0.56 (14.5)
0.66 (14.5)
2.16 (12.5)
9.26 (10.5)
0.24 (15)
0.56 (13.5)
0.65 (13.5)
1.5 (12)
6.13 (10)
–8–
REV. A
AD7707
Table II. Output RMS Noise/Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +5 V
AIN1 and AIN2 Buffered Mode Only
Filter First
Notch and O/P –3␣ dB
Gain of
Data Rate
Frequency 1
Typical Output RMS Noise in ␮V (Peak-to-Peak Resolution in Bits)
Gain of
Gain of
Gain of
Gain of
Gain of
Gain of
2
4
8
16
32
64
Gain of
128
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
1.47 (16)
4.2 (16)
4.9 (16)
104 (13)
572 (10.5)
0.95 (16)
2.6 (16)
3 (16)
52 (13)
293 (10.5)
0.88 (16)
1.6 (16)
1.8 (16)
26 (13)
125 (10.5)
0.55 (16)
1 (16)
1.1 (16)
14 (13)
69 (10.5)
0.42 (16)
0.89 (15.5)
1 (15.5)
6.5 (13)
40 (10.5)
0.42 (16)
0.94 (15)
1 (14.5)
4.1 (12.5)
19 (10.5)
0.42 (15)
0.9 (14)
0.94 (14)
2.7 (12.5)
10 (10.5)
0.41 (14)
0.9 (13)
0.94 (13)
2.3 (11.5)
5.9 (10)
MCLK IN = 1 MHz
4.05␣ Hz
1.06␣ Hz
20␣ Hz
5.24␣ Hz
25␣ Hz
6.55␣ Hz
100␣ Hz
26.2␣ Hz
200␣ Hz
52.4␣ Hz
1.48 (16)
3.9 (16)
5.37 (16)
98.9 (13)
596 (10.5)
8.95 (16)
2.46 (16)
3.05 (16)
52.4 (13)
298 (10.5)
0.87 (16)
1.77 (16)
1.89 (16)
26.1 (13)
133 (10.5)
0.67 (16)
1.19 (16)
1.33 (16)
12.7 (13)
69.3 (10.5)
0.41 (16)
0.94 (16)
1.11 (15.5)
6.08 (13)
34.7 (10.5)
0.40 (16)
0.93 (15)
1.06 (14.5)
4.01 (12.5)
16.9 (10.5)
0.40 (15)
0.95 (14)
1.04 (13.5)
2.62 (12.5)
9.67 (10.5)
0.40 (14)
0.9 (13)
1.02 (12.5)
2.33 (11.5)
6.34 (10)
OUTPUT NOISE FOR LOW LEVEL INPUT CHANNELS (3 V OPERATION)
Table III shows the AD7707 output rms noise and peak-to-peak resolution in unbuffered mode for the selectable notch and –3␣ dB
frequencies for the part, as selected by FS0, FS1 and FS2 of the Clock Register. The numbers given are for the bipolar input ranges
with a VREF of +1.225␣ V and an AVDD = 3 V. These numbers are typical and are generated at an analog input voltage of 0 V. Table
IV shows the rms noise and peak-to-peak resolution when operating in unbuffered mode. It is important to note that the peak-to-peak
numbers represent the resolution for which there will be no code flicker. They are not calculated based on rms noise but on peak-to-peak noise.
The numbers given are for bipolar input ranges with a VREF of +1.225 V and for either buffered or unbuffered mode. These numbers
are typical and are rounded to the nearest LSB. The numbers apply for the CLK DIV bit of the Clock Register set to 0. The first is
the electrical noise in the semiconductor devices (device noise) used in the implementation of the modulator. Secondly, when the
analog input is converted into the digital domain, quantization noise is added. The device noise is at a low level and is independent of
frequency. The quantization noise starts at an even lower level but rises rapidly with increasing frequency to become the dominant
noise source. The numbers in the tables are given for the bipolar input ranges. For the unipolar ranges the rms noise numbers will be
the same as the bipolar range but the peak-to-peak resolution is now based on half the signal range which effectively means losing
1 bit of resolution.
Table III. Output RMS Noise/Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +3 V
AIN1 and AIN2 Unbuffered Mode Only
Filter First
Notch and O/P –3␣ dB
Gain of
Data Rate
Frequency 1
Typical Output RMS Noise in ␮V (Peak-to-Peak Resolution in Bits)
Gain of
Gain of
Gain of
Gain of
Gain of
Gain of
2
4
8
16
32
64
Gain of
128
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
1.60 (16)
3.8 (16)
4.4 (16)
53 (13)
300 (10.5)
0.8 (16)
1.9 (16)
2.2 (16)
24 (13)
138 (10.5)
0.48 (16)
1.1 (16)
1.35 (16)
15 (13)
80 (10.5)
0.29 (16)
0.64 (16)
0.78 (16)
6.8 (13)
34 (10.5)
0.29 (16)
0.60 (15.5)
0.7 (15)
3.6 (12.5)
18 (10.5)
0.27 (15.5)
0.6 (14.5)
0.68 (14.5)
2.1 (12.5)
8.7 (10.5)
0.26 (14.5)
0.6 (13.5)
0.64 (13.5)
1.5 (12)
4.8 (10)
0.26 (13.5)
0.6 (12.5)
0.64 (12.5)
1.3 (11)
3.4 (10)
MCLK IN = 1 MHz
4.05␣ Hz
1.06␣ Hz
20␣ Hz
5.24␣ Hz
25␣ Hz
6.55␣ Hz
100␣ Hz
26.2␣ Hz
200␣ Hz
52.4␣ Hz
1.56 (16)
3.85 (16)
4.56 (16)
45.7 (13)
262 (10.5)
0.88 (16)
2.02 (16)
2.4 (16)
22 (13)
125 (10.5)
0.52 (16)
1.15 (16)
1.4 (16)
13.7 (13)
66 (10.5)
0.3 (16)
0.74 (16)
0.79 (16)
5.27 (13)
32.4 (10.5)
0.28 (16)
0.63 (15.5)
0.68 (15)
2.64 (13)
18.4 (10.5)
0.27 (15.5)
0.57 (14.5)
0.66 (14.5)
2 (12.5)
8.6 (10.5)
0.27 (14.5)
0.61 (13.5)
0.64 (13.5)
1.59 (12)
4.64 (10.5)
0.26 (13.5)
0.58 (12.5)
0.64 (12.5)
1.4 (11)
3.3 (10)
REV. A
–9–
AD7707
Table IV. Output RMS Noise/Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +3 V
AIN1 and AIN2 Buffered Mode Only
Filter First
Notch and O/P –3␣ dB
Gain of
Data Rate
Frequency 1
Typical Output RMS Noise in ␮V (Peak-to-Peak Resolution in Bits)
␣ Gain of
Gain of
Gain of
Gain of
Gain of
Gain of
␣␣2
4
8
16
32
64␣ ␣ ␣ ␣
Gain of
128
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
1.80 (16)
4.1 (16)
5.1 (16)
50 (13)
275 (10.5)
1 (16)
2.4 (16)
3 (16)
27 (13)
125 (10.5)
0.7 (16)
1.5 (16)
1.8 (16)
12.3 (13)
80 (10.5)
0.41 (16)
1 (15.5)
1.1 (15.5)
6.4 (13)
39 (10.5)
0.41 (16)
0.91 (15)
0.94 (14.5)
4 (12.5)
16 (10.5)
0.41 (15)
0.89 (14)
0.94 (13.5)
2.7 (12.5)
8.9 (10.5)
0.41 (14)
0.86 (13)
0.99 (13)
2.2 (11.5)
5.2 (10)
0.41 (13)
0.83 (12)
0.99 (11.5)
1.8 (11)
4.2 (9.5)
MCLK IN = 1 MHz
4.05␣ Hz
1.06␣ Hz
20␣ Hz
5.24␣ Hz
25␣ Hz
6.55␣ Hz
100␣ Hz
26.2␣ Hz
200␣ Hz
52.4␣ Hz
1.75 (16)
4.21 (16)
5.15 (16)
46.1 (13)
282 (10.5)
1.18 (16)
2.5 (16)
2.8 (16)
24.3 (13)
123 (10.5)
0.67 (16)
1.48 (16)
1.8 (16)
13.6 (13)
66 (10.5)
0.44 (16)
1 (15.5)
1.15 (15.5)
6.71 (13)
35.3 (10.5)
0.41 (16)
0.94 (15)
1 (14.5)
4.1 (12.5)
14.8 (10.5)
0.44 (15)
0.96 (14)
1.02 (13.5)
2.54 (12.5)
9.91 (10.5)
0.43 (14)
0.89 (13)
0.96 (13)
2.3 (11.5)
5.48 (10)
0.43 (13)
0.86 (12)
1.03 (11.5)
2.15 (10.5)
4.01 (9.5)
OUTPUT NOISE FOR HIGH LEVEL INPUT CHANNEL AIN3 (5 V OPERATION)
Table V shows the AD7707 output rms noise and peak-to-peak resolution in unbuffered for the selectable notch and –3␣ dB
frequencies for the part, as selected by FS0, FS1 and FS2 of the Clock Register. The numbers given are for the ± 10 V,
± 5 V, 0 to 5 V and 0 V to +10 V ranges with a VREF of +2.5 V, HBIAS = 2.5 V, HICOM = AGND and AVDD = 5 V. These
numbers are typical and are generated at an analog input voltage of 0 V. Table VI meanwhile shows the output rms noise and
peak-to-peak resolution in buffered mode. It is important to note that these numbers represent the resolution for which there will be
no code flicker. They are not calculated based on rms noise but on peak-to-peak noise. Operating the high level channel with a gain
of 2 in bipolar mode gives an operating range of ± 10 V. Operating at a gain of 2 in unipolar mode gives a range of 0 V to
+10 V. Operating the high level channel with a gain of 4 in bipolar mode gives the ± 5 V operating range. Operating at a gain
of 4 in unipolar mode gives an operating range of 0 V to +5 V. Noise for all input ranges is shown in Appendix 1. The output noise comes from two sources. The first is the electrical noise in the semiconductor devices (device noise) used in the
implementation of the modulator. Secondly, when the analog input is converted into the digital domain, quantization noise
is added. The device noise is at a low level and is independent of frequency. The quantization noise starts at an even lower
level but rises rapidly with increasing frequency to become the dominant noise source. The numbers in the tables are given
for the bipolar input ranges. For the unipolar ranges the rms noise numbers will be the same as the bipolar range but the
peak-to-peak resolution is now based on half the signal range which effectively means losing 1 bit of resolution.
Table V. Output RMS Noise/Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +5 V
AIN3 Unbuffered Mode Only
Filter First
Notch and O/P
Data Rate
ⴞ10 V Range
ⴞ5 V Range
0 V to +10 V Range
0 V to +5 V Range
–3␣ dB
RMS Noise P-P (Bits) RMS Noise P-P (Bits) RMS Noise P-P (Bits) RMS Noise P-P (Bits)
Frequency (␮V)
Resolution (␮V)
Resolution (␮V)
Resolution (␮V)
Resolution
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
5.10
15.82
20.36
430
2350
16
16
16
13
10
3.52
9.77
12.29
212
1287
16
16
16
13
10
5.10
15.82
20.36
430
2350
16
16
16
12
9
3.52
9.77
12.29
212
1287
16
16
16
12
9
MCLK IN = 1 MHz
4.05␣ Hz
1.06␣ Hz
20␣ Hz
5.24␣ Hz
25␣ Hz
6.55␣ Hz
100␣ Hz
26.2␣ Hz
200␣ Hz
52.4␣ Hz
5.13
18.9
23.7
406
2184
16
16
16
13
10.5
3.53
13.25
15.3
174
1144
16
16
16
13
10.5
5.13
18.9
23.7
406
2184
16
16
16
12
9.5
3.53
13.25
15.3
174
1144
16
16
15.5
12
9.5
–10–
REV. A
AD7707
Table VI. Output RMS Noise/ Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +5 V
AIN3 Buffered Mode Only
Filter First
Notch and O/P –3␣ dB
Data Rate
Frequency
ⴞ10 V Range
ⴞ5 V Range
0 V to +10 V Range
0 to +5 V Range
RMS Noise P-P (Bits) RMS Noise P-P (Bits) RMS Noise P-P (Bits) RMS Noise P-P (Bits)
(␮V)
Resolution (␮V)
Resolution (␮V)
Resolution (␮V)
Resolution
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
7.4
22.2
26.6
475
2423
16
16
16
13
10.5
5.2
14.3
15.85
187
1097
16
16
16
13
10.5
7.4
22.2
26.6
475
2423
16
16
16
12
9.5
5.2
14.3
15.85
187
1097
16
16
16
12
9.5
MCLK IN = 1 MHz
4.05␣ Hz
1.06␣ Hz
20␣ Hz
5.24␣ Hz
25␣ Hz
6.55␣ Hz
100␣ Hz
26.2␣ Hz
200␣ Hz
52.4␣ Hz
7.63
20.25
23.5
377
2226
16
16
16
13
10.5
5.45
13.3
14.6
210
1132
16
16
16
13
10.5
7.63
20.25
23.5
377
2226
16
16
16
12
9.5
5.45
13.3
14.6
210
1132
16
16
15.5
12
9.5
OUTPUT NOISE FOR HIGH LEVEL INPUT CHANNEL AIN3 (5 V OPERATION)
Table VII shows the AD7707 output rms noise and peak-to-peak resolution for the selectable notch and –3␣ dB frequencies for the
part, as selected by FS0, FS1 and FS2 of the Clock Register. The numbers given are for the ± 5 V, 0 V to +5 V and 0 V to +10 V
ranges with a VREF of +1.225 V, HBIAS = 1.225 V, HICOM = AGND and AVDD = 3 V. These numbers are typical and are generated at an analog input voltage of 0 V for unbuffered mode of operation. The above operating ranges are only achievable in unbuffered mode when operating at 3 V due to common-mode limitations on the input amplifier. It is important to note that these numbers
represent the resolution for which there will be no code flicker. They are not calculated based on rms noise but on peak-to-peak noise. Operating
at a gain of 1 in unipolar mode provides a range of 0 V to +10 V. Operating the high level channel with a gain of 2 in bipolar mode
provides a ± 5 V operating range. Operating at a gain of 2 in unipolar mode provides an operating range of 0 V to +5 V. The output
noise comes from two sources. The first is the electrical noise in the semiconductor devices (device noise) used in the implementation
of the modulator. Secondly, when the analog input is converted into the digital domain, quantization noise is added. The device
noise is at a low level and is independent of frequency. The quantization noise starts at an even lower level but rises rapidly with
increasing frequency to become the dominant noise source. The numbers in the tables are given for the bipolar input ranges. For the
unipolar ranges the rms noise numbers will be the same as the bipolar range but the peak-to-peak resolution is now based on half the
signal range which effectively means losing 1 bit of resolution.
Table VII. Output RMS Noise/ Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +3 V
AIN3 Unbuffered Mode Only
0 V to +10 V Range
RMS Noise P-P (Bits)
(␮V)
Resolution
ⴞ5 V Range
RMS Noise P-P (Bits)
(␮V)
Resolution
0 to +5 V Range
RMS Noise P-P (Bits)
(␮V)
Resolution
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
12.4
30.35
34.55
498
2266
16
16
16
12.5
10.5
7.02
16.4
19.13
204
1151
16
16
16
13
10.5
7.02
16.4
19.13
204
1151
16
15.5
15
12
9.5
MCLK IN = 1 MHz
4.05␣ Hz
1.06␣ Hz
20␣ Hz
5.24␣ Hz
25␣ Hz
6.55␣ Hz
100␣ Hz
26.2␣ Hz
200␣ Hz
52.4␣ Hz
13.9
32.2
33.4
430
2207
16
16
16
13
10.5
7.3
17.4
18.57
200
1048
16
16
16
13
10.5
7.3
17.4
18.57
200
1048
16
15
15
12
9.5
Filter First
Notch and O/P
Data Rate
REV. A
–3␣ dB
Frequency
–11–
AD7707
32771
VDD = 5V
VREF = 2.5V
GAIN = 128
50Hz UPDATE RATE
32770
400
TA = +258C
RMS NOISE = 600nV
300
OCCURRENCE
CODE READ
32769
32768
32767
32766
32765
200
100
32764
32763
0
100
200
300
400 500 600
READING NO.
700
800
0
900 1000
32764
Figure 2. Typical Noise Plot @ Gain = 128 with 50 Hz
Update Rate for Low Level Input Channel
32765
32766
32767
CODE
32768
32769
32770
Figure 5. Histogram of Data in Figure 2
800
32769
10Hz UPDATE RATE, UNBUFFERED MODE
GAIN = 2, (610V INPUT RANGE)
BIPOLAR MODE
ANALOG INPUT SET ON CODE TRANSITION
700
600
10Hz UPDATE RATE
UNBUFFERED MODE
BIPOLAR MODE
GAIN = 2,
(610V INPUT RANGE)
CODE
OCCURRENCE
32768
32767
500
400
300
200
100
32766
0
200
400
600
READING NO.
800
1000
0
1
32767
2
32768
CODE
Figure 3. Typical Noise Plot for AIN3, High Level Input
Channel
10
Figure 6. Histogram of Data in Figure 3
0.6
HIGH LEVEL INPUT CHANNEL
610V INPUT RANGE
10Hz UPDATE RATE
9
0.5
8
LOW LEVEL INPUT CHANNEL
GAIN = 128
10Hz UPDATE RATE
6
5
4
RMS NOISE – mV
RMS NOISE – mV
7
BUFFERED MODE
AVDD = DVDD = 5V
REFIN(+) = 2.5V
REFIN(–) = AGND
TA = +258C
3
0.4
0.3
UNBUFFERED MODE
0.2
UNBUFFERED MODE
2
0.1
1
0
–10
–6
–2
2
AIN3 – Volts
6
0
–20
10
Figure 4. Typical RMS Noise vs. Analog Input Voltage for
High Level Input Channel, AIN3
BUFFERED MODE
AVDD = DVDD = 5V
REFIN(+) = 2.5V
REFIN(–) = AGND
TA = +25 C
–15
–10
–5
0
5
INPUT VOLTAGE – mV
10
15
20
Figure 7. Typical RMS Noise vs. Analog Input Voltage for
Low Level Input Channels, AIN1 and AIN2
–12–
REV. A
AD7707
TEK STOP: SINGLE SEQ 50.0kS/s
20
16
STANDBY CURRENT – mA
VDD
1
2
OSCILLATOR = 4.9152 MHz
MCLK IN = 0V OR VDD
12
VDD = 5V
8
VDD = 3V
4
2
OSCILLATOR = 2.4576 MHz
CH1 5.00V
CH2 2.00V
0
–40 –30 –20 –10
5ms/DIV
Figure 8. Typical Crystal Oscillator Power-Up Time
0
10 20 30 40
TEMPERATURE – 8C
50
60
70
80
Figure 9. Standby Current vs. Temperature
ON-CHIP REGISTERS
The AD7707 contains eight on-chip registers which can be accessed via the serial port of the part. The first of these is a Communications Register that controls the channel selection, decides whether the next operation is a read or write operation and also decides
which register the next read or write operation accesses. All communications to the part must start with a write operation to the
Communications Register. After power-on or RESET, the device expects a write to its Communications Register. The data written
to this register determines whether the next operation to the part is a read or a write operation and also determines to which register
this read or write operation occurs. Therefore, write access to any of the other registers on the part starts with a write operation to the
Communications Register followed by a write to the selected register. A read operation from any other register on the part (including
the Communications Register itself and the output data register) starts with a write operation to the Communications Register followed by a read operation from the selected register. The Communications Register also controls the standby mode and channel
selection and the DRDY status is also available by reading from the Communications Register. The second register is a Setup Register that determines calibration mode, gain setting, bipolar/unipolar operation and buffered mode. The third register is labelled the
Clock Register and contains the filter selection bits and clock control bits. The fourth register is the Data Register from which the
output data from the part is accessed. The final registers are the calibration registers which store channel calibration data. The registers are discussed in more detail in the following sections.
Communications Register (RS2, RS1, RS0 = 0, 0, 0)
The Communications Register is an 8-bit register from which data can either be read or to which data can be written. All communications to the part must start with a write operation to the Communications Register. The data written to the Communications Register determines whether the next operation is a read or write operation and to which register this operation takes place. Once the
subsequent read or write operation to the selected register is complete, the interface returns to where it expects a write operation to
the Communications Register. This is the default state of the interface, and on power-up or after a RESET, the AD7707 is in this
default state waiting for a write operation to the Communications Register. In situations where the interface sequence is lost, if a
write operation of sufficient duration (containing at least 32 serial clock cycles) takes place with DIN high, the AD7707 returns to
this default state. Table VIII outlines the bit designations for the Communications Register.
Table VIII. Communications Register
0/DRDY (0)
RS2 (0)
RS1 (0)
RS0 (0)
R/W (0)
STBY (0)
CH1 (0)
CH0 (0)
0/DRDY
For a write operation, a “0” must be written to this bit so that the write operation to the Communications Register
actually takes place. If a “1” is written to this bit, the part will not clock on to subsequent bits in the register. It will
stay at this bit location until a “0” is written to this bit. Once a “0” is written to this bit, the next seven bits will be
loaded to the Communications Register. For a read operation, this bit provides the status of the DRDY flag from
the part. The status of this bit is the same as the DRDY output pin.
RS2–RS0
Register Selection Bits. These three bits select to which one of eight on-chip registers the next read or write operation takes place, as shown in Table IX, along with the register size. When the read or write operation to the selected register is complete, the part returns to where it is waiting for a write operation to the Communications
Register. It does not remain in a state where it will continue to access the register.
REV. A
–13–
AD7707
Table IX. Register Selection
RS2
RS1
RS0
Register
Register Size
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
Communications Register
Setup Register
Clock Register
Data Register
Test Register
No Operation
Offset Register
Gain Register
8 Bits
8 Bits
8 Bits
16 Bits
8 Bits
24 Bits
24 Bits
R/W
Read/Write Select. This bit selects whether the next operation is a read or write operation to the selected register.
A “0” indicates a write cycle for the next operation to the appropriate register, while a “1” indicates a read operation from the appropriate register.
STBY
Standby. Writing a “1” to this bit puts the part into its standby or power-down mode. In this mode, the part consumes only 8 µA of power supply current. The part retains its calibration coefficients and control word information
when in STANDBY. Writing a “0” to this bit places the part in its normal operating mode. The serial interface on
the AD7707 remains operational when the part is in STBY mode.
CH1–CH0
Channel Select. These two bits select a channel for conversion or for access to the calibration coefficients as outlined in Table X. Three pairs of calibration registers on the part are used to store the calibration coefficients following a calibration on a channel. They are shown in Tables VII for the AD7707 to indicate which channel
combinations have independent calibration coefficients. With CH1 at Logic 1 and CH0 at a Logic 0, the part looks
at the LOCOM input internally shorted to itself. This can be used as a test method to evaluate the noise performance of the part with no external noise sources. In this mode, the LOCOM input should be connected to an
external voltage within the allowable common-mode range for the part.
Table X. Channel Selection for AD7707
CH1
CH0
AIN
Reference
Calibration Register Pair
0
0
1
1
0
1
0
1
AIN1
AIN2
LOCOM
AIN3
LOCOM
LOCOM
LOCOM
HICOM
Register Pair 0
Register Pair 1
Register Pair 0
Register Pair 2
–14–
REV. A
AD7707
Setup Register (RS2, RS1, RS0 = 0, 0, 1); Power-On/Reset Status: 01␣ Hex
The Setup Register is an eight-bit register from which data can either be read or to which data can be written. Table XI outlines the
bit designations for the Setup Register.
Table XI. Setup Register
MD1 (0)
MD0 (0)
G2 (0)
G1 (0)
B/U (0)
G0 (0)
BUF (0)
FSYNC (1)
MD1
MD0
Operating Mode
0
0
Normal Mode: this is the normal mode of operation of the device whereby the device is performing normal
conversions.
0
1
Self-Calibration: this activates self-calibration on the channel selected by CH1 and CH0 of the Communications Register. This is a one-step calibration sequence and when complete the part returns to Normal Mode
with MD1 and MD0 returning to 0, 0. The DRDY output or bit goes high when calibration is initiated and
returns low when this self-calibration is complete and a new valid word is available in the data register. The
zero-scale calibration is performed at the selected gain on internally shorted (zeroed) inputs and the fullscale calibration is performed at the selected gain on an internally-generated VREF/Selected Gain.
1
0
Zero-Scale System Calibration: this activates zero scale system calibration on the channel selected by CH1
and CH0 of the Communications Register. Calibration is performed at the selected gain on the input voltage
provided at the analog input during this calibration sequence. This input voltage should remain stable for
the duration of the calibration. The DRDY output or bit goes high when calibration is initiated and returns
low when this zero-scale calibration is complete and a new valid word is available in the data register. At the
end of the calibration, the part returns to Normal Mode with MD1 and MD0 returning to 0, 0.
1
1
Full-Scale System Calibration: this activates full-scale system calibration on the selected input channel.
Calibration is performed at the selected gain on the input voltage provided at the analog input during this
calibration sequence. This input voltage should remain stable for the duration of the calibration. Once again,
the DRDY output or bit goes high when calibration is initiated and returns low when this full-scale calibration is complete and a new valid word is available in the data register. At the end of the calibration, the part
returns to Normal Mode with MD1 and MD0 returning to 0, 0.
G2–G0
Gain Selection Bits. These bits select the gain setting for the on-chip PGA as outlined in Table XII.
Table XII. Gain Selection
G2
G1
G0
Gain Setting
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
1
2
4
8
16
32
64
128
B/U
Bipolar/Unipolar Operation. A “0” in this bit selects Bipolar Operation. A “1” in this bit selects Unipolar Operation.
BUF
Buffer Control. With this bit at “0,” the on-chip buffer on the analog input is shorted out. With the buffer shorted
out, the current flowing in the VDD line is reduced. When this bit is high, the on-chip buffer is in series with the
analog input allowing the input to handle higher source impedances.
FSYNC
Filter Synchronization. When this bit is high, the nodes of the digital filter, the filter control logic and the calibration control logic are held in a reset state and the analog modulator is also held in its reset state. When this bit goes
low, the modulator and filter start to process data and a valid word is available in 3 × 1/ (output update rate), i.e.,
the settling time of the filter. This FSYNC bit does not affect the digital interface and does not reset the DRDY
output if it is low.
REV. A
–15–
AD7707
Clock Register (RS2, RS1, RS0 = 0, 1, 0); Power-On/Reset Status: 05␣ Hex
The Clock Register is an 8-bit register from which data can either be read or to which data can be written. Table XIII outlines the bit
designations for the Clock Register.
Table XIII. Clock Register
ZERO (0)
ZERO (0)
CLKDIS (0)
CLKDIV (0)
CLK (1)
FS2 (0)
FS1 (0)
FS0 (1)
ZERO
Zero. A zero MUST be written to these bits to ensure correct operation of the AD7707. Failure to do so may
result in unspecified operation of the device.
CLKDIS
Master Clock Disable Bit. A Logic 1 in this bit disables the master clock from appearing at the MCLK OUT pin.
When disabled, the MCLK OUT pin is forced low. This feature allows the user the flexibility of using the MCLK
OUT as a clock source for other devices in the system or of turning off the MCLK OUT as a power saving feature.
When using an external master clock on the MCLK IN pin, the AD7707 continues to have internal clocks and will
convert normally with the CLKDIS bit active. When using a crystal oscillator or ceramic resonator across the
MCLK IN and MCLK OUT pins, the AD7707 clock is stopped and no conversions take place when the CLKDIS
bit is active.
CLKDIV
Clock Divider Bit. With this bit at a Logic 1, the clock frequency appearing at the MCLK IN pin is divided by two
before being used internally by the AD7707. For example, when this bit is set to 1, the user can operate with a
4.9152 MHz crystal between MCLK IN and MCLK OUT and internally the part will operate with the specified
2.4576 MHz. With this bit at a Logic 0, the clock frequency appearing at the MCLK IN pin is the frequency used
internally by the part.
CLK
Clock Bit. This bit should be set in accordance with the operating frequency of the AD7707. If the device has a
master clock frequency of 2.4576 MHz (CLKDIV = 0) or 4.9152 MHz (CLKDIV = 1), then this bit should be set
to a “1.” If the device has a master clock frequency of 1 MHz (CLKDIV = 0) or 2 MHz (CLKDIV = 1), this bit
should be set to a “0.” This bit sets up the appropriate scaling currents for a given operating frequency and also
chooses (along with FS2, FS1 and FS0) the output update rate for the device. If this bit is not set correctly for the
master clock frequency of the device, then the AD7707 may not operate to specification.
FS2, FS1, FS0 Filter Selection Bits. Along with the CLK bit, FS2, FS1 and FS0 determine the output update rate, filter
first notch and –3 dB frequency as outlined in Table XIV. The on-chip digital filter provides a sinc3 (or
Sinx/x3) filter response. Placing the first notch at 10 Hz places notches at both 50 and 60 Hz giving better
than 150 dB rejection at these frequencies. In association with the gain selection the filter cutoff also determines the output noise of the device. Changing the filter notch frequency, as well as the selected gain, impacts resolution. Tables I to IV show the effect of filter notch frequency and gain on the output noise and
effective resolution of the part. The output data rate (or effective conversion time) for the device is equal to
the frequency selected for the first notch of the filter. For example, if the first notch of the filter is selected
at 50 Hz, a new word is available at a 50 Hz output rate or every 20 ms. If the first notch is at 500 Hz, a
new word is available every 2 ms. A calibration should be initiated when any of these bits are changed.
The settling time of the filter to a full-scale step input is worst case 4 × 1/(output data rate). For example,
with the filter first notch at 50 Hz, the settling time of the filter to a full-scale step input is 80 ms max. If
the first notch is at 500 Hz, the settling time is 8 ms max. This settling time can be reduced to 3 × 1/ (output data rate) by synchronizing the step input change to a reset of the digital filter. In other words, if the step
input takes place with the FSYNC bit high, the settling time will be 3 × 1/(output data rate) from when the
FSYNC bit returns low.
The –3 dB frequency is determined by the programmed first notch frequency according to the relationship:
filter –3 dB frequency = 0.262 × filter first notch frequency
–16–
REV. A
AD7707
Table XIV. Output Update Rates
CLK*
FS2
FS1
FS0
Output Update Rate
–3 dB Filter Cutoff
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
20 Hz
25 Hz
100 Hz
200 Hz
50 Hz
60 Hz
250 Hz
500 Hz
4.054 Hz
4.23 Hz
4.84 Hz
4.96 Hz
10 Hz
10.34 Hz
11.90 Hz
12.2 Hz
5.24 Hz
6.55 Hz
26.2 Hz
52.4 Hz
13.1 Hz
15.7 Hz
65.5 Hz
131 Hz
1.06 Hz
1.11 Hz
1.27 Hz
1.3 Hz
2.62 Hz
2.71 Hz
3.13 Hz
3.2 Hz
*Assumes correct clock frequency on MCLK IN pin with CLKDIV bit set appropriately.
Data Register (RS2, RS1, RS0 = 0, 1, 1)
The Data Register on the part is a 16-bit read-only register that contains the most up-to-date conversion result from the AD7707. If
the Communications Register sets up the part for a write operation to this register, a write operation must actually take place to return the part to where it is expecting a write operation to the Communications Register. However, the 16 bits of data written to the
part will be ignored by the AD7707.
Test Register (RS2, RS1, RS0 = 1, 0, 0); Power-On/Reset Status: 00␣ Hex
The part contains a Test Register that is used when testing the device. The user is advised not to change the status of any of the bits
in this register from the default (Power-on or RESET) status of all 0s as the part will be placed in one of its test modes and will not
operate correctly.
Zero-Scale Calibration Register (RS2, RS1, RS0 = 1, 1, 0); Power-On/Reset Status: 1F4000␣ Hex
The AD7707 contains independent sets of zero-scale registers, one for each of the input channels. Each of these registers is a 24-bit
read/write register; 24 bits of data must be written otherwise no data will be transferred to the register. This register is used in conjunction with its associated full-scale register to form a register pair. These register pairs are associated with input channel pairs as
outlined in Table VII. While the part is set up to allow access to these registers over the digital interface, the part itself no longer has
access to the register coefficients to correctly scale the output data. As a result, there is a possibility that after accessing the calibration registers (either read or write operation) the first output data read from the part may contain incorrect data. In addition, a write
to the calibration register should not be attempted while a calibration is in progress. These eventualities can be avoided by taking the
FSYNC bit in the mode register high before the calibration register operation and taking it low after the operation is complete.
Full-Scale Calibration Register (RS2, RS1, RS0 = 1, 1, 1); Power-On/Reset Status: 5761AB␣ Hex
The AD7707 contains independent sets of full-scale registers, one for each of the input channels. Each of these registers is a 24-bit
read/write register; 24 bits of data must be written otherwise no data will be transferred to the register. This register is used in conjunction with its associated zero-scale register to form a register pair. These register pairs are associated with input channel pairs as
outlined in Table X. While the part is set up to allow access to these registers over the digital interface, the part itself no longer has
access to the register coefficients to correctly scale the output data. As a result, there is a possibility that after accessing the calibration registers (either read or write operation) the first output data read from the part may contain incorrect data. In addition, a write
to the calibration register should not be attempted while a calibration is in progress. These eventualities can be avoided by taking
FSYNC bit in the mode register high before the calibration register operation and taking it low after the operation is complete.
REV. A
–17–
AD7707
CALIBRATION SEQUENCES
The AD7707 contains a number of calibration options as previously outlined. Table XV summarizes the calibration types, the operations involved and the duration of the operations. There are two methods of determining the end of calibration. The first is to monitor when DRDY returns low at the end of the sequence. DRDY not only indicates when the sequence is complete, but also that the
part has a valid new sample in its data register. This valid new sample is the result of a normal conversion which follows the calibration sequence. The second method of determining when calibration is complete is to monitor the MD1 and MD0 bits of the Setup
Register. When these bits return to 0 (0 following a calibration command), it indicates that the calibration sequence is complete.
This method does not give any indication of there being a valid new result in the data register. However, it gives an earlier indication
than DRDY that calibration is complete. The duration to when the Mode Bits (MD1 and MD0) return to 0 00 represents the duration of the calibration carried out). The sequence to when DRDY goes low also includes a normal conversion and a pipeline delay,
tP, to correctly scale the results of this first conversion. tP will never exceed 2000 × tCLKIN. The time for both methods is given in the
table.
Table XV. Calibration Sequences
Calibration Type
MD1, MD0
Calibration Sequence
Duration to Mode Bits
Duration to DRDY
Self-Calibration
0, 1
6 × 1/Output Rate
9 × 1/Output Rate + tP
ZS System Calibration
FS System Calibration
1, 0
1, 1
Internal ZS Cal @ Selected Gain +
Internal FS Cal @ Selected Gain
ZS Cal on AIN @ Selected Gain
FS Cal on AIN @ Selected Gain
3 × 1/Output Rate
3 × 1/Output Rate
4 × 1/Output Rate + tP
4 × 1/Output Rate + tP
–18–
REV. A
AD7707
(Sigma-Delta Modulator) converts the sampled signal into a
digital pulse train whose duty cycle contains the digital information. The programmable gain function on the analog input is
also incorporated in this sigma-delta modulator with the input
sampling frequency being modified to give the higher gains. A
sinc3 digital low-pass filter processes the output of the sigmadelta modulator and updates the output register at a rate determined by the first notch frequency of this filter. The output data
can be read from the serial port randomly or periodically at any
rate up to the output register update rate. The first notch of this
digital filter (and hence its –3␣ dB frequency) can be programmed
via the Setup Register bits FS0 and FS1. With a master clock
frequency of 2.4576 MHz, the programmable range for this first
notch frequency is from 10 Hz to 500 Hz, giving a programmable range for the –3␣ dB frequency of 2.62 Hz to 131␣ Hz.
With a master clock frequency of 1 MHz, the programmable
range for this first notch frequency is from 4 Hz to 200 Hz,
giving a programmable range for the –3␣ dB frequency of 1.06␣ Hz
to 52.4␣ Hz.
CIRCUIT DESCRIPTION
The AD7707 is a sigma-delta A/D converter with on-chip digital
filtering, intended for the measurement of wide dynamic range,
low frequency signals such as those in industrial control or process control applications. It contains a sigma-delta (or chargebalancing) ADC, a calibration microcontroller with on-chip
static RAM, a clock oscillator, a digital filter and a bidirectional
serial communications port. The part consumes only 320 µA of
power supply current, making it ideal for battery-powered or
loop-powered instruments. On-chip thin-film resistors allow
± 10 V, ± 5 V, 0 V to +10 V and 0 V to +5 V high level input
signals to be directly accommodated on the analog input without
requiring split supplies, dc-dc converters or charge pumps. This
part operates with a supply voltage of 2.7 V to 3.3 V or 4.75 V
to 5.25 V.
The AD7707 contains two low level (AIN1 and AIN2) programmable-gain pseudo-differential analog input channels and one
high level (AIN3) single-ended input channel. For the low level
input channels the selectable gains are 1, 2, 4, 8, 16, 32, 64 and
128 allowing the part to accept unipolar signals of between
0 mV to +20 mV and 0 V to +2.5 V, or bipolar signals in the
range from ± 20 mV to ± 2.5 V when the reference input voltage
equals +2.5 V. With a reference voltage of +1.225 V, the input
ranges are from 0 mV to +10 mV to 0 V to +1.225 V in unipolar
mode, and from ± 10 mV to ± 1.225 V in bipolar mode. Note
that the signals are with respect to the LOCOM input.
The basic connection diagram for the AD7707 is shown in
Figure 10. An AD780 or REF192, precision +2.5 V reference,
provides the reference source for the part. On the digital side,
the part is configured for three-wire operation with CS tied to
DGND. A quartz crystal or ceramic resonator provide the master clock source for the part. In most cases, it will be necessary
to connect capacitors on the crystal or resonator to ensure that it
does not oscillate at overtones of its fundamental operating
frequency. The values of capacitors will vary, depending on the
manufacturer’s specifications. A similar circuit is applicable for
operation with 3 V supplies, in this case a 1.225 V reference
(AD1580) should be used for specified performance.
The high level input channel can directly accept input signals of
± 10 V with respect to HICOM when operating with 5 V supplies and a reference of 2.5 V. With 3 V supplies ± 5 V can be
accommodated on the AIN3 input.
The input signal to the analog input is continuously sampled at a
rate determined by the frequency of the master clock, MCLK␣ IN,
and the selected gain. A charge-balancing A/D converter
ANALOG
+5V SUPPLY
10mF
0.1mF
0.1mF
AVDD
AD7707
AIN1
LOW LEVEL
ANALOG
INPUT
DVDD
AIN2
DRDY
DATA READY
DOUT
RECEIVE (READ)
LOCOM
AIN3
HIGH LEVEL
ANALOG
INPUT
VBIAS
DIN
SERIAL DATA
HICOM
SCLK
ANALOG +5V
SUPPLY
AGND
DGND
RESET
CS
VIN
VOUT
AD780/
REF192
SERIAL CLOCK
+5V
REF IN(+)
10mF
0.1mF
MCLK IN
REF IN(–)
MCLK OUT
GND
CRYSTAL OR
CERAMIC
RESONATOR
Figure 10. Basic Connection Diagram for 5 V Operation
REV. A
–19–
AD7707
CSAMP must be charged through RSW and any additional source
impedances every input sample cycle. Therefore, in unbuffered
mode, source impedances mean a longer charge time for CSAMP
and this may result in gain errors on the part. Table XVI shows
the allowable external resistance/capacitance values, for unbuffered
mode, such that no gain error to the 16-bit level is introduced
on the part. Note that these capacitances are total capacitances
on the analog input. This external capacitance includes 10 pF
from pins and lead frame of the device.
ANALOG INPUT
Analog Input Ranges
The AD7707 contains two low level pseudo-differential analog
input channels AIN1 and AIN2. These input pairs provide
programmable-gain, differential input channels that can handle
either unipolar or pseudo bipolar input signals. It should be
noted that the bipolar input signals are referenced to the
LOCOM input. The AD7707 also has a high level analog input
channel AIN 3 which is referenced to HICOM. Figure 11
shows the input structure on the high level input channel.
In normal 5 V operation VBIAS is normally connected to 2.5 V
and HICOM is connected to AGND. This arrangement ensures
that the voltages seen internally are within the common-mode
range of the buffer in buffered mode and within the supply
range in unbuffered mode. This device can be programmed to
operate in either buffered or unbuffered mode via the BUF bit
in the setup register. Note that the signals on AIN3 are with
respect to the HICOM input and not with respect to AGND or
DGND.
AIN(+)
CSAMP
(7pF)
AIN(–)
FIRST
INTEGRATOR
HIGH INPUT
IMPEDANCE
1G
VDD/2
SWITCHING FREQUENCY DEPENDS ON
fCLKIN AND SELECTED GAIN
Figure 12. Unbuffered Analog Input Structure
The differential voltage seen by the AD7707 when using the
high level input channel is the difference between AIN3(+) and
AIN3(–) on the mux as shown in Figure 11.
Table XVI. External R, C Combination for No 16-Bit Gain
Error on Low Level Input Channels (Unbuffered Mode Only)
AIN3(+) = (AIN3 + 6 × VBIAS+ V (HICOM))/8
6R
RSW (7kV TYP)
Gain
0
External Capacitance (pF)
50
100
500
1000
5000
1
2
4
8–128
368 kΩ
177.2 kΩ
82.8 kΩ
35.2 kΩ
90.6 kΩ
44.2 kΩ
21.2 kΩ
9.6 kΩ
2.2 kΩ
1.12 kΩ
540 Ω
240 Ω
AIN3
1R = 5kV
VBIAS
1R
AIN3(+)
AIN3(–)
MUX
54.2 kΩ
26.4 kΩ
12.6 kΩ
5.8 kΩ
14.6 kΩ
7.2 kΩ
3.4 kΩ
1.58 Ω
8.2 kΩ
4 kΩ
1.94 kΩ
880 Ω
3R
HICOM
6R
400
Figure 11. AIN3 Input Structure
350
In unbuffered mode, the common-mode range of the low level
input channels is from AGND – 100 mV to AVDD +␣ 30 mV.
This means that in unbuffered mode the part can handle both
unipolar and bipolar input ranges for all gains. Absolute voltages of AGND – 100 mV can be accommodated on the analog
inputs without degradation in performance, but leakage current
increases appreciably with increasing temperature. In buffered
mode, the analog inputs can handle much larger source impedances, but the absolute input voltage range is restricted to between AGND␣ + 50 mV to AVDD – 1.5 V which also places
restrictions on the common-mode range. This means that in
buffered mode there are some restrictions on the allowable
gains for bipolar input ranges. Care must be taken in setting up
the common-mode voltage and input voltage range so that the
above limits are not exceeded, otherwise there will be a degradation in linearity performance.
In unbuffered mode, the analog inputs look directly into the
7␣ pF input sampling capacitor, CSAMP. The dc input leakage
current in this unbuffered mode is 1␣ nA maximum. As a result,
the analog inputs see a dynamic load that is switched at the
input sample rate (see Figure 12). This sample rate depends on
master clock frequency and selected gain. CSAMP is charged to
AIN(+) and discharged to AIN(–) every input sample cycle.
The effective on-resistance of the switch, RSW, is typically 7 kΩ.
EXTERNAL RESISTANCE – kV
AIN3(–) = V (HICOM) + 0.75 × (VBIAS – V (HICOM))
GAIN = 1
300
250
200
GAIN = 2
150
GAIN = 8-128
100
GAIN = 4
50
0
0
10
100
1000
EXTERNAL CAPACITANCE – pF
10000
Figure 13. External R, C Combination for No 16-Bit Gain
Error on Low Level Input Channels (Unbuffered Mode Only)
In buffered mode, the analog inputs look into the high impedance inputs stage of the on-chip buffer amplifier. CSAMP is
charged via this buffer amplifier such that source impedances do
not affect the charging of CSAMP. This buffer amplifier has an
offset leakage current of 1 nA. In buffered mode, large source
impedances result in a small dc offset voltage developed across
the source impedance, but not in a gain error.
–20–
REV. A
AD7707
Input Sample Rate
The modulator sample frequency for the AD7707 remains at
fCLKIN/128 (19.2␣ kHz @ fCLKIN = 2.4576␣ MHz) regardless of the
selected gain. However, gains greater than 1 are achieved by a
combination of multiple input samples per modulator cycle and
a scaling of the ratio of reference capacitor to input capacitor. As
a result of the multiple sampling, the input sample rate of the
device varies with the selected gain (see Table XVII). In buffered mode, the input impedance is constant. In unbuffered
mode, where the analog input looks directly into the sampling
capacitor, the effective input impedance is 1/CSAMP × fS where
C SAMP is the input sampling capacitance and fS is the input
sample rate.
Table XVII. Input Sampling Frequency vs. Gain
Gain
Input Sampling Frequency (fS)
1
2
4
8–128
fCLKIN/64 (38.4␣ kHz @ fCLKIN = 2.4576␣ MHz)
2 × fCLKIN/64 (76.8␣ kHz @ fCLKIN =2.4576␣ MHz)
4 × fCLKIN/64 (76.8␣ kHz @ fCLKIN =2.4576␣ MHz)
8 × fCLKIN/64 (307.2␣ kHz @ fCLKIN = 2.4576␣ MHz)
DIGITAL FILTERING
Bipolar/Unipolar Inputs
The analog inputs on the low level input channels on the AD7707
can accept either unipolar or bipolar input voltage ranges with
respect to LOCOM.
The high level input channel handles true bipolar signals of
± 10 V max for guaranteed operation.
Bipolar or unipolar options are chosen by programming the B/U
bit of the Setup Register. This programs the channel for either
unipolar or bipolar operation. Programming the channel for
either unipolar or bipolar operation does not change any of the
channel conditions, it simply changes the data output coding
and the points on the transfer function where calibrations occur.
In unipolar operation the output coding is straight binary. In
bipolar mode the output coding is offset binary.
REFERENCE INPUT
The AD7707 reference inputs, REF␣ IN(+) and REF␣ IN(–),
provide a differential reference input capability. The commonmode range for these differential inputs is from GND to AVDD.
The nominal reference voltage, VREF REF␣ IN(+)␣ – REF␣ IN(–),
for specified operation, is +2.5␣ V for the AD7707 operated with
an AVDD of 5 V and +1.225 V for the AD7707 operated with an
AVDD of +3 V. The part is functional with VREF voltages down
to 1 V, but with degraded performance since the LSB size is
smaller. REF␣ IN(+) must always be greater than REF␣ IN(–) for
correct operation of the AD7707.
Both reference inputs provide a high impedance, dynamic load
similar to the analog inputs in unbuffered mode. The maximum
dc input leakage current is ± 1 nA over temperature, and source
resistance may result in gain errors on the part. In this case, the
sampling switch resistance is 5 kΩ typ and the reference capacitor (CREF) varies with gain. The sample rate on the reference
inputs is fCLKIN/64 and does not vary with gain. For gains of 1
and 2, CREF is 8␣ pF; for a gain of 16, it is 5.5␣ pF; for a gain of
32, it is 4.25 pF; for a gain of 64, it is 3.625 pF and for a gain of
128, it is 3.3125␣ pF.
REV. A
The output noise performance outlined in Tables I through IV
is for an analog input of 0 V, which effectively removes the effect
of noise from the reference. To obtain the same noise performance as shown in the noise tables over the full input range
requires a low noise reference source for the AD7707. If the
reference noise in the bandwidth of interest is excessive, it will
degrade the performance of the AD7707. In bridge transducer
applications where the reference voltage for the ADC is derived
from the excitation voltage the effect of the noise in the excitation voltage will be removed as the application is ratiometric.
Recommended reference voltage sources for the AD7707 with
an AVDD of 5 V include the AD780, REF43 and REF192, while
the recommended reference sources for the AD7707 operated
with an AVDD of 3 V include the AD589 and AD1580. It is
generally recommended to decouple the output of these references in order to further reduce the noise level.
The AD7707 contains an on-chip low-pass digital filter which
processes the output of the part’s sigma-delta modulator. Therefore, the part not only provides the analog-to-digital conversion
function but also provides a level of filtering. There are a number of system differences when the filtering function is provided
in the digital domain rather than the analog domain and the
user should be aware of these.
First, since digital filtering occurs after the A-to-D conversion
process, it can remove noise injected during the conversion
process. Analog filtering cannot do this. Also, the digital filter
can be made programmable far more readily than an analog
filter. Depending on the digital filter design, this gives the user
the capability of programming cutoff frequency and output
update rate.
On the other hand, analog filtering can remove noise superimposed on the analog signal before it reaches the ADC. Digital
filtering cannot do this and noise peaks riding on signals near
full scale have the potential to saturate the analog modulator
and digital filter, even though the average value of the signal is
within limits. To alleviate this problem, the AD7707 has overrange headroom built into the sigma-delta modulator and digital
filter, which allows overrange excursions of 5% above the analog
input range. If noise signals are larger than this, consideration
should be given to analog input filtering, or to reducing the
input channel voltage so that its full scale is half that of the
analog input channel full scale. This will provide an overrange
capability greater than 100% at the expense of reducing the
dynamic range by 1 bit (50%).
In addition, the digital filter does not provide any rejection at
integer multiples of the digital filter’s sample frequency. However, the input sampling on the part provides attenuation at
multiples of the digital filter’s sampling frequency so that the
unattenuated bands actually occur around multiples of the
sampling frequency fS (as defined in Table XV). Thus the
unattenuated bands occur at n × fS (where n = 1, 2, 3 . . .). At
these frequencies, there are frequency bands, ± f3 dB wide f3 dB is
the cutoff frequency of the digital filter) at either side where
noise passes unattenuated to the output.
–21–
AD7707
Since the AD7707 contains this on-chip, low-pass filtering, a
settling time is associated with step function inputs and data on
the output will be invalid after a step change until the settling
time has elapsed. The settling time depends upon the output
rate chosen for the filter. The settling time of the filter to a fullscale step input can be up to four times the output data period.
For a synchronized step input (using the FSYNC function), the
settling time is three times the output data period.
Filter Characteristics
The AD7707’s digital filter is a low-pass filter with a (sinx/x)3
response (also called sinc3). The transfer function for this filter
is described in the z-domain by:
H (z ) =
1
N
×
1 − Z –N
3
1 − Z –1
and in the frequency domain by:
3
SIN ( N × π × f / fS )
1
H( f ) =
×
N
SIN ( π × f / f S )
where N is the ratio of the modulator rate to the output rate.
Post-Filtering
The on-chip modulator provides samples at a 19.2 kHz output
rate with fCLKIN at 2.4576␣ MHz. The on-chip digital filter decimates these samples to provide data at an output rate that corresponds to the programmed output rate of the filter. Since the
output data rate is higher than the Nyquist criterion, the output
rate for a given bandwidth will satisfy most application requirements. There may, however, be some applications which require
a higher data rate for a given bandwidth and noise performance.
Applications that need this higher data rate will require some
post-filtering following the digital filter of the AD7707.
Phase Response:
∠H = –3 π ( N – 2) × f / fS Rad
Figure 14 shows the filter frequency response for a cutoff frequency of 2.62␣ Hz, which corresponds to a first filter notch
frequency of 10␣ Hz. The plot is shown from dc to 65 Hz. This
response is repeated at either side of the digital filter’s sample
frequency and at either side of multiples of the filter’s sample
frequency.
The response of the filter is similar to that of an averaging filter,
but with a sharper roll-off. The output rate for the digital filter
corresponds with the positioning of the first notch of the filter’s
frequency response. Thus, for the plot of Figure 14 where the
output rate is 10 Hz, the first notch of the filter is at 10 Hz. The
notches of this (sinx/x)3 filter are repeated at multiples of the
first notch. The filter provides attenuation of better than 100␣ dB
at these notches.
0
–20
–40
–60
GAIN – dB
–80
Post-filtering can also be used to reduce the output noise from
the device for bandwidths below 2.62 Hz. At a gain of 128 and a
bandwidth of 2.62 Hz, the output rms noise is 450 nV. This is
essentially device noise or white noise and since the input is
chopped, the noise has a primarily flat frequency response. By
reducing the bandwidth below 2.62 Hz, the noise in the resultant passband can be reduced. A reduction in bandwidth by a
factor of 2 results in a reduction of approximately 1.25 in the
output rms noise. This additional filtering will result in a longer
settling-time.
Analog Filtering
–100
The digital filter does not provide any rejection at integer multiples of the modulator sample frequency, as outlined earlier.
However, due to the AD7707’s high oversampling ratio, these
bands occupy only a small fraction of the spectrum and most
broadband noise is filtered. This means that the analog filtering
requirements in front of the AD7707 are considerably reduced
versus a conventional converter with no on-chip filtering. In
addition, because the part’s common-mode rejection performance of 100 dB extends out to several kHz, common-mode
noise in this frequency range will be substantially reduced.
–120
–140
–160
–180
–200
–220
–240
For example, if the required bandwidth is 7.86␣ Hz, but the
required update rate is 100 Hz, the data can be taken from the
AD7707 at the 100 Hz rate giving a –3 dB bandwidth of 26.2 Hz.
Post-filtering can be applied to this to reduce the bandwidth and
output noise, to the 7.86 Hz bandwidth level, while maintaining
an output rate of 100 Hz.
0
10
20
30
40
FREQUENCY – Hz
50
60
Figure 14. Frequency Response of AD7707 Filter
Simultaneous 50 Hz and 60 Hz rejection is obtained by placing
the first notch at 10 Hz. Operating with an update rate of 10 Hz
places notches at both 50 Hz and 60 Hz giving better than 100 dB
rejection at these frequencies.
The cutoff frequency of the digital filter is determined by the
value loaded to bits FS0 to FS2 in the CLOCK Register. Programming a different cutoff frequency via FS0 and FS1 does not
alter the profile of the filter response, it changes the frequency of
the notches. The output update of the part and the frequency of
the first notch correspond.
Depending on the application, however, it may be necessary to
provide attenuation prior to the AD7707 in order to eliminate
unwanted frequencies from these bands which the digital filter
will pass. It may also be necessary in some applications to
provide analog filtering in front of the AD7707 to ensure that
differential noise signals outside the band of interest do not
saturate the analog modulator.
If passive components are placed in front of the AD7707 in
unbuffered mode, care must be taken to ensure that the source
impedance is low enough not to introduce gain errors in the system. This significantly limits the amount of passive antialiasing
–22–
REV. A
AD7707
filtering which can be provided in front of the AD7707 when it
is used in unbuffered mode. However, when the part is used in
buffered mode, large source impedances will simply result in a
small dc offset error (a 10 kΩ source resistance will cause an
offset error of less than 10␣ µV). Therefore, if the system requires
any significant source impedances to provide passive analog
filtering in front of the AD7707, it is recommended that the part
be operated in buffered mode.
CALIBRATION
The AD7707 provides a number of calibration options which
can be programmed via the MD1 and MD0 bits of the Setup
Register. The different calibration options are outlined in the
Setup Register and Calibration Sequences sections. A calibration
cycle may be initiated at any time by writing to these bits of the
Setup Register. Calibration on the AD7707 removes offset and
gain errors from the device. A calibration routine should be
initiated on the device whenever there is a change in the ambient
operating temperature or supply voltage. It should also be initiated if there is a change in the selected gain, filter notch or bipolar/unipolar input range.
The AD7707 offers self-calibration and system calibration facilities. For full calibration to occur on the selected channel, the onchip microcontroller must record the modulator output for two
different input conditions. These are “zero-scale” and “fullscale” points. These points are derived by performing a conversion on the different input voltages provided to the input of the
modulator during calibration. As a result, the accuracy of the
calibration can only be as good as the noise level that it provides
in normal mode. The result of the “zero-scale” calibration conversion is stored in the Zero-Scale Calibration Register while the
result of the “full-scale” calibration conversion is stored in the
Full-Scale Calibration Register. With these readings, the microcontroller can calculate the offset and the gain slope for the
input-to-output transfer function of the converter.
Self-Calibration
A self-calibration is initiated on the AD7707 by writing the
appropriate values (0, 1) to the MD1 and MD0 bits of
the Setup Register. In the self-calibration mode with a unipolar
input range, the zero-scale point used in determining the calibration coefficients is with the inputs of the differential pair internally shorted on the part (i.e., AIN1 = LOCOM = Internal Bias
Voltage in the case of the AD7707. The PGA is set for the selected gain (as per G1 and G0 bits in the Communications Register) for this zero-scale calibration conversion. The full-scale
calibration conversion is performed at the selected gain on an
internally-generated voltage of VREF/Selected Gain.
The duration time for the calibration is 6 × 1/Output Rate. This
is made up of 3 × 1/Output Rate for the zero-scale calibration
and 3 × 1/Output Rate for the full-scale calibration. At this time
the MD1 and MD0 bits in the Setup Register return to 0, 0.
This gives the earliest indication that the calibration sequence is
complete. The DRDY line goes high when calibration is initiated and does not return low until there is a valid new word in
the data register. The duration time from the calibration command being issued to DRDY going low is 9 × 1/Output Rate.
This is made up of 3 × 1/Output Rate for the zero-scale calibration, 3 × 1/Output Rate for the full-scale calibration, 3 × 1/Output Rate for a conversion on the analog input and some overhead
REV. A
to correctly set up the coefficients. If DRDY is low before (or
goes low during) the calibration command write to the Setup
Register, it may take up to one modulator cycle (MCLK␣ IN/
128) before DRDY goes high to indicate that calibration is in
progress. Therefore, DRDY should be ignored for up to one
modulator cycle after the last bit is written to the Setup Register
in the calibration command.
For bipolar input ranges in the self-calibrating mode, the sequence is very similar to that just outlined. In this case, the two
points are exactly the same as above but, since the part is configured for bipolar operation, the shorted inputs point is actually
midscale of the transfer function.
Errors due to resistor mismatch in the attenuator on the high
level input channel AIN3 are not removed by a self-calibration.
System Calibration
System calibration allows the AD7707 to compensate for system
gain and offset errors as well as its own internal errors. System
calibration performs the same slope factor calculations as selfcalibration, but uses voltage values presented by the system to
the AIN inputs for the zero- and full-scale points. Full system
calibration requires a two-step process, a ZS System Calibration
followed by an FS System Calibration.
For a full system calibration, the zero-scale point must be presented to the converter first. It must be applied to the converter
before the calibration step is initiated and remain stable until the
step is complete. Once the system zero-scale voltage has been
set up, a ZS System Calibration is then initiated by writing the
appropriate values (1, 0) to the MD1 and MD0 bits of the
Setup Register. The zero-scale system calibration is performed
at the selected gain. The duration of the calibration is 3 × 1/
Output Rate. At this time, the MD1 and MD0 bits in the Setup
Register return to 0, 0. This gives the earliest indication that the
calibration sequence is complete. The DRDY line goes high
when calibration is initiated and does not return low until there
is a valid new word in the data register. The duration time from
the calibration command being issued to DRDY going low is 4
× 1/Output Rate as the part performs a normal conversion on
the AIN voltage before DRDY goes low. If DRDY is low before
(or goes low during) the calibration command write to the Setup
Register, it may take up to one modulator cycle (MCLK␣ IN/128)
before DRDY goes high to indicate that calibration is in progress.
Therefore, DRDY should be ignored for up to one modulator
cycle after the last bit is written to the Setup Register in the
calibration command.
After the zero-scale point is calibrated, the full-scale point is
applied to AIN and the second step of the calibration process is
initiated by again writing the appropriate values (1, 1) to MD1
and MD0. Again, the full-scale voltage must be set up before
the calibration is initiated and it must remain stable throughout
the calibration step. The full-scale system calibration is performed at the selected gain. The duration of the calibration is
3 × 1/Output Rate. At this time, the MD1 and MD0 bits in the
Setup Register return to 0, 0. This gives the earliest indication
that the calibration sequence is complete. The DRDY line goes
high when calibration is initiated and does not return low until
there is a valid new word in the data register. The duration time
from the calibration command being issued to DRDY going low
is 4 × 1/Output Rate as the part performs a normal conversion
–23–
AD7707
on the AIN voltage before DRDY goes low. If DRDY is low
before (or goes low during) the calibration command write to
the Setup Register, it may take up to one modulator cycle
(MCLK␣ IN/128) before DRDY goes high to indicate that calibration is in progress. Therefore, DRDY should be ignored for
up to one modulator cycle after the last bit is written to the
Setup Register in the calibration command.
the part is used in unipolar mode and required to remove an
offset of 0.2 × VREF/GAIN, the span range the system calibration can handle is 0.85 × VREF/GAIN.
1.05 3 VREF/GAIN
In the unipolar mode, the system calibration is performed between the two endpoints of the transfer function; in the bipolar
mode, it is performed between midscale (zero differential voltage) and positive full-scale.
GAIN CALIBRATIONS EXPAND
OR CONTRACT THE
AD7707 INPUT RANGE
–0V DIFFERENTIAL
NOMINAL ZERO
SCALE POINT
OFFSET CALIBRATIONS MOVE
INPUT RANGE UP OR DOWN
The fact that the system calibration is a two-step calibration
offers another feature. After the sequence of a full system calibration has been completed, additional offset or gain calibrations can be performed by themselves to adjust the system zero
reference point or the system gain. Calibrating one of the parameters, either system offset or system gain, will not affect the
other parameter.
LOWER LIMIT ON
AD7707 INPUT VOLTAGE
–1.05 3 VREF/GAIN
Figure 15. Span and Offset Limits for Low Level Input
Channels AIN1 and AIN2
System calibration can also be used to remove any errors from
source impedances on the analog input when the part is used in
unbuffered mode. A simple R, C antialiasing filter on the front
end may introduce a gain error on the analog input voltage, but
the system calibration can be used to remove this error.
Span and Offset Limits on the Low Level Input Channels
AIN1 and AIN2
Whenever a system calibration mode is used, there are limits on
the amount of offset and span which can be accommodated.
The overriding requirement in determining the amount of offset
and gain that can be accommodated by the part is the requirement that the positive full-scale calibration limit is < 1.05 ×
VREF/GAIN. This allows the input range to go 5% above the
nominal range. The built-in headroom in the AD7707’s analog
modulator ensures that the part will still operate correctly with a
positive full-scale voltage that is 5% beyond the nominal.
The input span in both the unipolar and bipolar modes has a
minimum value of 0.8 × VREF/GAIN and a maximum value of
2.1 × VREF/GAIN. However, the span (which is the difference
between the bottom of the AD7707’s input range and the top of
its input range) has to take into account the limitation on the
positive full-scale voltage. The amount of offset which can be
accommodated depends on whether the unipolar or bipolar
mode is being used. Once again, the offset has to take into account the limitation on the positive full-scale voltage. In unipolar mode, there is considerable flexibility in handling negative
offsets. In both unipolar and bipolar modes, the range of positive offsets that can be handled by the part depends on the
selected span. Therefore, in determining the limits for system
zero-scale and full-scale calibrations, the user has to ensure that
the offset range plus the span range does exceed 1.05 × VREF/
GAIN. This is best illustrated by looking at a few examples.
If the part is used in unipolar mode with a required span of
0.8 × VREF/GAIN, the offset range the system calibration can
handle is from –1.05 × VREF/GAIN to +0.25 × VREF/GAIN. If
the part is used in unipolar mode with a required span of VREF/
GAIN, the offset range the system calibration can handle is
from –1.05 × VREF/GAIN to +0.05 × VREF/GAIN. Similarly, if
UPPER LIMIT ON
AD7707 INPUT VOLTAGE
AD7707 LOW LEVEL
INPUT CHANNEL
INPUT RANGE
(0.8 3 VREF/GAIN TO
2.1 3 VREF/GAIN)
If the part is used in bipolar mode with a required span of ± 0.4
× VREF/GAIN, the offset range the system calibration can handle
is from –0.65 × VREF/GAIN to +0.65 × VREF/GAIN. If the part is
used in bipolar mode with a required span of ± VREF/GAIN,
then the offset range which the system calibration can handle is
from –0.05 × VREF/GAIN to +0.05 × VREF/GAIN. Similarly, if
the part is used in bipolar mode and required to remove an
offset of ± 0.2 × VREF/GAIN, the span range the system calibration can handle is ± 0.85 × VREF/GAIN. Figure 15 shows a
graphical representation of the span and offset limits for the low
level input channels.
Span and Offset Limits on the High Level Input Channel
AIN3
The exact same reasoning as above can be applied to the high
level input channel. When using the high level channel the attenuator provides an attenuation factor of 8. All span and offset
limits should be multiplied by a factor of 8. Therefore, the range
of input span in both the unipolar and bipolar modes has a
minimum value of 6.4 × VREF/GAIN and a maximum value of
16.8 × VREF/GAIN. The offset range plus the span range cannot
exceed 8.4 × VREF/GAIN.
Power-Up and Calibration
On power-up, the AD7707 performs an internal reset that sets
the contents of the internal registers to a known state. There are
default values loaded to all registers after power-on or reset.
The default values contain nominal calibration coefficients for
the calibration registers. However, to ensure correct calibration
for the device, a calibration routine should be performed after
power-up. A calibration should be performed if the update-rate
or gain are changed.
The power dissipation and temperature drift of the AD7707 are
low and no warm-up time is required before the initial calibration is performed. However, if an external reference is being
used, this reference must have stabilized before calibration is
initiated. Similarly, if the clock source for the part is generated
from a crystal or resonator across the MCLK pins, the start-up
time for the oscillator circuit should elapse before a calibration
is initiated on the part (see the following).
–24–
REV. A
AD7707
the current drain varies across crystal types. When using a crystal with an ESR of 700 Ω or when using a ceramic resonator, the
increase in the typical current over an externally-applied clock is
20 µA with DVDD = +3 V and 200 µA with DVDD = +5 V. When
using a crystal with an ESR of 3 kΩ, the increase in the typical
current over an externally applied clock is again 100 µA with
DVDD = +3 V but 400 µA with DVDD = +5␣ V.
C1
MCLK IN
C2
CRYSTAL OR
CERAMIC
RESONATOR
AD7707
MCLK OUT
Figure 16. Crystal/Resonator Connection for the AD7707
USING THE AD7707
Clocking and Oscillator Circuit
The AD7707 requires a master clock input, which may be an
external CMOS compatible clock signal applied to the MCLK␣ IN
pin with the MCLK␣ OUT pin left unconnected. Alternatively, a
crystal or ceramic resonator of the correct frequency can be
connected between MCLK␣ IN and MCLK␣ OUT as shown in
Figure 16, in which case the clock circuit will function as an
oscillator, providing the clock source for the part. The input
sampling frequency, the modulator sampling frequency, the
–3␣ dB frequency, output update rate and calibration time are all
directly related to the master clock frequency, fCLKIN. Reducing
the master clock frequency by a factor of 2 will halve the above
frequencies and update rate and double the calibration time.
The current drawn from the DVDD power supply is also related
to fCLKIN. Reducing fCLKIN by a factor of 2 will halve the DVDD
current but will not affect the current drawn from the AVDD.
Using the part with a crystal or ceramic resonator between the
MCLK IN and MCLK OUT pins generally causes more current to be drawn from DVDD than when the part is clocked from
a driven clock signal at the MCLK IN pin. This is because the
on-chip oscillator circuit is active in the case of the crystal or
ceramic resonator. Therefore, the lowest possible current on the
AD7707 is achieved with an externally applied clock at the
MCLK IN pin with MCLK OUT unconnected, unloaded and
disabled.
The amount of additional current taken by the oscillator depends on a number of factors—first, the larger the value of
capacitor (C1 and C2) placed on the MCLK␣ IN and MCLK␣ OUT
pins, the larger the current consumption on the AD7707. Care
should be taken not to exceed the capacitor values recommended by the crystal and ceramic resonator manufacturers to
avoid consuming unnecessary current. Typical values for C1
and C2 are recommended by crystal or ceramic resonator
manufacturers, these are in the range of 30 pF to 50 pF and if
the capacitor values on MCLK IN and MCLK OUT are kept in
this range they will not result in any excessive current. Another
factor that influences the current is the effective series resistance
(ESR) of the crystal that appears between the MCLK IN and
MCLK OUT pins of the AD7707. As a general rule, the lower
the ESR value the lower the current taken by the oscillator circuit.
When operating with a clock frequency of 2.4576 MHz, there is
50 µA difference in the current between an externally applied
clock and a crystal resonator when operating with a DVDD of
+3 V. With DVDD = +5 V and fCLKIN = 2.4576␣ MHz, the typical
current increases by 250␣ µA for a crystal/resonator supplied
clock versus an externally applied clock. The ESR values for
crystals and resonators at this frequency tend to be low and as a
result there tends to be little difference between different crystal
and resonator types.
When operating with a clock frequency of 1 MHz, the ESR
value for different crystal types varies significantly. As a result,
REV. A
The on-chip oscillator circuit also has a start-up time associated
with it before it is oscillating at its correct frequency and correct
voltage levels. Typical start-up times with DVDD = 5 V are 6 ms
using a 4.9512 MHz crystal, 16 ms with a 2.4576 MHz crystal
and 20 ms with a 1 MHz crystal oscillator. Start-up times are
typically 20% slower when the power supply voltage is reduced
to 3 V. At 3 V supplies, depending on the loading capacitances
on the MCLK pins, a 1 MΩ feedback resistor may be required
across the crystal or resonator in order to keep the start-up times
around the 20 ms duration.
The AD7707’s master clock appears on the MCLK OUT pin of
the device. The maximum recommended load on this pin is one
CMOS load. When using a crystal or ceramic resonator to generate the AD7707’s clock, it may be desirable to use this clock
as the clock source for the system. In this case, it is recommended that the MCLK OUT signal is buffered with a CMOS
buffer before being applied to the rest of the circuit.
System Synchronization
The FSYNC bit of the Setup Register allows the user to reset
the modulator and digital filter without affecting any of the
setup conditions on the part. This allows the user to start gathering samples of the analog input from a known point in time,
i.e., when the FSYNC is changed from 1 to 0.
With a 1 in the FSYNC bit of the Setup Register, the digital
filter and analog modulator are held in a known reset state and
the part is not processing any input samples. When a 0 is then
written to the FSYNC bit, the modulator and filter are taken
out of this reset state and the part starts to gather samples again
on the next master clock edge.
The FSYNC input can also be used as a software start convert
command allowing the AD7707 to be operated in a conventional converter fashion. In this mode, writing to the FSYNC bit
starts conversion and the falling edge of DRDY indicates when
conversion is complete. The disadvantage of this scheme is that
the settling time of the filter has to be taken into account for
every data register update. This means that the rate at which the
data register is updated is three times slower in this mode.
Since the FSYNC bit resets the digital filter, the full settling
time of 3 × 1/Output Rate has to elapse before there is a new
word loaded to the output register on the part. If the DRDY
signal is low when FSYNC goes to a 0, the DRDY signal will
not be reset high by the FSYNC command. This is because the
AD7707 recognizes that there is a word in the data register
which has not been read. The DRDY line will stay low until an
update of the data register takes place, at which time it will go
high for 500 × tCLKIN before returning low again. A read from
the data register resets the DRDY signal high and it will not
return low until the settling time of the filter has elapsed (from
the FSYNC command) and there is a valid new word in the
data register. If the DRDY line is high when the FSYNC
command is issued, the DRDY line will not return low until the
settling time of the filter has elapsed.
–25–
AD7707
Reset Input
Accuracy
The RESET input on the AD7707 resets all the logic, the digital
filter and the analog modulator, while all on-chip registers are
reset to their default state. DRDY is driven high and the AD7707
ignores all communications to any of its registers while the
RESET input is low. When the RESET input returns high, the
AD7707 starts to process data and DRDY will return low in 3 ×
1/Output Rate indicating a valid new word in the data register.
However, the AD7707 operates with its default setup conditions
after a RESET and it is generally necessary to set up all registers
and carry out a calibration after a RESET command.
Sigma-Delta ADCs, like VFCs and other integrating ADCs, do
not contain any source of nonmonotonicity and inherently offer
no missing codes performance. The AD7707 achieves excellent
linearity by the use of high quality, on-chip capacitors, which
have a very low capacitance/voltage coefficient. The device also
achieves low input drift through the use of chopper-stabilized
techniques in its input stage. To ensure excellent performance
over time and temperature, the AD7707 uses digital calibration
techniques that minimize offset and gain error.
The AD7707’s on-chip oscillator circuit continues to function
even when the RESET input is low. The master clock signal
continues to be available on the MCLK OUT pin. Therefore, in
applications where the system clock is provided by the AD7707’s
clock, the AD7707 produces an uninterrupted master clock
during RESET commands.
Charge injection in the analog switches and dc leakage currents
at the sampling modes are the primary sources of offset voltage
drift in the converter. The dc input leakage current is essentially
independent of the selected gain. Gain drift within the converter
depends primarily upon the temperature tracking of the internal
capacitors. It is not affected by leakage currents.
Standby Mode
Measurement errors due to offset drift or gain drift can be eliminated at any time by recalibrating the converter. Using the system calibration mode can also minimize offset and gain errors in
the signal conditioning circuitry. Integral and differential linearity errors are not significantly affected by temperature changes.
The STBY bit in the Communications Register of the AD7707
allows the user to place the part in a power-down mode when it
is not required to provide conversion results. The AD7707
retains the contents of all its on-chip registers (including the
data register) while in standby mode. When released from
standby mode, the part starts to process data and a new word is
available in the data register in 3 × 1/Output rate from when a 0
is written to the STBY bit.
Drift Considerations
POWER SUPPLIES
Placing the part in standby mode reduces the total current to
9␣ µA typical with 5 V supplies and 4 µA with 3 V supplies when
the part is operated from an external master clock provided this
master clock is stopped. If the external clock continues to drive
the MCLK IN pin in standby mode, the standby current increases to 150␣ µA typical with 5 V supplies and 75 µA typical
with 3 V supplies. If a crystal or ceramic resonator is used as the
clock source, the total current in standby mode is 400␣ µA typical
with 5 V supplies and 90 µA with 3 V supplies. This is because
the on-chip oscillator circuit continues to run when the part is in
its standby mode. This is important in applications where the
system clock is provided by the AD7707’s clock, so that the
AD7707 produces an uninterrupted master clock even when it is
in its standby mode. The serial interface remains operational
when in standby mode so that data can be read from the output
register in standby, regardless of whether or not the master clock
is stopped.
1600
1400
USING CRYSTAL OSCILLATOR
TA = +258C
UNBUFFERED MODE
GAIN = 128
1200
1000
IDD – mA
The STBY bit does not affect the digital interface, nor does it
affect the status of the DRDY line. If DRDY is high when the
STBY bit is brought low, it will remain high until there is a valid
new word in the data register. If DRDY is low when the STBY
bit is brought low, it will remain low until the data register is
updated, at which time the DRDY line will return high for
500␣ ×␣ tCLKIN before returning low again. If DRDY is low when
the part enters its standby mode (indicating a valid unread word
in the data register), the data register can be read while the part
is in standby. At the end of this read operation, the DRDY will
be reset high as normal.
The AD7707 operates with power supplies between 2.7 V and
5.25 V. There is no specific power supply sequence required for
the AD7707, either the AVDD or the DVDD supply can come up
first. In normal operation the DVDD must not exceed AVDD by
0.3 V. While the latch-up performance of the AD7707 is good,
it is important that power is applied to the AD7707 before signals at REF␣ IN, AIN or the logic input pins in order to avoid
excessive currents. If this is not possible, the current that flows
in any of these pins should be limited to less than 100 mA. If
separate supplies are used for the AD7707 and the system digital circuitry, the AD7707 should be powered up first. If it is not
possible to guarantee this, current limiting resistors should be
placed in series with the logic inputs to again limit the current.
Latch-up current is greater than 100 mA.
fCLK = 2.4576MHz
800
600
fCLK = 1MHz
400
200
0
2.5
3.0
3.5
4.0
VDD
4.5
5.0
5.5
Figure 17. IDD vs. Supply Voltage
–26–
REV. A
AD7707
Supply Current
The current consumption on the AD7707 is specified for supplies in the range +2.7 V to +3.3 V and in the range +4.75 V to
+5.25 V. The part operates over a +2.7 V to +5.25 V supply
range and the IDD for the part varies as the supply voltage varies
over this range. There is an internal current boost bit on the
AD7707 that is set internally in accordance with the operating
conditions. This affects the current drawn by the analog circuitry within these devices. Minimum power consumption is
achieved when the AD7707 is operated with an fCLKIN of 1 MHz
or at gains of 1 to 4 with fCLKIN = 2.4575 MHz as the internal
boost bit is off reducing the analog current consumption. Figure
17 shows the variation of the typical IDD with VDD voltage for
both a 1␣ MHz crystal oscillator and a 2.4576 MHz crystal oscillator at +25°C. The AD7707 is operated in unbuffered mode.
The relationship shows that the IDD is minimized by operating
the part with lower VDD voltages. IDD on the AD7707 is also
minimized by using an external master clock or by optimizing
external components when using the on-chip oscillator circuit.
Figures 3, 4, 6 and 7 show variations in IDD with gain, VDD and
clock frequency using an external clock.
Grounding and Layout
Since the analog inputs and reference input are differential,
most of the voltages in the analog modulator are common-mode
voltages. The excellent common-mode rejection of the part will
remove common-mode noise on these inputs. The digital filter
will provide rejection of broadband noise on the power supplies,
except at integer multiples of the modulator sampling frequency.
The digital filter also removes noise from the analog and reference inputs provided those noise sources do not saturate the
analog modulator. As a result, the AD7707 is more immune to
noise interference than a conventional high resolution converter.
However, because the resolution of the AD7707 is so high, and
the noise levels from the AD7707 so low, care must be taken
with regard to grounding and layout.
The printed circuit board that houses the AD7707 should be
designed so that the analog and digital sections are separated
and confined to certain areas of the board. This facilitates the
use of ground planes which can be separated easily. A minimum
etch technique is generally best for ground planes as it gives the
best shielding. Digital and analog ground planes should only be
joined in one place to avoid ground loops. If the AD7707 is in a
system where multiple devices require AGND-to-DGND connections, the connection should be made at one point only, a
star ground point which should be established as close as possible to the AD7707.
Avoid running digital lines under the device as these might
couple noise onto the analog circuitry within the AD7707. The
analog ground plane should be allowed to run under the AD7707
to reduce noise coupling. The power supply lines to the AD7707
should use wide traces to provide low impedance paths and
reduce the effects of glitches on the power supply line. Fast
switching signals like clocks should be shielded with digital
ground to avoid radiating noise to other sections of the board
and clock signals should never be run near the analog inputs.
Avoid crossover of digital and analog signals. Traces on opposite
sides of the board should run at right angles to each other. This
will reduce the effects of feedthrough through the board. A
microstrip technique is by far the best, but is not always possible
with a double-sided board. In this technique, the component
REV. A
side of the board is dedicated to ground planes while signals are
placed on the solder side.
Good decoupling is important when using high resolution ADCs.
All analog supplies should be decoupled with 10␣ µF tantalum in
parallel with 0.1␣ µF ceramic capacitors to GND. To achieve the
best from these decoupling components, they have to be placed
as close as possible to the device, ideally right up against the
device. All logic chips should be decoupled with 0.1␣ µF disc
ceramic capacitors to DGND.
Evaluating the AD7707 Performance
The recommended layout for the AD7707 is outlined in the
evaluation board. The evaluation board package include a fully
assembled and tested evaluation board, documentation, software for controlling the board over the printer port of a PC and
software for analyzing their performance on the PC.
Noise levels in the signals applied to the AD7707 may also
affect performance of the part. The AD7707 software evaluation
package allows the user to evaluate the true performance of the
part, independent of the analog input signal. The scheme involves using a test mode on the part where the inputs to the
AD7707 are internally shorted together to provide a zero differential voltage for the analog modulator. External to the device,
the LOCOM and HICOM inputs on the AD7707 should be
connected to voltages that are within the allowable commonmode range of the part. This scheme should be used after a
calibration has been performed on the part.
DIGITAL INTERFACE
As previously outlined, the AD7707’s programmable functions
are controlled using a set of on-chip registers. Data is written to
these registers via the part’s serial interface and read access to
the on-chip registers is also provided by this interface. All communications to the part must start with a write operation to the
Communications Register. After power-on or RESET, the device expects a write to its Communications Register. The data
written to this register determines whether the next operation to
the part is a read or a write operation and also determines to
which register this read or write operation occurs. Therefore,
write access to any of the other registers on the part starts with a
write operation to the Communications Register followed by a
write to the selected register. A read operation from any other
register on the part (including the output data register) starts
with a write operation to the Communications Register followed
by a read operation from the selected register.
The AD7707’s serial interface consists of five signals, CS,
SCLK, DIN, DOUT and DRDY. The DIN line is used for
transferring data into the on-chip registers while the DOUT line
is used for accessing data from the on-chip registers. SCLK is
the serial clock input for the device and all data transfers (either
on DIN or DOUT) take place with respect to this SCLK signal.
The DRDY line is used as a status signal to indicate when data
is ready to be read from the AD7707’s data register. DRDY
goes low when a new data word is available in the output register. It is reset high when a read operation from the data register
is complete. It also goes high prior to the updating of the output
register to indicate when not to read from the device to ensure
that a data read is not attempted while the register is being
updated. CS is used to select the device. It can be used to decode the AD7707 in systems where a number of parts are connected to the serial bus.
–27–
AD7707
Figures 18 and 19 show timing diagrams for interfacing to the
AD7707 with CS used to decode the part. Figure 17 is for a
read operation from the AD7707’s output shift register while
Figure 18 shows a write operation to the input shift register. It is
possible to read the same data twice from the output register
even though the DRDY line returns high after the first read
operation. Care must be taken, however, to ensure that the read
operations have been completed before the next output update
is about to take place.
The AD7707 serial interface can operate in three-wire mode by
tying the CS input low. In this case, the SCLK, DIN and DOUT
lines are used to communicate with the AD7707 and the status
of DRDY can be obtained by interrogating the MSB of the
Communications Register. This scheme is suitable for interfacing to microcontrollers. If CS is required as a decoding signal, it
can be generated from a port bit. For microcontroller interfaces,
it is recommended that the SCLK idles high between data
transfers.
The AD7707 can also be operated with CS used as a frame
synchronization signal. This scheme is suitable for DSP interfaces. In this case, the first bit (MSB) is effectively clocked out
by CS since CS would normally occur after the falling edge of
SCLK in DSPs. The SCLK can continue to run between data
transfers provided the timing numbers are obeyed.
The serial interface can be reset by exercising the RESET input
on the part. It can also be reset by writing a series of 1s on the
DIN input. If a Logic 1 is written to the AD7707 DIN line for
at least 32 serial clock cycles the serial interface is reset. This
ensures that in three-wire systems, if the interface gets lost either
via a software error or by some glitch in the system, it can be
reset back to a known state. This state returns the interface to
where the AD7707 is expecting a write operation to its Communications Register. This operation in itself does not reset the
contents of any registers but since the interface was lost, the
information written to any of the registers is unknown and it is
advisable to set up all registers again.
Some microprocessor or microcontroller serial interfaces have a
single serial data line. In this case, it is possible to connect the
AD7707’s DATA OUT and DATA IN lines together and connect them to the single data line of the processor. A 10 kΩ pullup resistor should be used on this single data line. In this case, if
the interface gets lost, because the read and write operations
share the same line the procedure to reset it back to a known
state is somewhat different than previously described. It requires
a read operation of 24 serial clocks followed by a write operation
where a Logic 1 is written for at least 32 serial clock cycles to
ensure that the serial interface is back into a known state.
CONFIGURING THE AD7707
The AD7707 contains six on-chip registers that the user can
accesses via the serial interface. Communication with any of
these registers is initiated by writing to the Communications
Register first. Figure 20 outlines a flow diagram of the sequence
used to configure all registers after a power-up or reset on the
AD7707. The flowchart also shows two different read options—
the first where the DRDY pin is polled to determine when an
update of the data register has taken place, the second where the
DRDY bit of the Communications Register is interrogated to
see if a data register update has taken place. Also included in the
flowing diagram is a series of words that should be written to the
registers for a particular set of operating conditions. These conditions are gain of one, no filter sync, bipolar mode, buffer off,
clock of 4.9512␣ MHz and an output rate of 50 Hz.
DRDY
t10
t3
CS
t4
t8
t6
SCLK
t9
t7
t5
DOUT
LSB
MSB
Figure 18. Read Cycle Timing Diagram
CS
t11
t16
t14
SCLK
t12
DIN
t13
t15
LSB
MSB
Figure 19. Write Cycle Timing Diagram
–28–
REV. A
AD7707
START
POWER-ON/RESET FOR AD7707
CONFIGURE & INITIALIZE mC/mP SERIAL PORT
WRITE TO COMMUNICATIONS REGISTER SELECTING
CHANNEL & SETTING UP NEXT OPERATION TO BE A
WRITE TO THE CLOCK REGISTER (20 HEX)
WRITE TO CLOCK REGISTER SETTING THE CLOCK
BITS IN ACCORDANCE WITH THE APPLIED MASTER
CLOCK SIGNAL AND SELECT UPDATE RATE FOR
SELECTED CHANNEL (0C HEX)
WRITE TO COMMUNICATIONS REGISTER SELECTING
CHANNEL & SETTING UP NEXT OPERATION TO BE A
WRITE TO THE SETUP REGISTER (10 HEX)
WRITE TO SETUP REGISTER CLEARING F SYNC,
SETTING UP GAIN, OPERATING CONDITIONS &
INITIATING A SELF-CALIBRATION ON SELECTED
CHANNEL (40 HEX)
POLL DRDY PIN
NO
WRITE TO COMMUNICATIONS REGISTER SETTING UP NEXT
OPERATION TO BE A READ FROM THE COMMUNICATIONS
REGISTER (08 HEX)
DRDY
LOW?
YES
READ FROM COMMUNICATIONS REGISTER
WRITE TO COMMUNICATIONS REGISTER SETTING UP
NEXT OPERATION TO BE A READ FROM THE DATA
REGISTER (38 HEX)
POLL DRDY BIT OF COMMUNICATIONS REGISTER
READ FROM DATA REGISTER
NO
DRDY
LOW?
YES
WRITE TO COMMUNICATIONS REGISTER SETTING UP
NEXT OPERATION TO BE A READ FROM THE DATA
REGISTER (38 HEX)
READ FROM DATA REGISTER
Figure 20. Flowchart for Setting Up and Reading from the AD7707
REV. A
–29–
AD7707
MICROCOMPUTER/MICROPROCESSOR INTERFACING
VDD
The AD7707’s flexible serial interface allows for easy interface
to most microcomputers and microprocessors. The flowchart of
Figure 20 outlines the sequence that should be followed when
interfacing a microcontroller or microprocessor to the AD7707.
Figures 21, 22 and 23 show some typical interface circuits.
VDD
RESET
68HC11
The serial interface on the AD7707 is capable of operating from
just three wires and is compatible with SPI interface protocols.
The three-wire operation makes the part ideal for isolated systems where minimizing the number of interface lines minimizes
the number of optoisolators required in the system. The serial
clock input is a Schmitt triggered input to accommodate slow
edges from optocouplers. The rise and fall times of other digital
inputs to the AD7707 should be no longer than 1␣ µs.
Most of the registers on the AD7707 are 8-bit registers, which
facilitates easy interfacing to the 8-bit serial ports of microcontrollers. The Data Register on the AD7707 is 16␣ bits, and the
offset and gain registers are 24-bit registers but data transfers to
these registers can consist of multiple 8-bit transfers to the
serial port of the microcontroller. DSP processors and microprocessors generally transfer 16 bits of data in a serial data
operation. Some of these processors, such as the ADSP-2105,
have the facility to program the amount of cycles in a serial
transfer. This allows the user to tailor the number of bits in any
transfer to match the register length of the required register in
the AD7707.
AD7707
SS
SCK
SCLK
MISO
DATA OUT
MOSI
DATA IN
CS
Figure 21. AD7707 to 68HC11 Interface
Even though some of the registers on the AD7707 are only
eight bits in length, communicating with two of these registers
in successive write operations can be handled as a single 16-bit
data transfer if required. For example, if the Setup Register is to
be updated, the processor must first write to the Communications Register (saying that the next operation is a write to the
Setup Register) and then write eight bits to the Setup Register.
If required, this can all be done in a single 16-bit transfer because once the eight serial clocks of the write operation to the
Communications Register have been completed, the part immediately sets itself up for a write operation to the Setup Register.
The 68HC11 is configured in the master mode with its CPOL
bit set to a logic one and its CPHA bit set to a logic one. When
the 68HC11 is configured like this, its SCLK line idles high
between data transfers. The AD7707 is not capable of full
duplex operation. If the AD7707 is configured for a write operation, no data appears on the DATA OUT lines even when
the SCLK input is active. Similarly, if the AD7707 is configured for a read operation, data presented to the part on the
DATA IN line is ignored even when SCLK is active.
Coding for an interface between the 68HC11 and the
AD7707 is given in Table XV. In this example, the DRDY
output line of the AD7707 is connected to the PC0 port bit of
the 68HC11 and is polled to determine its status.
AD7707 to 68HC11 Interface
Figure 21 shows an interface between the AD7707 and the
68HC11 microcontroller. The diagram shows the minimum
(three-wire) interface with CS on the AD7707 hard-wired low.
In this scheme, the DRDY bit of the Communications Register
is monitored to determine when the Data Register is updated.
An alternative scheme, which increases the number of interface
lines to four, is to monitor the DRDY output line from the
AD7707. The monitoring of the DRDY line can be done in two
ways. First, DRDY can be connected to one of the 68HC11’s
port bits (such as PC0), which is configured as an input. This
port bit is then polled to determine the status of DRDY. The
second scheme is to use an interrupt driven system, in which
case the DRDY output is connected to the IRQ input of the
68HC11. For interfaces that require control of the CS input
on the AD7707, one of the port bits of the 68HC11 (such as
PC1), which is configured as an output, can be used to drive
the CS input.
–30–
VDD
8XC51
VDD
P3.0
AD7707
RESET
DATA OUT
DATA IN
P3.1
SCLK
CS
Figure 22. AD7707 to 8XC51 Interface
REV. A
AD7707
AD7707 to 8051 Interface
AD7707 to ADSP-2103/ADSP-2105 Interface
An interface circuit between the AD7707 and the 8XC51
microcontroller is shown in Figure 22. The diagram shows the
minimum number of interface connections with CS on the
AD7707 hard-wired low. In the case of the 8XC51 interface the
minimum number of interconnects is just two. In this scheme,
the DRDY bit of the Communications Register is monitored to
determine when the Data Register is updated. The alternative
scheme, which increases the number of interface lines to three,
is to monitor the DRDY output line from the AD7707. The
monitoring of the DRDY line can be done in two ways. First,
DRDY can be connected to one of the 8XC51’s port bits (such
as P1.0) which is configured as an input. This port bit is then
polled to determine the status of DRDY. The second scheme is
to use an interrupt-driven system, in which case the DRDY
output is connected to the INT1 input of the 8XC51. For interfaces that require control of the CS input on the AD7707, one
of the port bits of the 8XC51 (such as P1.1), which is configured as an output, can be used to drive the CS input. The
8XC51 is configured in its Mode 0 serial interface mode. Its
serial interface contains a single data line. As a result, the
DATA OUT and DATA IN pins of the AD7707 should be
connected together with a 10 kΩ pull-up resistor. The serial
clock on the 8XC51 idles high between data transfers. The
8XC51 outputs the LSB first in a write operation, while the
AD7707 expects the MSB first so the data to be transmitted has
to be rearranged before being written to the output serial register. Similarly, the AD7707 outputs the MSB first during a read
operation while the 8XC51 expects the LSB first. Therefore, the
data read into the serial buffer needs to be rearranged before
the correct data word from the AD7707 is available in the
accumulator.
Figure 23 shows an interface between the AD7707 and the
ADSP-2103/ADSP-2105 DSP processor. In the interface
shown, the DRDY bit of the Communications Register is again
monitored to determine when the Data Register is updated. The
alternative scheme is to use an interrupt-driven system, in which
case the DRDY output is connected to the IRQ2 input of the
ADSP-2103/ADSP-2105. The serial interface of the ADSP2103/ADSP-2105 is set up for alternate framing mode. The
RFS and TFS pins of the ADSP-2103/ADSP-2105 are configured as active low outputs and the ADSP-2103/ADSP-2105
serial clock line, SCLK, is also configured as an output. The CS
for the AD7707 is active when either the RFS or TFS outputs
from the ADSP-2103/ADSP-2105 are active. The serial clock
rate on the ADSP-2103/ADSP-2105 should be limited to
3␣ MHz to ensure correct operation with the AD7707.
VDD
ADSP-2103/
ADSP-2105
AD7707
RESET
RFS
CS
TFS
DR
DATA OUT
DT
DATA IN
CODE FOR SETTING UP THE AD7707
Table XVII gives a set of read and write routines in C code for
interfacing the 68HC11 microcontroller to the AD7707. The
sample program sets up the various registers on the AD7707
and reads 1000 samples from the part into the 68HC11. The
setup conditions on the part are exactly the same as those outlined for the flowchart of Figure 20. In the example code given
here, the DRDY output is polled to determine if a new valid
word is available in the data register.
The sequence of the events in this program are as follows:
1. Write to the Communications Register, selecting channel one
as the active channel and setting the next operation to be a
write to the clock register.
2. Write to Clock Register setting the CLK DIV bit which
divides the external clock internally by two. This assumes
that the external crystal is 4.9512 MHz. The update rate is
selected to be 50 Hz.
3. Write to Communication Register selecting Channel 1 as the
active channel and setting the next operation to be a write to
the Setup Register.
4. Write to the Setup Register, setting the gain to 1, setting
bipolar mode, buffer off, clearing the filter synchronization
and initiating a self-calibration.
5. Poll the DRDY output.
SCLK
SCLK
6. Read the data from the Data Register.
7. Loop around doing Steps 5 and 6 until the specified number
of samples have been taken from the selected channel.
Figure 23. AD7707 to ADSP-2103/ADSP-2105 Interface
REV. A
–31–
AD7707
Table XVIII. C Code for Interfacing AD7707 to 68HC11
/* This program has read and write routines for the 68HC11 to interface to the AD7707 and the sample program sets the various
registers and then reads 1000 samples from one channel. */
#include <math.h>
#include <io6811.h>
#define NUM_SAMPLES 1000 /* change the number of data samples */
#define MAX_REG_LENGTH 2 /* this says that the max length of a register is 2 bytes */
Writetoreg (int);
Read (int,char);
char *datapointer = store;
char store[NUM_SAMPLES*MAX_REG_LENGTH + 30];
void main ()
{
/* the only pin that is programmed here from the 68HC11 is the /CS and this is why the PC2 bit of PORTC is made as an output */
char a;
DDRC = 0x04; /* PC2 is an output the rest of the port bits are inputs */
PORTC | = 0x04; /* make the /CS line high */
Writetoreg (0x20); /* Active Channel is AIN1/LOCOM, next operation as write to the clock register */
Writetoreg (0x18); /* master clock enabled, 4.9512MHz Clock, set output rate to 50Hz*/
Writetoreg (0x10); /* Active Channel is AIN1/LOCOM, next operation as write to the setup register */
Writetoreg (0x40); /* gain = 1, bipolar mode, buffer off, clear FSYNC and perform a Self Calibration*/
while (PORTC and 0x10); /* wait for /DRDY to go low */
for (a=0;a<NUM_SAMPLES;a++);
{
Writetoreg (0x38); /*set the next operation for 16 bit read from the data register */
Read (NUM_SAMPES,2);
}
}
Writetoreg (int byteword);
{
int q;
SPCR = 0x3f;
SPCR = 0X7f; /* this sets the WiredOR mode (DWOM=1), Master mode (MSTR=1), SCK idles high (CPOL=1), /SS can be low
always (CPHA=1), lowest clock speed (slowest speed which is master clock /32 */
DDRD = 0x18; /* SCK, MOSI outputs */
q = SPSR;
q = SPDR; /* the read of the staus register and of the data register is needed to clear the interrupt which tells the user that the data
transfer is complete */
PORTC &= 0xfb; /* /CS is low */
SPDR = byteword; /* put the byte into data register */
while (! (SPSR & 0x80)); /* wait for /DRDY to go low */
PORTC |= 0x4; /* /CS high */
}
Read (int amount, int reglength)
{
int q;
SPCR = 0x3f;
SPCR = 0x7f; /* clear the interupt */
DDRD = 0x10; /* MOSI output, MISO input, SCK output */
while (PORTC & 0x10); /* wait for /DRDY to go low */
PORTC & 0xfb ; /* /CS is low */
for (b=0;b<reglength;b++)
{
SPDR = 0;
while (! (SPSR & 0x80)); /* wait until port ready before reading */
*datapointer++=SPDR; /* read SPDR into store array via datapointer */
}
PORTC|=4; /* /CS is high */
}
–32–
REV. A
AD7707
signals up to ± 10 V in amplitude. This application shows the
high level input channel being used to convert a number of
input signals provided through an external mux controlled by
the system microcontroller. Switching channels on the external
multiplexer is equivalent to providing a step change on the
AIN3 input. It takes 3 or 4 updates before the correct output
code corresponding to the new analog input appears at the
output. Therefore, when switching between channels on the
external mux the first three outputs should be ignored following
the channel change or the FSYNC bit in the setup register
should be used to reset the digital filter and ensure that the
DRDY is set high until a valid result is available in the output
register.
APPLICATIONS
The AD7707 provides a low cost, high resolution analog-todigital function with two low level input channels and one high
level input channel. Because the analog-to-digital function is
provided by a sigma-delta architecture, it makes the part more
immune to noisy environments thus making the part ideal for
use in industrial and process control applications. It also provides
a programmable gain amplifier, a digital filter and calibration
options. Thus, it provides far more system level functionality
than off-the-shelf integrating ADCs without the disadvantage of
having to supply a high quality integrating capacitor. In addition, using the AD7707 in a system allows the system designer
to achieve a much higher level of resolution because noise performance of the AD7707 is better than that of the integrating
ADCs.
Smart Valve/Actuator Control
Another area where the low power, single supply and high voltage input capability is of benefit is in smart valve and and actuator control circuits. The AD7707 monitors the signal from the
control valve. The controller and the AD7707 form a closedloop control circuit. Figure 25 shows a block diagram of a smart
actual or control circuit which includes the AD7707. The AD7707
monitors the valve position via a high quality servo pot whose
output is ± 10 V.
The on-chip PGA allows the AD7707 to handle an analog input
voltage ranges as low as 10 mV full-scale with VREF = +1.25 V.
The pseudo-differential input capability of the low level channel
allows this analog input range to have an absolute value anywhere between AGND – 100 mV and AVDD + 30 mV when the
part is operated in unbuffered mode. It allows the user to connect the transducer directly to the input of the AD7707.
In addition, the three-wire digital interface on the AD7707
allows data acquisition front ends to be isolated with just three
wires. The AD7707 can be operated from a single 3 V or 5 V
and its low power operation ensures that very little power needs
to be brought across the isolation barrier in an isolated application.
Similar applications for the AD7707 would be in the area of
smart transmitters. Here, the entire smart transmitter must
operate from the 4 mA to 20 mA loop.
Tolerances in the loop mean that the amount of current available to power the transmitter is as low as 3.5 mA. The AD7707
consumes only 280 µA, leaving at least 3 mA available for the
rest of the transmitter. Figure 26 shows a block diagram of a
smart transmitter which includes the AD7707.
Data Acquisition
Figure 24 shows a data acquisition system showing in which the
low level input channel is used to digitize signals from a thermocouple and the high level input channel converts process control
+5V
+5V
CJC
AD590
AVDD
DVDD
AIN2
MCLK IN
8.2kV
AIN1
THERMOCOUPLE
JUNCTION
LOCOM
MCLK OUT
+15V
AD7707
610V INPUT
0V TO 10V INPUT
65V INPUT
0V TO 5V INPUT
4-20mA
IN1
IN2
IN3
IN4
IN5
IN6
250V
OUT
AIN3
MICROCONTROLLER
REF IN(+)
2.5V
VBIAS
HICOM
REF IN(–)
ANALOG
MULTIPLEXER
250V
0-20mA
VDD
AGND
DGND
VSS
A0
A1
SCLK
CS
DIN
DOUT
A2
–15V
Figure 24. Data Acquisition System Using the AD7707
REV. A
–33–
SCLK
PO
DOUT
DIN
P1 P2 P3
AD7707
+5V
SMART
VALVE/ACTUATOR
AVDD
DVDD
MCLK IN
AIN2
ACTUATOR/
VALVE
AD7707
610V
AIN3
MCLK OUT
REF IN(+)
VBIAS
2.5V
HICOM
REF IN(–)
MICROCONTROLLER
AGND
DGND
AIN1
AD420
DAC
RSENSE
LOCOM
CONTROL
ROOM
Figure 25. Smart Valve/Actuator Control Using the AD7707
MAIN TRANSMITTER ASSEMBLY
3V
DN25D
2.2mF 0.1mF
1.25V
4.7mF
AVDD AVDD REF IN
AD7707
SENSORS
RTD
mV
AIN1
THERMOCOUPLE
AIN2
6V
AIN3
MCLK IN
100kV
VCC
VCC
COMP
REF OUT2
DRIVE
MICROCONTROLLER UNIT
•PID
•RANGE SETTING
•CALIBRATION
•LINEARIZATION
•OUTPUT CONTROL
•SERIAL COMMUNICATION
•HART PROTOCOL
REF IN
0.01mF
4mA
TO
20mA
1kV
1000pF
4.7mF
AD421
C1 C2 C3
LOOP
RTN
COM
COM
MCLK OUT
0.01mF
AGND
BOOST
REF OUT1
0.0033mF
DGND
0.01mF
Figure 26. Smart Transmitter Using the AD7707
–34–
REV. A
AD7707
Pressure Measurement
Thermocouple Measurement
Other typical applications for the AD7707 include temperature
and pressure measurement. Figure 27 shows the AD7707 used
with a pressure transducer, the BP01 from Sensym. The
pressure transducer is arranged in a bridge network and gives a
differential output voltage between its OUT(+) and OUT(–)
terminals. With rated full-scale pressure (in this case 300␣ mmHg)
on the transducer, the differential output voltage is 3␣ mV/V of
the input voltage (i.e., the voltage between its IN(+) and IN(–)
terminals). Assuming a 5␣ V excitation voltage, the full-scale
output range from the transducer is 15 mV. The low level input
channels are ideal for this type of low signal measurement application. The excitation voltage for the bridge is also used to
generate the reference voltage for the AD7707. Therefore,
variations in the excitation voltage do not introduce errors in
the system. Choosing resistor values of 24 kΩ and 15 kΩ, as
per Figure 27, gives a 1.92 V reference voltage for the AD7707
when the excitation voltage is 5␣ V.
Another application area for the AD7707 is in temperature
measurement. Figure 28 outlines a connection from a thermocouple to the AD7707. In this application, the AD7707 is operated in its unbuffered mode to accommodate signals of ±100 mV
on the front end. Cold conjunction compensation is implemented using the AD590 temperature transducer that produces
an output current proportional to absolute temperature.
Using the part with a programmed gain of 128 results in the
full-scale input span of the AD7707 being 15␣ mV, which corresponds with the output span from the transducer.
+5V
+5V
CJC
AD590
AVDD
DVDD
AIN2
THERMOCOUPLE
JUNCTION
MCLK IN
8.2kV
AIN1
LOCOM
+5V
AD7707
REF192
GND
OUT
MCLK OUT
REF IN(+)
REF IN(–) DRDY
SCLK
+5V
EXCITATION VOLTAGE = +5V
AVDD
IN(+)
DVDD
LOCOM
24kV
AD7707
MCLK OUT
REF IN(+)
15kV
REF IN(–) DRDY
SCLK
AGND
DGND
DIN
CONTROLLER
DOUT
CS
Figure 27. Pressure Measurement Using the AD7707
REV. A
DGND
DOUT
CONTROLLER
Figure 28. Thermocouple Measurement Using the
AD7707
AIN1
IN(–)
DIN
CS
MCLK IN
OUT(+)
OUT(–)
AGND
–35–
AD7707
RTD Measurement
Chart Recorders
Figure 29 shows another temperature measurement application
for the AD7707. In this case, the transducer is an RTD (Resistive Temperature Device), a PT100. The arrangement is a 4lead RTD configuration. There are voltage drops across the lead
resistances RL1 and RL4 but these simply shift the commonmode voltage. There is no voltage drop across lead resistances
RL2 and RL3 as the input current to the AD7707 is very low. The
lead resistances present a small source impedance so it would
not generally be necessary to turn on the buffer on the AD7707.
Another area where both high and low level input channels are
usually found is in chart recorder applications. Circular chart
recorders generally have two requirements. The first which
would utilize the low level input channels of the AD7707 to
measure inputs from thermocouples, RTDs and pressure sensors. The second requirement is to be able to measure dc input
voltage ranges up to ± 10 V. The high level input channel is
ideally suited to this measurement as there is no external signal
conditioning required to accommodate these high level input
signals.
If the buffer is required, the common-mode voltage should be
set accordingly by inserting a small resistance between the bottom end of the RTD and GND of the AD7707. In the application shown, an external 400␣ µA current source provides the
excitation current for the PT100 and also generates the reference voltage for the AD7707 via the 6.25 kΩ resistor. Variations
in the excitation current do not affect the circuit as both the
input voltage and the reference voltage vary radiometrically with
the excitation current. However, the 6.25 kΩ resistor must have
a low temperature coefficient to avoid errors in the reference
voltage over temperature.
+5V
400mA
AVDD
DVDD
REF IN(+)
MCLK IN
REF IN(+)
6.25kV
REF IN(–)
RL1
RL2
REF IN(–)
MCLK OUT
AIN1
AD7707
RTD
RL3
RL4
LOCOM
DRDY
SCLK
AGND
DIN
DGND
DOUT
CONTROLLER
CS
Figure 29. RTD Measurement Using the AD7707
–36–
REV. A
AD7707
Accommodating Various High Level Input Ranges
The high level input channel, AIN3 can accommodate input signals from –11 V to +30 V on its input. This is achieved using
on-chip thin film resistors that map the signal on AIN3 into a usable range for the AD7707. The input structure is arranged
so that the sigma-delta converter sees the same impedance at its AIN(+) and AIN(–) inputs. The signal on the AIN3 input is
referenced to the HICOM input and the VBIAS signal is used to adjust the common-mode voltage at the modulator input.
In normal 5 V operation VBIAS is normally connected to 2.5 V and HICOM is connected to AGND. This arrangement
ensures that the voltages seen at the modulator input are within the common-mode range of the buffer.
The differential voltage, AIN, seen by the AD7707 when using the high level input channel is the difference between AIN3(+)
and AIN3(–) as shown in Figure 30 and must remain within the absolute common-mode range of the modulator.
AIN3(+) = (AIN3 + 6 × VBIAS+ V (HICOM))/8
AIN3(–) = 0.75 × VBIAS + 0.25 V (HICOM)
AIN = (AIN3 – V (HICOM))/8
30kV
AIN3
5kV
AIN3(+)
VBIAS
AIN
5kV
MUX
AIN3(–)
15kV
30kV
HICOM
Figure 30. AIN3, High Level Input Channel Structure
The VBIAS and HICOM inputs are used to tailor the input range on the high level input channel to suit a variety of input
ranges. The following table applies for operation with AVDD = 5 V, and REF(+) – REF(–) = 2.5 V.
Table XIX
AIN3 RANGE
VBIAS
HICOM
GAIN
BUF/UNBUF
AIN RANGE
± 10 V
±5 V
0 V to 10 V
0 V to 20 V
2.5 V
2.5 V
2.5 V
AGND
2.5 V
2.5 V
AGND
AGND
AGND
AGND
AGND
2.5 V
2
4
2
1
1
2
BUF/UNBUF
BUF/UNBUF
BUF/UNBUF
UNBUF
UNBUF
BUF/UNBUF
1.875 V ± 1.25 V
1.875 V ± 0.625 V
1.875 V to 3.125 V
0 V to 2.5 V
1.875 V to 4.375 V
2.5 V ± 0.9375 V
–5 V to +10 V
The following table applies for operation with AVDD = 3 V, and REF(+) – REF(–) = 1.25 V.
Table XX
AIN3 RANGE
VBIAS
HICOM
GAIN
BUF/UNBUF
AIN RANGE
±5 V
0 V to 10 V
–5 V to +10 V
–7.5 V to +10 V
± 10 V
1.25 V
1.25 V
1.25 V
1.25 V
1.666 V
AGND
AGND
2.5 V
0
AGND
2
1
1
1
1
UNBUF
UNBUF
UNBUF
UNBUF
UNBUF
0.9375 V ± 0.625 V
0.9375 V to 2.1875 V
1.5625 V ± 0.9375 V
0 V to 2.1875 V
1.25 V ± 1.25 V
Typical Input Currents
When using the high level input channel, power dissipation is determined by the currents flowing in the AIN3, VBIAS and
HICOM inputs. The voltage level applied to these inputs determines whether the external source driving these inputs needs
to sink of source current. The following table shows the currents associated with the ± 10 V input range. These inputs should
be driven from a low impedance source in all applications to prevent significant gain errors being introduced.
Table XXI
AIN3
VBIAS
HICOM
I (AIN3)
I (VBIAS)
I (HICOM)
–10 V
0V
+10 V
2.5 V
2.5 V
2.5 V
AGND
AGND
AGND
–354 µA
–62 µA
229 µA
500 µA
250 µA
0
–146 µA
–188 µA
–229 µA
REV. A
–37–
AD7707
APPENDIX 1. OUTPUT NOISE FOR HIGH LEVEL INPUT CHANNEL, AIN3 (5 V OPERATION)
Specified high level input voltage ranges of ± 10 V, ± 5 V, 0 V to +10 V and 0 V to +5 V only utilize two gain different gain
settings (gains of 2 and 4) out of the eight possible settings available within the PGA. The tables here show what the high
level channel performance actually is over the complete range of gain settings. Table XXII shows the AD7707 output rms
noise and peak-to-peak resolution for the selectable notch and –3␣ dB frequencies for the part, as selected by FS0, FS1 and
FS2 of the Clock Register. The numbers are given for all input ranges with a VREF of +2.5 V, HBIAS = 2.5 V, HICOM =
AGND and AVDD = 5 V. These numbers are typical and are generated at an analog input voltage of 0 V for buffered mode of
operation. Table XXIII meanwhile shows the rms and peak-to-peak resolution for buffered mode of operation. It is important
to note that these numbers represent the resolution for which there will be no code flicker. They are not calculated based on rms noise but
on peak-to-peak noise. The output noise comes from two sources. The first is the electrical noise in the semiconductor devices
(device noise) used in the implementation of the modulator. Secondly, when the analog input is converted into the digital
domain, quantization noise is added. The device noise is at a low level and is independent of frequency. The quantization
noise starts at an even lower level but rises rapidly with increasing frequency to become the dominant noise source. The
numbers in the tables are given for the bipolar input ranges. For the unipolar ranges the rms noise numbers will be the same
as the bipolar range but the peak-to-peak resolution is now based on half the signal range which effectively means losing 1
bit of resolution.
Table XXII. AIN3, Output RMS Noise/Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +5 V
Unbuffered Mode
Filter First
Notch and O/P
Data Rate
–3␣ dB
Gain of
Frequency 1
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
10.90 (16)
31.34 (16)
36.74 (16)
690 (13)
4679 (10)
Typical Output RMS Noise in ␮V (Peak-to-Peak Resolution)
Gain of
Gain of
Gain of
Gain of
Gain of
Gain of
2
4
8
16
32
64
Gain of
128
5.10 (16)
15.82 (16)
20.36 (16)
430 (13)
2350 (10)
2.30 (14)
4.75 (13)
5.3 (13)
13.8 (12)
53 (10)
3.52 (16)
9.77 (16)
12.29 (16)
212 (13)
1287 (10)
2.62 (16)
6.00 (16)
7.33 (16)
100 (13)
564 (10)
2.34 (16)
5.12 (16)
5.84 (16)
42 (13)
294 (10)
2.34 (16)
5.36 (15)
5.65 (15)
30 (13)
137 (10)
2.34 (15)
4.84 (14)
5.1 (14)
18.5 (12)
73 (10)
Table XXIII. AIN3, Output RMS Noise/Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +5 V
Buffered Mode
Filter First
Notch and O/P
Data Rate
–3␣ dB
Gain of
Frequency 1
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
14.28 (16)
37.4 (16)
48.8 (16)
778 (12.5)
4716 (10.5)
Typical Output RMS Noise in ␮V (Peak-to-Peak Resolution)
Gain of
Gain of
Gain of
Gain of
Gain of
Gain of
2
4
8
16
32
64
Gain of
128
7.4 (16)
22.2 (16)
26.6 (16)
475 (13)
2423 (10.5)
2.34 (14.5)
7.5 (12.5)
8.1 (12.5)
18.3 (11.5)
49 (10)
5.2 (16)
14.3 (16)
15.88 (16)
187 (13)
1097 (10.5)
3.35 (16)
8.7 (16)
10.17 (16)
98 (13)
551 (10.5)
3.35 (16)
7.33 (15.5)
8.78 (15.5)
60 (12.5)
288 (10.5)
3.34 (15.5)
7.7 (14.5)
8.1 (14.5)
31.7 (12.5)
150 (10)
3.34 (15)
7.6 (13.5)
8.1 (13.5)
23 (12)
81 (10)
Output Noise For High Level Input Channel, AIN3 (3 V Operation)
Table XXIV shows the AD7707 output rms noise and peak-to-peak resolution for the selectable notch and –3␣ dB frequencies for the part, as selected by FS0, FS1 and FS2 of the Clock Register. The numbers are given for all input ranges with a
VREF of +1.25 V, HBIAS = 1.25 V, HICOM = AGND and AVDD = 3 V. These numbers are typical and are generated at an
analog input voltage of 0 V for unbuffered mode of operation. Table XXV meanwhile shows the output rms noise and peak-topeak resolution for buffered mode of operation with the same operating conditions as above. It is important to note that these
numbers represent the resolution for which there will be no code flicker. They are not calculated based on rms noise but on peak-to-peak
noise. The output noise comes from two sources. The first is the electrical noise in the semiconductor devices (device noise)
used in the implementation of the modulator. Secondly, when the analog input is converted into the digital domain, quantization noise is added. The device noise is at a low level and is independent of frequency. The quantization noise starts at an
even lower level but rises rapidly with increasing frequency to become the dominant noise source. The numbers in the tables
are given for the bipolar input ranges. For the unipolar ranges the rms noise numbers will be the same as the bipolar range
but the peak-to-peak resolution is now based on half the signal range which effectively means losing 1 bit of resolution.
–38–
REV. A
AD7707
Table XXIV. AIN3, Output RMS Noise/Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +3 V
Unbuffered Mode
Filter First
Notch and O/P –3␣ dB
Gain of
Data Rate
Frequency 1
Typical Output RMS Noise in ␮V (Peak-to-Peak Resolution)
Gain of
Gain of
Gain of
Gain of
Gain of
Gain of
2
4
8
16
32
64
Gain of
128
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
7.02 (16)
16.4 (16)
19.13 (16)
204 (13)
1151 (10.5)
2.13 (13.5)
5.09 (12)
6.14 (12)
11.42 (11)
27.5 (9.5)
12.4 (16)
30.35 (16)
34.55 (16)
498 (13)
2266 (10.5)
3.87 (16)
9.4 (16)
10.9 (16)
105 (13)
554 (10.5)
2.41 (16)
5.85 (16)
6 (16)
57.5 (13)
280 (10.5)
2.39 (16)
5.2 (15)
5.8 (15)
27.5 (13)
136 (10.5)
2.3 (15.5)
4.5 (14.5)
5.62 (14)
17.4 (12.5)
83 (10)
2.29 (14.5)
4.5 (13.5)
5.2 (13)
12.7 (12)
39 (10)
Table XXV. AIN3, Output RMS Noise/Peak-to-Peak Resolution vs. Gain and Output Update Rate @ +3 V
Buffered Mode
Filter First
Notch and O/P –3␣ dB
Gain of
Data Rate
Frequency 1
Typical Output RMS Noise in ␮V (Peak-to-Peak Resolution)
Gain of
Gain of
Gain of
Gain of
Gain of
Gain of
2
4
8
16
32
64
Gain of
128
MCLK IN = 2.4576 MHz
10␣ Hz
2.62␣ Hz
50␣ Hz
13.1␣ Hz
60␣ Hz
15.72␣ Hz
250␣ Hz
65.5␣ Hz
500␣ Hz
131␣ Hz
8.39 (16)
18.8 (16)
21.55 (16)
194 (13)
1231 (10.5)
3.3 (13)
7 (12)
7.7 (12)
16.7 (10.5)
31 (9.5)
REV. A
14.84 (16)
36.1 (16)
38.8 (16)
420 (13)
2234 (10.5)
5.56 (16)
11.5 (16)
13.39 (16)
97.6 (13)
534 (10.5)
–39–
3.45 (16)
7.5 (15.5)
8.5 (15.5)
54.5 (12.5)
275 (10.5)
3.3 (16)
7.4 (14.5)
8.36 (14.5)
30 (12.5)
145 (10.5)
3.2 (15)
7.43 (13.5)
8 (13.5)
22 (12)
71 (10.5)
3.2 (14)
6.8 (12.5)
8.2 (12.5)
18 (11.5)
48 (10)
AD7707
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
C3580a–2.5–2/00 (rev. A)
20-Lead SOIC
(R-20)
0.5118 (13.00)
0.4961 (12.60)
20
11
0.2992 (7.60)
0.2914 (7.40)
1
0.4193 (10.65)
0.3937 (10.00)
10
PIN 1
0.1043 (2.65)
0.0926 (2.35)
0.0118 (0.30) 0.0500
0.0040 (0.10) (1.27)
BSC
0.0291 (0.74)
3 458
0.0098 (0.25)
88
08
0.0192 (0.49) SEATING
0.0125 (0.32)
PLANE
0.0138 (0.35)
0.0091 (0.23)
0.0500 (1.27)
0.0157 (0.40)
20-Lead TSSOP
(RU-20)
0.260 (6.60)
0.252 (6.40)
20
11
0.177 (4.50)
0.169 (4.30)
0.256 (6.50)
0.246 (6.25)
1
10
PIN 1
SEATING
PLANE
0.0433 (1.10)
MAX
0.0256 (0.65) 0.0118 (0.30)
BSC
0.0075 (0.19)
0.0079 (0.20)
0.0035 (0.090)
88
08
0.028 (0.70)
0.020 (0.50)
PRINTED IN U.S.A.
0.006 (0.15)
0.002 (0.05)
–40–
REV. A