AD AD8304ARUZ 160 db range (100 pa -10 ma) logarithmic converter Datasheet

a
FEATURES
Optimized for Fiber Optic Photodiode Interfacing
Eight Full Decades of Range
Law Conformance 0.1 dB from 1 nA to 1 mA
Single-Supply Operation (3.0 V– 5.5 V)
Complete and Temperature Stable
Accurate Laser-Trimmed Scaling:
Logarithmic Slope of 10 mV/dB (at VLOG Pin)
Basic Logarithmic Intercept at 100 pA
Easy Adjustment of Slope and Intercept
Output Bandwidth of 10 MHz, 15 V/s Slew Rate
1-, 2-, or 3-Pole Low-Pass Filtering at Output
Miniature 14-Lead Package (TSSOP)
Low Power: ~4.5 mA Quiescent Current (Enabled)
APPLICATIONS
High Accuracy Optical Power Measurement
Wide Range Baseband Log Compression
Versatile Detector for APC Loops
PRODUCT DESCRIPTION
The AD8304 is a monolithic logarithmic detector optimized for
the measurement of low frequency signal power in fiber optic
systems. It uses an advanced translinear technique to provide an
exceptionally large dynamic range in a versatile and easily used
form. Its wide measurement range and accuracy are achieved
using proprietary design techniques and precise laser trimming.
In most applications only a single positive supply, VP, of 5 V
will be required, but 3.0 V to 5.5 V can be used, and certain
applications benefit from the added use of a negative supply,
VN. When using low supply voltages, the log slope is readily
altered to fit the available span. The low quiescent current and
chip disable features facilitate use in battery-operated applications.
The input current, IPD , flows in the collector of an optimally
scaled NPN transistor, connected in a feedback path around a
low offset JFET amplifier. The current-summing input node
operates at a constant voltage, independent of current, with a
default value of 0.5 V; this may be adjusted over a wide range,
including ground or below, using an optional negative supply.
An adaptive biasing scheme is provided for reducing the dark
current at very low light input levels. The voltage at Pin VPDB
applies approximately 0.1 V across the diode for IPD = 100 pA,
rising linearly with current to 2.0 V of net bias at IPD = 10 mA.
The input pin INPT is flanked by the guard pins VSUM that
track the voltage at the summing node to minimize leakage.
160 dB Range (100 pA –10 mA)
Logarithmic Converter
AD8304
FUNCTIONAL BLOCK DIAGRAM
VPS2
PWDN
VPS1
10
2
12
PDB
BIAS
VREF
AD8304
7
VREF
8
VLOG
9
BFIN
13
BFNG
VPDB
6
3
IPD
4
0.5V
~10k
VSUM
INPT
TEMPERATURE
COMPENSATION
VSUM 5
5k
1
14
11
VNEG
ACOM
VOUT
The default value of the logarithmic slope at the output VLOG is
accurately scaled to 10 mV/dB (200 mV/decade). The resistance
at this output is laser-trimmed to 5 kΩ, allowing the slope to be
lowered by shunting it with an external resistance; the addition
of a capacitor at this pin provides a simple low-pass filter. The
intermediate voltage VLOG is buffered in an output stage that can
swing to within about 100 mV of ground (or VN) and the positive supply, VP, and provides a peak current drive capacity of
± 20 mA. The slope can be increased using the buffer and a pair
of external feedback resistors. An accurate voltage reference of
2 V is also provided to facilitate the repositioning of the intercept.
Many operational modes are possible. For example, low-pass filters
of up to three poles may be implemented, to reduce the output
noise at low input currents. The buffer may also serve as a comparator, with or without hysteresis, using the 2 V reference, for
example, in alarm applications. The incremental bandwidth of
a translinear logarithmic amplifier inherently diminishes for small
input currents. At the 1 nA level, the AD8304’s bandwidth is
about 2 kHz, but this increases in proportion to IPD up to a
maximum value of 10 MHz.
The AD8304 is available in a 14-lead TSSOP package and specified
for operation from –40°C to +85°C.
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 2002
AD8304–SPECIFICATIONS (V = 5 V, V = 0 V, T = 25C, unless otherwise noted.)
P
N
A
Parameter
Conditions
Min1
INPUT INTERFACE
Specified Current Range
Pin 4, INPT; Pin 3 and Pin 5, VSUM
Flows toward INPT Pin
100
Input Node Voltage
Temperature Drift
Input Guard Offset Voltage
Internally preset; may be altered
–40°C < TA < +85°C
VIN – VSUM
–20
PHOTODIODE BIAS
Minimum Value
Transresistance
Established between Pin 6, VPDB, and Pin 4
IPD = 100 pA
70
100
200
LOGARITHMIC OUTPUT
Slope
Pin 8, VLOG
Laser-trimmed at 25°C
0°C < TA < 70°C
Laser-trimmed at 25°C
0°C < TA < 70°C
10 nA < IPD < 1 mA, Peak Error
1 nA < IPD < 1 mA, Peak Error
196
194
60
35
200
2
Intercept
Law Conformance Error
Maximum Output Voltage
Minimum Output Voltage
Output Resistance
REFERENCE OUTPUT
Voltage WRT Ground
0.46
Typ
Limited by VN = 0 V
Laser-trimmed at 25°C
4.95
Pin 7, VREF
Laser-trimmed at 25°C
–40°C < TA < +85°C
1.98
1.92
Output Resistance
0.5
0.02
Max1
10
0.54
+20
100
0.05
0.1
1.6
0.1
5
2
Pin 9, BFIN; Pin 13, BFNG; Pin 11, VOUT
POWER-DOWN INPUT
Logic Level, HI State
Logic Level, LO State
Pin 2, PWDN
–40°C < TA < +85°C, 2.7 V < VP < 5.5 V
–40°C < TA < +85°C, 2.7 V < VP < 5.5 V
POWER SUPPLY
Positive Supply Voltage
Quiescent Current
In Disabled State
Negative Supply Voltage4
Pin 10 and Pin 12, VPS1 and VPS2; Pin 1, VNEG
–20
Flowing out of Pin 9 or Pin 13
204
207
140
175
0.25
0.7
5.05
IPD > 1 µA (see Typical Performance Characteristics)
IPD > 1 µA (see Typical Performance Characteristics)
0.2 V to 4.8 V output swing
V
V
Ω
+20
mV
µA
MΩ
V
Ω
µV/√Hz
MHz
V/µs
2
1
3.0
|1VP –VN| < 8V
5
4.5
60
0
mV/dec
mV/dec
pA
pA
dB
dB
V
V
kΩ
2.02
2.08
0.4
35
VP – 0.1
0.5
1
10
15
RL = 1 kΩ to ground
pA
mA
V
mV/°C
mV
mV
mV/mA
2
OUTPUT BUFFER
Input Offset Voltage
Input Bias Current
Incremental Input Resistance
Output Range
Output Resistance
Wide-Band Noise3
Small Signal Bandwidth3
Slew Rate
Unit
5.5
5.3
–5.5
V
V
V
mA
µA
V
NOTES
1
Minimum and maximum specified limits on parameters that are guaranteed but not tested are six sigma values.
2
This bias is internally arranged to track the input voltage at INPT; it is not specified relative to ground.
3
Output Noise and Incremental Bandwidth are functions of Input Current; see Typical Performance Characteristics.
4
Optional
Specifications subject to change without notice.
–2–
REV. A
AD8304
ABSOLUTE MAXIMUM RATINGS*
PIN FUNCTION DESCRIPTIONS
Supply Voltage VP – VN . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 V
Input Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 mA
Internal Power Dissipation . . . . . . . . . . . . . . . . . . . . 270 mW
␪JA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C/W
Maximum Junction Temperature . . . . . . . . . . . . . . . . 125°C
Operating Temperature Range . . . . . . . . . . . –40°C to +85°C
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . . 300°C
Pin No. Mnemonic Function
*Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
1
VNEG
2
PWDN
3, 5
VSUM
4
INPT
6
VPDB
7
8
VREF
VLOG
9
BFIN
10
11
12
13
14
VPS2
VOUT
VPS1
BFNG
ACOM
PIN CONFIGURATION
VNEG 1
14 ACOM
PWDN 2
13 BFNG
VSUM 3
AD8304
12 VPS1
INPT 4
TOP VIEW 11 VOUT
(Not to Scale)
10 VPS2
VSUM 5
VPDB 6
9 BFIN
VREF 7
8 VLOG
Optional Negative Supply, VN. This
pin is usually grounded; for details of
usage, see Applications section.
Power-Down Control Input. Device is
active when PWDN is taken LOW.
Guard Pins. Used to shield the INPT
current line.
Photodiode Current Input. Usually
connected to photodiode anode (the
photo current flows toward INPT).
Photodiode Biaser Output. May be
connected to photodiode cathode to
provide adaptive bias control.
Voltage Reference Output of 2 V
Output of the Logarithmic Front-End
Processor; ROUT = 5 kΩ to ground.
Buffer Amplifier Noninverting Input
(High Impedance)
Positive Supply, VP (3.0 V to 5.5 V)
Buffer Output; Low Impedance
Positive Supply, VP (3.0 V to 5.5 V)
Buffer Amplifier Inverting Input
Analog Ground
ORDERING GUIDE
Model
Temperature Range
Package Description
Package Option
AD8304ARU
AD8304ARU-REEL
AD8304ARU-REEL7
AD8304-EVAL
–40°C to +85°C
Tube, 14-Lead TSSOP
13" Tape and Reel
7" Tape and Reel
Evaluation Board
RU-14
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
AD8304 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended
to avoid performance degradation or loss of functionality.
REV. A
–3–
WARNING!
ESD SENSITIVE DEVICE
AD8304–Typical Performance Characteristics
(VP = 5 V, VN = 0 V, TA = 25C, unless otherwise noted.)
1.6
1.4
0.510
TA = –40C, +25C, +85C
VN = –0.5V
TA = –40C, +25C, +85C
0.508
1.2
VLOG – V
0C
+70C
VSUM – V
–40C
+25C
+85C
1.0
0.8
0.6
0.506
0.504
–40C
+25C
0.4
+85C
0.502
0.2
0
100p
1n
10n
100n
1
10
INPUT – A
100
1m
0.500
100p
10m
1n
10n
TPC 1. VLOG vs. IPD
1
10
INPUT – A
100
1m
10m
TPC 4. VSUM vs. IPD
2.0
1.5
100n
2.8
TA = –40C, +25C, +85C
VN = –0.5V
TA = –40C, +25C, +85C
2.6
–40C
+25C
+85C
2.2
+25C
0C
0.5
2.0
VPDB – V
ERROR – dB (10mV/dB)
2.4
–40C
1.0
0
–0.5
1.8
1.6
1.4
+85C
+70C
1.2
–1.0
1.0
–1.5
0.8
1n
10n
100n
1
10
INPUT – A
100
1m
0.6
10m
0
1
TPC 2. Logarithmic Conformance (Linearity) for VLOG
4
5
6
INPUT – mA
7
8
9
2.2
0
1.8
4.5V
5.0V
1.00
0.75
5.5V
–1.5
1n
10n
100n
1
10
INPUT – A
100
1m
0.25
TPC 3. Absolute Deviation from Nominal Specified Value of VLOG for Several Supply Voltages
0
+85C
–0.25
1.0
–0.50
0.8
–0.75
0.6
100p
10m
0.50
1.4 +25C
1.2
–1.0
–40C
1.6
–0.5
–2.0
100p
TA = –40C, +25C, +85C
VP = 3.0V
2.0
1.0
0.5
10
1.25
2.4
VP = 4.5V, 5.0V, 5.5V
VN = –0.1V
VOUT – V
ERROR FROM IDEAL OUTPUT – dB (10mV/dB)
3
TPC 5. VPDB vs. IPD
2.0
1.5
2
1n
10n
100n
1
10
INPUT – A
100
1m
ERROR – dB (10mV/dB)
–2.0
100p
–1.00
10m
TPC 6. Logarithmic Conformance (Linearity) for a
3 V Single Supply (See Figure 6)
–4–
REV. A
AD8304
10
10nA100nA 1A
10
10A
10mA
1nA
–10
WIDEBAND NOISE – mV rms
NORMALIZED RESPONSE – dB
9
100A
0
1mA
–20
–30
–40
–50
–60
8
7
6
5
4
3
2
1
–70
100
1k
10k
100k
1M
FREQUENCY – Hz
10M
0
1n
100M
10n
100n
1
10
100
INPUT CURRENT – A
1m
10m
TPC 10. Total Wideband Noise Voltage at VLOG vs. IPD
TPC 7. Small Signal AC Response, IPD to VLOG
(5% Sine Modulation of IPD at Frequency)
3
100
GAIN = 1, 2, 2.5, 5
NORMALIZED RESPONSE – dB
10kHz
100kHz
V rms/ Hz
10
1
100Hz
1kHz
1MHz
0.1
0.01
1n
10n
100n
1
10
IPD – A
100
1m
–3
AV = 2.5
–6
AV = 2
–9
1k
10k
100k
1M
FREQUENCY – Hz
10M
100M
TPC 11. Small Signal Response of Buffer
10
100
fC = 1kHz
1nA
0
NORMALIZED GAIN – dB
10
10nA
1A
V rms/ Hz
AV = 1
AV = 5
–12
100
10m
TPC 8. Spot Noise Spectral Density at VLOG vs. IPD
1
0
100nA
10A
>100A
0.1
–10
–20
–30
–40
–50
–60
0.01
100
1k
10k
100k
FREQUENCY – Hz
1M
–70
10
10M
1k
FREQUENCY – Hz
10k
TPC 12. Small Signal Response of Buffer
Operating as Two-Pole Filter
TPC 9. Spot Noise Spectral Density at VLOG vs. Frequency
REV. A
100
–5–
100k
AD8304
20
2.0
TA = 25C
15
1.5
MEAN + 3
1.0
VREF DRIFT – mV
5
0.5
MEAN + 3
0
MEAN – 3
–0.5
0
–5
–10
–15
–1.0
MEAN – 3
–20
–1.5
–25
–2.0
100p
1n
10n
100n
1
10
INPUT – A
100
1m
–30
–40 –30 –20 –10
10m
0
10 20 30 40 50
TEMPERATURE – C
60
70
80
90
TPC 16. VREF Drift vs. Temperature (3σ to Either
Side of Mean)
TPC 13. Logarithmic Conformance Error
Distribution (3σ to Either Side of Mean)
5
3
TA = 0C, 70C
SLOPE CHANGE FROM 25C – mV/dec
4
ERROR – dB (10mV/dB)
3
2
MEAN + 3 @ 70C
1
MEAN 3 @ 0C
0
–1
MEAN – 3 @ 70C
–2
–3
–4
–5
100p
1n
10n
100n
1
10
INPUT – A
100
1m
5
INTERCEPT CHANGE FROM 25C – pA
MEAN 3 @40C
2
1
MEAN 3 @ 85C
–1
–2
–3
MEAN 3 @40C
–4
–5
100p
1n
10n
100n
1
10
INPUT – A
100
1m
1
0
–1
–2
–3
MEAN – 3
–4
0
10 20 30 40 50
TEMPERATURE – C
60
70
80
90
40
3
0
MEAN + 3
TPC 17. Slope Drift vs. Temperature (3σ to Either
Side of Mean)
TA = 40C, 85C
4
2
–5
–40 –30 –20 –10
10m
TPC 14. Logarithmic Conformance Error
Distribution (3σ to Either Side of Mean)
ERROR – dB (10mV/dB)
ERROR – dB (10mV/dB)
10
30
MEAN + 3
20
10
0
–10
–20
–30
MEAN – 3
–40
–50
–40 –30 –20 –10
10m
0
10 20 30 40 50
TEMPERATURE – C
60
70
80
90
TPC 18. Intercept Drift vs. Temperature (3σ to
Either Side of Mean)
TPC 15. Logarithmic Conformance Error
Distribution (3σ to Either Side of Mean)
–6–
REV. A
AD8304
160
8
140
6
vOS DRIFT – mV
4
120
MEAN + 3
100
HITS
2
80
0
60
–2
–4
40
MEAN – 3
20
–6
–40 –30 –20 –10
0
10 20 30 40 50
TEMPERATURE – C
60
70
80
0
60
90
TPC 19. Output Buffer Offset vs. Temperature
(3σ to Either Side of Mean)
160
140
140
120
120
100
100
HITS
160
HITS
180
80
80
60
60
40
40
20
20
198
200
202
LOGARITHMIC SLOPE – mV/dec
0
–20
204
TPC 20. Distribution of Logarithmic Slope, Sample 1000
REV. A
140
TPC 21. Distribution of Logarithmic Intercept,
Sample 1000
180
0
196
80
100
120
LOGARITHMIC INTERCEPT – pA
–10
0
10
INPUT GUARD OFFSET – mV
20
TPC 22. Distribution of Input Guard Offset Voltage
(VINPT – VSUM), Sample 1000
–7–
AD8304
BASIC CONCEPTS
Optical Measurements
The AD8304 uses an advanced circuit implementation that
exploits the well known logarithmic relationship between the
base-to-emitter voltage, VBE, and collector current, IC, in a
bipolar transistor, which is the basis of the important class of
translinear circuits*:
When interpreting the current IPD in terms of optical power incident on a photodetector, it is necessary to be very clear about the
transducer properties of a biased photodiode. The units of this
transduction process are expressed as amps per watt. The parameter ␳, called the photodiode responsivity, is often used for this
purpose. For a typical InGaAs p-i-n photodiode, the responsivity
is about 0.9 A/W.
VBE = VT log( I C /I S )
(1)
There are two scaling quantities in this fundamental equation, namely
the thermal voltage VT = kT/q and the saturation current IS. These
are of key importance in determining the slope and intercept for this
class of log amp. VT has a process-invariant value of 25.69 mV
at T = 25°C and varies in direct proportion to absolute temperature,
while IS is very much a process- and device-dependent parameter,
and is typically 10–16 A at T = 25°C but exhibits a huge variation
over the temperature range, by a factor of about a billion.
It is also important to note that amps and watts are not usually
related in this proportional manner. In purely electrical circuits,
a current IPD applied to a resistive load RL results in a power
proportional to the square of the current (that is, IPD2 RL). The
reason for the difference in scaling for a photodiode interface is
that the current IPD flows in a diode biased to a fixed voltage,
VPDB. In this case, the power dissipated within the detector
diode is simply proportional to the current IPD (that is, IPDVPDB)
and the proportionality of IPD to the optical power, POPT, is
preserved.
While these variations pose challenges to the use of a transistor as
an accurate measurement device, the remarkable matching and
isothermal properties of the components in a monolithic process
can be applied to reduce them to insignificant proportions, as will
be shown. Logarithmic amplifiers based on this unique property
of the bipolar transistor are called translinear log amps to distinguish them from other Analog Devices products designed for RF
applications that use quite different principles.
I PD = ρPOPT
Accordingly, a reciprocal correspondence can be stated between the
intercept current, IZ, and an equivalent “intercept power,” PZ, thus:
I Z = ρPZ
VLOG = VY log10( POPT /PZ )
(6)
For the AD8304 operating in its default mode, its IZ of 100 pA
corresponds to a PZ of 110 picowatts, for a diode having a
responsivity of 0.9 A/W. Thus, an optical power of 3 mW would
generate:
VLOG = 0.2V log10 ( 3 mW /110 pW ) = 1.487 V
(2)
(7)
Note that when using the AD8304 in optical applications, the
interpretation of VLOG is in terms of the equivalent optical
power, the logarithmic slope remains 10 mV/dB at this output.
This can be a little confusing since a decibel change on the
optical side has a different meaning than on the electrical side.
In either case, the logarithmic slope can always be expressed in
units of mV per decade to help eliminate any confusion.
VY is called the slope voltage (in the case of base-10 logarithms,
it is also the “volts per decade”). The fixed current IZ is called
the intercept. The scaling is chosen so that VY is trimmed to
200 mV/decade (10 mV/dB). The intercept is positioned at
100 pA; the output voltage VLOG would cross zero when IPD is
of this value. However, when using a single supply the actual
VLOG must always be slightly above ground. On the other hand,
by using a negative supply, this voltage can actually cross zero at
the intercept value.
Decibel Scaling
In cases where the power levels are already expressed as so many
decibels above a reference level (in dBm, for a reference of 1 mW),
the logarithmic conversion has already been performed, and the
“log ratio” in the above expressions becomes a simple difference. One needs to be careful in assigning variable names here,
because “P” is often used to denote actual power as well as this
same power expressed in decibels, while clearly these are numerically different quantities.
Using Equation 2, one can calculate the output for any value of IPD.
Thus, for an input current of 25 nA,
VLOG = 0.2 V log10(25 nA/100 pA) = 0.4796 V
(5)
and Equation 2 may then be written as:
The very strong temperature variation of the saturation current
IS is readily corrected using a second reference transistor, having
an identical variation, to stabilize the intercept. Similarly, proprietary techniques are used to ensure that the logarithmic slope is
temperature-stable. Using these principles in a carefully scaled
design, the now accurate relationship between the input current,
IPD, applied to Pin INPT, and the voltage appearing at the intermediate output Pin VLOG is:
VLOG = VY log10( I PD /I Z )
(4)
(3)
In practice, both the slope and intercept may be altered, to either
higher or lower values, without any significant loss of calibration
accuracy, by using one or two external resistors, often in conjunction with the trimmed 2 V voltage reference at Pin VREF.
Such potential misunderstandings can be avoided by using “D”
to denote decibel powers. The quantity VY (“volts per decade”)
must now be converted to its decibel value, VY´ = VY/10, because
there are 10 dB per decade in the context of a power measurement.
Then it can be stated that:
VLOG = 20 ( DOPT − DZ ) mV /dB
(8)
where DOPT is the optical power in decibels above a reference level,
and DZ is the equivalent intercept power relative to the same level.
This convention will be used throughout this data sheet.
*For a basic discussion of the topic, see Translinear Circuits: An Historical Overview,
B. Gilbert, Analog Integrated Circuits and Signal Processing, 9, pp. 95–118, 1996.
–8–
REV. A
AD8304
To repeat the previous example: for a reference power level of
1 mW, a POPT of 3 mW would correspond to a DOPT of 10 log10(3) =
4.77 dBm, while the equivalent intercept power of 110 pW will
correspond to a DZ of –69.6 dBm; now using Equation 8:
VLOG = 20 mV {4.77 – (–69.9)} = 1.487 V
voltage is applied to a processing block—essentially an analog divider
that effectively puts a variable proportional to temperature
underneath the T in Equation 10. In this same block, IREF is transformed to the much smaller current IZ, to provide the previously
defined value for VLOG, that is,
(9)
VLOG = VY log10 ( IPD /I Z )
which is in agreement with the result from Equation 7.
Recall that VY is 200 mV/decade and IZ is 100 pA. Internally,
this is generated first as an output current of 40 µA/decade
(2 µA/dB) applied to an internal load resistor from VLOG to
ACOM that is laser-trimmed to 5 kΩ ± 1%. The slope may be
altered at this point by adding an external shunt resistor. This is
required when using the minimum supply voltage of 3.0 V,
because the span of VLOG for the full 160 dB (eight-decade)
range of IPD amounts to 8 ⫻ 0.2 V = 1.6 V, which exceeds the
internal headroom at this node. Using a shunt of 5 kΩ, this is
reduced to 800 mV, that is, the slope becomes 5 mV/dB. In
those applications needing a higher slope, the buffer can provide
voltage gain. For example, to raise the output swing to 2.4 V,
which can be accommodated by the rail-to-rail buffer when
using a 3.0 V supply, a gain of 3⫻ can be used which raises the
slope to 15 mV/dB. Slope variations implemented in these ways
do not affect the intercept. Keep in mind these measures to
address the limitations of a small positive supply voltage will not
be needed when IPD is limited to about 1 mA maximum. They
can also be avoided by using a negative supply that allows VLOG
to run below ground, which will be discussed later.
GENERAL STRUCTURE
The AD8304 addresses a wide variety of interfacing conditions
to meet the needs of fiber optic supervisory systems, and will also
be useful in many nonoptical applications. These notes explain
the structure of this unique translinear log amp. Figure 1 is a
simplified schematic showing the key elements.
VPDB
VBE1
0.5V
VPDB
PHOTODIODE
INPUT CURRENT
~10k
VBE2–
296mVP
200
INTERCEPT AND
TEMPERATURE
COMPENSATION
(SUBTRACT AND
DIVIDE BY TK)
IREF
(INTERNAL)
IPD
VSUM
INPT
40A/dec
0.5V
VLOG
VLOG
0.6V
C1
0.5V
Q1
R1
5k
QM
VBE1
Q2
VBE2
ACOM
Figure 1 shows how a sample of the input current is derived using
a very small monitoring transistor, QM, connected in parallel with
Q1. This is used to generate the photodiode bias, VPDB, at Pin VPDB,
which varies from 0.6 V when IPD = 100 pA, and reverse-biases
the diode by 0.1 V (after subtracting the fixed 0.5 V at INPT)
and rises to 2.6 V at IPD = 10 mA, for a net diode bias of 2 V.
The driver for this output is current-limited to about 20 mA.
VNEG (NORMALLY GROUNDED)
Figure 1. Simplified Schematic
The photodiode current IPD is received at input Pin INPT. The
summing voltage at this node is essentially equal to that on the
two adjacent guard pins, VSUM, due to the low offset voltage of
the ultralow bias J-FET op amp used to support the operation of
the transistor Q1, which converts the current to a logarithmic
voltage, as delineated in Equation 1. VSUM is needed to provide
the collector-emitter bias for Q1, and is internally set to 0.5 V,
using a quarter of the reference voltage of 2 V appearing on
Pin VREF.
The system is completed by the final buffer amplifier, which is
essentially an uncommitted op amp with a rail-to-rail output
capability, a 10 MHz bandwidth, and good load-driving capabilities, and may be used to implement multipole low-pass filters,
and a voltage reference for internal use in controlling the scaling,
but that is also made available at the 2.0 V level at Pin VREF.
Figure 2 shows the ideal output VLOG versus IPD.
In conventional translinear log amps, the summing node is generally held at ground potential, but that condition is not readily
realized in a single-supply part. To address this, the AD8304 also
supports the use of an optional negative supply voltage, VN, at
Pin VNEG. For a VN of at least –0.5 V the summing node can
be connected to ground potential. Larger negative voltages may
be used, with essentially no effect on scaling, up to a maximum
supply of 8 V between VPOS and VNEG. Note that the resistance
at the VSUM pins is approximately 10 kΩ to ground; this voltage
is not intended as a general bias source.
Bandwidth and Noise Considerations
The response time and wide-band noise of translinear log amps
are fundamentally a function of the signal current IPD. The
bandwidth becomes progressively lower as IPD is reduced,
largely due to the effects of junction capacitances in Q1. This is
easily understood by noting that the transconductance (gm) of a
bipolar transistor is a linear function of collector current, IC,
(hence, translinear), which in this case is just IPD. The corresponding incremental emitter resistance is:
1
kT
re =
=
(12)
gm qI PD
Basically, this resistance and the capacitance CJ of the transistor
generate a time constant of reCJ and thus a corresponding low-pass
corner frequency of:
qI PD
f3dB =
(13)
2 π kTC j
showing the proportionality of bandwidth to current.
The input-dependent VBE of Q1 is compared with the fixed VBE of
a second transistor, Q2, which operates at an accurate internally
generated current, IREF = 10 µA. The overall intercept is arranged
to be 100,000 times smaller than IREF, in later parts of the signal chain.
The difference between these two VBE values can be written as
VBE 1 – VBE 2 = kT /q log10 ( I PD /I REF )
(10)
Thus, the uncertain and temperature-dependent saturation current,
IS that appears in Equation 1, has been eliminated. Next, to
eliminate the temperature variation of kT/q, this difference
REV. A
(11)
–9–
AD8304
is thus 4.0 V, which can be accommodated by the rail-to-rail
output stage when using the recommended 5 V supply.
1.6
The capacitor from VLOG to ground forms an optional singlepole low-pass filter. Since the resistance at this pin is trimmed
to 5 kΩ, an accurate time constant can be realized. For example, with CFLT = 10 nF, the –3 dB corner frequency is
3.2 kHz. Such filtering is useful in minimizing the output noise,
particularly when IPD is small. Multipole filters are more effective in reducing noise, and are discussed below. A capacitor
between VSUM and ground is essential for minimizing the
noise on this node. When the bias voltage at either VPDB or
VREF is not needed these pins should be left unconnected.
VLOG – V
1.2
0.8
0.4
0
100p
Slope and Intercept Adjustments
1n
10n
100n
1
10
INPUT – A
100
1m
10m
Figure 2. Ideal Form of VLOG vs. IPD
Using a value of 0.3 pF for CJ evaluates to 20 MHz/mA. Therefore, the minimum bandwidth at IPD = 100 pA would be 2 kHz.
While this simple model is useful in making a point, it excludes
other effects that limit its usefulness. For example, the network
R1, C1 in Figure 1, which is necessary to stabilize the system over
the full range of currents, affects bandwidth at all values of IPD.
Later signal processing blocks also limit the maximum value.
TPC 7 shows ac response curves for the AD8304 at eight representative currents of 100 pA to 10 mA, using R1 = 750 Ω and
C1 = 1000 pF. The values for R1 and C1 ensure stability over
the full 160 dB dynamic range. More optimal values may be used
for smaller subranges. A certain amount of experimental trial and
error may be necessary to select the optimum input network
component values for a given application.
The choice of slope and intercept depends on the application.
The versatility of the AD8304 permits optimal choices to be
made in two common situations. First, it allows an input current
range of less than the full 160 dB to use the available voltage span
at the output. Second, it allows this output voltage range to be
optimally positioned to fit the input capacity of a subsequent
ADC. In special applications, very high slopes, such as 1 V/dec,
allow small subranges of IPD to be covered at high sensitivity.
The slope can be lowered without limit by the addition of a
shunt resistor, RS, from VLOG to ground. Since the resistance
at this pin is trimmed to 5 kΩ, the accuracy of the modified
slope will depend on the external resistor. It is calculated using:
VY =
PWDN
10
2
12
PDB
BIAS
VREF
(14)
~10k
3
where SNSD is nV/Hz, IPD is expressed in microamps and TA = 25°C.
For an input of 1 nA, SNSD evaluates to almost 0.5 µV/√Hz; assuming a 20 kHz bandwidth at this current, the integrated noise
voltage is 70 µV rms. However, the calculation is not complete.
The basic scaling of the VBE is approximately 3 mV/dB; translated
to 10 mV/dB, the noise predicted by Equation 14 must be multiplied by approximately 3.33. The additive noise effects associated
with the reference transistor, Q2, and the temperature compensation circuitry must also be included. The final voltage noise
spectral density presented at the VLOG Pin varies inversely with
IPD, but not as simple as square root. TPCS 8 and 9 show the
measured noise spectral density versus frequency at the VLOG
output, for the same nine-decade spaced values of IPD.
The AD8304 may be powered down by taking the PWDN Pin
to a high logic level. The residual supply current in the disabled
mode is typically 60 µA.
7
VREF
VPDB
NC
Chip Enable
VPS1
IPD
14.7
IPD
(15)
VP
VPS2
Turning now to the noise performance of a translinear log amp,
the relationship between IPD and the voltage noise spectral density,
SNSD, associated with the VBE of Q1, evaluates to the following:
SNSD =
VY RS
R'S +5 kΩ
4
C1
1nF
5
0.5V
200mV/DEC
VSUM
CFLT
VLOG
8
INPT
5k
VSUM
TEMPERATURE
COMPENSATION
BFIN
BFNG
9
RB
10k
13
10nF
R1
750
RA
15k
VNEG
1
ACOM
14
VOUT
11
VOUT
500mV/DEC
NC = NO CONNECT
Figure 3. Basic Connections (RA, RB, CFLT are
optional; R1 and C1 are the default values)
For example, using RS = 3 kΩ, the slope is lowered to 75 mV per
decade or 3.75 mV/dB. Table I provides a selection of suitable
values for RS and the resulting slopes.
Table I. Examples of Lowering the Slope
USING THE AD8304
The basic connections (Figure 3) include a 2.5:1 attenuator in
the feedback path around the buffer. This increases the basic slope
of 10 mV/dB at the VLOG Pin to 25 mV/dB at VOUT. For the
full dynamic range of 160 dB (80 dB optical), the output swing
–10–
RS (k)
VY (mV/dec)
3
5
15
75
100
150
REV. A
AD8304
In addition to uses in filter and comparator functions, the buffer
amplifier provides the means to adjust both the slope and intercept, which require a minimal number of external components.
The high input impedance at BFIN, low input offset voltage,
large output swing, and wide bandwidth of this amplifier permit
numerous transformations of the basic VLOG signal, using standard op amp circuit practices. For example, it has been noted
that to raise the gain of the buffer, and therefore the slope, a
feedback attenuator, RA and RB in Figure 3, should be inserted
between VLOG and the inverting input Pin BFNG.
A wide range of gains may be used and the resistor magnitudes
are not critical; their parallel sum should be about equal to the
net source resistance at the noninverting input. When high gains
are used, the output dynamic range will be reduced; for maximum swing of 4.8 V, it will amount to simply 4.8 V/VY decades.
Thus, using a ratio of 3⫻, to set up a slope 30 mV/dB (600 mV/
decade), eight decades can be handled, while with a ratio of 5⫻,
which sets up a slope of 50 mV/dB (1 V/decade), the dynamic
range is 4.8 decades, or 96 dB. When using a lower positive
supply voltage, the calculation proceeds in the same way,
remembering to first subtract 0.2 V to allow for 0.1 V upper and
lower headroom in the output swing.
Alteration of the logarithmic intercept is only slightly more tricky.
First note that it will rarely be necessary to lower the intercept
below a value of 100 pA, since this merely raises all output voltages further above ground. However, where this is required, the
first step is to raise the voltage VLOG by connecting a resistor, RZ,
from VLOG to VREF (2 V) as shown in Figure 4.
VP
VPS2
10
PWDN
VPS1
2
12
IPD
PDB
NC 6
3
4
C1
1nF
5
BIAS
VREF
VREF
7
VPDB
~10k
0.5V
Table II. Examples of Lowering the Intercept
VY (mV/decade)
IZ (pA)
RA (k)
RB (k)
RZ (k)
200
200
200
300
300
300
400
400
400
500
500
500
1
10
50
1
10
50
1
10
50
1
10
50
20.0
10.0
3.01
10.0
8.06
6.65
11.5
9.76
8.66
16.5
14.3
13.0
100
100
100
12.4
12.4
12.4
8.2
8.2
8.2
8.2
8.2
8.2
25
50
165
25
50
165
25
50
165
25
50
165
Equations for use with Table II:

I 
RZ
RLOG 
VOUT = G VY ×
× log10  PD  + VREF ×

R
+
R
I
R
 Z 
Z
LOG
LOG + RZ 


where
G = 1+
Generally, it will be useful to raise the intercept. Keep in mind
that this moves the VLOG line in Figure 2 to the right, lowering all
output values. Figure 5 shows how this is achieved. The feedback
resistors, RA and RB, around the buffer are now augmented with
a third resistor, RZ, placed between the Pins BFNG and VREF.
This raises the zero-signal voltage on BFNG, which has the effect
of pushing VOUT lower. Note that the addition of this resistor also
alters the feedback ratio. However, this is readily compensated
in the design of the network. Table III lists the resistor values
for representative intercepts.
RZ
VSUM
Table III. Examples of Raising the Intercept
VLOG
8
INPT
5k
VSUM
TEMPERATURE
COMPENSATION
BFIN
BFNG
9
13
10nF
R1
750
RA
VNEG
NC = NO CONNECT
RA
and RLOG = 5 kΩ
RB
1
ACOM
14
VOUT
RB
11
VOUT
Figure 4. Method for Lowering the Intercept
This has the effect of elevating VLOG for small inputs while lowering the slope to some extent because of the shunt effect of RZ
on the 5 kΩ output resistance. Then, if necessary, the slope may
be increased as before, using a feedback attenuator around the
buffer. Table II lists some examples of lowering the intercept
combined with various slope variations.
VY (mV/decade)
IZ (nA)
RA (k)
RB (k)
RC(k)
300
300
400
400
400
500
500
500
10
100
10
100
500
10
100
500
7.5
8.25
10
9.76
9.76
12.4
12.4
11.5
37.4
130
16.5
25.5
36.5
12.4
16.5
20.0
24.9
18.2
25.5
16.2
13.3
24.9
16.5
12.4
Equations for use with Table III:

RA RB 
I 
VOUT = G VY × log10  PD  – VREF ×

 IZ 
RA RB + RC 

where
G = 1+
REV. A
–11–
RA
R × RB
and RA RB = A
RA + RB
RB RC
AD8304
Using the Adaptive Bias
VP
VPS2
10
PWDN
VPS1
2
For most photodiode applications, the placement of the anode
somewhat above ground is acceptable, as long as the positive
bias on the cathode is adequate to support the peak current for a
particular diode, limited mainly by its series resistance. To address
this matter, the AD8304 provides for the diode a bias that varies
linearly with the current. This voltage appears at Pin VPDB, and
varies from 0.6 V (reverse-biasing the diode by 0.1 V) for IPD =
100 pA and rises to 2.6 V (for a diode bias of 1 V) at IPD = 10 mA.
This results in a constant internal junction bias of 0.1 V when the
series resistance of the photodiode is 200 Ω. For optical power
measurements over a wide dynamic range the adaptive biasing
function will be valuable in minimizing dark current while preventing the loss of photodiode bias at high currents. Use of the
adaptive bias feature is shown in Figure 7.
12
IPD
PDB
NC 6
3
4
C1
1nF
5
BIAS
VREF
VREF
7
VPDB
~10k
0.5V
VSUM
VLOG
8
INPT
5k
VSUM
BFIN
TEMPERATURE
COMPENSATION
BFNG
RC
9
13
10nF
R1
750
RA
1
11
14
VNEG
NC = NO CONNECT
RB
ACOM
VOUT
VOUT
Figure 5. Method for Raising the Intercept
VP
VPS2
Low Supply Slope and Intercept Adjustment
When using the device with a positive supply less than 4 V, it is
necessary to reduce the slope and intercept at the VLOG Pin in
order to preserve good log conformance over the entire 160 dB
operating range. The voltage at the VLOG Pin is generated by
an internal current source with an output current of 40 µA/decade
feeding the internal laser-trimmed output resistance of 5 kΩ. When
the voltage at the VLOG Pin exceeds VP – 2.3 V, the current
source ceases to respond linearly to logarithmic increases in current.
This headroom issue can be avoided by reducing the logarithmic
slope and intercept at the VLOG Pin. This is accomplished by
connecting an external resistor RS from the VLOG Pin to ground
in combination with an intercept lowering resistor RZ. The values
shown in Figure 6 illustrate a good solution for a 3.0 V positive
supply. The resulting logarithmic slope measured at VLOG is
62.5 mV/decade with a new intercept of 57 fA. The original
logarithmic slope of 200 mV/decade can be recovered using voltage
gain on the internal buffer amplifier.
VP
VPS2
10
PWDN
VPS1
2
12
IPD
PDB
NC 6
3
4
C1
1nF
5
BIAS
VREF
VREF
7
RZ
15.4k
VPDB
~10k
0.5V
VLOG
8
INPT
5k
VSUM
TEMPERATURE
COMPENSATION
BFIN
BFNG
9
62.5mV/DEC
13
10nF
R1
750
RA
4.98k
1
VNEG
NC = NO CONNECT
RB
2.26k
11
14
ACOM
VOUT
PDB
PWDN
VPS1
2
BIAS
12
7 VREF
VREF
VPDB
6
IPD
3
4
C1
1nF
5
~10k
0.5V
VSUM
CFILT
VLOG
8
INPT
5k
VSUM
TEMPERATURE
COMPENSATION
BFIN
BFNG
9
RB
13
10nF
R1
750
RA
VNEG
1
ACOM
14
VOUT
11
VOUT
Figure 7. Using the Adaptive Biasing
Capacitor CPB, between the photodiode cathode at Pin VPDB
and ground, is included to lower the impedance at this node and
thereby improve the high frequency accuracy at those current
levels where the AD8304 bandwidth is high. It also ensures an
HF path for any high frequency modulation on the optical signal
which might not otherwise be accurately averaged. It will not be
necessary in all cases, and experimentation may be required to find
an optimum value.
Changing the Voltage at the Summing Node
RS
2.67k
VSUM
CPB
10
VOUT
Figure 6. Recommended Low Supply Application Circuit
The default value of VSUM is determined by using a quarter of
VREF (2 V). This may be altered by applying an independent voltage source to VSUM, or by adding an external resistive divider
from VREF to VSUM. This network will operate in parallel with
the internal divider (40 kΩ and 13.3 kΩ), and the choice of external
resistors should take this into account. In practice, the total
resistance of the added string may be as low as 10 kΩ (consuming
400 µA from VREF). Low values of VSUM and thus VCE (see
Figure 13) are not advised when large values of IPD are expected.
Implementing Low-Pass Filters
Noise, leading to uncertainty in an observed value, is inherent to
all measurement systems. Translinear log amps exhibit significant
amounts of noise for reasons stated above, and are more troublesome at low current levels. The standard way of addressing this
problem is to average the measurement over an appropriate time
interval. This can be achieved in the digital domain, in post-ADC
DSP, or in analog form using a variety of low-pass structures.
–12–
REV. A
AD8304
The use of a capacitor at the VLOG Pin to create a single-pole
filter has already been mentioned. The small added cost of the few
external components needed to realize a multipole filter is often
justified in a high performance measurement system. Figure 8
shows a Sallen-Key filter structure. Here, the resistor needed at
the front of the network is provided entirely by the accurate 5 kΩ
present at the VLOG output; RB will have a similar value. The corner
frequency and Q (damping factor) are determined by the capacitors
CA and CB and the gain G = (RA + RB)/RB. A suggested starting
point for choosing these components using various gains is provided in Table IV; the values shown are for a 1 kHz corner (also
see TPC 12). This frequency can be increased or decreased by
scaling the capacitor values. Note that RD, G, and the capacitor ratio
CA/CB should not deviate from the suggested values to maintain the
shape of the ac amplitude response and pulse overshoot provided
by the values shown in this table. In all cases, the roll-off rate above
the corner is 40 dB/dec.
10
PWDN
VPS1
2
12
IPD
PDB
NC 6
3
4
C1
1nF
5
BIAS
VREF
0.5V
VSUM
8
INPT
5k
VSUM
BFIN
TEMPERATURE
COMPENSATION
BFNG
VLOG
RD
9
RB
13
10nF
CA
R1
750k
CB
RA
VNEG
NC = NO CONNECT
1
ACOM
14
PWDN
VPS1
2
12
IPD
PDB
NC 6
3
4
C1
1nF
5
BIAS
VREF
7
VREF
VPDB
~10k
0.5V
VSUM
8
INPT
5k
VSUM
TEMPERATURE
COMPENSATION
BFIN
BFNG
VLOG
RG
9
13
10nF
R1
750
RA
RH
VNEG
NC = NO CONNECT
1
ACOM
14
VOUT
11
VOUT
Figure 9. Using the Buffer as a Comparator
Using a Negative Supply
The use of a negative supply, VN, allows the summing node to
be placed exactly at ground level, because the input transistor
(Q1 in Figure 1) will have a negative bias on its emitter. VN may
be as small as –0.5 V, making the VCE the same as for the default
case. This bias need not be accurate, and a poorly defined source
can be used.
7 VREF
VPDB
~10k
10
Most applications of the AD8304 will require only a single supply
of 3.0 V to 5.5 V. However, to provide further versatility, dual
supplies may be employed, as illustrated in Figure 10.
VP
VPS2
VP
VPS2
11
VOUT
VOUT
Figure 8. Two-Pole Low-Pass Filter
Table IV. Two-Pole Filter Parameters for 1 kHz Cutoff
Frequency*
A larger supply of up to –5 V may be used. The effect on scaling
is minor. It merely moves the intercept by ~0.01 dB/V. Accordingly, an uncertainty of 0.2 V in VN would result in a negligible
error of 0.002 dB. The slope is unaffected by VN. The log linearity will be degraded at the extremes of the dynamic range as
indicated in Figure 11. The bias current, buffer output (and its
load) current, and the full IPD all have to be absorbed by this
negative supply, and its supply capacity must be ensured for the
maximum current condition.
VP
RA
(k)
RB
(k)
0
10
12
24
open
10
8
6
G
VY
(V/decade)
RD
(k)
CA
(nF)
CB
(nF)
1
2
2.5
5
0.2
0.4
0.5
1.0
11.3
6.02
12.1
10.0
12
33
33
33
12
22
18
18
VPS2
10
PWDN
VPS1
2
12
IPD
PDB
NC 6
3
4
The corner frequency can be adjusted by scaling capacitors C A and CB. For
example, to reduce the corner frequency to 100 Hz, raise the values of C A and
CB by 10 ⫻.
*See TPC 12.
C1
1nF
5
BIAS
VREF
7
~10k
0.5V
VSUM
8
INPT
5k
VSUM
TEMPERATURE
COMPENSATION
BFIN
BFNG
REV. A
VLOG
9
13
R1
750
RA
Operation in Comparator Modes
In certain applications, the need may arise to generate a logical
output when the input current has reached a certain value. This
can be easily addressed by using a fraction of the voltage reference to provide the setpoint (threshold) and using the buffer
without feedback in a comparator mode, as illustrated in Figure 9.
Since VLOG runs from ground up to 1.6 V maximum, the 2 V
reference is more than adequate to cover the full dynamic range
of IPD. Note that the threshold for an increasing IPD is unchanged,
while the release point for decreasing currents is 5 dB below
this. Raising RH to 5 MΩ reduces the hysteresis to 0.5 dB, or it
may be increased using a lower value for RH.
VREF
VPDB
RB
1
NC = NO CONNECT
VNEG
ACOM
14
VOUT
11
VOUT
VN (–0.5V TO –3V)
Figure 10. Using a Negative Supply
With the summing node at ground, the AD8304 may now be used
as a voltage-input log amp, simply by inserting a suitably scaled
resistor from the voltage source to the INPT Pin. The logarithmic accuracy for small voltages is limited by the offset of the JFET
op amp, appearing between this pin and VSUM.
The use of a negative supply also allows the output to swing below
ground, thereby allowing the intercept to correspond to a midrange
value of IPD. However, the voltage VLOG remains referenced to the
–13–
AD8304
ACOM Pin, and does not normally go negative with regard to this
pin, but is free to do so. Therefore, a resistor from VLOG to the
negative supply can lower VLOG, thus raising the intercept. A more
accurate method for repositioning the intercept is described below.
2.0
ERROR – dB (10mV/dB)
1.0
WITHOUT INTERCEPT ADJUST
VNEG = 0
0
–0.5
VNEG = –0.5
WITH INTERCEPT ADJUST
–1.0
–1.5
VNEG = –3
–2.0
100p
1n
10n
100n
1
10
INPUT – A
100
1m
The AD8304 incorporates features that improve its usefulness in
both fiber optic supervisory applications and in more general ones.
To aid in the exploration of these possibilities, a SPICE macromodel is provided and a versatile evaluation board is available.
The macromodel is shown in generalized schematic form (and thus
is independent of variations in SPICE programs) in Figure 12.
Q1, QM, and Q2 (here made equal in size) correspond to the
identical transistors in Figure 1. The model parameters for these
transistors are not critical; the default model provided in SPICE
libraries will be satisfactory. However, the AD8304 employs
compensation techniques to reduce errors caused by junction
resistances (notably, RB and RE) at high input currents. Therefore, it is advisable to set these to zero. While this will not model
the AD8304 precisely, it is safer than using possibly high default
values for these parameters. The low current model parameters
may also need consideration. Note that no attempt is made to
capture either dynamic behavior or the effects of temperature in this
simple macromodel; scaling is correct for 27°C.
1.5
0.5
APPLICATIONS
10m
Figure 11. Log Conformance (Linearity) vs. IPD for
Various Negative Supplies
E2
5
V1
V
+
I1
1
3k
2
3
IN
I1
IPD
C1
I1
C1
E1
V1
Q1
I2
Q2
I3
Q3
.MODEL
E2
E3
E4
V2
R1
C2
R2
RL
Q1
0
IN
2
1
IN
0
3
0
4
NPN
5
6
7
8
8
9
9
VLOG
Q2
IN
0
0
0
2
3
3
4
4
NPN
0
0
0
7
9
0
VLOG
0
DC
1.0N
IN
0.5
0
1
0
316.2
0
I2
E4
E3
4
100k
6
7
V2 R1
+
V
C2
R2
VLOG
RL
Q3
1A
1
3K
NPN
NPN
NPN
POLY (2) 2 3 1 0 0, 0, 0, 0, 1
POLY (2) 4 3 7 0 0, 0, 0, 0, 1
6
5
100K
0.8
100
163P
4.9K
1000K
Figure 12. Basic Macromodel
–14–
REV. A
AD8304
Summing Node at Ground and Voltage Inputs
A negative supply may be used to reposition the input node at
ground potential. A voltage as small as –0.5 V is sufficient. Figure 13
shows the use of this feature. An input current of up to 10 mA is
supported.
This connection mode will be useful in cases where the source is a
positive voltage VSIG referenced to ground, rather than for use with
photodiodes, or other “perfect” current sources. RIN scales the
input current and should be chosen to optimally position the range
of IPD, or provide a very high input resistance, thus minimizing
the loading of the signal source. For example, assume a voltage
source that spans the four-decade range from 100 mV to 1 kV and
is desired to maximize RIN. When set to 1 GΩ, IPD spans the range
100 pA to 1 mA. Using a value of 10 MΩ, the same four decades
of input voltage would span the central current range of 10 nA
to 100 mA.
is grounded. A negative supply capable of supporting the input
current IPD must be used, the fraction of quiescent bias that flows
out of the VNEG Pin, and the load current at VLOG. For the
example shown in Figure 14, this totals less than 20 mA when
driving a 1 kΩ load as far as –4 V.
The use of a much larger value for the intercept may be useful in
certain situations. In this example, it has been moved up four
decades, from the default value of 100 pA to the center of the full
eight-decade range at 1 mA. Using a voltage input as described
above, this corresponds to an altered voltage-mode intercept, VZ,
which would be 1 V for RIN = 1 MΩ. To take full advantage of the
larger output swing, the gain of the buffer has been increased to
4.53, resulting in a scaling of 900 mV/decade and a full-scale
output of ± 3.6 V.
VP
VPS2
Smaller input voltages can be measured accurately when aided by
a small offset-nulling voltage applied to VSUM. The optional
network shown in Figure 13 provides more than ± 20 mV for
this purpose.
10
PWDN
VPS1
2
PDB
NC 6
3
RIN
4
12
IPD
AD8304
5
VSIG
PDB
NC
VREF
7
6
RIN
4
IPD
5
~10k
2
12
BIAS
VREF
7
~10k
0.5V
VSUM
VLOG
8
INPT
5k
VSUM
TEMPERATURE
COMPENSATION
BFIN
BFNG
RC
12.4k
9
13
RA
13.3k
1k
0.5V
VSUM
8
INPT
5k
VSUM
VREF
VPDB
VREF
VPDB
3
VSIG
BIAS
VPS1
AD8304
VP
VPS2
10
PWDN
TEMPERATURE
COMPENSATION
BFIN
BFNG
RB
22.6k
VLOG
1
VP
9
VLOW
10k
11
14
VNEG
VN
ACOM
VOUT
VOUT
RL
1k
NC = NO CONNECT
13
RA
Figure 14. Using a Negative Supply to Allow the
Output to Swing Below Ground
1k
RB
1
VP
VLOW
10k
ACOM
Inverting the Slope
11
14
VNEG
VN
VOUT
VOUT
The buffer is essentially an uncommitted op amp that can be used
to support the operation of the AD8304 in a variety of ways. It
can be completely disconnected from the signal chain when not
needed. Figure 15 shows its use as an inverting amplifier; this
changes the polarity of the slope. The output can either be
repositioned to all positive values by applying a fraction of VREF
to the BFIN Pin, or range negative when using a negative supply.
The full design for a practical application is left undefined in this
brief illustration, but a few cases will be discussed.
NC = NO CONNECT
Figure 13. Using a Negative Supply and Placing VSUM at
Ground Permits Voltage-Mode Inputs
The minimum voltage that can be accurately measured is then
limited only by the drift in the input offset of the AD8304. The
specifications show the maximum spread over the full temperature and supply range. Over a limited temperature range, and with
a regulated supply, the offset drift will be lower; in this situation,
processing of inputs down to 5 mV is practicable.
The input system of the AD8304 is quasi-differential, so VSUM
can be placed at an arbitrary reference level VLOW, over a wide
range, and used as the “signal LO” of the source. For example,
using VP = 5 V and VN = –3 V, VLOW can be any voltage within
a ± 2.5 V range.
For example, suppose we need a slope of –30 mV/dB; this requires
the gain to be three. Since VLOG exhibits a source resistance of
5 kΩ, RB must be 15 kΩ. In cases where a small negative supply
is available, the output voltage can swing below ground, and the
BFIN Pin may be grounded. But a negative slope is still possible
when only a single supply is used; a positive offset, VOFS, is applied
to this pin, as indicated in Figure 15. In general, the resulting
output voltage can be expressed as:
Providing Negative Outputs and Rescaling
 R 
I 
VOUT = –  B  VY × log10  PD  + VOFS
 5 kΩ 
 IZ 
As noted, the AD8304 allows the buffer to drive a load to negative
voltages with respect to ACOM, the analog common pin, which
REV. A
–15–
(16)
AD8304
down by 1.6 V. Clearly, a higher slope (or gain) is desirable, in
which case VOFS should be set to a smaller voltage to avoid railing
the output at low currents. If VOFS = 1.2 V and G = 33, VOUT
now starts at 4.8 V and falls through this same voltage toward
ground with a slope of –0.6 V per decade, spanning the full
range of IPD.
VP
VPS2
10
PWDN
VPS1
2
12
AD8304
IPD
PDB
NC 6
3
4
C1
1nF
5
BIAS
VREF
7 VREF
VPDB
~10k
0.5V
VSUM
8
INPT
5k BFIN
VSUM
TEMPERATURE
COMPENSATION
BFNG
Programmable Level Comparator with Hysteresis
VLOG
The buffer amplifier and reference voltage permit a calibrated
level detector to be realized. Figure 16 shows the use of a 10-bit
MDAC to control the setpoint to within 0.1 dB of an exact value
over the 100 dB range of 1 nA ≤ IPD ≤ 100 µA when the fullscale output of the MDAC is equal to that of its reference. The
2 V VREF also sets the minimum value of VSPT to 0.2 V, corresponding to an input of 1 nA. Since 100 dB at the VLOG interface
corresponds to a 1 V span, the resistor network is calculated to
provide a maximum VSPT of 1.2 V while adding the required
10% of VREF.
VOFS
9
13
10nF
R1
750
RB
1
NC = NO CONNECT
11
14
VNEG
ACOM
VOUT
VOUT
VN (–0.5V TO –3V)
Figure 15. Using the Buffer to Invert the Polarity
of the Slope
In this example, the hysteresis range is arranged to be 0.1 dB,
(1 mV at VLOG) when using a 5 V supply. This will usually be
adequate to prevent noise that causes the comparator output to
thrash. That risk can be reduced further by using a low-pass filtering
capacitor at VLOG (shown dotted) to decrease the noise bandwidth.
When the gain is set to 13 (RB = 5 kΩ) the 2 V VREF can be tied
directly to BFIN, in which case the starting point for the output
response is at 4 V. However, since the slope in this case is only
–0.2 V/decade, the full current range will only take the output
VP
VPS2
PWDN
VPS1
10
2
12
PDB
BIAS
VREF
AD8304
IPD
NC 6
3
4
1nF
5
VREF
7
VPDB
~10k
0.5V
VSUM
VLOG
8
INPT
5k
VSUM
BFIN
TEMPERATURE
COMPENSATION
BFNG
9
VREF
VOUT
MDAC
13
49.9k
VSPT
10nF
100k
750
1
ACOM
50M
VOUT
RH
11
14
VNEG
NC = NO CONNECT
VOUT
Figure 16. Calibrated Level Comparator
VP
VPS2
PWDN
VPS1
10
2
12
PDB
BIAS
VREF
AD8304
ISRC
NC 6
3
4
25k
5
VREF
7
VPDB
~10k
0.5V
VSUM
VLOG
8
INPT
5k
VSUM
TEMPERATURE
COMPENSATION
BFIN
BFNG
VREF
VOUT
9
13
C1
10nF
1
VNEG
1k
C2
1nF
100k
11
14
ACOM
MDAC
VOUT
VN (–0.5V TO –5V)
NC = NO CONNECT
Figure 17. Multidecade Current Source
–16–
REV. A
AD8304
Programmable Multidecade Current Source
TRIAX
CONNECTOR*
The AD8304 supports a wide variety of general (nonoptical)
applications. For example, the need frequently arises in test
equipment to provide an accurate current that can be varied over
many decades. This can be achieved using a logarithmic amplifier
as the measuring device in an inverse function loop, as illustrated
in Figure 16. This circuit generates the current:
ISRC = 100 pA × 10 (
V SPT /0.2 )
PWDN
Transistor QA may be a single bipolar device, which will result in
a small alpha error in ISRC (the current is monitored in the emitter
branch), or a Darlington pair or an MOS device, either of which
ensure a negligible difference between IPD and ISRC. In this example,
the bipolar pair is used. The output voltage compliance is determined by the collector breakdown voltage of these transistors,
while the minimum voltage depends on where VSUM is placed.
Optional components could be added to put this node and VNEG
at a low enough bias to allow the voltage to go slightly below ground.
VOUT
KEITHLEY 236
CHARACTERIZATION
BOARD
BFIN
VLOG
VSUM
VPDB
VREF
RIBBON
CABLE
DC MATRIX, DC SUPPLIES, DMM
Figure 18. Primary Characterization Setup
The primary characterization setup shown in Figure 18 is used to
measure the static performance, logarithmic conformance, slope
and intercept, buffer offset and VREF drift with temperature, and
the performance of the VPDB Pin functions. For the dynamic tests,
such as noise and bandwidth, more specialized setups are used.
HP 3577A
NETWORK
ANALYZER
OUTPUT INPUT INPUTA INPUTB
AD8304
+IN
AD8138
B
EVALUATION
A
BOARD
Characterization Setups and Methods
REV. A
INPT
*SIGNAL: INPT;
GUARD: VSUM;
SHIELD: GROUND
Many variations of this basic circuit are possible. For example, the
current can be continuously controlled by a simple voltage, or
by a second current. Larger output currents can be controlled by
setting VSUM to zero and using a current shunt divider.
During the primary characterization of the AD8304, the device
was treated as a high precision current-in logarithmic amplifier
(converter). Rather than attempting to accurately generate photocurrents by illuminating a photodiode, precision current sources,
like the Keithley 236, were used as input sources. Great care was
taken when applying the low level input currents. The triax output
of the current source was used with the guard connected to VSUM
at the characterization board. On the board the input trace was
guarded by connecting adjacent traces and a portion of an internal
copper layer to the VSUM Pins. One obvious reason for the care
was leakage current. With 0.5 V as the nominal bias on the
INPT Pin, a resistance of 50 GΩ to ground would cause 10 pA
of leakage, or about one decibel of error at the low end of the
measurement range. Additionally, the high output resistance of
the current source and the long signal cable lengths commonly
needed in characterization make a good receiver for 60 Hz emissions. Good guarding techniques help to reduce the pickup of
unwanted signals.
VPOS
AD8304
(17)
The principle is as follows. The current in QA is forced to supply
a certain IPD by measuring the error between a setpoint VSPT and
VLOG, and nulling this error by integration. This is performed by
the internal op amp and capacitor C1, with a time constant formed
with the internal 5 kΩ resistor. The choice of C1 in this example
ensures loop stability over the full eight-decade range of output
currents; C2 reduces phase lag. The system is completed with a
10-bit MDAC using VREF as its reference, whose output is scaled
to 1.6 V FS by R1 and R2 (whose parallel sum is also 5 kΩ).
VNEG
POWER
SPLITTER
1 VNEG
ACOM 14
2 PWDN
BFNG 13
3 VSUM
VPS1 12
4 INPT
VOUT 11
5 VSUM
VPS2 10
6 VPDB
BFIN
9
7 VREF
VLOG
8
+VS
0.1F
49.9
Figure 19. Configuration for Buffer Amplifier
Bandwidth Measurement
Figure 19 shows the configuration used to measure the buffer
amplifier bandwidth. The AD8138 Evaluation Board provides a
dc offset at the buffer input, allowing measurement in single-supply
mode. The network analyzer input impedance was set to 1 MΩ.
–17–
AD8304
HP 3577A
NETWORK
ANALYZER
HP 89410A
SOURCE TRIGGER
OUTPUT INPUT INPUTA INPUTB
AD8304
POWER
SPLITTER
+IN
AD8138
CHANNEL CHANNEL
2
1
AD8304
B
EVALUATION
A
BOARD
VNEG
ACOM 14
BFNG 13
1 VNEG
ACOM 14
2
PWDN
2 PWDN
BFNG 13
3
VSUM
VPS1 12
3 VSUM
VPS1 12
4
INPT
VOUT 11
4 INPT
VOUT 11
5
VSUM
VPS2 10
5 VSUM
VPS2 10
6
VPDB
BFIN 9
6 VPDB
BFIN
9
7
VREF
VLOG 8
7 VREF
VLOG
8
R1
+VS
R1
750
1
ALKALINE
D CELL
0.1F
750
1nF
ALKALINE
D CELL
1nF
Figure 20. Configuration for Logarithmic
Amplifier Bandwidth Measurement
Figure 21. Configuration for Noise Spectral
Density Measurement
The setup shown in Figure 20 was used for frequency response
measurements of the logarithmic amplifier section. In this configuration, the AD8138 output was offset to 1.5 V and R1 was
adjusted to provide the appropriate operating current. The
buffer amplifier was then used; still any capacitance added at
the VLOG Pin during measurement would form a filter with the
on-chip 5 kΩ resistor.
The configuration illustrated in Figure 21 measures the device
noise. Batteries provide both the supply and the input signal to
remove the supplies as a possible noise source and to reduce
ground loop effects. The AD8304 Evaluation Board and the
current setting resistors are mounted in closed aluminum enclosures to provide additional shielding to external noise sources.
+VS
R10
10k
SW1
LK2 OPEN
INPUT
LK1
INSTALLED
R15
750
C11
1nF
BIASER
GND
Evaluation Board
An evaluation board is available for the AD8304, the schematic
for which is shown in Figure 22, and the two board sides are
shown in Figure 23 and Figure 24. It can be configured for a wide
variety of experiments. The board is factory set for Photoconductive Mode with a buffer gain of unity, providing a slope of
10 mV/dB and an intercept of 100 pA. By substituting resistor and
capacitor values, all of the application circuits presented in this
data sheet can be evaluated. Table V describes the various configuration options.
–VS
AD8304
C1
0.1nF
R7
OPEN
C2
1nF
ACOM 14
2 PWDN
BFNG 13
3 VSUM
VPS1 12
4 INPT
VOUT 11
5 VSUM
VPS2 10
R5
OPEN
R7
OPEN
R9
0.1F
1 VNEG
R6
OPEN
C9
10nF
R2
0
C3
1nF
C4
0.1F
C7
OPEN
R11
0
6 VPDB
C10
0.1F
R1
OPEN
R13
0
R12
OPEN
BFIN 9
7 VREF
VLOG 8
C8
OPEN
LOG
OUT
R14
0
C6
OPEN
BUFFER
OUT
C5
OPEN
R4
OPEN
R3
OPEN
Figure 22. Evaluation Board Schematic
–18–
REV. A
AD8304
Figure 23. Component Side Layout
Figure 24. Component Side Silkscreen
Table V. Evaluation Board Configuration Options
Component
Function
Default Condition
VP, VN, AGND
Positive and Negative Supply and Ground Pins
Not Applicable
SW1, R10
Device Enable: When SW1 is in the “0” position, the PWDN Pin is
connected to ground and the AD8304 is in its normal operating mode.
SW1 = Installed
R10 = 10 kΩ (Size 0603)
R1, R2
Buffer Amplifier Gain/Slope Adjustment: The logarithmic slope
of the AD8304 can be altered using the buffer’s gain-setting resistors,
R1 and R2.
R1 = Open (Size 0603)
R2 = 0 Ω (Size 0603)
R3, R4
Intercept Adjustment: A dc offset can be applied to the input terminals of the buffer amplifier to adjust the effective logarithmic intercept.
R3 = Open (Size 0603)
R4 = Open (Size 0603)
R5, R6, R7, R8, R9
Bias Adjustment: The voltage on the VSUM and INPT Pins can be
altered using appropriate resistor values. R9 is populated with a decoupling capacitor to reduce noise pickup. The decoupling capacitor can be
removed when a fixed bias is applied to VSUM.
R5 = R6 = Open (Size 0603)
R7 = R8 = Open (Size 0603)
R9 = 0.1 µF (Size 0603)
C1, C2, C3, C4, C9
Supply Decoupling Capacitors
C1 = C4 = 0.1 µF (Size 0603)
C2 = C3 = 1 nF (Size 0603)
C9 = 10 nF (Size 0603)
C10
Photodiode Biaser Decoupling: Provides high frequency decoupling
of the adaptive bias output at Pin VPDB.
C10 = 0.1 µF (Size 0603)
C5, C6, C7, C8, R11, Output Filtering: Allows implementation of a variety of filter configR12, R13, R14
urations, from simple RC low-pass filters to three-pole Sallen and Key.
R11 = R13 = 0 Ω (Size 0603)
R12 = Open (Size 0603)
R14 = 0 Ω (Size 0603)
C5 = C6 = Open (Size 0603)
C7 = C8 = Open (Size 0603)
R15, C11
Input Filtering: Provides essential HF compensation at the input
Pin INPT.
R15 = 750 Ω (Size 0603)
C11 = 1 nF (Size 0603)
LK1, LK2
Guard/Shield Options: The shells of the SMA connectors used
for the input and the photodiode bias can be set to the voltage on the
VSUM Pin or connected to ground.
LK1 = Installed
LK2 = Open
REV. A
–19–
AD8304
OUTLINE DIMENSIONS
14-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-14)
Dimensions shown in millimeters
14
C02743–0–8/02(A)
5.10
5.00
4.90
8
4.50
4.40
4.30
6.40
BSC
1
7
PIN 1
1.05
1.00
0.80
0.65
BSC
1.20
MAX
0.15
0.05
0.30
0.19
0.20
0.09
SEATING
PLANE
8
0
0.75
0.60
0.45
COMPLIANT TO JEDEC STANDARDS MO-153AB-1
Revision History
Location
Page
8/02—Data Sheet changed from REV. 0 to REV. A.
Edits to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
New TPC 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Edits to TPC 7 caption . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Changes to TPC 19 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Edits to USING THE AD8304 section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Changes to Figure 3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Edits to Table I . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Edits to Table III . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
New Figure 12 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Changes to Figure 22 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
PRINTED IN U.S.A.
Changes to Table V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
–20–
REV. A