TI ADS807 12-bit, 53mhz sampling analog-to-digital converter Datasheet

ADS807
ADS
807
E
SBAS072A – JANUARY 1999 – REVISED JULY 2002
12-Bit, 53MHz Sampling
ANALOG-TO-DIGITAL CONVERTER
FEATURES
DESCRIPTION
● SPURIOUS-FREE DYNAMIC RANGE:
82dB at 10MHz fIN
● HIGH SNR: 67.5dB (2Vp-p), 69dB (3Vp-p)
● LOW POWER: 335mW
● INTERNAL OR EXTERNAL REFERENCE
● LOW DNL: 0.5LSB
● FLEXIBLE INPUT RANGE: 2Vp-p to 3Vp-p
● SSOP-28 PACKAGE
The ADS807 is a high-speed, high dynamic range,
12-bit pipelined Analog-to-Digital (A/D) converter. This converter includes a high-bandwidth track-and-hold that gives
excellent spurious performance up to and beyond the Nyquist
rate. The differential nature of this track-and-hold and A/D
converter circuitry minimizes even-order harmonics and gives
excellent common-mode noise immunity. The track-and-hold
can also be operated single-ended.
APPLICATIONS
● COMMUNICATIONS IF PROCESSING
● COMMUNICATIONS BASESTATIONS
● TEST EQUIPMENT
● MEDICAL IMAGING
● VIDEO DIGITIZING
● CCD DIGITIZING
The ADS807 provides for setting the full-scale range of the
converter without any external reference circuitry. The internal reference can be disabled allowing low drive, internal
references to be used for improved tracking in multichannel
systems.
The ADS807 provides an over-range indicator flag to indicate
an input signal that exceeds the full-scale input range of the
converter. This flag can be used to reduce the gain of front
end gain control circuitry. There is also an output enable pin
to allow for multiplexing and testability on a PC board.
The ADS807 employs digital error correction techniques to
provide excellent differential linearity for demanding imaging
applications.
CLK
ADS807
Timing
Circuitry
+3V
+2.5V
IN
+2V
+3V
+2.5V
T/H
IN
(Opt.)
Pipelined
A/D
Converter
+2V
Error
Correction
Logic
3-State
Outputs
D0
•
•
•
D11
Internal
Reference
CM
Optional External
Reference
INT/EXT
FSSEL
OE
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Copyright © 1999, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
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ELECTROSTATIC
DISCHARGE SENSITIVITY
ABSOLUTE MAXIMUM RATINGS(1)
+VS ....................................................................................................... +6V
Analog Input ........................................................... (–0.3V) to (+VS + 0.3V)
Logic Input ............................................................. (–0.3V) to (+VS + 0.3V)
Case Temperature ......................................................................... +100°C
Junction Temperature .................................................................... +150°C
Storage Temperature ..................................................................... +150°C
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and
installation procedures can cause damage.
NOTE: (1) Stresses above those listed under “Absolute Maximum Ratings”
may cause permanent damage to the device. Exposure to absolute maximum
conditions for extended periods may affect device reliability.
ESD damage can range from subtle performance degradation to
complete device failure. Precision integrated circuits may be
more susceptible to damage because very small parametric
changes could cause the device not to meet its published
specifications.
PACKAGE/ORDERING INFORMATION
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
ADS807E
ADS807E/1K
Tube, 50
Tape and Reel, 1000
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR(1)
ADS807E
SSOP-28
DB
–40°C to +85°C
ADS807E
"
"
"
"
"
NOTE: (1) For the most current specifications and package information refer to our web site at www.ti.com.
PIN CONFIGURATION
PIN DESCRIPTIONS
Top View
SSOP
GND
1
28
VDRV
Bit 1 (MSB)
2
27
+VS
Bit 2
3
26
GND
Bit 3
4
25
IN
Bit 4
5
24
IN
Bit 5
6
23
CM
Bit 6
7
22
REFT
ADS807E
2
Bit 7
8
21
REFB
Bit 8
9
20
GND
Bit 9 10
19
OE
Bit 10 11
18
INT/EXT
Bit 11 12
17
OTR
Bit 12 (LSB) 13
16
FSSEL
CLK 14
15
+VS
PIN
DESIGNATOR
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
GND
Bit 1
Bit 2
Bit 3
Bit 4
Bit 5
Bit 6
Bit 7
Bit 8
Bit 9
Bit 10
Bit 11
Bit 12
CLK
+VS
FSSEL
OTR
INT/EXT
19
20
21
22
23
24
25
26
27
28
OE
GND
REFB
REFT
CM
IN
IN
GND
+VS
VDRV
DESCRIPTION
Ground
Data Bit 1 (MSB)
Data Bit 2
Data Bit 3
Data Bit 4
Data Bit 5
Data Bit 6
Data Bit 7
Data Bit 8
Data Bit 9
Data Bit 10
Data Bit 11
Data Bit 12 (LSB)
Convert Clock
+5V Supply
HI = 3V, LO = 2V
Out-of-Range Indicator
Reference Select: HIGH or Floating = External LOW = Internal 50kΩ pull-up.
Output Enable
Ground
Bottom Reference/Bypass
Top Reference/Bypass
Common-Mode Voltage Output
Complementary Analog Input
Analog Input
Ground
+5V Supply
Logic Driver Supply Voltage
ADS807
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SBAS072A
ELECTRICAL CHARACTERISTICS
At TA = full specified temperature range, VS = +5V, differential input range = 2V to 3V for each input, sampling rate = 50MHz, unless otherwise noted.
ADS807E
PARAMETER
CONDITIONS
MIN
Ambient Air
–40
+85
°C
2
1.5
1.75
1
3
3.5
3.25
4
V
V
V
V
µA
MHz
MΩ || pF
53M
Samples/s
Clock Cycles
±1.0
±1.0
LSB
LSB
±4.0
LSBs
RESOLUTION
MAX
12 Tested
SPECIFIED TEMPERATURE RANGE
ANALOG INPUT
2V Full-Scale Input Range
2V Full-Scale Input Range
3V Full-Scale Input Range
3V Full-Scale Input Range
Analog Input Bias Current
Analog Input Bandwidth
Input Impedance
TYP
(Differential)
(Single-Ended)
(Differential)
(Single-Ended)
2Vp-p,
2Vp-p,
3Vp-p,
3Vp-p,
INT
INT
INT
INT
or
or
or
or
EXT
EXT
EXT
EXT
Ref
Ref
Ref
Ref
DYNAMIC CHARACTERISTICS
Differential Linearity Error (largest code error)
f = 1MHz
f = 10MHz
No Missing Codes
No MIssing Codes
Integral Nonlinearity Error, f = 1MHz
Spurious-Free Dynamic Range(1)
f = 1MHz (–1dB input)
f = 10MHz (–1dB input)
f = 20MHz (–1dB input)
f = 40MHz (undersampling)
f = 1MHz to 10MHz, fS = 40MHz
10k
6
±0.5
±0.5
Tested
Tested
±2.0
fS = 40MHz
fS = 50MHz,TA = +25°C
fS = 40MHz, Full Temp
67
2Vp-p, Single-Ended Input
62
2-Tone Intermodulation Distortion(3)
f = 12MHz and 13MHz (–7dB each tone)
Signal-to-Noise Ratio (SNR)
f = 1MHz (–1dB input)
f = 10MHz (–1dB input)
f = 20MHz (–dB input)
f = 40MHz (undersampling)
f = 1MHz to 10MHz, fS = 40MHz
f = 1MHz to 10MHz, fS = 40MHz
f = 1MHz (–1dB input)
f = 10MHz (–1dB input)
Signal-to-(Noise + Distortion) (SINAD)(4)
f = 1MHz (–1dBFS input)
f = 10MHz (–1dBFS input)
f = 20MHz (–1dBFS input)
f = 1MHz to 10MHz, fS = 40MHz
f = 1MHz to 10MHz, fS = 40MHz
Bits
1
270
1.25 || 3
CONVERSION CHARACTERISTICS
Sample Rate
Data Latency
63
63
2Vp-p, Single-Ended Input
63
60
3Vp-p
3Vp-p
61
61
2Vp-p, Single-Ended Input
63
60
UNITS
83
82
76
76
69
dBFS(2)
dBFS
dBFS
dBFS
dBFS
71
dBc
68
68
66
67
67.5
67
dB
dB
dB
dB
dB
dB
69
69
dB
dB
67
67
67
67
64
dB
dB
dB
dB
dB
f = 1MHz (–1dBFS input)
f = 10MHz (–dBFS Input)
3Vp-p
3Vp-p
69
69
dB
dB
Output Noise
Aperture Delay Time
Aperture Jitter
Over-Voltage Recovery Time
Input Grounded
0.2
2
1.2
2
LSBs rms
ns
ps rms
ns
DIGITAL INPUTS
Logic Family
Convert Command
High Level Input Current(5) (VIN = 5V)
Low Level Input Current (VIN = 0V)
High Level Input Voltage
Low Level Input Voltage
Input Capacitance
+2.4
+1.0
5
ADS807
SBAS072A
CMOS
Rising Edge of Convert Clock
+50
+10
Start Conversion
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µA
µA
V
V
pF
3
ELECTRICAL CHARACTERISTICS (Cont.)
At TA = full specified temperature range, VS = +5V, differential input range = 2V to 3V for each input, sampling rate = 50MHz, unless otherwise noted.
ADS807E
PARAMETER
DIGITAL OUTPUTS
Logic Family
Logic Coding
Low Output Voltage (IOL = 50µA)
Low Output Voltage, (IOL = 1.6mA)
High Output Voltage, (IOH = 50µA)
High Output Voltage, (IOH = 0.5mA)
Low Output Voltage, (IOL = 50µA)
High Output Voltage, (IOH = 50µA)
3-State Enable Time
3-State Disable Time
Output Capacitance
CONDITIONS
POWER-SUPPLY REQUIREMENTS
Supply Voltage: +VS
Supply Current: +IS
Power Dissipation: VDRV = 5V
VDRV = 3V
VDRV = 5V
VDRV = 3V
Thermal Resistance, θJA
SSOP-28
TYP
MAX
UNITS
+0.1
+0.2
V
V
V
V
V
V
ns
ns
pF
CMOS
Straight Offset Binary
VDRV = 5V
VDRV = 5V
VDRV = 5V
VDRV = 5V
VDRV = 3V
VDRV = 3V
OE = L(5)
OE = H(5)
ACCURACY (Internal Reference, 2Vp-p, Unless Otherwise Noted)
Zero Error (Referred to –FS)
at 25°C
Zero Error Drift (Referred to –FS)
at 25°C
Gain Error(6)
Gain Error Drift(6)
Gain Error(7)
at 25°C
Gain Error Drift(7)
Power-Supply Rejection of Gain
∆VS = ±5%
REFT Tolerance
2V Full-Scale
Deviation From Ideal 3.0V
3V Full-Scale
Deviation From Ideal 3.25V
REFB Tolerance
2V Full-Scale
3V Full-Scale
External REFT Voltage Range
External REFB Voltage Range
Reference Input Resistance
MIN
+4.9
+4.8
+0.1
+2.8
50
Deviation From Ideal 2.0V
Deviation From Ideal 1.75V
REFB + 0.4
1.70
Operating
Operating
External Reference
External Reference
Internal Reference
Internal Reference
+4.75
20
2
5
40
10
±1.0
16
±1.5
66
±1.0
23
70
±2.0
±2.5
±1.5
%FS
ppm/°C
%FS
ppm/°C
%FS
ppm/°C
dB
±10
±20
±65
±100
mV
mV
±10
±20
3
2
1
±65
±100
VS – 1.70
REFT – 0.4
mV
mV
V
V
kΩ
+5.0
60
305
290
350
335
+5.25
V
mA
mW
mW
mW
mW
50
360
350
390
380
°C/W
NOTES: (1) Spurious-Free Dynamic Range refers to the magnitude of the largest harmonic. (2) dBFS means dB relative to Full-Scale. (3) 2-tone intermodulation
distortion is referred to the largest fundamental tone. This number will be 6dB higher if it is referred to the magnitude of the 2-tone fundamental envelope.
(4) Effective number of bits (ENOB) is defined by as (SINAD – 1.76)/6.02. (5) A 50kΩ pull-down resistor is inserted internally on OE pin. (6) Includes internal reference.
(7) Excludes internal reference.
4
ADS807
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SBAS072A
TIMING DIAGRAM
N+2
N+1
Analog In
N+4
N+3
N
tD
N+5
tL
tCONV
N+7
N+6
tH
Clock
6 Clock Cycles
t2
Data Out
N–6
N–5
N–4
N–3
N–2
N–1
N
Data Invalid
SYMBOL
tCONV
tL
tH
tD
t1 (1)
t2 (1)
N+1
t1
DESCRIPTION
MIN
Convert Clock Period
Clock Pulse LOW
Clock Pulse HIGH
Aperture Delay
Data Hold Time, CL = 0pF
New Data Delay Time, CL = 15pF max
18.87
9.4
9.4
TYP
MAX
UNITS
100µs
ns
ns
ns
ns
ns
ns
tCONV/2
tCONV/2
2
2.7
12
NOTE: (1) t1 and t2 times are valid for VDRV voltages of +2.7V to +5V.
ADS807
SBAS072A
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5
TYPICAL CHARACTERISTICS
At TA = full specified temperature range, differential input range = 2V to 3V, sampling rate = 50MHz, and internal reference, unless otherwise noted.
SPECTRAL PERFORMANCE
SPECTRAL PERFORMANCE
0
0
fIN = 1MHz
SNR = 68dBFS
SFDR = 83dBFS
–10
–20
Magnitude (dBFS)
Magnitude (dBFS)
–20
fIN = 10MHz
SNR = 68dBFS
SFDR = 82dBFS
–10
–30
–40
–50
–60
–70
–80
–30
–40
–50
–60
–70
–80
–90
–90
–100
–100
0
5
10
15
20
25
0
5
10
Frequency (MHz)
fIN = 21MHz
SNR = 68dBFS
SFDR = 77dBFS
f1 = 12MHz
f2 = 13MHz
IMD(3) = –71dBc
–20
Magnitude (dBc)
Magnitude (dBFS)
25
0
–20
–40
–60
–40
–60
–80
–80
–100
–100
0
5
10
15
20
0
25
5
10
15
20
Frequency (MHz)
Frequency (MHz)
SPECTRAL PERFORMANCE
(Sampling Frequency = 27MHz)
SPECTRAL PERFORMANCE
(Single-Ended, 2Vp-p)
0
25
0
fIN = 10MHz
SNR = 68dBFS
SFDR = 81dBFS
fIN = 10MHz
SNR = 68dBFS
SFDR = 62dBFS
–20
Magnitude (dBFS)
–20
Magnitude (dBFS)
20
2-TONE INTERMODULATION DISTORTION
SPECTRAL PERFORMANCE
0
–40
–60
–80
–40
–60
–80
–100
–100
0
4.5
9.0
13.5
0
Frequency (MHz)
6
15
Frequency (MHz)
4.5
9.0
13.5
Frequency (MHz)
ADS807
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SBAS072A
TYPICAL CHARACTERISTICS (Cont.)
At TA = full specified temperature range, differential input range = 2V to 3V, sampling rate = 50MHz, and internal reference, unless otherwise noted.
SPECTRAL PERFORMANCE
(Sampling Frequency = 53MHz)
UNDERSAMPLING
(Sampling Frequency = 27MHz)
0
0
fIN = 21MHz
SNR = 68dBFS
SFDR = 72dBFS
–20
Magnitude (dBFS)
Magnitude (dBFS)
–20
fIN = 50MHz
SNR = 65dBFS
SFDR = 73dBFS
–40
–60
–40
–60
–80
–80
–100
–100
0
5.3
10.6
15.9
21.2
26.5
0
4.5
Frequency (MHz)
9.0
13.5
Frequency (MHz)
SWEPT POWER SFDR
OUTPUT NOISE HISTOGRAM (DC INPUT)
100
800k
fIN = 10MHz
dBFS
3V Full-Scale
80
SFDR (dBFS, dBc)
Counts
600k
400k
200k
60
dBc
40
20
0
N–2
0
N–1
N
N+1
N+2
0
–10
–20
Code
–30
–40
–50
–60
Input Amplitude (dBFS)
DYNAMIC PERFORMANCE
vs SAMPLING FREQUENCY
(Differential Input)
SINAD vs SAMPLING FREQUENCY
(Differential Input)
75
90
fIN = 5MHz
fIN = 5MHz
85
70
80
SINAD (dB)
SFDR, SNR (dB)
SFDR (2Vp-p)
SFDR (3Vp-p)
75
SNR (3Vp-p)
3Vp-p
2Vp-p
65
70
60
65
SNR (2Vp-p)
55
60
35
40
45
50
55
60
ADS807
SBAS072A
35
40
45
50
55
60
Sampling Frequency (MHz)
Sampling Frequency (MHz)
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7
TYPICAL CHARACTERISTICS (Cont.)
At TA = full specified temperature range, differential input range = 2V to 3V, sampling rate = 50MHz, and internal reference, unless otherwise noted.
DIFFERENTIAL LINEARITY ERROR
INTEGRAL LINEARITY ERROR
2.0
2.0
fIN = 1MHz
fIN = 1MHz
1.5
1.0
1.0
0.5
0.5
ILE (LSB)
DLE (LSB)
1.5
0
–0.5
0
–0.5
–1.0
–1.0
–1.5
–1.5
–2.0
–2.0
0
1024
2048
3072
4096
0
1024
Output Codes
4096
2.0
fIN = 10MHz
fIN = 10MHz
1.5
1.5
1.0
1.0
0.5
0.5
ILE (LSB)
DLE (LSB)
3072
INTEGRAL LINEARITY ERROR
DIFFERENTIAL LINEARITY ERROR
2.0
0
–0.5
0
–0.5
–1.0
–1.0
–1.5
–1.5
–2.0
–2.0
0
1024
2048
3072
0
4096
1024
2048
3072
Output Codes
Output Codes
DIFFERENTIAL LINEARITY ERROR
(Single-Ended, Input Sampling Frequency = 40MHz)
INTEGRAL LINEARITY ERROR
(Sampling Frequency = 40MHz)
1.000
4096
2.0
fIN = 10MHz
fIN = 10MHz
1.5
0.500
1.0
ILE (LSB)
DLE (LSB)
2048
Output Codes
0
0.5
0
–0.5
–1.0
–0.500
–1.5
–2.0
–1.000
0
1024
2048
3072
4096
8
0
1024
2048
3072
4096
Output Codes
Output Codes
ADS807
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SBAS072A
• improves the noise immunity based on the converter’s
common-mode input rejection
APPLICATION INFORMATION
THEORY OF OPERATION
The ADS807 is a high-speed, CMOS A/D converter which
employs a pipelined converter architecture consisting of 12
internal stages. Each stage feeds its data into the digital
error correction logic ensuring excellent differential linearity
and no missing codes at the 12-bit level. The output data
becomes valid after the rising clock edge (see Timing Diagram). The pipeline architecture results in a data latency of
6 clock cycles.
The analog input of the ADS807 consists of a differential
track-and-hold circuit. The differential topology along with
tightly matched poly-poly capacitors produce a high level of
AC performance at high sampling rates and in undersampling
applications.
Both inputs (IN, IN) require external biasing using a common-mode voltage that is typically at the mid-supply level
(+VS/2).
DRIVING THE ANALOG INPUTS
The analog inputs of the ADS807 are a very high impedance.
They should be driven through an R-C network designed to
pass the highest frequency of interest. This prevents highfrequency noise in the input from affecting SFDR and SNR.
The ADS807 can be used in a wide variety of applications
and deciding on the best performing analog interface circuit
depends on the type of application. The circuit definition
should include considerations of input frequency spectrum
and amplitude, single-ended or differential drive, and available power supplies. For example, communication (frequency
domain) applications process frequency bands not including
DC. In imaging (time domain) applications, the input DC
component must be maintained into the A/D converter.
Features of the ADS807, including full-scale select (FSSEL),
external reference, and CM output provide flexibility to accommodate a wide range of applications. The ADS807
should be configured to meet application objectives while
observing the headroom requirements of the driving amplifiers to yield the best overall performance.
The ADS807 input structure allows it to be driven either
single-ended or differentially. Differential operation of the
ADS807 requires an in-phase input signal and a 180° out-ofphase part simultaneously applied to the inputs (IN, IN). The
differential operation offers a number of advantages which,
in most applications, will be instrumental in achieving the
best dynamic performance of the ADS807:
• the signal swing is half of that required for the singleended operation and therefore, is less demanding to
achieve while maintaining good linearity performance from
the signal source
• the reduced signal swing allows for more headroom in the
interface circuitry and therefore, a wider selection of the
best suitable driver op amp
• even-order harmonics are minimized
Using the single-ended mode, the signal is applied to one of
the inputs, while the other input is biased with a DC voltage
to the required common-mode level. Both inputs are equal in
terms of their impedance and performance, except that
applying the signal to the complementary input (IN) instead
of the IN input will invert the input signal relative to the output
code. For example, in case the input driver operates in
inverting mode, using IN as the signal input will restore the
phase of the signal to its original orientation. Time-domain
applications may benefit from a single-ended interface configuration and its reduced circuit complexity. While maintaining good SNR, driving the ADS807 with a single-ended
signal will result in a reduction of the distortion performance.
Employing dual-supply amplifiers and AC-coupling will usually yield the best results, while DC-coupling and/or singlesupply amplifiers impose additional design constraints due to
their headroom requirements, especially when selecting the
3Vp-p input range. However, single-supply amplifiers have
the advantage of inherently limiting their output swing to
within the supply rails. Alternatively, a voltage-limiting amplifier, like the OPA688, may be considered to set fixed-signal
limits and avoid any severe over-range condition for the A/D
converter.
The full-scale input range of the ADS807 is defined by the
reference voltages. For example, setting the range select pin
to FSSEL = LOW, and using the internal references
(REFT = +3.0V and REFTB = +2.0V), the full-scale range is
defined to: FSR = 2 • (REFT – REFB) = 2Vp-p.
The trade-off of the differential input configuration versus the
single-ended is its higher complexity. In either case, the
selection of the driver amplifier should be such that the
amplifier’s performance will not degrade the A/D converter’s
performance. The ADS807 operates on a single power
supply, which requires a level shift to a ground-based bipolar
input signals to comply with its input voltage range requirements.
The input of the ADS807 is of a capacitive nature and the
driving source needs to provide the current to charge or
discharge the input sampling capacitor while the track-andhold is in track mode. This effectively results in a dynamic
input impedance which depends on the sampling frequency.
It most applications, it is recommended to add a series
resistor, typically 20Ω to 50Ω, between the drive source and
the converter inputs. This will isolate the capacitive input
from the source, which can be crucial to avoid gain peaking
when using wideband operational amplifiers. Secondly, it will
create a 1st-order, low-pass filter in conjunction with the
specified input capacitance of the ADS807. Its cutoff frequency can be adjusted even further by adding an external
shunt capacitor from each signal input to ground. The optimum values of this R-C network depend on a variety of
factors which include the ADS807 sampling rate, the selected op amp, the interface configuration, and the particular
application (time domain versus frequency domain). Generally, increasing the size of the series resistor and/or capacitor
ADS807
SBAS072A
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9
will improve the SNR performance, but depending on the
signal source, large resistor values may be detrimental to
achieving good harmonic distortion. In any case, optimizing
the R-C values for the specific application is encouraged.
Transformer Coupled, Single-Ended to Differential
Configuration
If the application requires a signal conversion from a singleended source to drive the ADS807 differentially, an RF
transformer might be a good solution. The selected transformer must have a center tap in order to apply the commonmode DC voltage necessary to bias the converter inputs. ACgrounding the center tap will generate the differential signal
swing across the secondary winding. Consider a step-up
transformer to take advantage of a signal amplification without the introduction of another noise source. Furthermore,
the reduced signal swing from the source may lead to
improved distortion performance.
The differential input configuration provides a noticeable
advantage of achieving good SFDR over a wide range of
input frequencies. In this mode, both inputs of the ADS807
see matched impedances. Figure 1 shows the schematic for
the suggested transformer coupled interface circuit. The
component values of the R-C low-pass may be optimized
depending on the desired roll-off frequency. The resistor
across the secondary side (RT) should be calculated using
the equation RT = n2 • RG to match the source impedance
(RG) for good power transfer and VSWR.
The circuit example of Figure 1 shows the voltage-feedback
amplifier OPA680 driving the RF transformer, which converts
the single-ended signal into a differential one. The OPA680
can be employed for either single- or dual-supply operation.
For details on how to optimize its frequency response, refer
to the OPA680 data sheet (SBOS083), available at
www.ti.com. With the 49.9Ω series output resistor, the amplifier emulates a 50Ω source (RG). Any DC content of the
signal can be easily blocked by a capacitor (0.1µF) and to
also to avoid DC loading of the op amp’s output stage.
AC-Coupled, Single-Ended-to-Differential Interface
with Dual-Supply Op Amps
Communications applications, in particular, demand a very
high dynamic range and low levels of intermodulation distortion, but usually allow the input signal to be AC-coupled into
the A/D converter. Appropriate driver amplifiers need to be
selected to maintain the excellent distortion performance of
the ADS807. Often, these op amps deliver the lowest distortion with a small, ground-centered signal swing that requires
dual power supplies. Because of the AC-coupling, this requirement can be easily accomplished and the needed level
shifting of the input signal can be implemented without
affecting the driver circuit.
See Figure 2 for an example of such an interface circuit
specifically designed to maximize the dynamic performance.
The voltage feedback amplifier, OPA642, maintains an excellent distortion performance for input frequencies of up to
15MHz. The two amplifiers (A1, A2) are configured as an
inverting and noninverting gain stage to convert the input
signal from single-ended to differential. The nominal gain for
this stage is set to +2V/V. The outputs of the OPA642s are
AC-coupled to the converter’s differential inputs. This will
keep the distortion performance at its best since the signal
range stays within the linear region of the op amp and
sufficient headroom to the supply rails can be maintained.
Four resistors located between the top (REFT) and bottom
(REFB) reference shift the input signal to a common-mode
voltage of approximately +2.5V.
The interface circuit of Figure 2 can be modified to extend
the bandwidth to approximately 25MHz by replacing the
OPA642 with its decompensated version, the OPA643. The
OPA643 provides the necessary slew rate for a low distortion
front end to the ADS807. With a minimum gain stability of +3,
the gain resistors have to be modified, as well as optimizing
the series resistor and shunt capacitance at each of the
converter inputs.
RG
VIN
49.9Ω
0.1µF 1:n
24.9Ω
IN
OPA680
47pF
R1
ADS807E
RT
24.9Ω
R2
IN
CM
+2.5V
47pF
+
10µF
0.1µF
FIGURE 1. Converting a Single-Ended Input Signal into a Differential Signal Using a RF-Transformer.
10
ADS807
www.ti.com
SBAS072A
402Ω
200Ω
VIN
A1
OPA642
1.82kΩ
0.1µF
16.5Ω
1.82kΩ
REFT
IN
100pF
402Ω
ADS807E
402Ω
A2
OPA642
0.1µF
16.5Ω
IN
100pF
1.82kΩ
REFB
1.82kΩ
FIGURE 2. AC-Coupled Differential Driver Interface with OPA642.
RF
499Ω
0.1µF
RIN
249Ω
RS
24.9Ω
VIN
A1
IN
RP
499Ω
499Ω
68pF
VCM = +2.5V
ADS807E
CM
0.1µF
+5V
RS
24.9Ω
A2
OPA2681
IN
68pF
RF
499Ω
RG
499Ω
RP
499Ω
0.1µF
FIGURE 3. AC-Coupled, Differential Interface for Single-Supply Operation.
AC-Coupled, Single-Ended-to-Differential Interface
for Single-Supply Operation
The previously discussed interface circuit can be modified if
the system only allows for a single-supply operation, e.g.,
VS = +5V. Single-supply operation requires the driver amplifier to be biased as well in order to process a bipolar input
signal. Typically, single-supply amplifiers do not achieve
distortion performance as well as dual-supply op amps. The
driver amplifier’s output swing must exceed the full-scale
input range of the converter. In addition, dual op amps, such
as the current-feedback OPA2681, should be considered
since they provide the closest open-loop gain and phase
matching between the two channels. Shown in Figure 3 is a
single-supply interface circuit for an AC-coupled input signal.
With the ADS807 set to the 2Vp-p input range, the top and
bottom references (REFT, REFB) provide an output voltage
of +3.0V and +2.0V, respectively. The CM output of the
ADS807 is used to bias the inputs of the driving amplifiers.
Using the OPA2681 on a single +5V supply, its ideal common-mode point is +2.5V, which coincides with the recom-
mended common-mode input level for the ADS807, thus
obviating the need for coupling capacitors between the
amplifiers and the converter.
The addition of a small series resistor (RS) between the
output of the op amps and the input of the ADS807 will be
beneficial in almost all interface configurations. It will decouple
the op amp’s output from the capacitive load and avoid gain
peaking, which can result in increased noise. For best
spurious and distortion performance, the resistor value should
be kept below 100Ω. Furthermore, the series resistor in
combination with the shunt capacitor, establishes a passive
low-pass filter limiting the bandwidth for the wideband noise,
thus improving the SNR. The spurious-free dynamic range of
this single-supply front end is limited by the 2nd-harmonic
distortion. An improvement of several dB may be realized by
adding a pull-down resistor (RP) at the output of each
amplifier. This pulls a DC bias current out of the output stage
of the amplifier. It is set to approximately 5mA in Figure 3, but
will vary depending on the amplifier used.
ADS807
SBAS072A
www.ti.com
11
Single-Ended, AC-Coupled, Dual-Supply Interface
DC-Coupled, Differential Driver with Level Shift
The circuit provided in Figure 4 shows typical connections for
using the ADS807 in a single-ended input configuration. The
bias requirements for AC-coupling are provided by a single
resistor to the CM output lead. The single-ended mode of
operation should be considered for ease of interface complexity and applications where the dynamic performance can
be compromised. The series resistor RS, along with the shunt
capacitance, provide the means to adjust the bandwidth and
optimize the performance towards good signal-to-noise ratio.
In addition, the amplifier configuration can be easily modified
for an anti-aliasing filter based on a 2nd-order Sallen-Key or
Multiple-Feedback topology.
Several applications will require that the bandwidth of the
signal path include DC, in which case, the signal has to be
DC-coupled to the A/D converter. An op amp based interface
circuit can be configured to scale and level shift the input
signal to be compatible with the selected input range of the
A/D converter. The circuit shown in Figure 5 employs a dual
op amp, OPA2681, to drive the input of the ADS807 differentially. The single-supply, general-purpose op amp OPA234
is added to buffer the common-mode voltage of +2.5V,
available at the CM pin, and apply it to the input of the driver
amplifier. This sets the correct DC voltage to bias the inputs
of the ADS807. It should be noted that any DC voltage
differences between the IN and IN inputs of the ADS807 will
result in an offset error.
The interface example shown in Figure 4 operates with the
full-scale range of the ADS807 set to 2Vp-p, leaving sufficient headroom for the output of the OPA642 to drive the
converter and maintain low signal distortion.
Using the OPA2681, this circuit can be operated either with
a single or a dual ±5V supply.
+5V
RS
16.5Ω
VIN
0.1µF
IN
OPA642
68pF
–5V
ADS807E
RF
402Ω
1.82kΩ
CM
IN
0.1µF
RG
402Ω
FIGURE 4. AC-Coupling the Dual-Supply Amplifier OPA642 to the ADS807 for a 2Vp-p Full-Scale Input Range.
499Ω
249Ω
24.9Ω
VIN
IN
OPA2681
22pF
499Ω
249Ω
ADS807E
499Ω
IN
249Ω
24.9Ω
CM
22pF
499Ω
24.9Ω
OPA234
249Ω
0.1µF
0.1µF
0.1µF
1kΩ
FIGURE 5. DC-Coupled Input Driver with Level Shifting.
12
ADS807
www.ti.com
SBAS072A
REFERENCE OPERATION
USING EXTERNAL REFERENCES
The internal reference consists of a bandgap voltage reference, the drivers for the top and bottom reference, and the
resistive reference ladder. The bandgap reference circuit
includes logic functions that allow setting the analog input
swing of the ADS807 to a differential full-scale range of either
2Vp-p or 3Vp-p by simply tying the FSSEL pin to a LOW or
HIGH potential, respectively. While operating the ADS807 in
the external reference mode, the buffer amplifiers for the
REFT and REFB are disabled. The ADS807 has an internal
50kΩ pull-down resistor at the range select pin (RSEL).
Therefore, this pin can be either hardwired to ground or left
unconnected, which will default the converter to a 2Vp-p fullscale input range (FSR). While set for the 2Vp-p range, the
top and bottom reference voltages will be REFT = +3.0V and
REFB = +2.0V. Switching to the 3Vp-p range changes those
voltages to REFT = +3.25V and REFB = +1.75V. The
reference buffers can be utilized to supply up to 1mA (sink
and source) to external circuitry. To ensure proper operation
with any reference configuration, it is necessary to provide
solid bypassing at all reference pins in order to keep the
clock feedthrough to a minimum, as shown in Figure 6. Good
performance requires using 0.1µF low inductance capacitors. All bypassing capacitors should be located as close to
their respective pins as possible.
For even more design flexibility, the internal reference can
be disabled and an external reference voltage used. The
utilization of an external reference may be considered for
applications requiring higher accuracy, improved temperature performance, or a wide adjustment range of the
converter’s full-scale range. In multichannel applications, the
use of a common external reference has the benefit of
obtaining better matching and drift of the full-scale range
between converters. Figure 7 gives an example of an external reference circuit using a single-supply, low-power, dual
op amp (OPA2234).
(1)
CM
REFB
(1)
+
0.1µF
10µF
(1)
+
+
0.1µF
10µF
The logic level applied to the INT/EXT pin of the ADS807
determines if the converter operates with either the built-in
reference or external reference voltages. Because this function
pin has an internal 50kΩ pull-up resistor, the default configuration is external reference mode. Grounding this pin will
activate the internal reference option.
The input track-and-hold amplifier is differential. A positive
1Vp-p on the IN and its compliment, a negative
1Vp-p, on the IN (see Figure 3) results in 2Vp-p on the
output of the track-and-hold. Likewise, 2Vp-p on the IN and
0Vp-p on the IN (see Figure 4) results in 2Vp-p on the
output of the track-and-hold. Therefore, the reference voltages, REFT and REFB, are the same for both differential
and single-ended inputs, see Table I.
ADS807
REFT
The external references can vary as long as the value of the
external top reference (REFT EXT) stays within the range of
VS – 1.70V and REFB + 0.4V, and the external bottom
reference (REFB EXT) stays within 1.70V and REFT – 0.4V.
Note that the function of the range selector pin (FSSEL) is
disabled while the converter operates in external reference
mode. Setting the ADS807 for external reference mode requires the INT/EXT pin (pin 18) to be HIGH.
0.1µF
10µF
NOTE: (1) Optional.
FIGURE 6. Recommended Bypassing for the Reference Pins.
The external references may be changed for different tasks.
The ADS807 will follow the external references with a latency
of 8 to 10 clock cycles. If it is desired to use INT/EXT and FSSEL
to change the configuration of a circuit for different tasks, a
large amount of time must be allowed. This time could be
hundreds of microseconds. Refer to the diagram on the front
page. Note that there is no disconnect for external references.
+5V
+5V
OPA2234
A1
4.7kΩ
< 3.30V
Top Reference
R3
R4
R1
REF1004
+2.5V
+
10µF
R2
0.1µF
OPA2234
A2
> 1.70V
Bottom Reference
FIGURE 7. Example for an External Reference Driver Using the Dual, Single-Supply Op Amp, OPA2234.
ADS807
SBAS072A
www.ti.com
13
INPUT
REFERENCE
IN (Pin-25)
IN (Pin-24)
2Vp-p Differential
1Vp-p Times 2 Inputs
Internal
or External
2V to 3V
3V to 2V
REFT REFB
+3V
+2V
2Vp-p Single-Ended
2Vp-p Times 1 Input
Internal
or External
1.5V to 3.5V
2.5VDC
+3V
+2V
3Vp-p Differential
1.5Vp-p Times 2 Inputs
Internal
or External
3Vp-p Single-Ended
3Vp-p Times 1 Input
Internal
or External
1.75V to 3.35V 3.25V to 1.75V +3.25V +1.75V
1V to 4V
2.5VDC
SINGLE-ENDED INPUT
(IN = CM, Pin-23)
STRAIGHT OFFSET BINARY
(SOB)
+FS – 1LSB (IN = CMV + FSR/2)
1111 1111 1111
+1/2 FS
1100 0000 0000
Bipolar Zero (IN = VCM)
1000 0000 0000
–1/2 FS
0100 0000 0000
–FS (IN = CMV – FSR/2)
0000 0000 0000
+3.25V +1.75V
TABLE II. Coding Table for Single-Ended Input Configuration
with IN Tied to the Common-Mode Voltage.
TABLE I. Reference Voltages for Input Signal Ranges.
If it is desired to switch between internal and external references, disconnect switches must be added between the external references and the ADS807.
DIGITAL INPUTS AND OUTPUTS
Clock Input Requirements
Clock jitter is critical to the SNR performance of high-speed,
high-resolution A/D converters. Clock jitter leads to aperture
jitter (tA), which adds noise to the signal being converted. The
ADS807 samples the input signal on the rising edge of the CLK
input. Therefore, this edge should have the lowest possible
jitter. The jitter noise contribution to total SNR is given by the
following equation. If this value is near your system requirements, input clock jitter must be reduced.
Jitter SNR = 20 log
1
rms signal to rms noise
2π ƒ IN t A
where: ƒIN is input signal frequency
tA is rms clock jitter
STRAIGHT OFFSET BINARY
(SOB)
+FS – 1LSB (IN = +3V, IN = +2V)
1111 1111 1111
+1/2 FS
1100 0000 0000
Bipolar Zero (IN = IN = VCM)
1000 0000 0000
–1/2 FS
0100 0000 0000
–FS (IN = +2V, IN = +3V)
0000 0000 0000
TABLE III. Coding Table for Single-Ended Input Configuration
with IN Tied to the Common-Mode Voltage.
loading will cause larger dynamic currents as the digital
outputs are changing. Those high current surges can feed
back to the analog portion of the ADS807 and affect the
performance. If necessary, external buffers or latches close
to the converter’s output pins may be used to minimize the
capacitive loading. They also provide the added benefit of
isolating the ADS807 from high-frequency digital noise on
the bus coupling back into the converter.
Digital Output Driver Supply (VDRV)
Particularly in undersampling applications, special consideration should be given to clock jitter. The clock input should be
treated as an analog input in order to achieve the highest
level of performance. Any overshoot or undershoot of the
clock signal may cause degradation of the performance.
When digitizing at high sampling rates, the clock should have
50% duty cycle (tH = tL), along with fast rise and fall times of
2ns or less.
Over-Range Indicator (OTR)
If the analog input voltage exceeds the set full-scale range,
an over-range condition exists. The ‘OTR’ pin of the ADS807
can be used to monitor any such out-of-range condition. This
‘OTR’ output is updated along with the data output corresponding to the particular sampled analog input voltage.
Therefore, the OTR data is subject to the same pipeline
delay as the digital data. The OTR output is LOW when the
input voltage is within the defined input range. It will go to
HIGH if the applied signal exceeds the full-scale range.
Data Outputs
The output data format of the ADS807 is in positive Straight
Offset Binary code, as shown in Table II and Table III. This
format can easily be converted into the Binary Two’s Complement code by inverting the MSB.
It is recommended that the capacitive loading on the data
lines be as low as possible (< 15pF). Higher capacitive
14
DIFFERENTIAL INPUT
The ADS807 features a dedicated supply pin for the output
logic drivers, VDRV, which is not internally connected to the
other supply pins. Setting the voltage at VDRV to +5V or
+3V, the ADS807 produces corresponding logic levels and
can directly interface to the selected logic family. The output
stages are designed to supply sufficient current to drive a
variety of logic families. However, it is recommended to use
the ADS807 with +3V logic supply. This will lower the power
dissipation in the output stages due to the lower output swing
and reduce current glitches on the supply line which may
affect the AC performance of the converter. In some applications, it might be advantageous to decouple the VDRV pin
with additional capacitors or a pi-filter.
GROUNDING AND DECOUPLING
Proper grounding, bypassing, short trace lengths, and the
use of power and ground planes are particularly important for
high-frequency designs. Multilayer PC boards are recommended for best performance since they offer distinct advantages such as minimizing ground impedance, separation of
signal layers by ground layers, etc. The ADS807 should be
treated as an analog component. Whenever possible, the
supply pins should be powered by the analog supply. This
will ensure the most consistent results, since digital supply
lines often carry high levels of noise which otherwise would
be coupled into the converter and degrade the achievable
ADS807
www.ti.com
SBAS072A
performance. All ground connections on the ADS807 are
internally joined together eliminating the need for split ground
planes. The ground pins (1, 20, 26) should directly connect
to an analog ground plane which covers the PC board area
under the converter. While designing the layout, it is important to keep the analog signal traces separated from any
digital lines to prevent noise coupling onto the analog signal
path. Because of the its high sampling rate, the ADS807
generates high frequency current transients and noise (clock
feedthrough) that are fed back into the supply and reference
lines. This requires that all supply and reference pins are
sufficiently bypassed. Figure 8 shows the recommended
decoupling scheme for the ADS807. In most cases, 0.1µF
ceramic chip capacitors at each pin are adequate to keep the
impedance low over a wide frequency range. Their effectiveness largely depends on the proximity to the individual
supply pin. Therefore, they should be located as close to the
supply pins as possible. If system supplies are not a low
enough impedance, adding a small tantalum capacitor will
yield the best results.
ADS807
+VS
1, 20
+VS
GND
15
0.1µF
0.1µF
VDRV
26
28
0.1µF
10µF(1)
+
NOTE: (1) Optional.
+5V
+3V/+5V
FIGURE 8. Recommended Bypassing for the Supply Pins.
ADS807
SBAS072A
GND
27
www.ti.com
15
PACKAGE DRAWING
MSSO002E – JANUARY 1995 – REVISED DECEMBER 2001
DB (R-PDSO-G**)
PLASTIC SMALL-OUTLINE
28 PINS SHOWN
0,38
0,22
0,65
28
0,15 M
15
0,25
0,09
8,20
7,40
5,60
5,00
Gage Plane
1
14
0,25
A
0°– 8°
0,95
0,55
Seating Plane
2,00 MAX
0,10
0,05 MIN
PINS **
14
16
20
24
28
30
38
A MAX
6,50
6,50
7,50
8,50
10,50
10,50
12,90
A MIN
5,90
5,90
6,90
7,90
9,90
9,90
12,30
DIM
4040065 /E 12/01
NOTES: A.
B.
C.
D.
16
All linear dimensions are in millimeters.
This drawing is subject to change without notice.
Body dimensions do not include mold flash or protrusion not to exceed 0,15.
Falls within JEDEC MO-150
ADS807
www.ti.com
SBAS072A
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