AD AD636 Low level, true rms-to-dc converter Datasheet

a
FEATURES
True RMS-to-DC Conversion
200 mV Full Scale
Laser-Trimmed to High Accuracy
0.5% Max Error (AD636K)
1.0% Max Error (AD636J)
Wide Response Capability:
Computes RMS of AC and DC Signals
1 MHz –3 dB Bandwidth: V RMS >100 mV
Signal Crest Factor of 6 for 0.5% Error
dB Output with 50 dB Range
Low Power: 800 ␮A Quiescent Current
Single or Dual Supply Operation
Monolithic Integrated Circuit
Low Cost
Available in Chip Form
Low Level,
True RMS-to-DC Converter
AD636
PIN CONNECTIONS &
FUNCTIONAL BLOCK DIAGRAM
IOUT
VIN 1
NC 2
–VS 3
ABSOLUTE
VALUE
PRODUCT DESCRIPTION
The AD636 is a low power monolithic IC which performs true
rms-to-dc conversion on low level signals. It offers performance
which is comparable or superior to that of hybrid and modular
converters costing much more. The AD636 is specified for a
signal range of 0 mV to 200 mV rms. Crest factors up to 6 can
be accommodated with less than 0.5% additional error, allowing
accurate measurement of complex input waveforms.
The low power supply current requirement of the AD636, typically 800 µA, allows it to be used in battery-powered portable
instruments. A wide range of power supplies can be used, from
± 2.5 V to ±16.5 V or a single +5 V to +24 V supply. The input
and output terminals are fully protected; the input signal can
exceed the power supply with no damage to the device (allowing
the presence of input signals in the absence of supply voltage)
and the output buffer amplifier is short-circuit protected.
The AD636 includes an auxiliary dB output. This signal is
derived from an internal circuit point which represents the logarithm of the rms output. The 0 dB reference level is set by an
externally supplied current and can be selected by the user
to correspond to any input level from 0 dBm (774.6 mV) to
–20 dBm (77.46 mV). Frequency response ranges from 1.2 MHz
at a 0 dBm level to over 10 kHz at –50 dBm.
The AD636 is designed for ease of use. The device is factorytrimmed at the wafer level for input and output offset, positive
and negative waveform symmetry (dc reversal error), and fullscale accuracy at 200 mV rms. Thus no external trims are required to achieve full-rated accuracy.
AD636 is available in two accuracy grades; the AD636J total
error of ± 0.5 mV ± 0.06% of reading, and the AD636K
+
AD636
12 NC
+
BUF
–
10kV
10 COMMON
9
RL
8
IOUT
10kV
NC = NO CONNECT
–
BUF
BUF OUT
CURRENT
MIRROR
10kV
11 NC
CURRENT
MIRROR
BUF OUT 6
BUF IN 7
10kV
COMMON
SQUARER
DIVIDER
CAV 4
dB 5
14 +VS
13 NC
AD636
BUF IN
RL
SQUARER
DIVIDER
+VS
dB
ABSOLUTE
VALUE
CAV
VIN
–VS
is accurate within ± 0.2 mV to ± 0.3% of reading. Both versions
are specified for the 0°C to +70°C temperature range, and are
offered in either a hermetically sealed 14-pin DIP or a 10-lead
TO-100 metal can. Chips are also available.
PRODUCT HIGHLIGHTS
1. The AD636 computes the true root-mean-square of a complex ac (or ac plus dc) input signal and gives an equivalent dc
output level. The true rms value of a waveform is a more
useful quantity than the average rectified value since it is a
measure of the power in the signal. The rms value of an
ac-coupled signal is also its standard deviation.
2. The 200 millivolt full-scale range of the AD636 is compatible
with many popular display-oriented analog-to-digital converters. The low power supply current requirement permits
use in battery powered hand-held instruments.
3. The only external component required to perform measurements to the fully specified accuracy is the averaging capacitor. The value of this capacitor can be selected for the desired
trade-off of low frequency accuracy, ripple, and settling time.
4. The on-chip buffer amplifier can be used to buffer either the
input or the output. Used as an input buffer, it provides
accurate performance from standard 10 MΩ input attenuators. As an output buffer, it can supply up to 5 milliamps of
output current.
5. The AD636 will operate over a wide range of power supply
voltages, including single +5 V to +24 V or split ± 2.5 V to
± 16.5 V sources. A standard 9 V battery will provide several
hundred hours of continuous operation.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1999
AD636–SPECIFICATIONS (@ +25ⴗC, and +V = +3 V, –V = –5 V, unless otherwise noted)
S
Model
Min
AD636J
Typ
CONVERSION ACCURACY
Total Error, Internal Trim1, 2
vs. Temperature, 0°C to +70°C
vs. Supply Voltage
dc Reversal Error at 200 mV
Total Error, External Trim 1
± 0.1 ± 0.01
± 0.2
± 0.3 ± 0.3
ERROR VS. CREST FACTOR 3
Crest Factor 1 to 2
Crest Factor = 3
Crest Factor = 6
FREQUENCY RESPONSE2, 4
Bandwidth for 1% Additional Error (0.09 dB)
VIN = 10 mV
VIN = 100 mV
VIN = 200 mV
± 3 dB Bandwidth
VIN = 10 mV
VIN = 100 mV
VIN = 200 mV
OUTPUT CHARACTERISTICS2
Offset Voltage, VIN = COM
vs. Temperature
vs. Supply
Voltage Swing
+3 V, –5 V Supply
± 5 V to ± 16.5 V Supply
Output Impedance
0.3
0.3
8
I OUT TERMINAL
I OUT Scale Factor
I OUT Scale Factor Tolerance
Output Resistance
Voltage Compliance
BUFFER AMPLIFIER
Input and Output Voltage Range
Input Offset Voltage, RS = 10k
Input Bias Current
Input Resistance
Output Current
2
1
–20
8
–VS to (+VS
–2 V)
POWER SUPPLY
Voltage, Rated Performance
Dual Supply
Single Supply
Quiescent Current6
ⴞ0.2 ⴞ0.5
± 0.1 ± 0.005
mV ± % of Reading
mV ± % of Reading/°C
mV ± % of Reading/V
% of Reading
mV ± % of Reading
± 0.1 ± 0.01
± 0.1
± 0.1 ± 0.2
Specified Accuracy
–0.2
–0.5
% of Reading
% of Reading
25
ms/µF CAV
0 to 200
0 to 200
mV rms
6.67
± 12
8
± 0.5
5.33
6.67
± 2.8
± 2.0
± 5.0
V pk
V pk
V pk
± 12
8
± 0.2
V pk
kΩ
mV
14
90
130
14
90
130
kHz
kHz
kHz
100
900
1.5
100
900
1.5
kHz
kHz
MHz
0 to +1.0
0 to +1.0
10
± 0.3
–3.0
+0.33
–0.033
4
ⴞ0.5
12
ⴞ0.5
8
50
2
1
100
± 10
+20
10
12
–VS to (+VS
–2 V)
± 0.8
100
108
± 10
± 0.1
0.3
0.3
8
ⴞ2
300
(+5 mA,
–130 µA)
Short Circuit Current
Small Signal Bandwidth
Slew Rate 5
Units
25
± 10
± 0.1
dB OUTPUT
Error, VIN = 7 mV to 300 mV rms
Scale Factor
Scale Factor Temperature Coefficient
I REF for 0 dB = 0.1 V rms
I REF Range
ⴞ0.5 ⴞ1.0
± 0.1 ± 0.01
± 2.8
± 2.0
± 5.0
5.33
Max
V OUT = avg. ( V IN )2
Specified Accuracy
–0.2
–0.5
AVERAGING TIME CONSTANT
AD636K
Typ
Min
V OUT = avg. ( V IN )2
TRANSFER FUNCTION
INPUT CHARACTERISTICS
Signal Range, All Supplies
Continuous rms Level
Peak Transient Inputs
+3 V, –5 V Supply
± 2.5 V Supply
± 5 V Supply
Maximum Continuous Nondestructive
Input Level (All Supply Voltages)
Input Resistance
Input Offset Voltage
Max
S
–20
8
–VS to (+VS
–2 V)
0 to +1.0
0 to +1.0
10
± 0.1
–3.0
+0.33
–0.033
4
100
± 10
10
–VS to (+VS
–2 V)
± 0.5
100
108
ⴞ0.2
12
ⴞ0.2
mV
µV/°C
mV/ V
V
V
kΩ
8
50
dB
mV/dB
% of Reading/°C
dB/°C
µA
µA
+20
12
µA/V rms
%
kΩ
V
ⴞ1
300
V
mV
nA
Ω
(+5 mA,
–130 µA)
20
l
5
+3, –5
+2, –2.5
+5
0.80
20
l
5
± 16.5
+24
1.00
–2–
+3, –5
+2, –2.5
+5
0.80
mA
MHz
V/µs
± 16.5
+24
1.00
V
V
V
mA
REV. B
AD636
Model
AD636J
Typ
Min
TEMPERATURE RANGE
Rated Performance
Storage
0
–55
TRANSISTOR COUNT
Max
Min
+70
+150
0
–55
AD636K
Typ
62
Max
Units
+70
+150
°C
°C
62
NOTES
1
Accuracy specified for 0 mV to 200 mV rms, dc or 1 kHz sine wave input. Accuracy is degraded at higher rms signal levels.
2
Measured at Pin 8 of DIP (IOUT ), with Pin 9 tied to common.
3
Error vs. crest factor is specified as additional error for a 200 mV rms rectangular pulse trim, pulse width = 200 µs.
4
Input voltages are expressed in volts rms.
5
With 10 kΩ pull down resistor from Pin 6 (BUF OUT) to –V S.
6
With BUF input tied to Common.
Specifications subject to change without notice.
All min and max specifications are guaranteed. Specifications shown in boldface are tested on all production units at final electrical test and are used to calculate outgoing
quality levels.
ABSOLUTE MAXIMUM RATINGS 1
ORDERING GUIDE
Supply Voltage
Dual Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 16.5 V
Single Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +24 V
Internal Power Dissipation2 . . . . . . . . . . . . . . . . . . . . 500 mW
Maximum Input Voltage . . . . . . . . . . . . . . . . . . . . ± 12 V Peak
Storage Temperature Range N, R . . . . . . . . . –55°C to +150°C
Operating Temperature Range
AD636J/K . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . . +300°C
ESD Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1000 V
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
10-Lead Header: θJA = 150°C/Watt.
14-Lead Side Brazed Ceramic DIP: θJA = 95°C/Watt.
Package
Descriptions
Package
Options
AD636JD
AD636KD
AD636JH
AD636KH
AD636J Chip
AD636JD/+
0°C to +70°C
0°C to +70°C
0°C to +70°C
0°C to +70°C
0°C to +70°C
0°C to +70°C
Side Brazed Ceramic DIP
Side Brazed Ceramic DIP
Header
Header
Chip
Side Brazed Ceramic DIP
D-14
D-14
H-10A
H-10A
D-14
The AD636 is simple to connect for the majority of high accuracy rms measurements, requiring only an external capacitor to
set the averaging time constant. The standard connection is
shown in Figure 1. In this configuration, the AD636 will measure the rms of the ac and dc level present at the input, but will
show an error for low frequency inputs as a function of the filter
capacitor, CAV, as shown in Figure 5. Thus, if a 4 µF capacitor
is used, the additional average error at 10 Hz will be 0.1%, at
3 Hz it will be 1%. The accuracy at higher frequencies will be
according to specification. If it is desired to reject the dc input, a
capacitor is added in series with the input, as shown in Figure 3; the capacitor must be nonpolar. If the AD636 is driven
with power supplies with a considerable amount of high frequency
ripple, it is advisable to bypass both supplies to ground with
0.1 µF ceramic discs as near the device as possible. CF is an
optional output ripple filter, as discussed elsewhere in this data
sheet.
Contact factory for latest dimensions.
Dimensions shown in inches and (mm).
0.1315 (3.340)
RL
9
+VS 14
Temperature
Range
STANDARD CONNECTION
METALIZATION PHOTOGRAPH
COM
10
Model
8 IOUT
CF
CAV
– +
0.0807
(2.050)
VIN
1a*
1b*
VIN
7 BUF IN
2
6 BUF OUT
–VS
3
–VS
1
4
5
CAV dB
3
(OPTIONAL)
ABSOLUTE
VALUE
5
VOUT
NOTE
*BOTH PADS SHOWN MUST BE CONNECTED TO VIN.
AD636
SQUARER
DIVIDER
12
+VS
ABSOLUTE
VALUE
VIN
10kV
CF
(OPTIONAL)
CAV
+ –
–VS
Figure 1. Standard RMS Connection
REV. B
–3–
VOUT
10kV
SQUARER
DIVIDER
10
8
–
BUF
CURRENT
MIRROR
9
+
BUF
–
10kV
+
AD636
11
CURRENT
MIRROR
6
7
10kV
+VS
13
4
PAD NUMBERS CORRESPOND TO PIN NUMBERS
FOR THE TO-116 14-PIN CERAMIC DIP PACKAGE.
14
AD636
APPLYING THE AD636
The input and output signal ranges are a function of the supply
voltages as detailed in the specifications. The AD636 can also
be used in an unbuffered voltage output mode by disconnecting
the input to the buffer. The output then appears unbuffered
across the 10 kΩ resistor. The buffer amplifier can then be used
for other purposes. Further, the AD636 can be used in a current
output mode by disconnecting the 10 kΩ resistor from the
ground. The output current is available at Pin 8 (Pin 10 on the
“H” package) with a nominal scale of 100 µA per volt rms input,
positive out.
flows into Pin 10 (Pin 2 on the “H” package). Alternately, the
COM pin of some CMOS ADCs provides a suitable artificial
ground for the AD636. AC input coupling requires only capacitor C2 as shown; a dc return is not necessary as it is provided
internally. C2 is selected for the proper low frequency break
point with the input resistance of 6.7 kΩ; for a cut-off at 10 Hz,
C2 should be 3.3 µF. The signal ranges in this connection are
slightly more restricted than in the dual supply connection. The
load resistor, RL, is necessary to provide current sinking capability.
CAV
– +
C2
3.3mF
OPTIONAL TRIMS FOR HIGH ACCURACY
If it is desired to improve the accuracy of the AD636, the external trims shown in Figure 2 can be added. R4 is used to trim the
offset. The scale factor is trimmed by using R1 as shown. The
insertion of R2 allows R1 to either increase or decrease the scale
factor by ± 1.5%.
The trimming procedure is as follows:
1. Ground the input signal, VIN, and adjust R4 to give zero
volts output from Pin 6. Alternatively, R4 can be adjusted to
give the correct output with the lowest expected value of VIN.
2. Connect the desired full-scale input level to VIN, either dc or
a calibrated ac signal (1 kHz is the optimum frequency);
then trim R1 to give the correct output from Pin 6, i.e.,
200 mV dc input should give 200 mV dc output. Of course,
a ± 200 mV peak-to-peak sine wave should give a 141.4 mV
dc output. The remaining errors, as given in the specifications, are due to the nonlinearity.
CAV
– +
SCALE
FACTOR
ADJUST
VIN
1
R1
200V
61.5%
2
–VS
3
ABSOLUTE
VALUE
13
AD636
SQUARER
DIVIDER
CURRENT
MIRROR
5
6
7
2
+
RL
10kV to 1kV
7
0.1mF
12
20kV
11
CURRENT
MIRROR
10
0.1mF
9
+
10kV
BUF
–
10kV
8
39kV
Figure 3. Single Supply Connection
CHOOSING THE AVERAGING TIME CONSTANT
The AD636 will compute the rms of both ac and dc signals. If
the input is a slowly-varying dc voltage, the output of the AD636
will track the input exactly. At higher frequencies, the average
output of the AD636 will approach the rms value of the input
signal. The actual output of the AD636 will differ from the ideal
output by a dc (or average) error and some amount of ripple, as
demonstrated in Figure 4.
IDEAL
EO
R2
154V
10kV
8
SQUARER
DIVIDER
5
6
+VS
13
AD636
4
VOUT
14
EO
9
BUF
–
10kV
ABSOLUTE
VALUE
3
12
10
1
NONPOLARIZED
11
4
VOUT
+VS
14
VIN
R3
470kV
DC ERROR = EO – EO (IDEAL)
+VS
DOUBLE-FREQUENCY
RIPPLE
R4
500kV
TIME
–VS
OFFSET
ADJUST
Figure 2. Optional External Gain and Output Offset Trims
SINGLE SUPPLY CONNECTION
The applications in Figures 1 and 2 assume the use of dual
power supplies. The AD636 can also be used with only a single
positive supply down to +5 volts, as shown in Figure 3. Figure 3
is optimized for use with a 9 volt battery. The major limitation
of this connection is that only ac signals can be measured since
the input stage must be biased off ground for proper operation.
This biasing is done at Pin 10; thus it is critical that no extraneous signals be coupled into this point. Biasing can be accomplished by using a resistive divider between +VS and ground.
The values of the resistors can be increased in the interest of
lowered power consumption, since only 1 microamp of current
AVERAGE EO = EO
Figure 4. Typical Output Waveform for Sinusoidal Input
The dc error is dependent on the input signal frequency and the
value of CAV. Figure 5 can be used to determine the minimum
value of CAV which will yield a given % dc error above a given
frequency using the standard rms connection.
The ac component of the output signal is the ripple. There are
two ways to reduce the ripple. The first method involves using
a large value of CAV. Since the ripple is inversely proportional
to CAV, a tenfold increase in this capacitance will effect a tenfold
reduction in ripple. When measuring waveforms with high crest
factors, (such as low duty cycle pulse trains), the averaging time
constant should be at least ten times the signal period. For
example, a 100 Hz pulse rate requires a 100 ms time constant,
which corresponds to a 4 µF capacitor (time constant = 25 ms
per µF).
–4–
REV. B
AD636
100
100
%
01
0.
10
R
O
R
ER
R
O
R
ER
1.0
R
O
R
ER
VALUES FOR CAV AND
1% SETTLING TIME FOR
0.1 STATED % OF READING
AVERAGING ERROR*
ACCURACY 620% DUE TO
COMPONENT TOLERANCE
0.1
10
1
100
1k
INPUT FREQUENCY – Hz
10k
–VS
ABSOLUTE
VALUE
–
13
AD636
SQUARER
DIVIDER
3
+
12
11
4
CAV
CURRENT
MIRROR
5
10
9
6
+
7
BUF
–
10kV
+
–
10.0
+VS
14
10kV
8
Rx
10kV
C3
0.01
100k
Figure 5. Error/Settling Time Graph for Use with the
Standard rms Connection
The primary disadvantage in using a large CAV to remove ripple
is that the settling time for a step change in input level is increased proportionately. Figure 5 shows the relationship between CAV and 1% settling time is 115 milliseconds for each
microfarad of CAV. The settling time is twice as great for decreasing signals as for increasing signals (the values in Figure 5
are for decreasing signals). Settling time also increases for low
signal levels, as shown in Figure 6.
SETTLING TIME RELATIVE TO
SETTLING TIME @ 200mV rms
2
C2
*% dc ERROR + % RIPPLE (PEAK)
0.01
1
(FOR SINGLE POLE, SHORT Rx,
REMOVE C3)
–
+
Vrms OUT
Figure 7. 2 Pole ‘’Post’’ Filter
DC ERROR OR RIPPLE – % of Reading
%
10
REQUIRED CAV – mF
1%
0.
1%
1.0
R
O
R
ER
10
FOR 1% SETTLING TIME IN SECONDS
MULTIPLY READING BY 0.115
VIN
10
p-p RIPPLE
(ONE POLE)
CAV = 1mF
C2 = 4.7mF
p-p RIPPLE
CAV = 1mF (FIG 1)
DC ERROR
CAV = 1mF
(ALL FILTERS)
1
p-p RIPPLE
(TWO POLE)
CAV = 1mF, C2 = C3 = 4.7mF
0.1
10
7.5
100
1k
FREQUENCY – Hz
10k
Figure 8. Performance Features of Various Filter Types
5.0
RMS MEASUREMENTS
AD636 PRINCIPLE OF OPERATION
2.5
1.0
0
1mV
10mV
100mV
rms INPUT LEVEL
Figure 6. Settling Time vs. Input Level
1V
The AD636 embodies an implicit solution of the rms equation
that overcomes the dynamic range as well as other limitations
inherent in a straightforward computation of rms. The actual
computation performed by the AD636 follows the equation:
V 2 
V rms = Avg.  IN 
 V rms 
Figure 9 is a simplified schematic of the AD636; it is subdivided
into four major sections: absolute value circuit (active rectifier),
squarer/divider, current mirror, and buffer amplifier. The input
voltage, VIN, which can be ac or dc, is converted to a unipolar
current I1, by the active rectifier A1, A2. I1 drives one input of
the squarer/divider, which has the transfer function:
A better method for reducing output ripple is the use of a
“post-filter.” Figure 7 shows a suggested circuit. If a single pole
filter is used (C3 removed, RX shorted), and C2 is approximately 5 times the value of CAV, the ripple is reduced as shown
in Figure 8, and settling time is increased. For example, with
CAV = 1 µF and C2 = 4.7 µF, the ripple for a 60 Hz input is reduced from 10% of reading to approximately 0.3% of reading.
2
The settling time, however, is increased by approximately a
I
I4 = 1
factor of 3. The values of CAV and C2 can therefore be reduced
I3
to permit faster settling times while still providing substantial
The output current, I4, of the squarer/divider drives the current
ripple reduction.
mirror through a low-pass filter formed by R1 and the externally
The two-pole post-filter uses an active filter stage to provide
connected capacitor, CAV. If the R1, CAV time constant is much
even greater ripple reduction without substantially increasing
greater than the longest period of the input signal, then I4 is
the settling times over a circuit with a one-pole filter. The values
effectively averaged. The current mirror returns a current I3,
of CAV, C2, and C3 can then be reduced to allow extremely fast
which equals Avg. [I4], back to the squarer/divider to complete
settling times for a constant amount of ripple. Caution should
the implicit rms computation. Thus:
be exercised in choosing the value of CAV , since the dc error is
dependent upon this value and is independent of the post filter.
 I12 
  = I1 rms
I
=
Avg.
For a more detailed explanation of these topics refer to the
4
 I 4 
RMS-to-DC Conversion Application Guide, 2nd Edition, available
from Analog Devices.
REV. B
–5–
AD636
The current mirror also produces the output current, IOUT,
which equals 2I4. IOUT can be used directly or converted to a
voltage with R2 and buffered by A4 to provide a low impedance
voltage output. The transfer function of the AD636 thus results:
VOUT = 2 R2 I rms = V IN rms
The dB output is derived from the emitter of Q3, since the voltage at this point is proportional to –log VIN. Emitter follower,
Q5, buffers and level shifts this voltage, so that the dB output
voltage is zero when the externally supplied emitter current
(IREF) to Q5 approximates I3.
Addition of an external resistor in parallel with RE alters this
voltage divider such that increased negative swing is possible.
Figure 11 shows the value of REXTERNAL for a particular ratio of
VPEAK to –VS for several values of RLOAD. Addition, of REXTERNAL
increases the quiescent current of the buffer amplifier by an
amount equal to REXT/–VS. Nominal buffer quiescent current
with no REXTERNAL is 30 µA at –VS = –5 V.
1.0
RATIO OF VPEAK/VSUPPLY
CURRENT MIRROR
14 +VS
10 COM
20mA
FS
R1
25kV
10mA
FS
ABSOLUTE VALUE/
VOLTAGE –CURRENT
CONVERTER
I1
A3
Q1
R4
20kV
|VIN|
+
VIN 1
Q3
R4
8kV
A1
I3
Q2 Q4
4
8
9 RL
R2
C
I
I4 AV OUT 10kV
IREF
dB
5 OUT
BUF
IN BUFFER
7
A4
6 BUF
OUT
Q5
8kV
RL = 16.7kV
RL = 6.7kV
0
0
The buffer amplifier included in the AD636 offers the user
additional application flexibility. It is important to understand
some of the characteristics of this amplifier to obtain optimum
performance. Figure 10 shows a simplified schematic of the buffer.
Since the output of an rms-to-dc converter is always positive, it
is not necessary to use a traditional complementary Class AB
output stage. In the AD636 buffer, a Class A emitter follower is
used instead. In addition to excellent positive output voltage
swing, this configuration allows the output to swing fully down
to ground in single-supply applications without the problems
associated with most IC operational amplifiers.
1
VOUT – Volts
200m
100m
CURRENT
MIRROR
RE
40kV
–VS
1M
The AD636 utilizes a logarithmic circuit in performing the
implicit rms computation. As with any log circuit, bandwidth is
proportional to signal level. The solid lines in the graph below
represent the frequency response of the AD636 at input levels
from 1 millivolt to 1 volt rms. The dashed lines indicate the
upper frequency limits for 1%, 10%, and ± 3 dB of reading
additional error. For example, note that a 1 volt rms signal will
produce less than 1% of reading additional error up to 220 kHz.
A 10 millivolt signal can be measured with 1% of reading additional error (100 µV) up to 14 kHz.
+VS
10kV
100k
FREQUENCY RESPONSE
THE AD636 BUFFER AMPLIFIER
BUFFER
INPUT
10k
REXTERNAL – V
3 –VS
Figure 9. Simplified Schematic
5mA
1k
Figure 11. Ratio of Peak Negative Swing to –VS vs.
R EXTERNAL for Several/Load Resistances
ONE-QUADRANT
SQUARER/
DIVIDER
5mA
0.5
10kV
A2
R3
10kV
RL = 50kV
BUFFER
OUTPUT
1%
10%
63dB
200mV rms INPUT
100mV rms INPUT
30mV rms INPUT
30m
10m
10mV rms
INPUT
1m
RLOAD
1mV rms INPUT
100m
REXTERNAL
(OPTIONAL, SEE TEXT)
1k
Figure 10. AD636 Buffer Amplifier Simplified Schematic
When this amplifier is used in dual-supply applications as an
input buffer amplifier driving a load resistance referred to
ground, steps must be taken to insure an adequate negative
voltage swing. For negative outputs, current will flow from the
load resistor through the 40 kΩ emitter resistor, setting up a
voltage divider between –VS and ground. This reduced effective
–VS, will limit the available negative output swing of the buffer.
1 VOLT rms INPUT
10k
100k
FREQUENCY – Hz
1M
10M
Figure 12. AD636 Frequency Response
AC MEASUREMENT ACCURACY AND CREST FACTOR
Crest factor is often overlooked in determining the accuracy of
an ac measurement. Crest factor is defined as the ratio of the
peak signal amplitude to the rms value of the signal (C.F. = VP/
V rms) Most common waveforms, such as sine and triangle
waves, have relatively low crest factors (<2). Waveforms that
–6–
REV. B
AD636
resemble low duty cycle pulse trains, such as those occurring in
switching power supplies and SCR circuits, have high crest
factors. For example, a rectangular pulse train with a 1% duty
cycle has a crest factor of 10 (C.F. = 1 η ).
Figure 13 is a curve of reading error for the AD636 for a 200 mV
rms input signal with crest factors from 1 to 7. A rectangular
pulse train (pulse width 200 µs) was used for this test since it is
the worst-case waveform for rms measurement (all the energy is
contained in the peaks). The duty cycle and peak amplitude
were varied to produce crest factors from 1 to 7 while maintaining a constant 200 mV rms input amplitude.
0.5
h = DUTY CYCLE =
INCREASE IN ERROR – % of Reading
T
0
0
CF = 1/ h
VP
Circuit Description
The input voltage, VIN, is ac coupled by C4 while resistor R8,
together with diodes D1, and D2, provide high input voltage
protection.
The buffer’s output, Pin 6, is ac coupled to the rms converter’s
input (Pin 1) by capacitor C2. Resistor, R9, is connected between
the buffer’s output, a Class A output stage, and the negative output
swing. Resistor R1, is the amplifier’s “bootstrapping” resistor.
With this circuit, single supply operation is made possible by
setting “ground” at a point between the positive and negative
sides of the battery. This is accomplished by sending 250 µA
from the positive battery terminal through resistor R2, then
through the 1.2 volt AD589 bandgap reference, and finally back
to the negative side of the battery via resistor R10. This sets
ground at 1.2 volts +3.18 volts (250 µA × 12.7 kΩ) = 4.4 volts
below the positive battery terminal and 5.0 volts (250 µA × 20 kΩ)
above the negative battery terminal. Bypass capacitors C3 and
C5 keep both sides of the battery at a low ac impedance to
ground. The AD589 bandgap reference establishes the 1.2 volt
regulated reference voltage which together with resistor R3 and
trimming potentiometer R4 set the zero dB reference current IREF.
200ms
T
EIN (rms) = 200mV
200ms
EO
–0.5
Performance Data
–1.0
1
2
3
4
5
CREST FACTOR
6
0 dB Reference Range = 0 dBm (770 mV) to –20 dBm
(77 mV) rms
0 dBm = 1 milliwatt in 600 Ω
Input Range (at IREF = 770 mV) = 50 dBm
Input Impedance = approximately 1010 Ω
VSUPPLY Operating Range +5 V dc to +20 V dc
IQUIESCENT = 1. 8 mA typical
7
Figure 13. Error vs. Crest Factor
A COMPLETE AC DIGITAL VOLTMETER
Figure 14 shows a design for a complete low power ac digital
voltmeter circuit based on the AD636. The 10 MΩ input
attenuator allows full-scale ranges of 200 mV, 2 V, 20 V and
200 V rms. Signals are capacitively coupled to the AD636 buffer
amplifier, which is connected in an ac bootstrapped configuration to minimize loading. The buffer then drives the 6.7 kΩ
input impedance of the AD636. The COM terminal of the ADC
chip provides the false ground required by the AD636 for single
supply operation. An AD589 1.2 volt reference diode is used to
provide a stable 100 millivolt reference for the ADC in the linear rms mode; in the dB mode, a 1N4148 diode is inserted in
series to provide correction for the temperature coefficient of the
dB scale factor. Calibration of the meter is done by first adjusting offset pot R17 for a proper zero reading, then adjusting the
R13 for an accurate readout at full scale.
Calibration of the dB range is accomplished by adjusting R9 for
the desired 0 dB reference point, then adjusting R14 for the
desired dB scale factor (a scale of 10 counts per dB is convenient).
Total power supply current for this circuit is typically 2.8 mA
using a 7106-type ADC.
A LOW POWER, HIGH INPUT IMPEDANCE dB METER
Introduction
The portable dB meter circuit featured here combines the functions of the AD636 rms converter, the AD589 voltage reference,
and a µA776 low power operational amplifier. This meter offers
excellent bandwidth and superior high and low level accuracy
while consuming minimal power from a standard 9 volt transistor radio battery.
In this circuit, the built-in buffer amplifier of the AD636 is used
as a “bootstrapped” input stage increasing the normal 6.7 kΩ
input Z to an input impedance of approximately 1010 Ω.
REV. B
–7–
Accuracy with 1 kHz sine wave and 9 volt dc supply:
0 dB to –40 dBm ± 0.1 dBm
0 dBm to –50 dBm ± 0.15 dBm
+10 dBm to –50 dBm ± 0.5 dBm
Frequency Response ⴞ3 dBm
Input
0 dBm = 5 Hz to 380 kHz
–10 dBm = 5 Hz to 370 kHz
–20 dBm = 5 Hz to 240 kHz
–30 dBm = 5 Hz to 100 kHz
–40 dBm = 5 Hz to 45 kHz
–50 dBm = 5 Hz to 17 kHz
Calibration
1. First calibrate the zero dB reference level by applying a 1 kHz
sine wave from an audio oscillator at the desired zero dB
amplitude. This may be anywhere from zero dBm (770 mV
rms – 2.2 volts p-p) to –20 dBm (77 mV rms 220 mV – p-p).
Adjust the IREF cal trimmer for a zero indication on the analog
meter.
2. The final step is to calibrate the meter scale factor or gain.
Apply an input signal –40 dB below the set zero dB reference
and adjust the scale factor calibration trimmer for a 40 µA
reading on the analog meter.
The temperature compensation resistors for this circuit may be
purchased from: Tel Labs Inc., 154 Harvey Road, P.O. Box 375,
Londonderry, NH 03053, Part #Q332A 2 kΩ 1% +3500 ppm/°C
or from Precision Resistor Company, 109 U.S. Highway 22, Hillside, NJ 07205, Part #PT146 2 kΩ 1% +3500 ppm/°C.
AD636
D1
1N4148
C4
2.2mF
+
–
R6
1MV
1
C3
0.02mF
R1
9MV
2
+VS
14
ABSOLUTE
VALUE
R8
2.49kV
13
AD636
12
R9
100kV
0dB SET
11
R10
20kV
2V
3
R2
900kV
–
6.8mF +
20V
SQUARER
DIVIDER
4
CURRENT
MIRROR
5
R3
90kV
6
200V
R4
10kV
7
COM
R11
10kV
LIN
R12
1kV
D3
1.2V
AD589
10
OFF
+ ON
REF HI
dB
R14
10kV
dB
SCALE
R13
500V
+VDD
+VDD
1mF
3-1/2 DIGIT
7106 TYPE
A/D
–VSS
CONVERTER
–
+
9V
BATTERY
REF LO
9
+
BUF
–
COM
10kV
LIN
SCALE
LIN
8
10kV
R7
20kV
D2
1N4148
C651d–0–8/99
200mV
VIN
R5
47kV
1W
10%
dB
+
R15
1MV
D4
1N4148
ANALOG
IN
C6
0.01mF
LIN
C7
6.8mF
3-1/2
DIGIT
LCD
DISPLAY
HI
LO
dB
–VS
LXD 7543
–VSS
Figure 14. A Portable, High Z Input, RMS DPM and dB Meter Circuit
D1
1N6263
C1
3.3mF
+
R1
1MV
C2
6.8mF +
2
ABSOLUTE
VALUE
14
R2
12.7kV
13
AD636
3
SIGNAL
INPUT
SQUARER
DIVIDER
C3
10mF
+
12
11
CURRENT
MIRROR
R8
47kV
1 WATT
9
6
7
R9
10kV
D2
1N6263
10
+
BUF
–
10kV
*R7
2kV
+
8
–
mA776
+
C6
0.1mF
C5
10mF
R5
10kV
100mA
R6
100V
10kV
SCALE FACTOR
ADJUST
R4
500kV
IREF
ADJUST
R3
5kV
AD589J
250mA
5
–
9 VOLT
+1.2 VOLTS
+
4
C4
0.1mF
ON/OFF
+
+4.4 VOLTS
1
R10
20kV
+
–
0–50mA
R11
820kV
5%
+4.7 VOLTS
ALL RESISTORS 1/4 WATT 1% METAL FILM UNLESS OTHERWISE STATED EXCEPT
*WHICH IS 2kV +3500ppm 1% TC RESISTOR.
Figure 15. A Low Power, High Input Impedance dB Meter
OUTLINE DIMENSIONS
D Package (TO-116)
PRINTED IN U.S.A.
Dimensions shown in inches and (mm).
H Package (TO-100)
REFERENCE PLANE
0.005 (0.13) MIN
0.185 (4.70)
0.165 (4.19)
0.098 (2.49) MAX
14
0.750 (19.05)
0.500 (12.70)
0.160 (4.06)
0.110 (2.79)
0.250 (6.35) MIN
0.050 (1.27) MAX
8
0.310 (7.87)
0.220 (5.59)
6
PIN 1
0.785 (19.94) MAX
0.200 (5.08)
MAX
0.200 (5.08)
0.125 (3.18)
0.023 (0.58)
0.014 (0.36)
0.060 (1.52)
0.015 (0.38)
0.370 (9.40)
0.335 (8.51)
7
0.320 (8.13)
0.290 (7.37)
0.150
(3.81)
MAX
0.100 0.070 (1.78) SEATING
(2.54) 0.030 (0.76) PLANE
BSC
0.115
(2.92)
BSC
8
4
0.040 (1.02) MAX
0.045 (1.14)
0.010 (0.25)
–8–
0.045 (1.14)
0.027 (0.69)
9
3
2
0.015 (0.38)
0.008 (0.20)
7
5
0.335 (8.51)
0.305 (7.75)
1
0.019 (0.48)
0.016 (0.41)
0.230 (5.84)
BSC
0.021 (0.53)
0.016 (0.41)
10
1
0.034 (0.86)
0.027 (0.69)
36° BSC
BASE & SEATING PLANE
REV. B
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