AD AD8320-EB Serial digital controlled variable gain line driver Datasheet

a
FEATURES
8-Bit Serial Gain Control
V/V/LSB Linear Gain Response
36 dB Gain Range
60.20 dB Gain Accuracy
Upper Bandwidth: 150 MHz
22 dBm 1 dB Compression Point (75 V)
Drives Low Distortion Signals into 75 V Load:
–57 dBc SFDR at 42 MHz and 12 dBm Out
–46 dBc SFDR at 42 MHz and 18 dBm Out
Single Supply Operation from 5 V to 12 V
Maintains 75 V Output Impedance
Power-Up and Power-Down Condition
Supports SPI Input Control Standard
Serial Digital Controlled
Variable Gain Line Driver
AD8320
FUNCTIONAL BLOCK DIAGRAM
VCC
PWR AMP
REFERENCE
VREF
VIN
INV.
The AD8320 is made up of a digitally controlled variable attenuator of 0 dB to –36 dB, which is preceded by a low noise,
fixed gain buffer and followed by a low distortion high power
amplifier. The AD8320 has a 220 Ω input impedance and accepts a single-ended input signal with a specified analog input
level of up to 0.310 V p-p. The output is specified for driving a
75 Ω load, such as coaxial cable, although the AD8320 is capable of driving other loads. Distortion performance of –57 dBc
is achieved with an output level up to 12 dBm (3.1 V p-p) at
42 MHz, while –46 dBc distortion is achieved with an output
level up to 18 dBm (6.2 V p-p).
A key performance and cost advantage of the AD8320 results
from the ability to maintain a constant 75 Ω output impedance
during power-up and power-down conditions. This eliminates
the need for external 75 Ω back-termination, resulting in twice
the effective output voltage when compared to a standard operational amplifier. Additionally, the on-chip 75 Ω termination
REVERSE
AMP
ATTENUATOR CORE
BUF.
DATA LATCH
POWERDOWN/
SWITCH
INTER.
DATA SHIFT REGISTER
DATEN CLK
PD
SDATA
results in low glitch output during power-down and power-up
transitions, eliminating the need for an external switch.
The AD8320 is packaged in a 20-lead SOIC and operates from
a single +5 V through +12 V supply and has an operational
temperature range of –40°C to +85°C.
220
230
PO = 18dBm
DISTORTION – dBc
The AD8320 is a digitally controlled variable gain amplifier
optimized for coaxial line driving applications. An 8-bit serial
word determines the desired output gain over a 36 dB range
(256 gain levels). The AD8320 provides linear gain response.
AD8320
VOUT
APPLICATIONS
Coaxial Cable Driver
HFC Cable Telephony Systems
HFC High Speed Data Modems
Interactive Set-Top Boxes
PC Plug-In Modems
Interfaces with AD9853 I2C Controlled Digital Modulator
High Performance Digitally Controlled Variable Gain
Block
DESCRIPTION
GND
240
250
PO = 12dBm
PO = 8dBm
260
270
PO = 4dBm
280
1
10
FREQUENCY – MHz
100
Figure 1. Worst Harmonic Distortion vs. Frequency for
Various Output Levels at VCC = 12 V
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1998
(@ VCC = 12 V, TA = +258C, VIN = 0.310 V p-p, RL = 75 V, RS = 75 V unless
AD8320–SPECIFICATIONS otherwise noted)
Parameter
INPUT CHARACTERISTICS
Full-Scale Input Voltage
Conditions
Min
Max Gain, POUT = 18 dBm, VCC = 12 V
Max Gain, POUT = 12 dBm, VCC = 5 V
Input Resistance
Input Capacitance
GAIN CONTROL INTERFACE
Gain Range
Full Scale (Max) Gain
Gain Offset (Min) Gain
Gain Scaling Factor
OUTPUT CHARACTERISTICS
Bandwidth (–3 dB)
Bandwidth Roll-Off
Bandwidth Peaking
Output Offset Voltage
Output Offset Drift
Output Noise Spectral Density
1 dB Compression Point
Output Impedance
Overload Recovery
OVERALL PERFORMANCE
Worst Harmonic Distortion
3rd Order Intercept
Full-Scale (Max Gain) Accuracy
Gain Offset (Min Gain) Accuracy
Gain Accuracy
Gain Drift
Gain Variation w/Supply
Output Settling to 1 mV
Gain Change @ TDATEN = 1
Input Change
POWER CONTROL
Power-Down Settling Time to 1 mV
Power-Up Settling Time to 1 mV
Power-Down Pedestal Offset
Spectral Output Leakage
Maximum Reverse Power
POWER SUPPLY
Specified Operating Range
Quiescent Current
Power Down
Power Up, VCC = +12 V
Power Down, VCC =+12 V
All Gain Codes
F = 65 MHz
F = 65 MHz
All Gain Codes
Full Temperature Range
Max. Gain, Frequency = 10 MHz
Min. Gain, Frequency = 10 MHz
PD = 0, Frequency = 10 MHz
VCC = 12 V
VCC = 5 V
Power Up and Power Down
Max Gain, VIN = 500 mV p-p
F = 42 MHz, POUT = 12 dBm, VCC = 12 V
F = 42 MHz, POUT = 12 dBm, VCC = 5 V
F = 42 MHz, POUT = 18 dBm, VCC = 12 V
F = 65 MHz, POUT = 12 dBm, VCC = 12 V
F = 65 MHz, POUT = 12 dBm, VCC = 5 V
F = 65 MHz, POUT = 18 dBm, VCC = 12 V
F = 42 MHz, POUT = 18 dBm, VCC = 12 V
F = 42 MHz, POUT = 12 dBm, VCC = 5 V
F = 65 MHz, POUT = 18 dBm, VCC = 12 V
F = 65 MHz, POUT = 12 dBm, VCC = 5 V
F = 10 MHz
F = 10 MHz
F = 10 MHz, All Gain Codes
Full Temperature Range
VCC = +5 V to 12 V
65
–0.75
Typ
Max
Units
0.310
0.155
220
2.0
V p-p
V p-p
Ω
pF
36
26 (20)
–10.0 (0.316)
0.077
dB
dB (V/V)
dB (V/V)
V/V/LSB
150
0.7
0
± 40
± 0.25
73
53
4.5
22.5
16
75
40
MHz
dB
dB
mV
mV/°C
nV/√Hz
nV/√Hz
nV/√Hz
dBm
dBm
Ω
ns
–57.0
–43.0
–46.0
–57.0
–42.5
–43.0
34
32
32.5
28.5
± 0.1
± 0.2
± 0.2
± 0.5
35
85
–52.0
–39.0
–42.0
–52.0
–39.0
–40.0
0.75
dBc
dBc
dBc
dBc
dBc
dBc
dBm
dBm
dBm
dBm
dB
dB
dB
mdB/°C
mdB/V
Min to Max Gain, VIN = 0.31 V p-p
Max Gain, VIN = 0 V to 0.31 V p-p
30
25
ns
ns
Max Gain, VIN = 0
Max Gain, VIN = 0
Max Gain, VIN = 0
F (PD) = 400 Hz @ 15% Duty Cycle
5 MHz ≤ F ≤ 65 MHz
PD = 0
45
65
± 30
–70
ns
ns
mV
dBm
5
dBm
+5
PD = 1,
PD = 0,
PD = 1,
PD = 0,
VCC = +5 V
VCC = +5 V
VCC = +12 V
VCC = +12 V
–2–
80
25
97
32
+12
85
30
105
37
V
mA
mA
mA
mA
REV. 0
AD8320
LOGIC INPUTS (TTL/CMOS Logic) (DATEN, CLK, SDATA, 5 V ≤ V
CC
≤ 12 V; Full Temperature Range)
Parameter
Min
Logic “1” Voltage
Logic “0” Voltage
Logic “1” Current (VINH = 5 V) CLK, SDATA, DATEN
Logic “0” Current (VINL = 0 V) CLK, SDATA, DATEN
Logic “1” Current (VINH = 5 V) PD
Logic “0” Current (VINL = 0 V) PD
2.1
0
0
–450
0
–320
TIMING REQUIREMENTS (Full Temperature Range, V
CC
Typ
Min
Clock Pulse Width (TWH)
Clock Period (TC)
Setup Time SDATA vs. Clock (TDS)
Setup Time DATEN vs. Clock (TES)
Hold Time SDATA vs. Clock (TDH)
Hold Time DATEN vs. Clock (TEH)
Input Rise and Fall Times, SDATA, DATEN, Clock (TR, TF)
12.0
32.0
6.5
17.0
5.0
3.0
Typ
TDS
VALID DATA WORD G1
MSB. . . .LSB
VALID DATA WORD G2
TC
TWH
CLK
TES
TEH
8 CLOCK CYCLES
DATEN
GAIN TRANSFER (G1)
GAIN TRANSFER (G2)
TOFF
PD
TGS
TON
ANALOG
OUTPUT
PEDESTAL
SIGNAL AMPLITUDE (p-p)
Figure 2. Serial Interface Timing
VALID DATA BIT
MSB
MSB-1
TDS
MSB-2
TDH
CLK
Figure 3.
REV. 0
Units
5.0
0.8
20
–75
190
–70
V
V
nA
nA
µA
µA
Supply Range, TR = TF = 4 ns, FCLK = 8 MHz unless otherwise noted.)
Parameter
SDATA
Max
–3–
Max
Units
10
ns
ns
ns
ns
ns
ns
ns
AD8320
ABSOLUTE MAXIMUM RATINGS*
PIN CONFIGURATION
Supply Voltage +VS
Pins 7, 8, 9, 17, 20 . . . . . . . . . . . . . . . . . . . –0.8 V to +13 V
Input Voltages
Pins 19 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 3 V
Pins 1, 2, 3, 6 . . . . . . . . . . . . . . . . . . . . . . . . –0.8 V to +5 V
Internal Power Dissipation
Small Outline (RP) . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 W
Operating Temperature Range . . . . . . . . . . . –40°C to +85°C
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
Lead Temperature, Soldering 60 seconds . . . . . . . . . . +300°C
20 VCC
SDATA 1
CLK 2
19 VIN
DATEN 3
18 VREF
17 VCC
GND 4
VOCM 5
AD8320
16 GND
TOP VIEW 15 GND
PD
(Not to Scale)
14 BYP
7
VCC
6
*Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
VCC 8
13 GND
VCC
12 GND
9
VOUT 10
11 GND
ORDERING GUIDE
Model
Temperature Range
Package Description
uJA
Package Option
AD8320ARP
AD8320-EB
–40°C to +85°C
20-Lead Thermally Enhanced Power SOIC*
Evaluation Board
53°C/W
RP-20
*Shipped in tubes (38 pieces/tube) and dry packed per J-STD-020.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD8320 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
PIN FUNCTION DESCRIPTIONS
Pin
Function
Description
1
SDATA
Serial Data Input. This digital input allows for an 8-bit serial (gain) word to be loaded into the internal
register with the MSB (most significant bit) first.
2
CLK
Clock Input. The clock port controls the serial attenuator data transfer rate to the 8-bit master-slave
register. A Logic 0 to 1 transition latches the data bit and a 1 to 0 transfers the data bit to the slave.
This requires the input serial data word to be valid at or before this clock transition.
3
DATEN
Data Enable Low Input. This port controls the 8-bit parallel data latch and shift register. A Logic 0 to 1
transition transfers the latched data to the attenuator core (updates the gain) and simultaneously inhibits serial data transfer into the register. A 1 to 0 transition inhibits the data latch (holds the previous
gain state) and simultaneously enables the register for serial data load.
4, 11, 12,
13, 15, 16
GND
Common External Ground Reference.
5
VOCM
VCC/2 Reference Pin. A dc output reference level that is equal to 1/2 of the supply voltage (VCC).
This port should be externally ac decoupled (0.1 µF cap).
6
PD
Power-Down Low Logic Input. A Logic 0 powers down (shuts off) the power amplifier disabling the
output signal and enabling the reverse amplifier. A Logic 1 enables the output power amplifier and
disables the reverse amplifier.
7, 8, 9, 17, 20 VCC
Common Positive External Supply Voltage.
10
VOUT
Output Signal Port. DC biased to approximately VCC/2.
14
BYP
Internal Bypass. This pin must be externally ac decoupled (0.1 µF cap).
18
VREF
Input Reference Voltage (typically 1.9 V at 27°C). This port should be externally ac decoupled
(0.1 µF cap).
19
VIN
Analog Voltage Input Signal Port. DC biased to VREF voltage.
–4–
REV. 0
Typical Performance Characteristics–AD8320
0.6
VCC = 12V
F = 10MHz
T = +258C
0.1
0
T = 2408C
20.1
T = +858C
20.2
0
10MHz
20.3
42MHz
20.6
0
21.2
256
64
128
192
GAIN CONTROL – Decimal
Figure 4. Gain Error vs. Gain Control
at Various Temperatures
30
255D
170D
GAIN – dB
1M
10M
100M
FREQUENCY – Hz
1G
00D
Figure 7. AC Response
1M
F = 10MHz
VCC = 12V
+858C
65
+258C
60
2408C
55
50
250
260
270
VCC = 5V, PIN = 214dBm
280
10M
100M
FREQUENCY – Hz
1G
VCC = 12V, PIN = 28dBm
1M
10M
100M
FREQUENCY – Hz
1G
Figure 9. Input Signal Feedthrough
vs. Frequency
90
F = 10MHz
75
OUTPUT NOISE – nV/!Hz
70
240
2100
100k
80
75
256
MAX GAIN
PD = 0V
290
Figure 8. AC Response
80
OUTPUT NOISE – nV/!Hz
230
01D
220
100k
64
128
192
GAIN CONTROL – Decimal
220
0
210
0
Figure 6. Gain Error vs. Gain Control
at Different Supply Voltages
VCC = 12V
10
01D
00D
20.45
80
OUTPUT NOISE – nV/!Hz
GAIN – dB
0
220
100k
256
85D
10
20.15
170D
20
85D
210
64
128
192
GAIN CONTROL – Decimal
255D
VCC = 5V
VCC = 12V
0
20.30
Figure 5. Gain Error vs. Gain Control
at Various Frequencies
30
20
0
0.15
VCC = 5V
65MHz
20.9
F = 10MHz
T = +258C
0.30
FEEDTHROUGH – dB
20.3
VCC = 12V
T = +258C
0.3
GAIN ERROR – dB
GAIN ERROR – dB
0.2
0.45
GAIN ERROR – dB
0.3
70
VCC = 12V
65
60
VCC = 5V
55
50
MAX GAIN, VCC = 12V
70
MAX GAIN, VCC = 5V
60
MIN GAIN, VCC = 12V
50
MIN GAIN, VCC = 5V
40
45
45
40
0
128
192
64
GAIN CONTROL – Decimal
256
Figure 10. Output Referred Noise vs.
Gain Control at Various Temperatures
REV. 0
40
0
128
192
64
GAIN CONTROL – Decimal
256
Figure 11. Output Referred Noise vs.
Gain Control at Different Supply
Voltages
–5–
30
100k
10M
1M
FREQUENCY – Hz
100M
Figure 12. Output Referred Noise vs.
Frequency
AD8320
230
220
220
VCC = 12V
F = 65MHz, PO = 18dBm
230
250
260
270
230
PO = 12dBm
240
PO = 10dBm
250
260
280
270
F = 65MHz, PO = 12dBm
64
128
192
GAIN CONTROL – Decimal
220
220
230
230
PO = 18dBm
PO = 12dBm
250
250
260
PO = 8dBm
260
270
PO = 4dBm
PO = 8dBm
PO = 4dBm
280
1
10
FREQUENCY – MHz
100
Figure 14. Worst Harmonic Distortion vs. Frequency for Various Output
Levels at VCC = 5 V
DISTORTION – dBc
DISTORTION – dBc
280
256
Figure 13. Worst Harmonic Distortion vs. Gain Control
240
PO = 10dBm
1
10
FREQUENCY – MHz
100
Figure 15. Worst Harmonic Distortion vs. Frequency for Various Output
Levels at VCC = 6 V
40
PO = 18dBm
30
240
250
PERCENTAGE
0
PO = 12dBm
240
PO = 8dBm
F = 42MHz, PO = 12dBm
290
DISTORTION – dBc
F = 42MHz, PO = 18dBm
DISTORTION – dBc
DISTORTION – dBc
240
PO = 12dBm
PO = 8dBm
260
VCC = 12V
PO = 18dBm
F = 42MHz
N = 30
20
10
270
270
PO = 4dBm
PO = 4dBm
280
1
10
FREQUENCY – MHz
1
100
Figure 16. Worst Harmonic Distortion vs. Frequency for Various Output
Levels at VCC = 10 V
20
20
10
0
245
241
Figure 19. Distribution of Worst Harmonic Distortion
VCC = 5V
PO = 12dBm
F = 65MHz
N = 30
15.0
10
0
259
243
Figure 18. Distribution of Worst Harmonic Distortion
22.5
5
244
243
242
HARMONIC DISTORTION 2 dBc
246
245
244
HARMONIC DISTORTION – dBc
30.0
VCC = 12V
PO = 12dBm
F = 42MHz
N = 30
15
PERCENTAGE
PERCENTAGE
VCC = 12V
PO = 18dBm
F = 65MHz
N = 30
0
247
100
Figure 17. Worst Harmonic Distortion vs. Frequency for Various Output
Levels at VCC = 12 V
40
30
10
FREQUENCY – MHz
PERCENTAGE
280
7.5
258
257
256
HARMONIC DISTORTION 2 dBc
255
Figure 20. Distribution of Worst Harmonic Distortion
–6–
0
244
243
242
241
HARMONIC DISTORTION 2 dBc
240
Figure 21. Distribution of Worst Harmonic Distortion
REV. 0
AD8320
235
45
20
0
245
250
250
F = 42MHz
VCC = 12V
PO = 18dBm
255
POUT – dBm
DISTORTION 2 dBc
240
PO = 18dBm
MAX GAIN
VCC = 12V
250
F = 65MHz
VCC = 12V
PO = 18dBm
–20
–40
–60
260
VCC = 12V
3RD ORDER INTERCEPT – dBm
F = 42MHz, VCC = 5V, PO = 12dBm
40
PO = 12dBm
35
PO = 18dBm
30
25
F = 65MHz, VCC = 12V, PO = 12dBm
265
250
225
25
50
0
TEMPERATURE 2 8C
75
Figure 22. Harmonic Distortion vs.
Temperature
20
–80
100
41.2
42.0
42.4
41.6
FREQUENCY – MHz
42.8
Figure 23. Two-Tone Intermodulation Distortion
1
10
FREQUENCY – MHz
100
Figure 24. Third Order Intercept vs.
Frequency
30
VCC = 5V
MAX GAIN
VCC = 12V
VIN = 310mV p-p
MAX GAIN
F = 10MHz
27
GAIN – dB
VCC = 12V
VIN = 310mV p-p
MIN GAIN
F = 10MHz
CL = 0pF
24
CL = 22pF
21
CL = 100pF
CL = 150pF
18
1.2V
12.5nsec
20mV
12.5nsec
15
100k
Figure 26. Transient Response
Figure 25. Transient Response
1M
10M
100M
FREQUENCY – Hz
1G
Figure 27. AC Response for Various
Capacitive Loads
30
VCC = 5V
MAX GAIN
F = 1MHz
CL = 0pF
27
CL = 100pF
CL = 150pF
24
21
CL = 150pF
CL = 150pF
1V
Figure 28. Transient Response for
Various Capacitive Loads
15
100k
CL = 22pF
CL = 100pF
CL = 100pF
5nsec
TIME – Seconds
VCC = 12V
MAX GAIN
F = 1MHz
CL = 0pF
CL = 22pF
18
500mV
REV. 0
CL = 0pF
CL = 22pF
GAIN – dB
OUTPUT VOLTAGE – Volts
VCC = 12V
MAX GAIN
1M
10M
100M
FREQUENCY – Hz
1G
Figure 29. AC Response for Various
Capacitive Loads
–7–
5nsec
Figure 30. Transient Response for
Various Capacitive Loads
AD8320
100mV
VCC = 12V
MAX GAIN
VIN = 0V p-p
5mV
VCC = 12V
MAX GAIN
VIN = 0V p-p
VOUT
VCC = 5V
F = 40MHz
MAX GAIN
1.25V
VOUT
VOUT
CLK
VIN
PD
5V
DATEN
75nsec
5V
VCC = 12V
F = 40MHz
MAX GAIN
2.00V
VOUT
VCC = 12V, F = 40MHz
MAX GAIN
VCC = 12V, F = 40MHz
MIN TO MAX GAIN
VIN = .310V p-p
2.50V
VOUT
VOUT
VIN
20nsec
Figure 33. Overload Recovery
Figure 32. Clock Feedthrough
Figure 31. Power-Up/Power-Down
Glitch
2.50V
500mV
5V
250nsec
VIN
DATEN
1V
5V
250mV
20nsec
Figure 34. Overload Recovery
Figure 35. Output Settling Time Due
to Input Change
80
50
1M
10M
100M
FREQUENCY – Hz
1G
Figure 37. Input Impedance vs.
Frequency
VCC = 5V, PD = 1
SUPPLY CURRENT – mA
60
120
VCC = 5V, PD = 0
OUTPUT IMPEDANCE – V
INPUT IMPEDANCE – V
70
80
70
VCC = 12V, PD = 1
VCC = 12V, PD = 0
60
50
100k
20nsec
Figure 36. Output Settling Time Due
to Gain Change
90
VCC = 5V
RT = 115V
40
100k
5V
20nsec
1M
10M
FREQUENCY – Hz
100M
Figure 38. Output Impedance vs.
Frequency
–8–
100
VCC = 12V, PD = 1
80
VCC = 5V, PD = 1
60
40
VCC = 12V, PD = 0
20
VCC = 5V, PD = 0
0
–50
–25
0
25
50
TEMPERATURE – 8C
75
100
Figure 39. Supply Current vs.
Temperature
REV. 0
AD8320
OPERATIONAL DESCRIPTION
32
The AD8320 is a digitally controlled variable gain power amplifier that is optimized for driving 75 Ω cable. A multifunctional
bipolar device on single silicon, it incorporates all the analog
features necessary to accommodate reverse path (upstream) high
speed (5 MHz to 65 MHz) cable data modem and cable telephony requirements. The AD8320 has an overall gain range of
36 dB (–10 dB to 26 dB) and is capable of greater than 100 MHz
of operation at output signal levels exceeding 18 dBm. Overall,
when considering the device’s wide gain range, low distortion,
wide bandwidth and variable load drive, the device can be used
in many variable gain block applications.
24
GAIN – dB
16
8
0
AV = 20 3 LOG10 (0.316 + 0.077 3 CODE)
–8
The digitally programmable gain is controlled by the three wire
“SPI” compatible inputs. These inputs are called SDATA
(serial data input port), DATEN (data enable low input port)
and CLK (clock input port). See Pin Function Descriptions and
Functional Block diagram. The AD8320 is programmed by an
8-bit “attenuator” word. These eight bits determine the 256
programmable gain settings. See attenuator core description
below. The gain is linear in V/V/LSB and can be described by
the following equation:
–16
0
32
64
96
128
160
GAIN – Code – Decimal
192
224
256
Figure 41. Log Gain vs. Gain Control
The attenuator core can be viewed as eight binarily weighted
(differential in–differential out) transconductance (gm) stages
with the “in phase” current outputs of all eight stages connected
in parallel to their respective differential load resistors (not
shown). The core differential output signals are also 180 degrees
out of phase and equal in amplitude. The input stages are likewise parallel, connected to the inverting input amplifier and
buffer outputs as shown. Nine bits plus of accuracy is achieved
for all gain settings over the specified frequency, supply voltage
and temperature range. The actual total core GM × RL attenuation is determined by which combination of binarily weighted
gm stages are selected by the data latch. With 8 bits, 256 levels
of attenuation can be programmed. This results in a 36 dB
attenuation range (0 dB to –36 dB). See gain equation above.
AV = 0.316 + 0.077 × Code (RL = 75 Ω)
where code is the decimal equivalent of the 8-bit word. For example, if all 8 bits are at a logic “1,” the decimal equivalent is
255 and AV equals 19.95 V/V or 26 dB. The gain scaling factor
is 0.077 V/V/LSB, with an offset of 0.316 V/V (–10.0 dB). Figure 40 shows the linear gain versus decimal code and Figure 41
shows the gain in dB versus decimal code. Note the nonlinearity that results when viewed in dB versus code. The dB step
size increases as the attenuation increases (i.e., gain decreases)
and reaches a maximum step size of approximately 1.9 dB (gain
change between 01 and 00 decimal).
VCC
GND
PWR AMP
22
REFERENCE
AD8320
VOUT
20
18
14
GAIN – V/V
VREF
VIN
AV = 0.316 + 0.077 3 CODE
16
POWER-UP
INV.
REVERSE
AMP
ATTENUATOR CORE
BUF.
12
DATA LATCH
10
POWER–
DOWN
8
6
DATA SHIFT REGISTER
PD
SWITCH
INTER.
4
2
–2
DATEN CLK
POWER-DOWN
0
Figure 42. Functional Block Diagram
0
32
64
96
128
160
GAIN – Code – Decimal
192
224
256
To update the AD8320 gain, the following digital load sequence
is required. The attenuation setting is determined by the 8-bit
word in the data latch. This 8-bit word is serially loaded (MSB
first) into the shift register at each rising edge of the clock. See
Figure 43. During this data load time (T), DATEN is low and
the data latch is latched holding the previous (T – 1) data word
keeping the attenuation level unchanged. After eight clock
cycles the new data word is fully loaded and DATEN is
switched high. This enables the data latch (becomes transparent) and the loaded register data is passed to the attenuator with
the updated gain value. Also at this DATEN transition, the
internal clock is disabled, thus inhibiting new serial input data.
Figure 40. Linear Gain vs. Gain Control
The AD8320 is composed of three analog functions in the powerup or forward mode (Figure 42). The input inverter/buffer
amplifier provides single-ended to differential output conversion. The output signals are nominally 180 degrees out of phase
and equal in amplitude with a differential voltage gain of 2 (6 dB).
Maintaining close to 180 degrees and equal amplitude is required for proper gain accuracy of the attenuator core over the
specified operating frequency. The input buffer/inverter also
provides equal dc voltages to the core inputs via the internal
reference. This is required to ensure proper core linearity over
the full specified power supply range (5 V to 12 V).
REV. 0
SDATA
–9–
AD8320
TDS
SDATA
VALID DATA WORD G1
MSB. . . .LSB
TC
VALID DATA WORD G2
TWH
CLK
TES
DATEN
TEH
8 CLOCK CYCLES
GAIN TRANSFER (G1)
GAIN TRANSFER (G2)
TOFF
PD
TGS
TON
ANALOG
OUTPUT
SIGNAL AMPLITUDE (p-p)
PEDESTAL
Figure 43. Serial Interface Timing
The power amplifier has two basic modes of operation; forward
or power-up mode and reverse or power-down mode. In the
power-up mode (PD = 1), the power amplifier stage is enabled
and the differential output core signal is amplified by 20 dB.
With a core attenuation range of 0 dB to –36 dB and 6 dB of
input gain, the overall AD8320 gain range is 26 dB to –10 dB.
In this mode, the single-ended output signal maintains a dc
level of VCC/2. This dc output level provides for optimum large
signal linearity and allows for dc coupling the output if necessary. The output stage is unique in that it maintains a dynamic
output impedance of 75 Ω. This allows for a direct 75 Ω cable
connection and results in 6 dB of added load power versus using
a series 75 Ω back-termination resistor as required with traditional low output impedance amplifiers. The power amplifier
will also drive lower or higher output loads, although the device’s
gain (not gain range) will change accordingly (see Applications
section).
In the power-down mode (PD = 0), the power amplifier is turned
off and a “reverse” amplifier (the inner triangle in Figure 42) is
enabled. During this 1 to 0 transition, the output power amplifier’s
input stage is also disabled, resulting in no forward output signal
(S21 is 0), although the attenuator core and input amplifier/
buffer signals are not affected (S11 ≈ 0). The function of the
reverse amplifier is to maintain 75 Ω and VCC/2 at the output
port (VOUT) during power-down. This is required to minimize
line reflections (S22 ≈ 0) and ensures proper filter operation for
any forward mode device sharing the same bus (i.e., in a multiplexed configuration). (See Applications section.) In the time
domain, as PD switches states, a transitional glitch and pedestal
offset results. (See Figures 31 and 43.) The powered down
supply current drops to 32 mA versus 97 mA (VCC = 12 V) in
power-up mode.
Generally, using the power-down low input (PD) for switching
allows for multiple devices to be multiplexed via splitters (N-1
off, 1 on) and reduces overall total power consumption as required for cable data applications. For cable telephony, the
power-down current generally needs to be much lower during
what is referred to as sleep and standby modes, and VCC supply
switching via PFETS or equivalent, as described in the applications section, would be required.
APPLICATIONS
The AD8320 is primarily intended to be used as the return path
(also called upstream path) line driver in cable modem and
cable telephony applications. Data to be transmitted is modulated in either QPSK or QAM format. This is done either in
DSP or by a dedicated QPSK/QAM modulator such as the
AD9853.
The amplifier receives its input signal either from the dedicated
QPSK/QAM modulator or from a DAC. In both cases, the
signal must be low-pass filtered before being applied to the line
driving amplifier.
SUBSCRIBER
TO MODEM
RECEIVE
CIRCUITRY
CENTRAL
OFFICE
DIPLEXER
AD9853
7TH ORDER
ELLIPTIC
LOW PASS
FILTER
75V
AD8320
Figure 44. Block Diagram of Cable Modem’s Upstream
Driver Section
The amplifier drives the line through a diplexer. The insertion
loss of a diplexer is typically –3 dB. As a result, the line driver
must deliver a power level roughly 3 dB greater than required by
the applicable cable modem standard so that diplexer losses are
canceled out.
Because the distance to the central office varies from subscriber
to subscriber, signals from different subscribers will be attenuated by differing amounts. As a result, the line driver is required
to vary its gain so that all signals arriving at the central office
have the same amplitude.
–10–
REV. 0
AD8320
Basic Connection
The timing diagram for AD8320’s serial interface is shown in
Figure 43.
Figure 45 shows the basic schematic for operating the AD8320.
Because the amplifier operates from a single supply, the input
signal must be ac-coupled using a 0.1 µF capacitor. The input
pin has a bias level of about 1.9 V. This bias level is available on
the VREF pin (Pin 18) and can be used to externally bias signals
if dc-coupling is desired. Under all conditions, a 0.1 µF decoupling
capacitor must be connected to the VREF pin. If the VREF voltage is to be used externally, it should be buffered first.
The write cycle to the device is initiated by the falling edge of
DATEN. This is followed by eight clock pulses that clock in the
control word. Because the clock signal is level triggered, data is
effectively clocked on the falling edge of CLK.
After the control word has been clocked in, the DATEN line
goes back high, allowing the gain to be updated (this takes
about 30 ns).
The VIN pin of the AD8320 (Pin 19) has an input impedance
of 220 Ω. Typically, in video applications, 75 Ω termination is
favored. As a result, an external shunt resistance (R1) to ground
of 115 Ω is required to create an overall input impedance of
75 Ω. If 50 Ω termination is required, a 64.9 Ω shunt resistor
should be used. Note, to avoid dc loading of the VIN pin, the
ac-coupling capacitor should be placed between the input pin
and the shunt resistor as shown in Figure 45.
The relationship between gain and control word is given by the
equation:
Gain (V/V) = 0.077 × Code + 0.316
where code is the decimal equivalent of the gain control word
(0 to 255).
The gain in dB is given by the equation:
Gain (dB) = 20 log10 (0.077 × Code + 0.316)
On the output side, the VOUT pin also has a dc bias level. In
this case the bias level is midway between the supply voltage and
ground. The output signal must therefore be ac-coupled before
being applied to the load. The dc bias voltage is available on the
VOCM pin (Pin 5) and can be used in dc-coupled applications.
This node must be decoupled to ground using a 0.1 µF capacitor. If the VOCM voltage is to be used externally, it should be
buffered.
The digital interface also contains an asynchronous power-down
mode. The normally high PD line can be pulled low at any time.
This turns off the output signal after 45 ns, and reduces the
quiescent current to between 25 mA and 32 mA (depending
upon the power supply voltage). In this mode, the programmed
gain is maintained.
Since the AD8320 has a dynamic output impedance of 75 Ω, no
external back termination resistor is required. If the output
signal is being evaluated on 50 Ω test equipment such as a spectrum analyzer, a 75 Ω to 50 Ω adapter (commonly called a pad)
should be used to maintain a properly matched circuit.
Clock Line Feedthrough
Varying the Gain
Power Supply and Decoupling
Clock feedthrough results in a 5 mV p-p signal appearing superimposed on the output signal (see Figure 32). If this impinges
upon the dynamic range of the application, the clock signal
should be noncontinuous, i.e., should only be turned on for
eight cycles during programming.
The gain of the AD8320 can be varied over a range of 36 dB,
from –10 dB to +26 dB, by varying the 8-bit gain setting word.
The AD8320 should be powered with a good quality (i.e., low
noise) single supply of between +5 V and +12 V. In order to
achieve an output power level of +18 dBm (6.2 V p-p) into
VCC
+5V TO +12V
C7
10mF
C5
0.1mF
C6
0.1mF
VCC
C4
0.1mF
VCC
REFERENCE
C12
0.1mF
C1
0.1mF
INPUT
C2
0.1mF
VCC
C3
0.1mF
C11
0.1mF
VCC
VCC
BYP
VOUT
AD8320
VREF
VOCM
DATA LATCH
*FOR A 75V INPUT
IMPEDANCE
DATA SHIFT REGISTER
DATEN
POWERDOWN
/
SWITCH
INTER.
PD
SDATA GND GND GND GND GND
CLK
CLK
SDATA
DATEN
PD
Figure 45. Basic Connection
–11–
C10
0.1mF
TO DIPLEXER
RIN = 75V
C8
0.1mF
ATTENUATOR CORE
VIN
R1*
115V
REV. 0
GND
AD8320
75 Ω, a supply voltage of at least +10 V is required. To achieve
a signal level of +12 dBm (about 3.1 V p-p) into 75 Ω, a minimum supply level of +5 V is required. However, for the lowest
possible distortion, the power supply voltage should be raised as
high as possible. In varying the power supply from +5 V to
+12 V, the quiescent current increases from 80 mA to 97 mA.
A HEXFET power MOSFET (International Rectifier part number IRLML5103) is used to turn on and off the current to the
supply pins of the AD8320. Under normal operating conditions,
the gate (labeled POWER-DOWN) should be grounded. Pulling the gate to within 2 V of the supply will open the switch and
reduce the current to the amplifier to zero.
Careful attention must be paid to decoupling the power supply
pins. A 10 µF capacitor, located fairly close to the device, is
required to provide good decoupling for lower frequency signals.
In addition, five 0.1 µF decoupling capacitors should be located
close to each of the five power supply pins (7, 8, 9, 17 and 20).
A 0.1 µF capacitor must also be connected to the pin labeled
BYP (Pin 14), to provide decoupling to an internal node of the
device. All six ground pins should be connected to a low impedance ground plane.
In cable modem and cable telephony applications the modem
must always present an output impedance of 75 Ω to the line.
This forces the line driver to always present a 75 Ω impedance
to the diplexer. In this application, a single pole double throw
RF switch (AS103, Alpha Semiconductor) is used to switch in
an external 75 Ω impedance when the AD8320 is turned off.
This resistor then mimics the dynamic output impedance of the
AD8320. TTL or CMOS logic can be used to drive the two
voltages driving the RF switch (V1 and V2).
Alternative Power-Down Mode
Before the AD8320 is turned back on again, the gain needs to
be set to a known level. This can be done by holding the PD pin
of the AD8320 low after POWER-DOWN has gone high. While
PD is held low, the 8-bit serial data stream can be clocked into
the AD8320. During this time the quiescent current will increase to 32 mA. However, this time period can be as small as
about 1 µs. In this mode the output settles about 45 ns after the
rising edge of PD.
As previously mentioned, the AD8320 can be put into a low
power sleep mode by pulling the PD pin low. If lower power
consumption is required during power-down mode, an alternative
scheme can be used as shown in Figure 46.
VCC+12V
POWERDOWN
S
G
IRLML5103
0.1mF
D
PD
VOUT
J1
VDD
PD
DATEN
AD8320*
CLK
SDATA
75V VOUT
GND
+5V
AS103
VCC
0.1mF
0.1mF (SEETEXT)
J3
J2
Alternatively, if DATEN is held low as the AD8320 is powered
on, the device will power up in minimum gain. In this mode, the
output settles after about 200 µs. Note that for both cases, the
capacitor on VOCM has been reduced from 0.1 µF to 0.01 µF
to facilitate a faster turn-on time. All other capacitors in the
circuit should be connected as shown in Figure 45.
0.1mF
VOCM
75V
14
0.1mF
0.01mF
V2
V1
V2
V1
* ADDITIONAL PINS AND DECOUPLING CAPACITORS
OMITTED FOR CLARITY
10–12V
POWERDOWN
0V
3–5V
V1
0V
3–5V
V2
0V
PD
DATEN
CLK
0V
SDATA
OUTPUT
97mA
QUIESCENT
CURRENT
97mA
0mA
32mA
Figure 46. Alternative Power-Down Mode with Timing
–12–
REV. 0
AD8320
TO MODEM
RECEIVE
CHANNEL
58dBmV
DIPLEXER
+5V
VIN
VIN
11dBmV
AD603*
VPOS
VOUT
220V
AD8320*
0.1mF
75V
100V
FBDK
41dBmV
VNEG
COMM
GNEG
GPOS
15V
REFIN
PD
VDD
75V
+12V
20pF
61dBmV
220V
–5V
CLR
1.5V
AD7801*
AGND DGND CS WR LDAC D0–D7
+0.5V
*ADDITIONAL PINS AND
DECOUPLING CAPACITORS
OMITTED FOR CLARITY
1kV
8
D0–D7
CLK
74HCT164*
A B
DATA
CLK
ENABLE
PD
Figure 47. Enhanced Dynamic Range Circuit
Enhanced Dynamic Range Application
The AD8320 can be combined with the AD603 to give additional dynamic range as shown in Figure 47. The AD603 is a
voltage controlled variable gain amplifier. The gain of the AD603 is
determined by the difference in voltage between the GPOS and
GNEG pins. This differential voltage has a range of ± 0.5 V. In
this example, the voltage on GNEG is tied to +0.5 V. As the
voltage on GPOS is varied from 0 V to 1 V, the gain of the AD603
changes from –10 dB to +30 dB with a slope of 25 mV/dB (i.e.,
linear in dB). The gain control voltage is supplied by the AD7801
DAC. The output voltage of the DAC (0 V to +2.5 V) is divided
down to fit the 0 V to 1 V range of the AD603 using a resistor
attenuator network.
In order that the same gain control word can be used for both
the AD603 and the AD8320, the serial data stream is converted
to the parallel format of the AD7801 DAC using a serial-toparallel shift register. The rising edge of the enable pulse simultaneously updates both amplifiers.
Figure 48. Output Spectrum of Enhanced Dynamic Range
Circuit (Output Level = 61 dBmV, Frequency = 42 MHz)
60
50
As the control word is varied from 00Dec to 255Dec, the gain of
the signal chain varies from –26 dB to +50 dB (there is 6 dB of
attenuation between AD603 and AD8320). In practice, this
circuit is not usable at the lower end of the gain range due to the
small input signal (11 dBmV or about 10 mV p-p). Figure 48
shows the spectrum of the output signal at a frequency of
42 MHz and an output level of 61 dBmV (3.1 V p-p, max gain).
AD8320/AD603
40
GAIN
30
AD8320
20
10
AD603
0
The gain vs. code transfer function of the two amplifiers along
with the overall gain is shown in Figure 49. The overall gain
transfer function combines a linear in dB transfer function with
a linear in Volts/Volt transfer function. It is clear from Figure 49
that the overall gain transfer function can be considered to be
approximately linear in dB over the top 50 dB of its range.
–10
–20
–30
0
21
41
81 101 121 141 161 181 201 221 241
GAIN CONTROL WORD – Decimal
61
Figure 49. Gain Transfer Function of Enhanced Dynamic
Range Circuit
REV. 0
–13–
AD8320
Varying Gain by Varying Load Impedance
and tested to demonstrate the specified high speed performance
of the device. Figure 51 shows the schematic of the evaluation
board. The silkscreen for the component side layer is shown in
Figure 52. The layout of the board is shown in Figure 53 and
Figure 54.
As already mentioned, the AD8320 has a dynamic output impedance of 75 Ω. The specified gain range assumes that the
output is terminated with a 75 Ω load impedance. Varying the
load impedance allows the gain to be varied, up to a maximum
of twice the specified gain (for RL = `). The variation in gain
with load resistance is shown in Figure 50 for the case of a gain
control word of 255Dec (i.e., max gain).
The evaluation board package includes a fully populated board
with BNC-type connectors along with Windows®-based software for controlling the board from a PC’s printer port via a
standard printer cable.
32
A prototyping area is provided to allow for additional circuitry
on the board. The single supply and ground to the board are
brought over to this area and are available on two strips. There
are also two extra strips available on the prototyping area which
can be used for additional power supplies.
30
GAIN – dB
28
The board should be powered with a good quality (i.e., low
noise) single supply of between +5 V and +12 V. Extensive
decoupling is provided on the board. A 10 µF capacitor, located
fairly close to the device, provides good decoupling for lower
frequency signals. In addition, and more importantly, five
0.1 µF decoupling capacitors are located close to each of the
five power supply pins (7, 8, 9, 17 and 20).
26
24
22
20
0
100
1000
10000
Controlling the Evaluation Board from a PC
RLOAD – V
The evaluation board ships with Windows-based control software. A standard printer cable can be used to connect the
evaluation board to a PC’s printer port (also called parallel
port). The cable length should be kept to less than about 5 feet.
The wiring of a standard printer cable, with respect to the signal lines that are used in this application, is shown in Figure 55.
Although the software controls the evaluation board via the
PC’s parallel port, the AD8320 digital interface is serial. Three
of the parallel port’s eight bits (and one digital ground line) are
used to implement this serial interface. A fourth bit is used to
control the PD pin.
Figure 50. Gain vs. RLOAD (Gain Control Word = 255Dec)
The gain can be described by the following equation:
 2 RLOAD 

AV = 20 log 10 
 0.316 + 0.077 × Code 
R
+
75


LOAD


(
)
where Code is the decimal equivalent of the 8-bit word.
Evaluation Board
A two layer evaluation board for the AD8320 is available (part
number AD8320-EB). This board has been carefully laid out
VCC
TP2
C7
10mF
C6
0.1mF
C5
0.1mF
C4
0.1mF
C2
0.1mF
C3
0.1mF
C11
0.1mF
TP4
VCC
VCC
VREF
C12
0.1mF
TP1 C1
0.1mF
R1
115V
VCC
REFERENCE
VCC
VCC
GND
BYP
VOUT
AD8320
C10
0.1mF
OUTPUT
TP3
VREF
VOCM
ATTENUATOR CORE
VOCM
VIN
INPUT
POWERDOWN
/
SWITCH
INTER
DATA LATCH
C8
0.1mF
PD
DATA SHIFT REGISTER
DATEN GND GND GND GND GND
SDATA
CLK
R4
0
C9
OPTIONAL
6
5
2
16, 19-30, 33
3
36-PIN CENTRONICS CONNECTOR
Figure 51. Evaluation Board Schematic
All trademarks are the property of their respective holders.
–14–
REV. 0
AD8320
The control software requires Windows 3.1 or later to operate.
To install the software, insert the disk labeled “Disk # 1 of 2” in
the PC and run the file called SETUP.EXE. Additional installation instructions will be given on-screen. Before beginning installation, it is important to close any other Windows applications
that are running.
When you launch the installed control software from Windows,
you will be asked to select the printer port you are using. Most
modern PCs have only one printer port, usually called LPT1.
However, some laptop computers use the PRN port.
Figure 56 shows the main screen of the control software. Using
the slider, you can set any gain in the AD8320’s 36 dB range.
The gain is displayed on-screen in dB and V/V. The 8-bit gain
setting byte is also displayed, in binary, hexadecimal and decimal.
Each time the slider is moved, the software automatically sends
and latches the required 8-bit data stream to the AD8320. You
can power down or reset the device simply by clicking the
appropriate buttons. The software also offers one volatile storage location that can be used to store a particular gain. This
functions in the same way as the memory on a pocket calculator.
Overshoot on PC Printer Ports’ Data Lines
The data lines on some printer ports have excessive overshoot.
Overshoot on the pin used as the serial clock (Pin 6 on the DSub-25 connector) can cause communication problems. This
overshoot can be eliminated by applying mild filtering to the
CLK line on the evaluation board. This can be done by putting
a small series resistor on the CLK line, combined with a
capacitor to ground. Pads are provided (C9, R4) on the component side of the evaluation board to allow easy insertion of
these devices. Determining the size of these values will take
some experimentation. Depending upon the overshoot from the
printer port, this capacitor may need to be as large as 0.01 µF,
while the resistor is typically in the 50 Ω to 100 Ω range.
Figure 52. Evaluation Board Silkscreen (Component Side)
REV. 0
–15–
AD8320
Figure 53. Evaluation Board Layout (Component Side)
–16–
REV. 0
AD8320
Figure 54. Evaluation Board Layout (Solder Side)
36 PIN CENTRONICS
1
19
D-SUB 25 PIN (MALE)
14
1
DATEN
PD
SDATA
CLK
GND
25
13
SIGNAL
D-SUB-25
DATEN
2
36-PIN CENTRONICS
PD
3
1
DATA
CLK
DGND
5
6
25
5
2
16, 19–30, 33
3
36
18
EVALUATION BOARD
PC
Figure 55. Interconnection Between AD8320EB and PC Printer Port
REV. 0
–17–
AD8320
Figure 56. Screen Display of Windows-Based Control Software
–18–
REV. 0
AD8320
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
20-Lead Thermally Enhanced Power Small Outline Package
(RP-20)
0.5118 (13.00)
0.4961 (12.60)
20
11
0.1890 (4.80) 0.4193 (10.65)
0.1791 (4.55) 0.3937 (10.00)
HEAT
SINK
0.2992 (7.60)
0.2914 (7.40)
1
10
PIN 1
0.3340 (8.61)
0.3287 (8.35)
0.1043 (2.65)
0.0926 (2.35)
8°
0°
0.0118 (0.30) 0.0500
(1.27)
0.0040 (0.10)
BSC
STANDOFF
REV. 0
0.0201 (0.51)
SEATING 0.0500 (1.27)
0.0130 (0.33) PLANE
0.0057 (0.40)
–19–
0.0295 (0.75)
x 45°
0.0098 (0.25)
–20–
PRINTED IN U.S.A.
C3167–8–1/98