AD AD8311-EVAL 50 db gsm pa controller Datasheet

50 dB GSM PA Controller
AD8311
Its high sensitivity allows control at low signal levels, thus
reducing the amount of power that needs to be coupled to the
detector. For convenience, the signal is internally ac-coupled.
This high-pass coupling, with a corner at approximately
0.016 GHz, determines the lowest operating frequency. Thus,
the source can be dc-grounded.
FEATURES
Complete RF detector/controller function
>50 dB range at 0.9 GHz (−48 dBm to +3 dBm re 50 Ω)
Accurate scaling from 0.1 GHz to 2.5 GHz
Temperature-stable linear-in-dB response
Log slope of 23 mV/dB, intercept at −60 dBm at 0.9 GHz
True integration function in control loop
Low power: 20 mW at 2.7 V
The AD8311 provides a voltage output, VAPC, which has the
voltage range and current drive to directly connect to the gain
control pin of most handset power amplifiers. VAPC can swing
from 300 mV above ground to within 200 mV below the supply
voltage. Load currents of up to 6 mA can be supported.
APPLICATIONS
Single, dual, and triple band mobile handset (GSM, DCS,
EDGE)
Transmitter power control
The setpoint control input is applied to pin VSET and has an
operating range of 0.25 V to 1.4 V. The associated circuit
determines the slope and intercept of the linear-in-dB
measurement system; these are nominally 23.6 mV/dB and
−59.7 dBm at 0.9 GHz. Further simplifying the application of
the AD8311, the input resistance of the setpoint interface is over
35 MΩ, and the bias current is typically 0.26 µA.
GENERAL DESCRIPTION
The AD8311 is a complete low cost subsystem for the precise
control of RF power amplifiers operating in the frequency range
0.1 GHz to 2.5 GHz and over a typical dynamic range of 50 dB.
It is intended for use in cellular handsets and other batteryoperated wireless devices. The log amp technique provides a
much wider measurement range and better accuracy than
controllers using diode detectors. In particular, its temperature
stability is excellent over a specified range of −40°C to +85°C.
The AD8311 is available in a 6-ball wafer-level chip scale
package (WLCSP), 1.0 mm × 1.5 mm, and consumes 7.6 mA
from a 2.7 V to 5.5 V supply.
FUNCTIONAL BLOCK DIAGRAM
LOW NOISE
BAND GAP
REFERENCE
LOW NOISE
GAIN BIAS
VPOS
VAPC
⋅ 1.35
DET
DET
DET
DET
DET
HI-Z
LOW NOISE
RAIL-TO-RAIL BUFFER
RFIN
10dB
10dB
FLTR
10dB
V
OFFSET
COMPENSATION
INTERCEPT
POSITIONING
COMM
I
VSET
23mV/dB
250mV TO
1.4V = 50dB
05545-001
10dB
Figure 1.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
© 2005 Analog Devices, Inc. All rights reserved.
AD8311
TABLE OF CONTENTS
Specifications..................................................................................... 3
Mobile Handset Power Control Example ............................... 15
Absolute Maximum Ratings............................................................ 5
Power-On and Power-Off ......................................................... 16
ESD Caution.................................................................................. 5
Input Coupling Options ............................................................ 16
Pin Configuration and Function Descriptions............................. 6
Temperature Drift ...................................................................... 17
Typical Performance Characteristics ............................................. 7
Device Calibration and Error Calculation.............................. 17
Theory of Operation ...................................................................... 11
Selecting Calibration Points to Improve Accuracy over a
Reduced Range ........................................................................... 18
Basic Theory................................................................................ 11
Controller-Mode Log Amps ..................................................... 12
Control Loop Dynamics............................................................ 12
Basic Connections ...................................................................... 14
Range on VSET and RFIN......................................................... 14
Device Handling......................................................................... 19
Evaluation Board ............................................................................ 20
Outline Dimensions ....................................................................... 22
Ordering Guide .......................................................................... 22
Transient Response..................................................................... 15
REVISION HISTORY
6/05—Revision 0: Initial Version
Rev. 0 | Page 2 of 24
AD8311
SPECIFICATIONS
VPOS = 2.7 V, Frequency = 0.1 GHz, TA = 25°C, 52.3 Ω termination on RFIN, light condition = 600 lux, unless otherwise noted.
Table 1.
Parameter
SPECIFIED FREQUENCY RANGE
MEASUREMENT MODE1 (f = 0.1 GHz)
Input Impedance
±1 dB Dynamic Range
Maximum Input Level
Minimum Input Level
Slope2
Intercept2
VSET Voltage—High Power In
VSET Voltage—Low Power In
Temperature Sensitivity
MEASUREMENT MODE (f = 0.9 GHz)
Input Impedance
±1 dB Dynamic Range
Maximum Input Level
Minimum Input Level
Slope2
Intercept2
VSET Voltage—High Power In
VSET Voltage—Low Power In
Temperature Sensitivity
MEASUREMENT MODE (f = 1.9 GHz)
Input Impedance
±1 dB Dynamic Range
Maximum Input Level
Minimum Input Level
Slope2
Intercept2
VSET Voltage—High Power In
VSET Voltage—Low Power In
Temperature Sensitivity
Conditions
RFIN (Pin 6)
Min
0.1
No termination resistor on RFIN
TA = +25°C
−40°C < TA < +85°C
±1 dB error
±1 dB error
+2
21.5
−66
PIN = –10 dBm
PIN = –40 dBm
PIN = –10 dBm
25°C ≤ TA ≤ +85°C
−40°C ≤ TA ≤ +25°C
No termination resistor on RFIN
TA = +25°C
−40°C < TA < +85°C
±1 dB error
±1 dB error
PIN = –10 dBm
PIN = –40 dBm
PIN = –10 dBm
25°C ≤ TA ≤ +85°C
−40°C ≤ TA ≤ +25°C
No termination resistor on RFIN
TA = +25°C
−40°C < TA < +85°C
±1 dB error
±1 dB error
PIN = –10 dBm
PIN = –40 dBm
PIN = –10 dBm
25°C ≤ TA ≤ +85°C
−40°C ≤ TA ≤ +25°C
Rev. 0 | Page 3 of 24
Typ
2140 || 1.97
47
46
+2.6
−44.5
23.8
−58.9
1.16
0.45
Max
2.5
−44
25.5
−51
Unit
GHz
Ω || pF
dB
dB
dBm
dBm
mV/dB
dBm
V
V
−0.0020
+0.0121
dB/°C
dB/°C
370 || 1.58
51
50
+2.8
−47.9
23.6
−59.7
1.17
0.46
Ω || pF
dB
dB
dBm
dBm
mV/dB
dBm
V
V
0.0015
0.0094
dB/°C
dB/°C
180 || 1.67
42
41
−5.6
−48.0
22.7
−60.8
1.15
0.47
Ω || pF
dB
dB
dBm
dBm
mV/dB
dBm
V
V
0.0056
0.0077
dB/°C
dB/°C
AD8311
Parameter
MEASUREMENT MODE (f = 2.5 GHz)
Input Impedance
±1 dB Dynamic Range
Maximum Input Level
Minimum Input Level
Slope2
Intercept2
VSET Voltage—High Power In
VSET Voltage—Low Power In
Temperature Sensitivity
OUTPUT INTERFACE
Minimum Output Voltage
Maximum Output Voltage
vs. Temperature
General Limit
Output Current Drive
Output Noise
Small Signal Bandwidth
Fall Time
Rise Time
Slew Rate
Response Time
VSET INTERFACE
Nominal Input Range
Logarithmic Scale Factor
Bias Current Source
Input Resistance
Slew Rate
POWER INTERFACE
Supply Voltage
Quiescent Current
vs. Temperature
Power-On Time
Power-Off Time
1
2
Conditions
Min
No termination resistor on RFIN
TA = +25°C
−40°C < TA < +85°C
±1 dB error
±1 dB error
PIN = –10 dBm
PIN = –40 dBm
PIN = –10 dBm
25°C ≤ TA ≤ 85°C
−40°C ≤ TA ≤ +25°C
VAPC (Pin 2)
VSET ≤ 150mV
IOUT = 3 mA
RL = ∞
85°C, VPOS = 3 V, IOUT = 6 mA
2.7 V ≤ VPOS ≤ 5.5 V, RL = ∞
VSET = 1.5 V, RFIN = –50 dBm, source/sink
RF Input = 2 GHz, 0 dBm, fNOISE = 100 kHz, CFLT = 220 pF
RFIN = −10 dBm; from FLTR to VAPC
Input level = off to 0 dBm, 90% to 10%
Input level = 0 dBm to off, 10% to 90%
90% – 10%, VSET = 0.3 V, open loop
FLTR = Open
VSET (Pin 3)
RFIN = 0 dBm; measurement mode
RFIN = −50 dBm; measurement mode
0.2
2.3
2.4
2.54
RFIN = −10 dBm; VSET = 1.4 V
Typ
Max
Unit
164 || 1.55
42
41
−6.2
−47.7
22.5
−60.6
1.14
0.46
Ω || pF
dB
dB
dBm
dBm
mV/dB
dBm
V
V
−0.0004
+0.0090
dB/°C
dB/°C
0.3
0.4
2.55
2.65
VPOS – 0.1
5/200
170
30
120
270
15
130
V
V
V
V
V
mA/µA
nV/√Hz
MHz
ns
ns
V/µs
ns
1.4
0.27
0.04
0.26
36
14
V
V
dB/mV
µA
MΩ
V/µs
VPOS (Pin 1)
2.7
5
–40°C ≤ TA ≤ +85°C
Time from VPOS high to VAPC within 1% of final value,
VSET ≤ 200 mV
Time from VPOS low to VAPC within 1% of final value,
VSET ≤ 200 mV
VAPC (Pin 2) to VSET (Pin 3) with inversion stage, sinusoidal input signal.
Mean and standard deviation specifications are available in Table 4.
Rev. 0 | Page 4 of 24
7.6
8.2
3
5.5
10.7
12.9
10
V
mA
mA
µs
100
2000
ns
AD8311
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameters
Supply Voltage VPOS
VAPC, VSET
RFIN
Equivalent Voltage
Internal Power Dissipation
θJA (WLCSP)
Maximum Junction Temperature
Operating Temperature Range
Storage Temperature Range
Ratings
5.5 V
0 V, VPOS
17 dBm
1.6 V rms
60 mW
200°C/W
125°C
−40°C to +85°C
−65°C to +150°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. 0 | Page 5 of 24
AD8311
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
VPOS
1
6
RFIN
VAPC
2
5
COMM
VSET
3
4
FLTR
TOP VIEW
Not to Scale
05545-002
BUMP 1
INDICATOR
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No.
1
2
3
4
5
6
Mnemonic
VPOS
VAPC
VSET
FLTR
COMM
RFIN
Function
Positive Supply Voltage: 2.7 V to 5.5 V.
Output. Control voltage for gain control element.
Setpoint Input. Nominal input range 0.25 V to 1.4 V.
Integrator Capacitor. Connect between FLTR and COMM.
Device Common (Ground).
RF Input.
Rev. 0 | Page 6 of 24
AD8311
TYPICAL PERFORMANCE CHARACTERISTICS
VPOS = 2.7 V; TA = 25°C; CFLT = open; light condition = 600 lux, 52.3 Ω termination; unless otherwise noted. Colors: +25°C = black,
−40°C = blue, +85°C = red.
4
10
0.9GHz
1.9GHz
2.5GHz
3
0.1GHz
1.9GHz
2
–10
2.5GHz
ERROR (dB)
–30
–1
–50
–2
1.0
1.2
–3
0.2
1.6
1.4
0.4
0.6
0.8
1.0
Figure 3. Input Amplitude vs. VSET
10
4
0
3
2
–20
1
–30
0
–60
0.2
–1
+85°C
ERROR AT +85°C AND –40°C
BASED ON DEVIATION FROM
SLOPE AND INTERCEPT AT +25°C
0.4
0.6
0.8
1.0
ERROR (dB)
–10
RF INPUT AMPLITUDE (dBm)
+85°C
–40°C
1.4
–3
1.6
10
RF INPUT AMPLITUDE (dBm)
–30
0
–40
ERROR (dB)
1
–1
1.0
–40°C
+25°C
1.2
–1
1.4
ERROR AT +85°C AND –40°C
BASED ON DEVIATION FROM
SLOPE AND INTERCEPT AT +25°C
0.4
0.6
0.8
1.0
–2
+25°C
1.2
–3
1.6
1.4
4
+25°C
–40°C
–3
1.6
VSET (V)
Figure 5. Input Amplitude and Log Conformance vs. VSET at 0.9 GHz
−40°C, +25°C, and +85°C
3
+85°C
–10
2
–20
1
–30
0
–40
–1
+25°C
+85°C
–40°C
–50
–2
05545-019
RF INPUT AMPLITUDE (dBm)
–20
0.8
–40
0
2
+85°C
0.6
0
+85°C
10
3
–10
0.4
–30
Figure 7. Input Amplitude and Log Conformance vs. VSET at 1.9 GHz
−40°C, +25°C, and +85°C
+85°C
–60
0.2
1
VSET (V)
4
+25°C
–40°C
ERROR AT +85°C AND –40°C
BASED ON DEVIATION FROM
SLOPE AND INTERCEPT AT +25°C
–20
–60
0.2
Figure 4. Input Amplitude and Log Conformance vs. VSET at 0.1 GHz
−40°C, +25°C, and +85°C
–50
2
–50
–2
VSET (V)
0
–10
+25°C
1.2
3
+85°C
–40°C
05545-018
RF INPUT AMPLITUDE (dBm)
0
+25°C
–40°C
ERROR (dB)
4
+25°C
–40°C
–50
1.6
1.4
Figure 6. Log Conformance vs. VSET
10
–40
1.2
VSET (V)
VSET (V)
05545-021
0.8
–60
0.2
–2
ERROR AT +85°C AND –40°C
BASED ON DEVIATION FROM
SLOPE AND INTERCEPT AT +25°C
0.4
0.6
0.8
ERROR (dB)
0.6
1.0
1.2
1.4
–3
1.6
VSET (V)
Figure 8. Input Amplitude and Log Conformance vs. VSET at 2.5 GHz
−40°C, +25°C, and +85°C
Rev. 0 | Page 7 of 24
05545-022
0.4
0.9GHz
0
–40
–60
0.2
0.1GHz
1
05545-020
–20
05545-017
RF INPUT AMPLITUDE (dBm)
0
AD8311
3
3
2
2
+85°C
1
+85°C
ERROR (dB)
0
–1
0
–1
+25°C
+25°C
–2
–2
–3
–60
–50
–40
–30
–20
–10
–40°C
05545-023
–40°C
0
–3
–60
10
–50
RF INPUT AMPLITUDE (dBm)
–40
–30
–20
–10
05545-026
ERROR (dB)
1
0
10
RF INPUT AMPLITUDE (dBm)
Figure 9. Distribution of Error over Temperature After Ambient Normalization
vs. Input Amplitude at 0.1 GHz
Figure 12. Distribution of Error over Temperature After Ambient Normalization
vs. Input Amplitude at 1.9 GHz
3
3
2
2
+85°C
+85°C
1
ERROR (dB)
0
–1
0
–1
+25°C
+25°C
–2
05545-024
–2
–40°C
–3
–60
–50
–40
–30
–20
–10
0
05545-027
ERROR (dB)
1
–40°C
–3
–60
10
RF INPUT AMPLITUDE (dBm)
–50
–40
–30
–20
–10
0
10
RF INPUT AMPLITUDE (dBm)
Figure 10. Distribution of Error over Temperature after Ambient Normalization
vs. Input Amplitude at 0.9 GHz
Figure 13. Distribution of Error over Temperature after Ambient Normalization
vs. Input Amplitude at 2.5 GHz
3.5
3.3
3.1
VAPC (V)
0mA
2.9
100MHz
2mA
4mA
900MHz
2.7
6mA
05545-025
2.3
2.7
2.5GHz
2.8
2.9
3.0
3.1
3.2
3.3
3.4
3.5
VPOS (V)
START FREQUENCY = 0.05GHz
STOP FREQUENCY = 3.5GHz
Figure 11. Maximum VAPC Voltage vs. Supply Voltage by Load Current
1.9GHz
05545-028
2.5
Figure 14. Input Impedance vs. Frequency, No Termination Resistor on RFIN
Rev. 0 | Page 8 of 24
AD8311
VAPC
200mV PER
VERTICAL
DIVISION
VAPC
1V PER
VERTICAL
DIVISION
GND
GND
RF
INPUT
2µs PER
HORIZONTAL
DIVISION
GND
PULSED RF
0.1GHz, 0dBm 100ns PER
HORIZONTAL
DIVISION
05545-029
GND
Figure 15. Power-On and -Off Response with VSET Grounded
10MHz REF
R AND S SMT03 OUTPUT
SIGNAL
GENERATOR
EXT TRIG STANFORD DS345
PULSE
GENERATOR
Figure 18. VAPC Response Time, Full-Scale Amplitude Change, Open-Loop
TRIG
OUT
R AND S SMT03
SIGNAL
GENERATOR
PULSE
MODULATION
MODE
PULSE OUT
RF OUT
10MHz REF
OUTPUT
EXT TRIG
PULSE MODE IN
OUT
49.9Ω
2
VAPC
COMM 5
3
VSET
FLTR 4
2.7V
–3dB
52.3Ω
0.3V
TRIG
TEK P6205
FET PROBE
TEK TDS694C
SCOPE
2
VAPC
COMM 5
3
VSET
FLTR 4
10k
100k
FREQUENCY (Hz)
NC
TRIG
TEK P6205
FET PROBE
1M
10M
TEK TDS694C
SCOPE
Figure 19. Test Setup for VAPC Response Time
10k
NOISE SPECTRAL DENSITY (nV/ Hz)
CFLT = 220pF
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
–120
–130
PHASE (Degrees)
CFLT = 0pF
05545-039
AMPLITUDE (dB)
RFIN 6
NC = NO CONNECT
Figure 16. Test Setup for Power-On and -Off Response with VSET Grounded
1k
VPOS
05545-030
TEK P6205
FET PROBE
100
1
52.3Ω
220pF
45
40
35
30
25
20
15
10
5
0
–5
–10
–15
–20
–25
–30
–35
–40
10
AD8311
0.1µF
CFLT = 220pF, RF INPUT = 2GHz
–50dBm
1k
–40dBm
0dBm
–20dBm
100
10
100
–38dBm
–35dBm
–10dBm
–30dBm
05545-034
RFIN 6
TRIG
OUT
RF
SPLITTER –3dB
732Ω
AD8311
VPOS
PICOSECOND
PULSE LABS
PULSE
GENERATOR
RF OUT
AD811
1
05545-032
1V PER
VERTICAL
DIVISION
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 20. VAPC Noise Spectral Density
Figure 17. AC Response from VSET to VAPC
Rev. 0 | Page 9 of 24
10M
05545-033
VPOS
AD8311
25
–55
–57
INTERCEPT (dBm)
SLOPE (mV/dB)
23
–40°C
+25°C
23
–59
+25°C
–61
–40°C
+85°C
22
05545-035
21
0
0.5
1.0
1.5
2.0
05545-037
–63
+85°C
–65
0
2.5
0.5
FREQUENCY (GHz)
1.0
1.5
2.0
2.5
FREQUENCY (GHz)
Figure 21. Slope vs. Frequency
−40°C, +25°C, and +85°C
Figure 23. Intercept vs. Frequency
−40°C, +25°C, and +85°C
25
–58
–59
0.1GHz
0.1GHz
INTERCEPT (dBm)
SLOPE (mV/dB)
23
0.9GHz
23
1.9GHz
2.5GHz
0.9GHz
–60
1.9GHz
–61
2.5GHz
22
3.0
3.5
4.0
4.5
5.0
5.5
–63
2.5
VPOS (V)
05545-038
21
2.5
05545-036
–62
3.0
3.5
4.0
4.5
5.0
5.5
VPOS (V)
Figure 22. Slope vs. Supply Voltage
Figure 24. Intercept vs. Supply Voltage
Table 4. Typical Specifications at Selected Frequencies at 25°C (Mean and Sigma)
Frequency (GHz)
0.1
0.9
1.9
2.5
Slope (mV/dB)
Mean
Sigma
23.8
0.1
23.6
0.1
22.7
0.1
22.5
0.1
Intercept (dBm)
Mean
Sigma
−58.9
0.4
−59.7
0.4
−60.8
0.5
−60.6
0.5
Rev. 0 | Page 10 of 24
±1 dB Dynamic Range
Low Point (dBm)
High Point (dBm)
Mean
Sigma
Mean
Sigma
−44.5
0.8
+2.6
0.3
−47.9
0.3
+2.8
0.1
−48.0
0.6
−5.6
0.6
−47.7
0.6
−6.2
0.5
AD8311
THEORY OF OPERATION
corresponds to 20 dB, VSLP/20 represents the volts/dB. For the
AD8311, a nominal (low frequency) slope of 24 mV/dB was
chosen, and the intercept VZ was placed at −59 dBm for a sine
wave input (251 µV rms). However, both the slope and the
intercept are dependent on frequency.
The AD8311 is a wideband logarithmic amplifier (log amp)
similar in design to the AD8313, AD8314, and AD8315. Like
the AD8315, it is strictly optimized for use in power control
applications rather than as a measurement device. Figure 25
shows the main features in block schematic form. The output
(Pin 2, VAPC) is intended to be applied directly to the
automatic power-control (APC) pin of a power amplifier
module.
Keeping in mind that log amps do not respond to power but
only to voltages and that the calibration of the intercept is
waveform dependent and is only quoted for a sine wave signal,
the equivalent power response can be written as
BASIC THEORY
VOUT = VDB (PIN − PZ )
Logarithmic amplifiers provide a type of compression in which
a signal having a large range of amplitudes is converted to one
of a smaller range. The use of the logarithmic function uniquely
results in the output representing the decibel value of the input.
The fundamental mathematical form is
VOUT = VSLP log 10
VIN
VZ
(2)
where the input power PIN and the equivalent intercept PZ are
both expressed in dBm (thus, the quantity in parentheses is
simply a number of decibels), and VDB is the slope expressed in
mV/dB. For a log amp having a slope VDB of 24 mV/dB and an
intercept at −59 dBm, the output voltage for an input power of –
30 dBm is 0.024 [−30 − (−57)] = 0.696 V.
(1)
Further details about the structure and function of log amps can
be found in data sheets for other log amps produced by Analog
Devices. Refer to the data sheets for the AD640 and AD8307,
both of which include a detailed discussion of the basic
principles of operation and explain why the intercept depends
on waveform, an important consideration when complex
modulation is imposed on an RF carrier.
where:
VIN is the input voltage.
VZ is called the intercept (voltage) because when VIN = VZ the
argument of the logarithm is unity and thus the result is zero.
VSLP is called the slope (voltage), which is the amount by which
the output changes for a certain change in the ratio (VIN/VZ).
When BASE-10 logarithms are used, denoted by the function
log10, VSLP represents the volts/decade, and since a decade
(PRECISE GAIN
CONTROL)
(PRECISE SLOPE
CONTROL)
LOW NOISE
GAIN BIAS
LOW NOISE
BAND GAP
REFERENCE
VPOS
(CURRENT-MODE SIGNAL)
DET
DET
DET
DET
DET
HI-Z
LOW NOISE
RAIL-TO-RAIL BUFFER
10dB
10dB
OFFSET
COMPENSATION
10dB
FLTR
10dB
(CURRENTNULLING
MODE)
INTERCEPT
POSITIONING
(WEAK GM STAGE)
Figure 25. Block Schematic
Rev. 0 | Page 11 of 24
(CURRENT-MODE
FEEDBACK)
V
(SMALL INTERNAL
FILTER CAPACITOR
FOR GHz RIPPLE)
I
VSET
23mV/dB
250mV TO
1.4V = 50dB
05545-003
RFIN
COM
VAPC
⋅ 1.35
AD8311
The intercept need not correspond to a physically realizable
part of the signal range for the log amp. Thus, the specified
intercept is −58.9 dBm at 0.1 GHz, whereas the smallest input
for accurate measurement (a +1 dB error) at this frequency is
higher, about −44.5 dBm. At 2.5 GHz, the +1 dB error point
shifts to −47.7 dBm. This positioning of the intercept is
deliberate and ensures that the VSET voltage is within the
capabilities of certain digital-to-analog converters (DACs),
whose outputs cannot swing below 200 mV. Figure 26 shows the
100 MHz response of the AD8311; the vertical axis represents
not the output (at the VAPC pin) but the value required at the
power control pin (VSET) to null the control loop. This is
explained in the Controller-Mode Log Amps section.
1.5
1.211V @ –8dBm
SLOPE = 23.8mV/dB
ACTUAL
0.5
05545-040
448mV @ –40dBm
IDEAL
0
100µV
1mV
–67dBm
–47dBm
–58.9dBm
10mV
–27dBm
VIN, PIN
100mV
–7dBm
1V (RMS)
13dBm (RE 50Ω)
This is achieved by converting the difference between the sum
of the detector outputs (still in current form) and an internally
generated current proportional to VSET to a single-sided
current-mode signal. This, in turn, is converted to a voltage (at
Pin 4, FLTR, the low-pass filter capacitor node) to provide a
close approximation to an exact integration of the error
between the power present in the termination at the input of the
AD8311 and the setpoint voltage. Finally, the voltage developed
across the ground-referenced filter capacitor CFLT is buffered by
a special low noise amplifier of low voltage gain (×1.35) and
presented at Pin 2 (VAPC) for use as the control voltage for the
RF power amplifier. This buffer can provide rail-to-rail swings
and can drive a substantial load current, including large
capacitors. Note that the RF power amplifier is assumed to have
a positive slope with RF power increasing monotonically with
an increasing APC control voltage.
CONTROL LOOP DYNAMICS
Figure 26. Basic Calibration of the AD8311 at 0.1 GHz
CONTROLLER-MODE LOG AMPS
The AD8311 combines the two key functions required for the
measurement and control of the power level over a moderately
wide dynamic range. First, it provides the amplification needed
to respond to small signals in a chain of four amplifier/limiter
cells (see Figure 25), each having a small signal gain of 10 dB
and a bandwidth of approximately 3.5 GHz. At the output of
each of these amplifier stages is a full-wave rectifier, essentially a
square law detector cell that converts the RF signal voltages to a
fluctuating current having an average value that increases with
signal level. A further passive detector stage is added before the
first stage. These five detectors are separated by 10 dB, spanning
some 50 dB of dynamic range. Their outputs are each in the
form of a differential current, making summation a simple
matter. It is readily shown that the summed output can closely
approximate a logarithmic function. The log conformance
error, which is the overall accuracy at the extremes of this total
range viewed as the deviation from an ideal logarithmic
response, can be judged by reference to Figure 6, which shows
that errors across the central 40 dB are moderate.
In order to understand how the AD8311 behaves in a complete
control loop, an expression for the current in the integration
capacitor as a function of the input PIN and the setpoint voltage
VSET must be developed. Refer to Figure 27.
DIRECTIONAL
COUPLER
Rev. 0 | Page 12 of 24
POUT
RF PA
PCW
RF DRIVE:
UP TO
2.5GHz
IDET = ISLP PIN + IINT
RFIN
6
VIN
LOGARITHMIC
RF DETECTION
SUBSYSTEM IDET
FLTR
4
VAPC
1.35
2
IERR
CFLT
VSET
3
SETPOINT
VSET INTERFACE
ISET = VSET/RSET
Figure 27. Behavioral Model of the AD8311
05545-047
VSET (V)
1.0
In a device intended for measurement applications, this current
would then be converted to an equivalent voltage, to provide the
log(VIN) function shown in Equation 1. However, the design of
the AD8311 differs from standard practice in that its output
needs to be a low noise control voltage for an RF power
amplifier, not a direct measure of the input level. Further, it is
highly desirable that this voltage be proportional to the timeintegral of the error between the actual input VIN and a dc
voltage VSET (applied to Pin 3, VSET). VSET defines the setpoint,
a target value for the power level typically generated by a DAC.
AD8311
First, the summed detector currents are written as a function of
the input power.
I DET = I SLP × PIN + I INT
Equation 6 can be restated as
V APC ( s ) =
(3)
VSET − VSLP × PIN − V INT
(7)
sT
where:
where:
IDET is the partially filtered demodulated signal, whose steadystate average value is extracted through the subsequent
integration step.
ISLP is the slope, which has a value of 5.75 µA/dB.
PIN is the input power in dBm (assuming 50 Ω input match).
IINT is the current intercept which, as previously noted, is
dependent on the RF waveform (not the envelope). Assuming
a sinusoidal input, IINT is 350 µA.
VSLP is ISLP × RSET, which has a value of 24 mV/dB.
VINT is the voltage intercept given by IINT × RSET, which has a
value of 1.44 V.
T is the effective time constant for the integration and is equal
to RSET × CFLT/1.35. The factor of 1.35 arises because of the
voltage gain of the buffer.
So the open-loop integration time constant can be written as
TOpenLoop = RSET × CFLT 1.35
The current generated by the setpoint interface is simply
I SET = VSET RSET
(4)
where the RSET resistor is 4.1 kΩ. The difference between this
current and IDET is applied to the loop filter capacitor CFLT. At
this point note that the inclusion of a filter resistor, RFLT, can be
helpful in improving the phase margin at low powers where the
PA control gain (that is, ∂POUT/∂VAPC) is large, as is described
later in this section. For now assume that RFLT is zero. It follows
that the voltage appearing on this capacitor, VFLT, is the timeintegral of the difference current.
VFLT (s ) = (I SET − I DET ) sC FLT
POUT =
I SET + I SLP × 30 − I INT × 1.35 × (G PA (1 + sτ PA )) × (1 sC FLT )
[
(5)
VSET RSET − I SLP × PIN − I INT
sC FLT
(6)
The control output VAPC is slightly greater than this, since the gain
of the output buffer is ×1.35, plus a slight offset voltage. The
polarity is such that VAPC rises to its maximum value for any value
of VSET greater than the equivalent value of PIN. That is, the
AD8311 seeks to drive the RF power to its maximum value
whenever it falls below the setpoint. The use of exact integration
results in a dc error that is theoretically zero, and the logarithmic
detection law would ideally result in a constant response time
following a step change of either the setpoint or the power level if
the power-amplifier control function were likewise linear-in-dB.
This latter condition is rarely true, however, and it follows that in
practice the loop response time depends on the power level. This
effect can strongly influence the design of the control loop.
ISET
IERR
+
To assess the closed-loop performance, refer to the block
diagram in Figure 28 and calculate the loop transfer function.
In general, the buffer time constant (τBUFFER) and the log amp
time constant (τLOGAMP) can be neglected, except in the case of
very high PA control function gains (> than 500 dB/V) and/or
very wide PA control port bandwidths. Assuming that the
frequency response of the output buffer and the log amp can be
neglected, the overall transfer function can be expressed as
Equation 9 assumes that the next parasitic pole in the control
loop comes from the PA. For a typical PA, a 1 MHz pole is not
unusual, making this a good assumption. Therefore, except for
in the case of a very wide bandwidth on the PA control port
(>10 MHz), the response time and stability of the control loop is
mainly determined by the characteristics of the PA. This is true
for both the gain and the phase response. It is essential to
understand both the magnitude and frequency response of the
power amplifier control port.
ISLP PIN + IINT
PIN
VAPC
GPA (dB/V)
POUT
1 + sτPA
IDET
1 + sτLOGAMP
(9)
The input power to the log amp, PIN, is given in dBm and
therefore is simply POUT of the PA minus the coupler value,
typically −30 dB, or PIN = POUT − 30.
1.35
1 + sτ BUFFER
_
]
1 + I SLP × 1.35 × (G PA (1 + sτ PA )) sC FLT
Here, GPA is the PA control function gain ∂POUT/∂VAPC given in
dB/V, and the factor of −30 is due to the coupler.
VFLT
1
+ RFLT
sCFLT
[
]
–30dB
COUPLER
Figure 28. Control Loop Block Diagram
Rev. 0 | Page 13 of 24
05545-048
=
(8)
AD8311
Continuing with the stability analysis, the gain of the control
loop can be expressed as
∂POUT
∂I SET
=
k
1 + kA
DIRECTIONAL
COUPLER
(10)
POWER
AMP
RFIN
GAIN
CONTROL
VOLTAGE
ATTENUATOR
VAPC
where:
AD8311
VSET
RFIN
1 + sτ PA
(dB/A)
52.3Ω
(11)
CFLT
RFLT
A = I SLP (A/dB)
DAC
FLTR
(12)
05545-008
k=
1.35 × G PA × (1 sC FLT )
Figure 30. Typical Application
The effect of the zero resistor, RFLT, can be easily included by
replacing (1/sCFLT) with (RFLT + 1/sCFLT). The criteria for loop
stability can be derived by setting the denominator of
Equation 10 equal to 0, giving
0 = 1+
1.35 × (1 + sR FLT C FLT )
(13)
(1 + sτ PA ) × sC FLT (G PA × I SLP )
From Equation 13, the closed-loop integration time constant is
given by
TClosedLoop = C FLT (GPA × I SLP × 1.35 )
(14)
The gain and phase margins of the control loop can be deduced
from the Bode plots of Equation 13.
BASIC CONNECTIONS
Figure 29 shows the basic connections for operating the
AD8311, and Figure 30 shows a block diagram of a typical
application. The AD8311 is typically used in the RF power
control loop of a mobile handset.
A supply voltage of 2.7 V to 5.5 V is required for the AD8311.
The supply to the VPOS pin should be decoupled with a low
inductance 0.1 µF surface-mount ceramic capacitor, close to the
device. The AD8311 has an internal input coupling capacitor,
which negates the need for external ac-coupling. This capacitor,
along with the low frequency input impedance of the device of
approximately 2.14 kΩ, sets the minimum usable input
frequency to around 0.016 GHz. A broadband 50 Ω input
match is achieved in this example by connecting a 52.3 Ω
resistor between RFIN and ground. A Smith chart plot of input
impedance vs. frequency is shown in Figure 14. Other coupling
methods are also possible (see the Input Coupling Options
section).
AD8311
1 VPOS
+VS
(2.7V TO 5.5V)
2 VAPC
VAPC
VSET
3
VSET
RFIN
RFIN 6
A setpoint voltage is applied to VSET from the controlling
source (generally this is a DAC). Any imbalance between the RF
input level and the level corresponding to the setpoint voltage is
corrected by the AD8311’s VAPC output that drives the gain
control terminal of the PA. This restores a balance between the
actual power level sensed at the input of the AD8311 and the
value determined by the setpoint. This assumes that the gain
control sense of the variable gain element is positive, that is, an
increasing voltage from VAPC tends to increase gain.
VAPC can swing from 200 mV to within 100 mV of the supply
rail and can source up to 6 mA. If the control input of the PA
needs to source current, a suitable load resistor can be
connected between VAPC and COMM. The output swing and
current sourcing capability of VAPC is shown in Figure 11.
RANGE ON VSET AND RFIN
The relationship between the RF input level and the setpoint
voltage follows from the nominal transfer function of the device
(see Figure 4, Figure 5, Figure 7, and Figure 8). At 0.9 GHz, for
example, a voltage of 1 V on VSET indicates a demand for
−18 dBm at RFIN. The corresponding power level at the output
of the power amplifier is greater than this amount due to the
attenuation through the directional coupler.
For setpoint voltages of less than approximately 150 mV, VAPC
unconditionally remains at its minimum level of approximately
300 mV. This feature can be used to prevent any spurious
emissions during power-up and power-down phases.
R1
52.3Ω
COMM 5
FLTR 4
CFLT
05545-007
C1
0.1µF
In a power control loop, the AD8311 provides both the detector
and controller functions. A sample of the power amplifier’s (PA)
output power is coupled to the RF input of the AD8311, usually
via a directional coupler. In dual mode applications, where there
are two PAs and two directional couplers, the outputs of the
directional couplers can be passively combined (both PAs will
never be turned on simultaneously) before being applied to the
AD8311.
Above 250 mV, VSET has a linear control range up to 1.4 V,
corresponding to a dynamic range of 50 dB. This results in a
slope of 23.8 mV/dB, or approximately 42.0 dB/V.
Figure 29. Basic Connections
Rev. 0 | Page 14 of 24
AD8311
results in the characteristic poles in the ac loop equation moving
off the real axis and thus becoming complex (and somewhat
resonant). This is a classic aspect of control loop design. The
lowest permissible value of CFLT needs to be determined
experimentally for a particular amplifier. For GSM and DCS
power amplifiers, CFLT typically ranges from 150 pF to 300 pF.
TRANSIENT RESPONSE
The time domain response of power amplifier control loops,
using any kind of controller, is only partially determined by the
choice of filter. In the case of the AD8311, the filter has a true
integrator form 1/sT as shown in Equation 7, with a time
constant given by Equation 8. The large signal step response is
also strongly dependent on the form of the gain-control law.
Nevertheless, some simple rules can be applied. When the filter
capacitor CFLT is very large it dominates the time domain
response, but the incremental bandwidth of this loop still varies
as VAPC traverses the nonlinear gain-control function of the PA.
This bandwidth is highest at the point where the slope of the
tangent drawn on the PA power-control curve is greatest—that
is, for power outputs near the center of the PA’s range—and is
much reduced at both the minimum and the maximum power
levels, where the slope of the gain control curve is lowest due to
its S-shaped form.
In many cases, some improvement in the worst-case response
time can be achieved by including a small resistor in series with
CFLT; this generates an additional zero in the closed-loop
transfer function, which serves to cancel a higher order pole in
the overall loop. A more complex filter network can be used to
minimize the settling time of the loop—for example, a
combination of the main capacitor, CFLT, shunted by a second
capacitor and resistor series.
MOBILE HANDSET POWER CONTROL EXAMPLE
Figure 31 shows a complete power amplifier control circuit for a
dual mode handset. The PF08123B (Hitachi), a dual mode
(GSM, DCS) PA, is driven by a nominal power level of +3 dBm.
The PA has a single gain control line; the band to be used is
selected by applying either 0 V or 2 V to the PA’s VCTL input.
Using smaller values of CFLT, the loop bandwidth generally
increases in inverse proportion to its value. Eventually, however,
a secondary effect appears due to the inherent phase lag in the
power amplifier’s control path, some of which can be due to
parasitic or deliberately added capacitance at the VAPC pin. This
3.5V
4.7µ F
4.7µ F
F1000pF
p0001
1000pF
BAND
SELECT
0V/+2V
POUT GSM
35dBm MAX
VCTL
PIN GSM
3dBm
LDC15D190A0007A
1
7
49.9Ω 4
PF08123B
8
3
PIN DCS
3dBm
VAPC
POUT DCS
32dBm MAX
T
TOOANTENNA
5
6
2
500Ω
ATTN
20dB
(OPTIONAL,
SEE TEXT)
0.1µF
+VS
2.7V
R21
600Ω
VPOS
RFIN 6
2
VAPC
COMM 5
3
VSET
FLTR 4
R1
52.3Ω
R31
1kΩ
150pF
1R2,
1.5kΩ
R3 OPTIONAL, SEE TEXT
Figure 31. Dual Mode (GSM/DCS) PA Control Example
Rev. 0 | Page 15 of 24
05545-049
8-BIT
RAMP DAC
0V–2.55V
AD8311
1
AD8311
Figure 32 shows the relationship between VSET and output
power (POUT) at 0.9 GHz. The overall gain control function is
linear in dB for a dynamic range of over 40 dB. Note that for
VSET voltages below 300 mV, the output power drops off steeply
as VAPC drops toward its minimum level of 300 mV.
3
40
–40°C
2
30
+85°C
0
10
–40°C
+25°C
+85°C
0
–1
–2
–10
During initialization and completion of the transmit sequence,
VAPC should be held at its minimum level of 300 mV by keeping
VSET below 150 mV.
ERROR (dB)
1
20
+25°C
–20
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
–3
1.6
SETPOINT VOLTAGE (V)
In this example, VSET is supplied by an 8-bit DAC that has an
output range from 0 V to 2.55 V or 10 mV per bit. This sets the
control resolution of VSET to 0.4 dB/bit (0.04 dB/mV times
10 mV). If finer resolution is required, the DAC’s output voltage
can be scaled using two resistors as shown. This converts the
DAC’s maximum voltage of 2.55 V down to 1.6 V and increases
the control resolution to 0.25 dB/bit.
A filter capacitor (CFLT) must be used to stabilize the loop. The
choice of CFLT depends to a large degree on the gain control
dynamics of the power amplifier, something that is frequently
poorly characterized, so some trial and error might be
necessary.
In this example, a 150 pF capacitor is used and a 1.5 kΩ series
resistor is included. This adds a zero to the control loop and
increases the phase margin, which helps to make the step
response of the circuit more stable when the PA output power is
low and the slope of the PA’s power control function is the
steepest.
A smaller filter capacitor can be used by inserting a series
resistor between VAPC and the control input of the PA. A series
resistor works with the input impedance of the PA to create a
resistor divider, which reduces the loop gain. The size of the
resistor divider ratio depends on the available output swing of
VAPC and the required control voltage on the PA.
05545-041
The operational setpoint voltage, in the range 250 mV to 1.4 V,
is applied to the VSET pin of the AD8311. This typically is
supplied by a DAC. The AD8311’s VAPC output drives the level
control pin of the power amplifier directly. VAPC reaches a
maximum value of approximately 2.5 V on a 2.7 V supply while
delivering the 3 mA required by the level control input of the
PA. This is more than sufficient to exercise the gain control
range of the PA.
This technique can also be used to limit the control voltage in
situations where the PA cannot deliver the power level being
demanded by VAPC. Overdrive of the control input of some
PAs causes increased distortion. It should be noted, however,
that if the control loop opens (that is, VAPC goes to its maximum
value in an effort to balance the loop), the quiescent current of
the AD8311 increases somewhat, particularly at supply voltages
greater than 3 V.
OUTPUT POWER (dBm)
Some of the output power from the PA is coupled off using a
dual-band directional coupler (Murata part number
LDC15D190A0007A). This has a coupling factor of
approximately +19 dB for the GSM band and +14 dB for DCS
and an insertion loss of 0.38 dB and 0.45 dB, respectively.
Because the PF08107B transmits a maximum power level of
+35 dBm for GSM and +32 dBm for DCS, additional
attenuation of 20 dB is required before the coupled signal is
applied to the AD8311. This results in peak input levels to the
AD8311 of −4 dBm (GSM) and −2 dBm (DCS). While the
AD8311 gives a linear response for input levels up to +2 dBm,
for highly temperature-stable performance at maximum PA
output power the maximum input level should be limited to
approximately −2 dBm (see Figure 5 and Figure 7). This does,
however, reduce the sensitivity of the circuit at the low end.
Figure 32. POUT vs. VSET at 0.9 GHz for Dual Mode Handset
Power Amplifier Application;
−40°C, +25°C, and +85°C
POWER-ON AND POWER-OFF
The AD8311 can be completely disabled by pulling the supply
voltage to ground. The voltage on VSET should be kept below
150 mV during power-on and power-off to prevent any
unwanted transients on VAPC.
INPUT COUPLING OPTIONS
The internal 5 pF coupling capacitor of the AD8311 and the low
frequency input impedance of 2.14 kΩ give a high-pass input
corner frequency of approximately 16 MHz. This sets the
minimum operating frequency. Figure 33, Figure 34, and
Figure 35 shows three options for input coupling. A broadband
resistive match can be implemented by connecting a shunt
resistor to ground at RFIN (Figure 33). This 52.3 Ω resistor
(other values can also be used to select different overall input
impedances) combines with the input impedance of the
AD8311 to give a broadband input impedance of 50 Ω. While
the input resistance and capacitance (CIN and RIN) of the
AD8311 vary from device to device by approximately ±20%, as
well as in the same device over a range of frequencies
(Figure 14), the dominance of the external shunt resistor means
that the variation in the overall input impedance is close to the
Rev. 0 | Page 16 of 24
AD8311
10
0
–10
AD8311
RFIN
RSHUNT
52.3Ω
CC
3
2
–20
1
–30
0
–40
–1
–50
–2
–60
0.2
CIN
4
–40°C
–20°C
0°C
+25°C
+45°C
+65°C
+85°C
ERROR (dB)
In Figure 34, the matching components are drawn as generic
reactances. Depending on the frequency, the input impedance,
and the availability of standard value components either a
capacitor or an inductor is used. As in the previous case, the
input impedance at a particular frequency is plotted on a Smith
Chart and matching components are chosen (shunt or series L,
shunt or series C) to move the impedance to the center of the
chart.
Figure 36 shows the log slope and error over temperature for a
0.9 GHz input signal. Error due to drift over temperature
consistently remains within ±1 dB and only begins to exceed
this limit when the ambient temperature goes above +65 °C and
below −20 °C. For all frequencies using a reduced temperature
range, higher measurement accuracy is achievable.
0.4
0.6
0.8
1.0
1.2
–3
1.6
1.4
05545-042
A reactive match can also be implemented as shown in
Figure 34. This is not recommended at low frequencies because
device tolerances dramatically vary the quality of the match due
to the large input resistance. For low frequencies, Figure 33 or
Figure 35 is recommended.
TEMPERATURE DRIFT
PIN (dBm)
tolerance of the external resistor. This method of matching is
most useful in wideband applications or in multiband systems
where there is more than one operating frequency.
VSET (V)
RIN
05545-009
Figure 36. Typical Drift at 900 GHz for Various Temperatures
DEVICE CALIBRATION AND ERROR CALCULATION
The measured transfer function of the AD8311 at 0.9 GHz is
shown in Figure 37. The figure shows plots of both input power
and calculated error vs. setpoint voltage.
Figure 33. Broadband Resistive Input Coupling Option
AD8311
CIN
RIN
05545-010
x2
The vertical axis represents the input power required at the RFIN
pin to null the control loop when a VSET voltage is applied. As
the setpoint voltage varies from about 0.2 V to 1.5 V, the
corresponding input power varies from −60 dBm to +10 dBm.
Figure 34. Narrow Band Reactive Input Coupling Option
10
PINIDEAL = (VSET1/ SLOPE) + INTERCEPT
ERROR (dB) = (PINIDEAL – PIN)
SLOPE = (VSET2 – VSET1)/(PIN1 – PIN2)
INTERCEPT = PIN1 – (VSET1 / SLOPE)
3
0
ANTENNA
–40°C
AD8311
RATTN
PA
CIN
RFIN (dBm)
STRIPLINE
+25°C
+85°C
+85°C
RIN
05545-011
RFIN
PIN2
–10
CC
4
VSET2
–20
2
1
–40°C
0
–30
+25°C
PIN1
–40
ERROR (dB)
RFIN
–1
INTERCEPT
VSET1
Figure 35. Series Attention Input Coupling Option
–2
–50
Figure 35 shows a third method for coupling the input signal
into the AD8311. A series resistor connected to the RF source
combines with the input impedance of the AD8311 to
resistively divide the input signal being applied to the input.
This has the advantage of very little power being tapped off in
RF power transmission applications.
–60
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
VSET (V)
–3
1.6
05545-043
x1
CC
Figure 37. Transfer Function of AD8311 at 0.9 GHz
Because slope and intercept vary from device to device, boardlevel calibration must be performed to achieve high accuracy.
Rev. 0 | Page 17 of 24
AD8311
Once slope and intercept have been calculated, an equation can
be written which allows calculation of an (unknown) power
based on the setpoint voltage.
–40°C
2
PIN1
–20
1
VSET1
+85°C
–30
0
+25°C
–40
–1
–50
–2
–60
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
–3
1.6
VSET (V)
(17)
Using Equation 17 as a reference for the ideal input power, the
log conformance error of the measured data can be calculated:
ERROR(dB) = (PIN,IDEAL − PIN,MEASURED )
(18)
Figure 37 includes a plot of the error at 25°C, the temperature at
which the AD8311 is calibrated. Note that the error is not zero.
This is because the AD8311 does not perfectly follow the ideal
VSET vs. PIN equation, even within its operating region. The error
at the calibration points (0.45 V and 1.15 V in this case) is,
however, equal to zero by definition.
Figure 37 also includes error plots for the output voltage at
−40°C and +85 °C. These error plots are calculated using the
slope and intercept at +25°C. This is consistent with calibration
in a mass-production environment where calibration at
temperature is not practical.
Figure 38. Output Voltage and Error vs. PIN with 2-Point Calibration at
Approximately 0.975 V and 1.3 V
Calibration points should be chosen to suit the application at
hand. In general, though, the calibration points should never be
chosen in the nonlinear portion of the log amp’s transfer
function (above 1.4 V or below 0.35 V in this case).
Figure 39 shows how calibration points can be adjusted to
increase dynamic range, but at the expense of linearity. In this
case the calibration points for slope and intercept are set at
0.37 V and 1.37 V. These points are at the end of the device’s
linear range. Once again at 25°C we see an error of 0 dB at the
calibration points. Note also that the range over which the
AD8311 maintains an error of less than ±0.5 dB is extended to
more than 45 dB at 25°C and more than 40 dB over
temperature. The disadvantage of this approach is that linearity
suffers, especially in the middle of the range.
10
SELECTING CALIBRATION POINTS TO IMPROVE
ACCURACY OVER A REDUCED RANGE
0
Figure 38 shows the same measured data as Figure 37. Notice
that accuracy is very high from −15 dBm to 0 dBm. Below
−15 dBm the error increases to about −2 dB. This is because the
calibration points have been changed to approximately 0.975 V
and 1.3 V.
PIN2
3
VSET2
–10
RFIN (dBm)
In some applications very high accuracy is required at just one
power level or over a reduced input range. For example, in a
wireless transmitter, the accuracy of the high power amplifier
(HPA) is most critical at or close to full power.
4
+85°C
+25°C
–40°C
2
1
–20
–40°C
+85°C
0
–30
ERROR (dB)
PIN = (V SET / SLOPE ) + INTERCEPT
+25°C
–40
–1
PIN1
VSET1
–50
–2
–60
0
0.2
0.4
0.6
0.8
VSET (V)
1.0
1.2
1.4
–3
1.6
05545-045
(16)
RFIN (dBm)
INTERCEPT = PIN 1 − (VSET 1 / SLOPE )
3
VSET2
–10
(15)
PIN2
ERROR (dB)
0
SLOPE = (VSET 2 − VSET 1 ) /(PIN 2 − PIN 1 )
4
+85°C
+25°C
–40°C
05545-044
10
In a control loop, calibration is performed by applying two
levels to the AD8311’s setpoint voltage and measuring the
corresponding power. The calibration points are generally
chosen to be within the linear-in-dB operating range of the
device (see Figure 37). Calculation of slope and intercept is
done using the equations
Figure 39. Dynamic Range Extension by Choosing Calibration Points that are
Close to the End of the AD8311’s Linear Range
Rev. 0 | Page 18 of 24
AD8311
This would be valid if the device transfer function perfectly
followed the ideal PIN = VSET / SLOPE + INTERCEPT equation.
However since a log amp in practice never perfectly follows this
equation (especially outside of its linear operating range), this
plot tends to artificially improve linearity and extend the
dynamic range. This plot is a useful tool for estimating
temperature drift at a particular power level with respect to the
(nonideal) response at ambient. However, achieving this level of
accuracy in an end application requires calibration at multiple
points in the device’s operating range.
4
+85°C
+25°C
–40°C
0
3
2
–10
–20
1
–40°C
–30
0
+25°C
ERROR (dB)
+85°C
–1
–40
–2
–50
DOES NOT TAKE INTO ACCOUNT TRANSFER
FUNCTIONS’ NONLINEARITIES AT +25°C
–60
0
0.2
0.4
0.6
0.8
VSET (V)
1.0
1.2
1.4
–3
1.6
05545-046
When we use this alternative technique, the error at ambient
becomes by definition equal to 0 (see Figure 40).
10
RFIN (dBm)
Another way of presenting the error function of a log amp
detector is shown in Figure 40. In this case, the dB error at hot
and cold temperatures is calculated with respect to the transfer
function at ambient. This is a key difference in comparison to
the previous plots. Up to now, all errors have been calculated
with respect to the ideal transfer function at ambient.
Figure 40. Error vs. Temperature with respect to Output Voltage at 25 °C
DEVICE HANDLING
The wafer-level chip scale package consists of solder bumps
connected to the active side of the die. The part is lead-free with
95.5% tin, 4.0% silver, and 0.5% copper solder bump
composition. The WLCSP package can be mounted on printed
circuit boards using standard surface-mount assembly
techniques; however, caution should be taken to avoid
damaging the die. See the AN-617 application note for
additional information. WLCSP devices are bumped die, and
exposed die can be sensitive to light condition, which can
influence specified limits.
Rev. 0 | Page 19 of 24
AD8311
EVALUATION BOARD
by default). For GSM/DCS handset power amplifiers, this
capacitor should typically range from 150 pF to 300 pF.
Figure 41 shows the schematic of the AD8311 WLCSP
evaluation board. The layout and silkscreen of the component
and circuit sides are shown in Figure 42 to Figure 45. The board
is powered by a single supply in the range 2.7 V to 5.5 V. The
power supply is decoupled by a 0.1 µF capacitor. A 100 pF
capacitor provides additional supply decoupling, but is not
necessary for basic operation.
A quasi-measurement mode (where the AD8311 delivers an
output voltage that is proportional to the log of the input signal)
can be implemented, to establish the relationship between VSET
and RFIN, by installing the two jumpers J1 and J2. This mimics
an AGC loop. To establish the transfer function of the log amp,
the RF input should be swept while the voltage on VSET is
measured, that is, the SMA connector labeled VSET now acts as
an output. This is the simplest method to validate operation of
the evaluation board. When operated in this mode, a large
capacitor (0.01 µF or greater) must be installed in C4 (filter
capacitor) to ensure loop stability. The op amp must be powered
with a nominal voltage of 2.7 V to 5.5 V with the VS supply.
Alternately, J3 can be installed to power the op amp with the
AD8311’s VPOS power supply.
Table 5 details the various configuration options of the
evaluation board.
For operation in controller mode, both jumpers J1 and J2
should be removed. The setpoint voltage is applied to VSET,
RFIN is connected to the RF source (PA output or directional
coupler), and VAPC is connected to the gain control pin of the
PA. When used in controller mode, a capacitor must be
installed in C4 for loop stability (R2 must also be installed, 0 Ω
C1
0.1µF
52.3Ω
C2
100pF
R1
AD8311
VPOS
1
VPOS
RFIN 6
2
VAPC
COMM 5
3
VSET
FTLR 4
RFIN
R3
VAPC
C3
(OPEN)
0Ω
R4
(OPEN)
C4
(OPEN)
R2
0Ω
VSET
VPOS
C6
0.1µF
R6
10kΩ
R10
(OPEN)
J1
J3
R7
16.2kΩ
VS
C5
0.1µF
R9
(OPEN)
R8
17.8kΩ
TO EDGE
CONNECTOR
R5
10kΩ
TO EDGE
CONNECTOR
C7
(OPEN)
Figure 41. Evaluation Board Schematic
Rev. 0 | Page 20 of 24
05545-012
J2
05545-013
05545-015
AD8311
05545-013
05545-016
Figure 44. Silkscreen of Component Side (WLCSP)
Figure 42. Layout of Component Side (WLCSP)
Figure 45. Silkscreen of Circuit Side (WLCSP)
Figure 43. Layout of Circuit Side (WLCSP)
Table 5. Evaluation Board Configuration Options
Component
VPOS, GND
R1
R3, R4, C3
Function
Supply and Ground Vector Pins.
Input Interface. The 52.3 Ω resistor in Position R1 combines with the AD8311’s internal
input impedance to give a broadband input impedance of around 50 Ω. Note that the
AD8311’s RF input is internally ac-coupled.
Output Interface. R4 and C3 can be used to check the response of VAPC to capacitive and
resistive loading. R3/R4 can be used to reduce the slope of VAPC.
C1, C2
Power Supply Decoupling. The nominal supply decoupling consists of a 0.1 µF capacitor at
C1. C2 can be used for additional supply decoupling.
C4, R2
Filter Capacitor. The response time of VAPC can be modified by placing a capacitor between
FLTR (Pin 4) and ground. The control loop phase margin can be increased by adding a series
resistor.
Measurement Mode. A quasi-measurement mode can be implemented by installing J1 and
J2 (connecting an inverted VAPC to VSET) to yield the nominal relationship between RFIN and
VSET. In this mode, a large capacitor (0.01 µF or greater) must be installed in C4. J3 can be
installed to power the op-amp with the VPOS power supply. Alternately, the op-amp can be
powered with the VS supply pin.
Alternate Interface. R5 and R6 allow for VOUT and VSET to be accessible from the edge
connector, which is only used for characterization.
J1, J2, J3
R9, R10
Rev. 0 | Page 21 of 24
Default Condition
Not Applicable
R1 = 52.3 Ω (Size 0402)
R3 = 0 Ω (Size 0402)
R4 = C3 = open (Size 0402)
C1 = 0.1 µF (Size 0402)
C2 = 100 pF (Size 0402)
C4 = open (Size 0402)
R2 = 0 Ω (Size 0402)
J1, J2 = installed
J3 = installed
R9 = R10 = open
(Size 0402)
AD8311
OUTLINE DIMENSIONS
0.675
0.595
0.515
0.380
0.355
0.330
1.00
0.95
0.90
SEATING
PLANE
B
A
1
A1 BALL
CORNER
0.345
0.295
0.245
1.50
1.45
1.40
1.00
BSC
2
0.50
BSC
3
TOP VIEW
(BUMP SIDE DOWN)
0.270
0.240
0.210
0.075
COPLANARITY
0.50 BSC
BOTTOM VIEW
(BUMP SIDE UP)
Figure 46. 6-Ball Wafer Level Chip Scale Package [WLCSP]
(CB-6)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD8311ACBZ-P71
Temperature
Range
–40°C to +85°C
AD8311ACBZ-P21
–40°C to +85°C
AD8311-EVAL
1
Package Description
6-Ball Wafer Level Chip Scale Package [WLCSP],
7” Pocket Tape and Reel
6-Ball Wafer Level Chip Scale Package [WLCSP],
7” Pocket Tape and Reel
Evaluation Board
Z = Pb-free part.
Rev. 0 | Page 22 of 24
Package
Option
CB-6
Branding
Q04
Ordering
Quantity
3000
CB-6
Q04
250
AD8311
NOTES
Rev. 0 | Page 23 of 24
AD8311
NOTES
© 2005 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D05545-0-6/05(0)
Rev. 0 | Page 24 of 24