TI ADS5221RGZ 12-bit, 65msps sampling, 3.3v analog-to-digital converter Datasheet

ADS5221
ADS
522
1
SBAS262A – APRIL 2003 – REVISED DECEMBER 2003
12-Bit, 65MSPS Sampling, +3.3V
ANALOG-TO-DIGITAL CONVERTER
FEATURES
DESCRIPTION
●
●
●
●
●
The ADS5221 is a pipeline, CMOS Analog-to-Digital Converter (ADC) that operates from a single +3.3V power
supply. This converter provides excellent performance with a
single-ended input and can be operated with a differential
input for added spurious performance. This high-performance converter includes a 12-bit quantizer, high bandwidth
track-and-hold, and a high accuracy internal reference; it
also allows for the user to disable the internal reference and
utilize external references, providing excellent gain and offset matching in multi-channel applications or in applications
where full-scale range adjustment is required.
●
●
●
●
HIGH SNR: 70dB
HIGH SFDR: 88dBFS
LOW POWER: 285mW
INTERNAL/EXTERNAL REFERENCE OPTION
SINGLE-ENDED OR
FULLY DIFFERENTIAL ANALOG INPUT
FLEXIBLE DUTY CYCLE ADJUST CIRCUITRY
LOW DNL: 0.5LSB
SINGLE +3.3V SUPPLY OPERATION
TQFP-48
APPLICATIONS
● WIRELESS LOCAL LOOP
● COMMUNICATIONS
● MEDICAL IMAGING
● PORTABLE INSTRUMENTATION
The ADS5221 employs digital error correction techniques to
enable excellent differential linearity for demanding imaging
applications. Its low distortion and high SNR give the extra
margin needed for medical imaging, communications, video,
and test instrumentation. The ADS5221 offers power dissipation of 285mW and also provides two power-down modes.
The ADS5221 is specified at a maximum sampling frequency of 65MHz and a differential input range of 1V to 2V.
The ADS5221 is available in a TQFP-48.
AVDD
CLK
VDRV
ADS5221
Timing/Duty Cycle
Adjust
Circuitry
IN
VIN
S/H
IN
12-Bit
Pipelined
ADC
Error
Correction
Logic
3-State
Output
D0
•
•
•
D11
OVR
Internal
Reference
STPD QPD
REFT REFB
RSEL
VREF
OE
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright © 2003, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
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ABSOLUTE MAXIMUM RATINGS(1)
AVDD, DVDD, VDRV .......................................................................... +3.8V
Analog Input ........................................................... –0.3V to (AVDD + 0.3V)
Logic Input ............................................................. –0.3V to (AVDD + 0.3V)
Case Temperature ......................................................................... +100°C
Junction Temperature .................................................................... +150°C
Storage Temperature ..................................................................... +150°C
NOTE: (1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods may degrade
device reliability.
ELECTROSTATIC
DISCHARGE SENSITIVITY
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.
ESD damage can range from subtle performance degradation
to complete device failure. Precision integrated circuits may be
more susceptible to damage because very small parametric
changes could cause the device not to meet its published
specifications.
PACKAGE/ORDERING INFORMATION(1)
PACKAGE-LEAD
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
ADS5221
"
TQFP-48
"
PFB
"
–40°C to +85°C
"
ADS5221PFB
"
ADS5221PFBT
ADS5221PFBR
Tape and Reel, 250
Tape and Reel, 2000
ADS5221
"
QFN-48(2)
"
RGZ
"
–40°C to +85°C
"
ADS5221RGZ
"
ADS5221RGZ
ADS5221RGZR
Rails, 52
Tape and Reel, 1000
PRODUCT
NOTE: (1) For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet.
(2) This package available beginning Q1 2004.
ELECTRICAL CHARACTERISTICS: AVDD = 3.3V
TMIN = –40°C, TMAX = +85°C, typical values are at TA = +25°C, sampling rate = 65MSPS, 50% clock duty cycle, AVDD = 3.3V, VDRV = 2.5V, –1dBFS, internal reference
voltage, and 2VPP differential input, unless otherwise noted.
ADS5221
PARAMETER
CONDITIONS
MIN
RESOLUTION
SPECIFIED TEMPERATURE RANGE
ANALOG INPUT
Single-Ended Input Range
Optional Single-Ended Input Range
Differential Input Range
Analog Input Bias Current
Input Impedance
Analog Input Bandwidth
Ambient Air
2VPP
1VPP
2VPP
DYNAMIC CHARACTERISTICS
Differential Linearity Error (largest code error)
f = 2.4MHz
f = 9.7MHz
No Missing Codes
Integral Nonlinearity Error, f = 2.4MHz
Spurious-Free Dynamic Range(1)
f = 2.4MHz
f = 9.7MHz
f = 32.5MHz
2-Tone Intermodulation Distortion(3)
f = 9.5MHz and 10.5MHz (–7dB each tone)
Signal-to-Noise Ratio (SNR)
f = 2.4MHz
f = 9.7MHz
f = 32.5MHz
Signal-to-(Noise + Distortion) (SINAD)
f = 2.4MHz
f = 9.7MHz
f = 32.5MHz
Effective Number of Bits(4), f = 2.4MHz
Output Noise
Aperture Delay Time
Aperture Jitter
Over-Voltage Recovery Time
Full-Scale Step Acquisition Time
2
MAX
UNITS
12 Tested
Bits
–40 to +85
°C
0.5
1
1
2.5
2
2
V
V
V
µA
MΩ || pF
MHz
65M
Samples/s
Clock Cycle
±0.75
LSB
LSB
±1.75
LSBs
1
1.25 || 5
300
Static, No Clock
–3dBFS Input
CONVERSION CHARACTERISTICS
Sample Rate
Data Latency
TYP
1M
5
±0.3
±0.5
Tested
±0.5
Referred to Full-Scale
at 25°C
78
90
88
85
dBFS(2)
dBFS
dBFS
92
dBFS
70
70
69
dB
dB
dB
70
70
68
11.2
0.2
3.0
1.2
1.0
5
dB
dB
dB
Bits
LSBs rms
ns
ps rms
Clock Cycle
ns
Referred to Full-Scale
at 25°C
68
Referred to Full-Scale
at 25°C
Input Tied to Common-Mode
67
ADS5221
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SBAS262A
ELECTRICAL CHARACTERISTICS: AVDD = 3.3V (Cont.)
TMIN = –40°C, TMAX = +85°C, typical values are at TA = +25°C, sampling rate = 65MSPS, 50% clock duty cycle, AVDD = 3.3V, VDRV = 2.5V, –1dBFS, internal reference
voltage, and 2VPP differential input, unless otherwise noted.
ADS5221
PARAMETER
DIGITAL INPUTS
Logic Family
Convert Command
High Level Input Current(5) (VIN = 3VDD)
Low Level Input Current (VIN = 0V)
High Level Input Voltage
Low Level Input Voltage
Input Capacitance
DIGITAL OUTPUTS
Logic Family
Logic Coding
Low Output Voltage (IOL = 50µA to 1.5mA)
High Output Voltage (IOH = 50µA to 0.5mA)
3-State Enable Time
3-State Disable Time
Output Capacitance
ACCURACY (Internal Reference, 2VPP,
Unless Otherwise Noted)
Zero Error (referred to midscale)
Zero Error Drift (referred to midscale)
Gain Error(6)
Gain Error Drift(6)
Power-Supply Rejection of Gain
CONDITIONS
MIN
MAX
CMOS-Compatible
Rising Edge of Convert Clock
100
10
Start Conversion
+1.7
+0.7
5
CMOS-Compatible
Straight Offset Binary or BTC
+0.1
VDRV = 2.5V
+2.4
UNITS
µA
µA
V
V
pF
V
V
ns
ns
pF
OE = H
OE = L
20
2
5
40
10
fIN = 2.4MHz, at 25°C
±0.5
5
±0.4
38
56
±1.5
%FS
ppm/°C
%FS
ppm/°C
dB
±5
0.8
±2.5
0.1
±35
mV
mV
mV
mV
+3.3
+2.5
86
285
290
15
70
+3.6
+3.6
V
V
mA
mW
mW
mW
mW
fIN = 9.7MHz, at 25°C
∆AVDD = ±5%
INTERNAL VOLTAGE REFERENCE
Output Voltage Error (1V)
Load Regulation at 1.0mA
Output Voltage Error (0.5V)
Load Regulation at 0.5mA
POWER-SUPPLY REQUIREMENTS
Supply Voltage: AVDD, DVDD
Driver Supply Voltage: VDRV
Supply Current: +IS
Power Dissipation: VDRV = 2.5V
VDRV = 3.3V
Standard Power-Down
Quasi Power-Down
Thermal Resistance, θJA
TQFP-48
QFN-48
TYP
Operating
Operating (External Reference)
+3.0
+2.3
63.7
26.1
295
°C/W
°C/W
NOTES: (1) Spurious-Free Dynamic Range refers to the magnitude of the largest harmonic.
(2) dBFS means dB relative to Full-Scale.
(3) 2-tone intermodulation distortion is referred to the largest fundamental tone. This number will be 6dB higher if it is referred to the magnitude of the
2-tone fundamental envelope.
(4) Effective Number of Bits (ENOB) is defined by (SINAD – 1.76) /6.02.
(5) A 50kΩ pull-down resistor is inserted internally on the OE pin.
(6) Includes internal reference.
ADS5221
SBAS262A
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3
DVDD
DVDD
CLK
DGND
DGND
AGND
IN
IN
AGND
AGND
AGND
AVDD
PIN CONFIGURATION
48
47
46
45
44
43
42
41
40
39
38
37
MSBI
1
36 AVDD
OE
2
35 NC
Mode Select
3
34 REFT
STPD
4
33 NC
QPD
5
32 REFB
GDRV
6
GDRV
7
30 VREF
VDRV
8
29 AGND
VDRV
9
28 AGND
D11 (MSB) 10
27 AGND
31 RSEL
ADS5221
D4
D3
19
20
21
22
23
24
OVR
18
NC
17
NC
16
D0 (LSB)
15
D1
14
D2
13
D5
25 NC
D6
D9 12
D7
26 NC
D8
D10 11
PIN ASSIGNMENTS
PIN
I/O
1
4
DESCRIPTION
MSBI
Most Significant Bit Invert (HI = Binary Two’s
Complement, LO = Straight Offset Binary)
Tri-State (LO = Enabled, HI = Tri-State)
Duty Cycle Adjust (HI = Enabled,
LO = Normal Operation)
Standard Power-Down (LO = Normal
Operation, HI = Enabled)
Quasi Power-Down (LO = Normal Operation,
HI = Enabled)
Output Driver Ground
Output Driver Ground
Output Driver Supply
Output Driver Supply
Data Bit 12
Data Bit 11
Data Bit 10
Data Bit 9
Data Bit 8
Data Bit 7
Data Bit 6
Data Bit 5
Data Bit 4
Data Bit 3
Data Bit 2
Data Bit 1
No Internal Connection
No Internal Connection
Over-Range Indicator
2
3
OE
Mode Select
4
STPD
5
QPD
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
GDRV
GDRV
VDRV
VDRV
D11 (MSB)
D10
D9
D8
D7
D6
D5
D4
D3
D2
D1
D0 (LSB)
NC
NC
OVR
O
O
O
O
O
O
O
O
O
O
O
O
PIN
NAME
I/O
NAME
DESCRIPTION
25
NC
No Internal Connection
26
27
NC
AGND
No Internal Connection
Analog Ground
28
29
AGND
AGND
Analog Ground
Analog Ground
30
VREF
Internal Reference Voltage
(1/2V Reference)
31
RSEL
32
REFB
Reference Mode Select (see Table I for
settings)
Bottom Reference Bypass
33
34
NC
REFT
No Internal Connection
Top Reference Bypass
35
36
NC
AVDD
No Internal Connection
Analog Supply
37
38
AVDD
AGND
Analog Supply
Analog Ground
39
40
AGND
AGND
Analog Ground
Analog Ground
41
42
I
I
43
44
IN
IN
Analog Input
Complementary Analog Input
AGND
DGND
Analog Ground
Digital Ground
45
46
I
DGND
CLK
Digital Ground
Convert Clock Input
47
48
I
I
DVDD
DVDD
Digital Supply
Digital Supply
ADS5221
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SBAS262A
TIMING DIAGRAM
N+2
N+1
Analog In
N+3
N
tD
N+4
N+5
tL
tCONV
N+6
N+7
tH
Clock
5 Clock Cycles
t2
Data Out
N–5
N–4
N–3
N–2
N–1
Data Invalid
SYMBOL
tCONV
tL
tH
tD
t1
t2
N+1
N+2
t1
DESCRIPTION
MIN
Convert Clock Period
Clock Pulse LOW
Clock Pulse HIGH
Aperture Delay
Data Hold Time, CL = 0pF
Propagation Delay Time, CL = 5pF max
15.4ns
6.7
6.7
ADS5221
SBAS262A
N
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TYP
MAX
UNITS
100µs
7.7
7.7
3
6.0
6.5
ns
ns
ns
ns
ns
5
TYPICAL CHARACTERISTICS: AVDD = 3.3V
TMIN = –40°C, TMAX = +85°C, typical values are at TA = +25°C, sampling rate = 65MSPS, 50% clock duty cycle, AVDD = 3.3V, VDRV = 2.5V, –1dBFS, internal reference
voltage, and 2VPP differential input, unless otherwise noted.
SPECTRAL PERFORMANCE
(Differential, 2VPP)
SPECTRAL PERFORMANCE
(Differential, 2VPP)
0
0
fIN = 2.4MHz (–1dBFS)
SFDR = 90.2dBFS
SNR = 70.2dBFS
SINAD = 70.1dBFS
–40
–60
–80
–100
–40
–60
–80
–100
–120
–120
0
6.5
13
26
32.5
0
6.5
SPECTRAL PERFORMANCE
(Differential, 2VPP, External Reference)
SPECTRAL PERFORMANCE
(Differential, 1VPP)
0
fIN = 9.7MHz (–1dBFS)
SFDR = 89.3dBFS
SNR = 69.3dBFS
SINAD = 69.2dBFS
fIN = 2.4MHz (–1dBFS)
SFDR = 89.3dBFS
SNR = 66.9dBFS
SINAD = 66.9dBFS
–20
Amplitude (dB)
–40
–60
–80
–100
–40
–60
–80
–100
–120
–120
0
6.5
13
26
32.5
0
6.5
Frequency (MHz)
13
26
32.5
Frequency (MHz)
SPECTRAL PERFORMANCE
(Single-Ended, 2VPP)
TWO TONE INTERMODULATION DISTORTION
0
0
fIN = 2.4MHz (–1dBFS)
SFDR = 84.0dBFS
SNR = 69.6dBFS
SINAD = 69.3dBFS
–20
fIN = 9.5MHz (–7dBFS)
fIN = 10.5MHz (–7dBFS)
SFDR = 92.5dBFS
–20
–40
Amplitude (dB)
Amplitude (dB)
32.5
26
Frequency (MHz)
–20
–60
–80
–40
–60
–80
–100
–100
–120
–120
0
6.5
13
26
32.5
0
6.5
13
26
32.5
Frequency (MHz)
Frequency (MHz)
6
13
Frequency (MHz)
0
Amplitude (dB)
fIN = 9.7MHz (–1dBFS)
SFDR = 90.1dBFS
SNR = 70.1dBFS
SINAD = 69.9dBFS
–20
Amplitude (dB)
Amplitude (dB)
–20
ADS5221
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SBAS262A
TYPICAL CHARACTERISTICS: AVDD = 3.3V (Cont.)
TMIN = –40°C, TMAX = +85°C, typical values are at TA = +25°C, sampling rate = 65MSPS, 50% clock duty cycle, AVDD = 3.3V, VDRV = 2.5V, –1dBFS, internal reference
voltage, and 2VPP differential input, unless otherwise noted.
DIFFERENTIAL LINEARITY ERROR
EXTERNAL REFERENCE
TWO-TONE INTERMODULATION DISTORTION
0
1.0
fIN = 45MHz (–7dBFS)
fIN = 46MHz (–7dBFS)
SFDR = 86.7dBFS
0.6
0.4
–40
DLE (LSB)
Amplitude (dB)
–20
fIN = 9.7MHz
0.8
–60
–80
0.2
0
–0.2
–0.4
–0.6
–100
–0.8
–120
–1.0
0
6.5
13
32.5
26
0
1024
2048
Frequency (MHz)
DIFFERENTIAL LINEARITY ERROR
1.0
fIN = 9.7MHz
0.8
0.6
0.6
0.4
0.4
0.2
0.2
ILE (LSB)
DLE (LSB)
INTEGRAL LINEARITY ERROR
1.0
fIN = 9.7MHz
0.8
0
–0.2
0
–0.2
–0.4
–0.4
–0.6
–0.6
–0.8
–0.8
–1.0
–1.0
0
1024
2048
3072
4096
0
1024
2048
Code
1.0
SWEPT POWER (SNR)
80
fIN = 9.7MHz
0.8
4096
3072
Code
INTEGRAL LINEARITY ERROR
EXTERNAL REFERENCE
dBFS
70
0.6
fIN = 9.7MHz
60
SNR (dBc, dBFS)
0.4
ILE (LSB)
4096
3072
Code
0.2
0
–0.2
–0.4
50
dBc
40
30
20
–0.6
10
–0.8
0
–1.0
0
1024
2048
3072
4096
ADS5221
SBAS262A
–70
–60
–50
–40
–30
–20
–10
0
Analog Input Level (dBFS)
Code
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7
TYPICAL CHARACTERISTICS: AVDD = 3.3V (Cont.)
TMIN = –40°C, TMAX = +85°C, typical values are at TA = +25°C, sampling rate = 65MSPS, 50% clock duty cycle, AVDD = 3.3V, VDRV = 2.5V, –1dBFS, internal reference
voltage, and 2VPP differential input, unless otherwise noted.
SWEPT POWER (SNR)
(External Reference)
SWEPT POWER (SFDR)
100
80
fIN = 9.7MHz
dBFS
90
fIN = 9.7MHz
dBFS
70
60
70
SNR (dBc, dBFS)
SFDR (dBc, dBFS)
80
60
50
40
dBc
30
dBc
50
40
30
20
20
10
10
0
0
–80
–70
–60
–50
–40
–30
–20
–10
0
–70
–60
Analog Input Level (dBFS)
–20
–10
0
DYNAMIC PERFORMANCE vs INPUT FREQUENCY
90
dBc
80
SFDR
85
70
SFDR, SNR (dBFS)
SFDR (dBc, dBFS)
–30
95
fIN = 9.7MHz
90
–40
Analog Input Level (dBFS)
SWEPT POWER (SFDR)
(External Reference)
100
–50
dBFS
60
50
40
30
80
75
65
60
20
55
10
50
0
SNR
70
45
–80
–70
–60
–50
–40
–30
–20
–10
0
1
10
Analog Input Level (dBFS)
100
Frequency (MHz)
POWER DISSIPATION vs INPUT FREQUENCY
320
305
Power (mW)
290
275
260
245
230
215
200
1
10
100
Frequency (MHz)
8
ADS5221
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SBAS262A
ANALOG INPUT
APPLICATION INFORMATION
THEORY OF OPERATION
The ADS5221 is a 12-bit, 65MSPS, CMOS ADC designed
with a fully differential pipeline architecture. The pipeline
consists of three sections: a 3-bit quantizer, eight middle
stages with a 1.5-bit quantizer for each stage, and a 4-bit
flash. The output of each pipeline stage is processed and
formed into 12-bit data in the digital error correction logic
section to ensure good differential linearity of the ADC. The
converter includes a high bandwidth track-and-hold amplifier
in the input stage as shown in Figure 1. It provides excellent
performance for a single-ended input or differential input of IF
(Intermediate Frequency) or an under-sampling signal. The
rising edge of the input clock initiates the conversion process. Once the signal is captured by the input track-and-hold,
the bits are sequentially encoded starting with the Most
Significant Bit (MSB). This process results in a data latency
of 5 clock cycles. The ADS5221 includes a high accuracy
internal reference and also allows to the use of an external
reference. The input full-scale range is up to +2V and is
selectable based on the reference voltage setting. For normal operation, both analog inputs (IN, IN) require external
common-mode voltage as signal swing center. The output
data of ADS5221 is available as a 12-bit parallel word either
coded in a Straight Offset Binary or Binary Two’s Complement format. The ADS5221 includes an option of a duty cycle
stabilizer (DCS) that allows ADS5221 to operate with a nonsquare wave input clock, such as from a 30% to 70% duty
cycle. When the DCS is selected the ADS5221 maintains an
internal clock duty cycle at 50% with ±5% tolerance. The
ADS5221 has low power dissipation in normal mode and has
two power down modes for energy saving. The device
operates from a single +3.3V power supply and has a
separate digital output driver supply pin.
Depending on the application and the desired level of performance , the analog input of the ADS5221 can be configured
in various ways and driven with different circuits. The ADS5221
is particularly well-suited for communication systems that
digitize large scale and wideband signals. In any case, the
analog interface requirements should be carefully examined
before selecting the appropriate circuit configuration. The
circuit definition should include considerations on the input
frequency band and amplitude, as well as the available
power supplies.
INPUT IMPEDANCE
The input impedance of the ADS5221 is capacitive due to the
input stray and sampling capacitors. These capacitors effectively result in a dynamic input impedance that is a function
of the sampling and input frequency. Figure 2 depicts the
differential input impedance of the ADS5221 as a function of
the signal input frequency. At certain sampling rates, increasing the signal frequency will decrease the input impedance.
This factor needs to be considered when the signal source is
designed. For applications that use op amps to drive the
ADC, it is recommended that a series resistor be added
between the amplifier output and the converter inputs. This
will isolate the capacitive input of the converter from the
driving source and avoid gain peaking, or instability; furthermore, it will create a 1st-order, low-pass filter (LPF) in
conjunction with the specified input capacitance of the
ADS5221. The cutoff frequency of this LPF can be further
adjusted by adding an external shunt capacitor from each
signal input to ground. In any case, the use of the RC
network is optional, but optimizing the values to adapt to the
specific application is encouraged.
DIFFERENTIAL INPUT IMPEDANCE
50
S5
40
S3
VBIAS
CIN
S2
CIN
30
Ohms
S1
IN
T&H
20
IN
S4
S6
10
VBIAS
0
1
Tracking Phase: S1, S2, S3, S4 closed; S5, S6 open
Hold Phase:
10
100
Input Frequency (MHz)
S1, S2, S3, S4 open; S5, S6 closed
FIGURE 2. Differential Input Impedance vs Input Frequency.
FIGURE 1. Simplified Circuit of Input Track-and-Hold Amplifier
of ADS5221.
ADS5221
SBAS262A
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9
INPUT COMMON-MODE VOLTAGE
ANALOG INPUT DRIVEN BY TRANSFORMER
The ADS5221 operates from a single +3.3V supply, and
requires a external common-mode voltage. This allows a
symmetrical signal swing while maintaining sufficient headroom to the supply rail. The common-mode voltage can be
from an external DC voltage source, for example, an analog
+3.3V supply with a simple resister divider, or from the input
signal source with DC-coupling. When it is a single-ended
input configuration, the common-mode voltage is typically
+1.25V. When the input configuration is differential, the
common-mode voltage is +1.5V.
The ADS5221 can be driven by a transformer, which provides signal AC-coupling and allows a signal conversion from
single-ended input to differential output, or from single-ended
input to single-ended output. Using a transformer offers a
number of advantages. As a passive component, it does not
add to the total noise and has better harmonics in wide
frequency bands, compared to an op amp driver. By using a
step-up transformer, further signal amplification can be realized; as a result, the signal swing from the source can be
reduced. For transformer selection, it is important to carefully
examine the application requirements and determine the
correct model, the desired impedance ratio, and frequency
characteristics. Furthermore, the appropriate model must
support the targeted distortion level and should not exhibit
any core saturation at full-scale voltage levels. A variety of
miniature RF transformers from different manufacturers, (such
as, Mini-Circuits, Coilcraft, or Trak) can be selected.
INPUT FULL SCALE RANGE
The input full-scale range (FSR) of the ADS5221 is selectable from +1V to +2V and any value within this range, by
setting the reference select pin RSEL and reference voltage
pin VREF (see Table I). The input FSR (differential) is always
twice VREF (the voltage at the VREF pin) for all reference
modes.
By choosing different signal input ranges, trade-offs can be
made between noise and distortion performance. For maximizing SNR, the maximum signal input 2VPP range may be
selected. This range may also be selected for low-level
(–6dBFS to –40dBFS) but high-frequency multi-tone input.
The signal distortion at the output of the device could be
sensitive to the input signal with large full-scale. Reducing
signal amplitude will improve the distortion performance. The
large input signal amplitude of the converter will impose
additional design constraints to the op amp due to its headroom requirements when the op amp is from single-supply
and DC-coupling.
Figure 3 shows a transformer-coupled input configuration of
the ADS5221. The ADS5221 receives a differential AC signal
from the output of the transformer and common-mode voltage of +1.5V from the center tap. A source termination
resistor, RT , is required, which can be placed at the input or
output of the transformer to satisfy the termination requirements of the source impedance, RS. The circuit also shows
the use of an additional RC low-pass filter placed in series
with each converter input to attenuate some of the wideband
noise. The resistor values are typically in the range of 10Ω to
50Ω, and capacitors are in the range of 10pF to 100pF for
individual application requirements.
DIFFERENTIAL INPUTS
The ADS5221 input structure is designed to accept both a
single-end or differential analog signal. Distortion performance, however, can be improved by utilizing the differential
input configuration. Differential operation of the ADS5221
requires that an input signal at the inputs (IN, IN) has the
same amplitude and is 180 degrees out-of-phase. Differential signals offer a number of advantages:
T4-6T
24.9Ω
IN
RS
24.9Ω
IN
RT
0.1µF
• The signal amplitude is half that required for the singleended operation, and is therefore less demanding to achieve,
while maintaining good linearity performance from the
signal source.
1.7kΩ
3.3V
ADS5221
22pF
AVDD
1.5V
1.5kΩ
• The reduced signal swing allows for more headroom of the
interface circuitry, and therefore also allows a wider selection of the best suitable driver amplifier.
• Minimization of even-order harmonics.
• Improved noise immunity based on the common-mode
input rejection of the converter.
10
FIGURE 3. Transformer-Coupled Differential Input Configuration of ADS5221.
ADS5221
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SBAS262A
ANALOG INPUT DRIVEN BY OP AMPLIFIER
The ADS5221 can be driven by an operational amplifier with
DC or AC signal coupling, as shown in Figure 4 and Figure
5. In Figure 4, the THS4503, a differential amplifier, is used
to convert a single-ended input to differential output with a
gain of 2. The THS4503 provides an output common-mode
voltage set by VOCM pin, and is DC-coupled to the input of
ADS5221. A low-pass filter can be created by adding small
capacitors (for example, 10pF) in parallel with the feedback
resistors of the THS4503 as needed for some applications.
Due to the THS4503 driving a capacitive load, small series
resistors in the output ensure stable operation. Further details of this and other functions of the THS4503 may be found
in its product datasheet, located on the Texas Instruments
web site (www.ti.com). In general, differential amplifiers provide a higher performance driver solution as compared to a
single-ended amplifier.
+5V
As shown in Figure 5, an AC-coupled, single-ended input
configuration is realized with TI’s OPA695 for wideband
applications. For narrowband applications, the OPA2822 can
be used. In Figure 5, the OPA695 is configured at single
supply +5V and noninverting operation. The AC gain of the
amplifier is 2 and the DC offset of the amplifier is +2.5V, set
by the voltage divider from the op amp power supply. The
output of the amplifier can provide maximum full-scale voltage range for the ADS5221. The OPA695 is a very high
bandwidth, current-feedback op amp that combines 4200V/
µs slew rate and low input voltage noise. It is optimized for
high gain operation. Further details of the OPA695 can be
found in the OPA695 data sheet. The common-mode voltage
at the ADS5221 input is +1.25V, set by a voltage divider from
+3.3V power supply. The +3.3V power supply must be
decoupled, as shown in Figure 11.
10pF(1)
392Ω
187Ω
24.9Ω
IN
50Ω
VCM
60.4Ω
THS4503
ADS5221
22pF
24.9Ω
IN
0.1µF
392Ω
215Ω
10pF(1)
AVDD
–5V
1.7kΩ
1.5V
3.3V
1.5kΩ
FIGURE 4. Using the THS4503 Differential Amplifier (Gain = 2) to Drive the ADS5221 in a DC-Coupled Configuration.
+5V
0.1µF
6.8µF
1kΩ
1.6kΩ
806Ω
3.3V
+5V
0.1µF
0.1µF
50Ω
57.6Ω
806Ω
30Ω
IN
OPA695
47pF
0.1µF
487Ω
ADS5221
487Ω
IN
0.1µF
47pF
AVDD
1.6kΩ
3.3V
1.25V
1kΩ
FIGURE 5. Single-Ended Input of ADS5221 Driven by OPA695 with Gain = 2.
ADS5221
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11
CLOCK INPUT
The clock input of the ADS5221 is designed to receive a
single-ended pulse clock with CMOS/TTL level and DCcoupling. There is no external common-mode voltage requirement at the clock input pin (see Figure 6). The typical
tolerance of the internal clock duty cycle is ±5% at its nominal
value of 50% in order to maintain the high performance of the
device.
The ADS5221 contains an optional clock duty cycle adjust
(DCA). When the DCA is enabled (Mode Select = high), a
wide range of input clock duty cycle can be adjusted internally to 50% (±5% tolerance) by correcting the non-sampling
edge of the clock. In this case, the noise and distortion
performance will not be affected by the input clock duty cycle
within this range. When the DCA is disabled (Mode Select =
low), a clock duty cycle variation greater than ±5% will
degrade the dynamic performance of the device. In this case,
an input clock duty cycle less than 45% will automatically
request a reduction in the sampling rate in order to maintain
the track/hold period within the specified values for stable
performance. The DCA disabled mode is suitable for nonuniform sampling ADC applications.
CMOS/TTL
CLK
Clock Source
50Ω
50Ω
ADS5221
FIGURE 6. General Input Clock Interface of ADS5221.
In any case, a very low jitter clock is fundamental to preserving the excellent AC performance of the ADS5221. The
device itself is specified for a low jitter, characterizing the
outstanding capability of the internal clock and track-andhold circuitry. Generally, as input frequency increases, clock
jitter becomes more critical to maintain a good signal-tonoise ratio. The following equation can be used to calculate
the achievable SNR for a given input frequency and clock
jitter (tJA in ps rms):
Here, the tJA is the rms aperture jitter from all jitter sources,
such as clock edge, input signal and the device. The fIN is
input frequency. The crystal oscillator has very low jitter, but
if using a clock conditioning circuit (gate, divider, logic lever
converter, and so forth), an extra jitter and timing variation
must be limited. In addition, the input clock is treated as an
analog signal and its power supply should be separated from
the power supply of the digital output driver to limit the digital
noise.
MINIMUM SAMPLING RATE
The pipeline architecture of the ADS5221 uses a switchedcapacitor technique in its internal track-and-hold stages. The
high sampling speed necessitates the use of very small
capacitor values. In order to hold the droop errors low, the
capacitors require a minimum refresh rate. To maintain
accuracy of the acquired sample charge, the sampling clock
on the ADS5221 must not drop below the specified minimum
of 1MHz.
REFERENCE
ADS5221 provides both internal reference and external reference modes by setting pins RSEL and VREF (see Table 1).
The input full-scale range (FSR) of ADS5221 is always twice
the voltage VREF at the VREF pin. The REFT and REFB are
internally buffered, and drive the ADC core for both the
external reference and internal reference modes. The REFT
and REFB are designed for external by pass only. The output
resistance between the REFT and REFB pins is approximately 1Ω. When the internal reference mode is selected the
voltage at VREF is generated by an internal 0.5V bandgap
voltage through a VREF amplifier, and the VREF output can
supply 2.5mA source current. When the external reference is
selected, the internal VREF amplifier is powered down, the
external reference voltage is added at the VREF pin, and the
voltage is input to the internal REFT/REFB amplifier. The
voltage at pins REFT, REFB and VREF, the full scale range
(FSR) voltage at analog input for both external and internal
reference modes, and the differential/single-ended input
configurations are as follows:
VREFT = VREF/2 +1.5V
VREFB = 1.5V – VREF/2
SNR = 20 log [1/(2 • π • fIN • tJA)]
VREF = VREFT – VREFB
FSR (Differential) = 2 • VREF
SELECTED MODE
RSEL PIN
CONNECT TO
VREF PIN (V)
INPUT FSR (V)
(Differential)
REFT (V)
REFB (V)
Internal Fixed
Internal Fixed
Internal Program
External
GND to 0.2V
VREF Pin
0.2V to VREF
AVDD (3.3V)
1.0
0.5
0.5 • (1+R2/R1)
Ext. 0.5V to 1V
2
1
2 • VREF
2 • VREF
2
1.75
VREF/2 + 1.5
VREF/2 + 1.5
1
1.25
1.5 – VREF/2
1.5 – VREF/2
TABLE I. Reference Configuration.
12
ADS5221
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SBAS262A
The ADS5221 requires solid bypassing for all reference pins
to keep the effects of clock feedthrough to a minimum and to
achieve the specified levels of performance. Figure 7 to
Figure 10 show the recommended decoupling scheme. All
0.1µF capacitors must be located as close to the pins as
possible. In addition, pins REFT, VREF, and REFB must be
decoupled with tantalum surface-mount capacitors (2.2µF,
4.7µF, 10µF or higher).
RSEL
R2
VREF
0.1µF
15µF
3.3V
R1
ADS5221
402Ω
2Ω
REFT
INTERNAL REFERENCE
10µF
There are two internal fixed reference modes and one
internal programmable reference mode as shown in Table I
and Figure 7 through Figure 9. Setting RSEL to ground (or
< 0.2V) provides an internal reference voltage of +1.0V at
VREF pin, +2V at REFT, and +1V at REFB pin. In this case,
the input FSR is +2V peak-to-peak. Connecting RSEL to the
VREF pin provides an internal reference voltage of +0.5V at
VREF, +1.75V at REFT, and +1.25V at REFB. In this case, the
input FSR is +1V peak-to-peak. Setting the resister divider as
in Figure 9 provides an internal voltage between +0.5V and
+1V at VREF, which is as follows:
VREF = 0.5 • (1+R2/R1)
In this case, the voltage at REFT and REFB and input FSR
is calculated based on Table I.
RSEL
VREF
0.1µF
ADS5221
2.2µF
1V
Output
3.3V
402Ω
2Ω
REFT
10µF
2.2µF
15µF
2.2µF
15µF
2Ω
REFB
2.2µF
15µF
2.2µF
15µF
2Ω
REFB
402Ω
FIGURE 9. Internal Reference Mode for VREF = 0.5 • (1 + R2/R1).
EXTERNAL REFERENCE
For even more design flexibility, the ADS5221 can be operated with external references. Utilization of an external reference voltage may be considered for applications requiring
higher accuracy, improved temperature stability, or flexible
full-scale range. Particularly in multi-channel applications,
the use of a common external reference offers the benefit of
improving gain matching between converters. Setting RSEL
to AVDD (+3.3V) provides an external reference mode for the
ADS5221. In this case, the internal VREF amplifier is powered
down, and the VREF pin requires an external reference
voltage between +0.5V to +1V to provide an input FSR
voltage of +1V to +2V. The REFT and REFB will appear with
the voltage as shown in Table I and input FSR is always
twice the voltage at the VREF pin. A voltage reference
(REF1004 or TPS79225) and a single-supply amplifier
(OPA2234 or OPA4227) can be used to generate a precision
external reference.
402Ω
AVDD
FIGURE 7. Internal Reference Mode for VREF = 1V.
RSEL
RSEL
VREF
0.1µF
15µF
Input
0.5V to 1V
VREF
0.5V
Output
0.1µF
3.3V
3.3V
10µF
ADS5221
ADS5221
402Ω
402Ω
2Ω
2Ω
REFT
REFT
10µF
2.2µF
10µF
15µF
2.2µF
15µF
2.2µF
15µF
2Ω
2Ω
REFB
REFB
2.2µF
15µF
402Ω
402Ω
FIGURE 10. External Reference Configuration.
FIGURE 8. Internal Reference Mode for VREF = 0.5V.
ADS5221
SBAS262A
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13
DIGITAL OUTPUTS
DATA OUTPUT FORMAT
The ADS5221 makes two data output formats available,
either the Straight Offset Binary (SOB) code or the Binary
Two’s Complement (BTC) code. The selection of the output
coding is controlled through the MSBI pin. Applying a logic
high will enable the BTC coding, whereas a logic low will
enable the SOB code. The two code structures are identical
with one exception: the MSB is inverted for the BTC format,
as shown in Table II. If the input signal exceeds the FSR, the
output code will remain at all 1s or all 0s.
STRAIGHT OFFSET
BINARY (SOB)
BINARY TWO’S
COMPLEMENT
(BTC)
+FS – 1LSB
(+FS: VIP = 2V, VIN = 1V)
1111 1111 1111
0111 1111 1111
+1/2 FS
(VIP = 1.75V, VIN = 1.25V)
1100 0000 0000
0100 0000 0000
Bipolar Zero
(VIP = VIN = 1.5V)
1000 0000 0000
0000 0000 0000
–1/2 FS
(VIN = 1.75V, VIP = 1.25V)
0100 0000 0000
1100 0000 0000
–FS
(VIN = 2V, VIP = 1V)
0000 0000 0000
1000 0000 0000
DIFFERENTIAL INPUT
the input voltage is (+FS – 1 LSB) or above this value, the
ADS5221 outputs all 1s at 12 data bits; when the input
voltage is –FS or below this value, the ADS5221 outputs all
0s at 12 data bits. When the input voltage is 0 (middle scale)
or only the common-mode voltage at the input, the ADS5221
outputs 1 at MSB and 0s at the remaining 11 data bits.
Another over-range control function of the ADS5221 is overrange indication, which is output by the OVR pin. The OVR
pin is the function of the reference voltage and the output
data bits, and has the same pipeline delay as the output data
bits. OVR is at logic low if the input voltage is within the FSR,
and is at logic high if the input voltage is over full-scale or
under full-scale. OVR changes from logic low to high or logic
high to low immediately following the change of the output
data, when the input voltage changes from normal value to
over FS or from over FS to normal value. When the input
signal continues under full-scale or over full-scale, OVR
stays high.
TIMING
OUTPUT ENABLE (OE )
The ADS5221 samples the analog signal at the rising edge
of its input clock, and outputs the digital data at the rising
edge of the input clock after a pipeline delay of 5 clocks.
There is an aperture delay (typically 3ns) between the
sampling edge and the actual sampling time. There is also a
propagation delay between the rising edge of the clock and
the time that data is valid on the data bus (see the timing
diagram on page 5). The output data of the ADS5221 is
latched data.
The digital outputs of the ADS5221 can be set to output
enable or output high impedance (tri-state) by the OE pin.
For normal operation, this pin must be at a logic low, whereas
a logic high disables the outputs or sets the output tri-state.
POWER SUPPLIES AND
POWER DISSIPATION
TABLE II. Coding Table for Differential Input Configuration
with FSR of 2V.
ANALOG AND DIGITAL POWER SUPPLY
OUTPUT LOADING
It is recommended to keep the capacitive loading on the data
output lines as low as possible, preferably below 5pF. Higher
capacitive loading will cause larger dynamic currents as the
digital outputs are changing. These high current surges can
feed back to the analog portion of the ADC and adversely
affect device performance. If necessary, external buffers or
latches (for example, the SN74LVTH16374) close to the
converter output pins can be used to minimize capacitive
loading. Buffers or latches also provide the added benefit of
isolating the ADS5221 from any digital activities on the bus
to limit the high-frequency noise.
OVER-RANGE INDICATOR
The ADS5221 has control functions for the input voltage over
full-scale that includes output data code control and overrange indication. The output data code control of over fullscale is shown in Table II. In SOB format, for example, when
14
The ADS5221 includes power-supply pins of AVDD, DVDD
and VDRV. The analog supply AVDD is +3.3V and digital
supply DVDD is +3.3V. A digital output driver supply, VDRV,
can be +2.5V up to +3.3V. AVDD, DVDD and VDRV are not
tied together internally. Each of these supply pins must be
bypassed separately with at least one 0.1µF ceramic chip
capacitor. The analog supply (AVDD) and the digital supply
(DVDD or VDRV) may be tied together externally with a ferrite
bead or inductor between the supply pins. The digital output
driver supply, VDRV, of +2.5V is used commonly. It is highly
recommended to consider linear supplies instead of switching types. Even with good filtering, switching supplies can
radiate noise that could interfere with any high-frequency
input signal and cause unwanted modulation products. The
supply voltage should stay within the tolerance given in the
specification table. A basic application configuration with the
power supply decoupling is shown in Figure 11.
ADS5221
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SBAS262A
30
31
24.9Ω
42
1.5V
36
AVDD
AVDD
37
0.10µF
48
0.10µF
DVDD
47
DVDD
4
5
Q PD
3
2
10µF
+
VREF
NC
RSEL
NC
OVR
IN
2.2µF
2Ω
+3.3
10µF
402Ω
+
15µF
34
33
2Ω
2.2µF
32
38
39
40
43
44
45
27
28
29
46
NC
D11 (MSB)
D10
REFT
D9
NC
D8
REFB
D7
ADS5221
D6
AGND
D5
AGND
D4
AGND
D3
AGND
D2
DGND
D1
DGND
D0 (LSB)
AGND
26
25
24
23
22
LATCH
10
11
12
13
14
15
16
17
18
19
20
21
SN74LVTH16374
AGND
AGND
CLK
GNDRV
15µF
GNDRV
+
NC
VDRV
35
NC
IN
VDRV
41
AVDD
10µF
+
22pF
24.9Ω
1.5V
402Ω
ST PD
+
Mode Select
2.2µF
0.10µF
OE
MSBI
1
DVDD
–
7
6
49.9Ω
9
VPULSE
8
+
0.10µF
10µF
+
VDRV
NC = No Connection.
FIGURE 11. General Layout for the ADS5221.
POWER DISSIPATION
POWER DOWN (PD)
In normal operating mode (STPD = low and QPD = low), the
typical total power dissipation of the ADS5221 is 285mW.
The majority of the power consumption is due to biasing;
therefore, this part of the total power dissipation is independent of the applied clock frequency. The digital power dissipation from the output driver is less than 10% of the total
power dissipation. This portion of the power consumption is
proportional to the sampling rate, digital output load and
number of the bits. The current on the VDRV supply is
directly related to the capacitive loading of the data output
pins; care must be taken to minimize such loading.
The ADS5221 provides two power-down modes for different
application requirements. One is standard power down
(STPD); the second is quasi power down (QPD). Setting
STPD to logic high (and QPD to logic low or high) will shut
down the internal ADC core and power down the reference
circuit. In this case the power dissipation is typically 15mW.
With 10µF external decoupling capacitor at REFT and REFB,
it takes about 800µs to fully restore normal operation after the
normal mode is enabled. Setting QPD to logic high (and
STPD to logic low) will shut down the internal ADC core and
while the internal reference circuit power remains on. In this
ADS5221
SBAS262A
www.ti.com
15
case, power dissipation is typically 70mW. It takes about 2µs
to fully restore normal operation after the normal mode is
enabled. During power-down, data in the converter pipeline
will be lost and new valid data will be subject to the specified
pipeline delay.
LAYOUT AND DECOUPLING
Proper grounding and bypassing, short lead length, and the
use of ground planes are particularly important for highfrequency designs. Achieving optimum performance with a
fast sampling converter like the ADS5221 requires careful
attention to the PC board layout in order to minimize the
effect of board parasitics and optimize component placement. A multi-layer board usually ensures best results and
allows convenient component placement. The ADS5221 must
be treated as an analog component, and the AVDD pins
connected to a clean analog supply. This ensures the most
consistent results, because digital supplies often carry a high
level of switching noise that could couple into the converter
and degrade the performance. The driver supply pins (VDRV)
must also be connected to a low-noise supply. Supplies of
adjacent digital circuits can carry substantial current transients. The supply voltage must be thoroughly filtered before
connecting to the VDRV supply of the converter. All ground
connections on the ADS5221 are internally bonded to the
metal flag (bottom of package) that forms a large ground
plane. All ground pins must directly connect to an analog
ground plane that covers the PC board area under the
converter. Due to its high sampling frequency, the ADS5221
generates high frequency current transients and noise (clock
feedthrough) that are fed back into the supply and reference
lines. If not sufficiently bypassed, this adds noise to the
conversion process. See Figure 11 for the recommended
supply decoupling scheme for the ADS5221. All AVDD pins
should be bypassed with a combination of 0.1µF ceramic
chip capacitors (0805, low ESR) and a 10µF tantalum tank
capacitor. A similar approach may be used on the digital
16
supply pins DVDD and driver supply pins, VDRV. In order to
minimize the lead and trace inductance, the capacitors must
be located as close to the supply pins as possible. They are
best placed directly under the package where double-sided
component mounting is allowed. In addition, larger bipolar
decoupling capacitors (2.2µF to 10µF), effective at lower
frequencies, must also be used on the main supply pins.
They can be placed on the PC board in close proximity
(< 0.5 inches) to the ADC. If the analog inputs to the
ADS5221 are driven differentially, it is especially important to
optimize towards a highly symmetrical layout. Small trace
length differences can create phase shifts compromising a
good distortion performance. For this reason, the use of two
single op amps rather than one dual amplifier enables a more
symmetrical layout and a better match of parasitic capacitances. The pin orientation of the ADS5221 package follows
a flow-through design with the analog inputs located on one
side of the package, whereas the digital outputs are located
on the opposite side of the quad-flat package. This provides
a good physical isolation between the analog and digital
connections. While designing the layout, it is important to
keep the analog signal traces separated from any digital lines
to prevent noise coupling onto the analog portion. Try to
match trace length for the differential clock signal (if used) to
avoid mismatches in propagation delays. Single-ended clock
lines must be short and should not cross any other signal
traces. Short circuit traces on the digital outputs will minimize
capacitive loading. Trace length must be kept short to the
receiving gate (< 2 inches) with only one CMOS gate connected to one digital output. If possible, the digital data
outputs must be buffered (with the TI SN74LTH16374, for
example). Dynamic performance can also be improved with
the insertion of series resistors at each data output line. This
sets a defined time constant and reduces the slew rate that
would otherwise flow as the fast edge rate. The resistor value
may be chosen to give a time constant of 15% to 25% of the
used data.
ADS5221
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SBAS262A
PACKAGE OPTION ADDENDUM
www.ti.com
9-Dec-2004
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
ADS5221PFBR
ACTIVE
TQFP
PFB
48
2500
None
CU NIPDAU
Level-2-220C-1 YEAR
ADS5221PFBT
ACTIVE
TQFP
PFB
48
250
None
CU NIPDAU
Level-2-220C-1 YEAR
Lead/Ball Finish
MSL Peak Temp (3)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - May not be currently available - please check http://www.ti.com/productcontent for the latest availability information and additional
product content details.
None: Not yet available Lead (Pb-Free).
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean "Pb-Free" and in addition, uses package materials that do not contain halogens,
including bromine (Br) or antimony (Sb) above 0.1% of total product weight.
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDECindustry standard classifications, and peak solder
temperature.
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Addendum-Page 1
MECHANICAL DATA
MTQF019A – JANUARY 1995 – REVISED JANUARY 1998
PFB (S-PQFP-G48)
PLASTIC QUAD FLATPACK
0,27
0,17
0,50
36
0,08 M
25
37
24
48
13
0,13 NOM
1
12
5,50 TYP
7,20
SQ
6,80
9,20
SQ
8,80
Gage Plane
0,25
0,05 MIN
0°– 7°
1,05
0,95
Seating Plane
0,75
0,45
0,08
1,20 MAX
4073176 / B 10/96
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. Falls within JEDEC MS-026
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